JP2006020467A - Switching power circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To enable the tacking to a wider AC input voltage range and a wide range of load conditions, by achieving the downsizing and the cost reduction of a power circuit which is equipped with a power factor improving function, and the rise of its power conversion. <P>SOLUTION: A voltage resonance type converter is provided with a power regeneration system of power factor improving circuit. As regards a power isolation transformer, the ripple of commercial AC power cycle to be superposed on secondary DC output voltage is reduced by putting it at the low coupling degree by a specified coupling coefficient or under. Moreover, the transmission loss between the primary side and the secondary side caused by putting the power isolation transformer at low coupling degree is compensated for by the resonation of a secondary parallel resonance circuit. Moreover, the power conversion improving circuit is provided with a control transformer PRT using a high frequency inductor as controlled winding. The inductance of the above controlled winding is variable according to the level of rectified smoothed voltage Ei, whereby it controls a power factor to be constant to AC input voltage and load ripple. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit having a power factor correction function.

The present applicant has previously proposed various power supply circuits including a resonance type converter on the primary side. Various power supply circuits configured with a power factor correction circuit for improving the power factor of the resonant converter have also been proposed.
As the above power factor correction circuit, the applicant of the present invention previously interrupted the rectified current flowing in the rectified current path by feeding back the switching output of the resonant converter to the rectified current path, thereby increasing the conduction angle of the AC input current. Various configurations have been proposed to expand and improve power factor.
However, when such a power factor correction circuit has a configuration in which, for example, the switching output of the resonant converter is directly fed back to the rectified current path (see, for example, Patent Document 1 and FIG. 6), the secondary side DC It has been found that the ripple voltage of the commercial power cycle of the output voltage is significantly increased than before the power factor improvement.
In the case of such a power factor improving configuration, the primary side resonance circuit is directly connected to the rectification current path of the commercial AC power supply. For this reason, a current having a commercial AC current cycle is superimposed on the current flowing through the primary winding of the insulating converter transformer, which is the inductance of the resonance circuit. As a result of the operation of the rectifying / smoothing circuit on the secondary side in a state where this overlap is transmitted to the secondary side of the insulation converter transformer, the ripple voltage of the commercial power cycle of the secondary DC output voltage is increased as described above. Will do. For example, when the power factor PF is set to be about 0.8, the level of the ripple voltage increases to 5 to 6 times.

  One possible countermeasure is to set a high control gain for stabilization control. However, if the level of the ripple voltage increases to 5 to 6 times as described above, even if the control gain is increased to the limit, it is insufficient to effectively suppress the ripple. Therefore, it is conceivable that the capacitance of the smoothing capacitor for smoothing the secondary side DC output voltage is further increased 5 to 6 times after the control gain is set to a certain limit. However, the selection of such a smoothing capacitor greatly increases the cost, and practical use is not practical. Since such a countermeasure is not practical, it may be difficult to adopt it in a device that strictly requires, for example, a ripple voltage to be below a certain level.

Therefore, the applicant of the present invention has proposed various types of power factor correction circuits that feedback the switching output in which the ripple voltage is suppressed.
FIG. 11 is a circuit diagram showing an example of a switching power supply circuit that is configured based on the invention of the power factor improvement circuit of the switching output feedback type that has been applied by the present applicant and that has reduced ripple voltage. . This power supply circuit is configured such that a power factor correction circuit is provided for a voltage resonance type switching converter.

  In the switching power supply circuit shown in this figure, as a noise filter for removing common mode noise, a pair of common mode choke coils CMC and two across capacitors CL are provided for the line of the commercial AC power supply AC. Yes.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by a bridge rectifier circuit Di composed of four low-speed rectifier diodes, and the rectified output is charged to the smoothing capacitor Ci via the power factor correction circuit 20. The As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. The configuration and operation of the power factor correction circuit 20 will be described later.

  In this figure, a voltage resonance type switching converter that performs a switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage has, for example, one switching element Q1 as a high breakdown voltage bipolar transistor. That is, a so-called single-end system is used.

A switching drive signal output from the oscillation / drive control circuit 21 is input to the base of the switching element Q1.
The collector of switching element Q1 is connected to the positive terminal of smoothing capacitor Ei via primary winding N1 of insulating converter transformer PRT. That is, the DC input voltage (Ei) is supplied through the primary winding N1. The emitter is grounded.
A clamp diode DD is connected between the base and emitter of the switching element Q1.

A primary side voltage resonance capacitor Cr is connected between the collector and emitter of the switching element Q1.
The primary side voltage resonance capacitor Cr forms a voltage resonance circuit (parallel resonance circuit) with the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT. A voltage resonance type operation is obtained as the switching operation of the switching element Q1 by the resonance action of the voltage resonance circuit. For this reason, a sinusoidal pulse waveform is obtained as the voltage VQ1 between the collector and emitter of the switching element Q1 during the period when the switching element is turned off.

  The oscillation / drive control circuit 21 supplies a base current as a switching drive signal to the base of the switching element Q1 in order to drive the switching element Q1. As a result, the switching element Q1 performs a switching operation at a switching frequency corresponding to the period of the switching drive signal. The oscillation / drive control circuit 21 is also configured to vary the switching frequency of the switching element Q1 in order to stabilize the secondary side DC output voltage.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
As the structure of the insulating converter transformer PIT, for example, an EE type core in which an E type core made of a ferrite material is combined is provided. Then, after the winding part is divided on the primary side and the secondary side, the primary winding N1 (and tertiary winding N3) and the secondary winding N2 are wound around the central magnetic leg of the EE type core. Disguise.
In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT, whereby k = 0.80 between the primary side and the secondary side. A coupling coefficient k of about ˜0.85 is obtained. This degree of coupling coefficient k is a degree of coupling that can be regarded as loose coupling, and accordingly, a saturated state is hardly obtained. Further, the value of the coupling coefficient k becomes a setting element of the leakage inductance (L1).

  One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end is connected to the positive electrode (rectified smoothing voltage Ei) of the smoothing capacitor Ci.

  On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, so that it depends on the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2. A secondary side parallel resonant circuit is formed. By the resonance operation of the parallel resonance circuit, the alternating voltage excited in the secondary winding N2 becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side.

  That is, in this power supply circuit, the primary side is provided with a primary side voltage (parallel) resonance circuit for making the switching operation a voltage resonance type, and the secondary side is also provided with a parallel for obtaining a voltage resonance operation in the rectifier circuit system. A resonant circuit is provided. In the present specification, a switching converter configured to operate with a plurality of resonance circuits in this way is also referred to as a “composite resonance switching converter”.

  A full-wave rectifier circuit including a bridge rectifier circuit DBR and a smoothing capacitor Co is connected to the secondary side parallel resonant circuit formed as described above. This full-wave rectifier circuit generates a DC output voltage Eo as a voltage across the smoothing capacitor Co by inputting an alternating voltage as a resonant output of the secondary side parallel resonant circuit and performing a rectifying and smoothing operation. The DC output voltage Eo is supplied to the load and is input to the oscillation / drive control circuit 21 as a detection voltage for constant voltage control.

In the circuit shown in FIG. 11, the configuration of the full-wave rectifier circuit for the secondary side rectifier circuit is zero current switching (ZCS: Zero Current Switching) of the switching element Q1 in response to a wide variation in load power. ) Is ensured.
As described above, in the power supply circuit shown in FIG. 11, a coupling coefficient k of the insulating converter transformer PIT is 0.8 or more. Although this is a loose coupling, it can be said that a certain degree of coupling is maintained. Under such a degree of coupling, for example, if the secondary side rectifier circuit is a half-wave rectifier circuit, the above-described ZCS operation may not be properly maintained.

  In the oscillation / drive control circuit 2, as a constant voltage control operation, the switching frequency is varied according to the level of the DC output voltage Eo input as described above. The period during which the element Q1 is off is fixed, and the period during which the element Q1 is on is variably controlled. That is, in this power supply circuit, as the constant voltage control operation, the switching frequency is variably controlled to control the resonance impedance for the switching output, and at the same time, the conduction angle control (PWM control) of the switching element in the switching cycle is also performed. Can be seen. This complex control operation is realized by a set of control circuit systems.

The power factor improvement is performed by the power factor improvement circuit 20.
The power factor correction circuit 20 shown in this figure includes rectifier diodes D1 and D2, a filter capacitor CN, an inductor LS, and a tertiary winding N3. The rectifier diode D1 is selected as a low speed type, and the rectifier diode D2 is selected as a high speed type (high speed recovery type) diode element.
The tertiary winding N3 is formed so as to wind up the winding end end side of the primary winding N1 in the insulating converter transformer PIT.

As shown in the figure, the rectifier diode D1 is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Further, between the positive output terminal of the same bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci, a series connection circuit of rectifier diode D2-inductor Ls-tertiary winding N3 due to the parallel connection with the rectifier diode D1. Is inserted.
In this case, the filter capacitor CN is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. In other words, the filter capacitor CN is connected in parallel to both the rectifier diode D1 and the series connection circuit of the rectifier diode D2, the inductor Ls, and the tertiary winding N3.
The filter capacitor CN and the inductor Ls inserted as described above form a normal mode LC low-pass filter, which prevents high-frequency noise of the switching period from flowing into the commercial AC power supply line.

  Depending on the switching operation of the switching element Q1, a sinusoidal resonance pulse voltage can be obtained as the voltage across the switching element Q1 (collector-emitter voltage) VQ1 during the off period. The alternating voltage due to the resonance pulse voltage is transmitted (induced) from the primary winding N1 of the insulating converter transformer PIT to the tertiary winding N3 of the power factor correction circuit 20. The alternating voltage obtained at the tertiary winding N3 is also applied to the rectifier diode D2 via the inductor Ls.

In this case, the rectified current flowing from the positive output terminal of the bridge rectifier circuit Di flows into the smoothing capacitor Ci via the rectifier diode D1 (anode → cathode), the rectifier diode D2 (anode → cathode) → the inductor Ls → It flows through a path flowing into the smoothing capacitor Ci via the tertiary winding N3.
That is, during the period when the AC input voltage VAC is in the vicinity of the peak, the low-speed rectifier diode D1 is turned on, and the rectified current from the bridge rectifier circuit Di is passed through the rectifier diode D1 to the smoothing capacitor Ci. To flow into. Further, during the period in which the AC input voltage VAC is lower than the vicinity of the peak, the rectified current from the bridge rectifier circuit Di passes through the series connection circuit of the rectifier diode D2-inductor Ls-tertiary winding N3 to the smoothing capacitor Ci. However, as described above, an alternating voltage is applied to the rectifier diode D2, so that the rectified current flowing through the path is switched and intermittently operated. Become. In this way, by configuring so that the rectified current is divided in the two rectified current paths, the conduction loss in the rectifier diode in the power factor correction circuit is reduced, and the power conversion efficiency can be increased accordingly. .
As described above, the operation of switching the rectifier diode D2 during the period in which the AC input voltage VAC is at a predetermined level lower than the vicinity of the peak is, for example, the ratio of the winding of the tertiary winding N3 to the primary winding N1. It is obtained by setting.

  In this manner, the switching output (voltage resonance pulse) of the switching element Q1 is fed back as a voltage to the rectified current path in the power factor correction circuit 20 through the tertiary winding N3. As a result of switching the rectified current flowing through the rectified current path to be switched, the conduction angle of the AC input current IAC that is the source of the rectified current is expanded, and the power factor is improved. The method of the power factor correction circuit configured to feed back the switching output as a voltage and thereby improve the power factor is also referred to as a voltage feedback method.

Then, depending on the configuration of the power factor correction circuit 20 shown in FIG. 11, the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage is suppressed.
In the power factor correction circuit 20, the switching output is voltage-feedbacked by the tertiary winding N3. In other words, it can be said that when the switching output obtained in the voltage resonance circuit on the primary side is fed back to the rectification current path of the commercial AC power supply, it is indirectly performed by the alternating voltage excited by the tertiary winding N3. As a result, the ripple component of the commercial AC power supply period is less likely to be induced from the tertiary winding N3 to the primary winding N1. Further, by connecting the inductor Ls in series with the tertiary winding N3, the tertiary winding is equivalent to the primary winding with a low degree of coupling due to the combined inductance of the tertiary winding and the inductor. There will be. As a result, the ripple component of the commercial AC power supply period is further less likely to be induced in the primary winding. As a result, the ripple voltage of the commercial AC power supply period superimposed on the secondary side DC output voltage is effectively suppressed.

The circuit diagram of FIG. 12 shows another example of the switching power supply circuit configured based on the invention previously filed by the present applicant. The basic configuration of the power supply circuit shown in this figure is similar to the power supply circuit of FIG. 11 in that a primary side single-ended voltage resonance converter and a secondary side secondary parallel (voltage) resonance circuit are combined. A power factor correction circuit based on a voltage feedback system is added to the composite resonance type converter. In addition, the insulating converter transformer PIT is configured to obtain a degree of coupling of about a coupling coefficient k = 0.80 to 0.85, as in FIG.
In this figure, the same parts as those in FIG.

The power factor improvement circuit 20A shown in this figure includes a low speed type rectifier diode D1, a high speed type (high speed recovery type) rectifier diode D2, a filter capacitor CN, and a magnetic coupling for power factor improvement as shown in the figure. It comprises a transformer MCT.
The magnetically coupled transformer MCT has a structure in which a primary winding Np and a secondary winding Ns are wound around a core so as to be magnetically coupled. The magnetically coupled transformer MCT in this case has a so-called divided bobbin in which divided winding positions are formed, and the primary winding Np and the secondary winding Ns are wound around different winding positions. Thus, a predetermined coupling coefficient that is loosely coupled is obtained as the degree of coupling between the primary side and the secondary side.

The winding start end portion of the primary winding Np is connected to the collector of the switching element Q1 via the serial connection of the primary winding N1 of the insulating converter transformer PIT, and the winding end portion is connected to the cathode of the rectifier diode D1. The The anode of the rectifier diode D1 is connected to the positive terminal of the smoothing capacitor Ci.
The winding end end of the secondary winding Ns is connected to the cathode of the rectifier diode D2, and the winding start end is connected to the positive terminal of the smoothing capacitor Ci. The anode of the rectifier diode D2 is connected to the positive output terminal of the bridge rectifier circuit Di.
Further, the filter capacitor CN is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci, so that the rectifier diode D1, the rectifier diode D2 and the secondary winding Ns are connected in series. Are connected in parallel to both.

By forming the power factor correction circuit 20A according to the above connection mode, the path through which the rectified current output from the positive output terminal of the bridge rectifier circuit Di flows into the smoothing capacitor Ci is the path through the rectifier diode D1, and the rectifier There are two paths through the series connection circuit of the diode D2-secondary winding Ns. That is, also in the power factor correction circuit 20A, the rectified current can flow through two paths in the same manner as the power factor correction circuit 20 in FIG.
Then, by performing the switching operation of the switching element Q1, as described above, as the voltage VQ1 across the switching element Q1, a resonance pulse voltage can be obtained in the off period. However, in the power factor correction circuit 20A, the resonance frequency described above is obtained. The alternating voltage due to the pulse voltage is first transmitted to the primary winding Np of the magnetic coupling transformer MCT, and further transmitted to the secondary winding Ns by the magnetic coupling of the magnetic coupling transformer MCT. As a result, an alternating voltage is applied to the rectifier diode D2. That is, the power factor correction circuit 20A employs a configuration in which the switching output is fed back as a voltage to the rectified current path through the magnetic coupling by the magnetic coupling transformer MCT as a voltage feedback system.
Incidentally, in the power factor correction circuit 20A, the primary winding Np of the magnetic coupling transformer MCT corresponds to the tertiary winding N3 in the power factor improvement circuit 20 of FIG. 11, and the secondary winding Ns is the power factor of FIG. This corresponds to the inductor Ls in the improvement circuit 20.

  Also in the power factor correction circuit 20A, the low-speed rectifier diode D1 conducts during the period when the AC input voltage VAC is in the vicinity of the peak, and the rectified current from the bridge rectifier circuit Di is passed through the rectifier diode D1. It is made to flow through the smoothing capacitor Ci. Further, during the period in which the AC input voltage VAC is lower than the vicinity of the peak, the rectified current from the bridge rectifier circuit Di is switched (interrupted) by the rectifier diode D2, so that the rectifier diode D2-secondary winding Ns. Is passed through the smoothing capacitor Ci via a series connection circuit. As a result, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.

  In the power factor correction circuit 20A shown in FIG. 12, the switching output obtained in the voltage resonance circuit on the primary side is fed back to the rectification current path through the magnetic coupling of the magnetic coupling transformer MCT. . In this case, since the magnetic coupling transformer MCT is formed loosely coupled, the ripple component of the commercial AC power supply period induced from the secondary winding Ns to the primary winding Np is reduced. Thereby, also in the power supply circuit shown in FIG. 12, the effect of suppressing the ripple voltage of the commercial AC power supply period superimposed on the secondary side DC output voltage is obtained.

FIG. 13 shows, as the characteristics of the power supply circuit shown in FIG. 12, power factor (PF), rectified smoothing voltage Ei, AC → DC power corresponding to load fluctuations from load power Po = 0 to 200 W (maximum load power). The fluctuation characteristics of conversion efficiency (ηAC → DC) are shown. In addition, in this figure, for comparison, characteristics of a power supply circuit that does not have a power factor improvement function by omitting the power factor improvement circuit 20A from the circuit configuration shown in FIG. 12 are also shown. In FIG. 13, the characteristic of the power supply circuit including the power factor correction circuit 20A as shown in FIG. 12 is indicated by a solid line, and the characteristic of the power supply circuit having the configuration in which the power factor improvement circuit 20A is omitted is indicated by a broken line.
As shown in this figure, with regard to the power factor (PF), the power supply circuit as shown in FIG. 12 is configured such that the power factor correction circuit 20A is omitted by the power factor improvement operation by the power factor improvement circuit 20A. You can see that it is higher than This figure also shows that a characteristic that is almost constant at PF = 0.8 is obtained in the range of the load power Po = 50 W to 200 W.
Further, as a tendency in the entire load range, the rectified and smoothed voltage Ei is higher in the power supply circuit as shown in FIG. 12 than in the power supply circuit in which the power factor correction circuit 20A is omitted. This is because the smoothing capacitor Ci is charged while the switching output is fed back to the rectification current path by the power factor correction circuit 20A, and the charging current level to the smoothing capacitor Ci increases accordingly. .
As for the AC → DC power conversion efficiency (ηAC → DC), the power supply circuit as shown in FIG. 12 is about 1 as compared with the power supply circuit omitting the power factor correction circuit 20A as a tendency with respect to the entire load fluctuation range. It has decreased by about%. This is because the magnetic coupling transformer MCT for feeding back the switching output to the rectification current path in the power factor correction circuit 20A is loosely coupled, so that power can be transmitted between the primary winding Np and the secondary winding Ns. This is due to loss.

JP 2001-95247 A JP 2002-34250 A

The power supply circuit as shown in FIGS. 11 and 12 has the following problems.
First, as described with reference to FIGS. 11 and 12, these power supply circuits are provided with the tertiary winding N3 in order to eliminate the problem of an increase in the ripple of the commercial AC power supply period superimposed on the secondary DC output voltage. Is added and inductors are connected in series, or a magnetic coupling transformer MCT is added.
For this reason, as the power factor improving circuit configured to feed back the switching output to the rectified current, the number of parts increases, resulting in an increase in cost, an increase in circuit scale, and an increase in weight.

Further, as described with reference to FIG. 13, in the power supply circuit shown in FIG. 12, the AC → DC power conversion efficiency (ηAC → DC) is reduced by about 1% as compared with the configuration without the power factor correction circuit 20A. Has produced. Such a tendency of lowering the power conversion efficiency occurs similarly in the power supply circuit of FIG. In the power supply circuit of FIG. 11, transmission loss occurs in the process of feeding back the switching output of the primary winding N1 as a voltage via the tertiary winding N3.
That is, in any of the power supply circuits shown in FIG. 11 and FIG. 12, transmission loss occurs in an additional component for feeding back the switching output as a voltage, which is disadvantageous as a total power conversion efficiency. As a result, as described above, the power conversion efficiency is lower than that of the power supply circuit having no power factor correction circuit.

In view of the above-described problems, the present invention is configured as a switching power supply circuit as follows.
That is, a rectifying / smoothing means that generates a rectified and smoothed voltage by inputting a commercial AC power supply, a switching means that includes a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage, and a switching drive for the switching element. Switching driving means.
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. An insulating converter transformer is formed and is set so that the coupling coefficient between the primary side and the secondary side is not more than a predetermined value.
Further, a primary side voltage resonance circuit which is formed by at least a leakage inductance component including the primary winding of the insulating converter transformer and a capacitance of the primary side voltage resonance capacitor and which makes the operation of the switching means a voltage resonance type is provided.
Also, a secondary parallel resonant circuit formed by connecting a secondary side parallel resonant capacitor in parallel to the secondary winding of the insulating converter transformer, and an alternating voltage excited by the secondary winding of the insulating converter transformer And a secondary side DC output voltage generation unit configured to generate a secondary side DC output voltage by performing a rectification operation.
Moreover, a power factor improvement means is provided.
The power factor improving means is a series connection circuit comprising a power factor improving switching element and a high frequency inductor, and the power factor improving series connection inserted in series with respect to the rectified current path formed by the rectifying and smoothing means. A circuit, a controlled winding that is a high-frequency inductor, and a control winding are wound, and the inductance of the controlled winding is made variable according to the level of the control current that is a direct current flowing through the control winding. A control transformer that operates, and control current variable control means that operates so as to vary the level of the control current according to the level of the rectified smoothing voltage level, and for the connection point between the power factor improving switching element and the high-frequency inductor. Thus, the end of the primary side voltage resonance circuit is connected.

According to the above configuration, the switching power supply circuit of the present invention includes the voltage resonance type converter as the primary side switching converter. In addition, the power factor is improved using a power regeneration system.
The power regeneration type power factor correction circuit has a fundamental circuit configuration that greatly increases the ripple of the commercial AC power supply cycle superimposed on the switching output, and as a result, commercial power superimposed on the secondary DC output voltage. The ripple of the AC power supply cycle also tends to increase significantly.
However, in the present invention, the ripple appearing in the secondary DC output voltage is reduced by weakening the degree of coupling between the primary side and the secondary side so that the magnetic flux density of the insulating converter transformer becomes not more than a predetermined value as described above. Can be sufficiently suppressed.
In addition, if the coupling degree of the insulating converter transformer is weakened, the transmission loss between the primary side and the secondary side will increase, but in the present invention, the secondary winding and the secondary side on the secondary side. By providing a secondary side parallel resonance circuit composed of a parallel resonance capacitor, resonance operation can be obtained on the secondary side. By this resonance operation, power is transmitted from the primary side to the secondary side in such a way that transmission loss between the primary side and the secondary side is compensated.
In addition, the power factor improving means is provided with a control transformer having a high-frequency inductor as a controlled winding, and the inductance of the high-frequency inductor (controlled winding) can be varied according to the rectified and smoothed voltage level. Yes.
The fact that the inductance of the high-frequency inductor (controlled winding) is varied in the power factor improving means in this way controls the feedback amount of the switching output obtained in the power factor improving means in accordance with the DC input voltage fluctuation. This means that, as a result, the power factor value improved by the power factor improving means can be controlled to be constant with respect to fluctuations in the rectified smoothing voltage. The level of the rectified and smoothed voltage varies corresponding to the commercial AC power supply level and also varies corresponding to the load variation. Therefore, according to the present invention, the power factor is controlled to be constant with respect to both level fluctuation and load fluctuation of the commercial AC power supply.

In this way, the present invention makes it possible to obtain a power supply circuit in which the ripple voltage of the commercial AC power supply cycle of the secondary side DC output voltage is sufficiently suppressed while having a power factor improvement circuit by a power regeneration system. Is done.
This ripple suppression effect is obtained not by adding circuit components in particular but by adopting a structure that can obtain a coupling degree (coupling coefficient) of a predetermined value or less for the insulating converter transformer. In addition, the power factor improvement circuit itself by the power regeneration method can be formed by a smaller number of simple component elements than a power factor improvement circuit such as a voltage feedback method. As a result, according to the present invention, a power supply circuit having a power factor correction circuit can be reduced in size / weight and can be obtained at a low cost.
In addition, in the power factor improvement circuit using the power regeneration system, there is less loss in the system that feeds back the switching output than the power factor improvement circuit such as the voltage feedback system. Further, according to the present invention, the secondary side parallel resonance circuit is provided to compensate for the transmission loss between the primary side and the secondary side due to the reduced degree of coupling of the insulating converter transformer. As a result, good characteristics can be obtained in terms of power conversion efficiency.
Furthermore, according to the present invention, a constant power factor can be obtained with respect to level fluctuations and load fluctuations of commercial AC power supplies, so that a switching power supply circuit having a power factor improvement function can accept an AC input voltage allowable range. As a result, it is possible to deal with a wider range than before with respect to various conditions of the load range, and the degree of freedom of equipment to which the power supply circuit can be applied is expanded.

The circuit diagram of FIG. 1 shows a configuration example of a power supply circuit as the best mode (embodiment) for carrying out the present invention. The power supply circuit shown in this figure employs a basic configuration including a power factor correction circuit for a voltage resonance type switching converter.
In the switching power supply circuit shown in this figure, first, a pair of common mode choke coils CMC and two across capacitors CL are inserted into the commercial AC power supply AC line as shown. The common mode choke coil CMC and the across capacitors CL and CL form a noise filter that removes common mode noise superimposed on the line of the commercial AC power supply AC.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by a bridge rectifier circuit Di formed by connecting four low-speed rectifier diodes as shown in the figure, and the rectified output thereof passes through the power factor correction circuit 10. In this way, the smoothing capacitor Ci is charged. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent switching converter. The configuration and operation of the power factor correction circuit 10 will be described later.

  In this figure, a voltage resonance type switching converter that performs a switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage has a structure including, for example, one MOS-FET switching element Q1. That is, a so-called single-ended voltage resonance converter is obtained.

A switching drive signal (voltage) output from the oscillation / drive circuit 2 is applied to the gate of the switching element Q1.
The drain of the switching element Q1 is connected to the winding start end of a primary winding N1 of an insulating converter transformer PIT described later. The winding end end of the primary winding N1 is connected to the positive terminal of the smoothing capacitor Ei through a series connection of the controlled winding N R (high frequency inductor) of the control transformer PRT in the power factor correction circuit 10. That is, in this case, since the power factor correction circuit 10 is inserted, the DC input voltage (Ei) is supplied to the switching element Q1 via the series connection of the controlled winding NR and the primary winding N1. It has come to be. The source of the switching element Q1 is grounded to the primary side ground.

  Since the switching element Q1 in this case is a MOS-FET, as shown in the figure, it incorporates a body diode DD so as to be connected in parallel between the source and the drain. As the body diode DD, the anode is connected to the source of the switching element Q1, and the cathode is connected to the drain of the switching element Q1. The body diode DD forms a path for flowing a switching current in the reverse direction, which is generated by the on / off operation (switching operation) of the switching element Q1.

A primary side voltage (parallel) resonance capacitor Cr is connected in parallel between the drain and source of the switching element Q1.
The primary side voltage resonance capacitor Cr forms a parallel resonance circuit (voltage resonance circuit) for the switching current flowing through the switching element Q1 by its own capacitance and the leakage (leakage) inductance L1 of the primary winding N1 of the insulating converter transformer PIT. . When this voltage resonance circuit performs a resonance operation, a voltage resonance type operation can be obtained as a switching operation of the switching element Q1. In response to this, a sinusoidal pulse waveform is obtained during the off-period of the voltage across the switching element Q1 (drain-source voltage) VQ1.

  However, in the power supply circuit according to the embodiment shown in FIG. 1, a circuit to be wound around the control transformer PRT in the power factor correction circuit 10 described later is interposed between the primary winding N1 of the insulating converter transformer PIT and the smoothing capacitor Ci. The control winding N R is inserted in series. Therefore, in this embodiment, the primary side voltage resonance circuit includes the controlled winding NR in addition to the capacitance of the primary side voltage resonance capacitor Cr and the leakage (leakage) inductance L1 of the primary winding N1 of the insulating converter transformer PIT. The inductance LR is also included. In other words, the inductance component for forming the primary side series resonance circuit is a combined inductance represented by L1 + LR.

  The oscillation / drive circuit 21 is a switching voltage which is a voltage for switching and driving the MOS-FET based on an oscillation circuit and an oscillation signal obtained by the oscillation circuit, for example, in order to drive the switching element Q1 by a separate excitation type. A drive signal is generated and applied to the gate of the switching element Q1. As a result, the switching element Q1 continuously performs on / off operations according to the switching frequency corresponding to the cycle of the switching drive signal. That is, a switching operation is performed.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained. In this case, as an actual EE type core, for example, EER-40 is selected on the premise that it corresponds to a load condition as described later and a rated level condition of an AC input voltage.

In addition, a gap G having a gap length of about 1.4 mm or more is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with k = 0.8 or less is obtained. That is, the prior art is more loosely coupled than the isolated converter transformer PIT of the power supply circuit shown in FIGS. Note that k = 0.75 was set as the actual coupling coefficient k.
The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  As described above, the winding end of the primary winding N1 of the insulating converter transformer PIT is connected to the positive terminal of the smoothing capacitor Ci through the controlled winding N R of the control transformer PRT in the power factor correction circuit 10. The winding start end is connected to the drain of the switching element Q1. As a result, the switching output of the switching element Q1 is transmitted to the primary winding N1, and an alternating voltage is generated in the primary winding N1.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2.
A secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. Thus, a secondary side parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2. Due to the resonance operation of the secondary side parallel resonance circuit, the alternating voltage excited in the secondary winding N2 becomes a resonance voltage close to a sine wave. That is, a voltage resonance operation is obtained on the secondary side.

  According to the description so far, the power supply circuit of the present embodiment includes a primary side voltage (parallel) resonance circuit for making the switching operation a voltage resonance type on the primary side, and a voltage in the rectifier circuit system on the secondary side. A parallel resonance circuit for obtaining a resonance operation is provided. That is, the configuration as a composite resonance type switching converter is adopted.

In the present embodiment, a half-wave rectifier circuit including one rectifier diode Do and a smoothing capacitor Co is connected as the secondary-side rectifier circuit. In this case, the anode of the rectifier diode Do is connected to the winding end of the secondary winding N2, and the cathode of the rectifier diode Do is connected to the positive terminal of the smoothing capacitor Co. The winding start end of the secondary winding and the negative terminal of the smoothing capacitor Co are both connected to the secondary side ground.
This half-wave rectifier circuit performs rectification during the half-wave period of the alternating voltage obtained as the output of the secondary side parallel resonant circuit and charges the smoothing capacitor Co. Thereby, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co.
The secondary side DC output voltage Eo is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

The control circuit 1 supplies a detection output corresponding to the level change of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching element Q1 such that the switching frequency is varied in accordance with the input detection output of the control circuit 1.
The variable operation of the switching frequency in the control circuit 1 is an operation in which the period during which the switching element Q1 is off is constant and the period during which the switching element Q1 is on is variably controlled. Such a control operation is a constant voltage control operation, in which the switching frequency is variably controlled to control the resonance impedance for the switching output. At the same time, the conduction angle control (PWM (Pulse Width Moduration) of the switching element in the switching period is performed. ) Control) can also be seen as performing. This composite control operation is realized by a set of control circuit systems.

  By variably controlling the switching frequency and conduction angle of the switching element Q1 as described above, the primary side and secondary side resonance impedances and the power transmission effective period in the power supply circuit are changed, and the primary winding of the insulating converter transformer PIT is changed. The amount of power transmitted from the line N1 to the secondary winding N2 (N2A, N2B) side also changes, but this cancels the level fluctuation of the secondary side DC output voltage Eo, so that the secondary side DC output An operation for controlling the level of the voltage Eo is obtained. That is, the secondary side DC output voltage Eo is stabilized.

Next, the configuration of the power factor correction circuit 10 will be described.
The power factor correction circuit 10 is provided so as to be inserted into a rectification current path in a rectification smoothing circuit for obtaining a DC input voltage (Ei) from a commercial AC power supply AC. Take the form of the shape.

  In the power factor correction circuit 10, the anode of a switching diode (power factor improving switching element) D1 that is a high-speed type (high-speed recovery type) is connected to the positive output terminal of the bridge rectifier circuit Di. The anode of the switching diode D1 is connected to the positive terminal of the smoothing capacitor Ci via a series connection of the controlled winding NR of the control transformer PRT. In other words, in this case, in the rectified current path for generating the rectified and smoothed voltage Ei, the switching diode D1−the controlled winding is applied to the line between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. A series connection circuit of the line NR is inserted. Here, the controlled winding NR has a function as a high-frequency inductor that receives feedback of the primary-side resonance current (switching output current) regenerated by the power regeneration in the power factor correction circuit using the power regeneration method.

Here, a structural example of the control transformer PRT provided in the power factor correction circuit 10 will be described with reference to FIGS.
First, the absolute control transformer PRT shown in FIG. 3 includes two double U-shaped cores CR11 and CR12 having four magnetic legs. A solid core is formed by joining the ends of the magnetic legs of the double U-shaped cores CR11 and CR12. In this case, the double U-shaped cores CR11 and CR12 having the same size can be used.
When the three-dimensional core is formed in this way, there are four joint portions of the double U-shaped cores CR11 and CR12 corresponding to each of the four magnetic legs. In this case, gaps G and G having a predetermined length are formed for two adjacent joints in these four joints, and no gap is formed for the remaining two joints.

In the three-dimensional core formed in this way, first, for example, the controlled winding NR is set in a predetermined manner so as to be wound around two magnetic legs that do not form the gap G on the double U-shaped core CR11 side. Wind the number of turns.
As shown in the figure, the control winding Nc is divided into a magnetic leg portion in which the gap G is formed and a magnetic leg portion in which the gap G adjacent to the magnetic leg is not formed on the double U-shaped core CR12 side. A predetermined number of turns are wound so as to be wound.

By winding the controlled winding NR and the control winding Nc as described above, the winding direction of the controlled winding NR is orthogonal to the winding direction of the control winding Nc. That is, a structure as a so-called orthogonal transformer is obtained.
By adopting such an orthogonal transformer structure, the control transformer PRT is configured as a saturable reactor that becomes magnetically saturated due to an increase in the direct current (control current Ic) flowing through the control winding Nc. Then, by being a saturable reactor, the leakage inductance in itself is varied in accordance with the level of the control current Ic flowing through the control winding Nc, so that the inductance LR of the controlled winding NR changes. To be.

Further, as another structure of the control transformer PRT, as shown in FIG. 4, with respect to the three-dimensional core, one core remains a double U-shaped core CR12 having four magnetic legs. The core may be formed by combining as a single U-shaped core CR13 having an arbitrary U-shaped cross section instead of the double U-shaped core CR11. Also in this core structure, two gaps G and G are formed by the same positional relationship as the control transformer PRT of FIG.
In contrast to this core structure, the controlled winding NR is wound around the magnetic leg on the side where the gap is not formed in the single U-shaped core CR13, and the primary winding N1 is a single core. In the U-shaped core CR3, it is wound so as to be wound around the magnetic leg on the side where no gap is formed.
For the control winding Nc, as shown in the drawing, a magnetic leg portion where a gap G is formed at a position on the magnetic leg end side of the double U-shaped core CR12 and a gap G adjacent to the magnetic leg are It winds so that it may wind around the magnetic leg part which is not formed.
In this case as well, the control winding Nc and the set of controlled windings NR are set so that their winding directions are orthogonal to each other, and a configuration as an orthogonal transformer is obtained.

Further, the control transformer PRT may have a structure using a core having the shape shown in FIG. 5 in addition to the above-described double U-shaped core or single U-shaped core.
First, in the control transformer PRT shown in FIG. 5, two half-eye-shaped cores CR21 and CR22 are prepared and combined so that the magnetic legs of these cores face each other, so that one planar-type character-shaped core is obtained. Form. In addition, in the square core, a total of four magnetic legs, two outside and two inside, face each other. Of these, each surface facing the two inside magnetic legs has a predetermined length. Gaps G and G are formed.
The controlled winding NR is wound with a predetermined number of turns so as to straddle the two inner magnetic legs in one half-eye-shaped core CR21.
The control winding Nc winds a predetermined number of turns so as to straddle one outer magnetic leg in the other half-eye-shaped core CR22 and one inner magnetic leg adjacent to the outer magnetic leg.

  In such a structure of the control transformer PRT, the outer magnetic leg around which the controlled winding NR (and the detection winding ND) is wound is different from the outer magnetic leg around which the control winding Nc is wound. However, this relationship is equivalent to making the winding directions orthogonal as shown in FIGS. 3 and 4. Therefore, even with the structure shown in FIG. 5, the control transformer PRT functions as a saturable reactor.

  With these structures, the control transformer PRT is formed as a saturable reactor that varies the inductance LR of the controlled winding NR according to the level of the direct current (control current Ic) flowing through the control winding Nc.

The level of the control current Ic is varied by the control circuit 3 provided in the same power factor correction circuit 10.
The control circuit 3 inputs the potential obtained at the connection point (voltage dividing point) of the voltage dividing resistors R1-R2 as a detection input. The series connection circuit of the voltage dividing resistors R1 to R2 is connected in parallel to the smoothing capacitor Ci, so that a rectified smoothing voltage (DC input voltage) is provided at the voltage dividing point of the voltage dividing resistors R1 to R2. A potential corresponding to Ei is generated. That is, the control circuit 3 inputs the level of the rectified and smoothed voltage Ei as a detection input, and variably controls the level of the control current Ic according to the level change of the input rectified and smoothed voltage Ei. In this case, the control circuit 3 performs a variable operation of decreasing the level of the control current Ic as the level of the rectified and smoothed voltage Ei (that is, the potential at the voltage dividing point of the voltage dividing resistor R1-R2) increases. Is configured to be performed. In practice, the control circuit 3 can be formed as a class A amplifier circuit including a bipolar transistor as an amplifier element, for example.
Although not shown in FIG. 1, the power for the control circuit 3 to flow the control current Ic is supplied by branching and inputting a DC voltage for operating the oscillation / drive circuit 2. It has become so. Further, the DC voltage serving as the power source for the oscillation / drive circuit 2 and the control circuit 3 is wound on the primary side of the insulating converter transformer PIT with a power supply winding having a predetermined number of turns. The voltage is obtained from the rectified and smoothed voltage output from the rectifier circuit connected to the supply winding.

A winding end end which is one end of the primary winding N1 is connected to a connection point between the switching diode D1 and the controlled winding NR in the power factor correction circuit 10. That is, the end of the primary side voltage (parallel) resonance circuit is connected.
Further, by adopting such a connection mode with the primary side voltage resonance circuit, as described above, the controlled winding NR has a serial connection relationship with the primary winding N1 in the primary side voltage resonance circuit. Therefore, the inductance LR is included in the inductance component forming the primary side voltage resonance circuit together with the leakage inductance L1 of the primary winding N1.

  The filter capacitor CN is provided to suppress the normal mode noise by absorbing the alternating component of the switching period generated by the switching operation of the switching diode D1, and in this case, the switching diode D1-controlled winding N R. Are connected in parallel to the series connection circuit.

As an operation of the power supply circuit of FIG. 1 including the power factor correction circuit 10 configured as described above, first, a basic power factor correction operation will be described.
In the configuration of the power factor correction circuit 10 of the present embodiment, the switching output obtained in the primary side voltage resonance circuit is directly transmitted to the controlled winding NR that functions as a high frequency inductor in the power factor correction circuit 10. It has come to be. As a result, in the power factor correction circuit 10, the switching output (primary side resonance current) obtained in the primary side voltage resonance circuit is regenerated as electric power so as to pass through the magnetic coupling by the controlled winding NR which is a high frequency inductor. Thus, it can be said that an operation of returning to the smoothing capacitor Ci is obtained.

The waveform diagram of FIG. 6 shows the switching operation of the switching element Q1 by the switching period. Here, a case where the switching frequency fs = 100 KHz is shown.
In this figure, a period from the period t2 to t0 is an on period in which the switching element Q1 is turned on, and a period from the period t0 to t2 is an off period in which the switching element Q1 is turned off.
The both-ends voltage (drain-source voltage) VQ1 of the switching element Q1 has a waveform that generates a sine wave-like resonant voltage pulse in the on period (t2 to t0) and a zero level in the off period (t0 to t2).

Here, in the ON period (t2 to t0) of the switching element Q1, the primary side resonance current Io flowing through the primary winding N1 of the insulating converter transformer PIT is smoothing capacitor Ci → controlled winding NR (high frequency inductor) → primary. It flows in the forward direction through the path of winding N1 → switching element Q1 (drain → source). Due to this current, electromagnetic energy is accumulated in the series connection circuit of the primary winding N1 and the controlled winding NR.
Subsequently, when the switching element Q1 is turned off at time t0, the flow of the primary resonance current Io in the forward direction is maintained by the electromagnetic energy accumulated in the series connection circuit of the primary winding N1 and the controlled winding NR, The primary side voltage resonance capacitor Cr is charged. This charging operation is performed during the period t0 to t1, and this charging current raises the potential at both ends of the primary side voltage resonance capacitor Cr and also increases the voltage VQ1 across the switching element Q1. .

  The next period t1 to t2 is a period in which the charge stored in the primary side voltage resonance capacitor Cr accumulated in the period t0 to t1 is discharged. By this discharge operation, the primary side resonance current Io is changed to the primary side voltage resonance. The current flows in a reverse path by the capacitor Cr → the primary winding N1 → the controlled winding NR → the smoothing capacitor Ci. That is, the primary side resonance current Io obtained by discharging the charge of the primary side voltage resonance capacitor Cr is regenerated in the smoothing capacitor Ci. Further, when current flows through this path, electromagnetic energy is accumulated in the series connection circuit of the primary winding N1 and the controlled winding NR at this time as well.

  In the period t2 to t3 immediately after the switching element Q1 ends the off state, the body diode DD of the switching element Q1 is turned on. For this reason, the primary resonance current Io in the reverse direction maintained by the electromagnetic energy accumulated in the series connection circuit of the primary winding N1 and the controlled winding NR during the period t1 to t2 is the body diode DD (anode → Cathode) → Primary winding N 1 → Controlled winding N R → Smoothing capacitor Ci That is, even during this period, the operation of regenerating the primary side resonance current Io to the smoothing capacitor Ci is obtained.

In the period t3 to t0, the switching element Q1 is turned on, and the primary side resonance current Io flows from the smoothing capacitor Ci → the controlled winding NR → the primary winding N1 → the switching element Q1 (drain → source). To be. That is, the forward (positive polarity) switching output current IQ1 flows.
According to such a switching operation, when the voltage resonance type converter and the power regeneration type power factor correction circuit 10 are combined, the periods t1 to t3 in FIG. 6 are power regeneration periods in which the power regeneration operation can be obtained. That's true.

FIG. 7 shows the operation of the power factor correction circuit 10 by the commercial AC power supply cycle.
Here, it is assumed that an AC input voltage VAC of 50 Hz is input as shown. When the AC input voltage VAC is input and the power supply circuit shown in FIG. 1 operates, the power factor correction circuit 10 also applies the primary side resonance current Io to the smoothing capacitor Ci as described above. The operation of regenerating and returning as electric power is performed.

  As the power regeneration is performed as described above, the voltage V1 between the connection point between the primary winding N1, the controlled winding NR, and the cathode of the switching diode D1 and the primary side ground is the AC input current. During a period when IAC is non-conducting and no rectified current flows, the envelope waveform has a constant level in which an alternating voltage component due to the switching period is superimposed on the rectified smoothed voltage Ei. On the other hand, during the period in which the AC input current IAC conducts and the rectified current flows, the rectified output voltage is obtained as a waveform that is superimposed in a sine wave form from the constant level.

  The high-speed (high-speed recovery) switching diode D1 conducts when, for example, the positive / negative absolute value of the AC input voltage VAC is about ½ or more of the peak value, but the alternating voltage component of the voltage V1. As a result, the rectified current is switched and intermittently operated. That is, the rectified current I1 flowing from the positive output terminal of the bridge rectifier circuit Di to the switching diode D1 is obtained as an approximately sine wave alternating waveform.

  The conduction period of the rectified current I1 flowing with such a waveform also flows during a period when the rectified output voltage level output from the bridge rectifier circuit Di is lower than the voltage across the smoothing capacitor Ci. The conduction period of the AC input current IAC shown in FIG. 7 also substantially coincides with the conduction period of the rectified output current I1. That is, the conduction angle of the AC input current IAC is larger than that without the power factor correction circuit, and the waveform of the AC input current IAC approaches the waveform of the AC input voltage VAC. Yes. That is, the power factor is improved.

The power factor correction circuit 10 according to the present embodiment enables the power factor to be improved by the above-described operation, and the power factor to be improved is further changed with respect to the change in the AC input voltage. Can be controlled to be constant. The power factor stabilization operation will be described.
As described above, in the power factor correction circuit 10, the control circuit 3 varies the control current level Ic flowing through the control winding Nc of the control transformer PRT in accordance with the change in the DC input voltage Ei that is the detection input. Has been. The DC input voltage Ei is a DC voltage obtained by rectifying and smoothing the AC input voltage VAC. Therefore, the level corresponds to the level of the AC input voltage VAC. From this, the control circuit 3 detects the level of the AC input voltage VAC by using the DC input voltage Ei as a detection input, and sets the level of the control current Ic according to the detected level of the AC input voltage VAC. It can be said that it is variable.

  Here, it is assumed that the fluctuation occurs as the AC input voltage VAC increases. In response to this, the control circuit 3 varies the level of the control current Ic so as to increase to a required level corresponding to the increase level of the AC input voltage VAC.

The control transformer PRT that is a saturable reactor operates so as to increase the inductance LR of the controlled winding NR as the level of the control current Ic flowing through the control winding Nc decreases.
An increase in the inductance of the controlled winding NR that functions as a high-frequency inductor in the power factor correction circuit 10 means that the controlled winding according to a certain amount of power fed back from the primary side voltage resonance circuit. This means that the energy stored in the inductor as the line NR also increases, which leads to an increase in the amount of power feedback for power factor improvement. Due to the increase in the power feedback amount, for example, the lower limit of the level of the AC input voltage VAC required for the switching diode D1 to switch the rectified current is lowered, and the conduction angle of the AC input current IAC corresponding to that level is reduced. Thus, an effect of canceling the decrease in the power factor in accordance with the increase in the AC input voltage VAC is obtained. As a result of such an operation, the power factor is controlled to be constant regardless of the change in the AC input voltage VAC.

The waveform diagram of FIG. 7 described above shows the operation of the power supply circuit shown in FIG. 1 when the AC input voltage VAC = 100 V corresponding to the AC 100 V system. On the other hand, the waveform diagram shown in FIG. 8 shows the operation of the power supply circuit shown in FIG. 1 when the AC input voltage VAC = 230 V corresponding to the AC 200 V system. 7 and 8 have the same load condition.
Comparing FIG. 8 with FIG. 7, it can be seen that the level of the AC input voltage VAC has almost twice the amplitude corresponding to the change from VAC = 100V to VAC = 230V. Further, the AC input current IAC has a peak level of about ½ of the AC input current IAC of FIG. 7 in response to the level of the AC input voltage VAC being almost doubled. Further, the peak level of the rectified output current I1 obtained by switching the rectified current based on the AC input current IAC by the switching diode D1 is also about 1/2 of the peak level of the rectified output current I1 in FIG. It has become.
Also in FIG. 8, the period in which the rectified output current I1 flows intermittently corresponds to the conduction period of the AC input current IAC. This is due to the intermittent rectified output current in the power factor correction circuit 10. It shows that the conduction angle of the input current IAC is enlarged. That is, the power factor is improved.
A measurement result is obtained that the conduction angle of the AC input current IAC shown in FIG. 8 is substantially equal to the AC input current IAC of FIG. That is, FIG. 7 and FIG. 8 show that the power factor is kept constant regardless of the AC input voltage fluctuation.

Further, the power factor has a property of fluctuating depending on the load fluctuation.
That is, although the load current amount changes according to the load fluctuation, the change in the load current also appears as a change in the primary side switching current (primary side resonance current) amount. For example, if the load fluctuates so as to become light, the load current amount decreases, and accordingly, the primary side switching current amount also decreases. Then, by reducing the primary side switching current amount, the amount of current taken out from the smoothing capacitor Ci that is a DC input voltage source is reduced, and the rectified and smoothed voltage Ei that is the voltage across the smoothing capacitor Ci is increased. Become. If the rectified and smoothed voltage Ei increases, the amount of current to be charged to the rectified and smoothed voltage Ei can be reduced accordingly, and the conduction angle of the AC input current IAC becomes narrow and the power factor decreases.

  Here, the power factor correction circuit 10 of the present embodiment uses the rectified and smoothed voltage Ei as a detection input. According to the above, since the DC input voltage Ei changes according to the load fluctuation, the control operation for power factor improvement executed by the control circuit 3 in the power factor improvement circuit 10 is the AC input voltage. It can be said that the variable operation of the inductance LR of the controlled winding NR according to the change in VAC and the variable operation of the inductance LR of the controlled winding NR according to the load change are simultaneously performed.

Then, as described above, the control circuit 3 performs control so that the inductance LR of the controlled winding NR is increased as the level of the rectified smoothing voltage Ei that is the detection input increases. Therefore, the load fluctuation is controlled so that the inductance LR of the controlled winding NR is increased in accordance with the light load condition.
If the inductance LR of the controlled winding NR is increased, the induced voltage of the controlled winding NR increases accordingly, and the power feedback amount by the power factor correction circuit 10 is increased. As a result, the tendency to increase the power factor is controlled, so that an operation that cancels the decrease in the power factor due to the decrease in the switching current amount is obtained. That is, the control system composed of the control circuit 3 and the control transformer PRT also executes control for keeping the power factor constant with respect to load fluctuations.

In this way, the power factor correction circuit 10 of the present embodiment is designed to improve the power factor by the power regeneration method, and the power factor to be improved is constant with respect to the AC input voltage VAC and the load fluctuation. It is trying to become.
As a result, for example, depending on the setting of the variable range for the inductance LR of the controlled winding NR of the control transformer PRT, the improved power factor is stabilized over a wide range of AC input voltage VAC and load power fluctuation range. It is possible to obtain a switching power supply circuit that can be used.
In practice, the power supply circuit of the present embodiment shown in FIG. 1 has an AC input voltage with respect to a fluctuation range of VAC = 80 V to 280 V, and a load power with respect to a fluctuation range of Po = 30 W to 200 W. Thus, it is possible to maintain PF> 0.75 for the power factor (PF). This characteristic satisfies the specified value of the power supply harmonic distortion regulation.
Here, the above-described fluctuation range of the AC input voltage VAC = 80 V to 280 V sufficiently covers both AC 100 V system and AC 200 V system AC input voltage ranges. Therefore, the power supply circuit of this embodiment It can be said that the power supply circuit having the power factor correction function has a very advantageous characteristic for wide-range compatibility. Note that the wide-range compatibility means that the power supply circuit can operate in accordance with any input of the commercial AC power supply AC100V system and AC200V system. Since the power supply circuit of the present embodiment is compatible with a wide range and can cope with a relatively wide range of load fluctuations as described above, the power supply circuit according to the present embodiment is accordingly adapted to this embodiment. The number of electronic devices that can be equipped with a power circuit based on this will also increase.

Here, the power factor improvement circuit 10 of the power supply circuit according to the present embodiment is configured as the third winding N3 or the power factor improvement circuits 20 and 20A of the power supply circuit shown in FIGS. It does not employ a configuration in which a primary side resonance current by a loosely coupled magnetic coupling transformer MCT or the like is fed back as a voltage. That is, as a circuit form, in accordance with the power regeneration system, the end of the primary winding N1 through which the primary side resonance current Io flows is connected to the rectified current path (the switching diode D1 and the controlled winding in the power factor correction circuit 10). The line NR (connection point of the high frequency inductor)).
However, in the power supply circuit of the present embodiment, even if the configuration of the power factor correction circuit that performs direct feedback of the switching output is adopted, the commercial AC power supply cycle superimposed on the secondary side DC output voltage Eo The increase in ripple is greatly suppressed to the extent that there is no practical problem.
This is because the degree of coupling of the insulating converter transformer PIT is lowered to a coupling coefficient k = 0.8 or less (about k = 0.75), which is higher than that of the insulating converter transformer PIT of the power supply circuit shown in FIGS. This is because a loosely coupled state is set.

That is, when the degree of coupling is reduced to about the coupling coefficient k = 0.75 for the insulating converter transformer PIT, the power transmission rate from the primary side to the secondary side is also lowered accordingly.
Thereby, even if the ripple current of the commercial AC power supply cycle is superimposed on the primary side resonance current Io on the primary side, the ripple component of the commercial AC power supply cycle is greatly attenuated when power is transmitted to the secondary side. As a result, the ripple voltage of the commercial AC power supply period superimposed on the secondary side DC output voltage Eo is also greatly suppressed.

However, simply by promoting the degree of loose coupling for the insulating converter transformer PIT, the transmission power between the primary side and the secondary side as the insulating converter transformer PIT is also attenuated. Transmission loss occurs, which causes a significant reduction in power conversion efficiency.
However, in the present embodiment, a parallel resonance circuit (N2 // C2) is provided also on the secondary side. As a result, on the secondary side, an alternating output is obtained so as to be amplified by the resonance operation of the secondary side parallel resonance circuit. As a result, transmission loss from the primary side to the secondary side is compensated, and as a result, the power conversion efficiency is lowered due to the insulating converter transformer PIT being further loosely coupled. There will be no.

In this way, as the power supply circuit of the present embodiment, first, by actively combining the secondary side parallel resonant circuit, the insulation converter transformer PIT can be made more efficient than before without causing a decrease in power conversion efficiency. Can also set the state of loose coupling. This suppresses the ripple of the commercial AC power supply cycle superimposed on the secondary side DC output voltage Eo, and it is possible to adopt a power factor improvement circuit based on the power regeneration system regardless of the tertiary winding N3 or magnetic coupling transformer. It is said.
In the power factor correction circuit 10 of the present embodiment, the basic configuration for power factor improvement by the power regeneration method is as follows: a switching diode D1, a controlled winding N R of a control transformer PRT functioning as a high frequency inductor, and a filter capacitor CN. In this respect, the number of parts is reduced when compared with the power factor correction circuits 20 and 20A shown in FIGS. .

  In addition, the power factor correction circuit 10 includes the control transformer PRT and the control circuit 3 as described above, thereby controlling the power factor improved by the power regeneration method to be constant. The PRT is considerably small as will be described later, and the control circuit 3 can also be formed by a simple class A amplifier circuit including a bipolar transistor, for example, as described above. That is, the power supply circuit according to the present embodiment can minimize the increase in the number of parts, the circuit scale, and the cost increase when adding the power factor correction circuit having the power factor stabilization function. It will be.

Further, the secondary side rectifier circuit is a full-wave rectifier circuit provided with a bridge rectifier circuit DBR in the circuits shown in FIGS. 11 and 12, whereas in the present embodiment, the secondary side rectifier circuit is a half-wave. It is a rectifier circuit. This is because, in the power supply circuit of the present embodiment, the isolation converter transformer PIT is more loosely coupled than before, so that even a half-wave rectifier circuit can ensure ZCS regardless of load fluctuations. Because it became.
Thus, even if the secondary side rectifier circuit is a half-wave rectifier circuit, the number of parts and the cost can be reduced accordingly.

As for the power conversion efficiency, the power factor improving circuit 10 adopts a circuit form as a power regeneration system in which the primary winding N1 is directly connected as described above. As a result, the power loss in the system that feeds back the switching output as compared with the case of adopting the configuration in which the tertiary winding N3 and the loosely coupled magnetic coupling transformer MCT are interposed and fed back as a voltage as shown in FIGS. Will be reduced.
In addition, as described above, in the power supply circuit shown in FIG. 1, the primary side and the secondary side are connected by the added polarity, but this also makes it more than the case where the connection is made by the depolarized connection. Power loss is reduced. When the depolarized connection is used, it is difficult to obtain a waveform having a flat peak as shown in FIG. 3, for example, as shown in FIG. 3 for the on-period waveform of the switching current IQ1 flowing through the switching element Q1, resulting in a switching loss. become. When the secondary-side rectifier circuit is a half-wave rectifier circuit, the polarity of the peak waveform of the switching current IQ1 is maintained by applying the polarity, and the power loss can be reduced accordingly.
Then, when combined with the effect of reducing the power loss obtained by these configurations and the compensation effect of the transmission power loss in the insulating converter transformer PIT by combining the secondary parallel resonant circuit described above, The power supply circuit according to the embodiment has good characteristics with respect to power conversion efficiency. As a result, as described below, the power supply circuit according to the present embodiment can achieve higher power conversion efficiency than the configuration without the power factor correction circuit 10 as an actual experimental result. It became a thing.

9 and 10 show various characteristics of the power supply circuit of the present embodiment shown in FIG.
The characteristics shown in these figures are obtained by conducting experiments by selecting the main parts of the configuration of the power supply circuit shown in FIG. 1 as follows.
For the insulating converter transformer PIT, the core of EER-40 was selected, and the gap G was 1.6 mm. The number of turns of the primary winding N1 is 40T (turns), the number of turns of the secondary winding N2 is 50T, and k = 0.75 is obtained for the coupling coefficient k.
Further, for the control transformer PRT in the power factor correction circuit 10, for example, as a child adopting the structure shown in FIG. 3, the core size in this case is 16 mm × about the size of a × b × c shown in FIG. It shall be 16 mm x 20 mm. Thus, the size of the control transformer PRT is considerably small. Then, the gap G = 25 μm, the controlled winding N R = 12T, and the control winding Nc = 1000T are selected.
Further, the primary side voltage resonance capacitor Cr = 4700 pF, C2 = 0.01 μF, and the filter capacitor CN = 1 μF were selected.
Further, a fast recovery type switching diode D1 and a secondary side rectifier diode Do were each selected to have a rating of 3A / 600V, and a switching element Q1 having a rating of 10A / 800V was selected. .
In addition, after selecting each part in this way, the secondary side DC output voltage Eo has a specification of Eo = 200V.

FIG. 9 shows the fluctuation characteristics of the rectified smoothing voltage Ei (DC input voltage), power factor PF, and AC → DC power conversion efficiency (ηAC → DC) with respect to load power fluctuations for the power supply circuit shown in FIG. . In this figure, with respect to the load power Po, experimental results in a fluctuation range of Po = 30 W to 200 W (maximum load power) are shown. The characteristic when the AC input voltage VAC = 100 V is indicated by a solid line, and the characteristic when the AC input voltage VAC is 230 V is indicated by a wavy line.
As shown in this figure, the level of the DC input voltage (Ei) tends to increase slightly in the light load range, but the overall level is almost constant with respect to the load fluctuation. I understand. When the AC input voltage VAC = 100V, it is substantially constant in the range of Ei = 150V to 140V, and when the AC input voltage VAC = 230V, it is substantially constant in the range of Ei = 320V to 300V.
The power factor PF obtained according to the operation of the power factor correction circuit 10 is such that the load power Po = 50 W to 200 W in the range of the load power Po = 50 W to 200 W, regardless of whether the AC input voltage VAC = 100 V or 230 V. = It is a characteristic that a constant state is maintained at about 0.80. That is, it is shown as an experimental result that the power factor improvement circuit 10 is controlled so that the value of the power factor PF is constant even with respect to load fluctuation.
Further, the AC → DC power conversion efficiency (ηAC → DC) tends to increase as the load power Po becomes a heavy load, and the AC input voltage VAC = 200 W under the condition of the maximum load power Po = 200 W. The measurement result was ηAC → DC = 91.2% at 100 V, and ηAC → DC = 89% at the AC input voltage VAC = 230 V.

FIG. 10 shows fluctuation characteristics of the rectified smoothing voltage Ei (DC input voltage), power factor PF, and AC → DC power conversion efficiency (ηAC → DC) with respect to fluctuations in the AC input voltage VAC. The characteristics shown in this figure show the experimental results when the AC input voltage VAC is varied from 80 V to 280 V over the AC 100 V system to the AC 200 V system under the maximum load power Po = 200 W (maximum load condition).
First, the DC input voltage (Ei) tends to rise as it fluctuates so that the level of the AC input voltage VAC increases.

  The power factor PF is constant at PF = 0.80 in the range of AC input voltage VAC = 80V to 280V. That is, it can be seen that the power factor improvement circuit 10 cancels the reduction of the power factor due to the increase of the AC input voltage and is improved so as to be constant.

  Further, although the AC → DC power conversion efficiency (ηAC → DC) tends to decrease as the AC input voltage VAC increases, the AC input voltage VAC is 92 in the level range corresponding to the AC100V system. % Is maintained.

For example, in the configuration in which the power factor correction circuit 10 is omitted from the power supply circuit of the present embodiment shown in FIG. 1, the AC input voltage is the power factor PF and AC → DC power conversion efficiency (ηAC → DC), respectively. When VAC = 100 V and the maximum load power Po = 200 W, the experimental results of PF = 0.58 and ηAC → DC = 90.8% were obtained.
Therefore, the power supply circuit of the present embodiment adopting the circuit configuration as shown in FIG. 1 has the same AC input voltage and VAC = VAC = as compared with the case of adopting the circuit configuration in which the power factor correction circuit 10 is omitted. Under the conditions of 100 V and load power Po = 200 W, the AC → DC power conversion efficiency (ηAC → DC) is improved by 1.2%. Along with this, the AC input power was reduced by 1.0 W.

Although not shown in FIGS. 5 and 6, for the ripple voltage component ΔEo of the commercial AC power supply period superimposed on the secondary side DC output voltage Eo, the power supply circuit of FIG. 1 has the maximum load power Po = 200 W, It was measured that ΔEo = 80 mV when the AC input voltage VAC = 100V. Further, it was measured that ΔEo = 50 mV when the maximum load power Po = 200 W and the AC input voltage VAC = 230 V.
In contrast, a measurement result of ΔEo = 50 mV was obtained for the power supply circuit in which the power factor correction circuit 10 was omitted from the power supply circuit of FIG.
As described in the background art, for example, when a power factor correction circuit using a power regeneration system is provided for a power supply circuit having a coupling coefficient k of about 0.8 or more of the insulating converter transformer PIT, the ΔEo It increases to 5 to 6 times that of the configuration in which the rate improvement circuit is omitted. On the other hand, in this embodiment, as described above, when the AC input voltage VAC = 100 V, the expansion is about 1.6 times (80/50 (mV)), which is greatly suppressed. Has been. Furthermore, substantially the same characteristics are obtained when the AC input voltage VAC = 230V.
The increase rate of ΔEo, which is about 1.6 times, is within an allowable range that allows sufficient practical use without changing the design such as a significant increase in the capacity of the secondary-side smoothing capacitor. Therefore, the coupling coefficient k set in the insulating converter transformer PIT in the present embodiment should be set in consideration that ΔEo falls within the allowable range. Further, if the degree of coupling of the insulating converter transformer PIT is reduced, the transmission loss between the primary side and the secondary side also increases. Therefore, the coupling coefficient k is a compensation amount of the transmission loss by the secondary side parallel resonant circuit. It should be set in consideration of the trade-off.

The present invention is not limited to the configuration as the embodiment described so far. For example, details such as components such as a common mode noise filter connected to the commercial AC power supply AC, a connection mode, and the like may be appropriately changed.
In the embodiment, a half-wave rectifier circuit is provided for the secondary side parallel resonant circuit. However, under the present invention, for example, in addition to a full-wave rectifier circuit and a double-wave rectifier circuit, a voltage doubler rectifier circuit is provided. A rectifier circuit that generates a secondary side DC output voltage Eo that is a predetermined multiple of the induced voltage level of the secondary winding N2 can also be configured. However, as described above, depending on the configuration of the present invention, even if a half-wave rectifier circuit is provided for the secondary-side rectifier circuit, ZCS against a wide range of load fluctuations is ensured. By using a half-wave rectifier circuit for the secondary rectifier circuit, the effect of reducing the number of parts can be obtained.

In each of the above embodiments, a MOS-FET is selected as a switching element for forming a primary voltage resonant converter. For example, an IGBT (insulated gate bipolar transistor), SIT (electrostatic induction thyristor) is selected. ), Semiconductor elements called various power devices such as bipolar transistors can be employed.
In addition, depending on the selection of these switching elements, any of a system for driving the switching elements, either a self-excited system using a driving IC, or a self-excited system having a self-excited oscillation drive circuit is used. May be adopted.
Further, as a voltage resonance type converter, a push-pull method in which two stone switching elements are alternately turned on / off is known, but the present invention also includes such a push-pull type voltage resonance type converter. Applicable. However, as shown in FIG. 1 as an embodiment, if the single-ended form is adopted, the switching element becomes one stone, which is advantageous in terms of reducing the number of parts, reducing the size and weight, and reducing the cost. It becomes.

It is a circuit diagram which shows the structural example of the switching power supply circuit of embodiment of this invention. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a figure which shows the structural example of the control transformer with which the power supply circuit of embodiment is equipped. It is a figure which shows the structural example of the control transformer with which the power supply circuit of embodiment is equipped. It is a figure which shows the structural example of the control transformer with which the power supply circuit of embodiment is equipped. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the power supply circuit of embodiment by a switching period. It is a wave form diagram (at the time of VAC = 100V) which shows operation of the principal part corresponding to power factor improvement operation of a power circuit of an embodiment by commercial AC power cycle. It is a wave form diagram (at the time of VAC = 230V) which shows operation of the principal part corresponding to power factor improvement operation of a power circuit of an embodiment by commercial AC power cycle. It is a figure which shows the fluctuation characteristic of the rectification | straightening smoothing voltage (DC input voltage) with respect to load fluctuation | variation, power factor, and AC-> DC power conversion efficiency in the power supply circuit of embodiment. It is a figure which shows the fluctuation characteristic of the rectification | straightening voltage (direct current input voltage) with respect to alternating current input voltage fluctuation | variation, a power factor, and AC-> DC power conversion efficiency in the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art. It is a circuit diagram which shows the other structural example of the power supply circuit as a prior art. It is a figure which shows the fluctuation characteristic of the rectification smoothing voltage (DC input voltage) with respect to the load fluctuation | variation of the power supply circuit shown in FIG. 12, a power factor, and AC-> DC power conversion efficiency.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 10 power factor correction circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, Cr primary side voltage resonance capacitor, N1 primary winding, N2 secondary winding Wire, C2 secondary side parallel resonant capacitor, Do rectifier diode, Co (secondary side) smoothing capacitor, CN filter capacitor, PRT control transformer, NR controlled winding, Nc control winding, D1 switching diode

Claims (2)

Rectifying and smoothing means for inputting a commercial AC power supply and generating a rectified and smoothed voltage;
Switching means formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An insulating converter transformer that is set to be equal to or less than a predetermined value for the coupling coefficient between the primary side and the secondary side,
A primary voltage resonance circuit formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a primary side voltage resonance capacitor, the operation of the switching means being a voltage resonance type;
A secondary side parallel resonant circuit formed by connecting a secondary side parallel resonant capacitor in parallel to the secondary winding of the insulating converter transformer;
A secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectifying operation to generate a secondary side DC output voltage;
With power factor improvement means,
The power factor improving means is
A series connection circuit comprising a power factor improving switching element and a high frequency inductor, the power factor improving series connection circuit inserted in series with respect to the rectified current path formed by the rectifying and smoothing means;
The controlled winding, which is the high-frequency inductor, and the control winding are wound so that the inductance of the controlled winding is variable according to the level of the control current that is a direct current flowing through the control winding. An operating control transformer,
Control current variable control means that operates so as to vary the level of the control current according to the level of the rectified smoothing voltage level,
Formed by connecting the end of the primary side voltage resonance circuit to the connection point of the power factor improving switching element and the high frequency inductor,
A switching power supply circuit. A half-wave rectifier circuit is provided as a rectifier circuit that forms the secondary side DC output voltage generation means,
The switching power supply circuit according to claim 1.

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