JP4595403B2 - Switching power supply circuit - Google Patents

Description

  The present invention relates to a switching power supply circuit having a power factor correction function.

The present applicant has previously proposed various power supply circuits including a resonance type converter on the primary side. Various power supply circuits configured with a power factor correction circuit for improving the power factor of the resonant converter have also been proposed.
As the above power factor correction circuit, the applicant of the present invention previously interrupted the rectified current flowing in the rectified current path by feeding back the switching output of the resonant converter to the rectified current path, thereby increasing the conduction angle of the AC input current. Various configurations have been proposed to expand and improve power factor.
However, when such a power factor correction circuit has a configuration in which, for example, the switching output of the resonant converter is directly fed back to the rectified current path (see, for example, Patent Document 1 and FIG. 6), the secondary side DC It has been found that the ripple voltage of the commercial power cycle of the output voltage is significantly increased than before the power factor improvement.
In the case of such a power factor improving configuration, the primary side resonance circuit is directly connected to the rectification current path of the commercial AC power supply. For this reason, the current of the commercial AC current cycle is superimposed on the current flowing through the primary winding of the insulating converter transformer, which is the inductance of the resonance circuit. As a result of the operation of the rectifying / smoothing circuit on the secondary side in a state where this overlap is transmitted to the secondary side of the insulation converter transformer, the ripple voltage of the commercial power cycle of the secondary DC output voltage is increased as described above. Will do. For example, when the power factor PF is set to be about 0.8, the level of the ripple voltage increases to 5 to 6 times.

  One possible countermeasure is to set a high control gain for stabilization control. However, if the level of the ripple voltage increases to 5 to 6 times as described above, even if the control gain is increased to the limit, it is insufficient to effectively suppress the ripple. Therefore, it is conceivable that the capacitance of the smoothing capacitor for smoothing the secondary side DC output voltage is further increased 5 to 6 times after the control gain is set to a certain limit. However, the selection of such a smoothing capacitor greatly increases the cost, and practical use is not practical. Since such a countermeasure is not practical, it may be difficult to adopt it in a device that strictly requires, for example, a ripple voltage to be below a certain level.

Therefore, the applicant of the present invention has proposed various types of power factor correction circuits that feedback the switching output in which the ripple voltage is suppressed.
FIG. 7 is a circuit diagram showing an example of a switching power supply circuit configured based on the invention of the power factor improvement circuit of the switching output period type in which the ripple voltage reduction is attempted previously filed by the present applicant. is there. This power supply circuit is configured such that a power factor correction circuit is provided for a voltage resonance type switching converter.

  In the switching power supply circuit shown in this figure, as a noise filter for removing common mode noise, a pair of common mode choke coils CMC and two across capacitors CL are provided for the line of the commercial AC power supply AC. Yes.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by a bridge rectifier circuit Di composed of four low-speed rectifier diodes, and the rectified output is charged to the smoothing capacitor Ci via the power factor correction circuit 20. The As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. The configuration and operation of the power factor correction circuit 20 will be described later.

  In this figure, a voltage resonance type switching converter that performs a switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage has, for example, one switching element Q1 as a high breakdown voltage bipolar transistor. That is, a so-called single-end system is used.

A switching drive signal output from the oscillation / drive control circuit 21 is input to the base of the switching element Q1.
The collector of switching element Q1 is connected to the positive terminal of smoothing capacitor Ei via primary winding N1 of insulating converter transformer PRT. That is, the DC input voltage (Ei) is supplied through the primary winding N1. The emitter is grounded.
A clamp diode DD is connected between the base and emitter of the switching element Q1.

A primary side voltage resonance capacitor Cr is connected between the collector and emitter of the switching element Q1.
The primary side voltage resonance capacitor Cr forms a voltage resonance circuit (parallel resonance circuit) with the leakage inductance L1 of the primary winding N1 of the insulating converter transformer PIT. A voltage resonance type operation is obtained as the switching operation of the switching element Q1 by the resonance action of the voltage resonance circuit. For this reason, a sinusoidal pulse waveform is obtained as the voltage VQ1 between the collector and emitter of the switching element Q1 during the period when the switching element is turned off.

  The oscillation / drive control circuit 21 supplies a base current as a switching drive signal to the base of the switching element Q1 in order to drive the switching element Q1. As a result, the switching element Q1 performs a switching operation at a switching frequency corresponding to the period of the switching drive signal. The oscillation / drive control circuit 21 is also configured to vary the switching frequency of the switching element Q1 in order to stabilize the secondary side DC output voltage.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
As the structure of the insulating converter transformer PIT, for example, an EE type core in which an E type core made of a ferrite material is combined is provided. Then, after the winding part is divided on the primary side and the secondary side, the primary winding N1 (and tertiary winding N3) and the secondary winding N2 are wound around the central magnetic leg of the EE type core. Disguise.
In addition, a gap of about 1.0 mm is formed with respect to the central magnetic leg of the EE type core of the insulating converter transformer PIT, whereby k = 0.80 between the primary side and the secondary side. A coupling coefficient k of about ˜0.85 is obtained. This degree of coupling coefficient k is a degree of coupling that can be regarded as loose coupling, and accordingly, a saturated state is hardly obtained. Further, the value of the coupling coefficient k becomes a setting element of the leakage inductance (L1).

  One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end is connected to the positive electrode (rectified smoothing voltage Ei) of the smoothing capacitor Ci.

  On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, so that it depends on the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2. A secondary side parallel resonant circuit is formed. By the resonance operation of the parallel resonance circuit, the alternating voltage excited in the secondary winding N2 becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side.

  That is, in this power supply circuit, the primary side is provided with a primary side voltage (parallel) resonance circuit for making the switching operation a voltage resonance type, and the secondary side is also provided with a parallel for obtaining a voltage resonance operation in the rectifier circuit system. A resonant circuit is provided. In the present specification, a switching converter configured to operate with a plurality of resonance circuits in this way is also referred to as a “composite resonance switching converter”.

  A full-wave rectifier circuit including a bridge rectifier circuit DBR and a smoothing capacitor Co is connected to the secondary side parallel resonant circuit formed as described above. This full-wave rectifier circuit generates a DC output voltage Eo as a voltage across the smoothing capacitor Co by inputting an alternating voltage as a resonant output of the secondary side parallel resonant circuit and performing a rectifying and smoothing operation. The DC output voltage Eo is supplied to the load and is input to the oscillation / drive control circuit 21 as a detection voltage for constant voltage control.

In the circuit shown in FIG. 7, the configuration of the full-wave rectifier circuit for the secondary side rectifier circuit is that zero current switching (ZCS) of the switching element Q1 corresponds to a wide variation in load power. ) Is ensured.
As described above, in the power supply circuit of FIG. 7, a coupling coefficient k of the insulating converter transformer PIT is 0.8 or more. Although this is a loose coupling, it can be said that a certain degree of coupling is maintained. Under such a degree of coupling, for example, if the secondary side rectifier circuit is a half-wave rectifier circuit, the above-described ZCS operation may not be properly maintained.

  In the oscillation / drive control circuit 2, as a constant voltage control operation, the switching frequency is varied according to the level of the DC output voltage Eo input as described above. The period during which the element Q1 is off is fixed, and the period during which the element Q1 is on is variably controlled. That is, in this power supply circuit, as the constant voltage control operation, the switching frequency is variably controlled to control the resonance impedance for the switching output, and at the same time, the conduction angle control (PWM control) of the switching element in the switching cycle is also performed. Can be seen. This complex control operation is realized by a set of control circuit systems.

The power factor improvement is performed by the power factor improvement circuit 20.
The power factor correction circuit 20 shown in this figure includes rectifier diodes D1 and D2, a filter capacitor CN, an inductor LS, and a tertiary winding N3. The rectifier diode D1 is selected as a low speed type, and the rectifier diode D2 is selected as a high speed type (high speed recovery type) diode element.
The tertiary winding N3 is formed so as to wind up the winding end end side of the primary winding N1 in the insulating converter transformer PIT.

As shown in the figure, the rectifier diode D1 is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Further, between the positive output terminal of the same bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci, a series connection circuit of rectifier diode D2-inductor Ls-tertiary winding N3 due to the parallel connection with the rectifier diode D1. Is inserted.
In this case, the filter capacitor CN is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. In other words, the filter capacitor CN is connected in parallel to both the rectifier diode D1 and the series connection circuit of the rectifier diode D2, the inductor Ls, and the tertiary winding N3.
The filter capacitor CN and the inductor Ls inserted as described above form a normal mode LC low-pass filter, which prevents high-frequency noise of the switching period from flowing into the commercial AC power supply line.

  Depending on the switching operation of the switching element Q1, a sinusoidal resonance pulse voltage can be obtained as the voltage across the switching element Q1 (collector-emitter voltage) VQ1 during the off period. The alternating voltage due to the resonance pulse voltage is transmitted (induced) from the primary winding N1 of the insulating converter transformer PIT to the tertiary winding N3 of the power factor correction circuit 20. The alternating voltage obtained at the tertiary winding N3 is also applied to the rectifier diode D2 via the inductor Ls.

In this case, the rectified current flowing from the positive output terminal of the bridge rectifier circuit Di flows into the smoothing capacitor Ci via the rectifier diode D1 (anode → cathode), the rectifier diode D2 (anode → cathode) → the inductor Ls → It flows through a path flowing into the smoothing capacitor Ci via the tertiary winding N3.
That is, during the period when the AC input voltage VAC is in the vicinity of the peak, the low-speed rectifier diode D1 is turned on, and the rectified current from the bridge rectifier circuit Di is passed through the rectifier diode D1 to the smoothing capacitor Ci. To flow into. Further, during the period in which the AC input voltage VAC is lower than the vicinity of the peak, the rectified current from the bridge rectifier circuit Di passes through the series connection circuit of the rectifier diode D2-inductor Ls-tertiary winding N3 to the smoothing capacitor Ci. However, as described above, an alternating voltage is applied to the rectifier diode D2, so that the rectified current flowing through the path is switched and intermittently operated. Become. In this way, by configuring so that the rectified current is divided in the two rectified current paths, the conduction loss in the rectifier diode in the power factor correction circuit is reduced, and the power conversion efficiency can be increased accordingly. .
As described above, the operation of switching the rectifier diode D2 during the period in which the AC input voltage VAC is at a predetermined level lower than the vicinity of the peak is, for example, the ratio of the winding of the tertiary winding N3 to the primary winding N1. It is obtained by setting.

  In this manner, the switching output (voltage resonance pulse) of the switching element Q1 is fed back as a voltage to the rectified current path in the power factor correction circuit 20 through the tertiary winding N3. As a result of switching the rectified current flowing through the rectified current path to be switched, the conduction angle of the AC input current IAC that is the source of the rectified current is expanded, and the power factor is improved. The method of the power factor correction circuit configured to feed back the switching output as a voltage and thereby improve the power factor is also referred to as a voltage feedback method.

Then, depending on the configuration of the power factor correction circuit 20 shown in FIG. 7, the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage is suppressed.
In the power factor correction circuit 20, the switching output is voltage-feedbacked by the tertiary winding N3. In other words, it can be said that when the switching output obtained in the voltage resonance circuit on the primary side is fed back to the rectification current path of the commercial AC power supply, it is indirectly performed by the alternating voltage excited by the tertiary winding N3. As a result, the ripple component of the commercial AC power supply period is less likely to be induced from the tertiary winding N3 to the primary winding N1. Further, by connecting the inductor Ls in series with the tertiary winding N3, the tertiary winding is equivalent to the primary winding with a low degree of coupling due to the combined inductance of the tertiary winding and the inductor. There will be. As a result, the ripple component of the commercial AC power supply period is further less likely to be induced in the primary winding. As a result, the ripple voltage of the commercial AC power supply period superimposed on the secondary side DC output voltage is effectively suppressed.

The circuit diagram of FIG. 8 shows another example of the switching power supply circuit configured based on the invention previously filed by the present applicant. The basic configuration of the power supply circuit shown in this figure is the same as the power supply circuit of FIG. 7, in which a primary side single-ended voltage resonance converter and a secondary side parallel (voltage) resonance circuit are combined. A power factor correction circuit based on a voltage feedback system is added to the composite resonance type converter. In addition, the insulating converter transformer PIT is configured to obtain a degree of coupling of about a coupling coefficient k = 0.80 to 0.85, as in FIG.
In this figure, the same parts as those in FIG.

The power factor improvement circuit 20A shown in this figure includes a low speed type rectifier diode D1, a high speed type (high speed recovery type) rectifier diode D2, a filter capacitor CN, and a magnetic coupling for power factor improvement as shown in the figure. It comprises a transformer MCT.
The magnetically coupled transformer MCT has a structure in which a primary winding Np and a secondary winding Ns are wound around a core so as to be magnetically coupled. The magnetically coupled transformer MCT in this case has a so-called divided bobbin in which divided winding positions are formed, and the primary winding Np and the secondary winding Ns are wound around different winding positions. Thus, a predetermined coupling coefficient that is loosely coupled is obtained as the degree of coupling between the primary side and the secondary side.

The winding start end portion of the primary winding Np is connected to the collector of the switching element Q1 via the serial connection of the primary winding N1 of the insulating converter transformer PIT, and the winding end portion is connected to the cathode of the rectifier diode D1. The The anode of the rectifier diode D1 is connected to the positive terminal of the smoothing capacitor Ci.
The winding end end of the secondary winding Ns is connected to the cathode of the rectifier diode D2, and the winding start end is connected to the positive terminal of the smoothing capacitor Ci. The anode of the rectifier diode D2 is connected to the positive output terminal of the bridge rectifier circuit Di.
Further, the filter capacitor CN is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci, so that the rectifier diode D1, the rectifier diode D2 and the secondary winding Ns are connected in series. Are connected in parallel to both.

By forming the power factor correction circuit 20A according to the above connection mode, the path through which the rectified current output from the positive output terminal of the bridge rectifier circuit Di flows into the smoothing capacitor Ci is the path through the rectifier diode D1, and the rectifier There are two paths through the series connection circuit of the diode D2-secondary winding Ns. That is, also in the power factor improvement circuit 20A, the rectified current can flow through two paths in the same manner as the power factor improvement circuit 20 in FIG.
Then, by performing the switching operation of the switching element Q1, as described above, as the voltage VQ1 across the switching element Q1, a resonance pulse voltage can be obtained in the off period. However, in the power factor correction circuit 20A, the resonance frequency described above is obtained. The alternating voltage due to the pulse voltage is first transmitted to the primary winding Np of the magnetic coupling transformer MCT, and further transmitted to the secondary winding Ns by the magnetic coupling of the magnetic coupling transformer MCT. As a result, an alternating voltage is applied to the rectifier diode D2. That is, the power factor correction circuit 20A employs a configuration in which the switching output is fed back as a voltage to the rectified current path through the magnetic coupling by the magnetic coupling transformer MCT as a voltage feedback system.
Incidentally, in the power factor correction circuit 20A, the primary winding Np of the magnetic coupling transformer MCT corresponds to the tertiary winding N3 in the power factor improvement circuit 20 of FIG. 7, and the secondary winding Ns is the power factor of FIG. This corresponds to the inductor Ls in the improvement circuit 20.

  Also in the power factor correction circuit 20A, the low-speed rectifier diode D1 conducts during the period when the AC input voltage VAC is in the vicinity of the peak, and the rectified current from the bridge rectifier circuit Di is passed through the rectifier diode D1. It is made to flow through the smoothing capacitor Ci. Further, during the period in which the AC input voltage VAC is lower than the vicinity of the peak, the rectified current from the bridge rectifier circuit Di is switched (interrupted) by the rectifier diode D2, so that the rectifier diode D2-secondary winding Ns. Is passed through the smoothing capacitor Ci via a series connection circuit. As a result, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.

  In the power factor correction circuit 20A shown in FIG. 8, the switching output obtained in the voltage resonance circuit on the primary side is fed back to the rectification current path through the magnetic coupling of the magnetic coupling transformer MCT. . In this case, since the magnetic coupling transformer MCT is formed loosely coupled, the ripple component of the commercial AC power supply period induced from the secondary winding Ns to the primary winding Np is reduced. Thereby, also in the power supply circuit shown in FIG. 8, the effect of suppressing the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage is obtained.

FIG. 9 shows, as the characteristics of the power supply circuit shown in FIG. 8, power factor (PF), rectified smoothing voltage Ei, AC → DC power corresponding to load fluctuations from load power Po = 0 to 200 W (maximum load power). The fluctuation characteristics of conversion efficiency (ηAC → DC) are shown. In addition, in this figure, for comparison, characteristics of a power supply circuit that does not have a power factor improvement function by omitting the power factor improvement circuit 20A from the circuit configuration shown in FIG. 8 are also shown. In FIG. 9, the characteristic of the power supply circuit including the power factor correction circuit 20A as shown in FIG. 8 is indicated by a solid line, and the characteristic of a power supply circuit having a configuration in which the power factor improvement circuit 20A is omitted is indicated by a broken line.
As shown in this figure, with regard to the power factor (PF), the power supply circuit as shown in FIG. 8 has a configuration in which the power factor correction circuit 20A is omitted by the power factor improvement operation by the power factor improvement circuit 20A. You can see that it is higher than This figure also shows that a characteristic that is almost constant at PF = 0.8 is obtained in the range of the load power Po = 50 W to 200 W.
Further, as a tendency in the entire load range, the rectified and smoothed voltage Ei is higher in the power supply circuit as shown in FIG. 8 than in the power supply circuit in which the power factor correction circuit 20A is omitted. This is because the smoothing capacitor Ci is charged while the switching output is fed back to the rectification current path by the power factor correction circuit 20A, and the charging current level to the smoothing capacitor Ci increases accordingly. .
As for the AC → DC power conversion efficiency (ηAC → DC), the power supply circuit as shown in FIG. It has decreased by about%. This is because the magnetic coupling transformer MCT for feeding back the switching output to the rectification current path in the power factor correction circuit 20A is loosely coupled, so that power can be transmitted between the primary winding Np and the secondary winding Ns. This is due to loss.

JP 2001-95247 A JP 2002-34250 A

The power supply circuit as shown in FIGS. 7 and 8 has the following problems.
First, as described with reference to FIGS. 7 and 8, these power supply circuits are provided with the tertiary winding N3 in order to eliminate the problem of an increase in the ripple of the commercial AC power supply period superimposed on the secondary DC output voltage. Is added and inductors are connected in series, or a magnetic coupling transformer MCT is added.
For this reason, as the power factor improving circuit configured to feed back the switching output to the rectified current, the number of parts increases, resulting in an increase in cost, an increase in circuit scale, and an increase in weight.

Further, as described with reference to FIG. 9, in the power supply circuit shown in FIG. 8, the AC → DC power conversion efficiency (ηAC → DC) is reduced by about 1% as compared with the configuration without the power factor correction circuit 20A. Has produced. Such a tendency to decrease the power conversion efficiency occurs similarly in the power supply circuit of FIG. In the power supply circuit of FIG. 7, transmission loss occurs in the process of feeding back the switching output of the primary winding N1 as a voltage via the tertiary winding N3.
That is, in any of the power supply circuits shown in FIGS. 7 and 8, transmission loss occurs in an additional component for feeding back the switching output as a voltage, which is disadvantageous as a total power conversion efficiency. As a result, as described above, the power conversion efficiency is lower than that of the power supply circuit having no power factor correction circuit.

In view of the above-described problems, the present invention is configured as a switching power supply circuit as follows.
That is, a rectifying / smoothing means that generates a rectified and smoothed voltage by inputting a commercial AC power supply, a switching means that includes a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage, and a switching driving device Switching driving means.
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. An insulating converter transformer that is formed and set to be equal to or smaller than a predetermined coupling coefficient between the primary side and the secondary side, at least a leakage inductance component including a primary winding of the insulating converter transformer, and a capacitance of the primary side voltage resonance capacitor And a primary-side voltage resonance circuit having a voltage resonance type operation of the switching means.
In addition, a secondary side parallel resonant circuit formed by connecting a secondary side parallel resonant capacitor in parallel to the secondary winding of the insulating converter transformer is provided.
In addition, a secondary side DC output voltage generation unit configured to generate a secondary side DC output voltage by inputting an alternating voltage excited to the secondary winding of the insulating converter transformer and performing a rectification operation is provided. .
Then, the primary side resonance current obtained in the primary side voltage resonance circuit is regenerated as electric power by the switching operation of the switching means, and is fed back to the smoothing capacitor forming the rectifying and smoothing means. Power factor improving means comprising a power factor improving switching element that switches the rectified current obtained by the rectifying operation by the rectifying and smoothing means intermittently according to the electric power is provided.

According to the above configuration, the switching power supply circuit of the present invention includes the voltage resonance type converter as the primary side switching converter. In addition, the power factor is improved using a power regeneration system.
The power regeneration type power factor correction circuit has a fundamental circuit configuration that greatly increases the ripple of the commercial AC power supply cycle superimposed on the switching output, and as a result, commercial power superimposed on the secondary DC output voltage. The ripple of the AC power supply cycle also tends to increase significantly.
However, in the present invention, the ripple appearing in the secondary DC output voltage is reduced by weakening the degree of coupling between the primary side and the secondary side so that the magnetic flux density of the insulating converter transformer becomes not more than a predetermined value as described above. Can be sufficiently suppressed.
In addition, if the coupling degree of the insulating converter transformer is weakened, the transmission loss between the primary side and the secondary side will increase, but in the present invention, the secondary winding and the secondary side on the secondary side. By providing a secondary side parallel resonance circuit composed of a parallel resonance capacitor, resonance operation can be obtained on the secondary side. By this resonance operation, power is transmitted from the primary side to the secondary side in such a way that transmission loss between the primary side and the secondary side is compensated.

In this way, the present invention makes it possible to obtain a power supply circuit in which the ripple voltage of the commercial AC power supply cycle of the secondary side DC output voltage is sufficiently suppressed while having a power factor improvement circuit by a power regeneration system. Is done.
This ripple suppression effect is obtained not by adding circuit components in particular but by adopting a structure that can obtain a coupling degree (coupling coefficient) of a predetermined value or less for the insulating converter transformer. In addition, the power factor improvement circuit itself by the power regeneration method can be formed by a smaller number of simple component elements than a power factor improvement circuit such as a voltage feedback method. As a result, according to the present invention, a power supply circuit having a power factor correction circuit can be reduced in size / weight and can be obtained at a low cost.
In addition, in the power factor improvement circuit using the power regeneration system, there is less loss in the system that feeds back the switching output than in the power factor improvement circuit such as the voltage feedback system. Further, according to the present invention, the secondary side parallel resonance circuit is provided to compensate for the transmission loss between the primary side and the secondary side due to the reduced degree of coupling of the insulating converter transformer. As a result, good characteristics can be obtained in terms of power conversion efficiency.

The circuit diagram of FIG. 1 shows a configuration example of a power supply circuit as the best mode (embodiment) for carrying out the present invention. The power supply circuit shown in this figure employs a basic configuration including a power factor correction circuit for a voltage resonance type switching converter.
In the switching power supply circuit shown in this figure, first, a pair of common mode choke coils CMC and two across capacitors CL are inserted into the commercial AC power supply AC line as shown. The common mode choke coil CMC and the across capacitors CL and CL form a noise filter that removes common mode noise superimposed on the line of the commercial AC power supply AC.

  The commercial AC power supply AC (AC input voltage VAC) is rectified by a bridge rectifier circuit Di formed by connecting four low-speed rectifier diodes as shown in the figure, and the rectified output thereof passes through the power factor correction circuit 10. In this way, the smoothing capacitor Ci is charged. As a result, the rectified and smoothed voltage Ei is obtained as the voltage across the smoothing capacitor Ci. This rectified and smoothed voltage Ei becomes a DC input voltage for the subsequent switching converter. The configuration and operation of the power factor correction circuit 10 will be described later.

  In this figure, a voltage resonance type switching converter that performs a switching operation by inputting the rectified and smoothed voltage Ei as a DC input voltage has a structure including, for example, one MOS-FET switching element Q1. That is, a so-called single-ended voltage resonance converter is obtained.

A switching drive signal (voltage) output from the oscillation / drive circuit 2 is applied to the gate of the switching element Q1.
The drain of the switching element Q1 is connected to the winding start end of a primary winding N1 of an insulating converter transformer PIT described later. The winding end end of the primary winding N1 is connected to the positive terminal of the smoothing capacitor Ei through the series connection of the high frequency inductor L10 of the power factor correction circuit 10. That is, in this case, since the power factor correction circuit 10 is inserted, the DC input voltage (Ei) is supplied to the switching element Q1 via a series connection of the high frequency inductor L10 and the primary winding N1. It is like that. The source of the switching element Q1 is grounded to the primary side ground.

  Since the switching element Q1 in this case is a MOS-FET, as shown in the figure, it incorporates a body diode DD so as to be connected in parallel between the source and the drain. As the body diode DD, the anode is connected to the source of the switching element Q1, and the cathode is connected to the drain of the switching element Q1. The body diode DD forms a path for flowing a switching current in the reverse direction, which is generated by the on / off operation (switching operation) of the switching element Q1.

A primary side voltage resonant capacitor Cr is connected in parallel between the drain and source of the switching element Q1.
The primary side voltage resonance capacitor Cr forms a parallel resonance circuit (voltage resonance circuit) for the switching current flowing through the switching element Q1 by its own capacitance and the leakage (leakage) inductance L1 of the primary winding N1 of the insulating converter transformer PIT. . When this voltage resonance circuit performs a resonance operation, a voltage resonance type operation can be obtained as a switching operation of the switching element Q1. In response to this, a sinusoidal pulse waveform is obtained during the off-period of the voltage across the switching element Q1 (drain-source voltage) VQ1.

  The oscillation / drive circuit 21 is a switching voltage which is a voltage for switching and driving the MOS-FET based on an oscillation circuit and an oscillation signal obtained by the oscillation circuit, for example, in order to drive the switching element Q1 by a separate excitation type. A drive signal is generated and applied to the gate of the switching element Q1. As a result, the switching element Q1 continuously performs on / off operations according to the switching frequency corresponding to the cycle of the switching drive signal. That is, a switching operation is performed.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side.
FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained. In this case, as an actual EE type core, for example, EER-40 is selected on the premise that it corresponds to a load condition as described later and a rated level condition of an AC input voltage.

In addition, a gap G having a gap length of about 1.4 mm or more is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with k = 0.8 or less is obtained. That is, the prior art is more loosely coupled than the isolated converter transformer PIT of the power supply circuit shown in FIGS. Note that k = 0.75 was set as the actual coupling coefficient k.
The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  As described above, the winding end end of the primary winding N1 of the insulating converter transformer PIT is connected to the positive terminal of the smoothing capacitor Ci through the high frequency inductor L10 in the power factor correction circuit 10, and the winding start end is Connected to the drain of switching element Q1. As a result, the switching output of the switching element Q1 is transmitted to the primary winding N1, and an alternating voltage is generated in the primary winding N1.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2.
A secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2. Thus, a secondary side parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2. Due to the resonance operation of the secondary side parallel resonance circuit, the alternating voltage excited in the secondary winding N2 becomes a resonance voltage close to a sine wave. That is, a voltage resonance operation is obtained on the secondary side.

  According to the description so far, the power supply circuit of the present embodiment includes a primary side voltage (parallel) resonance circuit for making the switching operation a voltage resonance type on the primary side, and a voltage in the rectifier circuit system on the secondary side. A parallel resonance circuit for obtaining a resonance operation is provided. That is, the configuration as a composite resonance type switching converter is adopted.

In the present embodiment, a half-wave rectifier circuit including one rectifier diode Do and a smoothing capacitor Co is connected as the secondary-side rectifier circuit. In this case, the anode of the rectifier diode Do is connected to the winding end of the secondary winding N2, and the cathode of the rectifier diode Do is connected to the positive terminal of the smoothing capacitor Co. The winding start end of the secondary winding and the negative terminal of the smoothing capacitor Co are both connected to the secondary side ground.
This half-wave rectifier circuit performs rectification during the half-wave period of the alternating voltage obtained as the output of the secondary side parallel resonant circuit and charges the smoothing capacitor Co. Thereby, the secondary side DC output voltage Eo is obtained as the voltage across the smoothing capacitor Co.
The secondary side DC output voltage Eo is supplied to the load. Further, it branches and is output as a detection voltage to the control circuit 1.

  Further, in FIG. 1, the primary winding N1 and the secondary winding N2 of the insulating converter transformer PIT are described so that the respective winding start ends are at the same lower position with respect to the transformer symbol. However, this indicates that the mutual inductance of the primary winding N1 and the secondary winding N2 is a connection mode in which the polarity is positive. In the added polarity, the rectifier diode Do conducts on the secondary side to perform the rectification operation in correspondence with the switching element Q1 being on.

The control circuit 1 supplies a detection output corresponding to the level change of the input secondary side DC output voltage Eo to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching element Q1 such that the switching frequency is varied in accordance with the input detection output of the control circuit 1.
The variable operation of the switching frequency in the control circuit 1 is an operation in which the period during which the switching element Q1 is off is constant and the period during which the switching element Q1 is on is variably controlled. Such a control operation is a constant voltage control operation, in which the switching frequency is variably controlled to control the resonance impedance for the switching output. At the same time, the conduction angle control (PWM (Pulse Width Moduration) of the switching element in the switching period is performed. ) Control) can also be seen as performing. This composite control operation is realized by a set of control circuit systems.

  By variably controlling the switching frequency and conduction angle of the switching element Q1 as described above, the primary side and secondary side resonance impedances and the power transmission effective period in the power supply circuit are changed, and the primary winding of the insulating converter transformer PIT is changed. The amount of power transmitted from the line N1 to the secondary winding N2 (N2A, N2B) side also changes, but this cancels the level fluctuation of the secondary side DC output voltage Eo, so that the secondary side DC output An operation for controlling the level of the voltage Eo is obtained. That is, the secondary side DC output voltage Eo is stabilized.

Next, the power factor correction circuit 10 will be described.
The power factor correction circuit 10 of the present embodiment is inserted into a rectification current path between a bridge rectification circuit Di that generates a rectified smoothing voltage Ei (DC input voltage) from a commercial AC power supply AC and a smoothing capacitor Ci. And includes a filter capacitor CN, a high-frequency inductor L10, and a high-speed (high-speed recovery type) switching diode D1 (power factor improving switching element).

In the power factor correction circuit 10, first, the anode of the switching diode D1 is connected to the positive output terminal of the bridge rectifier circuit Di, and the cathode is connected to the positive terminal of the smoothing capacitor Ci via a series connection of the high frequency inductor L10. Connected to.
In this case, a series connection circuit of a switching diode D1 (anode → cathode) and a high frequency inductor L10 is inserted into the rectified current path between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Will be. A filter capacitor CN is connected in parallel to the series connection circuit of the switching diode D1 and the high frequency inductor L10. In this case, the filter capacitor CN serves to suppress normal mode noise generated in the rectified current path due to feedback of the switching output.

  In this case, the end of winding end of the primary winding N1 is connected to the connection point between the cathode of the switching diode D1 and the high-frequency inductor L10. The operation is such that the resonance current obtained in the resonance circuit (L1 // Cr) is regenerated as power and fed back to the smoothing capacitor Ci (rectification current path) through the magnetic coupling (ie, the high frequency inductor L10). . That is, the power factor correction circuit 10 employs a magnetic coupling type power regeneration system configuration.

The waveform diagram of FIG. 3 shows the switching operation of the switching element Q1 by the switching period. Here, a case where the switching frequency fs = 100 KHz is shown.
In this figure, a period from the period t2 to t0 is an on period in which the switching element Q1 is turned on, and a period from the period t0 to t2 is an off period in which the switching element Q1 is turned off.
The both-ends voltage (drain-source voltage) VQ1 of the switching element Q1 has a waveform that generates a sine wave-like resonant voltage pulse in the on period (t2 to t0) and a zero level in the off period (t0 to t2).

Here, in the ON period (t2 to t0) of the switching element Q1, the primary side resonance current Io flowing through the primary winding N1 of the insulating converter transformer PIT is smoothing capacitor Ci → high frequency inductor L10 → primary winding N1 → switching element. It flows in the forward direction through the path of Q1 (drain → source). Due to this current, electromagnetic energy is accumulated in the series connection circuit of the primary winding N1 and the high frequency inductor L10.
Subsequently, when the switching element Q1 is turned off at time t0, the flow of the primary resonance current Io in the forward direction is maintained by the electromagnetic energy accumulated in the series connection circuit of the primary winding N1 and the high frequency inductor L10, and the primary side The voltage resonance capacitor Cr is charged. This charging operation is performed during the period t0 to t1, and this charging current increases the potential at both ends of the primary side voltage resonance capacitor Cr and also increases the voltage VQ1 at both ends of the switching element Q1. .
The next period t1 to t2 is a period in which the charge stored in the primary side voltage resonance capacitor Cr accumulated in the period t0 to t1 is discharged. By this discharge operation, the primary side resonance current Io is changed to the primary side voltage resonance. The current flows in the reverse path of the capacitor Cr → the primary winding N1 → the high frequency inductor L10 → the smoothing capacitor Ci. That is, the primary side resonance current Io obtained by discharging the charge of the primary side voltage resonance capacitor Cr is regenerated in the smoothing capacitor Ci. In addition, since current flows through this path, electromagnetic energy is accumulated in the series connection circuit of the primary winding N1 and the high-frequency inductor L10 even at this time.
In the period t2 to t3 immediately after the switching element Q1 ends the off state, the body diode DD of the switching element Q1 is turned on. For this reason, the primary resonance current Io in the reverse direction maintained by the electromagnetic energy accumulated in the series connection circuit of the primary winding N1 and the high frequency inductor L10 during the period t1 to t2 is the body diode DD (anode → cathode). It flows through the path of the primary winding N1, the high frequency inductor L10, and the smoothing capacitor Ci. That is, even during this period, the operation of regenerating the primary side resonance current Io to the smoothing capacitor Ci is obtained.
In the period t3 to t0, the switching element Q1 is turned on, and the primary side resonance current Io flows from the smoothing capacitor Ci → the high frequency inductor L10 → the primary winding N1 → the switching element Q1 (drain → source). The That is, the forward (positive polarity) switching output current IQ1 flows.
According to such a switching operation, when the voltage resonance type converter and the power regeneration type power factor correction circuit 10 are combined, the periods t1 to t3 in FIG. 3 are power regeneration periods in which the power regeneration operation can be obtained. That's true.

FIG. 4 shows the operation of the power factor correction circuit 10 by the commercial AC power supply cycle.
Here, it is assumed that an AC input voltage VAC of 50 Hz is input as shown. When the AC input voltage VAC is input and the power supply circuit shown in FIG. 1 operates, the power factor correction circuit 10 also applies the primary side resonance current Io to the smoothing capacitor Ci as described above. The operation of regenerating and returning as electric power is performed.

As the power regeneration is performed as described above, the voltage V1 between the connection point between the primary winding N1, the controlled winding NR, and the cathode of the switching diode D1 and the primary side ground is the AC input current. During a period when IAC is non-conducting and no rectified current flows, the envelope waveform has a constant level in which an alternating voltage component due to the switching period is superimposed on the rectified smoothed voltage Ei. On the other hand, during the period in which the AC input current IAC conducts and the rectified current flows, the rectified output voltage is obtained as a waveform that is superimposed in a sine wave form from the constant level.
Accordingly, the current IL1 flowing into the smoothing capacitor Ci from the high-frequency inductor L10 flows as an alternating waveform with a constant amplitude according to the switching period during the non-conduction period of the AC input current IAC, and corresponds to the conduction period of the AC input current IAC. Therefore, it flows in a state where a sinusoidal peak is obtained.

  The high-speed (high-speed recovery) switching diode D1 conducts when, for example, the positive / negative absolute value of the AC input voltage VAC is about ½ or more of the peak value, but the alternating voltage component of the voltage V1. As a result, the rectified current is switched and intermittently operated. That is, the rectified current I1 flowing from the positive output terminal of the bridge rectifier circuit Di to the switching diode D1 is obtained as an approximately sine wave alternating waveform.

  The conduction period of the rectified current I1 flowing with such a waveform also flows during a period when the rectified output voltage level output from the bridge rectifier circuit Di is lower than the voltage across the smoothing capacitor Ci. The conduction period of the AC input current IAC shown in FIG. 4 also substantially coincides with the conduction period of the rectified output current I1. That is, the conduction angle of the AC input current IAC is larger than that without the power factor correction circuit, and the waveform of the AC input current IAC approaches the waveform of the AC input voltage VAC. Yes. That is, the power factor is improved.

Here, the power factor improvement circuit 10 of the power supply circuit according to the present embodiment is configured as the third winding N3 or the power factor improvement circuits 20 and 20A of the power supply circuit shown in FIGS. It does not employ a configuration in which a primary side resonance current by a loosely coupled magnetic coupling transformer MCT or the like is fed back as a voltage. That is, as a circuit form, in accordance with the power regeneration system, the end of the primary winding N1 through which the primary side resonance current Io flows is connected to the rectification current path (switching diode D1 and high frequency inductor L10 in the power factor correction circuit 10). Connection point).
However, in the power supply circuit of the present embodiment, even if the configuration of the power factor correction circuit that performs direct feedback of the switching output is adopted, the commercial AC power supply cycle superimposed on the secondary side DC output voltage Eo The increase in ripple is greatly suppressed to the extent that there is no practical problem.
This is because the degree of coupling of the insulating converter transformer PIT is lowered to a coupling coefficient k = 0.8 or less (k = about 0.75), compared to the insulating converter transformer PIT of the power supply circuit shown in FIGS. This is because a loosely coupled state is set.

That is, when the degree of coupling is reduced to about the coupling coefficient k = 0.75 for the insulating converter transformer PIT, the power transmission rate from the primary side to the secondary side is also lowered accordingly.
As a result, even if the ripple current of the commercial AC power supply period is superimposed on the primary side resonance current Io on the primary side, the ripple component of the commercial AC power supply period is greatly attenuated when power is transmitted to the secondary side. become. As a result, the ripple voltage of the commercial AC power supply period superimposed on the secondary side DC output voltage Eo is also greatly suppressed.

However, simply by promoting the degree of loose coupling for the insulating converter transformer PIT, the transmission power between the primary side and the secondary side as the insulating converter transformer PIT is also attenuated. Transmission loss occurs, which causes a significant reduction in power conversion efficiency.
However, in the present embodiment, a parallel resonance circuit (N2 // C2) is provided also on the secondary side. As a result, on the secondary side, an alternating output is obtained so as to be amplified by the resonance operation of the secondary side parallel resonance circuit. As a result, transmission loss from the primary side to the secondary side is compensated, and as a result, the power conversion efficiency is lowered due to the insulating converter transformer PIT being further loosely coupled. There will be no.

In this way, as the power supply circuit of the present embodiment, first, by actively combining the secondary side parallel resonant circuit, the insulation converter transformer PIT can be made more efficient than before without causing a decrease in power conversion efficiency. Can also set the state of loose coupling. This suppresses the ripple of the commercial AC power supply cycle superimposed on the secondary side DC output voltage Eo, and it is possible to adopt a power factor improvement circuit based on the power regeneration system regardless of the tertiary winding N3 or magnetic coupling transformer. It is said.
The power factor correction circuit 10 according to the present embodiment using the power regeneration system is composed of only three components: a switching diode D1, a high frequency inductor L10, and a filter capacitor CN. The power factor shown in FIG. 7 and FIG. When compared with the improvement circuits 20 and 20A, the number of parts is reduced. Further, for example, the high-frequency inductor L10 only needs to have a structure capable of obtaining an inductance of about 30 μH, and the size of these three parts is also small.
As a result, the power supply circuit according to the present embodiment only requires a very small increase in the number of components, an increase in circuit scale, and an increase in cost when the power factor correction function is added.

Further, the secondary side rectifier circuit is a full-wave rectifier circuit provided with a bridge rectifier circuit DBR in the circuits shown in FIGS. 7 and 8, whereas in the present embodiment, the secondary side rectifier circuit is a half-wave. It is a rectifier circuit. This is because, in the power supply circuit of the present embodiment, the isolation converter transformer PIT is more loosely coupled than before, so that even a half-wave rectifier circuit can ensure ZCS regardless of load fluctuations. Because it became.
Thus, even if the secondary side rectifier circuit is a half-wave rectifier circuit, the number of parts and the cost can be reduced accordingly.

As for the power conversion efficiency, the power factor improving circuit 10 adopts a circuit form as a power regeneration system in which the primary winding N1 is directly connected as described above. As a result, the power loss in the system that feeds back the switching output as compared to the case of adopting the configuration in which the tertiary winding N3 and the loosely coupled magnetic coupling transformer MCT are interposed and fed back as a voltage shown in FIGS. Will be reduced.
In addition, as described above, in the power supply circuit shown in FIG. 1, the primary side and the secondary side are connected by the added polarity, but this also makes it more than the case where the connection is made by the depolarized connection. Power loss is reduced. When the depolarized connection is used, it is difficult to obtain a waveform having a flat peak as shown in FIG. 3, for example, as shown in FIG. 3 for the on-period waveform of the switching current IQ1 flowing through the switching element Q1, resulting in a switching loss. become. When the secondary-side rectifier circuit is a half-wave rectifier circuit, the polarity of the peak waveform of the switching current IQ1 is maintained by applying the polarity, and the power loss can be reduced accordingly.
Then, together with the effect of reducing the power loss obtained by these configurations and the compensation effect of the transmission power loss in the insulating converter transformer PIT by combining the secondary parallel resonant circuit described above, The power supply circuit according to the embodiment has good characteristics with respect to power conversion efficiency. As a result, as described below, the power supply circuit according to the present embodiment can achieve higher power conversion efficiency than the configuration without the power factor correction circuit 10 as an actual experimental result. It became a thing.

5 and 6 show various characteristics of the power supply circuit of the present embodiment shown in FIG. The characteristics shown in these figures are obtained by conducting experiments by selecting the main parts of the configuration of the power supply circuit shown in FIG. 4 as follows.
For the insulating converter transformer PIT, the core of EER-40 was selected, and the gap G was 1.6 mm. The number of turns of the primary winding N1 is 40T (turns), the number of turns of the secondary winding N2 is 50T, and k = 0.75 is obtained for the coupling coefficient k.
Further, the high frequency inductor L10 = 30 μH, the primary side voltage resonance capacitor Cr = 4700 pF, C2 = 0.01 μF, and the filter capacitor CN = 1 μF were selected.
Further, a fast recovery type switching diode D1 and a secondary side rectifier diode Do were selected with a rating of 3A / 600V, respectively, and a switching element Q1 with a rating of 10A / 800V was selected. .
In addition, after selecting each part in this way, the secondary side DC output voltage Eo has a specification of Eo = 200V.

FIG. 5 shows the fluctuation characteristics of the switching frequency fs, the rectified smoothing voltage Ei (DC input voltage), the power factor PF, and the AC → DC power conversion efficiency (ηAC → DC) with respect to the load power fluctuation for the power supply circuit shown in FIG. Is shown. Here, the experimental result in the fluctuation range of Po = 0 W to 200 W (maximum load power) is shown for the load power Po with the AC input voltage VAC fixed at 100V.
First, the switching frequency fs fluctuates so as to increase as the tendency of heavy load increases. This indicates that the switching frequency control operation for making the voltage constant is executed with respect to the fluctuation of the secondary side DC output voltage Eo according to the load fluctuation.
Further, the level of the DC input voltage (Ei) tends to increase slightly in the light load range, but as a whole, it can be seen that about 140 V is maintained with respect to the load fluctuation.
Further, the power factor PF obtained according to the operation of the power factor correction circuit 10 is such that PF> 0.75 or more is obtained in the range of the load power Po = 30 W to 200 W, and at the maximum load power Po = 200 W. PF = 0.85 is obtained, and for example, the power supply harmonic distortion regulation value is satisfied.
Further, the AC → DC power conversion efficiency (ηAC → DC) tends to increase as the load power Po becomes a heavy load tendency, and at the time of the maximum load power Po = 200 W, ηAC → DC = 92. % Measurement results were obtained.

FIG. 6 shows the fluctuation characteristics of the switching frequency fs, the rectified smoothing voltage Ei (DC input voltage), the power factor PF, and the AC → DC power conversion efficiency (ηAC → DC) with respect to the fluctuation of the AC input voltage VAC. The characteristics shown in this figure show the experimental results when the AC input voltage VAC = 85V to 140V fluctuates under the maximum load power Po = 200 W (maximum load condition).
The switching frequency fs has a characteristic of increasing as the AC input voltage VAC increases due to the constant voltage control operation. That is, as the switching frequency control, if the secondary side DC output voltage Eo fluctuates as the AC input voltage VAC increases or the load power Po becomes lighter, the switching frequency is controlled to be higher. Thus, stabilization is achieved by lowering the level of the secondary side DC output voltage Eo.
Further, the DC input voltage (Ei) tends to rise as it fluctuates so that the level of the AC input voltage VAC increases.

  Further, the power factor PF tends to increase as the AC input voltage VAC increases. And when maximum load electric power Po = 200W, it turns out that it is improved even to about PF = 0.80.

  The AC → DC power conversion efficiency (ηAC → DC) is maintained at 90% or more although the AC input voltage VAC tends to decrease slightly in the low level range, and the AC input voltage VAC is particularly 100V. In the range higher than the hit, 92% is maintained. This is consistent with the characteristics shown in FIG.

For example, in the configuration in which the power factor correction circuit 10 is omitted from the power supply circuit of the present embodiment shown in FIG. 1, the AC input voltage is the power factor PF and AC → DC power conversion efficiency (ηAC → DC), respectively. When VAC = 100 V and the maximum load power Po = 200 W, the experimental results of PF = 0.58 and ηAC → DC = 90.8% were obtained.
Therefore, the power supply circuit of the present embodiment adopting the circuit configuration as shown in FIG. 1 has an AC → DC power conversion efficiency (as compared with the case of adopting a circuit configuration in which the power factor correction circuit 10 is omitted). (ηAC → DC) is improved by 1.2%. Along with this, the AC input power was reduced by 2.8 W.

Although not shown in FIGS. 5 and 6, regarding the ripple voltage component ΔEo of the commercial AC power supply cycle superimposed on the secondary side DC output voltage Eo, the power supply circuit of FIG. 1 may have ΔEo = 80 mV. Measured. In contrast, a measurement result of ΔEo = 50 mV was obtained for the power supply circuit in which the power factor correction circuit 10 was omitted from the power supply circuit of FIG.
As described in the background art, for example, when a power factor correction circuit using a power regeneration system is provided for a power supply circuit having a coupling coefficient k of about 0.8 or more of the insulating converter transformer PIT, the ΔEo It increases to 5 to 6 times that of the configuration in which the rate improvement circuit is omitted. On the other hand, in this Embodiment, as above-mentioned, it is about 1.6 times (80/50 (mV)), and is suppressed significantly. The increase rate of ΔEo to this extent is within an allowable range that allows sufficient practical use without changing the design such as a significant increase in the capacity of the secondary-side smoothing capacitor. Therefore, the coupling coefficient k set in the insulating converter transformer PIT in the present embodiment should be set in consideration that ΔEo falls within the allowable range. Further, if the degree of coupling of the insulating converter transformer PIT is reduced, the transmission loss between the primary side and the secondary side also increases. Therefore, the coupling coefficient k is a compensation amount of the transmission loss by the secondary side parallel resonant circuit. It should be set in consideration of the trade-off.

The present invention is not limited to the configuration as the embodiment described so far. For example, details such as components such as a common mode noise filter connected to the commercial AC power supply AC, a connection mode, and the like may be appropriately changed.
Further, in the embodiment, a half-wave rectifier circuit is provided for the secondary side parallel resonant circuit. However, under the present invention, for example, in addition to a full-wave rectifier circuit and a double-wave rectifier circuit, a voltage doubler rectifier circuit is provided. A rectifier circuit that generates a secondary side DC output voltage Eo that is a predetermined multiple of the induced voltage level of the secondary winding N2 can also be configured. However, as described above, depending on the configuration of the present invention, even if a half-wave rectifier circuit is provided for the secondary-side rectifier circuit, ZCS against a wide range of load fluctuations is ensured. By using a half-wave rectifier circuit for the secondary rectifier circuit, the effect of reducing the number of parts can be obtained.

In each of the above embodiments, a MOS-FET is selected as a switching element for forming a primary voltage resonant converter. For example, an IGBT (insulated gate bipolar transistor), SIT (electrostatic induction thyristor) is selected. ), Semiconductor elements called various power devices such as bipolar transistors can be employed.
In addition, depending on the selection of these switching elements, any of a system for driving the switching elements, either a self-excited system using a driving IC, or a self-excited system having a self-excited oscillation drive circuit is used. May be adopted.
Further, as a voltage resonance type converter, a push-pull method in which two stone switching elements are alternately turned on / off is known, but the present invention also includes such a push-pull type voltage resonance type converter. Applicable. However, as shown in FIG. 1 as an embodiment, if the single-ended form is adopted, the switching element becomes one stone, which is advantageous in terms of reducing the number of parts, reducing the size and weight, and reducing the cost. It becomes.

It is a circuit diagram which shows the structural example of the power supply circuit as embodiment of this invention. It is a figure which shows the structural example of the insulation converter transformer with which the power supply circuit of embodiment is equipped. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the power supply circuit of embodiment by a switching period. It is a wave form diagram which shows operation | movement of the principal part corresponding to the power factor improvement operation | movement of the power supply circuit of embodiment by a commercial alternating current power supply period. It is a figure which shows the fluctuation characteristic of the switching frequency with respect to load fluctuation | variation, a rectification smoothing voltage (DC input voltage), a power factor, and AC-> DC power conversion efficiency in the power supply circuit of embodiment. It is a figure which shows the fluctuation characteristic of the switching frequency with respect to alternating current input voltage fluctuation | variation, a rectification smoothing voltage (DC input voltage), a power factor, and AC-> DC power conversion efficiency in the power supply circuit of embodiment. It is a circuit diagram which shows the structural example of the power supply circuit as a prior art. It is a circuit diagram which shows the other structural example of the power supply circuit as a prior art. It is a figure which shows the fluctuation characteristic of the rectification smoothing voltage (DC input voltage) with respect to the load fluctuation | variation of the power supply circuit shown in FIG. 8, a power factor, and AC-> DC power conversion efficiency.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 10 power factor correction circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, Cr primary side voltage resonance capacitor, N1 primary winding, N2 secondary winding Wire, C2 secondary side parallel resonant capacitor, Do rectifier diode, Co (secondary side) smoothing capacitor, CN filter capacitor, L10 high frequency inductor, D1 switching diode

Claims (2)

Rectifying and smoothing means for inputting a commercial AC power supply and generating a rectified and smoothed voltage;
Switching means formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
Winding only one primary winding to which a switching output obtained by the switching operation of the switching means is supplied, and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding; An insulating converter transformer formed and set to be equal to or less than a predetermined with respect to the coupling coefficient of the primary side and the secondary side;
A primary-side voltage resonance circuit formed by a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a primary-side voltage resonance capacitor, the operation of the switching means being a voltage resonance type;
A secondary side parallel resonant circuit formed by connecting a secondary side parallel resonant capacitor in parallel to the secondary winding of the insulating converter transformer;
A secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage;
The primary side resonance current obtained in the primary side voltage resonance circuit is regenerated as electric power by the switching operation of the switching means, and is fed back to the smoothing capacitor forming the rectifying and smoothing means. Power factor improving means configured to include a power factor improving switching element that switches the rectified current obtained by the rectifying operation by the rectifying and smoothing means intermittently according to electric power, and
The power factor improving means is
Comprising only one diode element, one filter capacitor and only one high-frequency inductor;
The diode element is inserted in series with a rectification output terminal of a rectification circuit for rectifying a commercial AC power supply in the rectification smoothing means, and the primary winding,
The filter capacitor is inserted in series with the rectifying output terminal and the smoothing capacitor in the rectifying and smoothing means,
The high-frequency inductor is inserted in series with respect to a connection point between the primary winding and the diode element, and a connection point between the filter capacitor and the smoothing capacitor.
Formed by connecting the primary side voltage resonance circuit to the connection point between the high-frequency inductor and the diode element.
Switching power supply circuit. The switching power supply circuit according to claim 1, comprising a half-wave rectifier circuit as a rectifier circuit forming the secondary side DC output voltage generating means.

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