JP2005168236A - Switching power supply - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a wide-range power supply circuit capable of attaining stabilization by means of a switching frequency control system using circuitry as simple as possible. <P>SOLUTION: In a switching power supply circuit comprising a resonance type converter, variations in secondary DC output voltage caused by level variation of a commercial AC power supply is stabilized, by performing variable control of switching frequency depending on level variation of the commercial AC power supply. The level of secondary DC output voltage is detected, and a control current Ic is fed to the control winding Nc of an insulation converter transformer PIT employing the structure as a saturable inductor depending on the level error thus performing variable control of the leakage inductance of the insulation converter transformer PIT. Constant voltage control of the secondary DC output voltage can be carried out properly, under such conditions as the load variational width is relatively large in the range from AC 100V system to AC 200V system. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit provided as a power source for various electronic devices.

As a switching power supply circuit, a circuit using a switching converter such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit to suppression of switching noise. Moreover, it has been found that there is a limit in improving the power conversion efficiency due to its operating characteristics.
Therefore, various types of switching power supply circuits using various resonant converters have been previously proposed by the present applicant. The resonant converter can easily obtain high power conversion efficiency, and low noise is realized by making the switching operation waveform sinusoidal. In addition, there is an advantage that it can be configured with a relatively small number of parts.

  As the switching power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC 85 V to 288 V is performed so as to correspond to an AC input voltage AC 100 V region such as Japan and the United States and an AC 200 V region such as Europe. A so-called wide-range power supply circuit that can be configured is known.

Here, as the above-described resonance type converter, one configured to be stabilized by controlling the switching frequency of a switching element forming the converter (switching frequency control method) is known.
As such a resonant converter based on the switching frequency control system, for example, in a configuration in which the switching element is switched and driven by a general-purpose oscillation / drive circuit IC, for example, the variable range of the switching frequency fs is maximum, fs = 50 KHz to 250 KHz. It is about. In the case of such a variable range, for example, under a load condition where the fluctuation range of the load power Po is a relatively large fluctuation range from Po = 0 W to about 90 W, and further to about 150 W, a wide range of AC85V to 288V is used. Stabilization corresponding to the AC input voltage range is almost impossible.

  In order to solve the above problem, the present applicant has previously proposed a switching power supply circuit having the following configuration. That is, as a switching power supply circuit for a wide range including a resonant converter, the switching power supply circuit can be stabilized against load fluctuations in a relatively wide range while adopting a switching frequency control method. A configuration example of such a switching power supply circuit is shown in FIG.

  In the power supply circuit shown in FIG. 9, first, a common mode choke coil CMC and a common mode noise filter composed of two sets of across capacitors CL and CL are connected to the commercial AC power supply AC as shown in the figure. Has been.

In this case, the rectifier circuit system includes a bridge rectifier circuit Di connected to a commercial AC power supply AC (AC input voltage VAC) and two smoothing capacitors Ci1 to Ci2 connected in series as shown in the figure. Connected. The rectified and smoothed voltage Ei is obtained as a voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2.
Further, a relay switch S is inserted between the negative output terminal of the bridge rectifier circuit Di and the connection point of the smoothing capacitors Ci1 to Ci2. This relay switch S is turned on / off according to the driving state of the relay RL connected to the rectifier circuit switching module 5.

The rectifier circuit switching module 5 is provided to switch the operation of the rectifier circuit system between when the input of the commercial AC power supply AC is a 100V system and when it is a 200V system by driving the relay RL.
Therefore, the rectifier circuit switching module 5 is supplied with a detection voltage output from a detection circuit including a diode D11, a capacitor C11, and voltage dividing resistors R11 and R12. In this detection circuit, first, a commercial AC power supply AC is rectified and smoothed by a half-wave rectifier circuit including a diode D11 and a capacitor C11, so that a DC voltage at a level corresponding to the commercial AC power supply AC is obtained as a voltage across the capacitor C11. Have been to get. A voltage level obtained by dividing this DC voltage by a series connection circuit of voltage dividing resistors R11-R12 connected in parallel to the capacitor C11 is input to the rectifier circuit switching module 5 as a detection voltage. Like to do. This detection voltage is obtained by dividing a DC voltage corresponding to the level of the commercial AC power supply AC at a predetermined ratio by a voltage dividing value corresponding to the resistance value of the voltage dividing resistors R11 and R12. Indicates the level of the commercial AC power supply AC.

  Further, a relay RL is connected to the rectifier circuit switching module 5. The rectifier circuit switching module 5 can switch conduction / non-conduction of the relay RL. Then, the relay RL performs on / off control of the relay switch S according to its conduction state. Here, it is assumed that the relay switch S is turned on when the relay RL is in a conductive state, and the relay switch S is turned off when the relay RL is in a non-conductive state.

The switching operation of the rectifier circuit configured as described above is as follows.
In the rectifier circuit switching module 5, the detection voltage input from the voltage dividing point of the voltage dividing resistors R11-R12 is input and compared as described above. For confirmation, this detection voltage indicates the level of the commercial AC power supply AC.
Here, for example, when the AC input voltage VAC is equal to or higher than a predetermined level near 150 V (AC 200 V system), the detected voltage is equal to or higher than the reference voltage, and when the AC input voltage VAC is equal to or lower than a predetermined level near 150 V (AC 100 V system). The voltage dividing ratio is set so as to be equal to or lower than the reference voltage.
In the rectifier circuit switching module 5, the relay RL is turned on when the divided voltage level is equal to or lower than the reference voltage, and the relay RL is turned off when it is equal to or higher than the reference voltage.

Here, for example, assuming that the commercial AC power supply is an AC 200V system, in this case, since the detected voltage level is equal to or higher than the reference voltage, the rectifier circuit switching module 5 turns off the relay RL. In response to this, the relay switch S is also turned off (opened).
When the relay switch S is OFF, the AC input voltage VAC is rectified by the bridge rectifier circuit Di and the rectified current is charged to the series connection circuit of the smoothing capacitors Ci1 to Ci2 in each period in which the AC input voltage VAC is positive / negative. Is obtained. That is, a rectification operation by a full-wave rectifier circuit including a normal bridge rectifier circuit can be obtained. As a result, a rectified and smoothed voltage Ei corresponding to an equal multiple of the AC input voltage VAC is obtained as a voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2.

On the other hand, it is assumed that the rectified and smoothed voltage Ei having a level corresponding to the AC input voltage VAC = 150 V or less is generated in correspondence with the commercial AC power supply of the AC100V system.
In this case, the voltage division level becomes equal to or lower than the reference voltage, and the rectifier circuit switching module 5 turns on the relay RL, so that the relay switch S is controlled to be turned on (closed).
When the relay switch S is on, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci1 when the AC input voltage VAC is positive. On the other hand, when the AC input voltage VAC is negative, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci2.
As a result of the rectifying operation performed in this way, a level corresponding to the same multiple of the AC input voltage VAC is generated as the voltage across each of the smoothing capacitors Ci1 and Ci2. Therefore, a level corresponding to twice the AC input voltage VAC is obtained as the rectified and smoothed voltage Ei that is the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. That is, a so-called voltage doubler rectifier circuit is formed.

  Thus, in the circuit shown in FIG. 9, in the case of the commercial AC power supply AC100V system, the rectified and smoothed voltage Ei corresponding to twice the AC input voltage VAC is generated by the voltage doubler rectification operation, and the commercial AC power supply AC200V In the case of the system, a rectified and smoothed voltage Ei corresponding to an equal magnification of the AC input voltage VAC is generated by a normal full-wave rectification operation. That is, as a result, the same level of rectified and smoothed voltage Ei is obtained when the commercial power input is an AC 100V system and an AC 200V system. This rectified and smoothed voltage Ei is input as a DC input voltage to the subsequent current resonance type converter.

  As shown in the figure, the current resonance type converter that switches by inputting the DC input voltage is connected by connecting two switching elements Q1 (high side) and Q2 (low side) by MOS-FET by half bridge coupling. Yes. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2, respectively, in the illustrated direction.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, in order to switch the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 using a general-purpose IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. Then, a drive signal (gate voltage) having a required frequency is applied to each gate of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

  Further, this oscillation / drive circuit 2 is a half-wave rectifier circuit comprising a rectifier diode D3 and a capacitor C3 with respect to a tertiary winding N3 formed by providing a tap output to the primary winding N1 of the insulating converter transformer PIT. The low-voltage DC voltage obtained by the above is input as an operating power source. Moreover, at the time of starting, it starts by inputting the rectification smoothing voltage Ei via the starting resistance Rs.

The insulating converter transformer PIT transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. The winding start end of the primary winding N1 of the insulation transformer PIT is connected to the connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the series connection of the primary side parallel resonant capacitor C1. By being connected, a switching output is transmitted.
Further, the winding end of the primary winding N1 is connected to the primary side ground.
Here, depending on the capacitance of the series resonance capacitor C1 and the leakage inductance L1 of the insulating converter transformer PIT including the primary winding N1, a primary side series resonance circuit for making the operation of the primary side switching converter a current resonance type is formed. To do. That is, the basic configuration of the switching converter in the power supply circuit shown in FIG. 9 is a current resonance type converter using a half-bridge coupling method.

According to the above description, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the part by the partial voltage resonance circuit (Cp // L1) described above. A voltage resonance operation is obtained.
That is, the power supply circuit shown in this figure adopts a form in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit. In this specification, such a switching converter is referred to as a composite resonance type converter.

Although illustration explanation here is omitted, the structure of the insulating converter transformer PIT includes, for example, an EE type core in which an E type core made of a ferrite material is combined. Then, after dividing the winding part on the primary side and the secondary side, the set of the primary winding N1 (tertiary winding N3) and the secondary windings N2 and N2A described below are made up of the EE type core. Winding around the central magnetic leg.
A gap of about 1.0 mm to 1.5 mm is formed with respect to the central magnetic leg of the EE type core. As a result, a loosely coupled state with a coupling coefficient of about 0.7 to 0.8 is obtained.

In this case, a secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT. An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding.
The secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then, as shown in the figure, a double wave rectification comprising rectifier diodes DO1 and DO2 and a smoothing capacitor CO. The circuit is connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown) and is also branched and inputted as a detection voltage for the control circuit 1 described below.

The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage EO to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. Thus, the level of the secondary side DC output voltage is stabilized by varying the switching frequency of the switching elements Q1, Q2.
Here, in the power supply circuit shown in FIG. 9, switching between a voltage doubler rectification operation and a full-wave rectification operation is performed when the commercial power input is an AC100V system and an AC200V system, respectively. As a result, a DC input voltage (Ei) of the same level can be obtained regardless of whether the commercial power input is an AC 100V system or an AC 200V system, and the level of the secondary side DC output voltage Eo can be set accordingly. Similarly, the AC100V system and the AC200V system are equivalent.
For this reason, for example, even under the condition of a relatively large load fluctuation range where the load power Po = 0 W to 150 W, the AC 100 V system and the AC 200 V are controlled by the switching frequency control within the maximum variable range of 50 KHz to 250 KHz described above. In both cases, the secondary side DC output voltage can be effectively stabilized. In this way, the power supply circuit shown in FIG. 9 is compatible with a wide range.

A configuration in which switching is performed between the voltage doubler rectifier circuit and the full-wave rectifier circuit in accordance with the levels of the AC 100 V system and AC 200 V system commercial AC power as described above is described in, for example, the above-mentioned document.
JP-A-7-281770

As can be understood from the above description, the power supply circuit shown in FIG. 9 is configured to switch between full-wave rectification operation and voltage doubler rectification operation by an electromagnetic relay. For this configuration, two smoothing capacitors Ci1 and Ci2 are required as smoothing capacitors for obtaining the rectified and smoothed voltage Ei (DC input voltage).
That is, at least the number of smoothing capacitors is increased and a required number of electromagnetic relays are added for switching the rectifying operation. For this reason, the number of parts is increased and the cost is increased, and the mounting area of the power supply circuit board is increased and the size is increased. In particular, since these smoothing capacitors and electromagnetic relays are large among the components forming the power supply circuit, the substrate size becomes considerably large.

Further, as a circuit for switching between the full-wave rectification operation and the voltage doubler rectification operation, for example, in the basic configuration simply shown in FIG. 9, the following malfunction may occur. In other words, when AC200V commercial AC power is being input, if an instantaneous power failure occurs or the AC input voltage drops below the rated value, the AC100V level becomes lower than that corresponding to the AC200 system. An operation of switching to a voltage doubler rectifier circuit occurs as a system.
When such a malfunction occurs, voltage doubler rectification is performed on an AC input voltage of AC200V system level, so that there is a possibility that the switching elements Q1, Q2, etc. may be destroyed due to overvoltage.

Therefore, as an actual circuit, in order to prevent the above-described malfunction, not only the DC input voltage of the main switching converter but also the DC input voltage of the converter circuit on the standby power supply side is detected. It is made to take. For example, in the case of FIG. 9, the power supply circuit shown in this figure is the main switching converter. Actually, the rectifier circuit switching module 5 also receives a DC input voltage obtained by a converter of a standby power supply not shown in FIG. 9 as a detection voltage.
Further, as the standby power supply side converter circuit is detected as described above, a comparator IC is mounted as the rectifier circuit switching module 5. As a result, the number of parts increases and the above-described cost increase and circuit board size increase are further promoted.
When the rectifier circuit switching module 5 adopts such a configuration, for example, in order to improve parts management and work efficiency during manufacturing, the rectifier circuit switching module 5 is unitized by assembling a circuit system for switching the rectifying operation as a module. The However, in the case of unitization in this way, more parts such as adding pin terminals are required, which increases the cost.

  In addition, the detection of the DC input voltage of the converter on the standby power supply side for the purpose of preventing malfunctions means that a wide range compatible power supply circuit having a circuit for switching the rectifying operation includes a standby power supply in addition to the main power supply. This means that it cannot actually be used unless it is an electronic device. In other words, the types of electronic devices that can be equipped with a power supply are limited to those equipped with a standby power supply, and there is a problem that the use range is narrowed accordingly.

In view of the above-described problems, the present invention is configured as a switching power supply circuit as follows.
That is, a rectified and smoothed voltage generating means for generating a rectified and smoothed voltage at a level corresponding to the same size as the input commercial AC power supply, and a switching operation by inputting the rectified and smoothed voltage as a DC input voltage. It comprises switching means formed by half-bridge coupling of switching elements, and switching drive means for driving the switching elements.
Further, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied, a secondary winding in which an alternating voltage as a switching output obtained in the primary winding is excited, and a control winding And an insulating converter transformer formed so that its own leakage inductance is varied according to the level of the control current flowing through the control winding.
A primary-side resonance capacitor provided so that a primary-side resonance circuit having a resonance type operation of the switching means is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and its own capacitance; DC output voltage generating means configured to generate a required number of secondary side DC output voltages of one or more by inputting an alternating voltage obtained in the secondary winding of the insulating converter transformer and performing rectification operation Is provided.
Then, the switching drive means is controlled so that the switching frequency of the switching means can be varied according to the level of the commercial AC power supply, or the control current level to be passed through the control winding is variably controlled. 1st constant voltage control means comprised so that the operation | movement which makes a side DC output voltage constant may be provided.
Further, according to the level of one specific secondary side DC output voltage, the control current level to be passed through the control winding is variably controlled, or the switching drive means so that the switching frequency of the switching means is variable. It is assumed that a second constant voltage control unit configured to obtain a constant voltage operation for the specific secondary side DC output voltage is obtained by controlling.

In the power supply circuit having the above-described configuration, the switching frequency control and the leakage inductance control in the insulating converter transformer are combined in combination to stabilize the secondary side DC output voltage. In either one of these two controls, the secondary side DC output voltage is made constant according to the level of the commercial AC power supply, and in the other control, the level of the secondary side DC output voltage is set. In response to this, the secondary side DC output voltage is made constant. That is, the one control performs constant voltage control with respect to the fluctuation of the secondary side DC output voltage caused by the level change of the commercial AC power supply, and the other control is connected to the secondary side DC output voltage. Constant voltage control is performed with respect to fluctuations in the secondary side DC output voltage caused by load fluctuations.
For example, assuming that the maximum variable range of the switching frequency and the maximum variable range of the leakage inductance are equivalent, simply switching frequency control or leakage inductance control of the isolated converter transformer according to the level of the secondary side DC output voltage alone A comparison is made between a conventional configuration that achieves stabilization by performing the above and a configuration that simultaneously operates two constant voltage controls as in the present invention. Then, it can be said that the present invention can perform constant voltage control corresponding to a wider input level range of a commercial AC power supply and a wider load fluctuation range.

Therefore, according to the present invention, for example, the level range of a commercial AC power source that is actually set to AC100V system to AC200V system is input and the load fluctuation range is relatively large. A switching power supply circuit capable of stabilization can be obtained. In this case, it is not necessary to switch the DC input voltage level of the switching converter. In other words, the rectifier circuit that generates the DC input voltage (rectified and smoothed voltage) is not configured to switch between the voltage doubler rectification operation and the full-wave rectification operation according to the level of the commercial AC power supply. Therefore, a normal full-wave rectifier circuit that generates a rectified and smoothed voltage corresponding to the same magnification as that of the normal full-wave rectifier circuit may be used. For this reason, the number of smoothing capacitors forming the rectifier circuit for generating the DC input voltage is reduced from two to one, for example.
In addition, the minimum number of switching elements may be two, for example, for forming a half-bridge connection. In this respect as well, the number of parts can be reduced. By reducing the number of components in this way, the circuit board can be reduced in size and weight and cost can be reduced. Further, since the number of switching elements is the minimum necessary, the switching noise can be suppressed to the minimum level.

FIG. 1 shows a configuration example of a power supply circuit based on the power supply circuit previously proposed by the present applicant. The power supply circuit shown in this figure has a configuration that is compatible with a wide range and includes a current resonance type converter that employs a switching frequency control system.
The power supply circuit shown in FIG. 1 is provided with a full-bridge coupled current resonance type converter having four switching elements on the primary side. As described later, the switching operation is switched between the full-bridge coupling method and the half-bridge coupling method when the commercial AC power supply is rated for the AC100V system and the AC200V system. It is compatible with a wide range.
Further, a separate excitation type is adopted as the switching drive method. A partial voltage resonance circuit that performs voltage resonance only when the switching element of the current resonance type converter is turned on / off is combined.

In the circuit shown in FIG. 1, a commercial AC power supply AC is first connected to a common mode noise filter consisting of a pair of common mode choke coils CMC and two sets of across capacitors CL and CL as shown. ing.
The commercial AC power supply AC is provided with a full-wave rectifier circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci. In this case, a rectified and smoothed voltage Ei having a level corresponding to only the equal magnification of the AC input voltage VAC is obtained at both ends of the smoothing capacitor Ci. This rectified and smoothed voltage Ei is input as a DC input voltage to the subsequent current resonance converter.
Thus, it can be seen that the power supply circuit shown in FIG. 1 is not provided with a circuit system for switching the rectifying operation, for example, as shown in FIG.

  The current resonance type converter shown in this figure includes four switching elements Q1, Q2, Q3, and Q4 corresponding to the full-bridge coupling method. In this case, corresponding to the separately excited type, a MOS-FET which is a voltage drive type is selected for these switching elements Q1 to Q4.

The drain of the switching element Q1 is connected to the line of the rectified and smoothed voltage Ei (DC input voltage). The source of switching element Q1 is connected to the drain of switching element Q2. The source of the switching element Q2 is connected to the primary side ground.
That is, the switching Q1 and Q2 are connected in series so as to be half-bridged so that the switching element Q1 is on the high side and the switching element Q2 is on the low side, thereby forming a pair of half-bridge circuits. doing.

Similarly, the drain of the switching element Q3 is connected to the line of the rectified and smoothed voltage Ei (DC input voltage), and the source is connected to the drain of the switching element Q4. The source of the switching element Q4 is connected to the primary side ground.
That is, the switching elements Q3 and Q4 are connected by half-bridge coupling so that the switching element Q3 is on the high side and the switching element Q4 is on the low side, thereby forming another set of half-bridge circuits.
According to such a connection mode, two sets of half-bridge circuits including a set of switching elements [Q1, Q2] and a set of switching elements [Q3, Q4] are connected to the DC input voltage (Ei) line and the primary. It is inserted in parallel between the side grounds. As a result, a switching circuit system as a full bridge coupling method is formed.

A clamp diode DD1 is connected in parallel between the drain and source of the switching element Q1. The anode and cathode of the clamp diode DD1 are connected to the source and drain of the switching element Q1, respectively. The clamp diode DD1 forms a pair of switching circuits together with the switching element Q1, and forms a path through which a reverse current flows when the switching element Q1 is turned on.
In the same connection manner, clamp diodes DD2, DD3, and DD4 are connected in parallel to switching elements Q2, Q3, and Q4, respectively.

In addition, partial resonance capacitors Cp1 and Cp2 are connected in parallel between the drain and source of the low-side switching elements Q2 and Q4 in each half bridge circuit.
A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitances of the partial resonance capacitors Cp1 and Cp2 and the leakage inductance component L1 of the primary winding N1 of the insulating converter transformer PIT described later.
By forming the partial voltage resonance circuit in this way, a partial voltage resonance operation in which voltage resonance occurs only in a short period of time when the switching elements Q1 to Q4 are turned on / off is obtained.
The configuration of the switching drive circuit system for these switching elements Q1 to Q4 will be described later.

The insulating converter transformer PIT transmits the switching outputs of the switching elements Q1 to Q4 to the secondary side.
Although the illustration of the structure of the insulating converter transformer PIT is omitted here, the primary winding N1 and the secondary winding N2 are formed corresponding to the primary side and the secondary side, for example, for the EE type core. Each of the divided areas is wound around. Further, in the insulating converter transformer PIT in this case, as shown in the figure, the tertiary winding N3 is also wound on the primary side.

  One end of the primary winding N1 of the insulating converter transformer PIT is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the series resonant capacitor C1. The other end of the primary winding N1 is connected to a connection point (switching output point) between the source of the switching element Q3 and the drain of the switching element Q4.

A primary side series resonance circuit is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component (L1) of the insulating converter transformer PIT including the inductance component L1 of the primary winding N1.
In the full-bridge coupling method, as described later, a switching operation is performed at the timing when the pair of switching elements [Q1, Q4] and the pair of switching elements [Q2, Q3] are alternately turned on / off. Since the primary side series resonant circuit comprising the primary winding N1 and the series resonant capacitor C1 is connected to the switching output point, the switching outputs of the switching elements Q1 to Q4 are transmitted to the primary side series resonant circuit. Will be. Then, the primary side series resonance circuit performs a resonance operation in accordance with the switching output, whereby an operation as a current resonance type is obtained. In the primary winding N1, a primary winding current close to the resonance waveform is obtained in accordance with the operation as the current resonance type.

  In this way, the switching converter of the power supply circuit shown in FIG. 1 has a combined operation of the current resonance type corresponding to the full bridge coupling method and the partial voltage resonance operation described above. . That is, a configuration as a composite resonance type converter is adopted.

An alternating voltage excited in accordance with the switching output transmitted to the primary winding N1 is generated in the secondary winding N2 of the insulating converter transformer PIT.
In this case, a center tap is provided for the secondary winding N2. This center tap is connected to the secondary side ground. In addition, as shown in the figure, a two-wave rectifier circuit is formed by connecting two rectifier diodes D01 and D02 and a smoothing capacitor Co to the secondary winding N2. The double-wave rectifier circuit receives the alternating voltage excited by the secondary winding N2 and performs a rectification operation, whereby a secondary side DC output voltage EO is obtained as a voltage across the smoothing capacitor CO.
The secondary side DC output voltage EO is supplied to a load (not shown). Further, this secondary side DC output voltage EO is also branched and inputted as a detection voltage for the control circuit 1 as shown.

  A half-wave rectifier circuit composed of a diode D4 and a capacitor C4 is connected to the tertiary winding N3 wound on the primary side of the insulating converter transformer PIT as shown in the figure. The low-voltage DC voltage obtained by this half-wave rectifier circuit is supplied as an operating power source to each of an oscillation / drive circuit 2 and a drive circuit 3 to be described later.

The control circuit 1 obtains, as a control output, a current or voltage whose level is varied according to the level of the secondary side DC output voltage EO, for example. This control output is output to the oscillation / drive circuit 2.
The oscillation / drive circuit 2 generates an oscillation signal as will be described later, and uses this oscillation signal to output a high-side drive signal and a low-side drive signal for driving the switching element by separate excitation. Then, with this drive signal, the switching elements Q1 to Q4 are switched and driven at a required switching timing.
The oscillation / drive circuit 2 operates so as to vary the frequency of the oscillation signal generated internally in accordance with the control output level input from the control circuit 1. As a result, the frequency of the drive signal is varied according to the control output level. That is, the oscillation / drive circuit 2 operates to variably control the switching frequency of the switching elements Q1 to Q4 according to the control output level input to the control terminal Vc.
By changing the switching frequency, the resonance impedance in the series resonance circuit changes. As the resonance impedance changes in this way, the amount of current supplied to the primary winding N1 of the primary side series resonance circuit changes, and the power transmitted to the secondary side also changes. As a result, the secondary output voltage changes, and constant voltage control is achieved.

  Next, a switching drive circuit system for driving the switching elements Q1 to Q4 in the power supply circuit shown in FIG. 1 will be described. The power supply circuit shown in FIG. 1 includes a single oscillation / drive circuit 2 and a drive circuit 3 as a switching drive circuit system.

The oscillation / drive circuit 2 includes an oscillation circuit, a control circuit, a protection circuit, and the like for driving a current resonance type converter by a separate excitation type, and an analog IC (Integrated Circuit) having a bipolar transistor therein. ).
As described above, the oscillation / drive circuit 2 operates by inputting a DC voltage obtained by a half-wave rectifier circuit including the tertiary winding N3, the diode D3, and the capacitor C3. Further, at the time of power activation, the rectified and smoothed voltage Ei input via the activation resistor Rs can be input as the activation power source so that the operation can be started.

The oscillation / drive circuit 2 outputs two drive signals as terminals for outputting a drive signal (gate voltage) to the switching element. That is, the drive signal SD1 for switching the high-side switching element and the drive signal SD2 for switching the low-side switching element are output.
In this case, the drive signal SD1 is applied to the gate of the high-side switching element Q1. The drive signal SD2 is applied to the gate of the low-side switching element Q2. The drive signal SD1 is output after being level-shifted with respect to the drive signal SD2 so that the switching element Q1 on the high side can be driven appropriately. The drive signals SD1 and SD2 branch and are also input to the drive circuit 3.

In the drive circuit 3, the input drive signal SD1 is shifted to a voltage level for driving the low-side switching element and applied to the gate of the switching element Q4 as the drive signal SD4. Further, the drive signal SD2 is shifted to a voltage level for driving the high-side switching element, and applied to the gate of the switching element Q4 through the switch Sw as the drive signal SD3.
The drive circuit 3 is supplied with a detection voltage obtained by dividing the rectified and smoothed voltage Ei (DC input voltage) by a voltage dividing resistor R1-R2 at a predetermined voltage dividing ratio. In the drive circuit 3, as will be described later, the switch Sw is turned on / off in accordance with the level of the detected voltage. The switch Sw may be configured to use, for example, a low-power bipolar transistor as a switch element.

  In the power supply circuit shown in FIG. 1, switching is performed between a switching operation by a full bridge coupling method and a switching operation by a half bridge coupling method in accordance with the level of the AC input voltage VAC (commercial AC power supply AC) as described later. Done. However, here, as a basic operation, an operation when all of the switching elements Q1 to Q4 are driven to be switched will be described corresponding to the case of the full bridge coupling method.

As described above, the oscillation / drive circuit 2 outputs the high-side drive signal SD1 for driving the high-side switching element and the low-side drive signal SD2 for driving the low-side switching element. .
Here, the drive signal SD1 and the drive signal SD2 have a pulse waveform that rises at a predetermined level in response to turning on the switching element and falls at a predetermined level in response to turning off. These pulse waveforms are generated so as to have a phase difference of 180 ° from each other.
As a result, first, the switching elements Q1 and Q2 having a high-side and low-side relationship are driven to be switched on and off alternately.

  On the other hand, in the drive circuit 3, the input drive signal SD1 is level-shifted and applied to the low-side switching element Q4 as the drive signal SD4. Similarly, the input drive signal SD2 is level-shifted and then applied to the high-side switching element Q3 as the drive signal SD3. That is, as the waveform timing, the set of drive signals SD1 and SD4 is the same, and the set of drive signals SD2 and SD3 is the same. In addition, the drive signals SD1 and SD4 and the drive signals SD2 and SD3 are set to have a phase difference of 180 ° from each other.

  Therefore, the switching elements Q1 and Q4 are turned on / off at the same timing, and similarly, the switching elements Q2 and Q3 are also turned on / off at the same timing. In addition, the group of switching elements [Q1, Q4] and the group of switching elements [Q2, Q3] are driven to be switched on and off alternately.

As a switching operation at this time, when the set of the switching elements [Q1, Q4] is on, the output is the drain-source of the switching element Q1 → the series resonant capacitor C1 → the primary winding N1 → the drain of the switching element Q4. -Current flows through the source → primary side ground.
When the pair of switching elements [Q2, Q3] is on, the output is the drain-source of the switching element Q3 → the primary winding N1 → the series resonant capacitor C1 → the drain-source of the switching element Q2 → the primary side. Current flows through the ground path. Then, as this operation is repeated, the primary side series resonance circuit (C1-N1) obtains a resonance operation, and a switching output current close to the resonance current waveform is applied to the primary winding N1 of the insulating converter transformer. Will be supplied.
An operation in which the switching elements Q1, Q2, Q3, and Q4 perform switching in this way is a switching operation as a full bridge coupling method.

Further, at the timing when the set of the switching elements [Q1, Q4] is turned off / turned on as described above, the parallel resonant capacitor Cp2 connected to the switching element Q4 has its own capacitance and the leakage inductance of the primary winding N1. A parallel resonance circuit is formed by the (leakage inductance) component L1, and voltage resonance operation is performed. That is, a partial voltage resonance operation in which voltage resonance occurs only when the pair of switching elements [Q1, Q4] is turned off / turned on can be obtained.
Similarly, at the timing when the pair of switching elements [Q2, Q3] is turned off / turned on, the parallel resonant circuit is generated by the capacitance of the parallel resonant capacitor Cp1 connected to the switching element Q2 and the leakage inductance component L1 of the primary winding N1. Is formed. A partial voltage resonance operation can be obtained at the time of turn-off / turn-on of the set of the switching elements [Q2, Q3].

  In this way, in the circuit of FIG. 1, a full-bridge coupled current resonance type converter in which each set (switching circuit) of switching elements [Q1, Q4] [Q2, Q3] is alternately turned on / off, and a partial voltage A converter in which resonance circuits (Cp1, Cp2, N1) are combined is formed.

  The power supply circuit shown in FIG. 1 performs the switching operation (full bridge operation) by the full bridge coupling method in the AC 100V system as described below under the above-described configuration of the separately excited full bridge coupling method, and the AC 200V system. Then, the structure which switches switching operation | movement is taken so that it may become switching operation (half-bridge operation | movement) by a half-bridge coupling system.

  In the power supply circuit shown in FIG. 1, a voltage dividing circuit in which voltage dividing resistors R1-R2 are connected in series is connected in parallel to the smoothing capacitor Ci. The voltage obtained at the connection point (voltage dividing point) of the voltage dividing resistors R1-R2 is input to the drive circuit 3 as the detection voltage as described above. As can be seen from the fact that the detected voltage is connected in parallel to the smoothing capacitor Ci, the detected voltage is obtained by dividing the rectified and smoothed voltage Ei (DC input voltage Ei). Here, since the level of the rectified and smoothed voltage Ei corresponds to the level of the commercial AC power supply AC, the detected voltage also corresponds to the level of the commercial AC power supply AC.

In the drive circuit 3, for example, when it is detected that the AC input voltage VAC is at a level corresponding to the AC 100V system corresponding to the vicinity of 150V or less, the detected voltage is between the drive signal SD3 and the gate of the switching element Q3. The inserted switch Sw is controlled to be turned on. At this time, a state in which the drive signal SD3 is applied to the gate of the switching element Q3 is obtained.
As a result, for example, as described above with reference to FIG. 5, a state is obtained in which the four switching elements Q1 to Q4 are driven to be switched. That is, a full bridge operation is obtained.

On the other hand, when the drive circuit 3 detects that the AC input voltage VAC is at a level corresponding to the AC 200 V system corresponding to the vicinity of 150 V or higher, the drive circuit 3 turns off the switch Sw. To control.
As a result, the drive signal SD3 is not applied to the gate of the switching element Q3, so that the switching element Q3 is brought into a state where the switching operation is stopped, and the remaining switching elements Q1, Q2, Q4 are in a state where the switching operation is performed.
At this time, the set of the switching elements [Q1, Q4] is turned on / off at the same timing, whereas only the switching element Q2 is switched at the timing that is alternate with respect to the set of the switching elements [Q1, Q4]. It will be turned on / off.

  In this way, the fact that the three switching elements Q1, Q2, Q4 perform the switching operation means that the switching elements [Q2, Q4] are turned on / off at the timing when the pairs of the switching elements [Q2, Q4] alternate. It can be said that the half-bridge operation is obtained.

As will be understood from the above description, in the power supply circuit shown in FIG. 1, first, a full-bridge coupling type current resonance type converter including four switching elements Q1 to Q4 is formed. In addition, the switching operation is switched so that a full-bridge operation is performed in the AC100V system and a half-bridge operation is performed in the AC200V system.
In this way, the current levels obtained by the switching outputs of the switching elements Q1 to Q4 are almost equal between the AC100V system and the AC200V system. Thereby, for example, even in the maximum variable range of the switching frequency of about 50 KHz to 250 KHz, it is possible to effectively stabilize the secondary side DC output voltage in both the AC100V system and the AC200V system. . In other words, wide range is supported by switching between full bridge operation and half bridge operation.

Then, by achieving compatibility with a wide range in this way, a normal full-wave rectifier circuit can be used as a rectifier circuit system that generates a DC input voltage (rectified and smoothed voltage Ei) from a commercial AC power supply AC. That is, it is not necessary to adopt a configuration for switching the rectification operation as in the circuit shown in FIG.
Therefore, in the power supply circuit shown in FIG. 1, only one smoothing capacitor for DC input voltage is required. Also, no electromagnetic relay is required.

Further, even if the commercial AC power supply of nominal AC 220V or 240V drops below around 150V due to, for example, an instantaneous power failure, the switching operation only changes from the half-bridge operation to the full-bridge operation. That is, since the DC input voltage level does not increase due to switching to the voltage doubler rectification operation as in the circuit shown in FIG. 9, the smoothing capacitor Ci and the switching element do not exceed the withstand voltage.
Further, by omitting the circuit system for switching the rectifying operation provided with the comparator IC and the like, it is not necessary to detect the DC input voltage on the standby power supply side in order to cope with the wide range. Therefore, the power supply circuit shown in FIG. 1 can be used for an electronic device that does not include a standby power supply.

Next, FIG. 2 shows a configuration of a switching power supply circuit as an embodiment of the present invention.
In the power supply circuit shown in this figure, first, as in the case of FIG. 1, a common mode noise filter formed by a common mode choke coil CMC and filter capacitors CL and CL is connected to a commercial AC power supply AC. The

  Further, as a rectifier circuit for rectifying and smoothing the commercial AC power supply AC to generate a DC input voltage (rectified and smoothed voltage Ei), as in the circuit of FIG. 1, all of the bridge rectifier circuit Di and the smoothing capacitor Ci are used. A wave rectifier circuit is provided, and a DC input voltage (rectified and smoothed voltage Ei) having a level corresponding to the same value as the effective value level of the commercial AC power supply AC is obtained as the voltage across the smoothing capacitor Ci.

  As a switching converter that performs switching by inputting the DC input voltage, a current resonance type converter using a half-bridge coupling system including two switching elements Q1 and Q2 is provided.

  As the switching elements Q1 and Q2, the drain of the switching element Q1 is connected to the positive terminal side of the smoothing capacitor Ci, and the source of the switching element Q1 is connected to the drain of the switching element Q2. The source of the switching element Q2 is connected to the negative terminal (primary side ground) of the smoothing capacitor Ci. That is, as half-bridge coupling, the switching element Q1 is connected in series so that the switching element Q1 is on the high side and the switching element Q2 is on the low side, and then connected in parallel to the smoothing capacitor Ci.

Damper diodes DD1 and DD2 are connected in parallel to each other between the drains and sources of the switching elements Q1 and Q2, respectively, so that the drain side is a cathode and the anode side is a drain.
Also in this circuit, a partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  Also in this case, in order to perform switching driving of the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 configured to include a general-purpose driving IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit for driving a switching element. Depending on the oscillation circuit and the drive circuit, a drive signal (gate voltage) having a required frequency is applied to each gate of the switching elements Q1 and Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

Further, in this case, the voltage obtained at the connection point of the voltage dividing resistors R21 to R22 connected in parallel to the smoothing capacitor Ci is input to the oscillation / drive circuit 2 as a control voltage. The oscillation / drive circuit 2 performs switching driving of the switching elements Q1, Q2 by variably controlling the switching frequency according to the level of the control voltage.
Here, the control voltage is a DC voltage obtained by dividing the rectified and smoothed voltage Ei (DC input voltage), but the rectified and smoothed voltage Ei is generated by rectifying and smoothing the commercial AC power supply AC. The level of the rectified and smoothed voltage Ei corresponds to the level of the commercial AC power supply AC (AC input voltage VAC). Therefore, the oscillation / drive circuit 2 in the present embodiment is configured to variably control the switching frequency according to the level of the commercial AC power supply AC (AC input voltage VAC).
It should be noted here that the commercial AC power supply AC is normally switched discontinuously between predetermined rated levels of AC 100 V system and AC 200 V system. That is, the variable control operation of the switching frequency according to the level of the commercial AC power supply AC (AC input voltage VAC) is continuous. In other words, the commercial AC power supply AC (AC input voltage VAC) is continuously changed between a predetermined level for the AC100V system and a predetermined level for the AC200V system (example: VAC = 85V to 288V). For example, in response to this, the oscillation / drive circuit 2 performs switching driving so as to continuously change the switching frequency.

Although not shown here, with respect to power supply driving of the oscillation / drive circuit 2, a tertiary winding wound around the primary side of the insulating converter transformer PIT, and this tertiary winding, as in FIG. A rectifier circuit that rectifies and smoothes the alternating voltage excited by the rectifier circuit may be provided, and a low-voltage DC voltage obtained by the rectifier circuit may be supplied as an operating power source.
In addition, the configuration for starting the oscillation / drive circuit 2 may be a configuration in which the rectified and smoothed voltage Ei obtained through the start resistor Rs is input as the start power supply in the same manner as in FIG.

The insulating converter transformer PIT according to the present embodiment transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side and varies the leakage inductance for constant voltage control.
In this case, the primary winding N1 is wound on the primary side of the insulating converter transformer PIT, and the secondary winding N2 is wound on the secondary side. Further, the control winding Nc is wound.

Here, an example of the structure of the insulating converter transformer PIT is shown in FIGS.
First, the insulating converter transformer PIT shown in FIG. 3 includes two double U-shaped cores CR1 and CR2 each having four magnetic legs. A solid core is formed by joining the ends of the magnetic legs of the double U-shaped cores CR1 and CR2.
When the three-dimensional core is formed in this way, there are four joint portions of the double U-shaped cores CR1 and CR2 corresponding to each of the four magnetic legs. In this case, gaps G and G having a predetermined length are formed for two adjacent joints in these four joints, and no gap is formed for the remaining two joints.
In the three-dimensional core thus formed, for example, a magnetic leg in which a gap G on the side of the double U-shaped core CR1 is formed and a magnetic leg in which a gap adjacent to the magnetic leg is not formed. The control winding Nc is wound with a predetermined number of turns (number of turns) so as to straddle.
In addition, the other double U-shaped core CR2 side has two winding straddling two adjacent magnetic legs so as to be in a winding direction orthogonal to the winding direction of the control winding NC. The secondary winding N2 and the primary winding N1 are wound by a predetermined number of turns. In this case, the secondary winding N2 is located closer to the magnetic leg joint than the primary winding N1. The two magnetic legs on which the primary winding N1 and the secondary winding N2 are wound in this way also have a relationship in which one has a gap G and the other has no gap. Become.
With such a structure, it is configured as a saturable reactor that is in a magnetic saturation state due to an increase in the direct current (control current Ic) flowing through the control winding Nc. By adopting such a configuration as a saturable reactor, the leakage inductance of the primary winding N1 (and the secondary winding N2) changes according to the level of the control current Ic flowing through the control winding Nc. That is, the insulating converter transformer PIT shown in this figure includes a control winding Nc, a primary winding N1 as a controlled winding whose secondary winding direction is orthogonal to the control winding Nc, and a secondary winding. It is configured as an orthogonal control transformer around which the line N2 is wound.

Further, as another structure of the insulating converter transformer PIT, as shown in FIG. 4, a three-dimensional core has a double U-shaped core CR1 having one magnetic core and four cores. Can be formed by combining as a single U-shaped core CR3 having a U-shaped arbitrary cross section instead of the double U-shaped core CR2.
Also in this case, the two gaps G and G are formed by the same positional relationship as that of the insulating converter transformer PIT shown in FIG. The control winding Nc is wound around the two magnetic legs of the double U-shaped core CR1 in the same manner as in FIG. 3, and the single U-shaped core CR3 is wound as shown in the figure. Thus, the primary winding N1 and the secondary winding N2 are wound. Also in this case, the control winding Nc and the set of the primary winding N1 and the secondary winding N2 are similar to the case of FIG. 3 in that the winding directions are orthogonal to each other.

In addition, the insulating converter transformer PIT may have a structure using a core having the shape shown in FIGS. 5 and 6 in addition to the above-described double U-shaped core or single U-shaped core.
First, in the insulating converter transformer PIT shown in FIG. 5, two half-eye-shaped cores CR11 and CR12 are prepared, and one core-shaped core is formed by combining these cores so that their magnetic legs face each other. To do. In addition, in the square core, a total of four magnetic legs, two outside and two inside, face each other. Of these, a gap of a predetermined length is formed on the face where one outer magnetic leg faces. G is formed.
The control winding Nc is formed with a predetermined number of turns (number of turns) so as to straddle one outer magnetic leg in one half-eye-shaped core CR11 and one inner magnetic leg adjacent to the outer magnetic leg. Wrap it. Further, the primary winding N1 and the secondary winding N2 are wound by a predetermined number of turns (number of turns) so as to straddle the two inner magnetic legs in the other half-eye-shaped core CR12. Also in this case, the winding position is such that the secondary winding N2 is closer to the magnetic leg joint than the primary winding N1.

  Further, in the insulating converter transformer PIT shown in FIG. 6, the six magnetic leg E-shaped core CR21 and the normal three magnetic leg E-shaped core CR22 are combined and illustrated such that their magnetic legs face each other. Forming the core of the structure. The control winding Nc is wound around the two outer magnetic legs on the lower side in the figure in the six magnetic leg E-shaped core CR21. On the other hand, the primary winding N1 and the secondary winding N2 are wound around the central magnetic leg of the E-shaped core CR22. In this case, the winding directions of the primary winding N1 and the secondary winding N2 are orthogonal to the winding direction of the control winding Nc. In this case, the relationship between the core and the winding is a so-called outer iron shape.

Returning to FIG.
The primary winding N1 of the insulating transformer PIT in this case also has one end connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the primary side series resonant capacitor C1. And the other end is connected to the primary side ground.

As understood from the above description, the series connection circuit of the series resonance capacitor C1 and the primary winding N1 connected to the switching output points of the switching elements Q1 and Q2 is the capacitance of the series resonance capacitor C1. And a primary side series resonance circuit including a leakage inductance L1 of the insulating converter transformer PIT including the primary winding N1. Then, the primary side series resonant circuit performs a resonance operation by the switching outputs of the switching elements Q1, Q2, thereby making the operation of the primary side switching converter a current resonance type.
In the case of the present embodiment, as illustrated in FIGS. 3 to 6, the insulating converter transformer PIT has a structure as a control transformer using a saturable reactor. As the level of the control current Nc flowing through the control winding Nc changes, the leakage inductances of the primary winding N1 and the secondary winding N2 of the insulating converter transformer PIT change. That is, in the present embodiment, the leakage inductance L1 forming the primary side series resonance circuit is variable rather than fixed.

  In addition, a center tap tpc is first provided for the secondary winding N2 of the insulating converter transformer PIT in this case. The center tap tpc is connected to the secondary side ground. The anodes of the rectifier diodes Do1 and Do2 are connected to both ends of the secondary winding N2, respectively, and the cathodes of the rectifier diodes Do1 and Do2 are connected to the positive terminal of the smoothing capacitor Co. That is, the rectifier diodes Do1 and Do2 and the smoothing capacitor Co form a double-wave rectifier circuit, and the rectifying operation of the double-wave rectifier circuit generates the secondary side DC output voltage Eo as the voltage across the smoothing capacitor Co. Is done. The secondary side DC output voltage Eo is supplied to a load side (not shown) and is also branched and input as a detection voltage for the control circuit 1.

In this case, the control circuit 1 causes a DC current having a variable level to flow as a detection output to the control winding Nc of the insulating converter transformer PIT in accordance with the level change of the secondary side DC output voltage Eo. Composed. The control circuit 1 in this case may be formed as a class A amplifier circuit that amplifies current according to the level of the secondary side DC output voltage Eo.
As described above, in the isolated converter transformer PIT, the leakage inductance of the primary winding N1 and the secondary winding N2 can be varied according to the level of the control current Ic flowing through the control winding Nc. In this form, the leakage inductance of the primary winding N1 and the secondary winding N2 is variably controlled in accordance with the level change of the secondary side DC output voltage Eo.

Further, in the secondary winding N2, taps tp1 and tp2 are respectively provided at both side winding positions corresponding to a predetermined number of turns from the center tap tpc. The end of the tap tp1 is connected to the anode of the rectifier diode Do3 via the series connection of the controlled winding N R1 of the orthogonal control transformer PRT-1, and the end of the tap tp2 is connected to the orthogonal control transformer PRT-1 It is connected to the anode of the rectifier diode Do4 through a series connection of the controlled winding NR2. The cathodes of the rectifier diodes Do3 and Do4 are connected to the positive terminal of the smoothing capacitor Co1. As a result, the rectifier diodes Do3 and Do4 and the smoothing capacitor Co1 form a double-wave rectifier circuit, and the rectifying operation of the double-wave rectifier circuit generates the secondary side DC output voltage Eo1 as the voltage across the smoothing capacitor Co1. To do.
As for the controlled winding NR1 in the orthogonal control transformer PRT-1, the winding end end is connected to the secondary winding N2 side and the winding start end is connected to the rectifier diode Do3 side. On the other hand, for the controlled winding NR2, the winding start end is connected to the secondary winding N2 side, and the winding end end is connected to the rectifier diode Do4 side. That is, in the rectifier circuit composed of the rectifier diodes Do3 and Do4 and the smoothing capacitor Co1, the controlled windings NR1 and NR2 are inserted so as to have opposite polarities so as to be adapted to the rectification operation of the rectifier circuit. ing.

  Further, in this case, in the secondary winding N2, a tap tp3 is provided at a predetermined winding position between the center tap tpc and the tap tp2, and the end of the tap tp3 is connected to the controlled winding. It is connected to the anode of the rectifier diode Do5 through a series connection of the line NR. The cathode of the rectifier diode Do5 is connected to the cathode of the smoothing capacitor Co2. The rectifier circuit formed by including the rectifier diode Do5 and the smoothing capacitor Co2 is a half-wave rectifier circuit, and generates the secondary side DC output voltage Eo2 as the voltage across the smoothing capacitor Co2.

Here, the stabilization with respect to the secondary side DC output voltage Eo is a constant configuration in which the inductance variable control in the insulating converter transformer PIT is performed mainly by flowing the control current Ic from the control circuit 1 to the control winding Nc. This is performed by the voltage control circuit system.
On the other hand, the stabilization of the secondary side DC output voltage Eo1 is performed by the constant voltage control circuit system including the control circuit 4-1 and the orthogonal control transformer PRT-1, and the secondary side DC output voltage Eo2 is controlled. Is stabilized by a constant voltage control circuit system including the control circuit 4-2 and the orthogonal control transformer PRT-2.
The constant voltage control operation by these constant voltage control circuit systems will be described in order. The internal configuration of the control circuits 4-1 and 4-2 will also be described together with the description of the constant voltage control operation for the secondary side DC output voltages Eo1 and Eo2.

  Subsequently, constant voltage control in the power supply circuit as the embodiment configured as described above will be described. First, referring to FIG. 7, the control current Ic with respect to the AC input voltage VAC and the load variation is described. The change characteristics of the switching frequency fs will be described. The characteristics shown in FIG. 7 are mainly related to constant voltage control of the secondary side DC output voltage Eo.

As described above with reference to FIG. 2, the oscillation / drive circuit 2 of the present embodiment variably controls the switching frequency of the switching elements Q1 and Q2 in accordance with the level of the AC input voltage VAC (commercial AC power supply AC). To do. The switching frequency fs in FIG. 7 indicates this switching frequency control characteristic.
According to this figure, the switching frequency fs is variably controlled so as to increase almost proportionally as the AC input voltage VAC shown on the horizontal axis changes from 80V to 280V. I understand that. It can also be seen that under the condition that the AC input voltage VAC is at the same level, the switching frequency fs is higher at the minimum load power (Pomax) and at the minimum load power (Pomin).

The following operation is obtained by such variable control of the switching frequency fs.
To make the explanation easy to understand, the resonance frequency fo of the primary side series resonance circuit in the power supply circuit shown in FIG. 2 is constant regardless of the level change (load fluctuation) of the secondary side DC output voltage. And In this case, first, the variable range of the switching frequency of the characteristic shown in FIG. 7 corresponds to, for example, the minimum level (for example, VAC = 85V) as the AC100V system to the maximum level (for example, VAC = 288V) as the AC200V system. Is approximately 50 KHz to 105 KHz. On the other hand, the resonance frequency fo is assumed to be a value smaller than 50 KHz. That is, the variable control of the switching frequency fs is performed in a frequency range higher than the resonance frequency. Such switching frequency control is also referred to as upper side control.
When variable control of the switching frequency by such upper side control is performed, the resonance impedance increases as the switching frequency fs increases and the distance from the resonance frequency fo increases. Here, the fact that the resonance impedance becomes higher corresponds to the fact that the amount of power of the switching output transmitted as being excited from the primary side to the secondary side becomes smaller. The level of the secondary side direct-current output voltage generated by the above will also decrease.

  Therefore, in the power supply circuit of the embodiment shown in FIG. 2, the oscillation / drive circuit 2 is variably controlled so as to increase the switching frequency fs as the level of the commercial AC power supply AC (AC input voltage VAC) increases. This means that the control is performed so that the level of the secondary side DC output voltage is decreased in accordance with the level increase of the commercial AC power supply AC (AC input voltage VAC). For example, when considering that factors such as load fluctuations connected to the secondary side DC output voltage are not taken into account, the actual switching frequency control of the oscillation / drive circuit 2 is in the range of AC100V system to AC200V system. In this operation, the level change of the secondary side DC output voltage in accordance with the change of the commercial AC power supply AC (AC input voltage VAC) is suppressed, and for example, the operation is controlled so as to be substantially constant.

  Further, as described above, the control current Ic is output by the control circuit 1 shown in FIG. 2 so as to be variable according to the level of the secondary side DC output voltage Eo. As the change characteristic, as shown in FIG. 7, first, as the level of the AC input voltage VAC (commercial AC power supply AC) increases, it gradually decreases. Further, when comparing the maximum load power (Pomax) and the minimum load power (Pomin), under the condition of the AC input voltage VAC of the same level, as shown in FIG. It increases by shifting by a predetermined level from the time.

In the insulating converter transformer PIT as a saturable reactor, the leakage inductance components of the primary winding N1 and the secondary winding N2 are increased as the control current Ic flowing through the control winding Nc decreases. To work.
Here, when viewed as the primary side series resonance circuit (C1-L1), the resonance frequency fo decreases as the leakage inductance L1 of the primary winding N1 increases. In this case as well, assuming that the switching frequency fs is in a higher frequency region than the resonance frequency fo, and assuming that the switching frequency fs is fixed to simplify the explanation, the resonance frequency As fo decreases, the frequency difference between the switching frequency fs and the resonance frequency fo increases. Therefore, the resonance impedance increases. As the resonance impedance increases, the level of the secondary side DC output voltage is controlled to decrease as described above.
Note that, as the control current Ic increases, the operation is opposite to that described above. That is, the secondary side DC output voltage level is finally controlled to increase.

As the change characteristic of the control current Ic shown in FIG. 7, under the condition that the AC input voltage VAC is at the same level, as described above, the control current Ic is greater at the maximum load power than at the minimum load power. It increases with the level indicated by ΔIc. Here, since the level of the control current Ic is variable and output by the control circuit 1 in accordance with the level of the secondary side DC output voltage Eo, the load power here is the secondary side DC output voltage. It refers to the power consumption of the load connected to Eo.
From the above, the following can be said. That is, in order to make the explanation easy to understand, when it is assumed that the level of the AC input voltage VAC is constant, the secondary side DC output voltage Eo accompanying the fluctuation of the load connected to the secondary side DC output voltage Eo. As the level changes, the level of the control current Ic is also variably controlled. The variable range is indicated by ΔIc according to the range between the maximum load power and the minimum load power. Within this range, the level of the control current Ic increases as the heavy load condition is met. In this way, it becomes variable.

In this case, when the load connected to the secondary side DC output voltage Eo becomes heavy and its level is lowered, the control current Ic is variably controlled so as to increase. As described above, as the level of the control current Ic increases, the level of the secondary side DC output voltage Eo is controlled to increase.
On the other hand, when the load connected to the secondary side DC output voltage Eo is lightened and the level is increased, the control current Ic is variably controlled so as to decrease. As the level of the control current Ic decreases, the level of the secondary side DC output voltage Eo is controlled to decrease.
Therefore, in the control circuit 1, the level of the control current Ic is varied in accordance with the load state of the secondary side DC output voltage Eo, whereby the level of the secondary side DC output voltage Eo is variably controlled. Will be. That is, the constant voltage control circuit system including the control circuit 1 is configured to perform constant voltage control with respect to load fluctuations with respect to the secondary side DC output voltage Eo.

  On the secondary side of the power supply circuit of the embodiment shown in FIG. 2, in addition to the secondary side DC output voltage Eo stabilized as described above, the secondary side DC output voltage Eo1, the secondary side DC output The voltage Eo2 is output. In this embodiment, the secondary side DC output voltage Eo1 and the secondary side DC output voltage Eo2 are also output in a stabilized state. Subsequently, a configuration for stabilizing the secondary side DC output voltage Eo1 and the secondary side DC output voltage Eo2 will be described.

  As a configuration for stabilizing the secondary side DC output voltage Eo1, the control circuit 4-1 is combined with the orthogonal control transformer PRT-1. Similarly, the configuration for stabilizing the secondary side DC output voltage Eo2 is such that the control circuit 4-2 is combined with the orthogonal control transformer PRT-2. First, the structure of these orthogonal control transformers PRT-1 and PRT-2 will be described.

As the orthogonal control transformers PRT-1 and PRT-2, for example, the core shape and winding structure shown in FIG. 3 or FIG. 4 can be employed.
In the case of the orthogonal control transformer PRT-1, for example, in FIG. 3 or FIG. 4, the primary winding N1 and the secondary winding with respect to the position where the primary winding N1 and the secondary winding N2 are wound. Instead of N2, controlled windings NR1 and NR2 are wound. In the case of the orthogonal type control transformer PRT-2, a structure in which a set of controlled windings NR is wound instead of the primary winding N1 and the secondary winding N2 may be used.

Then, the control circuit 4-1 for stabilizing the secondary side DC output voltage Eo1 operates by inputting the secondary side DC output voltage Eo, which is the voltage across the smoothing capacitor Co, as a power source. It is formed as an error amplifier that obtains an amplified output corresponding to the level error of the output voltage Eo1.
In the control circuit 4-1, the secondary side DC output voltage Eo1 as the detection voltage is divided by the voltage dividing resistors R31 to R32, and this divided level is input to the control terminal of the shunt regulator Q11. It has become. In the shunt regulator Q11, a current of a level corresponding to the error of the voltage division level (secondary DC output voltage Eo1 level) input to the control terminal is supplied from the power supply line (secondary DC output voltage Eo line). The current flows through the resistor R33.

  Here, the base current of the PNP transistor Q12 is flowing from the power supply line (secondary DC output voltage Eo line) through the series connection of the resistor R33 and the resistor R34. The collector of the transistor Q12 is connected to the power supply line (secondary DC output voltage Eo line) via the control winding Nc of the orthogonal control transformer PRT-1. The emitter is grounded to the secondary side ground. As a result, a collector current at a level amplified according to the base current flows through the control winding Nc of the orthogonal control transformer PRT-1 as the control current Ic1.

Here, if the load is reduced and the secondary side DC output voltage Eo1 is changed so as to increase, the control circuit 4-1 applies it to the control terminal of the shunt regulator Q11 accordingly. Since the voltage dividing level of the voltage dividing resistors R31 to R32 also increases, the current flowing through the shunt regulator Q11 also increases. In response to this, the base potential obtained as the potential across the resistor R35 inserted between the base and emitter of the transistor Q12 is lowered and the base current is reduced, so that the level of the control current Ic1 is also lowered. become. In the orthogonal control transformer PRT-1, the inductances of the controlled windings NR1 and NR2 are increased as the level of the control current Ic1 decreases.
That is, depending on the control circuit 4-1, the inductance of the controlled windings NR1 and NR2 of the orthogonal control transformer PRT-1 is increased as the secondary side DC output voltage Eo1 increases. As the secondary DC output voltage Eo1 decreases, an operation for increasing the inductance of the controlled winding NR of the orthogonal control transformer PRT-1 is obtained.

Here, the controlled windings NR1 and NR2 of the orthogonal control transformer PRT-1 are inserted into the rectified current paths of the rectifier diodes Do3 and Do4 for generating the secondary side DC output voltage Eo1, When the inductance is variably controlled as described above, the rectified output voltage level is changed. As a result, the voltage across the smoothing capacitor Co1 charged with this rectified output voltage, that is, the level of the secondary side DC output voltage Eo1 changes. The level change of the secondary side DC output voltage Eo1 tends to decrease as the inductances of the controlled windings NR1 and NR2 increase. As a result of the operation of the control circuit 4-1, the secondary side DC output voltage Eo1 is controlled to be stabilized.
Note that the control circuit 4-1 includes a start-up circuit including the transistor Q14. The configuration and operation of this start-up circuit portion will be described later.

Also, the control circuit 4-2 for stabilizing the secondary side DC output voltage Eo2 includes the voltage dividing resistors R41-R42, the shunt regulator Q21, the resistor R43, the resistor R44, the resistor R45, and the transistor Q22. -1 voltage dividing resistors R31-R32, shunt regulator Q11, resistor R33, resistor R34, resistor R35, and transistor Q12 are connected in the same connection manner to form an error amplifier.
However, in this case, the secondary side DC output voltage Eo2 which is the voltage across the smoothing capacitor Co2 is divided by the voltage dividing resistors R41-R42 and inputted to the control terminal of the shunt regulator Q21. The collector current of the transistor Q22, which is an error amplification output, is caused to flow as a control current Ic2 to the control winding Nc of the orthogonal control transformer PRT-2. Here, the controlled winding NR of the orthogonal control transformer PRT-2 is inserted in the rectified current path of the half-wave rectifier circuit for generating the secondary side DC output voltage Eo2.
That is, the control circuit 4-2 is an error amplifier that obtains an amplified output corresponding to the level error of the secondary side DC output voltage Eo2, and is provided to stabilize the secondary side DC output voltage Eo2. In addition, it is the same as that of the control circuit 4-1 in that it operates by inputting the secondary side DC output voltage Eo that is the voltage across the smoothing capacitor Co as a power source.
The control circuit 4-1 configured as described above performs the same operation as that described as the control circuit 4-1 in response to the level fluctuation of the secondary side DC output voltage Eo 2. As a result, the inductance of the controlled winding NR of the orthogonal control transformer PRT-2 is variably controlled according to the level of the secondary side DC output voltage Eo2. That is, a constant voltage for the secondary side DC output voltage Eo2 is achieved by inductance control in the orthogonal control transformer.

  In addition, the control circuit 4-1 described above is provided with a startup circuit. This starting circuit comprises a PNP transistor Q14, resistors R36, R37, and ZD1. The collector of the transistor Q14 is connected to the positive terminal of the smoothing capacitor Co (secondary DC output voltage Eo line) via the resistor R36. The base of the transistor Q14 is also connected to the positive terminal of the smoothing capacitor Co (secondary DC output voltage Eo line) through the resistor R37, and from the cathode of the Zener diode ZD1 to the secondary through the anode. Connected to side ground. The emitter of the transistor Q14 is connected to the connection point between the end of the control winding Nc of the orthogonal control transformer PRT-1, the end of the control winding Nc of the orthogonal control transformer PRT-2, and the positive terminal of the smoothing capacitor Co1. Connected to each other.

As a practical example of the power supply circuit shown in FIG. 2, even if the secondary side DC output voltages Eo, Eo1, and Eo2 rise immediately after the commercial AC power supply AC is turned on, the control circuits 4-1 and 4-2 start operating. The control currents Ic of the control windings Nc of the orthogonal control transformers PRT-1 and PRT-2 are difficult to flow.
However, since the start-up circuit is provided as described above, first, in response to the rise of the secondary side DC output voltage Eo, the transistor Q14 in the start-up circuit is turned on, forcing the secondary side DC output voltage. Current flows from the line Eo to the control windings Nc of the orthogonal control transformers PRT-1 and PRT-2. When a certain time elapses after the power is turned on and the voltage across the smoothing capacitor Co rises above a certain level, the Zener diode ZD1 becomes conductive and the transistor Q14 is turned off. At this time, the secondary side DC output voltages Eo1 and Eo2 have risen to a level higher than necessary, and hence the level control of the normal control current Ic is performed by the control circuits 4-1 and 4-2 thereafter. Transition to the state.

As can be seen from the above description, in the power supply circuit of the present embodiment shown in FIG. 2, the switching frequency is variably controlled in response to the level change of the AC input voltage VAC corresponding to the AC100V system to AC200V system. Thus, for example, under the condition where the load condition is constant, the level of the secondary side DC output voltage is made substantially constant.
Of the secondary side DC output voltages Eo1, Eo2, and Eo3, the secondary side DC output voltage Eo1 that has the largest load fluctuation is variably controlled by controlling the leakage inductance of the insulating converter transformer PIT. Is to be stabilized.
Further, the remaining secondary side DC output voltages Eo1 and Eo2 are respectively provided with orthogonal control transformers PRT-1 and PRT-2 and inserted into the rectified current path. Similarly, by varying the inductance of the controlled winding NR, stabilization against load fluctuation is achieved.

  That is, in the power supply circuit of the present embodiment, first, switching frequency control is performed for fluctuations in the secondary side DC output voltage caused by the change in the input level of the commercial AC power supply in the range of AC100V system to AC200V system. As for the level change corresponding to the load fluctuation for each secondary side DC output voltage (Eo, Eo1, Eo2), variable control of the leakage inductance of the insulating converter transformer PIT or the rectified current Stabilization is achieved by variable inductance control of the inductor inserted in the path. In other words, the level change of the secondary side DC output voltage is regarded as having two factors: the change of the input level of the commercial AC power supply AC and the change of the load state connected to the secondary side DC output voltage, Corresponding to each of these factors, it is configured to be stabilized by different control systems.

For example, as described in the prior art, when only the switching frequency control is used, the change in the commercial AC power supply AC ranging from the AC 100V system to the AC 200V range and the load fluctuation under the corresponding heavy load condition are attempted. In some cases, the maximum variable width of the switching frequency is insufficient with respect to the change amount of the secondary side DC output voltage, and proper constant voltage control cannot be performed.
On the other hand, in the configuration of the constant voltage control as the present embodiment described above, stabilization corresponding to fluctuations in the commercial AC power supply AC and stabilization corresponding to load fluctuations are shared by different control methods. In addition, these control methods are performed simultaneously. As a result, the secondary side DC output corresponding to the input level range of a wide range of commercial AC power sources, for example, AC100V system to AC200V system (for example, VAC = 85V to 288V), and the load fluctuation range which is correspondingly large. It will be guaranteed that the voltage Eo is stabilized. That is, a sufficiently practical power supply circuit can be obtained as a power supply circuit that is compatible with a wide range and has a large load fluctuation range as a load condition.

As an actual example of the power supply circuit shown in FIG. 2, when the insulating converter transformer PIT adopts the structure shown in FIG. 4, for example, the size of the three-dimensional core is 20 mm × 20 mm × 20 mm, and the gap G The length is 50 μm to 100 μm. Further, the number of turns for the primary winding portion N1 was 50T (turns), the secondary winding N2 was 10T, and the control winding Nc was 1000T.
When the component is selected in this way, the switching frequency fs under the condition that the load input range Po is 0 W to 90 W and the load frequency range is the range of the AC input voltage VAC = 85 to 288 V, the control current The change characteristics of Ic are as follows.
fs = 50 KHz to 105 KHz
Ic = 60mA ~ 0

Here, as the load conditions, the minimum load power Pomin = 0 W and the maximum load power Pomax = 90 W are set because it is assumed that the power supply circuit shown in FIG. is there. The printer device here refers to a device that is connected to, for example, a personal computer and has a function of performing printing in accordance with an instruction of an application of the personal computer.
In such a printer device, for example, when the printer is not performing a printing operation, the load is almost open and unloaded. That is, as the load condition, the minimum load power is in a state that the minimum load power Pomin = 0 W as described above. Therefore, for example, even if the maximum load power is about 90 W, the difference between the maximum load power and the minimum load power tends to be correspondingly large. In addition, for example, when trying to cope with a wide range of commercial AC power input of AC input voltage VAC = 85 to 288 V, for example, as described in the related art, if stabilization is simply achieved by switching frequency control, switching is performed. This makes it difficult to cover the fluctuation range of the secondary side DC output voltage in the maximum frequency variable range.
From this, it can be said that the merit becomes more effective and useful when applied as a power supply device such as a printer device as the power supply circuit of the present embodiment.

For example, as a practical example of the oscillation / drive circuit 2, the maximum variable range of the switching frequency fs when a specific general-purpose driving IC is used is 50 KHz to 250 KHz, but the required control range Δfs (60 KHz to 195 KHz) ), The control range of the switching frequency described above is within the maximum variable range of 50 KHz to 250 KHz.
Here, for confirmation, the load fluctuation range of the load power Po = 0W to 90W described above is a load condition arbitrarily set for obtaining the experimental result, and therefore, in the present embodiment. As a power supply circuit based on it, it is possible to achieve appropriate stabilization in response to a wider load fluctuation range by changing part selection and the like. Further, it is possible to cope with a wider input level range of a commercial AC power supply.

By adopting such a configuration, the power supply circuit of the present embodiment is, for example, similar to the circuit shown in FIG. The rectifying circuit system that generates Ei) does not need to adopt a configuration for switching the rectifying operation, and can be a normal full-wave rectifying circuit. Therefore, one smoothing capacitor for generating the DC input voltage is sufficient. Further, since the rectifier circuit system is switched, the electromagnetic relay can be made unnecessary.
In addition, the power supply circuit shown in FIG. 1 requires a total of four switching elements, and accordingly, it is necessary to add a circuit configuration as the drive circuit 2, and the circuit configuration is relatively complicated. It can be said that there is.
On the other hand, in this embodiment, since the switching converter is fixed by the half-bridge coupling method, the switching element may be two stones, and the configuration of the drive circuit system is simpler. It can be. For this reason, as a whole, the board mounting area can be reduced as compared with the power supply circuit shown in FIG. 1, and a low-cost power supply circuit can be obtained.
In the circuit shown in FIG. 1, four stone switching elements perform switching operation in the full bridge operation, and three stone switching elements perform switching operation in the half bridge operation. On the other hand, in this embodiment, since the switching operation of the two stone switching elements is always performed, it can be said that the switching noise is reduced accordingly.

Further, in this embodiment, as constant voltage control for wide range, continuous variable control of switching frequency, continuous variable control of leakage inductance of insulation converter transformer PIT, or inductance inserted in a rectification current path The continuous variable control is performed. In other words, for example, it does not have a switching function by an electromagnetic relay. In the present embodiment, the switching drive is of the separately excited type, so that the variable frequency control of the switching frequency and the leakage inductance of the insulating converter transformer PIT can be performed by the configuration including one set of the insulating converter transformer PIT. .
For this reason, for example, even if an abnormality such as an instantaneous power failure or a steep drop occurs in the commercial AC power supply AC, the level of the rectified smoothing voltage Ei and the secondary side DC output voltage according to the abnormality is followed. The switching frequency and the control currents Ic, Ic1, and Ic2 are simultaneously varied to protect the circuit and prevent malfunction of the electromagnetic relay. Therefore, even if a protective circuit is not particularly provided, the power supply circuit according to the present embodiment does not break down due to an abnormality in the commercial AC power supply AC.

Further, in the configuration of the power supply circuit as the embodiment shown in FIG. 2, the switching frequency control is executed when the oscillation / drive circuit 2 provided on the primary side detects the rectified and smoothed voltage Ei on the same primary side. . That is, the switching frequency control is completed on the primary side.
The leakage inductance of the insulating converter transformer PIT is controlled by feeding back the level of the secondary side DC output voltage Eo so that the control winding of the insulating converter transformer PIT is controlled from the control circuit 1 that is supposed to be on the secondary side. This is done by passing a control current Ic through Nc. Here, as understood from the structures shown in FIGS. 3 to 6, the control winding Nc of the insulating converter transformer PIT is galvanically isolated from the primary winding N1. Therefore, the leakage inductance control system of the insulating converter transformer PIT is completed on the secondary side. From this, it can be said that at least the constant voltage control system of the secondary side DC output voltage Eo is not fed back between the primary side and the secondary side. Thereby, for example, for constant voltage control, a component element such as a photocoupler for DC-insulating the primary side and the secondary side can be eliminated. For example, when switching frequency control is performed with the configuration shown in FIG. 1, a system that feeds back from the secondary control circuit 1 to the primary oscillation / drive circuit 2 is necessary. It is necessary to insulate the secondary side in a DC manner.
In addition, the configuration of the embodiment shown in FIG. 2 is to stabilize the remaining secondary side DC output voltages Eo1 and Eo2 by providing orthogonal control transformers PRT-1 and PRT-2 on the secondary side. I am trying to plan. That is, the constant voltage control system for the secondary side DC output voltages Eo1 and Eo2 is completed on the secondary side, and therefore it is not necessary to insulate the primary side and the secondary side in a DC manner. Has been.

In addition, the power supply circuit shown in FIG. 2 is configured to generate and take out a plurality (multiple) of secondary side DC output voltages from one power supply circuit, as can be understood from the above description. I can say that. Usually, the plurality of secondary side DC output voltages are generated corresponding to different load conditions.
In such a configuration, for example, one secondary side DC output voltage (Eo in FIG. 2) is stabilized by switching frequency control or inductance control of the insulating converter transformer PIT, and the remaining secondary side DC output voltage ( In FIG. 2, in order to stabilize the individual secondary side DC output voltages for Eo1, Eo2), it is known that, for example, a constant voltage control system including a series regulator, a step-down converter, or a magnetic amplifier is used. It has been.

However, as is well known, in the configuration stabilized by the series regulator, the power loss is very large, which is disadvantageous in terms of power conversion efficiency, and the problem of heat generation cannot be ignored.
In addition, it can be said that the step-down converter also has a large power loss. The step-down converter is a single switching converter, and performs an independent switching operation with respect to the main primary side switching converter. For this reason, there is a problem that interference of different switching frequencies is likely to occur, and the amount of noise generated in the power supply circuit is increased accordingly.

On the other hand, as in the present embodiment shown in FIG. 2, the orthogonal control transformers (PRT-1, PRT-2) in which the controlled winding NR is inserted in the rectified current path and the control circuit (4- When the configuration of the constant voltage control system consisting of (1,4-2) is adopted, the control power required for changing the inductance of the controlled winding NR by the control circuit is very small. From this, the power loss can be significantly reduced as compared with the case where the secondary side DC output voltage is stabilized by the above-described series regulator, step-down converter, and magnetic amplifier. Thereby, for example, it is not necessary to provide a heat sink.
Further, in FIG. 2, the actual circuit for stabilizing the secondary side DC output voltages Eo1 and Eo2 includes a control transformer as a saturable reactor, and a DC current (control current) level to be passed through the control winding of the control transformer. It is sufficient to provide a circuit for varying the voltage according to the secondary DC output voltage level to be controlled. As a result, compared with the case where the secondary side DC output voltage is stabilized by a series regulator, a step-down converter, or a magnetic amplifier, the cost is very low.
Furthermore, the constant voltage control operation using the control transformer is a control of variably controlling the DC current (control current) level to be passed through the control winding of the control transformer, and the switching operation independent of the primary side switching converter is performed. Absent. Accordingly, there is no interference with different switching frequencies as in the case of adopting the step-down converter, and the amount of noise generated in the power supply circuit is reduced accordingly.

  Further, the circuit system for stabilizing the secondary side DC output voltages Eo1 and Eo2 shown in FIG. 2 is configured by adding a small number of component circuits to the control circuits 4-1 and 4-2, thereby preventing a load short circuit. It also functions as a protection circuit. This point will be described with reference to FIG. 8 using the control circuit 4-1 as an example. In FIG. 8, the same parts as those in FIG.

In the control circuit 4-1, shown in FIG. 8, a capacitor C2 is inserted between the secondary DC output voltage Eo and the control winding Nc of the orthogonal control transformer PRT-1 with the polarity shown in the figure. The capacitor C2 in this case is an electrolytic capacitor, the positive electrode of the capacitor C2 is connected to the secondary side DC output voltage Eo line, and the negative electrode is connected to the end of the control winding Nc.
The emitter of the transistor Q5 is connected to the positive electrode of the capacitor C2, and the collector of the transistor Q5 is connected to the negative electrode. The resistor R47 is a base-emitter resistor of the transistor Q5.
The base of the transistor Q5 is connected to the collector of the transistor Q6 via the resistor R48. The emitter of the transistor Q6 is connected to the secondary side ground. The base of the transistor Q6 is connected to the line of the secondary side DC output voltage Eo1 through the resistor R50. The resistor R19 is a base-emitter resistor of the transistor Q6.

In such a configuration of the control circuit 4-1, for example, when the commercial AC power supply AC is turned on and the secondary side DC output voltage Eo rises to a specified level, first, from the line of the secondary side DC output voltage Eo. A control current Ic1 is supplied to the control winding Nc via the capacitor C2.
At this time, the secondary side DC output voltage Eo1 also rises in response to the turning on of the commercial AC power supply AC. However, when the secondary side DC output voltage Eo1 rises to a level equal to or higher than a predetermined level (for example, 2V), the transistor Q6 is A base-emitter voltage sufficient for conduction is obtained, and the transistor Q6 is turned on. In response, transistor Q5 is also turned on.
After the transistor Q5 is turned on, the control current Ic flows from the line of the secondary side DC output voltage Eo to the path passing through the emitter collector of the transistor Q5. Thereafter, as described with reference to FIG. 2, the level of the control current Ic1 is controlled by the operation of the error amplifier including the shunt regulator Q11 and the transistor Q12 according to the level of the secondary side DC output voltage Eo2. As a result, the secondary side DC output voltage Eo1 is stabilized.

Here, it is assumed that a load short circuit has occurred in the secondary side DC output voltage Eo1. In this case, the secondary side DC output voltage Eo1 drops to the 0 level, but in response to this, the transistor Q6 is controlled to be turned off from the previous on state, and accordingly the transistor Q5 also transitions to the off state.
Thus, when the transistor Q5 transitions to the OFF state, the path for supplying the control current Ic from the line of the secondary side DC output voltage Eo is cut off, so that the control current Ic becomes 0 level. As a result, the inductances of the controlled windings NR1 and NR2 increase.

At this time, with respect to the secondary side DC output voltage Eo2, the zero level continues due to the occurrence of a load short circuit.
The rectified current that is about to flow to the smoothing capacitor Co1 is normally at a very high level due to a load short-circuit, but as described above, the control current Ic is set to the 0 level and is covered. Since the inductances of the control windings NR1 and NR2 are increased, the level of the rectified current flowing at a high frequency according to the switching period is suppressed to a level lower than that in the steady state.
That is, the control circuit 4-1 shown in FIG. 8 has a function as a protection circuit corresponding to a load short circuit of the secondary side DC output voltage Eo1. The load short-circuit protection function can be realized by a simple and low-cost circuit mainly including the transistors Q5 and Q6 and the electrolytic capacitor C2 and several other resistance elements. ing.
For example, in a configuration in which constant voltage control is performed by a step-down converter or a magnetic domain amplifier circuit, it is necessary to form and connect a more complicated load short circuit protection circuit if a protection function for load short circuit is to be provided. As a result, the circuit scale becomes larger than that of the present embodiment, and the cost increases.

Although not shown here, in a constant voltage control circuit system having an orthogonal control transformer PRT on the secondary side, each end of one resistance element is connected to the controlled winding NR1 and the rectifier diode Do3. It is also possible to provide the point and a connection point between the controlled winding NR2 and the rectifier diode Do4. By connecting the resistance elements in this way, noise as parasitic vibration that occurs at the timing when the rectifier diode turns off is removed. Thus, the withstand voltage of the rectifier diode can be reduced, and the cost of the rectifier diode can be reduced accordingly. In addition, since the low-voltage product can obtain better characteristics, the reliability of the circuit is also improved.
The same applies to the constant voltage control circuit system for the secondary side DC output voltage Eo2 when the control circuit 4-2 is configured as shown in FIG.

Here, as a comparison with the present embodiment, FIG. 10 shows a configuration example of a power supply circuit including a constant voltage control system using a step-down converter on the secondary side. Further, FIG. 11 shows a configuration example of a power supply circuit including a constant voltage control system using a magnetic amplifier on the secondary side.
Note that the power supply circuit shown in FIGS. 10 and 11 is mainly intended to show a constant voltage control system using a step-down converter on the secondary side or a magnetic amplifier, and therefore a configuration corresponding to a wide range is taken. No. 10 and FIG. 11, the same parts as those in FIG. 1, FIG. 2 and FIG.

First, on the primary side of the power supply circuit shown in FIG. 10, the commercial AC power supply AC is rectified and smoothed by a full-wave rectifier circuit, and the rectified and smoothed voltage Ei corresponding to the same level of the AC input voltage VAC at both ends of the smoothing capacitor Ci. Has been generated. As a switching converter for inputting and switching the rectified and smoothed voltage Ei (DC input voltage), the MOS-FET switching elements Q1 and Q2 are half-bridge coupled and the leakage of the primary winding N1 of the insulating converter transformer PIT. A current resonance type converter is constituted by the inductance L1 and the capacitance of the series resonance capacitor C1. In addition, a partial voltage resonance capacitor Cp is connected to the switching element Q2 side, so that a partial voltage resonance operation can be obtained when the switching element is turned off. That is, it is a composite resonance type converter capable of obtaining a current resonance type operation and a partial voltage resonance operation.
The switching elements Q1, Q2 are driven by the oscillation / drive circuit 2 by a separate excitation method.

In this case, the secondary side of the insulating converter transformer PIT generates and outputs three secondary side DC output voltages Eo, Eo1, and Eo2 as a plurality of secondary side DC output voltages.
A center tap connected to the secondary side ground is applied to the secondary winding N2 of the insulating converter transformer PIT, and the secondary side DC output voltage Eo is rectified connected to both ends of the secondary winding N2. It is generated by a double-wave rectifier circuit composed of diodes Do1 and Do2 and a smoothing capacitor Co.

  In this case, the control circuit 1 obtains, as a control output, a current or voltage whose level is variable according to the level of the secondary side DC output voltage Eo, and outputs it to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1, Q2 such that the switching frequency is variably controlled according to the input control output level. As a result, the secondary side DC output voltage Eo is made constant.

In this case, a step-down converter comprising a MOS-FET switching element Q41, a rectifier diode Do3, a choke coil L11, and a smoothing capacitor Co1 as shown in the figure with respect to the secondary side DC output voltage Eo. It is connected.
In this step-down converter, the secondary DC output voltage Eo is inputted and the alternating voltage obtained by switching by the switching element Q41 is half-wave rectified by the rectifier diode Do3 and the choke coil L11 to charge the smoothing capacitor Co1. By doing so, the secondary side DC output voltage Eo1 stepped down from the secondary side DC output voltage Eo is generated.

The constant voltage control for the secondary side DC output voltage Eo1 is performed by the control circuit 6-1.
The secondary DC output voltage Eo1 is input to the control circuit 6-1, and a drive signal output to the gate of the switching element Q41 in accordance with the level of the input secondary DC output voltage Eo1, for example, While keeping the switching frequency constant, the pulse width within one period is varied. That is, PWM control is performed. As a result, the conduction angle of the switching element Q41 within one switching cycle is varied, and as a result, the level of the secondary side DC output voltage Eo1 also changes. In this way, the secondary side DC output voltage Eo2 is stabilized by variably controlling the level of the secondary side DC output voltage Eo1.

  In addition, as a constant voltage control circuit system that generates the secondary side DC output voltage Eo2 and performs constant voltage control, the MOS− that forms the constant voltage control circuit system corresponding to the secondary side DC output voltage Eo1 is used. The switching elements Q42, rectifier diodes Do5, choke coils L12, smoothing capacitors Co2 of MOS-FETs are connected in the same manner as the FET switching elements Q41, rectifier diodes Do3, choke coils L11, smoothing capacitors Co1, and control circuit 6-1. The step-down converter is formed by the control circuit 6-2. As a result, the secondary side DC output voltage Eo2 is also stabilized by the PWM control operation of the step-down converter.

  Next, a configuration that stabilizes the secondary side DC output voltage by providing a magnetic amplifier on the secondary side will be described with reference to FIG. In this figure, the configuration on the primary side is the same as in FIG. Further, as the secondary side DC output voltage, three secondary side DC output voltages Eo, Eo1, Eo2 are generated, and the secondary side DC output voltage Eo includes a control circuit 1 and an oscillation / drive circuit 2. The stabilization by the switching frequency control by the constant voltage control system is the same as in FIG. In the power supply circuit shown in FIG. 11, the secondary side DC output voltages Eo1 and Eo2 are each stabilized by a magnetic amplifier circuit. Hereinafter, this point will be described.

First, the circuit system for generating the stabilized secondary side DC output voltage Eo1 is configured as follows.
The secondary winding N2 of the insulating converter transformer PIT is provided with a center tap that is grounded to the secondary side ground, and on both sides of the winding position corresponding to a predetermined number of turns starting from the center tap tpc. Taps tp1 and tp2 are provided. Then, the anodes of the rectifier diodes Do3 and Do4 are connected to the taps tp1 and tp2, respectively, as shown in the figure, and the cathode sides of the rectifier diodes Do3 and Do4 are connected to the positive terminal of the smoothing capacitor Co1. A double-wave rectifier circuit is formed. Then, the secondary side DC output voltage Eo1 is generated as the voltage across the smoothing capacitor Co1.

In addition, the double-wave rectifier circuit includes a constant voltage circuit (magnetic amplifier constant voltage circuit) including a magnetic amplifier.
In this magnetic amplifier constant voltage circuit, first, a saturable inductor (choke coil) SR1 is inserted between the tap tp1 of the secondary winding N2 and the anode of the rectifier diode Do3, and the tap tp2 and the anode of the rectifier diode Do4 are connected. A saturable inductor SR2 is inserted between them. Further, the cathode of the reset voltage varying diode DV1 is connected to the anode of the rectifier diode Do3, and the cathode of the reset voltage varying diode DV2 is connected to the anode of the rectifier diode Do4. The anodes of diodes DV1 and DV2 are connected to the collector of transistor Q31. The emitter of the transistor Q31 is connected to the positive line of the secondary side DC output voltage Eo1 through the resistor R51.

  In this case, the control circuit 7-1 controls the magnetic flux of the saturable inductors SR1 and SR2 in order to stabilize the secondary side DC output voltage Eo1. The control circuit 7-1 is formed as an error amplifier having a shunt regulator or the like, and variably controls the base current level of the transistor Q31 according to the level of the input secondary side DC output voltage Eo1. Accordingly, the collector current level of transistor Q31 is varied. Since the collector of the transistor Q31 is connected to the connection point of the anodes of the reset voltage variable diodes DV1 and DV2, the magnetic flux reset voltage in the saturable inductors SR1 and SR2 can be changed by changing the collector current level. The control voltage for changing is changed.

  Here, the above-described saturable inductor SR (SR1, SR2) is configured by winding a single wire Ln with a required number of turns around a circular toroidal core CR as shown in FIG. 12, for example. The

  FIG. 13 shows a BH curve diagram when cobalt-based amorphous is selected as the core material of the saturable inductor SR configured as described above. As can be seen from this figure, the BH characteristic of the saturable inductor SR has a large squareness ratio as a hysteresis characteristic.

  The operation of the magnetic amplifier provided with such a saturable inductor SR is as shown in FIG. In FIG. 14, a voltage V3 indicates a potential between the tap tp1 of the secondary winding N2 (a connection point between the saturable inductor SR1 and the secondary winding N2) and the center tap of the secondary winding N2. The voltage VL1 indicates the voltage across the saturable inductor SR1. A current ID1 indicates a rectified current flowing into the rectifier diode Do3.

Corresponding to the period t0 to t1, the voltage V3 is in a positive polarity state. At this time, the saturable inductor SR1 is in an unsaturated state (B0>B> B1). At this time, since the relationship between the voltages V3 and VL1 is V3≈VL1, the rectified current ID1 does not flow through the rectifier diode Do3.
In the next period t1 to t2, the saturable inductor SR1 is in a saturated state (B = B1), so that the voltage VL1 becomes almost 0 level. As a result, the relationship between the voltages V3 and VL1 is V3> VL1, and the rectified current ID1 begins to flow through the rectifier diode Do3.

In the next period t2 to t3, the output voltage adjustment circuit 11 shown equivalently in FIG. 15 operates. In FIG. 11, the output voltage adjustment circuit 11 is a control circuit 7-1 to which the secondary side DC output voltage Eo1 is input. As can be seen from FIG. 15, the control circuit 7-1 has a configuration as an error amplifier. That is, the level of the secondary side DC output voltage Eo1 divided by the voltage dividing resistors Ro1 and Ro2 is compared with the reference voltage Vref, and the error is amplified by the amplifier circuit composed of the operational amplifier OP and the feedback circuit (Ca, Ra). Thus, the signal is output via the resistor Rb.
Then, in response to the output from the output voltage adjustment circuit 11 obtained as described above, the reset circuit 10 passes a reset current to the saturable inductor SR1. The reset circuit 10 equally shows a function as a reset circuit including the resistor R51, the transistor Q31, the diodes DV1 and DV2, and the saturable inductors SR1 and SR2 in FIG.
At this time, the reset circuit 10 supplies the reset current by causing a current having a level corresponding to the output level from the output voltage adjustment circuit 11 to flow through the saturable inductor SR1 via the resistor Rc → the transistor Q31 → the diode DV1. It is obtained by. With this reset current, the saturable inductor SR1 is reset so as to return the magnetic flux density to B0.

The time length of the period t0 to t1 in which the saturable inductor SR1 is in the unsaturated state is determined by the reset amount (reset current level) in the period t2 to t3.
Therefore, the reset amount is increased in response to an increase in the level of the secondary side DC output voltage Eo1 in response to a light load tendency. As a result, the residual magnetic flux density B0 becomes B0A as shown in FIG. 17, so that the periods t0 to t1, which are unsaturated periods, also become the periods t0A to t1A as shown in FIG. Can be long. In this way, if the period of the unsaturated state becomes longer, the period during which the rectified current ID1 does not flow also becomes longer, so the power supply time to the load per unit time is also shortened, and the secondary side DC output The level of the voltage Eo1 also decreases accordingly.
Such an operation is also performed on the saturable inductor SR2 side at a timing when the waveform shown in FIG. 14 has a phase difference of 180 °.
In this way, in the circuit shown in FIG. 11, the secondary side DC output voltage Eo1 obtained by the two-wave rectification is stabilized.

Further, the secondary side DC output voltage Eo2 shown in FIG. 15 is also configured to perform constant voltage control by a magnetic amplifier constant voltage circuit in the same manner as the secondary side DC output voltage Eo1.
However, as a basic configuration for generating the secondary side DC output voltage Eo2, a tap tp3 is further provided for the secondary winding N2 at a winding position corresponding to a predetermined number of turns from the center tap. A half-wave rectifier circuit including a rectifier diode Do5 and a smoothing capacitor Co2 is formed for tp3.
Then, for this half-wave rectifier circuit, a saturable inductor (choke coil) SR3, a reset voltage variable diode DV3, a reset current output transistor Q32, a resistor R52, and a control circuit 7-2 are illustrated as shown. They are connected to form a magnetic amplifier constant voltage circuit.

The present invention is not limited to the configuration as the above-described embodiment.
For example, the basic configuration of the power supply circuit as the embodiment shown in FIG. 2 is not a configuration that outputs a plurality of secondary side DC output voltages, but a configuration that outputs one secondary side DC output voltage (Eo). It is. That is, the present invention is characterized in that the switching frequency control and the leakage inductance control of the insulating converter transformer PIT are combined to stabilize the secondary side DC output voltage.

Further, in the configuration as the embodiment shown in FIG. 2, the change in the commercial AC power supply AC (AC input voltage VAC) is used as a constant voltage control configuration that combines the switching frequency control and the leakage inductance control of the insulating converter transformer PIT. The fluctuation of the secondary DC output voltage caused by the above is stabilized by switching frequency control, and the fluctuation according to the load fluctuation is stabilized by leakage inductance control of the insulating converter transformer PIT.
However, according to the present invention, this relationship is exchanged to stabilize the secondary side DC output voltage due to the change of the commercial AC power supply AC (AC input voltage VAC) by the leakage inductance control of the insulating converter transformer PIT. It is also possible to adopt a configuration in which fluctuation caused by one load fluctuation is stabilized by switching frequency control.
For this purpose, a control output corresponding to the level of the secondary side DC output voltage Eo is input from the control circuit to the oscillation / drive circuit 2, and the oscillation / drive circuit 2 is set to the level of the secondary side DC output voltage Eo. The switching frequency is controlled according to the change. In addition, in this case, another control circuit (error amplifier) is provided, and this control circuit allows a control current (DC current) whose level is variable according to the level of the AC input voltage VAC to be orthogonal. It is configured to flow through a control winding of an insulating converter transformer PIT configured as a saturable inductor.

In addition, for example, as a switching element, an IGBT (Insulated Gate Bipolar Transistor) or the like that can be used by another excitation type, the constants of each component element described above are also changed according to actual conditions. It does not matter. Further, for example, the circuit configuration for generating the secondary side DC output voltage on the secondary side of the insulating converter transformer PIT may be appropriately changed.
Furthermore, the wide-range configuration according to the present invention can also be applied to a self-excited resonant converter.

It is a circuit diagram which shows the structural example of the switching power supply circuit based on the structure which the present applicant has proposed previously. It is a circuit diagram which shows the structural example of the power supply circuit as embodiment of this invention. It is a perspective view which shows the structural example of the insulation converter transformer PIT with which the power supply circuit of this Embodiment is equipped. It is a perspective view which shows the structural example of the insulation converter transformer PIT with which the power supply circuit of this Embodiment is equipped. It is a perspective view which shows the structural example of the insulation converter transformer PIT with which the power supply circuit of this Embodiment is equipped. It is a perspective view which shows the structural example of the insulation converter transformer PIT with which the power supply circuit of this Embodiment is equipped. It is a figure which shows the change characteristic of the control current Ic with respect to alternating current input voltage VAC and load fluctuation | variation, and the switching frequency fs. It is a circuit diagram which shows the other structural example of the control circuit of the secondary side in embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art example. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art example. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art example. It is a figure which shows the structural example of a saturable inductor. It is a characteristic view which shows the BH characteristic of a saturable inductor. It is a wave form diagram for demonstrating the constant voltage control operation | movement by a magnetic amplifier provided with a saturable inductor. FIG. 12 is a circuit diagram equivalently showing the magnetic amplifier constant voltage circuit shown in FIG. 11.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 4-1 and 4-2 control circuit, PRT-1, PRT-2 orthogonal control transformer, Di bridge rectifier circuit, Ci smoothing capacitor, Q1, Q2 switching element, Cp partial resonance Capacitor, PIT isolated converter transformer, N1 primary winding, C1 primary side series resonant capacitor, N2 secondary winding, Do1, Do2, Do3, Do4, Do5 (secondary side) rectifier diode, Co, Co1, Co2 (secondary) Side) Smoothing capacitor, Q11, Q21 shunt regulator, Q12, Q14, Q22 transistor

Claims (2)

Rectified and smoothed voltage generating means for generating a rectified and smoothed voltage at a level corresponding to the same size as the input commercial AC power supply;
A switching means that performs the switching operation by inputting the rectified and smoothed voltage as a direct-current input voltage, and a switching means that is formed by half-bridge coupling two switching elements;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied, a secondary winding in which an alternating voltage as a switching output obtained in the primary winding is excited, and a control winding; And an insulating converter transformer formed such that its leakage inductance is varied according to the level of the control current flowing through the control winding,
A primary side resonance capacitor provided so as to form a primary side resonance circuit in which the operation of the switching means is a resonance type by at least the leakage inductance component of the primary winding of the insulating converter transformer and its own capacitance; ,
DC output voltage generating means configured to generate a required number of secondary side DC output voltages of one or more by inputting an alternating voltage obtained in the secondary winding of the insulating converter transformer and performing a rectifying operation. When,
By controlling the switching drive means so that the switching frequency of the switching means can be varied according to the level of the commercial AC power supply, or by variably controlling the control current level to be passed through the control winding, First constant voltage control means configured to obtain an operation of making the secondary side DC output voltage constant;
In accordance with the level of one specific secondary side DC output voltage, the control current level to be supplied to the control winding is variably controlled, or the switching drive means is changed so that the switching frequency of the switching means is variable. A second constant voltage control means configured to obtain an operation of making the specific secondary side DC output voltage constant by controlling
A switching power supply circuit comprising: When a plurality of secondary side DC output voltages are generated, each secondary side DC output voltage is provided corresponding to each secondary side DC output voltage other than the constant voltage controlled by the second constant voltage control means. And the controlled winding of the control transformer as a saturable reactor wound with the controlled winding is inserted into the secondary side rectified current path for generating the secondary side DC output voltage to be controlled, Depending on the secondary DC output voltage level to be controlled, the control current level to be passed through the control winding is varied to vary the inductance of the controlled winding, thereby controlling the secondary DC output to be controlled. An inductance control type constant voltage control means configured to perform constant voltage control on the voltage;
The switching power supply circuit according to claim 1.

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