JP2005168079A - Switching power supply - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a wide range power supply circuit which is stabilized by a switching frequency control system, with as simple a circuitry as possible. <P>SOLUTION: The switching power supply circuit, comprising a resonance converter, is provided with a crossed control transformer PRT having a controlled winding NR included in the components of a primary series resonance circuit. Inductance LR of the controlled winding NR is controlled variably, depending on the level of a commercial AC power supply. Consequently, resonance frequency of the primary resonance circuit is varied depending on the level of a commercial AC power supply. As a result, a switching frequency control range required for stabilization is set depending on the rated level of the commercial AC power supply, for example. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit provided as a power source for various electronic devices.

As a switching power supply circuit, a circuit using a switching converter such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit to suppression of switching noise. Moreover, it has been found that there is a limit in improving the power conversion efficiency due to its operating characteristics.
Therefore, various types of switching power supply circuits using various resonant converters have been previously proposed by the present applicant. The resonant converter can easily obtain high power conversion efficiency, and low noise is realized by making the switching operation waveform sinusoidal. In addition, there is an advantage that it can be configured with a relatively small number of parts.

  As the switching power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC 85 V to 288 V is performed so as to correspond to an AC input voltage AC 100 V region such as Japan and the United States and an AC 200 V region such as Europe. A so-called wide-range power supply circuit that can be configured is known.

Here, as the above-described resonance type converter, one configured to be stabilized by controlling the switching frequency of a switching element forming the converter (switching frequency control method) is known.
As such a resonant converter based on the switching frequency control system, for example, in a configuration in which the switching element is switched and driven by a general-purpose oscillation / drive circuit IC, for example, the variable range of the switching frequency fs is maximum, fs = 50 KHz to 250 KHz. It is about. In the case of such a variable range, for example, under a load condition where the fluctuation range of the load power Po is a relatively large fluctuation range from Po = 0 W to about 90 W, and further to about 150 W, a wide range of AC85V to 288V is used. Stabilization corresponding to the AC input voltage range is almost impossible.

  In order to solve the above problem, the present applicant has previously proposed a switching power supply circuit having the following configuration. That is, as a switching power supply circuit for a wide range including a resonant converter, the switching power supply circuit can be stabilized against load fluctuations in a relatively wide range while adopting a switching frequency control method. A configuration example of such a switching power supply circuit is shown in FIG.

  In the power supply circuit shown in FIG. 9, first, a common mode choke coil CMC and a common mode noise filter composed of two sets of across capacitors CL and CL are connected to the commercial AC power supply AC as shown in the figure. Has been.

In this case, the rectifier circuit system includes a bridge rectifier circuit Di connected to a commercial AC power supply AC (AC input voltage VAC) and two smoothing capacitors Ci1 to Ci2 connected in series as shown in the figure. Connected. The rectified and smoothed voltage Ei is obtained as a voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2.
Further, a relay switch S is inserted between the negative output terminal of the bridge rectifier circuit Di and the connection point of the smoothing capacitors Ci1 to Ci2. This relay switch S is turned on / off according to the driving state of the relay RL connected to the rectifier circuit switching module 5.

The rectifier circuit switching module 5 is provided to switch the operation of the rectifier circuit system between when the input of the commercial AC power supply AC is a 100V system and when it is a 200V system by driving the relay RL.
Therefore, the rectifier circuit switching module 5 is supplied with a detection voltage output from a detection circuit including a diode D11, a capacitor C11, and voltage dividing resistors R11 and R12. In this detection circuit, first, a commercial AC power supply AC is rectified and smoothed by a half-wave rectifier circuit including a diode D11 and a capacitor C11, so that a DC voltage at a level corresponding to the commercial AC power supply AC is obtained as a voltage across the capacitor C11. Have been to get. A voltage level obtained by dividing this DC voltage by a series connection circuit of voltage dividing resistors R11-R12 connected in parallel to the capacitor C11 is input to the rectifier circuit switching module 5 as a detection voltage. Like to do. This detection voltage is obtained by dividing a DC voltage corresponding to the level of the commercial AC power supply AC at a predetermined ratio by a voltage dividing value corresponding to the resistance value of the voltage dividing resistors R11 and R12. Indicates the level of the commercial AC power supply AC.

  Further, a relay RL is connected to the rectifier circuit switching module 5. The rectifier circuit switching module 5 can switch conduction / non-conduction of the relay RL. Then, the relay RL performs on / off control of the relay switch S according to its conduction state. Here, it is assumed that the relay switch S is turned on when the relay RL is in a conductive state, and the relay switch S is turned off when the relay RL is in a non-conductive state.

The switching operation of the rectifier circuit configured as described above is as follows.
In the rectifier circuit switching module 5, the detection voltage input from the voltage dividing point of the voltage dividing resistors R11-R12 is input and compared as described above. For confirmation, this detection voltage indicates the level of the commercial AC power supply AC.
Here, for example, when the AC input voltage VAC is equal to or higher than a predetermined level near 150 V (AC 200 V system), the detected voltage is equal to or higher than the reference voltage, and when the AC input voltage VAC is equal to or lower than a predetermined level near 150 V (AC 100 V system). The voltage dividing ratio is set so as to be equal to or lower than the reference voltage.
In the rectifier circuit switching module 5, the relay RL is turned on when the divided voltage level is equal to or lower than the reference voltage, and the relay RL is turned off when it is equal to or higher than the reference voltage.

Here, for example, assuming that the commercial AC power supply is an AC 200V system, in this case, since the detected voltage level is equal to or higher than the reference voltage, the rectifier circuit switching module 5 turns off the relay RL. In response to this, the relay switch S is also turned off (opened).
When the relay switch S is OFF, the AC input voltage VAC is rectified by the bridge rectifier circuit Di and the rectified current is charged to the series connection circuit of the smoothing capacitors Ci1 to Ci2 in each period in which the AC input voltage VAC is positive / negative. Is obtained. That is, a rectification operation by a full-wave rectifier circuit including a normal bridge rectifier circuit can be obtained. As a result, a rectified and smoothed voltage Ei corresponding to an equal multiple of the AC input voltage VAC is obtained as a voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2.

On the other hand, it is assumed that the rectified and smoothed voltage Ei having a level corresponding to the AC input voltage VAC = 150 V or less is generated in correspondence with the commercial AC power supply of the AC100V system.
In this case, the voltage division level becomes equal to or lower than the reference voltage, and the rectifier circuit switching module 5 turns on the relay RL, so that the relay switch S is controlled to be turned on (closed).
When the relay switch S is on, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci1 when the AC input voltage VAC is positive. On the other hand, when the AC input voltage VAC is negative, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci2.
As a result of the rectifying operation performed in this way, a level corresponding to the same multiple of the AC input voltage VAC is generated as the voltage across each of the smoothing capacitors Ci1 and Ci2. Therefore, a level corresponding to twice the AC input voltage VAC is obtained as the rectified and smoothed voltage Ei that is the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. That is, a so-called voltage doubler rectifier circuit is formed.

  Thus, in the circuit shown in FIG. 9, in the case of the commercial AC power supply AC100V system, the rectified and smoothed voltage Ei corresponding to twice the AC input voltage VAC is generated by the voltage doubler rectification operation, and the commercial AC power supply AC200V In the case of the system, a rectified and smoothed voltage Ei corresponding to an equal magnification of the AC input voltage VAC is generated by a normal full-wave rectification operation. That is, as a result, the same level of rectified and smoothed voltage Ei is obtained when the commercial power input is an AC 100V system and an AC 200V system. This rectified and smoothed voltage Ei is input as a DC input voltage to the subsequent current resonance type converter.

  As shown in the figure, the current resonance type converter that switches by inputting the DC input voltage is connected by connecting two switching elements Q1 (high side) and Q2 (low side) by MOS-FET by half bridge coupling. Yes. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2, respectively, in the illustrated direction.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, in order to switch the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 using a general-purpose IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit. Then, a drive signal (gate voltage) having a required frequency is applied to each gate of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

  Further, this oscillation / drive circuit 2 is a half-wave rectifier circuit comprising a rectifier diode D3 and a capacitor C3 with respect to a tertiary winding N3 formed by providing a tap output to the primary winding N1 of the insulating converter transformer PIT. The low-voltage DC voltage obtained by the above is input as an operating power source. Moreover, at the time of starting, it starts by inputting the rectification smoothing voltage Ei via the starting resistance Rs.

The insulating converter transformer PIT transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. The winding start end of the primary winding N1 of the insulation transformer PIT is connected to the connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the series connection of the primary side parallel resonant capacitor C1. By being connected, a switching output is transmitted.
Further, the winding end of the primary winding N1 is connected to the primary side ground.
Here, depending on the capacitance of the series resonance capacitor C1 and the leakage inductance L1 of the insulating converter transformer PIT including the primary winding N1, a primary side series resonance circuit for making the operation of the primary side switching converter a current resonance type is formed. To do. That is, the basic configuration of the switching converter in the power supply circuit shown in FIG. 9 is a current resonance type converter using a half-bridge coupling method.

According to the above description, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the part by the partial voltage resonance circuit (Cp // L1) described above. A voltage resonance operation is obtained.
That is, the power supply circuit shown in this figure adopts a form in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit. In this specification, such a switching converter is referred to as a composite resonance type converter.

Although illustration explanation here is omitted, the structure of the insulating converter transformer PIT includes, for example, an EE type core in which an E type core made of a ferrite material is combined. Then, after dividing the winding part on the primary side and the secondary side, the set of the primary winding N1 (tertiary winding N3) and the secondary windings N2 and N2A described below are made up of the EE type core. Winding around the central magnetic leg.
A gap of about 1.0 mm to 1.5 mm is formed with respect to the central magnetic leg of the EE type core. As a result, a loosely coupled state with a coupling coefficient of about 0.7 to 0.8 is obtained.

In this case, a secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT. An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding.
The secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then, as shown in the figure, a double wave rectification comprising rectifier diodes DO1 and DO2 and a smoothing capacitor CO. The circuit is connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown) and is also branched and inputted as a detection voltage for the control circuit 1 described below.

The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage EO to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. Thus, the level of the secondary side DC output voltage is stabilized by varying the switching frequency of the switching elements Q1, Q2.
Here, in the power supply circuit shown in FIG. 9, switching between a voltage doubler rectification operation and a full-wave rectification operation is performed when the commercial power input is an AC100V system and an AC200V system, respectively. As a result, a DC input voltage (Ei) of the same level can be obtained regardless of whether the commercial power input is an AC 100V system or an AC 200V system, and the level of the secondary side DC output voltage Eo can be set accordingly. Similarly, the AC100V system and the AC200V system are equivalent.
For this reason, for example, even under the condition of a relatively large load fluctuation range where the load power Po = 0 W to 150 W, the AC 100 V system and the AC 200 V are controlled by the switching frequency control within the maximum variable range of 50 KHz to 250 KHz described above. In both cases, the secondary side DC output voltage can be effectively stabilized. In this way, the power supply circuit shown in FIG. 9 is compatible with a wide range.

A configuration in which switching is performed between the voltage doubler rectifier circuit and the full-wave rectifier circuit in accordance with the levels of the AC 100 V system and AC 200 V system commercial AC power as described above is described in, for example, the above-mentioned document.
JP-A-7-281770

As can be understood from the above description, the power supply circuit shown in FIG. 9 is configured to switch between full-wave rectification operation and voltage doubler rectification operation by an electromagnetic relay. For this configuration, two smoothing capacitors Ci1 and Ci2 are required as smoothing capacitors for obtaining the rectified and smoothed voltage Ei (DC input voltage).
That is, at least the number of smoothing capacitors is increased and a required number of electromagnetic relays are added for switching the rectifying operation. For this reason, the number of parts is increased and the cost is increased, and the mounting area of the power supply circuit board is increased and the size is increased. In particular, since these smoothing capacitors and electromagnetic relays are large among the components forming the power supply circuit, the substrate size becomes considerably large.

Further, as a circuit for switching between the full-wave rectification operation and the voltage doubler rectification operation, for example, in the basic configuration simply shown in FIG. 9, the following malfunction may occur. In other words, when AC200V commercial AC power is being input, if an instantaneous power failure occurs or the AC input voltage drops below the rated value, the AC100V level becomes lower than that corresponding to the AC200 system. An operation of switching to a voltage doubler rectifier circuit occurs as a system.
When such a malfunction occurs, voltage doubler rectification is performed on an AC input voltage of AC200V system level, so that there is a possibility that the switching elements Q1, Q2, etc. may be destroyed due to overvoltage.

Therefore, as an actual circuit, in order to prevent the above-described malfunction, not only the DC input voltage of the main switching converter but also the DC input voltage of the converter circuit on the standby power supply side is detected. It is made to take. For example, in the case of FIG. 9, the power supply circuit shown in this figure is the main switching converter. Actually, the rectifier circuit switching module 5 also receives a DC input voltage obtained by a converter of a standby power supply not shown in FIG. 9 as a detection voltage.
Further, as the standby power supply side converter circuit is detected as described above, a comparator IC is mounted as the rectifier circuit switching module 5. As a result, the number of parts increases and the above-described cost increase and circuit board size increase are further promoted.
When the rectifier circuit switching module 5 adopts such a configuration, for example, in order to improve parts management and work efficiency during manufacturing, the rectifier circuit switching module 5 is unitized by assembling a circuit system for switching the rectifying operation as a module. The However, in the case of unitization in this way, more parts such as adding pin terminals are required, which increases the cost.

  In addition, the detection of the DC input voltage of the converter on the standby power supply side for the purpose of preventing malfunctions means that a wide range compatible power supply circuit having a circuit for switching the rectifying operation includes a standby power supply in addition to the main power supply. This means that it cannot actually be used unless it is an electronic device. In other words, the types of electronic devices that can be equipped with a power supply are limited to those equipped with a standby power supply, and there is a problem that the use range is narrowed accordingly.

In view of the above-described problems, the present invention is configured as a switching power supply circuit as follows.
That is, a rectified and smoothed voltage generating means for generating a rectified and smoothed voltage at a level corresponding to the same size as the input commercial AC power supply is provided.
In addition, a switching means formed by including a switching element that performs a switching operation by inputting a rectified and smoothed voltage as a DC input voltage, and a switching drive means that switches the switching element.
Further, at least a primary winding supplied with a switching output obtained by the switching operation of the switching means and a secondary winding excited with an alternating voltage obtained as a switching output obtained in the primary winding are wound. An insulated converter transformer is provided. In addition, a primary side resonance capacitor provided to form a primary side resonance circuit that makes the operation of the switching means a resonance type by at least the leakage inductance component of the primary side winding of the insulating converter transformer and its own capacitance. Prepare.
In addition, a controlled winding connected so that its own inductance becomes an inductance component forming a primary side resonance circuit and a control winding are wound, and depending on the level of control current flowing in the control winding A control transformer formed such that the inductance of the controlled winding is variable, and a current control means provided so as to flow a control current of a variable level according to the level of the commercial AC power source to the control winding. .
DC output voltage generating means configured to generate a secondary side DC output voltage by inputting an alternating voltage obtained in the secondary winding of the insulating converter transformer and performing a rectification operation; and a secondary side Constant voltage control means configured to perform constant voltage control on the secondary side DC output voltage by controlling the switching drive means in accordance with the level of the DC output voltage to vary the switching frequency of the switching means, and Prepare and configure.

According to the above configuration, the inductance of the controlled winding is varied according to the level of the commercial AC power supply. The inductance of the controlled winding is an inductance component for forming the primary side resonance circuit. is there. Therefore, the resonance frequency of the primary side resonance circuit is varied according to the level of the commercial AC power supply.
In this way, if the resonance frequency of the primary side resonance circuit is made variable according to the level of the commercial AC power supply, the control range of the switching frequency required for stabilization can be set according to the level of the commercial AC power supply. It becomes possible to set.

Thus, according to the present invention, for example, it is possible to set the control range of the switching frequency required for stabilization in conformity with the rated level of the commercial AC power supply. Accordingly, for example, regardless of the rated level of the commercial AC power supply, the control range of the switching frequency required for stabilization can be set within the variable range of the switching frequency in the switching power supply circuit, for example. That is, a wide range compatible switching power supply circuit can be obtained that can be properly stabilized regardless of the level change of the commercial AC power supply.
This eliminates the need for a rectifier circuit that generates a DC input voltage (rectified and smoothed voltage) to switch between a voltage doubler rectification operation and a full-wave rectification operation according to the commercial AC power supply level. For this reason, the number of smoothing capacitors forming the rectifier circuit for generating the DC input voltage is reduced from two to one, for example.
In addition, the minimum number of switching elements may be two, for example, for forming a half-bridge connection. In this respect as well, the number of parts can be reduced. By reducing the number of components in this way, the circuit board can be reduced in size and weight and cost can be reduced. Further, since the number of switching elements is the minimum necessary, the switching noise can be suppressed to the minimum level.

FIG. 1 shows a configuration example of a power supply circuit based on the power supply circuit previously proposed by the present applicant. The power supply circuit shown in this figure has a configuration that is compatible with a wide range and includes a current resonance type converter that employs a switching frequency control system.
The power supply circuit shown in FIG. 1 is provided with a full-bridge coupled current resonance type converter having four switching elements on the primary side. As described later, the switching operation is switched between the full-bridge coupling method and the half-bridge coupling method when the commercial AC power supply is rated for the AC100V system and the AC200V system. It is compatible with a wide range.
Further, a separate excitation type is adopted as the switching drive method. A partial voltage resonance circuit that performs voltage resonance only when the switching element of the current resonance type converter is turned on / off is combined.

In the circuit shown in FIG. 1, a commercial AC power supply AC is first connected to a common mode noise filter consisting of a pair of common mode choke coils CMC and two sets of across capacitors CL and CL as shown. ing.
The commercial AC power supply AC is provided with a full-wave rectifier circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci. In this case, a rectified and smoothed voltage Ei having a level corresponding to only the equal magnification of the AC input voltage VAC is obtained at both ends of the smoothing capacitor Ci. This rectified and smoothed voltage Ei is input as a DC input voltage to the subsequent current resonance converter.
Thus, it can be seen that the power supply circuit shown in FIG. 1 is not provided with a circuit system for switching the rectifying operation, for example, as shown in FIG.

  The current resonance type converter shown in this figure includes four switching elements Q1, Q2, Q3, and Q4 corresponding to the full-bridge coupling method. In this case, corresponding to the separately excited type, a MOS-FET which is a voltage drive type is selected for these switching elements Q1 to Q4.

The drain of the switching element Q1 is connected to the line of the rectified and smoothed voltage Ei (DC input voltage). The source of switching element Q1 is connected to the drain of switching element Q2. The source of the switching element Q2 is connected to the primary side ground.
That is, the switching Q1 and Q2 are connected in series so as to be half-bridged so that the switching element Q1 is on the high side and the switching element Q2 is on the low side, thereby forming a pair of half-bridge circuits. doing.

Similarly, the drain of the switching element Q3 is connected to the line of the rectified and smoothed voltage Ei (DC input voltage), and the source is connected to the drain of the switching element Q4. The source of the switching element Q4 is connected to the primary side ground.
That is, the switching elements Q3 and Q4 are connected by half-bridge coupling so that the switching element Q3 is on the high side and the switching element Q4 is on the low side, thereby forming another set of half-bridge circuits.
According to such a connection mode, two sets of half-bridge circuits including a set of switching elements [Q1, Q2] and a set of switching elements [Q3, Q4] are connected to the DC input voltage (Ei) line and the primary. It is inserted in parallel between the side grounds. As a result, a switching circuit system as a full bridge coupling method is formed.

A clamp diode DD1 is connected in parallel between the drain and source of the switching element Q1. The anode and cathode of the clamp diode DD1 are connected to the source and drain of the switching element Q1, respectively. The clamp diode DD1 forms a pair of switching circuits together with the switching element Q1, and forms a path through which a reverse current flows when the switching element Q1 is turned on.
In the same connection manner, clamp diodes DD2, DD3, and DD4 are connected in parallel to switching elements Q2, Q3, and Q4, respectively.

In addition, partial resonance capacitors Cp1 and Cp2 are connected in parallel between the drain and source of the low-side switching elements Q2 and Q4 in each half bridge circuit.
A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitances of the partial resonance capacitors Cp1 and Cp2 and the leakage inductance component L1 of the primary winding N1 of the insulating converter transformer PIT described later.
By forming the partial voltage resonance circuit in this way, a partial voltage resonance operation in which voltage resonance occurs only in a short period of time when the switching elements Q1 to Q4 are turned on / off is obtained.
The configuration of the switching drive circuit system for these switching elements Q1 to Q4 will be described later.

The insulating converter transformer PIT transmits the switching outputs of the switching elements Q1 to Q4 to the secondary side.
Although the illustration of the structure of the insulating converter transformer PIT is omitted here, the primary winding N1 and the secondary winding N2 are formed corresponding to the primary side and the secondary side, for example, for the EE type core. Each of the divided areas is wound around. Further, in the insulating converter transformer PIT in this case, as shown in the figure, the tertiary winding N3 is also wound on the primary side.

  One end of the primary winding N1 of the insulating converter transformer PIT is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the series resonant capacitor C1. The other end of the primary winding N1 is connected to a connection point (switching output point) between the source of the switching element Q3 and the drain of the switching element Q4.

A primary side series resonance circuit is formed by the capacitance of the series resonance capacitor C1 and the leakage inductance component (L1) of the insulating converter transformer PIT including the inductance component L1 of the primary winding N1.
In the full-bridge coupling method, as described later, a switching operation is performed at the timing when the pair of switching elements [Q1, Q4] and the pair of switching elements [Q2, Q3] are alternately turned on / off. Since the primary side series resonant circuit comprising the primary winding N1 and the series resonant capacitor C1 is connected to the switching output point, the switching outputs of the switching elements Q1 to Q4 are transmitted to the primary side series resonant circuit. Will be. Then, the primary side series resonance circuit performs a resonance operation in accordance with the switching output, whereby an operation as a current resonance type is obtained. In the primary winding N1, a primary winding current I1 close to the resonance waveform is obtained according to the operation as the current resonance type.

  In this way, the switching converter of the power supply circuit shown in FIG. 1 has a combined operation of the current resonance type corresponding to the full bridge coupling method and the partial voltage resonance operation described above. . That is, a configuration as a composite resonance type converter is adopted.

An alternating voltage excited in accordance with the switching output transmitted to the primary winding N1 is generated in the secondary winding N2 of the insulating converter transformer PIT.
In this case, a center tap is provided for the secondary winding N2. This center tap is connected to the secondary side ground. In addition, as shown in the figure, a two-wave rectifier circuit is formed by connecting two rectifier diodes D01 and D02 and a smoothing capacitor Co to the secondary winding N2. The double-wave rectifier circuit receives the alternating voltage excited by the secondary winding N2 and performs a rectification operation, whereby a secondary side DC output voltage EO is obtained as a voltage across the smoothing capacitor CO.
The secondary side DC output voltage EO is supplied to a load (not shown). Further, this secondary side DC output voltage EO is also branched and inputted as a detection voltage for the control circuit 1 as shown.

  A half-wave rectifier circuit composed of a diode D4 and a capacitor C4 is connected to the tertiary winding N3 wound on the primary side of the insulating converter transformer PIT as shown in the figure. The low-voltage DC voltage obtained by this half-wave rectifier circuit is supplied as an operating power source to each of an oscillation / drive circuit 2 and a drive circuit 3 to be described later.

The control circuit 1 obtains, as a control output, a current or voltage whose level is varied according to the level of the secondary side DC output voltage EO, for example. This control output is output to the oscillation / drive circuit 2.
The oscillation / drive circuit 2 generates an oscillation signal as will be described later, and uses this oscillation signal to output a high-side drive signal and a low-side drive signal for driving the switching element by separate excitation. Then, with this drive signal, the switching elements Q1 to Q4 are switched and driven at a required switching timing.
The oscillation / drive circuit 2 operates so as to vary the frequency of the oscillation signal generated internally in accordance with the control output level input from the control circuit 1. As a result, the frequency of the drive signal is varied according to the control output level. That is, the oscillation / drive circuit 2 operates to variably control the switching frequency of the switching elements Q1 to Q4 according to the control output level input to the control terminal Vc.
By changing the switching frequency, the resonance impedance in the series resonance circuit changes. As the resonance impedance changes in this way, the amount of current supplied to the primary winding N1 of the primary side series resonance circuit changes, and the power transmitted to the secondary side also changes. As a result, the secondary output voltage changes, and constant voltage control is achieved.

  Next, a switching drive circuit system for driving the switching elements Q1 to Q4 in the power supply circuit shown in FIG. 1 will be described. The power supply circuit shown in FIG. 1 includes a single oscillation / drive circuit 2 and a drive circuit 3 as a switching drive circuit system.

The oscillation / drive circuit 2 includes an oscillation circuit, a control circuit, a protection circuit, and the like for driving a current resonance type converter by a separate excitation type, and an analog IC (Integrated Circuit) having a bipolar transistor therein. ).
As described above, the oscillation / drive circuit 2 operates by inputting a DC voltage obtained by a half-wave rectifier circuit including the tertiary winding N3, the diode D3, and the capacitor C3. Further, at the time of power activation, the rectified and smoothed voltage Ei input via the activation resistor Rs can be input as the activation power source so that the operation can be started.

The oscillation / drive circuit 2 outputs two drive signals as terminals for outputting a drive signal (gate voltage) to the switching element. That is, the drive signal SD1 for switching the high-side switching element and the drive signal SD2 for switching the low-side switching element are output.
In this case, the drive signal SD1 is applied to the gate of the high-side switching element Q1. The drive signal SD2 is applied to the gate of the low-side switching element Q2. The drive signal SD1 is output after being level-shifted with respect to the drive signal SD2 so that the switching element Q1 on the high side can be driven appropriately. The drive signals SD1 and SD2 branch and are also input to the drive circuit 3.

In the drive circuit 3, the input drive signal SD1 is shifted to a voltage level for driving the low-side switching element and applied to the gate of the switching element Q4 as the drive signal SD4. Further, the drive signal SD2 is shifted to a voltage level for driving the high-side switching element, and applied to the gate of the switching element Q4 through the switch Sw as the drive signal SD3.
The drive circuit 3 is supplied with a detection voltage obtained by dividing the rectified and smoothed voltage Ei (DC input voltage) by a voltage dividing resistor R1-R2 at a predetermined voltage dividing ratio. In the drive circuit 3, as will be described later, the switch Sw is turned on / off in accordance with the level of the detected voltage. The switch Sw may be configured to use, for example, a low-power bipolar transistor as a switch element.

  In the power supply circuit shown in FIG. 1, switching is performed between a switching operation by a full bridge coupling method and a switching operation by a half bridge coupling method in accordance with the level of the AC input voltage VAC (commercial AC power supply AC) as described later. Done. However, here, as a basic operation, an operation when all of the switching elements Q1 to Q4 are driven to be switched will be described corresponding to the case of the full bridge coupling method.

As described above, the oscillation / drive circuit 2 outputs the high-side drive signal SD1 for driving the high-side switching element and the low-side drive signal SD2 for driving the low-side switching element. .
Here, the drive signal SD1 and the drive signal SD2 have a pulse waveform that rises at a predetermined level in response to turning on the switching element and falls at a predetermined level in response to turning off. These pulse waveforms are generated so as to have a phase difference of 180 ° from each other.
As a result, first, the switching elements Q1 and Q2 having a high-side and low-side relationship are driven to be switched on and off alternately.

  On the other hand, in the drive circuit 3, the input drive signal SD1 is level-shifted and applied to the low-side switching element Q4 as the drive signal SD4. Similarly, the input drive signal SD2 is level-shifted and then applied to the high-side switching element Q3 as the drive signal SD3. That is, as the waveform timing, the set of drive signals SD1 and SD4 is the same, and the set of drive signals SD2 and SD3 is the same. In addition, the drive signals SD1 and SD4 and the drive signals SD2 and SD3 are set to have a phase difference of 180 ° from each other.

  Therefore, the switching elements Q1 and Q4 are turned on / off at the same timing, and similarly, the switching elements Q2 and Q3 are also turned on / off at the same timing. In addition, the group of switching elements [Q1, Q4] and the group of switching elements [Q2, Q3] are driven to be switched on and off alternately.

As a switching operation at this time, when the set of the switching elements [Q1, Q4] is on, the output is the drain-source of the switching element Q1 → the series resonant capacitor C1 → the primary winding N1 → the drain of the switching element Q4. -Current flows through the source → primary side ground.
When the pair of switching elements [Q2, Q3] is on, the output is the drain-source of the switching element Q3 → the primary winding N1 → the series resonant capacitor C1 → the drain-source of the switching element Q2 → the primary side. Current flows through the ground path. As this operation is repeated, a resonance operation is obtained in the primary side series resonance circuit (C1-N1), and a drive current close to the resonance current waveform is applied to the primary winding N1 of the insulating converter transformer. Will be supplied.
An operation in which the switching elements Q1, Q2, Q3, and Q4 perform switching in this way is a switching operation as a full bridge coupling method.

Further, at the timing when the pair of switching elements [Q1, Q4] is turned off / turned on as described above, the parallel resonant capacitor Cp2 connected to the switching element Q4 has its own capacitance and the leakage inductance of the primary winding N1. A parallel resonance circuit is formed by the component L1, and a voltage resonance operation is performed. That is, a partial voltage resonance operation in which voltage resonance occurs only when the pair of switching elements [Q1, Q4] is turned off / turned on can be obtained.
Similarly, at the timing when the pair of switching elements [Q2, Q3] is turned off / turned on, the parallel resonant circuit is generated by the capacitance of the parallel resonant capacitor Cp1 connected to the switching element Q2 and the leakage inductance component L1 of the primary winding N1. Is formed. A partial voltage resonance operation can be obtained at the time of turn-off / turn-on of the set of the switching elements [Q2, Q3].

  In this way, in the circuit of FIG. 1, a full-bridge coupled current resonance type converter in which each set (switching circuit) of switching elements [Q1, Q4] [Q2, Q3] is alternately turned on / off, and a partial voltage A converter in which resonance circuits (Cp1, Cp2, N1) are combined is formed.

  The power supply circuit shown in FIG. 1 performs the switching operation (full bridge operation) by the full bridge coupling method in the AC 100V system as described below under the above-described configuration of the separately excited full bridge coupling method, and the AC 200V system. Then, the structure which switches switching operation | movement is taken so that it may become switching operation (half-bridge operation | movement) by a half-bridge coupling system.

  In the power supply circuit shown in FIG. 1, a voltage dividing circuit in which voltage dividing resistors R1-R2 are connected in series is connected in parallel to the smoothing capacitor Ci. The voltage obtained at the connection point (voltage dividing point) of the voltage dividing resistors R1-R2 is input to the drive circuit 3 as the detection voltage as described above. As can be seen from the fact that the detected voltage is connected in parallel to the smoothing capacitor Ci, the detected voltage is obtained by dividing the rectified and smoothed voltage Ei (DC input voltage Ei). Here, since the level of the rectified and smoothed voltage Ei corresponds to the level of the commercial AC power supply AC, the detected voltage also corresponds to the level of the commercial AC power supply AC.

In the drive circuit 3, for example, when it is detected that the AC input voltage VAC is at a level corresponding to the AC 100V system corresponding to the vicinity of 150V or less, the detected voltage is between the drive signal SD3 and the gate of the switching element Q3. The inserted switch Sw is controlled to be turned on. At this time, a state in which the drive signal SD3 is applied to the gate of the switching element Q3 is obtained.
As a result, for example, as described above with reference to FIG. 8, a state in which the four switching elements Q1 to Q4 are driven to be switched is obtained. That is, a full bridge operation is obtained.

On the other hand, when the drive circuit 3 detects that the AC input voltage VAC is at a level corresponding to the AC 200 V system corresponding to the vicinity of 150 V or higher, the drive circuit 3 turns off the switch Sw. To control.
As a result, the drive signal SD3 is not applied to the gate of the switching element Q3, so that the switching element Q3 is brought into a state where the switching operation is stopped, and the remaining switching elements Q1, Q2, Q4 are in a state where the switching operation is performed.
At this time, the set of the switching elements [Q1, Q4] is turned on / off at the same timing, whereas only the switching element Q2 is switched at the timing that is alternate with respect to the set of the switching elements [Q1, Q4]. It will be turned on / off.

  In this way, the fact that the three switching elements Q1, Q2, Q4 perform the switching operation means that the switching elements [Q2, Q4] are turned on / off at the timing when the pairs of the switching elements [Q2, Q4] alternate. It can be said that the half-bridge operation is obtained.

As will be understood from the above description, in the power supply circuit shown in FIG. 1, first, a full-bridge coupling type current resonance type converter including four switching elements Q1 to Q4 is formed. In addition, the switching operation is switched so that a full-bridge operation is performed in the AC100V system and a half-bridge operation is performed in the AC200V system.
In this way, the current levels obtained by the switching outputs of the switching elements Q1 to Q4 are almost equal between the AC100V system and the AC200V system. Thereby, for example, even in the maximum variable range of the switching frequency of about 50 KHz to 250 KHz, it is possible to effectively stabilize the secondary side DC output voltage in both the AC100V system and the AC200V system. . In other words, wide range is supported by switching between full bridge operation and half bridge operation.

Then, by achieving compatibility with a wide range in this way, a normal full-wave rectifier circuit can be used as a rectifier circuit system that generates a DC input voltage (rectified and smoothed voltage Ei) from a commercial AC power supply AC. That is, it is not necessary to adopt a configuration for switching the rectification operation as in the circuit shown in FIG.
Therefore, in the power supply circuit shown in FIG. 1, only one smoothing capacitor for DC input voltage is required. Also, no electromagnetic relay is required.

Further, even if the commercial AC power supply of nominal AC 220V or 240V drops below around 150V due to, for example, an instantaneous power failure, the switching operation only changes from the half-bridge operation to the full-bridge operation. That is, since the DC input voltage level does not increase due to switching to the voltage doubler rectification operation as in the circuit shown in FIG. 9, the smoothing capacitor Ci and the switching element do not exceed the withstand voltage.
Further, by omitting the circuit system for switching the rectifying operation provided with the comparator IC and the like, it is not necessary to detect the DC input voltage on the standby power supply side in order to cope with the wide range. Therefore, the power supply circuit shown in FIG. 1 can be used for an electronic device that does not include a standby power supply.

Next, FIG. 2 shows a configuration of a switching power supply circuit as an embodiment of the present invention.
In the power supply circuit shown in this figure, first, as in the case of FIG. 1, a common mode noise filter formed by a common mode choke coil CMC and filter capacitors CL and CL is connected to a commercial AC power supply AC. The

  Further, as a rectifier circuit for rectifying and smoothing the commercial AC power supply AC to generate a DC input voltage (rectified and smoothed voltage Ei), as in the circuit of FIG. 1, all of the bridge rectifier circuit Di and the smoothing capacitor Ci are used. A wave rectifier circuit is provided, and a DC input voltage (rectified and smoothed voltage Ei) having a level corresponding to the same value as the effective value level of the commercial AC power supply AC is obtained as the voltage across the smoothing capacitor Ci.

  As a switching converter that performs switching by inputting the DC input voltage, a current resonance type converter using a half-bridge coupling system including two switching elements Q1 and Q2 is provided.

  As the switching elements Q1 and Q2, the drain of the switching element Q1 is connected to the positive terminal side of the smoothing capacitor Ci, and the source of the switching element Q1 is connected to the drain of the switching element Q2. The source of the switching element Q2 is connected to the negative terminal (primary side ground) of the smoothing capacitor Ci. That is, as half-bridge coupling, the switching element Q1 is connected in series so that the switching element Q1 is on the high side and the switching element Q2 is on the low side, and then connected in parallel to the smoothing capacitor Ci.

Damper diodes DD1 and DD2 are connected in parallel to each other between the drains and sources of the switching elements Q1 and Q2, respectively, so that the drain side is a cathode and the anode side is a drain.
Also in this circuit, a partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  Also in this case, in order to perform switching driving of the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 configured to include a general-purpose driving IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit for driving a switching element. Depending on the oscillation circuit and the drive circuit, a drive signal (gate voltage) having a required frequency is applied to each gate of the switching elements Q1 and Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

The oscillation / drive circuit 2 operates by inputting a low-voltage DC voltage E10 generated on the primary side.
The DC voltage E10 is a DC voltage obtained by a half-wave rectifier circuit comprising a tertiary winding N3 wound around the primary side of the insulating converter transformer PIT, a diode D3, and a capacitor C3, and comprises a resistor R4 and a Zener diode ZD. It is obtained by being stabilized at a predetermined level by a stabilization circuit. Although not shown in the figure, it is assumed that a starting resistor is provided, and when starting up the power supply, the rectified and smoothed voltage Ei input through this starting resistor is input as a starting power source to What is necessary is just to comprise so that the drive circuit 2 can start operation | movement.

The insulating converter transformer PIT transmits the switching output of the switching elements Q1 and Q2 to the secondary side, and in this case, each set of primary winding N1 and secondary winding N2 is wound.
In the case of the present embodiment, one end of the primary winding N1 is connected to the source of the switching element Q1 and the switching element Q2 via a series connection of a controlled winding NR of the orthogonal control transformer PRT described later and the primary side series resonance capacitor C1. Connected to the drain connection point (switching output point). The other end of the primary winding N1 is connected to the primary side ground (source of the switching element Q2).

As understood from the above description, depending on the series connection circuit of the series resonance capacitor C1 and the primary winding N1 connected to the switching output points of the switching elements Q1 and Q2, the capacitance of the series resonance capacitor C1 may be used. And a primary side series resonance circuit composed of the leakage inductance L1 of the insulating converter transformer PIT including the primary winding N1. This primary side series resonance circuit performs a resonance operation with the switching outputs of the switching elements Q1 and Q2, thereby making the operation of the primary side switching converter a current resonance type.
In addition, in the power supply circuit of the embodiment shown in FIG. 2, the controlled winding NR of the orthogonal control transformer PRT is further inserted in series between the series connection of the series resonant capacitor C1 and the primary winding N1.
Therefore, in this embodiment, the primary side series resonance circuit includes the inductance LR of the controlled winding NR in addition to the capacitance of the series resonance capacitor C1 and the leakage inductance L1 on the insulating converter transformer PIT side. Will be formed. . In other words, the inductance component for forming the primary side series resonance circuit is represented by L1 + LR.

For example, the structure of the insulating converter transformer PIT is as follows. That is, an EE type core that is a combination of E type cores made of ferrite material is provided. And after dividing | segmenting a winding site | part by the primary side and a secondary side, the primary side coil | winding and the secondary side coil | winding are wound with respect to the center magnetic leg of an EE type | mold core.
A gap having a predetermined length is formed with respect to the central magnetic leg of the EE type core. Accordingly, for example, as described later, a predetermined coupling coefficient that is loosely coupled is given.

In this case, a secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT. An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding.
The secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then, as shown in the figure, a double wave rectification comprising rectifier diodes DO1 and DO2 and a smoothing capacitor CO. The circuit is connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown). The detection voltage for the control circuit 1 is also branched and input.

  The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage EO to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. That is, the level of the secondary side DC output voltage is stabilized by the switching frequency control method.

An orthogonal control transformer PRT and an amplifier circuit 3 are provided on the primary side of the power supply circuit as the embodiment shown in FIG.
Here, an example of the structure of the orthogonal control transformer PRT is shown in FIGS.
First, the orthogonal control transformer PRT shown in FIG. 3 includes two double U-shaped cores CR11 and CR12 having four magnetic legs. A solid core is formed by joining the ends of the magnetic legs of the double U-shaped cores CR11 and CR12.
When the three-dimensional core is formed in this way, there are four joint portions of the double U-shaped cores CR11 and CR12 corresponding to each of the four magnetic legs. In this case, gaps G and G having a predetermined length are formed for two adjacent joints in these four joints, and no gap is formed for the remaining two joints.
In the three-dimensional core thus formed, for example, a magnetic leg in which a gap G on the side of the double U-shaped core CR11 is formed and a magnetic leg in which a gap adjacent to the magnetic leg is not formed. The control winding Nc is wound with a predetermined number of turns (number of turns) so as to straddle.
Further, the other double U-shaped core CR12 side is covered with two adjacent magnetic legs so that the winding direction is perpendicular to the winding direction of the control winding NC. The control winding NR is wound with a predetermined number of turns. In this way, the two magnetic legs around which the controlled winding NR is wound have a relationship in which one has a gap G and the other has no gap.
With such a structure, the orthogonal control transformer PRT is configured as a saturable reactor that becomes saturated due to an increase in the current flowing through the control winding Nc.

Further, as another structure of the orthogonal control transformer PRT, as shown in FIG. 4, a three-dimensional core has one core of a double U-shaped core CR11 having four magnetic legs. The core may be formed by combining as a single U-shaped core CR21 having an arbitrary cross-section having a U shape instead of the double U-shaped core CR12.
Also in this case, the two gaps G and G are formed by the same positional relationship as the orthogonal control transformer PRT of FIG. The control winding Nc is wound around the two magnetic legs of the double U-shaped core CR11 in the same manner as in FIG. 3, but for the single U-shaped core CR21, In this way, the controlled winding NR is wound.

The controlled winding N R of the orthogonal control transformer PRT configured as a saturable reactor as described above is inserted in series between the primary winding N 1 and the series resonant capacitor C 1 as described above. Thus, the inductance LR of the controlled winding NR becomes an inductance component for forming a primary side series resonance circuit.
One end of the control winding Nc of the orthogonal control transformer PRT is connected to the primary side ground, and the other end is connected to the collector of the PNP type bipolar transistor Q10 in the amplifier circuit 3.

The amplifier circuit 3 includes the transistor Q10, a base resistor R1, a bias resistor R2, and an emitter resistor R3.
The collector of the transistor Q10 is connected to the primary side ground via the control winding Nc. The base resistor R1 is inserted between the base of the transistor Q10 and the line of the rectified and smoothed voltage Ei (the positive terminal of the smoothing capacitor Ci). The bias resistor R2 is inserted between the base of the transistor Q10 and the primary side ground.
The emitter resistor R3 is inserted between the emitter of the transistor Q10 and the line of the DC voltage E10.

  In the amplifier circuit 3 having such a configuration, a base current corresponding to the level of the rectified and smoothed voltage Ei is supplied to the base of the transistor Q10 as follows. In other words, as the level of the rectified and smoothed voltage Ei increases, the base potential of the transistor Q10 increases and the potential difference between the base and the emitter decreases, so that the base current is reduced and the collector current is controlled. The current Ic will be reduced. Conversely, as the level of the rectified smoothing voltage Ei increases, the base potential of the transistor Q10 decreases, the potential difference between the base and the emitter increases, and the base current increases, resulting in an increase in the control current. Ic (collector current) is reduced.

  Here, since the rectified and smoothed voltage Ei is a DC voltage obtained by rectifying and smoothing the commercial AC power supply AC (AC input voltage VAC) by full-wave rectification, the level of the rectified and smoothed voltage Ei is the level of the commercial AC power supply AC (AC This corresponds to the level of the input voltage VAC). Therefore, it can be said that the amplifier circuit 3 operates to vary the level of the control current Ic according to the level of the commercial AC power supply AC. That is, it operates so as to decrease the level of the control current Ic as the level of the commercial AC power supply AC increases.

In explaining the operation corresponding to the wide range of the power supply circuit as the embodiment configured as described above, first, regarding the inductance control for the controlled winding NR by the orthogonal control transformer PRT, FIG. The description will be given with reference.
FIG. 5 shows the inductance characteristics of the controlled winding NR, and the horizontal axis represents the level of the primary side series resonance current I1 flowing through the primary side series resonance circuit (C1-NR-N1). The primary side series resonance current I1 is obtained by combining the switching output currents of the switching elements Q1, Q2. The vertical axis represents the inductance LR of the control winding NR. In addition, as the inductance characteristics, the cases of Ic = 0 mA, Ic = 10 mA, Ic = 30 mA, Ic = 50 mA, Ic = 70 mA, and Ic = 90 mA are shown for the level of the control current Ic.

As shown in FIG. 5, since the orthogonal control transformer PRT is a saturable reactor, the inductance LR of the controlled winding NR is controlled under the condition that the amplitude level of the primary side series resonance current I1 is constant. It increases as the level of Ic decreases.
Under the condition that the control current Ic is constant, the inductance LR tends to increase as the absolute value of the amplitude level of the primary side series resonance current I1 decreases as a whole. Depending on the level of the control current Ic, since the amplitude level of the primary side series resonance current I1 decreases in the vicinity of 0, the characteristic may be indicated by a substantially M-shaped curve. In the case of FIG. 5, the characteristic becomes a quadratic curve when Ic = 0 when no control current Ic is passed and when the maximum control current Ic = 90 mA in FIG. In the case of the control current level (Ic = 10 mA, Ic = 30 mA, Ic = 50 mA, Ic = 70 mA), the curve is substantially M-shaped. This is because the gap G having a predetermined length is formed as shown in FIGS.

As described above with reference to FIG. 5, the inductance LR of the controlled winding NR of the orthogonal control transformer PRT increases as the level of the control current Ic flowing through the control winding Nc decreases. Is done. Then, according to the operation of the amplifier circuit 3 shown in FIG. 2, the control current Ic is controlled so as to be reduced as the level of the commercial AC power supply AC (AC input voltage VAC) increases.
Therefore, in the circuit shown in FIG. 2, the inductance LR of the controlled winding NR is variably controlled in accordance with the level of the commercial AC power supply AC (AC input voltage VAC). The inductance LR increases as the (AC input voltage VAC) increases, and the inductance LR decreases as the commercial AC power supply AC (AC input voltage VAC) decreases.

  Further, as described above, the inductance LR is an inductance component that forms a primary side series resonance circuit together with the leakage inductance L1 of the primary winding N1. Therefore, changing the inductance LR according to the level of the commercial AC power supply AC (AC input voltage VAC) as described above means that the primary side changes according to the level change of the commercial AC power supply AC (AC input voltage VAC). This means that the inductance value forming the series resonance circuit is variable. From the relationship between the commercial AC power supply AC and the inductance LR, the inductance forming the primary side series resonance circuit increases with an increase in the commercial AC power supply AC (AC input voltage VAC). The control is variably controlled so as to decrease in accordance with a decrease in AC (AC input voltage VAC).

FIG. 6 shows the secondary side DC output voltage Eo obtained when the configuration in which the inductance of the primary side series resonance circuit is varied according to the level of the commercial AC power supply AC (AC input voltage VAC) as described above. The constant voltage control characteristic of is illustratively shown by the relationship between the switching frequency and the level of the secondary side DC output voltage Eo.
Here, the secondary side DC output voltage Eo is supposed to be stabilized at 21.5V. As a switching frequency control method, the switching frequency is variably controlled in a frequency range higher than the resonance frequency fo of the primary side series resonance circuit (C1-L1), and a change in resonance impedance caused thereby is used. So-called upper side control is adopted. Further, in this figure, as a condition of the level of the commercial AC power supply, an AC input voltage VAC = 100V is input for the AC100V system, and an AC input voltage VAC = 220V is input for the AC200V system.

Here, the fact that the inductance value in the primary side series resonance circuit is varied as described above means that the resonance frequency fo of the primary side series resonance circuit changes. Then, as described above, the change in inductance of the primary side series resonance circuit in the present embodiment increases as the commercial AC power supply level increases and decreases as it decreases.
Therefore, when the resonance frequency fo1 in the AC 100V system and the resonance frequency fo2 in the AC 200V system are compared with each other in the primary side series resonance circuit, as shown in FIG. 6, the resonance frequency fo1 is more than the resonance frequency fo2. It will be higher.

  As a general matter, the series resonance circuit has the smallest resonance impedance at the resonance frequency fo. Thereby, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs, the level of the secondary side DC output voltage Eo becomes the highest when the switching frequency fs is the same as the resonance frequency fo, and is separated from the resonance frequency fo. It goes down as you go.

  That is, in FIG. 6, first, in the AC 100V system (VAC = 100V), as shown by the solid line, the secondary side DC output voltage Eo is obtained when the switching frequency fs is the resonance frequency fo1 of the primary side series resonance circuit. It shows a quadratic curve-like change in which the level decreases as the peak value moves away from the resonance frequency fo1. In addition, the level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs can be shifted so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  When the secondary side DC output voltage Eo is to be stabilized at 21.5 V by upper side control under the characteristics of the AC 100 V system shown by the solid line in FIG. 6 as described above, this is necessary. The variable range (necessary control range) of the switching frequency is the range indicated as Δfs1.

In one AC 200V system, as described above, the resonance frequency fo of the primary side series resonance circuit is set to a predetermined frequency indicated by fo2 lower than fo1 in the AC 100V system. The characteristic at this time is shown by a broken line in FIG. That is, in this case as well, the secondary side DC output voltage Eo is maximized when the switching frequency fs is the resonance frequency fo2, and as the switching frequency fs moves away from the resonance frequency fo2. It will decrease. Also in this case, if the switching frequency fs is fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.
The required control range when the secondary side DC output voltage Eo is to be stabilized at 21.5 V by upper side control under the characteristics of the AC 200 V system is the range indicated by Δfs2 in FIG. .

That is, in the present embodiment, the inductance value of the primary side series resonance circuit is variably controlled according to the level of the commercial AC power supply AC (AC input voltage VAC), so that the input of the commercial AC power supply is AC100V. As shown in FIG. 6, the required control ranges Δfs1 and Δfs2 are set in accordance with the AC100V system and the AC200V system, respectively, according to switching between the system and the AC200V system. ing.
In FIG. 6, only the characteristics when the AC input voltage VAC = 100V (AC100V system) and the AC input voltage VAC = 220V (AC200V system) are shown. Therefore, the necessary control ranges Δfs1 and Δfs2 are switched. Is shown as if it was changed and set. However, in the orthogonal control transformer PRT of the present embodiment, if the control current Ic is continuously varied, the inductance LR can be continuously variably controlled. Therefore, assuming that the AC input voltage VAC continuously changes between a predetermined level as the AC100V system and a predetermined level as the AC200V system, the resonance frequency of the primary side series resonance circuit and the necessary control corresponding thereto The range Δfs can also change continuously.

  For example, when the orthogonal type control transformer PRT and the amplifier circuit 3 are omitted from the power supply circuit shown in FIG. 1 and the inductance value of the primary side series resonance circuit is not variable, the primary side series is independent of the commercial AC power supply level. Since the inductance value of the resonance circuit is fixed, the resonance frequency fo of the primary side series resonance circuit is also fixed with respect to the commercial AC power supply level. In this case, a region having a switching frequency higher than the fixed resonance frequency fo is set as a necessary control range Δfs, and the range corresponding to the AC100V system to AC200 is very large corresponding to a predetermined load fluctuation range. It is necessary to stabilize the secondary side DC output voltage Eo having a fluctuation range. In this case, for example, under conditions where the load fluctuation range is large, the necessary control range Δfs must be very wide, and proper stabilization is achieved within the maximum variable range of the switching frequency determined by the specifications of the switching power supply circuit. Can be difficult.

  In contrast, in the present embodiment, as described above, for example, when the input level of the commercial AC power source changes between the AC 100V system and the AC 200V system, it is necessary to correspond to the input level of each commercial AC power source. The control ranges Δfs1 and Δfs2 can be set. This means that by selecting the number of turns of the primary winding N1, each of the necessary control ranges Δfs1 and Δfs2 can be set to fall within the maximum variable range of the switching frequency. As a result, it is guaranteed that the secondary side DC output voltage Eo is stabilized corresponding to a predetermined load fluctuation range in each of the AC100V system and the AC200V system. That is, a wide-range power supply circuit is realized.

As an actual example of the power supply circuit shown in FIG. 2, when the load power Po = 0W to 90W is to be supported, the insulating converter transformer PIT is an EER-28 ferrite core and a central magnetic leg is used. Is designed to form a 1.0 mm gap. The primary winding N1 has 35T (turns) and the secondary winding N2 has 10T. As the insulating converter transformer PIT configured as described above, a state of loose coupling represented by a coupling coefficient k = 0.85 is obtained.
Further, 0.027 μF is selected as the capacitance of the primary side series resonance capacitor C1.
In addition, the orthogonal control transformer PRT has a three-dimensional core size of 20 mm × 20 mm × 20 mm and a gap G of 0.8 mm in any structure shown in FIG. 3 and FIG. did. The controlled winding NR is 40T, and the control winding Nc is 1000T.

After selecting the components of the power supply circuit as described above, the experiment was performed by setting the conditions for the load fluctuation range from the load power Po = 0W to 90W within the fluctuation range of the AC input voltage VAC = 85-288V. As a result, the required control range Δfs was 60 KHz (lower limit value of Δfs2) to 195 KHz (upper limit value of Δfs1), and the variable range of the control current Ic was 60 mA to 10 mA.
For example, as a practical example of the oscillation / drive circuit 2, the maximum variable range of the switching frequency fs when a specific general-purpose driving IC is used is 50 KHz to 250 KHz, but the required control range Δfs (60 KHz to 195 KHz). ) Is well within the above-mentioned maximum variable range of 50 KHz to 250 KHz. Therefore, the secondary side DC output voltage Eo is appropriate for the commercial AC power supply input from AC100V system to AC200V system. It will be stabilized.
Here, for confirmation, the load fluctuation range of the load power Po = 0W to 90W described above is a load condition arbitrarily set for obtaining the experimental result, and therefore, in the present embodiment. As a power supply circuit based thereon, it is possible to appropriately stabilize in response to a load of a heavier load and a wider load fluctuation range.

By adopting such a configuration, the power supply circuit of the present embodiment is, for example, similar to the circuit shown in FIG. The rectifying circuit system that generates Ei) does not need to adopt a configuration for switching the rectifying operation, and can be a normal full-wave rectifying circuit. Therefore, one smoothing capacitor for generating the DC input voltage is sufficient. Further, since the rectifier circuit system is switched, the electromagnetic relay can be made unnecessary.
In addition, the power supply circuit shown in FIG. 1 requires a total of four switching elements, and accordingly, it is necessary to add a circuit configuration as the drive circuit 2, and the circuit configuration is relatively complicated. It can be said that there is.
On the other hand, in this embodiment, since the switching converter is fixed by the half-bridge coupling method, the switching element may be two stones, and the configuration of the drive circuit system is simpler. It can be. For this reason, as a whole, the board mounting area can be reduced as compared with the power supply circuit shown in FIG. 1, and a low-cost power supply circuit can be obtained.
In the circuit shown in FIG. 1, four stone switching elements perform switching operation in the full bridge operation, and three stone switching elements perform switching operation in the half bridge operation. On the other hand, in this embodiment, since the switching operation of the two stone switching elements is always performed, it can be said that the switching noise is reduced accordingly.

In the present embodiment, a wide range is supported by a configuration in which the inductance of the primary side series resonance circuit (inductance LR of the controlled winding NR) can be continuously changed by the orthogonal control transformer PRT. In other words, for example, it does not have a switching function by an electromagnetic relay.
For this reason, even if an abnormality such as an instantaneous power failure or a steep drop occurs in the commercial AC power supply AC, for example, the orthogonal control transformer PRT is adapted to follow the level change of the rectified smoothing voltage Ei according to the abnormality. Circuit protection is achieved by varying the inductance LR (resonance impedance) of the controlled winding NR, and no malfunction of the electromagnetic relay occurs. Therefore, even if a protective circuit is not particularly provided, the power supply circuit according to the present embodiment does not break down due to an abnormality in the commercial AC power supply AC.

In this embodiment, the inductance component in the primary side series resonance circuit is a combination of the leakage inductance L1 of the primary winding N1 and the inductance LR of the controlled winding NR of the orthogonal control transformer PRT. . This is because, for example, the leakage of the primary winding N1 is obtained in order to obtain an inductance value necessary for the primary side series resonance circuit as compared with the case where the primary side series resonance circuit is formed only by the primary winding N1 and the series resonance capacitor C1. It means that the inductance L1 is smaller. This is because the remaining shortage of inductance is compensated by the inductance LR of the controlled winding NR.
In this case, since the leakage inductance L1 can be reduced as the insulating converter transformer PIT, the insulating converter transformer PIT can also reduce the number of turns of the primary winding N1 and further reduce the gap length. Accordingly, as the core of the insulating converter transformer PIT, for example, the place where the size of the EER-35 should be used can be made the size of the EER-28 as described above. It will also be illustrated.

Here, the operation of the main part of the power supply circuit shown in FIG. 2 is shown in the waveform diagrams of FIGS. FIG. 7 shows an operation when an AC input voltage VAC = 100 V (AC 100 V system) is input, and FIG. 8 shows an operation when an AC input voltage VAC = 220 V (AC 200 V system) is input.
As shown in FIGS. 7 (a) and 8 (a), the voltage VQ2 across the switching element Q2 is clamped at a predetermined level when the switching element Q2 is off, and has a waveform that is 0 level when the switching element Q2 is on. can get. The clamp level of the both-end voltage VQ2 corresponds to the level of the rectified and smoothed voltage Ei. Therefore, in actuality, the clamp level is higher in FIG. 8A corresponding to when the AC input voltage VAC = 220V than in FIG. 7A corresponding to when the AC input voltage VAC = 100V.
Further, the switching current IQ2 flowing through the switching element Q2 first flows through the clamp diode DD2 at the turn-on timing, and then reverses in accordance with the waveforms shown in FIGS. 7B and 8B. The switching element Q2 flows from the drain to the source.
Although not shown here, the voltage across the other high-side switching element Q1 and the switching current are AC input voltage VAC = 100V and 220V, respectively, and are shown in FIGS. (A) The both-end voltage VQ2 and switching current IQ2 of the switching element Q2 shown in (b) have the same waveform and have a phase difference of 180 °.
The switching output current I1 assumed to flow through the primary side series resonant circuit is a combination of the switching currents flowing through the switching elements Q1 and Q2, and is therefore shown in FIGS. 7 (c) and 8 (c). It will flow depending on the waveform.

  Further, as a characteristic of the power supply circuit of the embodiment shown in FIG. 2, the AC → DC power conversion efficiency (ηAC-DC) is, for example, an AC input voltage VAC = V.sub. A measurement result was obtained that was almost constant at 89.0% at 100 V, and almost constant at 89.5% at AC input voltage VAC = 220 V. This measurement is based on the selection of parts as described above.

Further, the present invention need not be limited to the configuration as the above-described embodiment.
For example, as a switching element, an IGBT (Insulated Gate Bipolar Transistor) or the like that can be used in a separate excitation type, the constants of each component element described above are also changed according to actual conditions. I do not care. Further, for example, the circuit configuration for generating the secondary side DC output voltage on the secondary side of the insulating converter transformer PIT may be appropriately changed.
Furthermore, the wide-range configuration according to the present invention can be applied to a self-excited resonance type converter.

It is a circuit diagram which shows the structural example of the switching power supply circuit based on the structure which the present applicant has proposed previously. It is a circuit diagram which shows the structural example of the power supply circuit as embodiment of this invention. It is a perspective view which shows the structural example of the orthogonal control transformer with which the power supply circuit as this Embodiment is equipped. It is a perspective view which shows the other example of the structure of the orthogonal control transformer with which the power supply circuit as this Embodiment is equipped. It is a figure which shows the inductance characteristic of the to-be-controlled winding wound by the orthogonal control transformer of embodiment. It is a figure which shows the constant voltage control characteristic of the power supply circuit of this Embodiment. It is a wave form diagram which shows operation | movement of the principal part of the power supply circuit of this Embodiment. It is a wave form diagram which shows operation | movement of the principal part of the power supply circuit of this Embodiment. It is a circuit diagram which shows the structural example of the switching power supply circuit as a prior art example.

Explanation of symbols

1 control circuit, 2 oscillation / drive circuit, 3 amplifier circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1, Q2 switching element, Cp partial resonance capacitor, PIT isolation converter transformer, N1 primary winding, C1 primary side series resonance capacitor , N2 secondary winding, Do1, Do2 (secondary side) rectifier diode, Co smoothing capacitor, PRT orthogonal control transformer, NR controlled winding, Nc control winding

Claims (1)

Rectified and smoothed voltage generating means for generating a rectified and smoothed voltage at a level corresponding to the same size as the input commercial AC power supply;
Switching means formed with a switching element that performs the switching operation by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
Winding at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage obtained as a switching output obtained in the primary winding is excited An insulating converter transformer formed by
A primary-side resonance capacitor provided so that a primary-side resonance circuit having a resonance type operation of the switching means is formed by at least the leakage inductance component of the primary-side winding of the insulating converter transformer and its own capacitance. When,
A controlled winding connected so that its own inductance becomes an inductance component forming the primary resonance circuit and a control winding are wound, and according to the level of the control current flowing in the control winding A control transformer formed such that the inductance of the controlled winding is variable;
Current control means provided so as to flow the control current at a level variable according to the level of the commercial AC power supply to the control winding;
DC output voltage generation means configured to input an alternating voltage obtained in the secondary winding of the insulation converter transformer and generate a secondary side DC output voltage by performing a rectification operation;
The switching drive means is controlled according to the level of the secondary side DC output voltage, and the switching frequency of the switching means is varied to perform constant voltage control on the secondary side DC output voltage. Constant voltage control means;
A switching power supply circuit comprising:

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