JP2005168188A - Switching power supply - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce the number of circuit components and the cost for manufacturing the circuit coping with a wide-range switching power supply circuit, where power factor is improved by a voltage-feedback system. <P>SOLUTION: A required control range of stabilization operation through switching frequency control of an AC 100V system and an AC 200V system is confined within the maximum control range of switching frequency, by switching the number of turns of the primary winding N1 of an insulated converter transformer PIT, for example, depending on the level of a commercial AC power supply AC. Consequently, stabilization can be carried out depending on the secondary DC output voltage Eo at the time of both AC 100V system and AC 200V system, and a wide-range switching power supply circuit is realized. Since a conventional arrangement for switching the rectification operation is not required for realizing a wide range switching power supply circuit, design of the rectifying/smoothing circuit and the power factor improving circuit is not limited. Consequently, the number of components is decreased, breakdown voltage of elements is lowered, and the circuit cost can be reduced. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit used as a power supply for various electronic devices, for example.

In recent years, the development of switching elements that can withstand relatively high currents and voltages at high frequencies has led to the majority of power supply circuits that rectify commercial power supplies to obtain desired DC voltages. .
The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

By the way, in general, when a commercial power source is rectified, the current flowing through the smoothing circuit becomes a distorted waveform, which causes a problem that the power factor indicating the power source utilization efficiency is impaired.
Further, there is a need for measures for suppressing harmonics generated by such a distorted current waveform.

  As the switching power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC85V to 288V is adopted so as to correspond to an AC input voltage AC100V region such as Japan and the United States and an AC200V system region such as Europe. A so-called wide-range power supply circuit that is configured so as to be able to be used is known.

Here, as the above-described resonance type converter, one configured to be stabilized by controlling the switching frequency of a switching element forming the converter (switching frequency control method) is known.
As such a resonant converter based on the switching frequency control system, for example, in a configuration in which the switching element is switched and driven by a general-purpose oscillation / drive circuit IC, for example, the variable range of the switching frequency fs is maximum, fs = 50 KHz to 250 KHz. It is about. In the case of such a variable range, for example, under a load condition where the fluctuation range of the load power Po is a relatively large fluctuation range from Po = 0 W to about 90 W, and further to about 150 W, a wide range of AC85V to 288V is used. Stabilization corresponding to the AC input voltage range is almost impossible.

  In order to solve each of these problems, the present applicant has previously proposed the following configuration as a switching power supply circuit that has a configuration for power factor improvement and is compatible with a wide range.

FIG. 10 shows a configuration of a conventional switching power supply circuit that can be configured based on the invention previously proposed by the present applicant.
First, in the power supply circuit shown in FIG. 10, a noise filter including an across capacitor CL and a common mode choke coil CMC is connected to the line of the commercial AC power supply AC. In the line of the commercial AC power supply AC, one set of filter capacitors CN is inserted in the subsequent stage of the noise filter. The filter capacitor CN is for suppressing normal mode noise generated in the rectified output line of the bridge rectifier circuit Di described below.

  In this case, the rectifier circuit system that generates a rectified smoothed voltage (DC input voltage) Ei from a commercial AC power supply includes a bridge rectifier circuit Di and two smoothing capacitors Ci1 and Ci2. The smoothing capacitors Ci1 and Ci2 have the same capacitance.

The bridge rectifier circuit Di is composed of four rectifier diodes D1 to D4 as shown in the figure. In this case, the connection point between the rectifier diodes D1 and D2 of the bridge rectifier circuit Di is a negative electrode of the commercial AC power supply AC through a series connection of a tertiary winding N3 and a high-frequency choke coil LS in the power factor correction circuit 10 described later. Connected with line.
Further, the connection point between the rectifier diodes D4 and D3 is connected to the connection point of the smoothing capacitors Ci1 to Ci2 via a relay switch S10 described later. The connection point between the rectifier diodes D2 and D4 of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci1, and the connection point between the rectifier diode D3 and the rectifier diode D1 is connected to the primary side ground.

  In the case of the power supply circuit shown in FIG. 10, the rectifier diode that forms the bridge rectifier circuit Di is used to cause the rectifier current to flow as the operation of the power factor correction circuit 10 corresponding to the switching period. As a fast recovery diode.

The smoothing capacitors Ci1 and Ci2 are connected in series so that the negative terminal of the smoothing capacitor Ci1 and the positive terminal of the smoothing capacitor Ci2 are connected as shown. In the series connection circuit using the smoothing capacitors Ci1 to Ci2, the positive terminal on the smoothing capacitor Ci1 side is connected to the output terminal (the connection point of D2 and D4) of the bridge rectifier circuit Di as described above. On the other hand, the negative terminal on the smoothing capacitor Ci2 side is connected to the primary side ground.
A rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the series connection circuit of the smoothing capacitors Ci1 to Ci2.

The relay switch S10 is provided to switch the rectifying operation of the rectifying / smoothing circuit between the AC 100V system and the AC 200V system.
As can be understood from the above description, the relay switch S10 is inserted between the connection point of the rectifier diodes D3 and D4 of the bridge rectifier circuit Di and the connection point of the smoothing capacitors Ci1 to Ci2.

The relay switch S10 is turned on / off according to the driving state of the relay RL connected to the rectifier circuit switching module 7 as shown in the figure.
The rectifier circuit switching module 7 performs an operation for switching on / off of the relay switch S10 between the AC 100V system and the AC 200V system by driving the relay RL.
This rectifier circuit switching module 7 is provided with a rectifying and smoothing circuit comprising a rectifying diode D5 and a smoothing capacitor C5 as shown in the figure. In this case, the anode of the rectifier diode D5 is connected to the negative line of the commercial AC power supply AC. The cathode is connected to the positive terminal of the smoothing capacitor C5 whose negative terminal is grounded to the primary side ground, and the connection point between the smoothing capacitor C5 and the diode D5 is a voltage dividing resistor as shown in the figure. It is connected to the detection terminal of the rectifier circuit switching module 7 via a connection point R1-R2.
As a result, a DC voltage of a level corresponding to the AC input voltage VAC is obtained at the detection terminal of the rectifier circuit switching module 7, and the rectifier circuit switching module 7 uses the commercial AC power supply AC based on the voltage level thus obtained. Can be detected.

A relay RL is provided for the rectifier circuit switching module 7. This relay RL performs on / off control of the relay switch S10 according to its conduction state.
In this case, for example, switching is performed so that the relay switch S10 is turned on when the relay RL is in a conductive state and is turned off when the relay RL is in a non-conductive state.

The rectifier circuit switching module 7 compares the voltage level of the commercial AC power supply AC input from the detection terminal as described above with a predetermined reference voltage (for example, 150 V). Based on the comparison result, the relay RL is turned on when the input voltage level to the detection terminal is lower than the reference voltage level, and the relay RL is turned off when the input voltage level is higher than the reference voltage.
That is, in this case, when the level of the commercial AC power supply AC is equal to or lower than the reference voltage and the AC = 100V system, the relay RL is turned on and the relay switch S10 is turned on. Further, when the level of the commercial AC power supply AC is equal to or higher than the reference voltage and the AC = 200V system, the relay RL is disabled and the relay switch S10 is turned off.

Here, for example, when the relay switch S10 is turned on in response to the AC 100V system by the operation of the rectifier circuit switching module 7 as described above, the positive line of the commercial AC power supply AC and the smoothing capacitor Ci1− The connection point of Ci2 is connected.
Therefore, during the period when the AC input voltage VAC is positive, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci2, as will be described later. Further, during the period when the AC input voltage VAC is negative, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci1.
As a result of the rectifying operation performed in this way, a level corresponding to the same multiple of the AC input voltage VAC is generated as the voltage across each of the smoothing capacitors Ci1 and Ci2. Therefore, a level corresponding to twice the AC input voltage VAC is obtained as the DC input voltage Ei, which is the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. That is, a so-called voltage doubler rectifier circuit is formed.

When the relay switch S10 is turned off corresponding to the AC 200V system, the positive line of the commercial AC power supply AC and the connection point between the smoothing capacitor Ci1 and the smoothing capacitor Ci2 are not connected.
According to this, in the rectifying / smoothing circuit in this case, the AC input voltage VAC is rectified by the bridge rectifier circuit Di in each period in which the AC input voltage VAC is positive / negative, and the smoothing capacitors Ci1-Ci2 are connected in series. The operation of charging the circuit with the rectified current is obtained. That is, a rectification operation by a full-wave rectifier circuit including a normal bridge rectifier circuit can be obtained.
Therefore, in this case, the DC input voltage Ei having a level corresponding to the same multiple of the AC input voltage VAC can be obtained as the voltage across the series connection circuit of the smoothing capacitors Ci1-Ci2.

As described above, in the circuit of FIG. 1, the operation of the rectifier circuit switching module 7, the relay RL, and the relay switch S <b> 10 described above causes the AC input voltage VAC to be 2 by the voltage doubler rectification operation in the case of the commercial AC power supply AC100V system. The DC input voltage Ei corresponding to the double is generated. In the case of the commercial AC power supply AC200V system, the DC input voltage Ei corresponding to the AC input voltage VAC is generated by the equal voltage rectification operation by the full-wave rectifier circuit.
That is, in the case of the commercial AC power supply AC100V system and the AC200V system, as a result, the DC input voltage Ei of the same level can be obtained, and thereby the wide range is supported.

  In this case, a basic configuration as a current resonance type converter is adopted as the switching converter that operates by inputting the DC input voltage Ei generated by the operation of the rectifying and smoothing circuit as described above. Here, as shown in the figure, two switching elements Q1 (high side) and Q2 (low side) by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2, respectively, in the illustrated direction.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, in order to switch the switching elements Q1 and Q2, for example, an oscillation / drive circuit 2 using a general-purpose IC is provided. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit circuit. Then, a drive signal (gate voltage) having a required frequency is applied to each gate of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

The insulating converter transformer PIT transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side. One end of the primary winding N1 of the insulation transformer PIT is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via a series connection of the primary side resonance capacitor C1. Thus, the switching output is transmitted.
The other end of the primary winding N1 is connected to the primary side ground.

Here, depending on the capacitance of the primary side resonance capacitor C1 and the leakage inductance L1 of the insulating converter transformer PIT including the primary winding N1, a primary side series resonance circuit for making the operation of the primary side switching converter a current resonance type can be obtained. To be formed.
That is, according to the above description, the primary side switching converter shown in this figure has a current resonance type operation by the primary side series resonance circuit (L1-C1) and the above-described partial voltage resonance circuit (Cp // L1). ) And partial voltage resonance operation.
That is, the power supply circuit shown in this figure adopts a form in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit. In this specification, such a switching converter is referred to as a composite resonance type converter.

  Although not shown in the figure, the structure of the insulating converter transformer PIT includes an EE type core in which an E type core made of, for example, a ferrite material is combined. Then, after dividing the winding part on the primary side and the secondary side, the primary winding N1 on the primary side, in this case, the tertiary winding N3 described later, and the secondary winding described next on the secondary side N2 is wound around the central magnetic leg of the EE core.

An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding N2 of the insulating converter transformer PIT.
In this case, the secondary winding N2 connects the center tap output to the secondary side ground. In addition, a full-wave rectifier circuit composed of rectifier diodes D01 and D02 and a smoothing capacitor C0 is connected to the secondary winding N2 as shown in the figure. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown) and is also branched and inputted as a detection voltage for the control circuit 1 described below.

  The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage EO to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. Thus, the level of the secondary side DC output voltage is stabilized by varying the switching frequency of the switching elements Q1, Q2.

Further, the power supply circuit shown in FIG. 10 includes a power factor correction circuit 10 as shown in the figure as a configuration for improving the power factor.
The power factor correction circuit 10 includes a tertiary winding N3 wound around the primary side of the insulating converter transformer PIT and a high-frequency choke coil LS connected in series with the tertiary winding N3.
Further, it includes a bridge rectifier circuit Di including the above-described fast recovery type rectifier diode and a filter capacitor CN inserted in parallel with the line of the commercial AC power supply AC.

  The high-frequency choke coil LS is connected to a filter capacitor CN having one end of the winding N10 inserted in the line of the commercial AC power supply AC. The other end of the winding N10 is connected to one end of the tertiary winding N3. Further, the other end of the tertiary winding N3 is connected to a connection point between the rectifier diodes D1 and D2 of the bridge rectifier circuit Di.

According to the configuration of the power factor correction circuit 10 as described above, the primary winding N1 that is also wound on the primary side as well as the tertiary winding N3 that is wound on the primary side of the insulating converter transformer PIT as described above. The alternating voltage corresponding to the primary side switching output obtained is excited.
That is, according to such a configuration, the alternating voltage corresponding to the primary side switching output is fed back to the rectification current path through the magnetic coupling between the primary winding N1 and the tertiary winding N3. It is what has become.

Here, in the power factor correction circuit 3 having the above configuration, when the AC input voltage VAC is obtained at a level corresponding to the AC 100V system, the relay is switched by the operation of the rectifier circuit switching module 7 as described above. The switch S10 is turned on.
In such a state, in a half cycle in which the AC input voltage VAC is negative, the rectified current is [high-frequency choke coil LS → third winding N3 → rectifier diode D2 → smoothing capacitor Ci1 → relay switch S10 → filter capacitor CN. ] Flow.
In the AC100V system, the path [relay switch S10 → smoothing capacitor Ci2 → rectifier diode D1 → tertiary winding N3 → high frequency choke coil LS → filter capacitor CN] in the half cycle in which the AC input voltage VAC is positive. Causes a rectified current to flow.

From such a rectified current path, it can be seen that in the AC 100V system, the rectification operation is performed by the rectifier diode D2 which is a high-speed recovery type diode in the negative half cycle of the AC input voltage VAC. It can also be seen that the rectification operation is performed by the rectifier diode D1, which is a fast recovery diode, in the half cycle in which the AC input voltage VAC is positive.
As described above, the rectified current at this time is obtained by feeding back the primary side switching output via the magnetic coupling between the primary winding N1 and the tertiary winding N3 in the insulating converter transformer PIT. Such a high-frequency component based on the switching output is obtained as a superimposed waveform.

In the power factor correction circuit 10 shown in FIG. 10, the primary side switching output is voltage-feedback in this way, and the fast recovery type rectifier diode is switched by the rectified current superimposed with the high frequency component as the switching output. It is what. As a result, even in a period in which the rectified output voltage level is originally lower than the voltage across the smoothing capacitor Ci (Ci1-Ci2), the rectifier diode conducts in response to the superimposed high-frequency component, and the smoothing capacitor Ci The charging current is allowed to flow.
As a result, the average waveform of the AC input current component approaches the waveform of the AC input voltage, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.
For confirmation, the high-frequency component superimposed on the rectified current as described above is removed by the filter capacitor CN provided in the AC line, so that the waveform of the AC input current IAC has such a high-frequency component. Ingredients will not appear.

On the other hand, in the circuit of FIG. 10, when the AC input voltage VAC is input at a level corresponding to the AC 200V system, the relay switch S10 is turned off as described above.
In the AC 200V system in which the relay switch S10 is thus turned off, the rectified current is [high-frequency choke coil LS → tertiary winding N3 → rectifier diode D2 → smoothing capacitor Ci1 when the AC input voltage VAC is a negative half cycle. -Ci2 → rectifier diode D3 → filter capacitor CN].
Further, in the positive half cycle, the rectified current flows along the path [rectifier diode D4 → smoothing capacitor Ci1-Ci2 → rectifier diode D1 → tertiary winding N3 → high frequency choke coil LS → filter capacitor CN].
That is, in response to the AC 200 V system, the rectification operation is performed by the combination of the rectifier diode D2 and the rectifier diode D3 of the fast recovery type in the half cycle in which the AC input voltage VAC is negative.
Further, in the positive half cycle, the rectification operation is performed by the combination of the rectification diode D1 and the rectification diode D4 of the fast recovery type.

As described above, even when the AC 200 V system is used, the fast recovery type diode is configured to perform a rectifying operation in both periods where the AC input voltage VAC is positive / negative.
As a result, the switching operation is performed in accordance with the high-frequency component based on the switching output to which these rectifier diodes are voltage-feedback as in the case of the AC100V system described above. In other words, this also allows the charging current to flow to the smoothing capacitor Ci during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci.

  From the above description, by providing the power factor correction circuit 10 shown in FIG. 10, the conduction angle of the AC input current IAC is expanded in both the AC100V system and the AC200V system, thereby improving the power factor. It will be a thing.

Further, FIG. 11 shows another example of the configuration of the power supply circuit for wide range having a configuration for power factor improvement.
In FIG. 11, the parts described in FIG. 10 are given the same reference numerals, and the description thereof is omitted.
As another example of the conventional configuration shown in FIG. 11, instead of the tertiary winding N3 for voltage feedback provided in the circuit of FIG. 10, the primary winding N1 of the insulating converter transformer PIT as shown in the figure. And the primary side series resonance capacitor C1 are configured in a power factor correction circuit 11 including a primary winding N4 inserted in series and a voltage feedback transformer VFT around which a secondary winding N5 is wound. Is.
In this power factor correction circuit 11, the high frequency choke coil LS provided in the power factor correction circuit 10 of FIG. 10 is omitted.
Then, depending on the voltage feedback transformer VFT provided as described above, a primary side switching output is obtained in the primary winding N4, and an alternating voltage corresponding thereto is excited in the secondary winding N5. Also, the primary side switching output is voltage-feedbacked to the rectified current path.

  As a result, also in the configuration shown in FIG. 11, the high-speed recovery type diode that constitutes the bridge rectifier circuit Di is intermittently based on the high-frequency component that is voltage-feedback in this way. The charging current to the smoothing capacitor Ci can also flow during a period that is lower than the voltage across the terminals. As a result, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.

A configuration for improving the power factor by the voltage feedback method as provided in the circuits of FIGS. 10 and 11 is shown in Patent Document 1 below. Patent Document 2 discloses a technique for realizing a wide range by switching the rectifying operation.
Japanese Patent Laid-Open No. 2003-189617 JP-A-7-281770

In this way, in the conventional configuration shown in FIGS. 10 and 11, the power factor improvement circuit by the voltage feedback system is provided, and the power factor is improved by expanding the conduction angle of the AC input current IAC. It is done.
Further, as described above, in the circuits shown in FIGS. 10 and 11, the DC input voltage Ei according to the voltage level detected from the line of the commercial AC power supply AC by the operation of the rectifier circuit switching module 7 and the relay switch S10. By switching the rectifying / smoothing circuit that generates the voltage by voltage doubler rectification / full wave rectification, a wide-range configuration that can operate in both the AC100V system and the AC200V system is realized.

However, the conventional power supply circuit as shown in FIGS. 10 and 11 has the following problems.
First, in the circuits shown in FIGS. 10 and 11, the degree of freedom in circuit design is limited because the wide range is supported by switching the rectifying operation as described above.
For example, in the circuits of FIGS. 10 and 11, two smoothing capacitors Ci1 and Ci2 are configured as smoothing capacitors for obtaining the DC input voltage Ei when switching such a rectifying operation. .
However, in order to switch the rectifying operation in this way, two smoothing capacitors must be provided, which increases the number of circuit components and increases the cost, and also increases the mounting area of the power circuit board. And it will become larger. In particular, since the smoothing capacitor Ci is in a large category among the components constituting the power supply circuit, such an increase in the substrate size is further promoted.

In the circuits of FIGS. 10 and 11, when switching the rectification operation, the relay switch S10 needs to be inserted between the connection point of the smoothing capacitors Ci1 to Ci2 and the bridge rectification circuit Di as illustrated above. was there.
However, in this case, when the power is turned on, an inrush current to the smoothing capacitor Ci flows. And depending on this, since the insertion position is as described above, this inrush current flows through the relay switch S10, and accordingly, a relatively high level is required as a contact capacity in the relay RL. It was.
In this case, the rush current is suppressed to, for example, 30 A or less by a limiting resistor or a power thermistor although not shown in the figure, but in order to cope with this, the contact capacity of the relay RL in this case is required to be 30 A or more. Will be.
In order to obtain a relatively high contact capacity in this way, the components in this case are expensive, and this also promotes an increase in cost.

In the circuits of FIGS. 10 and 11, as understood from the above description, a high-frequency pulse voltage based on the switching period is superimposed on the AC input voltage component. For this reason, as described above, the filter capacitor CN is inserted into the line of the commercial AC power supply AC.
However, since the circuits of FIGS. 10 and 11 are configured to switch the rectification operation between voltage doubler and full-wave rectification in order to support a wide range, the insertion position of such a filter capacitor CN is the commercial AC power supply AC. It was limited to the position parallel to the line. In other words, when a power supply circuit having a configuration for power factor improvement by the voltage feedback method is adopted for a wide range configuration by switching the conventional rectifying operation, the insertion position of the filter capacitor CN for removing such noise is It will be limited.
Since the filter capacitor CN in this case must be inserted in parallel with the AC line in this way, a high breakdown voltage of 200 V or more is required.
For example, the filter capacitor CN in the circuits of FIGS. 10 and 11 uses a relatively expensive component of 1 μF / 250 V, for example, as a withstand voltage product that satisfies the safety standards of each country. Cost increase was encouraged.

Similarly, depending on the configuration of FIGS. 10 and 11, a fast recovery diode that can be turned on / off according to the switching period is inserted so as to operate in common in the rectified current path in the AC 100V system and the 200V system. Must be inserted into the bridge rectifier circuit Di as described above.
For this reason, in the circuits of FIGS. 10 and 11, it is necessary to use, for example, a 3A / 600V withstand voltage / withstand voltage product that can guarantee an inrush current 30A when the power is turned on as the fast recovery diode constituting the bridge rectifier circuit Di This also helped increase costs.

  Furthermore, as a diode constituting the bridge rectifier circuit Di as described above, there is no commercially available product in which such a fast recovery type diode is bridge-coupled at present. Will have to. That is, in this case, it is difficult to configure the bridge rectifier circuit Di with one component, and this also causes a problem that the number of components increases.

Furthermore, as another problem, as described above, as a circuit for switching between full-wave rectification operation and voltage doubler rectification operation, for example, when the basic configuration shown in FIGS. Such a malfunction may occur. In other words, when AC200V commercial AC power is being input, if an instantaneous power failure occurs or the AC input voltage drops below the rated value, the AC100V level becomes lower than that corresponding to the AC200 system. An operation of switching to a voltage doubler rectifier circuit occurs as a system.
When such a malfunction occurs, voltage doubler rectification is performed on an AC input voltage of AC200V system level, so that there is a possibility that the switching elements Q1, Q2, etc. may be destroyed due to overvoltage.

10 and 11, in order not to cause the above-described malfunction, not only the DC input voltage of the main switching converter but also the DC of the converter circuit on the standby power supply side. A configuration for detecting the input voltage is also adopted. For example, in the case of FIG. 10, the illustrated power supply circuit is the main switching converter. Actually, the rectifier circuit switching module 7 also inputs a DC input voltage obtained by a converter of a standby power supply not shown in FIG. 10 as a detection voltage.
Further, as the standby power supply side converter circuit is detected as described above, a comparator IC is mounted as the rectifier circuit switching module 7. For this reason, for example, an external component of IC or a peripheral circuit is mounted. As a result, the number of parts increases and the above-described cost increase and circuit board size increase are further promoted.
In the case of adopting such a configuration, the rectifier circuit switching module 7 is formed as a unit by assembling a circuit system for switching the rectification operation as a module, for example, in order to improve parts management and work efficiency during manufacturing. The However, in the case of unitization in this way, more parts such as adding pin terminals are required, so that an increase in cost is promoted.

  In addition, the detection of the DC input voltage of the converter on the standby power supply side for the purpose of preventing malfunctions means that a wide range compatible power supply circuit having a circuit for switching the rectifying operation includes a standby power supply in addition to the main power supply. This means that it cannot actually be used unless it is an electronic device. In other words, the types of electronic devices that can be equipped with a power supply are limited to those equipped with a standby power supply, and there is a problem that the use range is narrowed accordingly.

Therefore, in the present invention, in view of the various problems described above, the switching power supply circuit is configured as follows.
That is, first, a rectified and smoothed voltage generating means for generating a rectified and smoothed voltage having a level corresponding to the same size as the input commercial AC power supply, and the rectified and smoothed voltage is input as a DC input voltage to perform a switching operation. A switching means formed by half-bridge coupling two switching elements; and a switching driving means for driving each of the switching elements.
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding to which an alternating voltage as a switching output obtained in the primary winding is excited are wound. The insulating converter transformer formed at least, the leakage inductance component of the primary winding of the insulating converter transformer, and the capacitance of the insulating converter transformer form a primary side resonance circuit that resonates the operation of the switching means. A primary-side resonance capacitor provided in this manner.
A DC output voltage generating means configured to generate a secondary side DC output voltage by inputting an alternating voltage obtained in the secondary winding of the insulating converter transformer and performing a rectifying operation; The switching driving means is controlled according to the level of the secondary side DC output voltage, and the switching frequency of the switching means is variably controlled, thereby performing constant voltage control on the secondary side DC output voltage. Constant voltage control means.
Then, the alternating voltage based on the switching output by the switching means is fed back to the rectified current path, and the rectified current component is converted by the diode element provided in the rectifying and smoothing means using the alternating voltage based on the switching output. A power factor correction circuit configured to intermittently improve the power factor, and a switching configured to switch at least an inductance value forming the primary side resonance circuit according to a level of the commercial AC power supply And means.

According to the above configuration, the switching power supply circuit according to the present invention increases the conduction angle of the AC input current by switching the rectifier diode using the alternating voltage based on the switching output fed back to the rectified current path. Therefore, a configuration for improving the power factor by a so-called voltage feedback system is provided.
In addition, according to the above configuration, at least the value of the inductance component forming the primary side resonance circuit is switched according to the level of the commercial AC power supply by the operation of the switching means. In this way, if at least the inductance value forming the primary resonance circuit is switched according to the level of the commercial AC power supply, the control range of the switching frequency required for stabilization can be set, for example, at the level of the commercial AC power supply. It becomes possible to set according to.

Thus, according to the present invention, it is possible to set the control range of the switching frequency required for stabilization in conformity with the rated level of the commercial AC power supply.
Accordingly, for example, regardless of the rated level of the commercial AC power supply, the control range of the switching frequency required for stabilization can be set within the variable range of the switching frequency in the switching power supply circuit, for example. That is, a wide range compatible switching power supply circuit can be obtained that can be properly stabilized regardless of the level change of the commercial AC power supply.

  This eliminates the need for a rectifier circuit that generates a DC input voltage (rectified and smoothed voltage) to adopt a conventional configuration in which switching is performed between a voltage doubler rectification operation and a full-wave rectification operation according to the commercial AC power supply level. In other words, as a switching power supply circuit having a configuration for improving the power factor by the voltage feedback method, a configuration for switching the rectifier circuit as in the conventional case for making it compatible with a wide range can be made unnecessary. .

In this way, if the configuration for switching the rectifying operation for adapting to the wide range can be eliminated, the degree of freedom in circuit design can be increased.
For this reason, the number of smoothing capacitors forming the rectifier circuit for generating the DC input voltage is reduced from, for example, two to one.

  In addition, the degree of freedom increases as the configuration of the power factor correction circuit using the voltage feedback method. For example, a filter capacitor for removing noise from a commercial AC power line need not be inserted in parallel with the commercial AC power line as in the past. By reducing the breakdown voltage, a cheaper element can be used.

Similarly, a rectifier diode used for power factor improvement need not be inserted so as to operate in common in a rectification current path formed by a 100V system and a 200V system of a commercial AC power supply.
Depending on this, it becomes possible to insert the high-speed recovery type necessary for realizing the power factor improvement operation by the voltage feedback method into other parts instead of being incorporated in the bridge rectifier circuit, for example. It becomes possible to use a cheaper element by reducing the withstand voltage of the diode.

  In addition, if it is not necessary to incorporate a high-speed rectifier diode used for power factor improvement in the bridge rectifier circuit as described above, a commercially available bridge rectifier circuit can be used, and the number of parts can be reduced compared to the conventional one. Can be achieved.

Hereinafter, the best mode for carrying out the invention (hereinafter referred to as an embodiment) will be described.
FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to the first embodiment of the present invention.
The switching power supply circuit according to the present embodiment is assumed to be used as a power supply for a printer device, for example. In the printer device, the load power when the power is turned off is almost 0 W. Therefore, the power source provided for this needs to cover a relatively wide variation range from the load power 0 W to the maximum load. In this example, since the maximum load Po = 100 W is assumed, the switching power supply circuit in this case is adapted to cope with load fluctuations of load power Po = 100 W to 0 W.

First, in FIG. 1, in the power supply circuit of this example as well as the circuits shown in FIGS. 10 and 11, the noise composed of the across capacitor CL and the common mode choke coil CMC is applied to the line of the commercial AC power supply AC. A filter is connected.
In this case, the filter capacitor CN is not inserted into the subsequent stage of such a noise filter. That is, in the line of the commercial AC power supply AC, as shown in the figure, the bridge rectifier circuit Di is directly connected to the subsequent stage of the noise filter.
In this case, as a rectifier diode forming the bridge rectifier circuit Di, all diodes of a normal low speed recovery type are selected.

In this case, the rectifier circuit system that generates the rectified and smoothed voltage (DC input voltage) Ei from the commercial AC power supply AC includes the bridge rectifier circuit Di and a set of smoothing capacitors Ci.
The filter capacitor CN is provided so as to be inserted in series between the output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci as shown in the figure.

Further, as a switching converter that operates by inputting the DC input voltage Ei generated by the rectifying and smoothing circuit constituted by the bridge rectifying circuit Di and the smoothing capacitor Ci as described above, this case is also a current resonance type converter. The basic configuration is taken.
Also here, two switching elements Q1 (high side) and Q2 (low side) by MOS-FETs are connected by half bridge coupling as shown in the figure. In this case, damper diodes DD1 and DD2 are connected in parallel to the drains and sources of the switching elements Q1 and Q2, respectively.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. That is, as in the circuits of FIGS. 10 and 11, the parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1, thereby forming the switching element Q1. , Q2 so as to obtain a partial voltage resonance operation in which voltage resonance occurs only when Q2 is turned off.

The power supply circuit shown in this figure also includes, for example, an oscillation / drive circuit 2 using a general-purpose IC in order to drive the switching elements Q1 and Q2. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit for driving a switching element, similar to those shown in FIGS. Further, in the case of the present embodiment, a protection circuit 2a is provided as shown in the figure.
Depending on the oscillation circuit and the drive circuit, a drive signal (gate voltage) having a required frequency is applied to each gate of the switching elements Q1 and Q2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.
Further, the protection circuit 2a can detect, for example, overcurrent and overvoltage states in the power supply circuit. When an overcurrent or overvoltage state is detected, the switching operation of the switching elements Q1 and Q2 is controlled so that the circuit is protected.
Here, the protection circuit 2 a is provided so as to be built in the oscillation / drive circuit 2, but may be provided in the power supply circuit as a circuit unit independent of the oscillation / drive circuit 2. Good.

In this figure, the line of the operating power supply to be supplied to the oscillation / drive circuit 2 is omitted, but in practice, for example, the winding wound around the primary side of the insulating converter transformer PIT, A rectifier circuit that rectifies and smoothes the alternating voltage excited by the winding may be provided, and a low-voltage DC voltage obtained by the rectifier circuit may be supplied as an operating power source.
In addition, the configuration for starting the oscillation / drive circuit 2 may be a configuration in which, for example, the rectified and smoothed voltage Ei obtained through the startup resistor Rs is input as the startup power supply.

The insulating converter transformer PIT transmits the switching outputs of the switching elements Q1 and Q2 to the secondary side.
In the case of the present embodiment, the primary winding N1 of the insulating transformer PIT is formed by connecting two winding portions of the primary winding portions N1A and N1B in series. One end of the primary winding N1A is connected to the primary side ground. The other end of the primary winding N1A is connected to a terminal t2 of the relay switch S1 shown in the figure.
One end portion of the primary winding portion N1B is connected to the other end portion of the primary winding portion N1A. That is, the connection point between the ends of the primary windings N1B and N1A is the tap output of the primary winding N1, and is connected to the terminal t2 of the relay switch S1.
The other end of the primary winding N1B is connected to the terminal t3 of the relay switch S1.

The relay switch S1 in this case is switched so that either the terminal t2 or the terminal t3 is selectively connected to the terminal t1, and this terminal switching operation is performed by the relay RL. It is controlled according to conduction / non-conduction. That is, the relay RL and the relay switch S1 are configured as so-called one-circuit two-contact electromagnetic relays.
The terminal t1 of the relay switch S1 is connected to a connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2 via the primary side resonance capacitor C1.

In the case where the relay switch S1 is provided as described above, when the terminal t3 is connected to the terminal t1, the primary winding of the insulating converter transformer PIT connected in series to the primary side resonance capacitor C1. As N1, primary winding portions N1A-N1B are connected in series.
On the other hand, when the terminal t2 is connected to the terminal t1, the primary winding N1 of the insulating converter transformer PIT connected in series to the primary side resonance capacitor C1 is only the primary winding N1A. Will be.

  As understood from the above description, the series connection circuit of the primary side resonance capacitor C1 and the primary winding N1 connected to the switching output points of the switching elements Q1 and Q2 is the capacitance of the resonance capacitor C1. And a primary side series resonance circuit including a leakage inductance L1 of the insulating converter transformer PIT including the primary winding N1. Then, the primary side series resonant circuit performs a resonance operation by the switching outputs of the switching elements Q1 and Q2, thereby making the operation of the primary side switching converter the current resonance type.

  In the case of the present embodiment, switching of the relay switch S1 causes the primary winding N1 to be a series connection circuit of the primary winding portions N1A-N1B or only the primary winding portion N1A. In either case, the primary winding N1 and the primary side resonance capacitor C1 are connected in series, so that the primary side series resonance circuit is formed. However, the number of turns as the primary winding N1 varies depending on whether the primary winding N1 is a series connection circuit of the primary winding N1A-N1B or only the primary winding N1A. Thus, the leakage inductance L1 on the primary side of the insulating converter transformer PIT also changes.

  Further, on the primary side of the power supply circuit shown in FIG. 1, the switching between the relay switch S1 and a relay switch S2 described later is controlled by controlling the relay RL and the conduction / non-conduction of the relay RL. A relay switching circuit 5 configured as described above is provided.

  The relay switching circuit 5 receives the rectified and smoothed voltage Ei as a detection voltage. In practice, the detected voltage may be a voltage obtained by dividing the rectified and smoothed voltage Ei by a predetermined ratio with a voltage dividing resistor having a predetermined resistance value, for example. Here, since the rectified and smoothed voltage Ei is obtained by rectifying and smoothing the commercial AC power supply AC, a level corresponding to the same level as that of the commercial AC power supply AC is obtained. Therefore, the detection voltage obtained based on the rectified and smoothed voltage Ei also corresponds to the level of the commercial AC power supply AC.

In the relay switching circuit 5, the detected voltage is within a rated level range where the level of the commercial AC power supply AC is AC100V, and within a rated level range where the level of the commercial AC power supply AC is AC200V. At some point, the relay RL is switched between conduction and non-conduction. In this way, according to the switching of the conduction / non-conduction of the relay RL, in the relay switch S1 described above and the relay switch S2 described later, the state where the terminal t1 is connected to the terminal t3, and the terminal t2 The connected state and switching are performed.
In this case, for the AC100V system, switching is performed so that the terminal t2 is selected in both the relay switches S1 and S2. Further, in response to the AC200V system, switching is performed so that the terminal t3 is selected.

Moreover, as a structure of the insulating converter transformer PIT, for example, an EE type core that is a combination of an E type core made of a ferrite material is provided. And after dividing | segmenting a winding site | part by the primary side and a secondary side, the primary winding N1 mentioned above on the primary side, the tertiary winding N3 mentioned later in this case, and the secondary side are secondary Winding N2 is wound around the central magnetic leg of the EE core.
A gap having a predetermined length is formed with respect to the central magnetic leg of the EE type core. Thus, for example, a predetermined coupling coefficient that is loosely coupled is given.

In this case, a secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT. An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding N2.
The secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then a double-wave rectifier circuit comprising rectifier diodes DO1 and DO2 and a smoothing capacitor CO as shown in the figure. Is connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown). The detection voltage for the control circuit 1 is also branched and input.

  The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage EO to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. That is, the level of the secondary side DC output voltage is stabilized by the switching frequency control method.

In addition to the configuration described above, the power supply circuit shown in FIG. 1 is provided with a power factor correction circuit 3 for improving the power factor by a voltage feedback method.
Also in the configuration of the power factor correction circuit 3 in this case, as in the case shown in FIG. 10, the tertiary winding N3 wound on the primary side of the insulating converter transformer PIT and the tertiary winding N3 are connected. A configuration is adopted in which the primary side switching output is fed back to the rectified current path by the high frequency choke coil LS.

  As the high-frequency choke coil LS, for example, a winding N10 is wound around an EE type core (EE-shaped core) in which an E type core made of a ferrite material is combined so that magnetic legs of each other face each other. That's fine. As the inductance value of such a high frequency choke coil LS, a relatively low value such as about 100 μH may be set.

In this case, as described above, the power factor improving circuit 3 includes a filter capacitor CN for removing a high frequency component due to a voltage feedback switching period, an output terminal of the bridge rectifier circuit Di, and a smoothing capacitor. It is assumed that it is inserted between the positive electrode terminal of Ci.
Further, as a fast recovery type diode for intermittently rectifying current component based on the voltage feedback component, as shown as diode D10 in the figure, it is not incorporated in the bridge rectifier circuit Di in this case. It is said. That is, in such a fast recovery type diode D10, its anode is connected to the connection point between the output terminal of the bridge rectifier circuit Di and the filter capacitor CN, and its cathode is the tertiary winding of the high frequency choke coil LS. It is connected to the end that is not on the N3 side.

Further, as the power factor improving circuit 3 in this case, the tertiary winding N3 is divided into two winding parts, a tertiary winding part N3A and a tertiary winding part N3B. In addition, for such a tertiary winding N3, switching is performed between a state where the number of turns of the entire tertiary winding N3 is valid and a state where only the tertiary winding portion N3A is valid. Relay switch S2 is connected. The relay switch S2 is also a two-contact type in which the terminal t2 or the terminal t3 is alternatively selected with respect to the terminal t1.
One end of the tertiary winding N3A is connected to the winding N10 of the high-frequency choke coil LS, and the other end is connected to the terminal t2 of the relay switch S2. One end portion of the tertiary winding portion N3B is connected to the terminal t2 by being connected to the other end portion of the tertiary winding portion N3A, and the other end portion of the tertiary winding portion N3B is Connected to terminal t3 of relay switch S2.

Here, as described above, depending on the operation of the relay switching circuit 5, the terminal t2 is selected in each of the relay switches S1 and S2 corresponding to the AC 100V system. Further, the terminal t3 is selected corresponding to the AC200V system.
According to such an operation, in the tertiary winding N3 in the above connection form, switching is performed so that only the tertiary winding portion N3A is effective in the AC 100V system. In the AC200V system, the series connection of the tertiary winding portion N3A and the tertiary winding portion N3B is effective, and switching is performed so that the entire tertiary winding N3 is effective. That is, by such a switching operation, the number of turns of the tertiary winding N3 is increased in the AC 200V system.

By providing the power factor correction circuit 3 having the above configuration, in the circuit shown in FIG. 1, the rectified current is [bridge rectifier circuit Di → diode D10 → high frequency choke coil in a half cycle in which the AC input voltage VAC is positive. It flows through the path of L S → tertiary winding N 3 → relay switch S 2 → smoothing capacitor Ci].
In this case, the rectified current branches and also flows through the path [bridge rectifier circuit Di → filter capacitor CN → smoothing capacitor Ci].
Even in a half cycle in which the AC input voltage VAC is negative, the rectified current flows through the same path as described above.

  From such a rectified current path, it can be seen that the diode D10, which is a fast recovery diode, performs a rectifying operation in the circuit of FIG. As described above, the rectified current at this time includes a component based on the primary-side switching output that is fed back through the magnetic coupling between the primary winding N1 and the tertiary winding N3 in the insulating converter transformer PIT. The waveform is superimposed.

In the power factor correction circuit 3 shown in FIG. 1, the primary side switching output is voltage-feedback in this way, and the fast recovery type diode is switched by the rectified current superimposed with the high frequency component as the switching output. It is what. As a result, even in a period in which the rectified output voltage level is originally lower than the voltage across the smoothing capacitor Ci, the fast recovery diode (D10) conducts in response to the superimposed high frequency component, and the smoothing capacitor Ci. The charging current flows into.
As a result, the average waveform of the AC input current component approaches the waveform of the AC input voltage, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.
In this case, since the filter capacitor CN is provided so as to be connected in parallel with the tertiary winding N3, a noise component generated in the rectified output line of the bridge rectifier circuit Di is removed. It is what.

  In the power factor correction circuit 3 of this example, as described above, the number of turns of the tertiary winding N3 is switched. This will be described later.

Next, the operation corresponding to the wide range of the power supply circuit having the configuration shown in FIG. 1 will be described.
First, when a rated AC 100V system is input as a commercial AC power supply, the detection voltage input to the relay switching circuit 5 is in a level range corresponding to the rated AC 100V system. In this state, as described above, the relay switching circuit 5 switches the relay RL between the conductive state and the non-conductive state so that the relay switch S1 selects the terminal t2.
Thus, when the AC100 system is input as the commercial AC power supply, only the primary winding portion N1A is effective as the primary winding N1 of the insulating converter transformer PIT.

On the other hand, when the rated AC 200V system is input as the commercial AC power supply, and the detection voltage input to the relay switching circuit 5 is in the level range corresponding to the rated AC 200V system, the relay In the switching circuit 5, the relay RL is controlled so that the relay switch S1 selects the terminal t3. That is, when the AC200 system is input as the commercial AC power supply, the primary winding N1B-N1A in series connection is effective as the primary winding N1 of the insulating converter transformer PIT.
That is, the number of turns (the number of turns) as the primary winding N1 of the insulating converter transformer PIT is switched between the case where the commercial AC power supply is the AC 100V system and the AC 200V system.

As a configuration in the relay switching circuit 5, as a threshold level of the detection voltage corresponding to switching between conduction / non-conduction of the relay RL, for example, in the range of the AC input voltage VAC 144V to 170V, the actual use conditions, etc. It may be set in consideration.
Although not shown here, the power for driving the relay RL may be supplied from the oscillation / drive circuit 2.

FIG. 2 is a diagram illustrating the secondary-side DC output voltage Eo obtained when the configuration in which the number of turns as the primary winding N1 is switched between when the AC100V system is input and when the AC200V system is input as described above. The voltage control characteristic is exemplarily shown by the relationship between the switching frequency and the level of the secondary side DC output voltage Eo.
Here, the secondary side DC output voltage Eo is supposed to be stabilized at 21.5 V, for example. As a switching frequency control method, the switching frequency is variably controlled in a frequency range higher than the resonance frequency fo of the primary side series resonance circuit (C1-L1), and a change in resonance impedance caused thereby is used. So-called upper side control is adopted. In this figure, the AC input voltage VAC = 100 V for the AC 100 V system, and the AC input voltage VAC = 220 V for the AC 200 V system.

Here, as the number of turns as the primary winding N1 is switched as in the power supply circuit of FIG. 1, the leakage inductance L1 on the primary side of the insulating converter transformer PIT changes so as to be switched. .
As a method of changing the leakage inductance L1, for example, when the number of turns of the primary winding N1 increases as in the circuit of FIG. 1, the level of the secondary side DC output voltage Eo is fixed accordingly. It will change so as to be lower than the case. That is, in the circuit of FIG. 1, the level of the secondary side DC output voltage Eo at the time of AC200V system is made lower than when the number of turns is fixed.

On the other hand, the leakage inductance L1 forms a primary side series resonance circuit (C1-L1) together with the capacitance of the primary side resonance capacitor C1, and therefore, when the leakage inductance L1 changes, the primary side series resonance. The resonance frequency fo of the circuit (C1-L1) changes.
That is, in the circuit of FIG. 1, as understood from the above description, the leakage inductance L1 is set to be smaller in the AC100V system than in the AC200V system, so the resonance frequency fo1 in the AC100V system is Comparing with the resonance frequency fo2 in the AC 200V system, the resonance frequency fo1 is higher than the resonance frequency fo2 as shown in FIG.

  As a general matter, the series resonance circuit has the smallest resonance impedance at the resonance frequency fo. Thereby, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs, the level of the secondary side DC output voltage Eo becomes the highest when the switching frequency fs is the same as the resonance frequency fo, and is separated from the resonance frequency fo. It goes down as you go.

  That is, in FIG. 2, first, in the AC 100V system (VAC = 100V), as shown by the solid line, the secondary side DC output voltage Eo has a switching frequency fs of resonance of the primary side series resonance circuit (C1-L1). A peak at the frequency fo1 shows a quadratic curve-like change in which the level decreases as the frequency fo1 moves away from the resonance frequency fo1. In addition, the level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs can be shifted so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  If the secondary side DC output voltage Eo is to be stabilized at 21.5V by upper side control under the characteristics of the AC100V system shown by the solid line in FIG. 2 as described above, this is necessary. The variable range (necessary control range) of the switching frequency is the range indicated as Δfs1.

On the other hand, in the AC200V system, as described above, the resonance frequency fo of the primary side series resonance circuit (C1-L1) is set to a predetermined frequency indicated by fo2 lower than fo1 in the AC100V system. The characteristic at this time is shown by a broken line in FIG. That is, in this case as well, the secondary side DC output voltage Eo is maximized when the switching frequency fs is the resonance frequency fo2, and as the switching frequency fs moves away from the resonance frequency fo2. It will decrease. Also in this case, if the switching frequency fs is fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.
The required control range when the secondary side DC output voltage Eo is to be stabilized at 21.5 V by upper side control under the characteristics of the AC 200 V system is the range indicated by Δfs2 in FIG. .

In the case of the circuit of FIG. 1, the leakage on the primary winding N1 side of the insulating converter transformer PIT is changed by changing the number of turns as the primary winding N1 between the AC 100V system and the AC 200V system. The inductance L1 is also changed and set. By switching the setting of the leakage inductance L1 in this way, for example, the level of the secondary side DC output voltage Eo obtained corresponding to the AC 200 V system is changed.
In this way, by changing the setting of the leakage inductance L1, the level of the secondary side DC output voltage Eo as shown in FIG. 2 is changed by changing the level of the secondary side DC output voltage Eo. Can also be changed.
And if the characteristic about the secondary side direct-current output voltage Eo is changed in this way, the necessary control range Δfs also changes accordingly.

For example, when the number of turns of the primary winding N1 is not switched under the configuration of the power supply circuit shown in FIG. 1 and is fixed regardless of the commercial AC power supply level, it is proportional to the level of the commercial AC power supply AC. In this way, the level of the secondary side DC output voltage Eo is also increased.
Further, when the number of turns of the primary winding N1 is not switched as described above, the resonance frequency fo of the primary side series resonance circuit (C1-L1) is also fixed with respect to the commercial AC power supply level.
For this reason, in this case, a region having a switching frequency higher than the fixed resonance frequency fo is set as a necessary control range Δfs, and in a range corresponding to the AC100V system to AC200, it corresponds to a predetermined load fluctuation range. Therefore, it is necessary to stabilize the secondary side DC output voltage Eo having a very large fluctuation range. At this time, for example, under a condition where the load fluctuation range is large, the necessary control range Δfs must be very wide, and proper stabilization is achieved within the maximum variable range of the switching frequency determined by the specifications of the switching power supply circuit. Can be difficult.

  On the other hand, in the circuit of FIG. 1, the characteristics of the secondary side DC output voltage Eo shown in FIG. 2 in the 200 V system, for example, according to the input level change of the commercial AC power source in the AC 100 V system and the AC 200 V system are shown. The required control range Δfs2 can be set to a position as shown in the figure by changing it. This is because the required control range is selected by selecting the number of turns of the primary winding N1 in the AC 100V system and the 200V system. This means that each of Δfs1 and Δfs2 can be set to fall within the maximum variable range of the switching frequency. As a result, it is guaranteed that the secondary side DC output voltage Eo is stabilized corresponding to a predetermined load fluctuation range in each of the AC100V system and the AC200V system. That is, it is possible to realize wide range compatibility as a power supply circuit that is stabilized by the switching frequency control method.

As an actual example of the circuit shown in FIG. 1, the insulation converter transformer PIT has an EER-35 ferrite core and a central magnetic leg with respect to the load condition Po = 0W to 90W. In this case, a gap of 1.6 mm is formed. Further, the number of turns of the primary winding portion N1A is 50T (turns), and the primary winding portion N1B is 17T.
In this case, the secondary winding N2 is set to 20T (10T + 10T with the center tap as a division position). As the insulating converter transformer PIT, a loose coupling state represented by a coupling coefficient k = 0.72 is obtained.
Further, 0.033 μF is selected as the capacitance of the primary side series resonance capacitor C1.

The experimental results shown in FIG. 2 indicate that the required control ranges Δfs1 and Δfs2 are different frequency ranges. However, the actual required control ranges Δfs1 and Δfs2 may be different from the required control range Δfs1. The frequency ranges are obtained by overlapping so as to be within the frequency range of the necessary control range Δfs2.
That is, the result shown in FIG. 2 is a result when a value different from the maximum load power under the actual load condition is set, and the necessary control range Δfs is also different from the actual value. That is, the result shown in FIG. 2 is merely a conceptual diagram as an explanatory diagram for setting the required control range Δfs within the maximum variable range of the switching frequency fs by changing the leakage inductance L1. There is nothing.

  Further, with respect to the necessary control ranges Δfs1 and Δfs2 (that is, the number of turns as the primary winding N1), for example, the AC 100V system is guaranteed to be stabilized in the range of the AC input voltage VAC = 85V to 144V, and the AC 200V system Is set in such a manner that it is guaranteed to be stabilized in the range of AC input voltage VAC = 170V to 288V. In response to this, the switching operation of the number of turns as the primary winding N1 is performed by setting a threshold within the range of the AC input voltage VAC = 144V to 170V as described above.

3 and 4 show the characteristics of the power supply circuit of the embodiment shown in FIG. 1 as follows: AC → DC power conversion efficiency (ηAC-DC), power factor PF, fluctuation range ΔEi of DC input voltage Ei. The characteristics will be described.
FIG. 3 shows the characteristics of AC → DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range ΔEi of DC input voltage Ei with respect to fluctuations in load power Po = 100 W to 0 W in the circuit of FIG. FIG. 4 shows each characteristic of AC → DC power conversion efficiency (ηAC−DC), power factor PF, and fluctuation range ΔEi of DC input voltage Ei with respect to fluctuations of AC input voltage VAC = 85V to 288V.
In these figures, for each characteristic, in FIG. 3, the result when the AC input voltage VAC = 100V is constant is shown by a solid line, and the result when the AC input voltage VAC = 230V is constant is shown by a broken line. Yes. FIG. 4 shows the result when the load power Po = 100 W is constant. Further, the result of the AC input voltage VAC = 100 V system is indicated by a solid line, and the result of the AC input voltage VAC = 200 V system is indicated by a broken line. Show.

  In these figures, the AC → DC power conversion efficiency in the circuit of FIG. 1 is a solid line when the AC input voltage VAC = 100V (AC100V system) and the AC input voltage VAC = 230V (AC200V system), respectively. And a broken line. Incidentally, the AC-to-DC power conversion efficiency (ηAC-DC) when the load power Po = 100 W is ηAC-DC = 90.0% when the AC input voltage VAC = 100 V, and ηAC-DC when the AC input voltage VAC = 230 V. = 91.2%.

  Regarding the fluctuation range ΔEi of the DC input voltage Ei, a result is obtained that ΔEi = 24V with respect to the fluctuation of the load power Po = 100 W to 0 W when the AC input voltage VAC = 100V. Further, the fluctuation range ΔEi when the AC input voltage VAC = 230 V is ΔEi = 30 V with respect to the fluctuation of the load power Po = 100 W to 0 W.

The power factor PF is PF = 0.80 under the condition of AC input voltage VAC = 100 V and load power Po = 100 W, and PF = 0.81 under the conditions of AC input voltage VAC = 230 V and load power Po = 100 W. The result is obtained.
From these results, it can be seen that the power factor PF in this case has almost the same value for the AC100 system and the AC200V system.

Here, in the circuit shown in FIG. 1, the reason why the power factor is almost equal between the AC100 system and the AC200V system is as follows.
That is, as previously shown in FIG. 1, the power factor improving circuit 3 in this case is provided with a relay switch S2 for switching the number of turns of the tertiary winding N3. According to the above description, in response to the case where the AC input voltage VAC = AC 100V system, the terminal t2 is selected by the relay switch S2, and in response to the VAC = 200V system. , Terminal t3 is connected.
By selecting the terminal t2 corresponding to the AC 100V system, only the tertiary winding portion N3A is made effective in the tertiary winding N3. On the other hand, when the terminal t3 is selected corresponding to the AC200V system, the entire tertiary winding N3 including the tertiary winding portion N3A and the tertiary winding portion N3B is made effective.

If the number of turns of the tertiary winding N3 changes in this way, the inductance value of the tertiary winding N3 changes according to the difference in the number of turns, and accordingly, the alternating voltage level to be fed back to the rectified current path Will also change.
As the number of turns of the tertiary winding N3 increases in the AC200V system as described above, the alternating voltage level fed back to the rectified current path increases as the inductance value increases. Depending on this, since the energy fed back in the power factor correction circuit 3 increases, a higher power factor can be obtained.
That is, when the number of turns of the tertiary winding N3 is the same for the AC 100V system and the 200V system, the power factor of the AC 200V system is lower. This characteristic is improved by switching the number of turns by the relay switch S2.
The alternating voltage level fed back to the power factor correction circuit 3 can be changed by switching the number of turns of the winding N10 of the high frequency choke coil LS.

Depending on the configuration of the switching power supply circuit of the present embodiment described above, for example, the DC input voltage (rectified and smoothed) is supplied from the commercial AC power supply AC while being compatible with a wide range in the same manner as the circuits shown in FIGS. The rectifier circuit system that generates the voltage Ei) does not need to adopt a configuration for switching the rectification operation.
As a result, the configuration of the rectifier circuit can be a normal full-wave rectifier circuit as shown in FIG. 1, and therefore only one smoothing capacitor for generating the DC input voltage is required. Become.
That is, since the number of smoothing capacitors for generating the DC input voltage can be reduced in this way, the circuit manufacturing cost and the circuit board area can be reduced.

Further, if the configuration for switching the rectifying operation is not necessary as described above, the relay switch S10 for switching the rectifying operation shown in FIGS.
In this example, instead of such a relay switch S10, a relay switch S1 and a relay switch S2 for switching the number of turns of the primary winding N1 of the insulating converter transformer PIT and the number of turns of the tertiary winding N3 are provided. This switch may not be inserted in a path through which an inrush current flows into the smoothing capacitor Ci.
Depending on this, there is no need to set the contact capacity of the relay RL high as a countermeasure against the inrush current as in the relay switch S10 shown in FIGS. 10 and 11, and accordingly, inexpensive components are used as components. Therefore, the circuit cost can be reduced.

Further, as described above, since the configuration of switching the rectifying operation for wide range is not required, the flexibility of the configuration of the power factor improvement circuit 3 by the voltage feedback method can be ensured.
For example, as shown in FIG. 1, the filter capacitor CN required for power factor improvement by the voltage feedback method does not need to be inserted in parallel with the line of the commercial AC power supply AC as in the prior art. It becomes possible to reduce the breakdown voltage of the filter capacitor CN.
For example, when it is inserted between the bridge rectifier circuit Di and the smoothing capacitor Ci as in the circuit of FIG. 1, a 1 μF / 200V withstand voltage product can be selected as the filter capacitor CN, as shown in FIGS. It is possible to use an element that is cheaper than the 250V withstand voltage product included in the circuit. That is, this can also reduce the circuit manufacturing cost.

Further, even in the case of this example, it is not necessary to incorporate the high-speed recovery type diode required for power factor improvement by the voltage feedback method into the bridge rectifier circuit Di. Such a high-speed recovery type diode is designated as the diode D10 in FIG. It can be inserted relative to other parts in the rectified current path as shown.
As a result, in the fast recovery type diode D10 in this case, it is not necessary to guarantee the inrush current 30A when the power is turned on as in the case of the circuits of FIGS. 10 and 11, for example. An inexpensive element can be selected.
In the case of the circuit of FIG. 1, a 3A / 400V product is selected as such a diode D10.

  Furthermore, since it is not necessary to incorporate a high-speed recovery type diode in the bridge rectifier circuit Di as described above, it becomes possible to use a commercial product as the bridge rectifier circuit Di, which also reduces the number of components and reduces the circuit cost. Reduction can be achieved.

In the power supply circuit of the present embodiment, the relay switching circuit 5 is provided for switching the number of turns of the primary winding N1, and this relay switching circuit 5 is, for example, an instantaneous power failure of the commercial AC power supply AC. There is a possibility of malfunction due to or degradation. In particular, when the AC200V system is malfunctioned and the winding is switched so that only the primary winding N1A is effective as the primary winding N1, the AC100V system is input regardless of the AC200V system being input. Therefore, there is a possibility that an overcurrent flows through the circuit components or an overvoltage is applied.
However, as described above, the oscillation / drive circuit 2 in the present embodiment includes the protection circuit 2a. For this reason, even if the relay switching circuit 5 malfunctions, the protection operation by the protection circuit 2a works and the components of the power supply circuit are prevented from being destroyed. Therefore, even in this embodiment, it is not necessary to select a high withstand voltage product for the components of the power supply circuit. Further, it is not necessary to detect the DC input voltage on the standby power supply side in order to avoid malfunction. Furthermore, according to such a power supply circuit of the present embodiment, it can be adopted even for an electronic device that does not include a standby power supply.

Next, a configuration example of a switching power supply circuit according to the second embodiment of the present invention is shown in the circuit diagram of FIG.
The switching power supply circuit according to the second embodiment is based on the circuit configuration of FIG. 1 and improves the power factor using the voltage feedback transformer VFT in the same manner as the circuit shown in FIG. A power factor correction circuit 4 is provided.
In FIG. 5, parts already described in FIG. 1 are denoted by the same reference numerals and description thereof is omitted.

  First, in the power factor correction circuit 4 in this case, the voltage feedback transformer VFT is also formed by winding the primary winding N4 on the primary side and the secondary winding N5 on the secondary side. The primary winding N4 is configured such that a primary side switching output is obtained by connecting the primary side resonance capacitor C1 and one end thereof. Further, an alternating voltage corresponding to the switching output obtained in the primary winding N4 is excited in the secondary winding N5.

  In this case, the primary winding N4 and the secondary winding N5 of the voltage feedback transformer VFT are respectively subjected to tap output, so that the primary winding portion N4A, the primary winding N4B, the secondary winding portion N2A, Two winding portions of the secondary winding portion N2B are formed. The primary winding N4 and the secondary winding N5 are respectively provided with a relay switch S1 and a relay switch S2 for switching the number of turns.

In this case, the primary winding N4A of the primary winding N4 has one end connected to the primary side resonance capacitor C1 and the other end connected to the terminal t2 of the relay switch S1. One end of the primary winding portion N4B is connected to the terminal t2 of the relay switch S1 so as to be connected to the other end of the primary winding portion N4A. Further, the other end of the primary winding portion N4B is connected to the terminal t3 of the relay switch S1.
Terminal t1 of relay switch S1 is connected to the end of primary winding N1 of insulating converter transformer PIT that is not grounded with the primary side ground.

  One end of the secondary winding N5A is connected to the connection point between the filter capacitor CN and the smoothing capacitor Ci, and the other end is connected to the terminal t2 of the relay switch S2. One end of the secondary winding portion N5B is connected to the terminal t2 of the relay switch S2 so as to be connected to the other end of the secondary winding portion N5A, and the other end of the secondary winding portion N5B is a relay. Connected to terminal t3 of switch S2. Further, the terminal t1 of the relay switch S2 is connected to the cathode of the diode D10.

Also in this case, the relay switches S1 and S2 are configured to perform the same operation as in FIG. 1 based on the operation of the relay switching circuit 5.
That is, for the AC 100V system, the relay switch S1 and S2 both select the terminal t2, and the primary winding N4A and the secondary winding N5A are effective in the primary winding N4 and the secondary winding N5, respectively. Is obtained. Then, the terminal t3 is selected corresponding to the AC200V system, and switching is performed so that the total number of turns of the primary winding N4 and the secondary winding N5 becomes effective.
Therefore, also in this case, in response to the AC 200V system, switching is performed so that the number of turns increases in both the primary winding N4 and the secondary winding N5. In this case, as will be described later, it is considered that the winding ratio between the primary winding N4 and the secondary winding N5 does not change even if the number of turns increases.

In the power supply circuit of the second embodiment including the power factor correction circuit 4 having the above configuration, the primary side is connected to the rectification current path through the magnetic coupling of the primary winding N4 and the secondary winding N5 of the voltage feedback transformer VFT. The switching output is voltage fed back.
Also in this case, the fast recovery type diode D10 is intermittently based on the high-frequency component that is voltage-feedback in this way, and as a result, the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. In addition, the charging current to the smoothing capacitor Ci can flow. Along with this, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.

Further, in the circuit shown in FIG. 5, when the number of turns of the primary winding N4 of the voltage feedback transformer VFT is switched by the relay switch S1 as described above, the level of the leakage inductance L1 also changes accordingly. It becomes.
That is, in this case, the inductance component of the primary winding N4 connected in series with the primary winding N1 of the insulating converter transformer PIT can be equivalently viewed as the leakage inductance component of the primary winding N1 of the insulating converter transformer PIT. As the number of turns of the primary winding N4 increases, the leakage inductance L1 also increases.
Therefore, as described above, the circuit of FIG. 5 in which the number of turns of the primary winding N4 is increased corresponding to the AC 200V system and the leakage inductance L1 in the AC 200V system is increased, is the same as in the case of FIG. You can get action. That is, the required control range Δfs2 in the AC200 system as shown in FIG. 2 is set by the change in the leakage inductance L1, and accordingly, the required control range Δfs2 and the required control in the AC100V system are set. The range Δfs1 can be set within the maximum variable range of the switching frequency.
As a result, a configuration for realizing a wide range is realized by the stabilizing operation in the switching frequency control.

6 and 7 show the characteristics of the power supply circuit according to the second embodiment, such as AC → DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range ΔEi of DC input voltage Ei. Shows about.
Also in this case, FIG. 6 shows characteristics of AC → DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range ΔEi of DC input voltage Ei with respect to fluctuations in load power Po = 100 W to 0 W, and FIG. FIG. 4 shows characteristics of AC → DC power conversion efficiency (ηAC−DC), power factor PF, and fluctuation range ΔEi of DC input voltage Ei with respect to fluctuations of AC input voltage VAC = 85V to 288V.
Also in this case, for each characteristic, in FIG. 6, the result when the AC input voltage VAC = 100 V is constant is shown by a solid line, and the result when the AC input voltage VAC = 230 V is constant is shown by a broken line. In the case of FIG. 7, the characteristics for the AC 100 V system are indicated by a solid line, and the characteristics corresponding to the AC 200 V system are indicated by a broken line.

As for the circuit of FIG. 5, in order to obtain the experimental results shown in these figures, each part is selected as follows.
-Ferrite core of insulation converter transformer PIT EER-28, gap length G = 0.9 mm,
Primary winding N1 = 35T (turn)
Secondary winding N2 = 20T (10T + 10T)
-Voltage feedback transformer VFT EER-28 ferrite core, gap length G = 0.9 mm,
Primary winding N4 = Primary winding portion N4A + Primary winding portion N4B = 18T + 18T
Secondary winding N5 = secondary winding portion N5A + secondary winding portion N5B = 18T + 18T
・ Primary resonance capacitor C1 = 0.039μF
・ Filter capacitor CN = 1μF / 200V
・ Diode D10 = 3A / 400V

  As shown in these figures, in the case of the power supply circuit of the second embodiment, the AC → DC power conversion efficiency (ηAC-DC) at the load power Po = 100 W is ηAC at the AC input voltage VAC = 100V. When dc = 90.0% and AC input voltage VAC = 230V, ηAC−DC = 85.0%.

  Regarding the fluctuation range ΔEi of the DC input voltage Ei, a result is obtained that ΔEi = 19V with respect to the fluctuation of the load power Po = 100 W to 0 W when the AC input voltage VAC = 100V. Further, the fluctuation range ΔEi when the AC input voltage VAC = 230 V is ΔEi = 45 V with respect to the fluctuation of the load power Po = 100 W to 0 W.

  The power factor PF is PF = 0.83 under the condition of AC input voltage VAC = 100 V and load power Po = 100 W, and PF = 0.84 under the conditions of AC input voltage VAC = 230 V and load power Po = 100 W. The result is obtained.

In this case as well, the same power factor is obtained in the AC100V system and the AC200V system as described above because the number of turns of the primary winding N4 and the secondary winding N5 of the voltage feedback transformer VFT is switched. By doing so.
That is, in this case, by switching the number of turns of the primary winding N4 and the secondary winding N5 of the voltage feedback transformer VFT, the amount of voltage feedback in the AC200V system is increased to improve the power factor in the AC200V system. It is what you are doing.

Here, FIGS. 8 and 9 show a configuration of a modified example of the power supply circuit according to each embodiment.
In these drawings, FIG. 8 shows a modification example based on the circuit configuration of the first embodiment, and FIG. 9 shows a modification example configuration based on the circuit of the second embodiment. Shows about.
As a modification of the embodiment, not only the primary side leakage inductance L1 but also the primary side series resonance circuit itself is switched in accordance with the terminal switching of the relay switch S1.

As a configuration for this, first, in the case of FIG. 8, the primary side resonance is used as a capacitance for forming one primary side series resonance circuit between the terminal t2 of the relay switch S1 and the dividing point of the primary winding N1. It is assumed that a capacitor C1a is inserted.
Also, a primary side resonance capacitor C1b is inserted between the terminal t3 of the relay switch S1 and the end of the primary winding N1 as a capacitance for forming the other primary side series resonance circuit.
That is, according to such a configuration, when only the primary winding N1A is effective corresponding to the AC100V system, the primary side series resonance circuit by the primary winding portion N1A and the primary winding side resonance capacitor C1a is provided. It will be formed. Further, when the entire primary winding N1 becomes effective corresponding to the AC 200V system, a primary side series resonant circuit is formed by the entire primary winding N1 and the primary side resonance capacitor C1b.

In addition, in this case, the resonance frequency is set to the same value for the series resonant circuit formed in the AC 100V system and the AC 200V system as described above.
That is, in this case, since the capacitance of the primary side resonance capacitor C1 is switched together with the number of turns of the primary winding N1, the leakage inductance L1 can be switched, and C1a and C1b at this time can be switched. By setting the capacitance, the resonance frequencies of the two resonance circuits can be set to the same value.

As a result, by switching the leakage inductance L1, for example, the variation curve (see FIG. 2) for the secondary side DC output voltage Eo in the AC 200V system can be changed as in the case of FIG. In addition, in this case, since the resonance frequency of the primary side series resonance circuit can be set to the same value in the AC100V system and the AC200V system as described above, the secondary side as shown in FIG. The fluctuation curve for the DC output voltage Eo can be made to substantially coincide between the AC 100 V system and the AC 200 V system.
Since the fluctuation curves for the secondary DC output voltage Eo can be matched between the AC100V system and the AC200V system in this way, the necessary control ranges Δfs1 and Δfs2 for the AC100V system and the 200V system can be obtained. Within the maximum range of the switching frequency, substantially the same range can be achieved.
This makes it possible to set the switching frequency control range required for, for example, the AC100V system and the AC200V system to be narrower than in the case of FIG. 1, for example, a switching power supply circuit having a narrow maximum switching frequency range. There are advantages such as being more advantageous.

Similarly, in the case of FIG. 9, between the terminal t2 of the relay switch S1 and the dividing point of the primary winding N4 of the voltage feedback transformer VFT, and between the terminal t3 and the end of the primary winding N4. The primary side resonance capacitors C1a and C1b are inserted, respectively, so that another primary side series resonance circuit is formed in the AC 100 / 200V system.
By adopting such a configuration, in this case as well as in the case of FIG. 8, it is possible to set the same resonance frequency while enabling switching of the leakage inductance L1 in the AC100 / 200V system. I am doing so.
That is, in the circuit of FIG. 9 as well, the necessary control ranges Δfs1 and Δfs2 in the AC 100V system and the 200V system can be made substantially the same range within the maximum switching frequency range.

Further, in the modified examples shown in FIGS. 8 and 9, both the diode D10 is omitted and a fast recovery type diode is incorporated in the bridge rectifier circuit Di. The filter capacitor CN is inserted in parallel with the line of the commercial AC power supply AC as shown in the figure.
With such a configuration, the additional components necessary for improving the power factor can be substantially only the high-frequency choke coil LS in the case of the circuit of FIG. In the case of the circuit of FIG. 9, only the voltage feedback transformer VFT can be provided.

The present invention is not necessarily limited to the configuration as the above-described embodiment.
For example, in the circuit of FIG. 5, the number of turns of the primary winding N4 of the voltage feedback transformer VFT is switched for switching the leakage inductance L1, but the primary of the insulating converter transformer PIT is similar to the case of the circuit of FIG. The number of turns of the winding N1 may be switched.

  Further, in the configuration shown in FIG. 1, the number of turns of the primary winding N1 is switched by selecting the end of the side connected to the switching output point of the switching elements Q1, Q2 via the primary side resonance capacitor C1. On the contrary, the end of the primary winding N1 on the side to be connected to the other primary side ground is selected to switch the number of turns of the primary winding N1. I do not care. In the case of the circuit of FIG. 5, the primary winding N4 may be selected for the end portion on the primary side resonance capacitor C1 side.

  Further, as understood from the above description, the number of turns for the primary winding N1 or the primary winding N4 is set according to, for example, the stability of the secondary side DC output voltage Eo. An appropriate numerical value should be selected as appropriate according to the conversion level, load condition, and the like, and should not be limited to the value of the number of turns exemplified as the embodiment.

  In addition, an electromagnetic relay is used for switching the number of turns of the primary windings N1 and N4. However, other configurations for switching the number of turns may be adopted, for example, a switch circuit including an electronic switch may be employed.

In addition, for example, as a switching element, an IGBT (Insulated Gate Bipolar Transistor) or the like that can be used by another excitation type, the constants of each component element described above are also changed according to actual conditions. It does not matter. Further, for example, the circuit configuration for generating the secondary side DC output voltage on the secondary side of the insulating converter transformer PIT may be appropriately changed.
Furthermore, the wide-range configuration according to the present invention can be applied to a self-excited resonance type converter.

It is a circuit diagram shown about the structural example of the switching power supply circuit as 1st Embodiment in this invention. It is the figure which showed illustratively the constant voltage control characteristic about a secondary side DC output voltage by the relationship between a switching frequency and the level of a secondary side DC output voltage. It is a characteristic view about each characteristic of the fluctuation range of AC-> DC power conversion efficiency with respect to the load fluctuation of the power supply circuit of 1st Embodiment, a power factor, and DC input voltage. It is a characteristic view about each characteristic of the AC-> DC power conversion efficiency with respect to the fluctuation | variation of the alternating current input voltage of the power supply circuit of 1st Embodiment, a power factor, and the fluctuation width of a direct current input voltage. It is a circuit diagram shown about the structural example of the switching power supply circuit as 2nd Embodiment in this invention. It is a characteristic view about each characteristic of the fluctuation range of AC-> DC power conversion efficiency with respect to the load fluctuation of the power supply circuit of 2nd Embodiment, a power factor, and DC input voltage. It is a characteristic view about each characteristic of the AC-> DC power conversion efficiency with respect to the fluctuation | variation of the alternating current input voltage of the power supply circuit of 2nd Embodiment, a power factor, and the fluctuation range of a direct current input voltage. It is a circuit diagram shown about the modification of 1st Embodiment. It is a circuit diagram shown about the modification of 2nd Embodiment. FIG. 5 is a circuit diagram showing a configuration of a conventional example of a wide range compatible switching power supply circuit that improves power factor by a voltage feedback method. FIG. 10 is a circuit diagram showing another configuration of a conventional switching power supply circuit for a wide range in which a power factor is improved by a voltage feedback method.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 2a protection circuit, 3, 4 power factor correction circuit, 5 relay switching circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1, Q2 switching element, PIT isolation converter transformer, C1 primary side Resonance capacitor, Cp partial resonance capacitor, N1 primary winding, NIA, N1B primary winding, N2 secondary winding, N3 tertiary winding, RL relay, S1, S2 relay switch, D10 diode, CN filter capacitor, LS high frequency Choke coil, N10 winding, VFT voltage feedback transformer, N4 primary winding, N4A, N4B, primary winding, N5 secondary winding, N5A, N5B secondary winding

Claims (6)

Rectified and smoothed voltage generating means for generating a rectified and smoothed voltage at a level corresponding to the same size as the input commercial AC power supply;
A switching means that performs the switching operation by inputting the rectified and smoothed voltage as a direct-current input voltage, and a switching means that is formed by half-bridge coupling two switching elements;
Switching driving means for switching and driving each of the switching elements;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding to which an alternating voltage as a switching output obtained in the primary winding is excited are wound. An insulated converter transformer formed;
A primary side resonance capacitor provided so as to form a primary side resonance circuit in which the operation of the switching means is a resonance type by at least the leakage inductance component of the primary winding of the insulating converter transformer and its own capacitance; ,
DC output voltage generation means configured to input an alternating voltage obtained in the secondary winding of the insulation converter transformer and generate a secondary side DC output voltage by performing a rectification operation;
By controlling the switching drive means according to the level of the secondary side DC output voltage and variably controlling the switching frequency of the switching means, constant voltage control is performed on the secondary side DC output voltage. Constant voltage control means,
The alternating voltage based on the switching output by the switching means is fed back to the rectified current path, and the alternating voltage based on the switching output is used to intermittently rectify the rectified current component by the diode element provided in the rectifying and smoothing means. A power factor correction circuit configured to improve the rate;
Switching means configured to switch at least an inductance value forming the primary-side resonance circuit according to the level of the commercial AC power supply;
A switching power supply circuit comprising: The insulation converter transformer is
Winding the primary winding composed of two winding portions and the tertiary winding to the primary side,
The power factor correction circuit is
Using the alternating voltage according to the switching output by the switching means excited by the tertiary winding via the primary winding, the rectified current component is interrupted by the diode element provided in the rectifying and smoothing means. Configured to improve power factor,
The switching means is
By switching at least between a state where both of the two winding portions of the primary winding are effective and a state where only one of the two winding portions is effective, It is configured to switch the inductance value forming the primary side resonance circuit,
The switching power supply circuit according to claim 1. The insulating converter transformer is formed by winding a tertiary winding composed of two winding portions on the primary side,
The power factor correction circuit is
A rectified current component is generated by a diode element provided in the rectifying and smoothing means using an alternating voltage according to a switching output by the switching means, which is excited by the tertiary winding through the primary winding of the insulating converter transformer. Is configured to improve the power factor by intermittently,
It is configured to switch between a state where both of the two winding portions of the tertiary winding are effective and a state where only one of the two winding portions is effective. , Further comprising tertiary winding switching means,
The switching power supply circuit according to claim 1. The power factor correction circuit is
A voltage feedback transformer formed by a primary winding to which a switching output by the switching means is input and a secondary winding in which an alternating voltage corresponding to the switching output obtained in the primary winding is excited; Using the alternating voltage excited in the secondary winding of the voltage feedback transformer, the power factor is improved by intermittently rectifying current component by the diode element provided in the rectifying and smoothing means. With
The switching means is
Regarding the primary winding of the voltage feedback transformer composed of two winding portions, a state in which both of the two winding portions are effective and a predetermined one of the two winding portions. It is configured to switch the inductance value forming the primary side resonance circuit by performing switching in a state where only the line portion is effective,
The switching power supply circuit according to claim 1. Furthermore, a protection circuit that performs a protection operation against a state of overcurrent or overvoltage in the switching power supply circuit is provided.
The switching power supply circuit according to claim 1. The switching means is
According to the level of the commercial AC power supply, the inductance value forming the primary side resonance circuit is switched, and the capacitance of the primary side resonance capacitor is also switched at the same time.
The switching power supply circuit according to claim 1.

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