JP2005204451A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To make a switching power supply circuit compatible over a wide range for power factor improvement, and to reduce the number of circuit component parts and the manufacturing cost of the circuit. <P>SOLUTION: For example, the number of turns of the primary winding N1 of an isolation converter transformer PIT is changed, according to the level of commercial alternating-current power supply AC. Thus, the range of control required for stabilizing operation, by switching frequency control is thereby nestled within the maximum control range of switching frequency in the 100 V AC system and in the 200 V AC system. Thus, stabilization can be carried out in correspondence both with the secondary direct-current output voltage Eo in the 100 V AC system and with that in the 200 V AC system for coping with a wider range. Power factor improvement is carried out by a power regeneration (feedback) method, and the magnetic flux density of the insulating converter transformer is controlled to a required value or smaller to expand the continuous mode of secondary-side rectifying operation. Thus, the ripple component superposed on secondary direct-current output voltage, which has posed problems in a power factor improvement by a power regeneration method, is suppressed. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit used as a power supply for various electronic devices, for example.

JP-A-6-327246 (FIG. 11)

In recent years, the development of switching elements that can withstand relatively high currents and voltages at high frequencies has led to the majority of power supply circuits that rectify commercial power supplies to obtain desired DC voltages. .
The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

By the way, in general, when a commercial power source is rectified, the current flowing through the smoothing circuit becomes a distorted waveform, which causes a problem that the power factor indicating the power source utilization efficiency is impaired.
Further, there is a need for measures for suppressing harmonics generated by such a distorted current waveform.

  As the switching power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC85V to 288V is adopted so as to correspond to an AC input voltage AC100V region such as Japan and the United States and an AC200V system region such as Europe. A so-called wide-range power supply circuit that is configured so as to be able to be used is known.

Here, as the above-described resonance type converter, one configured to be stabilized by controlling the switching frequency of a switching element forming the converter (switching frequency control method) is known.
As such a resonant converter based on the switching frequency control system, for example, in a configuration in which the switching element is switched and driven by a general-purpose oscillation / drive circuit IC, for example, the variable range of the switching frequency fs is maximum, fs = 50 KHz to 250 KHz. It is about. In the case of such a variable range, for example, under a load condition where the fluctuation range of the load power Po is a relatively large fluctuation range from Po = 0 W to about 90 W, and further to about 150 W, a wide range of AC85V to 288V is used. Stabilization corresponding to the AC input voltage range is almost impossible.

  In order to solve each of these problems, a method using a so-called active filter is known as a conventional technique for realizing a power factor improvement and a wide-range configuration (see, for example, Patent Document 1).

The basic configuration of such an active filter is, for example, as shown in FIG.
In FIG. 9, a bridge rectifier circuit Di is connected to a commercial AC power source AC. An output capacitor Cout is connected in parallel to the positive / negative line of the bridge rectifier circuit Di. By supplying the rectified output of the bridge rectifier circuit Di to the output capacitor Cout, a DC voltage Vout is obtained as a voltage across the output capacitor Cout. This direct-current voltage Vout is supplied as an input voltage to a load 10 such as a subsequent DC-DC converter.

As shown in the figure, the power factor improvement includes an inductor L, a fast recovery diode D, a resistor Ri, a switching element Q, and a multiplier 11.
The inductor L and the diode D are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout.
The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectifier circuit Di and the negative terminal of the output capacitor Cout.
In this case, the switching element Q1 is a MOS-FET, and is inserted between the connection point of the inductor L and the diode D and the primary side ground as shown in the figure.

To the multiplier 11, a current detection line LI and a waveform input line Lw are connected as a feedforward circuit, and a voltage detection line LV is connected as a feedback circuit.
The multiplier 11 detects the level of the rectified current that flows from the current detection line LI and flows to the negative output terminal of the bridge rectifier circuit Di.
Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.
Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV.
The multiplier 11 outputs a drive signal for driving the switching element Q.

  A rectified current flowing through the negative output terminal of the bridge rectifier circuit Di is input from the current detection line LI to the multiplier 11. The multiplier 11 detects the rectified current level input from the current detection line LI. Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV. Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.

  The multiplier 11 first multiplies the rectified current level detected from the current detection line LI as described above by the fluctuation difference of the DC input voltage detected from the voltage detection line LV. Then, a current command value having the same waveform as the AC input voltage VAC is generated based on the multiplication result and the waveform of the AC input voltage detected from the waveform input line Lw.

  Further, the multiplier 11 in this case compares the current command value with the actual AC input current level (detected based on the input from the current detection line L1), and performs PWM control on the PWM signal according to this difference. To generate a drive signal based on the PWM signal. The switching element Q is switched by this drive signal. As a result, the AC input current is controlled to have the same waveform as the AC input voltage, and the power factor is improved so that the power factor approaches one. In this case, since the current command value generated by the multiplier is controlled so that the amplitude changes according to the fluctuation difference of the rectified smoothing voltage, the fluctuation of the rectified smoothing voltage is also suppressed. .

  FIG. 10A shows the input voltage Vin and the input current Iin input to the active filter circuit shown in FIG. The voltage Vin corresponds to the voltage waveform as the rectified output of the bridge rectifier circuit Di, and the current Iin corresponds to the current waveform as the rectified output of the bridge rectifier circuit Di. Here, the waveform of the current Iin has the same conduction angle as the rectified output voltage (voltage Vin) of the bridge rectifier circuit Di. This is the waveform of the AC input current flowing from the commercial AC power supply AC to the bridge rectifier circuit Di. Also indicates that the conduction angle is the same as that of the current Iin. That is, a power factor close to 1 is obtained.

FIG. 10B shows a change in energy (power) Pchg input / output to / from the output capacitor Cout. The output capacitor Cout stores energy when the input voltage Vin is high, and releases energy when the input voltage Vin is low to maintain the flow of output power.
FIG. 10C shows the waveform of the charge / discharge current Ichg with respect to the output capacitor Cout. The charge / discharge current Ichg flows corresponding to the energy Pchg accumulation / discharge operation in the output capacitor Cout, as can be seen from the fact that it is in phase with the waveform of the input / output energy Pchg in FIG. Current.

  Unlike the input current Vin, the charge / discharge current Ichg has substantially the same waveform as the second harmonic of the AC line voltage (commercial AC power supply AC). In the AC line voltage, a ripple voltage Vd is generated in the second harmonic component as shown in FIG. 10D due to the flow of energy with the output capacitor Cout. The ripple voltage Vd has a phase difference of 90 ° with respect to the charge / discharge current Ichg shown in FIG. The rating of the output capacitor Cout is determined in consideration of processing the second harmonic ripple current and the high frequency ripple current from the boost converter switch that modulates the current.

FIG. 11 shows a configuration example of an active filter including a basic control circuit system based on the circuit configuration of FIG. In addition, about the part made the same as FIG. 9, the same code | symbol is attached | subjected and description is abbreviate | omitted.
A switching pre-regulator 15 is provided between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout. In FIG. 9, the switching preregulator 15 is a part formed by the switching element Q, the inductor L, the diode D, and the like.

The control circuit system including the multiplier 11 includes a voltage error amplifier 12, a divider 13, and a squarer 14 in addition.
In the voltage error amplifier 12, the DC voltage Vout of the output capacitor Cout is divided by the voltage dividing resistor Rvo-Rvd and input to the non-inverting input of the operational amplifier 15. A reference voltage Vref is input to the inverting input of the operational amplifier 15. The operational amplifier 15 amplifies a voltage of a level corresponding to the error of the divided DC voltage Vout with respect to the reference voltage Vref by an amplification factor determined by the feedback resistor Rvl and the capacitor Cvl, and outputs the error output voltage Vvea to the divider 13. Output to.

  Further, a so-called feedforward voltage Vff is input to the squarer 14. The feedforward voltage Vff is an output (average input voltage) obtained by averaging the input voltage Vin by the averaging circuit 16 (Rf11, Rf12, Rf13, Cf11, Cf12). The squarer 14 squares the feedforward voltage Vff and outputs it to the divider 13.

In the divider 13, the error output voltage Vvea from the voltage error amplifier 12 is divided by the square value of the average input voltage output from the squarer 14. A signal as a result of the division is output to the multiplier 11.
That is, the voltage loop is made up of a system of a squarer 14, a divider 13, and a multiplier 11. The error output voltage Vvea output from the voltage error amplifier 12 is divided by the square of the average input voltage (Vff) before being multiplied by the rectified input signal Ivac in the multiplier 11. With this circuit, the gain of the voltage loop is kept constant without changing as the square of the average input voltage (Vff). The average input voltage (Vff) has a function of open loop correction that is sent forward in the voltage loop.

An output obtained by dividing the error output voltage Vvea by the divider 11 and a rectified output (Iac) of the positive output terminal (rectified output line) of the bridge rectifier circuit Di via the resistor Rvac are input to the multiplier 11. Here, the rectified output is shown not as voltage but as current (Iac). The multiplier 11 multiplies these inputs to generate and output a current programming signal (multiplier output signal) Imo. This corresponds to the current command value described in FIG. The output voltage Vout is controlled by varying the average amplitude of this current programming signal. That is, a PWM signal corresponding to a change in the average amplitude of the current programming signal is generated, and switching drive is performed by a drive signal based on the PWM signal, thereby controlling the level of the output voltage Vout.
Thus, the current programming signal has an average amplitude waveform that controls the input and output voltages. Note that the active filter controls not only the output voltage Vout but also the input current Vin. Since the current loop in the feedforward circuit can be said to be programmed by the rectified line voltage, the input to the subsequent converter (load 10) becomes resistive.

  FIG. 12 shows a configuration example of a power supply circuit in which a current resonance type converter is connected to the subsequent stage of the active filter based on the configuration shown in FIG. The power supply circuit shown in this figure is so-called wide range compatible with AC input voltage of both AC100V system and AC200V system. Moreover, the structure which can respond to the conditions of load electric power 0-150W is taken. Further, the current resonance type converter adopts a configuration of a separately excited half bridge coupling method.

In the power supply circuit shown in FIG. 12, two sets of line filter transformers LFT and LFT and three sets of across capacitors CL are connected to the commercial AC power supply AC in accordance with the connection mode shown in the figure. A circuit Di is connected.
Also connected to the rectified output line of the bridge rectifier circuit Di is a normal mode noise filter 4 formed by connecting one set of choke coils LN and two sets of filter capacitors (film capacitors) CN and CN as shown in the figure. Is done.

The positive output terminal of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci through a series connection of the choke coil LN, the inductor Lpc of the power choke coil PCC, and the fast recovery type rectifier diode D10. The smoothing capacitor Ci corresponds to the output capacitor Cout in FIGS. Further, the inductor Lpc and the diode D10 of the power choke coil PCC correspond to the inductor L and the diode D shown in FIG.
In addition, an RC snubber circuit including a capacitor Csn and a resistor Rsn is connected in parallel to the rectifier diode D10 in this figure.

The switching element Q6 corresponds to the switching element Q10 in FIG. That is, in actually mounting the switching element of the active filter, in this case, the switching element Q6 is connected to the connection point between the power choke coil Lpc and the fast recovery type rectifier diode D10, and the primary side ground (negative rectified output line). It is inserted between.
In this case, a MOS-FET is selected as the switching element Q6. A gate-source resistor R52 is connected between the gate and source of the switching element Q6.

In this case, the active filter control circuit 20 controls the operation of the active filter for improving the power factor so that the power factor approaches 1, and is an integrated circuit (IC) having one stone, for example.
In this case, the active filter control circuit 20 includes a multiplier, a divider, an error voltage amplifier, a PWM control circuit, a drive circuit that outputs a drive signal for switching the switching element, and the like. Circuit units corresponding to the multiplier 11, the error voltage amplifier 12, the divider 13, the squarer 14, and the like shown in FIG. 11 are mounted in the active filter control circuit 20.

  In this case, the feedback circuit is formed so that the voltage value obtained by dividing the voltage across the smoothing capacitor Ci (rectified smoothing voltage Ei) by the voltage dividing resistors R56 and R57 is input to the terminal T1 of the active filter control circuit 20. .

As the feedforward circuit, first, the rectified output is input to the terminal T3 via the resistor R58. This forms an AC input voltage waveform detection and corresponding feedforward circuit for the averaging circuit.
Further, the rectified current level is input to the terminal T6 through the resistor R60 from the connection point between the negative output terminal of the bridge rectifier circuit Di and the resistor R61 inserted between the primary side ground. That is, a feedforward circuit is formed as a line corresponding to the current detection line LI in FIG.

Further, the positive rectified output of the bridge rectifier circuit Di via the starting resistor Rs is input to the terminal T4 as the starting voltage. The active filter control circuit 20 is activated by the activation voltage input to the terminal T4 when the power supply is activated.
In the power choke coil PCC, a winding N5 that is transformer-coupled with the inductor Lpc is wound. The alternating voltage excited in the winding N5 is converted into a predetermined low-voltage DC voltage by a half-wave rectifier circuit comprising a diode D11 and a capacitor C11. This low-voltage DC voltage is also input to the terminal T4. Yes. After being activated by the activation voltage, the active filter control circuit 20 is operated by inputting this low-voltage DC voltage as a power source.
The terminal T5 is connected to the primary side ground via a resistor R59.

A drive signal for driving the switching element is output from the terminal T2. The drive signal output from the terminal T2 is output to the gate of the switching element Q6 via the resistor R51.
In the switching element Q6, a gate voltage is generated at both ends of the gate-source resistor R52 in accordance with the applied drive signal. Then, the switching operation is performed so that the gate voltage is turned on when the voltage is equal to or higher than the threshold value, and is turned off when the gate voltage is equal to or lower than the threshold value.

  The switching drive of the switching element Q6 is a drive based on PWM control so that the conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform as described with reference to FIGS. Done by signal. The conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform. That is, the conduction angle of the AC input current flowing from the commercial AC power supply AC is substantially the same as the waveform of the AC input voltage VAC. As a result, the power factor is controlled to be approximately 1. That is, power factor improvement is achieved. In practice, a characteristic of power factor PF = 0.99 to 0.98 is obtained.

Further, depending on the active filter control circuit 20 shown in FIG. 12, the average value of the rectified and smoothed voltage Ei (corresponding to Vout in FIG. 11) = 375 V is constant in the range of AC input voltage VAC = 85 V to 264 V. Also works. That is, a DC input voltage stabilized at 375 V is supplied to the subsequent stage current resonance type converter regardless of the fluctuation range of AC input voltage VAC = 85 V to 264 V.
The range of the AC input voltage VAC = 85V to 264V continuously covers the commercial AC power supply AC100V system and the 200V system. Therefore, the switching converter in the subsequent stage includes the commercial AC power supply AC100V system and the 200V system. The DC input voltage (Ei) stabilized at the same level is supplied. That is, the power supply circuit shown in FIG. 12 is configured as a wide-range power supply circuit by including an active filter.

  As shown in the figure, the current resonance converter at the latter stage of the active filter includes two switching elements Q1 and Q2. In this case, a half-bridge connection is made such that the switching element Q1 is on the high side and the switching element Q2 is on the low side, and the switching element Q1 is connected in parallel with the rectified and smoothed voltage Ei (DC input voltage). That is, a current resonance type converter by a half bridge coupling method is formed.

The current resonance type converter in this case is a separately excited type, and correspondingly, MOS-FETs are used for the switching elements Q1 and Q2. Clamp diodes DD1 and DD2 are connected in parallel to these switching elements Q1 and Q2, respectively, thereby forming a switching circuit. These clamp diodes DD1 and DD2 form a path through which a reverse current flows when the switching elements Q1 and Q2 are turned off.
The switching elements Q1 and Q2 are driven to be switched by the drive circuit 21 at a required switching frequency at the timing when they are alternately turned on / off. The drive circuit 21 variably controls the switching frequency in accordance with the level of a secondary side DC output voltage Eo described later, thereby stabilizing the secondary side DC output voltage Eo.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1, Q2 from the primary side to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the switching elements Q1 and Q2, and the other end is connected to the primary side via the series resonance capacitor C1. Connected to ground. Here, the series resonant capacitor C1 forms a series resonant circuit by its own capacitance and the leakage inductance (L1) of the primary winding N1. This series resonance circuit generates a resonance operation when the switching outputs of the switching elements Q1 and Q2 are supplied. By this, the operation of the switching circuit composed of the switching elements Q1 and Q2 is made a current resonance type.

A secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT.
In this case, the secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then comprises rectifier diodes DO1 and DO2 and a smoothing capacitor CO as shown in the figure. Both wave rectifier circuits are connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown) and is also branched and input as a detection voltage for the drive circuit 21. As described above, the drive circuit 21 switches the switching frequency so that the secondary side DC output voltage EO is stabilized based on the level of the input secondary side DC output voltage EO. Elements Q1 and Q2 are driven. That is, stabilization by the switching frequency control method is performed.

  As can be seen from the above description, the power supply circuit shown in FIG. 12 is configured by mounting the conventionally known active filter shown in FIGS. 9 and 11. By adopting such a configuration, the power factor is improved. In addition, it is compatible with a so-called wide range that operates on a commercial AC power supply AC100V system and AC200V system under a load power of 150 W or less.

However, the power supply circuit having the configuration shown in FIG. 12 has the following problems.
First, as shown in the figure, the power conversion efficiency in the power supply circuit shown in FIG. 12 is the AC-DC power conversion efficiency (ηAC → DC) corresponding to the active filter in the previous stage, and the DC of the current resonance converter in the subsequent stage. -DC power conversion efficiency (ηDC → DC).
Under the condition of the AC input voltage VAC = 100 V corresponding to the AC 100 V system, ηAC → DC = 93% and ηDC → DC = 94%, and the total efficiency is 87.4%. On the other hand, under the condition of AC input voltage VAC = 230 V corresponding to the AC 200 V system, ηAC → DC = 95%, ηDC → DC = 94%, and the total efficiency is 89.3%. That is, when the AC input voltage VAC is 230 V, the power conversion efficiency on the active filter circuit side is reduced and the overall efficiency is reduced when the AC input voltage VAC is 100 V.

In addition, since the active filter circuit is a hard switching operation, the level of noise generation is very high, and thus a relatively severe noise suppression measure is required.
For this reason, in the circuit shown in FIG. 12, a noise filter is formed by two sets of line filter transformers LFT and three sets of across capacitors for the line of the commercial AC power supply AC. That is, two or more stages of line noise filters are required.
Further, a normal mode noise filter including one set of choke coils LN and two sets of filter capacitors CN is provided for the rectified output line. Further, an RC snubber circuit is provided for the fast recovery diode D10 for rectification.
Thus, as an actual circuit, countermeasures against noise due to an extremely large number of parts are necessary, which leads to an increase in cost and an increase in the mounting area of the power supply circuit board.

  Furthermore, the switching frequency of the switching element Q6 operated by the active filter control circuit 20 as a general-purpose IC is 50 KHz, whereas the switching frequency of the subsequent current resonance type converter is in the range of 70 KHz to 150 KHz. As a result, there is also a problem that the operation as a power supply circuit tends to become unstable, such as the primary side ground potential interfering to induce abnormal oscillation.

Therefore, in the present invention, in view of the various problems described above, the switching power supply circuit is configured as follows.
That is, first, a rectifying / smoothing means for generating a rectified and smoothed voltage by inputting a commercial AC power supply, a switching means formed by including a switching element for performing switching by inputting the rectified and smoothed voltage as a DC input voltage, and the switching Switching drive means for switching and driving the element.
Then, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. An insulating converter transformer formed at least, a leakage inductance component of a primary winding of the insulating converter transformer, and a capacitance of a primary side series resonant capacitor connected in series to the primary winding, A primary-side series resonance circuit whose operation is a current resonance type.
A secondary-side DC output voltage generating means configured to generate a secondary-side DC output voltage by performing an rectifying operation by inputting an alternating voltage excited to the secondary winding of the insulating converter transformer; The constant voltage control for the secondary side DC output voltage is performed by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. Constant voltage control means.
Further, the primary side series resonance current obtained in the primary side series resonance circuit by the switching operation of the switching means is regenerated to feed back to the smoothing capacitor that forms the rectifying and smoothing means, Power factor improving means comprising a power factor improving switching element that switches the rectified current obtained by the rectifying operation by the rectifying and smoothing means intermittently according to the fed back electric power.
In addition, switching means configured to switch at least the number of turns of the primary winding in the insulating converter transformer according to the level of the commercial AC power supply is provided, and the magnetic flux density of the insulating converter transformer is set to the secondary voltage. Regardless of the fluctuation of the side DC output voltage, the secondary side rectification operation is set to be a predetermined mode or less so as to be in the continuous mode.

The switching power supply circuit of the present invention having the above configuration includes a current resonance type converter as the primary side switching converter. In addition, the power factor is improved using a power regeneration system.
Then, as described above, the magnetic flux density of the insulating converter transformer is set to be lower than the required value, so that the fluctuation of the secondary side DC output voltage, that is, the fluctuation of the load and the level fluctuation of the commercial AC power supply (AC input voltage). Regardless of the above, the secondary side rectification operation is always in a continuous mode that does not produce a period in which the secondary rectification current becomes discontinuous.
In this way, if the magnetic flux density of the insulating converter transformer is kept below the required level and the secondary side rectification operation is set to the continuous mode, it is superimposed on the primary side series resonance circuit, which was supposed to occur when the power factor is improved by the power regeneration method. The ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage, which is generated along with the ripple of the commercial AC power supply cycle, is suppressed.

In addition, according to the above configuration, at least the number of turns of the primary winding is switched according to the level of the commercial AC power supply by the operation of the switching unit. In this way, if at least the number of primary turns is switched according to the level of the commercial AC power supply, the control range of the switching frequency required for stabilization is set according to the level of the commercial AC power supply, for example. It becomes possible.
If the control range of the switching frequency required for stabilization can be set according to the level of the commercial AC power supply, the control range of the switching frequency required for stabilization can be set regardless of the rated level of the commercial AC power supply. It becomes possible to be within the variable range of the switching frequency in the switching power supply circuit. That is, this makes it possible to properly stabilize regardless of the level change of the commercial AC power supply.

  Thus, according to the present invention, the magnetic flux density of the insulating converter transformer is set to a predetermined value or less so that the secondary side rectification operation is always in the continuous mode, thereby adopting a configuration for improving the power factor by the power regeneration method. However, the ripple voltage of the secondary side DC output voltage can be suppressed. For example, the capacitance of the secondary side rectifying and smoothing capacitor for generating the secondary side DC output voltage can be within a practical range. In other words, it is possible to more easily realize and push forward the practical use of a power supply circuit having a power factor improving configuration by a power regeneration method.

  In addition, according to the present invention, since at least the number of turns of the primary winding is switched according to the level of the commercial AC power supply as described above, it is possible to properly stabilize regardless of the level change of the commercial AC power supply. This makes it possible to realize a wide-range switching power supply circuit configuration.

  For these reasons, according to the configuration of the switching power supply circuit of the present invention, it is possible to eliminate the need for a conventional active filter in realizing a wide-range configuration that improves power factor.

Thereby, the fall of power conversion efficiency in the case of aiming at a power factor improvement with the conventional active filter can be suppressed. Also, a large number of component elements for configuring the active filter can be eliminated.
Furthermore, according to the present invention, the current resonance type converter and the power factor correction means that constitute the switching power supply circuit can be set to a soft switching operation, whereby the switching noise is greatly reduced. Thus, there is no need to strengthen the noise filter.
For these reasons, the number of components is greatly reduced compared with a circuit that also uses a conventional active filter, and the size and weight of the power supply circuit can be reduced. In addition, the cost can be reduced accordingly.
Further, since the active filter is omitted, the interference of the primary side ground potential is eliminated, so that the primary side ground potential is also stabilized and the reliability is improved.

Furthermore, in the present invention, as described above, the magnetic flux density of the insulating converter transformer is set to a predetermined value or less, so that the secondary side rectification operation can always be in the continuous mode.
If the secondary side rectification operation is in the continuous mode in this way, the peak level of the secondary side rectification current can be reduced, and thereby the conduction of the secondary side rectification diode due to the discontinuous mode is achieved. Loss can be reduced. That is, since the conduction loss of the secondary side rectifier diode can be reduced in this way, the secondary side power loss that has occurred in the discontinuous mode can be reduced, and the power conversion efficiency can be further improved. Is.

Hereinafter, the best mode for carrying out the invention (hereinafter referred to as an embodiment) will be described.
FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to the first embodiment of the present invention. The power supply circuit shown in this figure employs a configuration in which a partial voltage resonance circuit is combined with a current resonance type converter using a half-bridge coupling method by a separate excitation type as a basic configuration on the primary side.

In the power supply circuit shown in FIG. 1, first, a noise filter is formed by filter capacitors CL and CL and a common mode choke coil CMC with respect to the commercial AC power supply AC.
A full-wave rectifying / smoothing circuit including a bridge rectifier circuit Di and one smoothing capacitor Ci is connected to the commercial AC power supply AC that is a subsequent stage of the noise filter. However, in the present embodiment, the power factor correction circuit 10 is provided between the positive output line of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The configuration and operation of the power factor correction circuit 10 will be described later.
When this full-wave rectifying / smoothing circuit inputs a commercial AC power supply AC and performs a full-wave rectifying operation, a rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the smoothing capacitor Ci. In this case, the rectified and smoothed voltage Ei is at a level corresponding to an equal magnification of the AC input voltage VAC. Further, a low speed recovery type is selected for the four rectifier diodes forming the bridge rectifier circuit Di.

  As shown in the figure, the current resonance type converter for switching (intermittently) by inputting the DC input voltage includes a switching circuit in which two switching elements Q1 and Q2 by MOS-FETs are connected by half bridge coupling. Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, an oscillation / drive circuit 2 is provided to drive the switching elements Q1, Q2 in a switching manner. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and for example, a general-purpose IC can be used. Then, a drive signal (gate voltage) having a required frequency is applied to the gates of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit in the oscillation / drive circuit 2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side. The primary winding N1 of the insulating transformer PIT has one end connected to the connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, and the switching output is transmitted. It is like that.
In this case, the primary winding N1 is formed by connecting two winding portions of a primary winding portion N1A and a primary winding portion N1B in series as shown. One end of the primary winding N1A is connected to the switching output point of the switching element Q1 and the switching element Q2. The other end of the primary winding N1A is connected to the terminal t2 of the relay switch S1 via the illustrated primary side series resonant capacitor C1A.
One end portion of the primary winding portion N1B is connected to the other end portion of the primary winding portion N1A. That is, the connection point between the primary winding portion N1B and the primary winding portion N1A is the tap output of the primary winding N1, and is connected to the terminal t2 of the relay switch S1 via the primary side series resonant capacitor C1A. It has become.
The other end of the primary winding N1B is connected to the terminal t3 of the relay switch S1 via the primary side series resonance capacitor C1B.

The relay switch S1 is switched so that either the terminal t2 or the terminal t3 is alternatively connected to the terminal t1, and this terminal switching operation is performed when the relay RL is turned on. / Controlled according to non-conduction. That is, the relay RL and the relay switch S1 are configured as so-called one-circuit two-contact electromagnetic relays.
The terminal t1 of the relay switch S1 is connected to a high frequency inductor L10 in the power factor correction circuit 10 via a relay switch S2 described later.

When the relay switch S1 is thus provided and the terminal t3 is connected to the terminal t1, the primary series resonance capacitor C1B and the primary winding portion N1A-N1B are connected in series. The entire winding N1 becomes effective.
On the other hand, when the terminal t2 is connected to the terminal t1, only the primary winding N1A is effective as the primary side series resonance capacitor C1A and the primary winding N1.

  Here, the insulating converter transformer PIT generates a required leakage inductance L1 in the primary winding N1 of the insulating converter transformer PIT by a structure described later. A primary side series resonant circuit is formed by the capacitance of the primary side series resonant capacitor C1 (C1A, C1B) and the leakage inductance L1. According to this, the switching outputs of the switching elements Q1, Q2 are transmitted to the primary side series resonance circuit. The primary side series resonance circuit performs a resonance operation by the transmitted switching output, thereby making the operation of the primary side switching converter a current resonance type.

  In the present embodiment, switching of the relay switch S1 enables the primary winding N1A and the primary side series resonant capacitor C1A to be effective as described above, and the entire primary winding N1 (N1A− N1B) and the primary side series resonant capacitor C1B are switched, but in either case, the primary winding N1 and the primary side series resonant capacitor C1 are connected in series. The primary side series resonant circuit is still formed. However, the number of turns as the primary winding N1 varies depending on whether the primary winding N1 is a series connection circuit of the primary winding N1A-N1B or only the primary winding N1A. Thus, the leakage inductance L1 on the primary side of the insulating converter transformer PIT also changes.

Further, depending on the switching of the relay switch S1 as described above, the series resonance circuit by the primary winding portion N1A-primary side series resonance capacitor C1A and the series resonance circuit by the primary winding portion N1A / N1B-primary side series resonance capacitor C1B. It can be seen that switching takes place.
That is, in this case, depending on the switching of the relay switch S1, it is possible to switch the leakage inductance L1 of the insulating converter transformer PIT and also to switch the primary side series resonance circuit.

  Further, on the primary side of the power supply circuit shown in FIG. 1, as a configuration for controlling switching between the relay switch S1 and a relay switch S2 described later, a relay RL and conduction / non-conduction of the relay RL are provided. There is provided a relay switching circuit 5 adapted to control.

  This relay switching circuit 5 divides the rectified and smoothed voltage Ei by a voltage dividing resistor R1 to a voltage dividing resistor R2 shown in the figure and inputs it as a detection voltage. Since the rectified and smoothed voltage Ei is obtained by rectifying and smoothing the commercial AC power source AC, a level corresponding to the same level as that of the commercial AC power source AC is obtained. Therefore, the detection voltage obtained based on the rectified and smoothed voltage Ei also corresponds to the level of the commercial AC power supply AC.

In the relay switching circuit 5, the detected voltage is within a rated level range where the level of the commercial AC power supply AC is AC100V, and within a rated level range where the level of the commercial AC power supply AC is AC200V. At some point, the relay RL is switched between conduction and non-conduction. In this way, according to the switching of the conduction / non-conduction of the relay RL, in the relay switch S1 described above and the relay switch S2 described later, the state where the terminal t1 is connected to the terminal t3, and the terminal t2 The connected state and switching are performed.
In this case, for the AC100V system, switching is performed so that the terminal t2 is selected in both the relay switches S1 and S2. Further, in response to the AC200V system, switching is performed so that the terminal t3 is selected.

A secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT. An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding N2.
The secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then a double-wave rectifier circuit comprising rectifier diodes DO1 and DO2 and a smoothing capacitor CO as shown in the figure. Is connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown). The detection voltage for the control circuit 1 is also branched and input.

The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage EO to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. That is, the level of the secondary side DC output voltage is stabilized by the switching frequency control method.
For example, when the secondary-side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. However, this reduces the resonant impedance. The secondary side DC output voltage Eo is raised. On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the resonance impedance is increased by controlling the switching frequency to be increased, and the secondary side The DC output voltage Eo is reduced.

Next, the configuration of the power factor correction circuit 10 will be described.
As described above, the power factor correction circuit 10 is provided so as to be inserted into the rectification current path in the rectification smoothing circuit for obtaining the DC input voltage (Ei) from the commercial AC power supply AC. A power factor correction circuit using electrostatic coupling is adopted as the regenerative system.

In the power factor correction circuit 10, first, one end of the high frequency inductor L10 is connected to the positive output terminal of the bridge rectifier circuit Di. The other end of the high-frequency inductor L10 is connected to the anode of the switching diode D1, which is a high-speed recovery type, via the relay switch S2, and the cathode of the switching diode D1 is connected to the positive terminal of the smoothing capacitor Ci. That is, equivalently, a series connection circuit of a high-frequency inductor L10 and a switching diode D1 (anode → cathode) is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. .
A filter capacitor CN is connected in parallel to the series connection circuit of the high-frequency inductor L10-relay switch S2-switching diode D1. The filter capacitor CN is provided to suppress normal mode noise.
Further, a power factor improving series resonance capacitor C20 is connected in parallel to the switching diode D1. In this case, the power factor improving series resonance capacitor C20 is in series with the high frequency inductor L10, but due to the capacitance of the power factor improving series resonance capacitor C20 and the inductance of the high frequency inductor L10, A series resonance circuit is formed in the power factor correction circuit 10 (in the rectified current path of the commercial AC power supply).

  As described above, the high frequency inductor L10 is connected to the relay switch S1 for switching the primary side series resonant circuit via the relay switch S2. That is, for this reason, the power factor correction circuit 10 is connected to the primary side series resonance circuit (L1-C1).

  According to such a circuit configuration of the power factor correction circuit 10, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit is regenerated as electric power, and the switching diode D1 // series resonance capacitor for power factor improvement An operation of returning to the smoothing capacitor Ci through the parallel connection of C20 is obtained. In this case, since the capacitance (capacitance) of the power factor improving series resonance capacitor C20 is interposed between the smoothing capacitor Ci and the primary side series resonance circuit, power regeneration is performed by electrostatic coupling. Can be seen.

In the power factor correction circuit 10, a voltage based on the primary series resonance current fed back in this way is obtained in the high frequency inductor L 10, whereby the switching diode D 1 that is a fast recovery type has the primary side series resonance circuit. Is interrupted by a relatively high-frequency voltage based on the switching output.
In this way, if the switching diode D1 is intermittent based on the high frequency component based on the switching output by the primary side series resonant circuit, the rectified output voltage level is originally lower than the voltage across the smoothing capacitor Ci. However, the charging current to the smoothing capacitor Ci flows.
As a result, the average waveform of the AC input current component approaches the waveform of the AC input voltage, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.
In this case, as described above, the filter capacitor CN is provided so as to be connected in parallel to the series connection circuit of the high frequency inductor L10 and the switching diode D1, so that the rectified output of the bridge rectifier circuit Di is provided. The noise component generated in the line is removed.

In this case, the power factor correction circuit 10 is provided with the relay switch S2 described above.
As this relay switch S2, a terminal t1 is connected to the terminal t1 of the relay switch S1 described above. The terminal t3 of the relay switch S2 is connected to the other end of the high frequency inductor L10. The terminal t2 is connected to a tap output point provided in the winding of the high frequency inductor L10.
Here, the winding portion from the end connected to the terminal t3 in the high-frequency inductor L10 to the tap output point is defined as a winding portion N10B. The remaining winding portion from the tap output point is the winding portion N10A.

Here, as described above, depending on the operation of the relay switching circuit 5, the terminal t2 is selected in each of the relay switches S1 and S2 corresponding to the AC 100V system. Further, the terminal t3 is selected corresponding to the AC200V system.
According to such an operation, in the high-frequency inductor L10 having the above connection configuration, switching is performed so that only the winding portion N10A is effective in the AC100V system. In the AC200V system, the series connection of the winding portion N10A and the winding portion N10B is effective, and switching is performed so that the entire winding is effective. That is, the high frequency inductor L10 is configured to increase the number of turns in the AC 200V system by such switching operation.
Note that the winding switching operation of the high-frequency inductor L10 by the relay switch S2 will be described later.

Next, the operation corresponding to the wide range of the power supply circuit having the configuration shown in FIG. 1 will be described.
First, when a rated AC 100V system is input as a commercial AC power supply, the detection voltage input to the relay switching circuit 5 is in a level range corresponding to the rated AC 100V system. In this state, as described above, the relay switching circuit 5 switches the relay RL between the conductive state and the non-conductive state so that the relay switch S1 selects the terminal t2.
Thus, when the AC100 system is input as the commercial AC power supply, only the primary winding portion N1A is effective as the primary winding N1 of the insulating converter transformer PIT.

On the other hand, when the rated AC 200V system is input as the commercial AC power supply, and the detection voltage input to the relay switching circuit 5 is in the level range corresponding to the rated AC 200V system, the relay switching is performed. In the circuit 5, the relay RL is controlled so that the relay switch S1 selects the terminal t3. Thereby, when the AC200 system is input as the commercial AC power supply, the serial connection of the primary winding portions N1B-N1A is effective as the primary winding N1 of the insulating converter transformer PIT.
That is, the number of turns (the number of turns) as the primary winding N1 of the insulating converter transformer PIT is switched between the case where the commercial AC power supply is the AC 100V system and the AC 200V system.

As a configuration in the relay switching circuit 5, as a threshold level of the detection voltage corresponding to switching between conduction / non-conduction of the relay RL, for example, in the range of the AC input voltage VAC 144V to 170V, the actual use conditions, etc. It may be set in consideration.
Although not shown, the power for driving the relay RL may be configured to be supplied from, for example, the oscillation / drive circuit 2.

FIG. 2 is a diagram illustrating the secondary-side DC output voltage Eo obtained when the configuration in which the number of turns as the primary winding N1 is switched between when the AC100V system is input and when the AC200V system is input as described above. The voltage control characteristic is exemplarily shown by the relationship between the switching frequency and the level of the secondary side DC output voltage Eo.
Here, it is assumed that the secondary side DC output voltage Eo should be stabilized at a potential ETG as shown in the figure, for example. As a switching frequency control method, the switching frequency is variably controlled in a frequency range higher than the resonance frequency fo of the primary side series resonance circuit (C1-L1), and a change in resonance impedance caused thereby is used. So-called upper side control is adopted. In this figure, the AC input voltage VAC = 100 V for the AC 100 V system, and the AC input voltage VAC = 220 V for the AC 200 V system.

Here, when the number of turns as the primary winding N1 is switched as in the power supply circuit of FIG. 1, the leakage inductance L1 on the primary side of the insulating converter transformer PIT is switched as described above. Will change.
As a method of changing the leakage inductance L1, for example, when the number of turns of the primary winding N1 increases as in the circuit of FIG. 1, the level of the secondary side DC output voltage Eo is fixed accordingly. It will change so as to be lower than the case. That is, in the circuit of FIG. 1, the level of the secondary side DC output voltage Eo at the time of AC200V system is made lower than when the number of turns is fixed.

On the other hand, the leakage inductance L1 forms the primary side series resonance circuit (C1-L1) together with the capacitance of the primary side series resonance capacitor C1, so that, for example, the capacitances of the primary side series resonance capacitors C1A and C1B have the same value. If the leakage inductance L1 changes, the resonance frequency fo of the primary side series resonance circuit (C1-L1) changes.
That is, in the circuit of FIG. 1, as can be understood from the above description, the leakage inductance L1 is set to be smaller in the AC 100V system than in the AC 200V system. When the capacitors C1A and C1B are considered to have the same value, when the resonance frequency fo1 in the AC100V system is compared with the resonance frequency fo2 in the AC200V system, the resonance frequency fo1 is more than the resonance frequency fo2 as shown in FIG. Will also be higher.

  As a general matter, the series resonance circuit has the smallest resonance impedance at the resonance frequency fo. Thereby, as the relationship between the secondary side DC output voltage Eo and the switching frequency fs, the level of the secondary side DC output voltage Eo becomes the highest when the switching frequency fs is the same as the resonance frequency fo, and is separated from the resonance frequency fo. It goes down as you go.

  That is, in FIG. 2, first, in the AC 100V system (VAC = 100V), as shown by the solid line, the secondary side DC output voltage Eo has a switching frequency fs of resonance of the primary side series resonance circuit (C1-L1). A peak at the frequency fo1 shows a quadratic curve-like change in which the level decreases as the frequency fo1 moves away from the resonance frequency fo1. In addition, the level of the secondary side DC output voltage Eo corresponding to the same switching frequency fs can be shifted so as to decrease by a predetermined amount at the maximum load power Pomax than at the minimum load power Pomin. That is, when the switching frequency fs is considered as being fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.

  When the secondary side DC output voltage Eo is to be stabilized by the potential ETG shown in the figure by the upper side control under the characteristics of the AC100V system shown by the solid line in FIG. 2 as described above, this is necessary. The switching frequency variable range (necessary control range) becomes a range indicated as Δfs1.

On the other hand, in the AC 200V system, the resonance frequency fo of the primary side series resonance circuit (C1-L1) is set to a predetermined frequency indicated by fo2 lower than fo1 in the AC 100V system as described above. The characteristic at this time is shown by a broken line in FIG. That is, in this case as well, the secondary side DC output voltage Eo is maximized when the switching frequency fs is the resonance frequency fo2, and as the switching frequency fs moves away from the resonance frequency fo2. It will decrease. Also in this case, if the switching frequency fs is fixed, the level of the secondary side DC output voltage Eo decreases as the heavy load condition is reached.
The required control range when the secondary side DC output voltage Eo is to be stabilized by the potential ETG by the upper side control under the characteristics of the AC200V system is a range shown as Δfs2 in FIG. .

In the case of the circuit of FIG. 1, the leakage on the primary winding N1 side of the insulating converter transformer PIT is changed by changing the number of turns as the primary winding N1 between the AC 100V system and the AC 200V system. The inductance L1 is changed and set. By switching the setting of the leakage inductance L1 in this way, for example, the level of the secondary side DC output voltage Eo obtained corresponding to the AC 200 V system is changed.
In this way, by changing the setting of the leakage inductance L1, the level of the secondary side DC output voltage Eo as shown in FIG. 2 is changed by changing the level of the secondary side DC output voltage Eo. Can also be changed.
And if the characteristic about the secondary side direct-current output voltage Eo is changed in this way, the necessary control range Δfs also changes accordingly.

For example, in the configuration of the power supply circuit shown in FIG. 1, when the number of turns of the primary winding N1 is not changed and fixed regardless of the commercial AC power supply level, it is proportional to the level of the commercial AC power supply AC. In this way, the level of the secondary side DC output voltage Eo is also increased.
Depending on this, it is necessary to stabilize the secondary side DC output voltage Eo having a very large fluctuation range corresponding to a predetermined load fluctuation range in a range corresponding to the AC100V system to AC200. At this time, for example, under the condition where the load fluctuation range is large, the above-mentioned required control range Δfs must be very wide, and it is properly stabilized within the maximum variable range of the switching frequency determined by the specifications of the switching power supply circuit. It may be difficult to plan.

On the other hand, in the circuit of FIG. 1, by switching the primary winding N1 as described above according to the change in the input level of the commercial AC power supply in the AC 100V system and the AC 200V system, for example, the secondary side DC in the 200V system is used. The required control range Δfs2 can be set to the position shown in FIG. 2 by changing the characteristics of the output voltage Eo. This is the primary in the AC100V system and the 200V system. By selecting the number of turns of the winding N1, it means that each of the necessary control ranges Δfs1, Δfs2 can be set so as to be within the maximum variable range of the switching frequency.
As a result, it is guaranteed that the secondary side DC output voltage Eo is stabilized corresponding to a predetermined load fluctuation range in each of the AC100V system and the AC200V system.

  In this way, in the circuit of FIG. 1, by changing the number of turns of the primary winding N1, the characteristic of the secondary side DC output voltage Eo in the AC200V system as shown in FIG. , Δfs2 can be set so as to be within the maximum variable range of the switching frequency. As a result, a wide-range configuration is realized by a configuration that is stabilized by the switching frequency control method.

  For confirmation, the experimental results shown in FIG. 2 do not show the results when actual values such as load conditions are set, for example, secondary DC The actual fluctuation curve of the output voltage Eo is not shown. That is, the result shown in FIG. 2 is an explanatory diagram for setting the required control range Δfs within the maximum variable range of the switching frequency fs by changing the leakage inductance L1 (switching the number of turns of the primary winding N1). It is only a conceptual diagram.

  Here, as described above, in the circuit of FIG. 1, according to the switching of the relay switch S1, the primary side series resonance circuit can be switched simultaneously with the switching of the number of turns of the primary winding N1 as described above. Composed. That is, the primary winding N1 and the number of turns of the primary series resonant capacitors C1A and C1B can be switched to switch the capacitance forming the primary series resonant circuit.

In this example, under such a configuration, by setting each capacitance of the primary side series resonance capacitors C1A and C1B, the resonance frequency in the resonance circuit including the primary winding N1A and the primary side series resonance capacitor C1A, and the primary The resonance frequency in the resonance circuit composed of the winding portion N1A-primary winding portion N1B (the entire primary winding N1) and the primary side series resonance capacitor C1B is set to a substantially equivalent value.
That is, by this, the resonance frequency fo1 of the primary side series resonance circuit formed by the primary winding N1A and the primary side series resonance capacitor C1A, which is formed in the AC100 system by switching the relay switch S1, and the primary winding formed in the AC200V system. The resonance frequency fo2 in the primary side series resonance circuit by the line portion N1A-primary winding portion N1B and the primary side series resonance capacitor C1B is set to be substantially the same value.

According to such a circuit configuration of FIG. 1, according to the switching of the relay switch S1, first, by switching the number of turns of the primary winding N1, as described above, for example, the secondary side in the AC 200V system The fluctuation curve (see FIG. 2) for the DC output voltage Eo can be changed. In addition, since the resonance frequency of the primary side series resonance circuit can be made substantially the same between the AC 100V system and the AC 200V system, the fluctuation curve for the secondary side DC output voltage Eo as shown in FIG. Can be substantially matched between the AC 100V system and the AC 200V system.
If the fluctuation curves for the secondary DC output voltage Eo can be matched between the AC100V system and the AC200V system in this way, the necessary control ranges Δfs1 and Δfs2 in the AC100V system and the 200V system can be switched. Within the maximum frequency range, the same range can be obtained.
This makes it possible to set the switching frequency control range required for, for example, the AC100V system and the AC200V system to be narrower than in the case of FIG. 1, for example, a switching power supply circuit having a narrow maximum switching frequency range. There are advantages such as being more advantageous.

  In this case, the above required control ranges Δfs1 and Δfs2 are guaranteed to stabilize at least in the range of AC input voltage VAC = 85V to 144V, for example, at least for the AC 100V system, and for AC 200V system, for example, the AC input voltage VAC It may be set so as to ensure stabilization in the range of = 170V to 288V.

  As can be understood from the description in FIG. 2, the switching frequency control range required for the AC100V system and the AC200V system can be reduced only by switching the primary winding N1, depending on the configuration of this example. Can be achieved. Therefore, in this example, it is possible to adopt a configuration corresponding to a wide range only by switching the primary winding N1.

By the way, in the circuit of FIG. 1 described so far, the secondary side DC output voltage is stabilized by variable control of the switching frequency of the primary side switching element. In the case of adopting such a configuration, for example, in a state where the load tends to be light, stabilization is achieved by controlling the switching frequency to be high. In this state, the secondary-side rectifier circuit operates in a so-called continuous mode in which the period during which the secondary-side rectified current flows through the secondary-side smoothing capacitor is continuous and there is no period of pause.
On the other hand, if the primary side switching frequency is controlled to decrease as the secondary side DC output voltage drops due to heavy load, the secondary side smoothing capacitor will The so-called discontinuous mode is entered, in which the rectified current stops flowing continuously and a current discontinuity period occurs. That is, there is a state in which the discontinuous mode is set according to the load variation as the secondary-side double-wave rectification operation.
The secondary side DC output voltage varies depending on the commercial AC power supply AC (AC input voltage VAC), and a constant voltage control operation corresponding to this is also performed, so that the secondary side DC output voltage also depends on the level of the AC input voltage VAC. There will be a state of discontinuous mode.

Further, as described with reference to FIG. 1, in the configuration in which the power factor is improved by the power regeneration method (including the electrostatic coupling type and the magnetic coupling method described later), the commercial AC is used as the primary side series resonance current. It is known that the ripple voltage of the power supply cycle is superimposed, the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage is significantly increased compared to the configuration without the power factor correction circuit. ing.
It is known that this is mainly caused by the secondary side rectification operation being in the discontinuous mode due to, for example, load fluctuations and fluctuations in the AC input voltage VAC as described above.

  As a countermeasure against such a ripple voltage, the capacitance of the smoothing capacitor (Co) for smoothing the DC output voltage must be increased 5 to 6 times. That is, even if the gain of the control circuit 1 is improved as much as possible, it is necessary to increase the capacitance of the smoothing capacitor 5 to 6 times in order to make it equivalent to the circuit before the power factor improvement, which greatly increases the cost. The practical application becomes unrealistic. Since such a measure is not realistic, when a power factor correction circuit based on a power regeneration method is simply provided, for example, it is strict that the ripple voltage is kept below a certain level, and it is very difficult to adopt it in a required device. It is difficult.

  Therefore, as the present embodiment, the ripple voltage of the secondary side DC output voltage Eo as described above is suppressed, and a power factor correction circuit using a power regeneration system can be practically used. For this reason, the circuit of FIG. 1 prevents the discontinuous mode described above, which is the main cause of the ripple voltage, and regardless of load fluctuations and fluctuations in the AC input voltage VAC. It is assumed that the continuous mode is maintained as the secondary side rectification operation.

FIG. 3 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 (primary winding portion N1A, primary winding portion N1B) is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained. In this case, for example, EER-35 is selected as the actual EE type core.

  In addition, a gap G having a gap length of about 1.6 mm is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with k = 0.8 or less is obtained. Note that k = 0.76 was set as the actual coupling coefficient k. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  Further, the number of turns (number of turns) of the primary winding N1 and the secondary winding N2 is set so that the induced voltage level per 1T (turn) of the secondary winding is lower than required. For example, by setting primary winding N1 = primary winding portion N1A + primary winding portion N1B = 48T + 32T = 80T and secondary winding N2 = 10T + 10T = 20T, induced voltage per 1T (turn) of the secondary winding The level is 2.5 V / T or less.

By setting the structure of the insulating converter transformer PIT and the number of turns of the primary winding N1 and the secondary winding N2, the leakage inductance of the insulating converter transformer PIT in this case tends to increase, and the magnetic flux density in the core Decreases below the required level.
If the magnetic flux density generated in the core of the insulating converter transformer PIT can be reduced to a required level or less in this way, the electric power excited on the secondary side also changes, thereby causing a heavy load and a low AC input voltage. Even in this condition, the secondary side rectification operation can be set to the continuous mode.

  In this way, the continuous mode is obtained even under heavy load and low AC input voltage states, regardless of changes in the secondary side DC output voltage Eo due to load fluctuations, AC input voltage fluctuations, etc. This means that the secondary side rectification operation is performed in the continuous mode. As a result, in the case of adopting the configuration including the power factor correction circuit based on the power regeneration system as in the present embodiment, it is assumed that the secondary side DC output voltage Eo is superposed by the ripple superposed on the primary side series resonance current. In addition, an increase in the ripple of the commercial AC power supply cycle is greatly suppressed.

  As a result, the level of the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage also decreases, and it is not necessary to increase the capacitance of the secondary side smoothing capacitor. That is, the practical use of a switching power supply circuit including a power factor correction circuit based on a power regeneration system can be easily realized. For example, in this embodiment, the increase in the capacitance of the secondary-side smoothing capacitor (Co) can be suppressed to about twice that in the case where no power factor correction circuit is provided.

  Note that the ripple of the commercial AC power supply cycle, which was supposed to be superimposed on the secondary side DC output voltage Eo in this way, can be suppressed because the secondary side rectification operation becomes a continuous mode as described above. This is because the peak level of the secondary side rectified current is suppressed. In addition, as described with reference to FIG. 3 above, the magnetic flux density of the insulating converter transformer is reduced to a required level or less, so that the power transmission from the primary side to the secondary side also changes, and the secondary side accordingly. This is because the influence of the ripple on the commercial AC power supply cycle generated in the primary side series resonance current is reduced.

4 and 5 show the characteristics of the power supply circuit of the first embodiment shown in FIG. 1 as AC to DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range of DC input voltage Ei. Each characteristic of ΔEi will be described.
4, AC → DC power conversion efficiency (ηAC-DC), power factor PF, DC input voltage with respect to fluctuations in load power Po = 100 W to 0 W (Eo = 25 V × (4 A to 0 A)) in the circuit of FIG. Each characteristic of the fluctuation range ΔEi of Ei is shown. In FIG. 5, the AC → DC power conversion efficiency (ηAC-DC), the power factor PF, and the fluctuation range ΔEi of the DC input voltage Ei with respect to the fluctuation of the AC input voltage VAC = 85V to 288V. Each characteristic is shown.
In these figures, in FIG. 4, for each characteristic, the result when the AC input voltage VAC = 100 V is constant is shown by a solid line, and the result when the AC input voltage VAC = 230 V is constant is shown by a broken line. Yes. FIG. 5 shows the result when the load power Po is constant at 100 W, and the result of the AC input voltage VAC = 100 V system is shown by a solid line, and the result of the AC input voltage VAC = 200 V system is shown by a broken line. Show.

Further, in obtaining the experimental results shown in these figures, each part in the circuit of FIG. 1 was selected as follows.
Insulating converter transformer PIT: EER-35 ferrite core, gap G = 1.6 mm, primary winding N1 = primary winding portion N1A + primary winding portion N1B = 48T + 32T = 80T, secondary winding N2 = 10T + 10T = 20T, coupling coefficient k = 0.76
Primary side series resonant capacitor C1A = 0.033μF
Primary side series resonant capacitor C1B = 0.022μF
Winding portion N10A = 47 μH, winding portion N10B = 147 μH of high frequency inductor L10
Series resonant capacitor for power factor improvement C20 = 0.039μF

In these figures, the AC → DC power conversion efficiency in the circuit of FIG. 1 is a solid line when the AC input voltage VAC = 100V (AC100V system) and the AC input voltage VAC = 230V (AC200V system), respectively. And a broken line. According to the experiment, the AC to DC power conversion efficiency (ηAC-DC) when the load power Po = 100 W is ηAC-DC = 90.0% when the AC input voltage VAC = 100 V, and when the AC input voltage VAC = 230 V. ηAC-DC = 92.9%.
This is an improvement of 2.6% in the case of the circuit of FIG. 12 compared to the overall efficiency at VAC = 100 V = 87.4%, and the overall efficiency at the time of VAC = 230 V in the case of the circuit of FIG. Compared to 89.3%, this is an improvement of 3.6%. In this case, even in the range of VAC = 85V to 264V, an improvement of about 2 to 3% is achieved as compared with the case of FIG.

  Regarding the fluctuation range ΔEi of the DC input voltage Ei, a result is obtained that ΔEi = 4.5V with respect to the fluctuation of the load power Po = 100 W to 0 W when the AC input voltage VAC = 100V. Further, the fluctuation range ΔEi when the AC input voltage VAC = 230 V is similarly ΔEi = 3 V with respect to the fluctuation of the load power Po = 100 W to 0 W.

The power factor PF is PF = 0.80 under the condition of AC input voltage VAC = 100 V and load power Po = 100 W, and PF = 0.83 under the conditions of AC input voltage VAC = 230 V and load power Po = 100 W. The result is obtained.
From these results, it can be seen that the power factor PF in this case has almost the same value for the AC100 system and the AC200V system.

Here, in the circuit shown in FIG. 1, the almost same power factor can be obtained in the AC100 system and the AC200V system in the following manner.
In other words, as shown in FIG. 1, the power factor correction circuit 10 in this case is provided with a relay switch S2 for switching the number of turns of the high-frequency inductor L10. According to the above description, only the winding part N10A in the high-frequency inductor L10 is effective in correspondence with the case where the AC input voltage VAC = AC100V system. On the other hand, for the AC200V system, the entire winding including winding portion N10A and winding portion N10B is made effective.

If the number of windings of the high frequency inductor L10 changes in this way, the inductance value of the high frequency inductor L10 changes according to the difference in the number of turns, and accordingly, the potential obtained by the power factor correction circuit 10 also changes. It will be. That is, the power level to be fed back to the rectified current path for power factor improvement changes.
As the number of turns of the high-frequency inductor L10 increases in the AC200V system as described above, the power level fed back to the rectified current path increases as the inductance value increases. Depending on this, since the energy returned in the power factor correction circuit 10 increases, a higher power factor can be obtained.

Here, according to the circuit configuration of FIG. 1, in the AC200V system, the number of turns of the primary winding N1 of the insulating converter transformer PIT is increased by the relay switch S1. When the number of turns is equal, the power level fed back from the primary side series resonance circuit in the AC200V system is lowered. From this, the power factor is lower in the AC200V system than in the AC100V system.
Therefore, as described above, this characteristic is improved by increasing the inductance value of the high-frequency inductor L10 in the AC200V system and increasing the feedback energy.

As described above, depending on the configuration of the switching power supply circuit of the first embodiment shown in FIG. 1, the structure of the insulating converter transformer PIT described above and the windings of the primary winding N1 and the secondary winding N2 are described. By setting the number of lines, it is possible to suppress the ripple of the commercial AC power supply cycle superimposed on the secondary side DC voltage Eo, which has been a cause of hindering the practical use of the switching power supply circuit including the power factor correction circuit based on the power regeneration method.
Thereby, the configuration for power factor improvement by the power regeneration method can be realized at a practical level.

  In addition, in the present embodiment, the primary side series resonant capacitors C1A and C1B are switched together with the number of turns of the primary winding N1 of the insulating converter transformer PIT in accordance with the rated level of the commercial AC power supply. Thus, each of the necessary control ranges Δfs1 and Δfs2 in the AC 100V system and the AC 200V system can be set so as to be within the maximum variable range of the switching frequency. As a result, a wide-range configuration can be realized by stabilizing the switching frequency control method.

Here, when the power supply circuit of this embodiment is compared with the power supply circuit provided with the active filter shown in FIG. 12 as a prior art which similarly aims for wide range compatibility with improvement of the power factor, The following can be said.
First, as is clear from the experimental results shown in FIGS. 4 and 5, the power conversion efficiency is improved by the circuit shown in FIG. 1 as compared with the circuit shown in FIG.
This is a configuration including the power factor improvement circuit 10 based on the power regeneration method and a configuration corresponding to a wide range for stabilization by the switching frequency control method, so that an active filter can be eliminated. It depends.
Further, as described in FIG. 3 above, the peak density of the secondary side rectified current is increased by reducing the magnetic flux density of the insulating converter transformer PIT to a required level and expanding the continuous mode of the secondary side rectification operation. It depends on being suppressed. That is, by suppressing the peak level of the secondary side rectified current in this way, the conduction loss in the secondary side rectifier diodes (DO1, DO2) is reduced, and the power loss is reduced accordingly. .

In the circuit of this example shown in FIG. 1, the number of circuit components can be reduced by eliminating the need for an active filter.
In other words, the active filter constitutes a set of converters. As can be seen from the description with reference to FIG. 12, in practice, there are many switching elements and ICs for driving them, and so on. Consists of the number of parts.
On the other hand, in the power supply circuit shown in FIG. 1, relay switches (S1, S2), a high-frequency inductor L10, a filter capacitor CN, a switching diode D1, a power factor are added as additional components necessary for power factor improvement and wide range compatibility. Since only the improvement series resonance capacitor C20 and the relay switching circuit 5 are provided, the number of parts is very small as compared with the active circuit.
As a result, the power supply circuit shown in FIG. 1 can be manufactured at a much lower cost than the circuit shown in FIG. 12 as a wide-range power supply circuit having a power factor correction function. Further, since the number of parts is greatly reduced, the circuit board can be effectively reduced in size and weight.

In the power supply circuit shown in FIG. 1, since the operation of the resonant converter and the power factor correction circuit 10 is a so-called soft switching operation, the level of switching noise is greatly reduced as compared with the active filter shown in FIG. Is done.
For this reason, as shown in FIG. 1, if a one-stage noise filter comprising a pair of common mode choke coils CMC and an across capacitor CL is provided, it is possible to sufficiently satisfy the power disturbance standard. The Further, as shown in FIG. 1, the normal mode noise of the rectified output line is taken by only one filter capacitor CN.
By reducing the number of components as a noise filter in this way, the cost reduction of the power supply circuit and the reduction in size and weight of the circuit board are promoted.

Further, in the case of the power supply circuit shown in FIG. 1, the switching frequency of the primary side switching converter varies in the range of, for example, 70 KHz to 150 KHz in order to make the voltage constant according to changes in the AC input voltage VAC and load power. However, the switching elements Q1 and Q2 forming the switching converter perform a switching operation in synchronization. Accordingly, as the primary side ground potential, as in the power supply circuit of FIG. 12, there is no interference between the active filter side and the subsequent switching converter, and the primary side ground potential can be stabilized regardless of the change of the switching frequency. it can.
Thereby, it is possible to suppress abnormal oscillation, which has been a problem in the circuit of FIG. 12 having the conventional active filter.

  From these comparisons, in the circuit of this example shown in FIG. 1, after solving various problems of the circuit of FIG. 12 having the active filter, it is practically sufficient, equivalent to the case of having the active filter. A high power factor can be obtained, and higher power conversion efficiency can be obtained.

Next, the circuit diagram of FIG. 6 shows a configuration example of the switching power supply circuit as the second embodiment of the present invention. In FIG. 6, parts already described in FIG. 1 are denoted by the same reference numerals and description thereof is omitted.
The switching power supply circuit according to the second embodiment includes a power coupling type power factor correction circuit 11 instead of the electrostatic coupling type power factor correction circuit 10 as a power regeneration type power factor correction circuit.
This magnetic coupling type power factor correction circuit 11 is also inserted into a rectification current path between a bridge rectification circuit Di that generates a rectification smoothing voltage Ei (DC input voltage) from a commercial AC power supply AC and a smoothing capacitor Ci. And comprises a filter capacitor CN, a high-frequency inductor L10, and a fast recovery type switching diode D1. When this point is compared with the power factor improving circuit 10 shown in FIG. 1, the power factor improving series resonance capacitor C20 is omitted, and the number of parts is reduced accordingly.

In the power factor correction circuit 11, first, the anode of the switching diode D1 is connected to the positive output terminal of the bridge rectifier circuit Di, and the cathode is connected to the terminal t1 of the relay switch S1 and the terminal t1 of the relay switch S2. Connected to the connection point.
Also in this case, the relay switch S2 is provided for switching the winding of the high-frequency inductor L10. Similarly to the case of FIG. 1, the terminal t2 is connected to the tap output point of the high-frequency inductor L10, and the terminal t3 is high-frequency. It is connected to one end of the inductor L10.
The end of the high-frequency inductor L10 on the side where the terminal t3 is not connected is connected to the positive terminal of the smoothing capacitor Ci. That is, in this case, equivalently, a series connection circuit of the switching diode D1 (anode → cathode) and the high-frequency inductor L10 is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. become. A filter capacitor CN is connected in parallel to the series connection circuit of the switching diode D1-relay switch S2-high frequency inductor L10. The filter capacitor CN in this case also serves to suppress normal mode noise.

  Also in this case, the primary side series resonant circuit (L1-C1) is connected to the power factor correction circuit 11 by connecting the terminal t1 of the relay switch S1 and the terminal t1 of the relay switch S2 as described above. As a result, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit is regenerated as power and fed back to the smoothing capacitor Ci via the magnetic coupling (ie, the high frequency inductor L10). To be.

  Also in the power factor correction circuit 11 having the above configuration, a voltage corresponding to the primary side series resonance current fed back from the primary side series resonance circuit is obtained in the high frequency inductor L10. Thereby, the switching diode D1 is intermittently interrupted by the high frequency component based on the switching output by the primary side series resonant circuit, and the power factor is improved in the same manner as in the circuit of FIG.

Also in the power supply circuit of the second embodiment having such a circuit configuration, the insulating converter transformer PIT adopts the same structure as that of the first embodiment described above, so that the coupling coefficient k For example, k is reduced to about 0.76, and a continuous mode is always obtained as a secondary side rectification operation regardless of load fluctuations, AC input voltage fluctuations, and the like.
As a result, in the same manner as the power supply circuit of the first embodiment, an increase in the ripple voltage superimposed on the secondary side DC output voltage Eo is suppressed, and a magnetic coupling type configuration is used as a power regeneration system. Even if it is assumed, the practical use is easily realized.

7 and 8 show the characteristics of the power supply circuit according to the second embodiment, such as AC → DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range ΔEi of DC input voltage Ei. Shows about.
Also in this case, in FIG. 7, the AC → DC power conversion efficiency (ηAC-DC), the power factor PF, and the DC input voltage Ei with respect to the variation of the load power Po = 100W to 0W (Eo = 25V × (4A to 0A)). Each characteristic of the fluctuation range ΔEi is shown. In FIG. 8, each of the AC → DC power conversion efficiency (ηAC-DC), the power factor PF, and the fluctuation range ΔEi of the DC input voltage Ei with respect to the fluctuation of the AC input voltage VAC = 85V to 288V. It shows the characteristics.
Also in this case, for each characteristic, in FIG. 7, the result when the AC input voltage VAC = 100V is constant is shown by a solid line, and the result when the AC input voltage VAC = 230V is constant is shown by a broken line. In the case of FIG. 8, the characteristics for the AC 100V system are indicated by a solid line, and the characteristics corresponding to the AC 200V system are indicated by a broken line.

Note that each part of the circuit shown in FIG. 6 was selected as follows in order to obtain the experimental results shown in these figures.
Insulating converter transformer PIT: EER-35 ferrite core, gap G = 1.6 mm, primary winding N1 = primary winding portion N1A + primary winding portion N1B = 48T + 32T = 80T, secondary winding N2 = 10T + 10T = 20T, coupling coefficient k = 0.76
Primary side series resonant capacitor C1A = 0.033μF
Primary side series resonant capacitor C1B = 0.015 μF
Winding portion N10A = 47 μH, winding portion N10B = 147 μH of high frequency inductor L10
Filter capacitor CN = 1μF / 200V

As shown in these figures, in the case of the power supply circuit of the second embodiment, the AC → DC power conversion efficiency (ηAC-DC) at the load power Po = 100 W is ηAC at the AC input voltage VAC = 100V. When dc = 90.0% and AC input voltage VAC = 230V, ηAC−DC = 92.5%.
From these results, it can be understood that the circuit shown in FIG. 6 can obtain higher efficiency than the case where the active filter shown in FIG. 12 is provided. That is, an improvement of 2.6% is achieved when VAC = 100V, an improvement of 3.2% is achieved when VAC = 230V, and an improvement of 2-3% is achieved even in the range of VAC = 85V to 264V.

  Regarding the fluctuation range ΔEi of the DC input voltage Ei, a result is obtained that ΔEi = 17.5V with respect to the fluctuation of the load power Po = 100 W to 0 W when the AC input voltage VAC = 100V. Further, the fluctuation range ΔEi when the AC input voltage VAC = 230 V is similarly ΔEi = 25 V with respect to the fluctuation of the load power Po = 100 W to 0 W.

As for the power factor PF, PF = 0.83 under the condition of AC input voltage VAC = 100 V and load power Po = 100 W, and PF = 0.0.3 under the conditions of AC input voltage VAC = 230 V and load power Po = 100 W. A result of 83 is obtained.
Also in this case, the same power factor is obtained in the AC100V system and the AC200V system as described above because the number of turns of the high-frequency inductor L10 is switched. That is, by switching the inductance of the high frequency inductor L10, the feedback energy in the AC 200V system is increased, and the power factor in the AC 200V system is improved.

  From these results of FIG. 7 and FIG. 8, depending on the second embodiment, as a configuration corresponding to a wide range for improving the power factor, a practically sufficient force as in the case of the first embodiment is obtained. It can be understood that the power conversion efficiency can be higher than that of the circuit shown in FIG.

The present invention is not necessarily limited to the configuration as the above-described embodiment.
For example, as a switching element, an element other than a MOS-FET may be employed as long as it is an element that can be used in an separately excited type, such as an IGBT (Insulated Gate Bipolar Transistor). Also, the constants of the component elements described above may be changed according to actual conditions. Further, for example, the circuit configuration for generating the secondary side DC output voltage on the secondary side of the insulating converter transformer PIT may be appropriately changed.
Furthermore, the wide-range configuration according to the present invention can be applied to a self-excited current resonance converter.

  As can be understood from the above description, the number of turns for switching the primary winding N1 can be determined by, for example, the stabilization level of the secondary side DC output voltage Eo or the load condition. An appropriate numerical value should be selected according to the above, and should not be limited to the value of the number of turns exemplified in the embodiment.

Further, although the electromagnetic relay is used for switching the number of turns of the primary winding N1, other configurations for switching the number of turns may be employed such as adopting a switch circuit including an electronic switch or the like.
In addition, as for the insulating converter transformer PIT, for example, the core type and the like, the structure may be appropriately changed as long as the magnetic flux density is lower than required.

  Furthermore, the power factor correction circuit is not limited to the one shown in the embodiment, and circuit configurations based on various power regeneration (feedback) methods proposed by the present applicant may be adopted. Is possible.

It is a circuit diagram shown about the structural example of the switching power supply circuit as 1st Embodiment in this invention. It is the figure which showed illustratively the constant voltage control characteristic about a secondary side DC output voltage by the relationship between a switching frequency and the level of a secondary side DC output voltage. It is sectional drawing which shows the structural example of an insulating converter transformer. It is a characteristic view about each characteristic of the fluctuation range of AC-> DC power conversion efficiency with respect to the load fluctuation of the power supply circuit of 1st Embodiment, a power factor, and DC input voltage. It is a characteristic view about each characteristic of the AC-> DC power conversion efficiency with respect to the fluctuation | variation of the alternating current input voltage of the power supply circuit of 1st Embodiment, a power factor, and the fluctuation width of a direct current input voltage. It is a circuit diagram shown about the structural example of the switching power supply circuit as 2nd Embodiment in this invention. It is a characteristic view about each characteristic of the fluctuation range of AC-> DC power conversion efficiency with respect to the load fluctuation of the power supply circuit of 2nd Embodiment, a power factor, and DC input voltage. It is a characteristic view about each characteristic of the AC-> DC power conversion efficiency with respect to the fluctuation | variation of the alternating current input voltage of the power supply circuit of 2nd Embodiment, a power factor, and the fluctuation range of a direct current input voltage. It is a circuit diagram which shows the basic circuit structure of an active filter. It is a wave form diagram which shows the operation | movement in the active filter shown in FIG. It is a circuit diagram which shows the structure of the control circuit system of an active filter. It is a circuit diagram which shows the structural example of the power supply circuit which mounted the active filter as a prior art.

Explanation of symbols

  1 control circuit, 2 oscillation / drive circuit, 5 relay switching circuit, 10, 11 power factor correction circuit, Di bridge rectifier circuit, Ci smoothing capacitor, Q1, Q2 switching element, PIT isolation converter transformer, C1A, C1B primary side series resonance Capacitor, Cp partial resonance capacitor, N1 primary winding, NIA, N1B primary winding, N2 secondary winding, RL relay, S1, S2 relay switch, CN filter capacitor, L10 high frequency inductor, series resonance for C20 power factor improvement Capacitor, D1 switching diode

Claims (5)

Rectifying and smoothing means for inputting a commercial AC power supply and generating a rectified and smoothed voltage;
Switching means formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An isolated converter transformer,
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage;
The switching drive means is controlled according to the level of the secondary side DC output voltage, and the switching frequency of the switching means is varied to perform constant voltage control on the secondary side DC output voltage. Constant voltage control means;
The primary side series resonance current obtained in the primary side series resonance circuit by the switching operation of the switching means is used to regenerate power, and is fed back to the smoothing capacitor forming the rectifying and smoothing means. Power factor improving means configured to include a power factor improving switching element that switches the rectified current obtained by the rectifying operation by the rectifying and smoothing means intermittently according to electric power;
Switching means configured to switch at least the number of turns of the primary winding in the insulating converter transformer according to the level of the commercial AC power supply,
The magnetic flux density of the insulating converter transformer is set to be equal to or less than a predetermined value so that the secondary side rectification operation becomes a continuous mode regardless of the fluctuation of the secondary side DC output voltage.
A switching power supply circuit. The primary winding in the insulating converter transformer is composed of two winding portions,
The switching means is
It is configured to switch between a state where both of the two winding portions of the primary winding are effective and a state where only one of the two winding portions is effective. Yes,
The switching power supply circuit according to claim 1. The switching means is
According to the level of the commercial AC power supply, the number of turns of the primary winding is switched, and the capacitance of the primary side series resonant capacitor forming the primary side series resonant circuit is also switched simultaneously.
The switching power supply circuit according to claim 1. The power factor improving means is
A high-frequency inductor inserted in series between a rectification output terminal of a rectification circuit for rectifying a commercial AC power supply and a positive electrode terminal of a smoothing capacitor in the rectifying and smoothing means; and a diode element serving as the power factor improving switching element; A series connection circuit in which
A power factor improving series resonance capacitor connected in parallel to the diode element and forming a series resonance circuit with the high frequency inductor;
A filter capacitor connected in parallel to the series connection circuit,
Formed by connecting the primary side series resonant circuit to the connection point between the high frequency inductor and the diode element in the series connection circuit,
The switching power supply circuit according to claim 1. The power factor improving means is
A high-frequency inductor inserted in series between a rectification output terminal of a rectification circuit for rectifying a commercial AC power supply and a positive electrode terminal of a smoothing capacitor in the rectifying and smoothing means; and a diode element serving as the power factor improving switching element; A series connection circuit in which
A filter capacitor connected in parallel to the series connection circuit,
Formed by connecting the primary side series resonant circuit to the connection point between the high frequency inductor and the diode element in the series connection circuit,
The switching power supply circuit according to claim 1.

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