JP2005210799A - Switching power supply circuit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To cut down on the cost of a circuit component and the manufacturing cost of a circuit by using a switching power supply circuit improved in power factor and adapted to a wide range . <P>SOLUTION: According to a commercial AC power source level, a constitution adapted to a wide range is achieved by switching a rectifying operation for the generation of a DC input voltage to a voltage equal doubler rectifying operation and a voltage doubler rectifying operation. The improvement of the power factor is performed by a power regeneration (return) method, and the continuous mode of a secondary-side rectifying operation is enlarged by setting the magnetic flux density of an insulating converter transformer not higher than a prescribed value, whereby the generation of a ripple component superposed to a secondary-side DC output voltage and having caused a trouble in the power factor improvement by the power regeneration method can be suppressed. A conventional active filter can be dispensed with, the number of the components is thereby largely reduced, and the cost reduction can be achieved. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a switching power supply circuit used as a power supply for various electronic devices, for example.

JP-A-6-327246 (FIG. 11)

In recent years, the development of switching elements that can withstand relatively high currents and voltages at high frequencies has led to the majority of power supply circuits that rectify commercial power supplies to obtain desired DC voltages. .
The switching power supply circuit reduces the size of the transformer and other devices by increasing the switching frequency, and is used as a power source for various electronic devices as a high-power DC-DC converter.

By the way, in general, when a commercial power source is rectified, the current flowing through the smoothing circuit becomes a distorted waveform, which causes a problem that the power factor indicating the power source utilization efficiency is impaired.
Further, there is a need for measures for suppressing harmonics generated by such a distorted current waveform.

  As the switching power supply circuit, for example, an operation corresponding to an AC input voltage range of about AC85V to 288V is adopted so as to correspond to an AC input voltage AC100V region such as Japan and the United States and an AC200V system region such as Europe. A so-called wide-range power supply circuit that is configured so as to be able to be used is known.

Here, as the above-described resonance type converter, one configured to be stabilized by controlling the switching frequency of a switching element forming the converter (switching frequency control method) is known.
As such a resonant converter based on the switching frequency control system, for example, in a configuration in which the switching element is switched and driven by a general-purpose oscillation / drive circuit IC, for example, the variable range of the switching frequency fs is maximum, fs = 50 KHz to 250 KHz. It is about. In the case of such a variable range, for example, under a load condition where the fluctuation range of the load power Po is a relatively large fluctuation range from Po = 0 W to about 90 W, and further to about 150 W, a wide range of AC85V to 288V is used. Stabilization corresponding to the AC input voltage range is almost impossible.

  In order to solve each of these problems, a method using a so-called active filter is known as a conventional technique for realizing a power factor improvement and a wide-range configuration (see, for example, Patent Document 1).

The basic configuration of such an active filter is, for example, as shown in FIG.
In FIG. 8, a bridge rectifier circuit Di is connected to a commercial AC power source AC. An output capacitor Cout is connected in parallel to the positive / negative line of the bridge rectifier circuit Di. By supplying the rectified output of the bridge rectifier circuit Di to the output capacitor Cout, a DC voltage Vout is obtained as a voltage across the output capacitor Cout. This direct-current voltage Vout is supplied as an input voltage to a load 10 such as a subsequent DC-DC converter.

As shown in the figure, the power factor improvement includes an inductor L, a fast recovery diode D, a resistor Ri, a switching element Q, and a multiplier 11.
The inductor L and the diode D are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout.
The resistor Ri is inserted between the negative output terminal (primary side ground) of the bridge rectifier circuit Di and the negative terminal of the output capacitor Cout.
In this case, the switching element Q1 is a MOS-FET, and is inserted between the connection point of the inductor L and the diode D and the primary side ground as shown in the figure.

To the multiplier 11, a current detection line LI and a waveform input line Lw are connected as a feedforward circuit, and a voltage detection line LV is connected as a feedback circuit.
The multiplier 11 detects the level of the rectified current that flows from the current detection line LI and flows to the negative output terminal of the bridge rectifier circuit Di.
Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.
Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV.
The multiplier 11 outputs a drive signal for driving the switching element Q.

  A rectified current flowing through the negative output terminal of the bridge rectifier circuit Di is input from the current detection line LI to the multiplier 11. The multiplier 11 detects the rectified current level input from the current detection line LI. Further, the fluctuation difference of the DC input voltage is detected based on the DC voltage Vout of the output capacitor Cout input from the voltage detection line LV. Further, the rectified voltage waveform at the positive output terminal of the bridge rectifier circuit Di input from the waveform input line Lw is detected. This corresponds to detecting the waveform of the commercial AC power supply AC (AC input voltage) as an absolute value.

  The multiplier 11 first multiplies the rectified current level detected from the current detection line LI as described above by the fluctuation difference of the DC input voltage detected from the voltage detection line LV. Then, a current command value having the same waveform as the AC input voltage VAC is generated based on the multiplication result and the waveform of the AC input voltage detected from the waveform input line Lw.

  Further, the multiplier 11 in this case compares the current command value with the actual AC input current level (detected based on the input from the current detection line L1), and performs PWM control on the PWM signal according to this difference. To generate a drive signal based on the PWM signal. The switching element Q is switched by this drive signal. As a result, the AC input current is controlled to have the same waveform as the AC input voltage, and the power factor is improved so that the power factor approaches one. In this case, since the current command value generated by the multiplier is controlled so that the amplitude changes according to the fluctuation difference of the rectified smoothing voltage, the fluctuation of the rectified smoothing voltage is also suppressed. .

  FIG. 9A shows the input voltage Vin and the input current Iin input to the active filter circuit shown in FIG. The voltage Vin corresponds to the voltage waveform as the rectified output of the bridge rectifier circuit Di, and the current Iin corresponds to the current waveform as the rectified output of the bridge rectifier circuit Di. Here, the waveform of the current Iin has the same conduction angle as the rectified output voltage (voltage Vin) of the bridge rectifier circuit Di. This is the waveform of the AC input current flowing from the commercial AC power supply AC to the bridge rectifier circuit Di. Also indicates that the conduction angle is the same as that of the current Iin. That is, a power factor close to 1 is obtained.

FIG. 9B shows a change in energy (power) Pchg input / output to / from the output capacitor Cout. The output capacitor Cout stores energy when the input voltage Vin is high, and releases energy when the input voltage Vin is low to maintain the flow of output power.
FIG. 9C shows a waveform of the charge / discharge current Ichg with respect to the output capacitor Cout. The charge / discharge current Ichg flows corresponding to the energy Pchg accumulation / discharge operation in the output capacitor Cout, as can be seen from the fact that it is in phase with the waveform of the input / output energy Pchg in FIG. Current.

  Unlike the input current Vin, the charge / discharge current Ichg has substantially the same waveform as the second harmonic of the AC line voltage (commercial AC power supply AC). In the AC line voltage, a ripple voltage Vd is generated in the second harmonic component as shown in FIG. 9D due to the flow of energy with the output capacitor Cout. The ripple voltage Vd has a phase difference of 90 ° with respect to the charge / discharge current Ichg shown in FIG. The rating of the output capacitor Cout is determined in consideration of processing the second harmonic ripple current and the high frequency ripple current from the boost converter switch that modulates the current.

FIG. 10 shows a configuration example of an active filter having a basic control circuit system based on the circuit configuration of FIG. In addition, about the part made the same as FIG. 8, the same code | symbol is attached | subjected and description is abbreviate | omitted.
A switching pre-regulator 15 is provided between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the output capacitor Cout. In FIG. 8, the switching pre-regulator 15 is a part formed by the switching element Q, the inductor L, the diode D, and the like.

The control circuit system including the multiplier 11 includes a voltage error amplifier 12, a divider 13, and a squarer 14 in addition.
In the voltage error amplifier 12, the DC voltage Vout of the output capacitor Cout is divided by the voltage dividing resistor Rvo-Rvd and input to the non-inverting input of the operational amplifier 15. A reference voltage Vref is input to the inverting input of the operational amplifier 15. The operational amplifier 15 amplifies a voltage of a level corresponding to the error of the divided DC voltage Vout with respect to the reference voltage Vref by an amplification factor determined by the feedback resistor Rvl and the capacitor Cvl, and outputs the error output voltage Vvea to the divider 13. Output to.

  Further, a so-called feedforward voltage Vff is input to the squarer 14. The feedforward voltage Vff is an output (average input voltage) obtained by averaging the input voltage Vin by the averaging circuit 16 (Rf11, Rf12, Rf13, Cf11, Cf12). The squarer 14 squares the feedforward voltage Vff and outputs it to the divider 13.

In the divider 13, the error output voltage Vvea from the voltage error amplifier 12 is divided by the square value of the average input voltage output from the squarer 14. A signal as a result of the division is output to the multiplier 11.
That is, the voltage loop is made up of a system of a squarer 14, a divider 13, and a multiplier 11. The error output voltage Vvea output from the voltage error amplifier 12 is divided by the square of the average input voltage (Vff) before being multiplied by the rectified input signal Ivac in the multiplier 11. With this circuit, the gain of the voltage loop is kept constant without changing as the square of the average input voltage (Vff). The average input voltage (Vff) has a function of open loop correction that is sent forward in the voltage loop.

An output obtained by dividing the error output voltage Vvea by the divider 11 and a rectified output (Iac) of the positive output terminal (rectified output line) of the bridge rectifier circuit Di via the resistor Rvac are input to the multiplier 11. Here, the rectified output is shown not as voltage but as current (Iac). The multiplier 11 multiplies these inputs to generate and output a current programming signal (multiplier output signal) Imo. This corresponds to the current command value described in FIG. The output voltage Vout is controlled by varying the average amplitude of this current programming signal. That is, a PWM signal corresponding to a change in the average amplitude of the current programming signal is generated, and switching drive is performed by a drive signal based on the PWM signal, thereby controlling the level of the output voltage Vout.
Thus, the current programming signal has an average amplitude waveform that controls the input and output voltages. Note that the active filter controls not only the output voltage Vout but also the input current Vin. Since the current loop in the feedforward circuit can be said to be programmed by the rectified line voltage, the input to the subsequent converter (load 10) becomes resistive.

  FIG. 11 shows a configuration example of a power supply circuit in which a current resonance type converter is connected to the subsequent stage of the active filter based on the configuration shown in FIG. The power supply circuit shown in this figure is so-called wide range compatible with AC input voltage of both AC100V system and AC200V system. Moreover, the structure which can respond to the conditions of load electric power 0-150W is taken. Further, the current resonance type converter adopts a configuration of a separately excited half bridge coupling method.

In the power supply circuit shown in FIG. 11, two sets of line filter transformers LFT and LFT and three sets of across capacitors CL are connected to the commercial AC power supply AC in accordance with the connection mode shown in the figure. A circuit Di is connected.
Also connected to the rectified output line of the bridge rectifier circuit Di is a normal mode noise filter 4 formed by connecting one set of choke coils LN and two sets of filter capacitors (film capacitors) CN and CN as shown in the figure. Is done.

The positive output terminal of the bridge rectifier circuit Di is connected to the positive terminal of the smoothing capacitor Ci through a series connection of the choke coil LN, the inductor Lpc of the power choke coil PCC, and the fast recovery type rectifier diode D10. The smoothing capacitor Ci corresponds to the output capacitor Cout in FIGS. Further, the inductor Lpc and the diode D10 of the power choke coil PCC correspond to the inductor L and the diode D shown in FIG.
In addition, an RC snubber circuit including a capacitor Csn and a resistor Rsn is connected in parallel to the rectifier diode D10 in this figure.

Switching element Q6 corresponds to switching element Q10 in FIG. That is, in actually mounting the switching element of the active filter, in this case, the switching element Q6 is connected to the connection point between the power choke coil Lpc and the fast recovery type rectifier diode D10, and the primary side ground (negative rectified output line). It is inserted between.
In this case, a MOS-FET is selected as the switching element Q6. A gate-source resistor R52 is connected between the gate and source of the switching element Q6.

In this case, the active filter control circuit 20 controls the operation of the active filter for improving the power factor so that the power factor approaches 1, and is an integrated circuit (IC) having one stone, for example.
In this case, the active filter control circuit 20 includes a multiplier, a divider, an error voltage amplifier, a PWM control circuit, a drive circuit that outputs a drive signal for switching the switching element, and the like. Circuit units corresponding to the multiplier 11, the error voltage amplifier 12, the divider 13, the squarer 14, and the like shown in FIG. 10 are mounted in the active filter control circuit 20.

  In this case, the feedback circuit is formed so that the voltage value obtained by dividing the voltage across the smoothing capacitor Ci (rectified smoothing voltage Ei) by the voltage dividing resistors R56 and R57 is input to the terminal T1 of the active filter control circuit 20. .

As the feedforward circuit, first, the rectified output is input to the terminal T3 via the resistor R58. This forms an AC input voltage waveform detection and corresponding feedforward circuit for the averaging circuit.
Further, the rectified current level is input to the terminal T6 through the resistor R60 from the connection point between the negative output terminal of the bridge rectifier circuit Di and the resistor R61 inserted between the primary side ground. That is, a feedforward circuit is formed as a line corresponding to the current detection line LI in FIG.

Further, the positive rectified output of the bridge rectifier circuit Di via the starting resistor Rs is input to the terminal T4 as the starting voltage. The active filter control circuit 20 is activated by the activation voltage input to the terminal T4 when the power supply is activated.
In the power choke coil PCC, a winding N5 that is transformer-coupled with the inductor Lpc is wound. The alternating voltage excited in the winding N5 is converted into a predetermined low-voltage DC voltage by a half-wave rectifier circuit comprising a diode D11 and a capacitor C11. This low-voltage DC voltage is also input to the terminal T4. Yes. After being activated by the activation voltage, the active filter control circuit 20 is operated by inputting this low-voltage DC voltage as a power source.
The terminal T5 is connected to the primary side ground via a resistor R59.

A drive signal for driving the switching element is output from the terminal T2. The drive signal output from the terminal T2 is output to the gate of the switching element Q6 via the resistor R51.
In the switching element Q6, a gate voltage is generated at both ends of the gate-source resistor R52 in accordance with the applied drive signal. Then, the switching operation is performed so that the gate voltage is turned on when the voltage is equal to or higher than the threshold value, and is turned off when the gate voltage is equal to or lower than the threshold value.

  The switching drive of the switching element Q6 is performed based on PWM control so that the conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform as described with reference to FIGS. Done by signal. The conduction angle of the rectified output current is substantially the same as the rectified output voltage waveform. That is, the conduction angle of the AC input current flowing from the commercial AC power supply AC is substantially the same as the waveform of the AC input voltage VAC. As a result, the power factor is controlled to be approximately 1. That is, power factor improvement is achieved. In practice, a characteristic of power factor PF = 0.99 to 0.98 is obtained.

Further, depending on the active filter control circuit 20 shown in FIG. 11, the average value of the rectified and smoothed voltage Ei (corresponding to Vout in FIG. 10) = 375 V is constant in the range of AC input voltage VAC = 85 V to 264 V. Also works. That is, a DC input voltage stabilized at 375 V is supplied to the subsequent stage current resonance type converter regardless of the fluctuation range of AC input voltage VAC = 85 V to 264 V.
The range of the AC input voltage VAC = 85V to 264V continuously covers the commercial AC power supply AC100V system and the 200V system. Therefore, the switching converter in the subsequent stage includes the commercial AC power supply AC100V system and the 200V system. The DC input voltage (Ei) stabilized at the same level is supplied. That is, the power supply circuit shown in FIG. 11 is configured as a wide-range power supply circuit by including an active filter.

  As shown in the figure, the current resonance converter at the latter stage of the active filter includes two switching elements Q1 and Q2. In this case, a half-bridge connection is made such that the switching element Q1 is on the high side and the switching element Q2 is on the low side, and the switching element Q1 is connected in parallel with the rectified and smoothed voltage Ei (DC input voltage). That is, a current resonance type converter by a half bridge coupling method is formed.

The current resonance type converter in this case is a separately excited type, and correspondingly, MOS-FETs are used for the switching elements Q1 and Q2. Clamp diodes DD1 and DD2 are connected in parallel to these switching elements Q1 and Q2, respectively, thereby forming a switching circuit. These clamp diodes DD1 and DD2 form a path through which a reverse current flows when the switching elements Q1 and Q2 are turned off.
The switching elements Q1 and Q2 are driven to be switched by the drive circuit 21 at a required switching frequency at the timing when they are alternately turned on / off. The drive circuit 21 variably controls the switching frequency in accordance with the level of a secondary side DC output voltage Eo described later, thereby stabilizing the secondary side DC output voltage Eo.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1, Q2 from the primary side to the secondary side.
One end of the primary winding N1 of the insulating converter transformer PIT is connected to the connection point (switching output point) of the switching elements Q1 and Q2, and the other end is connected to the primary side via the series resonance capacitor C1. Connected to ground. Here, the series resonant capacitor C1 forms a series resonant circuit by its own capacitance and the leakage inductance (L1) of the primary winding N1. This series resonance circuit generates a resonance operation when the switching outputs of the switching elements Q1 and Q2 are supplied. By this, the operation of the switching circuit composed of the switching elements Q1 and Q2 is made a current resonance type.

A secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT.
In this case, the secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then comprises rectifier diodes DO1 and DO2 and a smoothing capacitor CO as shown in the figure. Both wave rectifier circuits are connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown) and is also branched and input as a detection voltage for the drive circuit 21. As described above, the drive circuit 21 switches the switching frequency so that the secondary side DC output voltage EO is stabilized based on the level of the input secondary side DC output voltage EO. Elements Q1 and Q2 are driven. That is, stabilization by the switching frequency control method is performed.

  As can be seen from the above description, the power supply circuit shown in FIG. 11 is configured by mounting the conventionally known active filter shown in FIGS. 8 and 10. By adopting such a configuration, the power factor is improved. In addition, it is compatible with a so-called wide range that operates on a commercial AC power supply AC100V system and AC200V system under a load power of 150 W or less.

However, the power supply circuit having the configuration shown in FIG. 11 has the following problems.
First, as shown in the figure, the power conversion efficiency in the power supply circuit shown in FIG. 11 is the AC-DC power conversion efficiency (ηAC → DC) corresponding to the front-stage active filter and the DC of the rear-stage current resonance converter. -DC power conversion efficiency (ηDC → DC).
Under the condition of the AC input voltage VAC = 100 V corresponding to the AC 100 V system, ηAC → DC = 93% and ηDC → DC = 94%, and the total efficiency is 87.4%. On the other hand, under the condition of AC input voltage VAC = 230 V corresponding to the AC 200 V system, ηAC → DC = 95%, ηDC → DC = 94%, and the total efficiency is 89.3%. That is, when the AC input voltage VAC is 230 V, the power conversion efficiency on the active filter circuit side is reduced and the overall efficiency is reduced when the AC input voltage VAC is 100 V.

In addition, since the active filter circuit is a hard switching operation, the level of noise generation is very high, and thus a relatively severe noise suppression measure is required.
For this reason, in the circuit shown in FIG. 11, a noise filter including two sets of line filter transformers LFT and three sets of across capacitors is formed for the line of the commercial AC power supply AC. That is, two or more stages of line noise filters are required.
Further, a normal mode noise filter including one set of choke coils LN and two sets of filter capacitors CN is provided for the rectified output line. Further, an RC snubber circuit is provided for the fast recovery diode D10 for rectification.
Thus, as an actual circuit, countermeasures against noise due to an extremely large number of parts are necessary, which leads to an increase in cost and an increase in the mounting area of the power supply circuit board.

  Furthermore, the switching frequency of the switching element Q6 operated by the active filter control circuit 20 as a general-purpose IC is 50 KHz, whereas the switching frequency of the subsequent current resonance type converter is in the range of 70 KHz to 150 KHz. As a result, there is also a problem that the operation as a power supply circuit tends to become unstable, such as the primary side ground potential interfering to induce abnormal oscillation.

Therefore, in the present invention, in view of the various problems described above, the switching power supply circuit is configured as follows.
That is, first, a commercial AC power supply is input to generate a rectified and smoothed voltage, and the rectified and smoothed voltage having a level corresponding to the commercial AC power supply level equal to the commercial AC power supply level is set according to the level of the commercial AC power input. A rectifying / smoothing means that is switched between a generated equal voltage rectifying operation and a voltage rectifying operation that generates the rectified and smoothed voltage at a level corresponding to twice the commercial AC power supply level is provided.
In addition, a switching unit formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage, and a switching drive unit that performs switching driving of the switching element.
Then, at least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained in the primary winding are wound. An insulating converter transformer formed at least, a leakage inductance component of a primary winding of the insulating converter transformer, and a capacitance of a primary side series resonant capacitor connected in series to the primary winding, A primary-side series resonance circuit whose operation is a current resonance type.
A secondary-side DC output voltage generating means configured to generate a secondary-side DC output voltage by performing an rectifying operation by inputting an alternating voltage excited to the secondary winding of the insulating converter transformer; The constant voltage control for the secondary side DC output voltage is performed by controlling the switching drive means according to the level of the secondary side DC output voltage and varying the switching frequency of the switching means. Constant voltage control means.
Further, the primary side series resonance current obtained in the primary side series resonance circuit by the switching operation of the switching means is regenerated to feed back to the smoothing capacitor that forms the rectifying and smoothing means, Power factor improving means is provided for improving the power factor by intermittently rectifying current components by the diode element provided in the rectifying and smoothing means based on the fed back electric power.
In addition, the magnetic flux density of the insulating converter transformer is set to be equal to or less than a predetermined value so that the secondary side rectification operation becomes a continuous mode regardless of the fluctuation of the secondary side DC output voltage. did.

The switching power supply circuit of the present invention having the above configuration includes a current resonance type converter as the primary side switching converter. In addition, the power factor is improved using a power regeneration system.
Then, as described above, the magnetic flux density of the insulating converter transformer is set to be lower than the required value, so that the fluctuation of the secondary side DC output voltage, that is, the fluctuation of the load and the level fluctuation of the commercial AC power supply (AC input voltage). Regardless of the above, the secondary side rectification operation is always in a continuous mode that does not produce a period in which the secondary rectification current becomes discontinuous.
In this way, if the magnetic flux density of the insulating converter transformer is kept below the required level and the secondary side rectification operation is set to the continuous mode, it is superimposed on the primary side series resonance circuit, which was supposed to occur when the power factor is improved by the power regeneration method. The ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage, which is generated along with the ripple of the commercial AC power supply cycle, is suppressed.

  For wide range compatibility, the rectifying / smoothing means for generating the rectifying / smoothing voltage (DC input voltage) is switched between the equal voltage rectifying operation and the voltage rectifying operation according to the commercial AC power supply level. Is configured to be performed.

  Thus, according to the present invention, the magnetic flux density of the insulating converter transformer is set to a predetermined value or less so that the secondary side rectification operation is always in the continuous mode, thereby adopting a configuration for improving the power factor by the power regeneration method. However, the ripple voltage of the secondary side DC output voltage can be suppressed. For example, the capacitance of the secondary side rectifying and smoothing capacitor for generating the secondary side DC output voltage can be within a practical range. In other words, it is possible to more easily realize and push forward the practical use of a power supply circuit having a power factor improving configuration by a power regeneration method.

  In addition, according to the present invention, switching of the rectification operation between the equal voltage rectification operation and the voltage double rectification operation is performed according to the level of the commercial AC power supply as described above, so that a switching power supply circuit compatible with a wide range. Can be realized.

  For these reasons, according to the configuration of the switching power supply circuit of the present invention, it is possible to eliminate the need for a conventional active filter in realizing a wide-range configuration that improves power factor.

Thereby, the fall of power conversion efficiency in the case of aiming at a power factor improvement with the conventional active filter can be suppressed. Also, a large number of component elements for configuring the active filter can be eliminated.
Furthermore, according to the present invention, the current resonance type converter and the power factor correction means that constitute the switching power supply circuit can be set to a soft switching operation, whereby the switching noise is greatly reduced. Thus, there is no need to strengthen the noise filter.
For these reasons, the number of parts is greatly reduced compared with a circuit using a conventional active filter, and the size and weight of the power supply circuit can be reduced. In addition, the cost can be reduced accordingly.
Further, since the active filter is omitted, the interference of the primary side ground potential is eliminated, so that the primary side ground potential is also stabilized and the reliability is improved.

Furthermore, in the present invention, as described above, the magnetic flux density of the insulating converter transformer is set to a predetermined value or less, so that the secondary side rectification operation can always be in the continuous mode.
If the secondary side rectification operation is in the continuous mode in this way, the peak level of the secondary side rectification current can be reduced, and thereby the conduction of the secondary side rectification diode due to the discontinuous mode is achieved. Loss can be reduced. That is, since the conduction loss of the secondary side rectifier diode can be reduced in this way, the secondary side power loss that has occurred in the discontinuous mode can be reduced, and the power conversion efficiency can be further improved. Is.

Hereinafter, the best mode for carrying out the invention (hereinafter referred to as an embodiment) will be described.
FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to the first embodiment of the present invention. The power supply circuit shown in this figure employs a configuration in which a partial voltage resonance circuit is combined with a current resonance type converter using a half-bridge coupling method by a separate excitation type as a basic configuration on the primary side.

In the power supply circuit shown in FIG. 1, first, a noise filter is formed by a filter capacitor CL and a common mode choke coil CMC with respect to the commercial AC power supply AC.
The rectifier circuit system for generating a rectified and smoothed voltage (DC input voltage) Ei from the commercial AC power supply AC includes a bridge rectifier circuit Di, a smoothing capacitor Ci1, and a smoothing capacitor Ci2. The smoothing capacitors Ci1 and Ci2 have the same capacitance.

The bridge rectifier circuit Di is composed of four rectifier diodes D1 to D4 as shown in the figure. In this case, the connection point between the rectification diode D1 and the rectification diode D2 is the first terminal, the connection point between the rectification diode D2 and the rectification diode D4 is the second terminal, and the connection point between the rectification diode D4 and the rectification diode D3 is the third terminal. Assuming that the connection point between the rectifier diode D3 and the rectifier diode D1 is the fourth terminal, the first terminal of the bridge rectifier circuit Di is connected via a series connection of a relay switch S2 and a high-frequency inductor L10, which will be described later, as shown in the figure. , Connected to the positive line of the commercial AC power supply AC.
The second terminal is connected to the primary side ground, and the third terminal is connected to the negative electrode line of the commercial AC power supply AC. The fourth terminal is connected to the positive terminal of the smoothing capacitor Ci1.

In the case of the embodiment, as an operation of the power factor correction circuit 10 to be described later, a rectifier diode that forms the bridge rectifier circuit Di is used to switch the rectifier current so as to perform switching in accordance with the switching cycle. Suppose that at least two or more rectifier diodes are selected as fast recovery diodes.
In the figure, the fast recovery type rectifier diode is shown in black, and the low speed diode is shown in white. That is, here, an example in which a fast recovery diode is selected as the rectifier diodes D1 and D2 in the bridge rectifier circuit Di is shown.

The smoothing capacitors Ci1 and Ci2 are connected in series so that the negative terminal of the smoothing capacitor Ci1 and the positive terminal of the smoothing capacitor Ci2 are connected as shown. In the series connection circuit using the smoothing capacitors Ci1 to Ci2, the positive terminal on the smoothing capacitor Ci1 side is connected to the fourth terminal of the bridge rectifier circuit Di as described above. On the other hand, the negative terminal on the smoothing capacitor Ci2 side is connected to the primary side ground.
A rectified and smoothed voltage Ei (DC input voltage) is obtained at both ends of the series connection circuit of the smoothing capacitors Ci1 to Ci2.

The relay switch S1 is provided to switch the rectifying operation of the rectifying / smoothing circuit between the AC 100V system and the AC 200V system.
The relay switch S1 is a so-called two-contact switch in which the terminal t2 or the terminal t3 is alternatively connected to the terminal t1. The terminal t1 is connected to the third terminal of the bridge rectifier circuit Di. The terminal t2 is connected to a connection point between the smoothing capacitor Ci1 and the smoothing capacitor Ci2. Further, the terminal t3 is open.
Such terminal switching of the relay switch S1 is performed according to the driving state of the relay RL connected to the rectifier circuit switching module 5 as described below.

The rectifier circuit switching module 5 operates to switch the relay switch S1 and the relay switch S2 described above between the AC100V system and the AC200V system by driving the relay RL.
This rectifier circuit switching module 5 is provided with a rectifying and smoothing circuit comprising a rectifying diode D5 and a smoothing capacitor C5 as shown in the figure. In this case, the anode of the rectifier diode D5 is connected to the negative line of the commercial AC power supply AC. The cathode is connected to the positive terminal of the smoothing capacitor C5 whose negative terminal is grounded to the primary side ground, and the connection point between the smoothing capacitor C5 and the diode D5 is a voltage dividing resistor R5- It is connected to the detection terminal of the rectifier circuit switching module 5 via the connection point of R6.
Thereby, a DC voltage of a level corresponding to the AC input voltage VAC is obtained at the detection terminal of the rectifier circuit switching module 5, and the rectifier circuit switching module 5 uses the commercial AC power supply AC based on the voltage level thus obtained. Can be detected.

A relay RL is provided for the rectifier circuit switching module 5. This relay RL controls the contact switching operation of the relay switch S1 and the relay switch S2 according to its own conduction state.
In this case, when the relay RL is in the conductive state, switching is performed so that the relay switch S1 and the relay switch S2 select the terminal t2. Further, when the relay RL is in a non-conduction state, switching is performed so that the terminal t3 is selected by the relay switch S1 and the relay switch S2.

The rectifier circuit switching module 5 compares the voltage level of the commercial AC power supply AC input from the detection terminal as described above with a predetermined reference voltage (for example, 150 V). Based on the comparison result, the relay RL is turned on when the input voltage level to the detection terminal is lower than the reference voltage level, and the relay RL is turned off when the input voltage level is higher than the reference voltage.
That is, in this case, when the level of the commercial AC power supply AC is equal to or lower than the reference voltage and the AC = 100V system, the relay RL is turned on and the terminal t2 is selected in the relay switch S1 and the relay switch S2. It becomes like this. In addition, when the level of the commercial AC power supply AC is equal to or higher than the reference voltage and the AC = 200V system, the relay RL is disabled and each terminal t3 is selected.

Here, for example, when the terminal t2 is selected by the relay switch S1 corresponding to the AC 100V system by the operation of the rectifier circuit switching module 5 as described above, the negative line of the commercial AC power supply AC and the smoothing The connection point of the capacitors Ci1-Ci2 is connected.
For this reason, in the half cycle in which the AC input voltage VAC is positive, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci1. In the half cycle in which the AC input voltage VAC is negative, a rectified current path is formed in which the rectified output from the bridge rectifier circuit Di is charged only to the smoothing capacitor Ci2.
As a result of the rectifying operation performed in this way, a level corresponding to the same multiple of the AC input voltage VAC is generated as the voltage across each of the smoothing capacitors Ci1 and Ci2. Therefore, a level corresponding to twice the AC input voltage VAC is obtained as the DC input voltage Ei, which is the voltage across the series connection circuit of the smoothing capacitors Ci1 to Ci2. That is, a so-called voltage doubler rectifier circuit is formed.

When the terminal t3 is selected by the relay switch S1 corresponding to the AC200V system by the operation of the rectifier circuit switching module 5, the third terminal of the bridge rectifier circuit Di, the smoothing capacitor Ci1, and the smoothing capacitor are selected. The connection point with Ci2 is not connected.
According to this, in the rectifying / smoothing circuit in this case, in each half cycle in which the AC input voltage VAC is positive / negative, the AC input voltage VAC is rectified by the bridge rectifier circuit Di, and the smoothing capacitors Ci1-Ci2 are connected in series. The operation of charging the rectified current to the connection circuit is obtained. That is, a rectification operation by a full-wave rectifier circuit including a normal bridge rectifier circuit can be obtained.
Therefore, in this case, the DC input voltage Ei having a level corresponding to the same multiple of the AC input voltage VAC can be obtained as the voltage across the series connection circuit of the smoothing capacitors Ci1-Ci2.

In this way, in the switching power supply circuit of the present embodiment, the operation of the rectifier circuit switching module 5, the relay RL, and the relay switch S1, and in the case of the commercial AC power supply AC100V system, the AC input voltage VAC is performed by the voltage doubler rectification operation. The DC input voltage Ei corresponding to twice this is generated. In the case of the commercial AC power supply AC200V system, the DC input voltage Ei corresponding to the AC input voltage VAC is generated by the equal voltage rectification operation by the full-wave rectifier circuit.
That is, in the case of the commercial AC power supply AC100V system and the AC200V system, as a result, the DC input voltage Ei at the same level can be obtained, thereby realizing the configuration corresponding to the wide range. is there.

  As a switching converter that operates by inputting the DC input voltage Ei generated by the operation of the rectifying and smoothing circuit as described above, two switching elements Q1 and Q2 by MOS-FETs are half-bridge coupled as shown in the figure. A switching circuit connected by Damper diodes DD1 and DD2 are connected in parallel between the drains and sources of the switching elements Q1 and Q2. The anode and cathode of the damper diode DD1 are connected to the source and drain of the switching element Q1, respectively. Similarly, the anode and cathode of the damper diode DD2 are connected to the source and drain of the switching element Q2, respectively. The damper diodes DD1 and DD2 are body diodes provided in the switching elements Q1 and Q2, respectively.

  A partial resonance capacitor Cp is connected in parallel between the drain and source of the switching element Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed by the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of the primary winding N1. A partial voltage resonance operation in which voltage resonance occurs only when the switching elements Q1, Q2 are turned off is obtained.

  In this power supply circuit, an oscillation / drive circuit 2 is provided to drive the switching elements Q1, Q2 in a switching manner. The oscillation / drive circuit 2 includes an oscillation circuit and a drive circuit, and for example, a general-purpose IC can be used. Then, a drive signal (gate voltage) having a required frequency is applied to the gates of the switching elements Q1 and Q2 by the oscillation circuit and the drive circuit in the oscillation / drive circuit 2. Thereby, the switching elements Q1 and Q2 perform the switching operation so as to be alternately turned on / off at a required switching frequency.

The insulating converter transformer PIT is provided to transmit the switching outputs of the switching elements Q1 and Q2 to the secondary side. The primary winding N1 of the insulating transformer PIT has one end connected to the connection point (switching output point) between the source of the switching element Q1 and the drain of the switching element Q2, and the switching output is transmitted. It is like that.
A primary side series resonant capacitor C1 is connected in series to the other end of the primary winding N1 as shown in the figure. Further, the primary side series resonance capacitor C1 is connected to the high frequency inductor L10 in the power factor correction circuit 10 via the relay switch S2 described above.

  Here, the insulating converter transformer PIT generates a required leakage inductance L1 in the primary winding N1 of the insulating converter transformer PIT by a structure described later. A primary side series resonant circuit is formed by the capacitance of the primary side series resonant capacitor C1 and the leakage inductance L1. According to the connection mode described above, the switching outputs of the switching elements Q1, Q2 are transmitted to the primary side series resonance circuit. The primary side series resonance circuit performs a resonance operation by the transmitted switching output, thereby making the operation of the primary side switching converter a current resonance type.

According to the above description, the primary side switching converter shown in this figure has the operation as the current resonance type by the primary side series resonance circuit (L1-C1) and the part by the partial voltage resonance circuit (Cp // L1) described above. A voltage resonance operation is obtained.
That is, the power supply circuit shown in this figure has a configuration as a complex resonance type converter in which a resonance circuit for making the primary side switching converter a resonance type is combined with another resonance circuit.

A secondary winding N2 is wound on the secondary side of the insulating converter transformer PIT. An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is excited in the secondary winding N2.
The secondary winding N2 is provided with a center tap as shown in the figure and connected to the secondary side ground, and then a double-wave rectifier circuit comprising rectifier diodes DO1 and DO2 and a smoothing capacitor CO as shown in the figure. Is connected. As a result, the secondary side DC output voltage EO is obtained as the voltage across the smoothing capacitor CO. The secondary side DC output voltage EO is supplied to a load side (not shown). The detection voltage for the control circuit 1 is also branched and input.

The control circuit 1 supplies a detection output corresponding to the level change of the secondary side DC output voltage EO to the oscillation / drive circuit 2. The oscillation / drive circuit 2 drives the switching elements Q1 and Q2 such that the switching frequency is varied according to the input detection output of the control circuit 1. That is, the level of the secondary side DC output voltage is stabilized by the switching frequency control method.
For example, when the secondary-side DC output voltage Eo decreases due to a heavy load tendency, the switching frequency is controlled to be lowered. However, this reduces the resonant impedance. The secondary side DC output voltage Eo is raised. On the other hand, when the secondary side DC output voltage Eo rises due to a light load tendency, the resonance impedance is increased by controlling the switching frequency to be increased, and the secondary side The DC output voltage Eo is reduced.

Next, the configuration of the power factor correction circuit 10 will be described.
This power factor correction circuit 10 is provided so as to be inserted into a rectification current path in a rectification smoothing circuit for obtaining a DC input voltage (Ei) from a commercial AC power supply AC. The power factor correction circuit is configured with a combined type.
In FIG. 1, the power factor correction circuit 10 includes the bridge rectifier circuit Di, the high-frequency inductor L10, and the relay switch S2 described above. Further, the filter capacitor CN inserted in the line of the commercial AC power supply AC and the power factor improving series resonance capacitors C20A and C20B connected in parallel to the rectifier diodes D1 and D2 are included.

In the power factor correction circuit 10, as described above, the first terminal of the bridge rectifier circuit Di (the connection point between D1 and D2) is connected to one end (described later) of the high-frequency inductor L10 via the series connection of the relay switch S2. Terminal t2 and t3 side). The other end of the high-frequency inductor L10 is connected to the positive line of the commercial AC power supply AC. That is, in this case, the high-frequency inductor L10 is equivalently inserted in series between the first terminal of the bridge rectifier circuit Di and the commercial AC power supply AC.
The other end of the high frequency inductor L10 is connected to one terminal of the filter capacitor CN. The other terminal of the filter capacitor CN is connected to the third terminal of the bridge rectifier circuit Di. That is, the filter capacitor CN is provided in parallel with the bridge rectifier circuit Di with respect to the commercial AC power supply AC, thereby suppressing normal mode noise generated in the rectified current path.

  Further, as described above, the power factor improving series resonant capacitor C20A and the power factor improving series resonant capacitor C20B are connected in parallel to the rectifying diode D1 and the rectifying diode D2 in the bridge rectifier circuit Di, respectively, as shown in the figure. Connected. In this case, the power factor improving series resonance capacitors C20A and C20B are in series with the high frequency inductor L10. A series resonance circuit is formed in the power factor correction circuit 10 (within the rectified current path of the commercial AC power supply) by the inductance.

  As described above, the high-frequency inductor L10 is connected to the primary winding N1 via the series connection of the relay switch S2 and the primary side series resonant capacitor C1. That is, equivalently, a series connection circuit including a primary side series resonance capacitor C1 and a primary winding N1 is connected to a connection point between the high frequency inductor L10 and the high speed recovery type diode (D1, D2) in the bridge rectifier circuit Di. From this, it can be seen that the power factor correction circuit 10 is connected to the primary side series resonance circuit (L1-C1).

  According to such a circuit configuration of the power factor correction circuit 10, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit is regenerated as electric power, and the rectifier diode D1 // the series resonance capacitor for power factor improvement is recovered. An operation of feeding back to the smoothing capacitor Ci through the parallel connection of C20A or the rectifier diode D2 // power factor improving series resonant capacitor C20B is obtained. In this case, since the capacitance (capacitance) of the power factor improving series resonance capacitor C20 is interposed between the smoothing capacitor Ci and the primary side series resonance circuit, power regeneration is performed by electrostatic coupling. Can be seen.

In the power factor correction circuit 10 shown in FIG. 1, a relay switch S2 is provided for the high-frequency inductor L10.
As this relay switch S2, a terminal t1 is connected to the first terminal of the bridge rectifier circuit Di described above. The terminal t3 of the relay switch S2 is connected to one end of the high frequency inductor L10. The terminal t2 is connected to a tap output point provided in the winding of the high frequency inductor L10. Here, the winding portion from the end connected to the terminal t3 in the high-frequency inductor L10 to the tap output point is defined as a winding portion N10B. The remaining winding portion from the tap output point is the winding portion N10A.
As described above, depending on the operation of the relay switching circuit 5, the terminal t2 is selected in each of the relay switches S1 and S2 corresponding to the AC 100V system. Further, the terminal t3 is selected corresponding to the AC200V system.
According to such an operation, in the high-frequency inductor L10 having the above connection configuration, switching is performed so that only the winding portion N10A is effective in the AC100V system. In the AC200V system, the series connection of the winding portion N10A and the winding portion N10B is effective, and switching is performed so that the entire winding is effective. That is, the high frequency inductor L10 is configured to increase the number of turns in the AC 200V system by such switching operation.
The switching operation for the high-frequency inductor L10 accompanying such switching of the relay switch S2 will be described later.

Here, in the configuration shown in FIG. 1, when a level corresponding to the AC 100V system is obtained as the AC input voltage VAC, the terminal of the relay switch S1 is operated by the operation of the rectifier circuit switching module 5 described above. t2 is selected. In the AC100V system in which the terminal t2 is selected in this way, in the half cycle in which the AC input voltage VAC is positive, the rectified current is [high-frequency inductor L10 → relay switch S2 → rectifier diode D1 → smoothing capacitor Ci1 → It flows through the path of relay switch S1 (terminal t2-t1) → filter capacitor CN].
Further, in the AC100V system, the half cycle in which the AC input voltage VAC is negative is [relay switch S1 (terminal t1-t2) → smoothing capacitor Ci2 → rectifier diode D2 → relay switch S2 → high frequency inductor L10 → filter capacitor. The rectified current flows through the path CN].

From such a rectified current path, it can be seen that the rectification operation is performed by the rectifier diode D1, which is a high-speed recovery type diode, in the AC 100V system and in the half cycle in which the AC input voltage VAC is positive. In addition, it can be seen that the rectification operation is performed by the rectifier diode D2 which is a fast recovery diode in the half cycle in which the AC input voltage VAC is negative.
Then, as the rectified current at this time, the power factor correction circuit 10 and the primary side series resonance circuit are connected as described above, so that a relatively high frequency component as the primary side series resonance current is superimposed. Obtained as a waveform.

In the power factor correction circuit 10, a voltage having a relatively high frequency based on the primary side series resonance current as described above is obtained in the high frequency inductor L10, and a high speed recovery type rectifier diode is switched by this high frequency component. It is what. As a result, even in a period in which the rectified output voltage level is originally lower than the voltage across the smoothing capacitor Ci (Ci1-Ci2), the rectifier diodes D1 and D2 conduct in response to the superimposed high-frequency component, A charging current is supplied to the smoothing capacitor Ci.
As a result, the average waveform of the AC input current component approaches the waveform of the AC input voltage, the conduction angle of the AC input current IAC is expanded, and the power factor is improved.

On the other hand, when a level corresponding to the AC 200V system is input as the AC input voltage VAC, the terminal t3 is selected by the relay switch S1 as described above.
In the AC200V system in which the terminals t3 are both selected in this way, in the half cycle in which the AC input voltage VAC is positive, the rectified current is [high frequency inductor L10 → relay switch S2 → rectifier diode D1 → smoothing capacitor Ci1− It flows through a path of Ci2 → rectifier diode D4 → filter capacitor CN].
Further, in the negative half cycle, the rectified current flows along the path [rectifier diode D3 → smoothing capacitor Ci1-Ci2 → rectifier diode D2 → relay switch S2 → high frequency inductor L10 → filter capacitor CN].
That is, in response to the AC 200 V system, in the half cycle in which the AC input voltage VAC is positive, the rectification operation is performed by the set of the rectifier diode D1 (fast recovery type) and the rectifier diode D4. In the negative half cycle, the rectification operation is performed by a set of the rectifier diode D2 (high-speed recovery type) and the rectifier diode D3.

As described above, when AC is a 200V system, at least one fast recovery diode performs a rectifying operation in both periods where the AC input voltage VAC is positive / negative.
In this way, the rectification operation is performed by one of the high-speed recovery type rectifier diodes, so that the rectification is performed based on the primary series resonance current fed back as in the case of the AC100V system described above. An operation for interrupting the current is obtained. In other words, in this case as well, the charging current to the smoothing capacitor Ci can flow during the period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci.

  From the above description, depending on the power factor correction circuit 10 shown in FIG. 1, the conduction angle of the AC input current IAC is expanded in both the AC 100V system and the AC 200V system, thereby improving the power factor. According to this, the active filter can be omitted in order to improve the power factor.

In this case, since the two rectifier diodes perform rectification operation in each half cycle for the AC200V system as described above, the other rectifier diode is also a fast recovery type, and a bridge rectifier circuit. It is conceivable that all of the rectifier diodes constituting Di are of a high-speed recovery type. However, since such a fast recovery diode is relatively expensive, the circuit manufacturing cost is increased accordingly.
In this example, as described above, at least one of the rectifier diodes to be inserted in each half-cycle rectifier current path is a fast recovery diode, so that the other rectifier diode is configured. (Low speed diode) can be turned on, thereby realizing the power factor improving operation described above. In other words, according to the present example, it is possible to reduce the circuit cost of the bridge rectifier circuit Di by using only at least two rectifier diodes as a fast recovery type.
Then, according to the rectified current path described above, only the rectifier diodes D1 and D2 perform the rectification operation corresponding to the AC 100V system. In other words, in the bridge rectifier circuit Di, only the rectifier diodes D1 and D2 perform the rectification operation in common in the AC100V system and the AC200V system. In order to realize this, it is necessary to select a fast recovery diode as at least the rectifier diodes D1 and D2.

Here, as understood from the above description, in the circuit of FIG. 1, the secondary side DC output voltage is stabilized by variable control of the switching frequency of the primary side switching element. . In the case of adopting such a configuration, for example, in a state where the load tends to be light, stabilization is achieved by controlling the switching frequency to be high. In this state, the secondary-side rectifier circuit operates in a so-called continuous mode in which the period during which the secondary-side rectified current flows through the secondary-side smoothing capacitor is continuous and there is no period of pause.
On the other hand, if the primary side switching frequency is controlled to decrease as the secondary side DC output voltage drops due to heavy load, the secondary side smoothing capacitor will The so-called discontinuous mode is entered, in which the rectified current stops flowing continuously and a current discontinuity period occurs. That is, there is a state in which the discontinuous mode is set according to the load variation as the secondary-side double-wave rectification operation.
The secondary side DC output voltage varies depending on the commercial AC power supply AC (AC input voltage VAC), and a constant voltage control operation corresponding to this is also performed, so that the secondary side DC output voltage also depends on the level of the AC input voltage VAC. There will be a state of discontinuous mode.

Further, as in the circuit shown in FIG. 1, in the configuration in which the power factor is improved by the power regeneration method (including the electrostatic coupling type and the magnetic coupling method described later), the commercial AC is used as the primary side series resonance current. It is known that the ripple voltage of the power supply cycle is superimposed, the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage is significantly increased compared to the configuration without the power factor correction circuit. ing.
It is known that this is mainly caused by the secondary side rectification operation being in the discontinuous mode due to, for example, load fluctuations and fluctuations in the AC input voltage VAC as described above.

  As a countermeasure against such a ripple voltage, the capacitance of the smoothing capacitor (Co) for smoothing the DC output voltage must be increased 5 to 6 times. That is, even if the gain of the control circuit 1 is improved as much as possible, it is necessary to increase the capacitance of the smoothing capacitor 5 to 6 times in order to make it equivalent to the circuit before the power factor improvement, which greatly increases the cost. The practical application becomes unrealistic. Since such a measure is not realistic, when a power factor correction circuit based on a power regeneration method is simply provided, for example, it is strict that the ripple voltage is kept below a certain level, and it is very difficult to adopt it in a required device. It is difficult.

  Therefore, as the present embodiment, the ripple voltage of the secondary side DC output voltage Eo as described above is suppressed, and a power factor correction circuit using a power regeneration system can be practically used. For this reason, the circuit shown in FIG. 1 prevents the discontinuous mode described above, which is the main cause of the ripple voltage, and prevents the ripple voltage from being changed regardless of load fluctuations and AC input voltage VAC fluctuations. It is assumed that the continuous mode is maintained as the secondary side rectification operation.

FIG. 2 is a cross-sectional view showing a structural example of an insulating converter transformer PIT included in the power supply circuit of FIG.
As shown in this figure, the insulating converter transformer PIT includes an EE type core (EE-shaped core) in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs face each other.
And the bobbin B formed with the shape which divided | segmented so that it might mutually become independent about the winding part of a primary side and a secondary side, for example with a resin etc. is provided. The primary winding N1 is wound around one winding portion of the bobbin B. Further, the secondary winding N2 is wound around the other winding portion. By attaching the bobbin B on which the primary side winding and the secondary side winding are wound in this way to the EE type cores (CR1, CR2), the primary side winding and the secondary side winding are different from each other. By the winding area, the center magnetic leg of the EE core is wound. In this way, the structure of the insulating converter transformer PIT as a whole is obtained. In this case, for example, EER-40 is selected as the actual EE type core.

  In addition, a gap G having a gap length of about 1.4 mm is formed on the central magnetic leg of the EE type core as shown in the figure. Thereby, as the coupling coefficient k, for example, a loosely coupled state with k = 0.8 or less is obtained. Note that k = 0.75 was set as the actual coupling coefficient k. The gap G can be formed by making the central magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.

  Further, the number of turns (number of turns) of the primary winding N1 and the secondary winding N2 is set so that the induced voltage level per 1T (turn) of the secondary winding is lower than required. For example, by setting the primary winding N1 = 60T and the secondary winding N2 = 2T + 2T = 4T, the induced voltage level per 1T (turn) of the secondary winding is 2.5 V / T or less.

By setting the structure of the insulating converter transformer PIT and the number of turns of the primary winding N1 and the secondary winding N2, the leakage inductance of the insulating converter transformer PIT in this case tends to increase, and the magnetic flux density in the core Decreases below the required level.
If the magnetic flux density generated in the core of the insulating converter transformer PIT can be reduced to a required level or less in this way, the electric power excited on the secondary side also changes, thereby causing a heavy load and a low AC input voltage. Even in this condition, the secondary side rectification operation can be set to the continuous mode.

  In this way, the continuous mode is obtained even under heavy load and low AC input voltage states, regardless of changes in the secondary side DC output voltage Eo due to load fluctuations, AC input voltage fluctuations, etc. This means that the secondary side rectification operation is performed in the continuous mode. As a result, in the case of adopting the configuration including the power factor correction circuit based on the power regeneration system as in the present embodiment, it is assumed that the secondary side DC output voltage Eo is superposed by the ripple superposed on the primary side series resonance current. In addition, an increase in the ripple of the commercial AC power supply cycle is greatly suppressed.

  As a result, the level of the ripple voltage of the commercial AC power supply cycle superimposed on the secondary side DC output voltage also decreases, and it is not necessary to increase the capacitance of the secondary side smoothing capacitor. That is, the practical use of a switching power supply circuit including a power factor correction circuit based on a power regeneration system can be easily realized. For example, in this embodiment, the increase in the capacitance of the secondary-side smoothing capacitor (Co) can be suppressed to about twice that in the case where no power factor correction circuit is provided.

  Note that the ripple of the commercial AC power supply cycle, which was supposed to be superimposed on the secondary side DC output voltage Eo in this way, can be suppressed because the secondary side rectification operation becomes a continuous mode as described above. This is because the peak level of the secondary side rectified current is suppressed. In addition, as described above with reference to FIG. 2, the magnetic flux density of the insulating converter transformer is reduced to a required level or less, so that the power transmission from the primary side to the secondary side also changes, and the secondary side accordingly. This is also due to the fact that the influence of the ripple on the commercial AC power supply cycle generated in the primary side series resonance current is reduced.

3 and 4 show the characteristics of the power supply circuit of the first embodiment shown in FIG. 1 as AC to DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range of DC input voltage Ei. Each characteristic of ΔEi will be described.
3, AC → DC power conversion efficiency (ηAC-DC), power factor PF, DC input voltage with respect to fluctuations in load power Po = 150 W to 0 W (Eo = 5 V × (30 A to 0 A)) in the circuit of FIG. Each characteristic of the fluctuation range ΔEi of Ei is shown. In FIG. 4, the AC → DC power conversion efficiency (ηAC-DC), the power factor PF, and the fluctuation range ΔEi of the DC input voltage Ei with respect to the fluctuation of the AC input voltage VAC = 85V to 288V. Each characteristic is shown.
In these figures, for each characteristic, in FIG. 3, the result when the AC input voltage VAC = 100V is constant is shown by a solid line, and the result when the AC input voltage VAC = 230V is constant is shown by a broken line. Yes. FIG. 4 shows the result when the load power Po is constant at 150 W, and the result for the AC input voltage VAC = 100 V system is shown by a solid line, and the result for the AC input voltage VAC = 200 V system is shown by a broken line. Show.

Further, in obtaining the experimental results shown in these figures, each part in the circuit of FIG. 1 was selected as follows.
Insulating converter transformer PIT: EER-40 ferrite core, gap G = 1.4 mm, primary winding N1 = 60T, secondary winding N2 = 2T + 2T = 4T, coupling coefficient k = 0.75
Primary side series resonant capacitor C1 = 0.022 μF
High frequency inductor L10 = winding part N10A + winding part N10B = 56 μH + 64 μH = 120 μH
Series resonant capacitor for power factor improvement C20A = C20B = 0.022μF

In these figures, the AC → DC power conversion efficiency in the circuit of FIG. 1 is a solid line when the AC input voltage VAC = 100V (AC100V system) and the AC input voltage VAC = 230V (AC200V system), respectively. And a broken line. According to the experiment, the AC → DC power conversion efficiency (ηAC-DC) when the load power Po = 150 W is ηAC-DC = 91.5% when the AC input voltage VAC = 100 V, and when the AC input voltage VAC = 230 V. ηAC-DC = 90.7%.
This is an improvement of 4.1% in the case of the circuit of FIG. 11 compared with the overall efficiency at VAC = 100 V = 87.4%, and the overall efficiency at the time of VAC = 230 V in the case of the circuit of FIG. Compared to 89.3%, this is an improvement of 1.4%.
Further, according to experiments, the AC input power in the circuit of FIG. 1 is reduced by 7.7 W when VAC = 100 V and 2.6 W when VAC = 230 V, compared to the circuit of FIG. The result is obtained.

  Regarding the fluctuation range ΔEi of the DC input voltage Ei, a result is obtained that ΔEi = 37 V with respect to the fluctuation of the load power Po = 150 W to 0 W when the AC input voltage VAC = 100V. Further, the fluctuation range ΔEi when the AC input voltage VAC = 230 V is ΔEi = 2.0 V with respect to the fluctuation of the load power Po = 150 W to 0 W.

As for the power factor PF, PF = 0.835 under the condition of AC input voltage VAC = 100 V and load power Po = 150 W, and PF = 0.832 under the conditions of AC input voltage VAC = 230 V and load power Po = 150 W. The result is obtained.
From these results, it can be seen that the power factor PF in this case has almost the same value for the AC100 system and the AC200V system.

Here, in the circuit shown in FIG. 1, the almost same power factor can be obtained in the AC100 system and the AC200V system in the following manner.
In other words, as shown in FIG. 1, the power factor correction circuit 10 in this case is provided with a relay switch S2 for switching the number of turns of the high-frequency inductor L10. According to the above description, only the winding part N10A in the high-frequency inductor L10 is effective in correspondence with the case where the AC input voltage VAC = AC100V system. On the other hand, for the AC200V system, the entire winding including winding portion N10A and winding portion N10B is made effective.

If the number of windings of the high frequency inductor L10 changes in this way, the inductance value of the high frequency inductor L10 changes according to the difference in the number of turns, and accordingly, the potential obtained by the power factor correction circuit 10 also changes. It will be. That is, the power level to be fed back to the rectified current path for power factor improvement changes.
As the number of turns of the high-frequency inductor L10 increases in the AC200V system as described above, the power level fed back to the rectified current path increases as the inductance value increases. Depending on this, since the energy returned in the power factor correction circuit 10 increases, a higher power factor can be obtained.
That is, in the circuit configuration of FIG. 1, when the inductance of the high-frequency inductor L10 is fixed, the power factor is lower in the AC200V system than in the AC100V system, but as described above, in the AC200V system. This characteristic is improved by increasing the power factor.

As described above, depending on the configuration of the switching power supply circuit according to the first embodiment shown in FIG. 1, the rectification operation for generating the DC input voltage is performed at the same voltage rectification according to the rated level of the commercial AC power supply. By switching between the operation and the voltage doubler rectification operation, a configuration compatible with a wide range is realized.
In addition, in the present embodiment, a switching power supply including a power factor correction circuit based on a power regeneration system is set by the structure of the insulating converter transformer PIT described above and the number of turns of the primary winding N1 and the secondary winding N2. It is possible to suppress the ripple of the commercial AC power supply cycle superimposed on the secondary side DC voltage Eo, which has been a cause of hindering the practical use of the circuit.
Thereby, the configuration for power factor improvement by the power regeneration method can be realized at a practical level.

Here, when the power supply circuit of this embodiment is compared with the power supply circuit provided with the active filter shown in FIG. 11 as a prior art which also aims at wide-range compatibility while improving the power factor. The following can be said.
First, as is clear from the experimental results shown in FIGS. 3 and 4, the power conversion efficiency is improved by the circuit shown in FIG. 1 as compared with the circuit shown in FIG.
This is because the active filter is not required because the power factor improvement circuit 10 using the power regeneration system is provided and the wide range is supported by switching the rectification operation.
Further, as explained in FIG. 2 above, the peak density of the secondary side rectified current is reduced by reducing the magnetic flux density of the insulating converter transformer PIT to a required level and expanding the continuous mode of the secondary side rectification operation. It depends on being suppressed. That is, by suppressing the peak level of the secondary side rectified current in this way, the conduction loss in the secondary side rectifier diodes (DO1, DO2) is reduced, and the power loss is reduced accordingly. .

In the circuit of this example shown in FIG. 1, since the active filter can be eliminated as described above, the number of circuit components can be reduced.
In other words, the active filter constitutes a set of converters, and as can be seen from the description with reference to FIG. 11, in practice, there are many switching elements and ICs for driving them, and so on. Consists of the number of parts.
On the other hand, in the power supply circuit shown in FIG. 1, relay switches (S1, S2), a high-frequency inductor L10, a filter capacitor CN, and a series resonance for power factor improvement are added as additional components necessary for power factor improvement and wide range correspondence. Since only the capacitors C20A and C20B, the rectifier circuit switching module 5 and the like are provided, the number of parts can be reduced with a simple configuration as compared with the active circuit.
As a result, the power supply circuit shown in FIG. 1 can be manufactured at a much lower cost than the circuit shown in FIG. 11 as a wide-range power supply circuit having a power factor correction function. Further, since the number of parts is greatly reduced, the circuit board can be effectively reduced in size and weight.

In the power supply circuit shown in FIG. 1, since the operation of the resonant converter and the power factor correction circuit 10 is a so-called soft switching operation, the level of switching noise is greatly reduced as compared with the active filter shown in FIG. Is done.
For this reason, as shown in FIG. 1, if a one-stage noise filter comprising a pair of common mode choke coils CMC and an across capacitor CL is provided, it is possible to sufficiently satisfy the power disturbance standard. The Further, as shown in FIG. 1, the normal mode noise of the rectified output line is taken by only one filter capacitor CN.
By reducing the number of components as a noise filter in this way, the cost reduction of the power supply circuit and the reduction in size and weight of the circuit board are promoted.

Further, in the case of the power supply circuit shown in FIG. 1, the switching frequency of the primary side switching converter varies in the range of, for example, 70 KHz to 150 KHz in order to make the voltage constant according to changes in the AC input voltage VAC and load power. However, the switching elements Q1 and Q2 forming the switching converter perform a switching operation in synchronization. Therefore, as the primary side ground potential, as in the power supply circuit of FIG. 11, there is no interference between the active filter side and the subsequent switching converter, and the primary side ground potential can be stable regardless of the change of the switching frequency. it can.
Thereby, the abnormal oscillation which has been a problem in the circuit of FIG. 11 having the conventional active filter is suppressed, and a more stable operation as a power supply circuit can be obtained.

  From these comparisons, the circuit of the present example shown in FIG. 1 solves various problems of the circuit of FIG. 11 having the active filter, and is practically sufficient as equivalent to the case of having the active filter. A high power factor can be obtained, and furthermore, higher power conversion efficiency can be obtained.

Next, a configuration example of a switching power supply circuit as a second embodiment of the present invention is shown in the circuit diagram of FIG. In FIG. 5, parts already described in FIG. 1 are denoted by the same reference numerals and description thereof is omitted.
The switching power supply circuit according to the second embodiment includes a power coupling type power factor correction circuit 11 instead of the electrostatic coupling type power factor correction circuit 10 as a power regeneration type power factor correction circuit.
The magnetic coupling type power factor correction circuit 11 is composed of a bridge rectifier circuit Di, a filter capacitor CN, and a high frequency inductor L10. When this point is compared with the power factor correction circuit 10 shown in FIG. 1, the power factor correction series resonance capacitors C20A and C20B and the relay switch S2 are omitted, and the number of parts is reduced accordingly. It will be.

According to such a power factor correction circuit 11, the first terminal of the bridge rectifier circuit Di is connected to one end of the high-frequency inductor L10. Even in this case, the high-frequency inductor L10 is equivalently Is inserted in series between the first terminal of the bridge rectifier circuit Di and the commercial AC power supply AC.
Furthermore, also in this case, the primary side series resonant circuit (L1-C1) is connected to the connection point between the first terminal of the bridge rectifier circuit Di and the high frequency inductor L10. In other words, in this case, the switching output (primary side series resonance current) obtained in the primary side series resonance circuit is regenerated as power and fed back to the smoothing capacitor Ci via the magnetic coupling (ie, the high frequency inductor L10). Is done.

  Also in the power factor correction circuit 11 having the above configuration, a voltage based on the primary side series resonance current fed back from the primary side series resonance circuit is obtained in the high frequency inductor L10. As a result, the fast recovery diodes (D1, D2) in the bridge rectifier circuit Di perform switching based on the high-frequency component based on the switching output from the primary side series resonant circuit, as in the case of the circuit of FIG. The power factor can be improved.

Also in the power supply circuit of the second embodiment having such a circuit configuration, the insulating converter transformer PIT adopts the same structure as that of the first embodiment described above, so that the coupling coefficient k For example, k is reduced to about 0.75, and a continuous mode is always obtained as a secondary side rectification operation regardless of load fluctuations, AC input voltage fluctuations, and the like.
As a result, in the same manner as the power supply circuit of the first embodiment, an increase in the ripple voltage superimposed on the secondary side DC output voltage Eo is suppressed, and a magnetic coupling type configuration is used as a power regeneration system. Even if it is assumed, the practical use is easily realized.

As shown in the figure, the circuit of FIG. 5 also includes a relay switch S1, a relay RL, and a rectifying circuit switching module 5 for switching the rectifying operation. This realizes a wide-range configuration that switches the double voltage rectification operation / the equal voltage rectification operation between the AC 100 V system and the AC 200 V system.
As the relay switch S1 in this case, unlike the circuit of FIG. 1, a relay having a single contact for on / off switching by the terminal t1 and the terminal t2 is used. That is, in this case, the rectifier circuit switching module 5 is operated so that the relay switch S1 is turned on corresponding to the AC100V system and the relay switch S1 is turned off corresponding to the AC200V system. As a result, the same rectifying operation switching as in the case of FIG. 1 is possible, and the operation corresponding to the wide range is realized.

6 and 7 show the characteristics of the power supply circuit according to the second embodiment, such as AC → DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range ΔEi of DC input voltage Ei. Shows about.
Also in this case, in FIG. 6, the AC → DC power conversion efficiency (ηAC-DC), power factor PF, and DC input voltage Ei with respect to fluctuations in load power Po = 150 W to 0 W (Eo = 5 V × (30 A to 0 A)) FIG. 7 shows characteristics of the fluctuation range ΔEi. In FIG. 7, the AC → DC power conversion efficiency (ηAC-DC), power factor PF, and fluctuation range ΔEi of the DC input voltage Ei with respect to fluctuations of the AC input voltage VAC = 85V to 288V are shown. It shows the characteristics.
Also in this case, for each characteristic, in FIG. 6, the result when the AC input voltage VAC = 100 V is constant is shown by a solid line, and the result when the AC input voltage VAC = 230 V is constant is shown by a broken line. In the case of FIG. 7, the characteristics for the AC 100 V system are indicated by a solid line, and the characteristics corresponding to the AC 200 V system are indicated by a broken line.

Note that each part of the circuit shown in FIG. 5 was selected as follows in order to obtain the experimental results shown in these figures.
Insulating converter transformer PIT: EER-40 ferrite core, gap G = 1.4 mm, primary winding N1 = 60T, secondary winding N2 = 2T + 2T = 4T, coupling coefficient k = 0.75
Primary side series resonant capacitor C1 = 0.018 μF
High frequency inductor L10 = 120μH
Filter capacitor CN = 1μF / 200V
Smoothing capacitor Ci1 = Ci2 = 1000 μF

As shown in these figures, in the case of the power supply circuit of the second embodiment, the AC → DC power conversion efficiency (ηAC-DC) when the load power Po = 150 W is ηAC when the AC input voltage VAC = 100 V. When dc = 91.4% and AC input voltage VAC = 230V, ηAC−DC = 90.5%.
From these results, it can be understood that the circuit shown in FIG. 5 is also more efficient than the case where the active filter shown in FIG. 11 is provided. That is, an improvement of 4.0% is achieved when VAC = 100V, and an improvement of 1.2% is achieved when VAC = 230V.
Further, according to the experiment, the AC input power when compared with the circuit of FIG. 11 is reduced by 7.5 W when VAC = 100 V and 2.3 W when VAC = 230 V.

  Regarding the fluctuation range ΔEi of the DC input voltage Ei, a result is obtained that ΔEi = 48V with respect to the fluctuation of the load power Po = 150 W to 0 W when the AC input voltage VAC = 100V. Further, the fluctuation range ΔEi when the AC input voltage VAC = 230 V is similarly ΔEi = 14 V with respect to the fluctuation of the load power Po = 150 W to 0 W.

  The power factor PF is PF = 0.835 under the conditions of AC input voltage VAC = 100 V and load power Po = 150 W, and PF = 0.0.3 under the conditions of AC input voltage VAC = 230 V and load power Po = 150 W. A result of 834 is obtained. That is, it can be seen that even in the circuit shown in FIG. 5, almost the same power factor is obtained in the AC200V system and the AC100V system.

Here, according to the above description, the circuit shown in FIG. 5 is not provided with the relay switch S2 for switching the number of turns of the high-frequency inductor L10. However, in the configuration shown in FIG. Under the selection conditions and load conditions of each part as illustrated in Fig. 1, it is possible to obtain the same power factor in the AC 100V system and the AC 200V system by setting the inductance of the high frequency inductor L10.
In this case, by setting the inductance of the high-frequency inductor L10 to 120 μH, as shown in FIG. 6, the power factor with respect to the fluctuation from the load power Po = 150 W to around 80 W is as shown in FIG. It was possible to obtain almost the same characteristics.
As a matter of course, the circuit shown in FIG. 5 can also be used in the same way as the circuit shown in FIG. 1 when the power factor in the AC 200 V system decreases due to, for example, changes in selection conditions or load conditions of each part. It is also possible to adopt a configuration in which the switch S2 is provided to switch the inductance of the high frequency inductor L10.

As described above, in the second embodiment, the relay switch S1 for switching the rectifying operation is selected as having one contact, but this is because the circuit of FIG. This is because the relay switch S2 controlled by the same rectifier circuit switching module 5 need not be provided.
That is, in the circuit of FIG. 1, the relay switch S2 for switching the winding of the high-frequency inductor L10 needs to be a two-contact type of terminals t1, t2, and t3. However, the same two-contact type is used. On the other hand, in the circuit of FIG. 5, the relay switch S2 does not have to be provided as described above, so that there is no such inconvenience in component selection, and the relay switch S1 is simply a bridge rectifier circuit. One point of contact that can be switched between connection / non-connection between the third terminal of Di and the connection point of the smoothing capacitors Ci1-Ci2 can be selected.
Of course, in the case of FIG. 5, the relay switch S1 may have two contacts.

  From the experimental results of FIGS. 6 and 7 described above, depending on the second embodiment, as a configuration corresponding to a wide range for improving the power factor, it is practically sufficient as in the case of the first embodiment. It can be understood that a high power factor can be obtained, and that the power conversion efficiency is higher than that of the circuit shown in FIG.

Note that the present invention is not necessarily limited to the configuration as the embodiment described above.
For example, as a switching element, an element other than a MOS-FET may be employed as long as it is an element that can be used in an separately excited type, such as an IGBT (Insulated Gate Bipolar Transistor). Also, the constants of the component elements described above may be changed according to actual conditions. Further, for example, the circuit configuration for generating the secondary side DC output voltage on the secondary side of the insulating converter transformer PIT may be appropriately changed.
Furthermore, the wide-range configuration according to the present invention can be applied to a self-excited current resonance converter.

In the embodiment, as the operation of the power factor correction circuit, a fast recovery type diode for intermittently rectifying current based on the primary side switching output fed back from the primary side series resonance circuit is used as the bridge rectifier circuit Di. The power factor of the present invention can also be obtained by inserting such a fast recovery diode into the rectification current path independently of the bridge rectifier circuit Di. Improved operation is possible.
However, if it is also used as a rectifier diode constituting the bridge rectifier circuit Di as in the case of the embodiment, there is an advantage that the number of parts can be reduced accordingly.

In addition, although an electromagnetic relay is used for switching the rectifying operation, other switching configurations may be employed, for example, a switch circuit including an electronic switch or the like may be employed.
In addition, as for the insulating converter transformer PIT, for example, the core type and the like, the structure may be appropriately changed as long as the magnetic flux density is lower than required.

  Furthermore, the power factor correction circuit is not limited to the one shown in the embodiment, and circuit configurations based on various power regeneration (feedback) methods proposed by the present applicant may be adopted. Is possible.

It is a circuit diagram shown about the structural example of the switching power supply circuit as 1st Embodiment in this invention. It is sectional drawing which shows the structural example of an insulating converter transformer. It is a characteristic view about each characteristic of the fluctuation range of AC-> DC power conversion efficiency with respect to the load fluctuation of the power supply circuit of 1st Embodiment, a power factor, and DC input voltage. It is a characteristic view about each characteristic of the AC-> DC power conversion efficiency with respect to the fluctuation | variation of the alternating current input voltage of the power supply circuit of 1st Embodiment, a power factor, and the fluctuation width of a direct current input voltage. It is a circuit diagram shown about the structural example of the switching power supply circuit as 2nd Embodiment in this invention. It is a characteristic view about each characteristic of the fluctuation range of AC-> DC power conversion efficiency with respect to the load fluctuation of the power supply circuit of 2nd Embodiment, a power factor, and DC input voltage. It is a characteristic view about each characteristic of the AC-> DC power conversion efficiency with respect to the fluctuation | variation of the alternating current input voltage of the power supply circuit of 2nd Embodiment, a power factor, and the fluctuation range of a direct current input voltage. It is a circuit diagram which shows the basic circuit structure of an active filter. It is a wave form diagram which shows the operation | movement in the active filter shown in FIG. It is a circuit diagram which shows the structure of the control circuit system of an active filter. It is a circuit diagram which shows the structural example of the power supply circuit which mounted the active filter as a prior art.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Control circuit, 2 Oscillation drive circuit, 5 Rectifier circuit switching module, 10, 11 Power factor improvement circuit, Di bridge rectifier circuit, D1-D4 rectifier diode, Ci1, Ci2 smoothing capacitor, Q1, Q2 switching element, PIT isolation converter Transformer, C1 primary side series resonance capacitor, Cp partial resonance capacitor, N1 primary winding, N2 secondary winding, RL relay, S1, S2 relay switch, CN filter capacitor, L10 high frequency inductor, N10A, N10B winding part, C20A , C20B series resonant capacitor for power factor improvement

Claims (4)

A commercial AC power supply is input to generate a rectified and smoothed voltage. According to the level of the input commercial AC power, the same rectified and smoothed voltage is generated at a level corresponding to the same level as the commercial AC power supply level. Rectifying and smoothing means that is switched between a voltage rectifying operation and a voltage doubler rectifying operation that generates the rectified and smoothed voltage having a level corresponding to twice the level of the commercial AC power supply;
Switching means formed by including a switching element that performs switching by inputting the rectified and smoothed voltage as a DC input voltage;
Switching driving means for switching and driving the switching element;
At least a primary winding to which a switching output obtained by the switching operation of the switching means is supplied and a secondary winding in which an alternating voltage is excited by the switching output obtained by the primary winding are wound. An isolated converter transformer,
The primary side which is formed by at least the leakage inductance component of the primary winding of the insulating converter transformer and the capacitance of the primary side series resonance capacitor connected in series to the primary winding, and the operation of the switching means is a current resonance type A series resonant circuit;
A secondary side DC output voltage generating means configured to input an alternating voltage excited to the secondary winding of the insulating converter transformer and perform a rectification operation to generate a secondary side DC output voltage;
The switching drive means is controlled according to the level of the secondary side DC output voltage, and the switching frequency of the switching means is varied to perform constant voltage control on the secondary side DC output voltage. Constant voltage control means;
The primary side series resonance current obtained in the primary side series resonance circuit by the switching operation of the switching means is used to regenerate power, and is fed back to the smoothing capacitor forming the rectifying and smoothing means. Power factor improvement means for improving the power factor by intermittently rectifying current component by the diode element provided in the rectification smoothing means based on the power, and
The magnetic flux density of the insulating converter transformer is set to be equal to or less than a predetermined value so that the secondary side rectification operation becomes a continuous mode regardless of the fluctuation of the secondary side DC output voltage.
A switching power supply circuit. The rectifying and smoothing means is
As a bridge rectifier circuit formed by bridging a plurality of the above diode elements,
When the voltage doubler rectification operation and the voltage equalization voltage rectification operation are performed, a first diode element inserted in common in each rectification current path formed in one half cycle of the commercial AC power supply, and the voltage doubler When the voltage rectification operation and the equal voltage rectification operation are performed, the second diode element inserted in common in each rectification current path formed in the other half cycle of the commercial AC power supply is a fast recovery type And a bridge rectifier circuit.
The power factor improving means is
In the rectifying and smoothing means, an inductor element connected in series with respect to a connection point between the first diode element and the second diode element;
A power factor improving series resonance capacitor connected in parallel to each of the first diode element and the second diode element to form a series resonance circuit with the inductor element;
A filter capacitor connected in parallel to the bridge rectifier circuit,
Formed by connecting the primary side series resonant circuit to a connection point between the first diode element and the second diode element and a connection point between the inductor element,
The switching power supply circuit according to claim 1. The rectifying and smoothing means is
As a bridge rectifier circuit formed by bridging a plurality of the above diode elements,
When the voltage doubler rectification operation and the voltage equalization voltage rectification operation are performed, a first diode element inserted in common in each rectification current path formed in one half cycle of the commercial AC power supply, and the voltage doubler When the voltage rectification operation and the equal voltage rectification operation are performed, the second diode element inserted in common in each rectification current path formed in the other half cycle of the commercial AC power supply is a fast recovery type And a bridge rectifier circuit.
The power factor improving means is
In the rectifying and smoothing means, an inductor element connected in series with respect to a connection point between the first diode element and the second diode element;
A filter capacitor connected in parallel to the bridge rectifier circuit,
Formed by connecting the primary side series resonant circuit to a connection point between the first diode element and the second diode element and a connection point between the inductor element,
The switching power supply circuit according to claim 1. The rectifying and smoothing means is
As a bridge rectifier circuit formed by bridging a plurality of the above diode elements,
When the voltage doubler rectification operation and the voltage equalization voltage rectification operation are performed, a first diode element inserted in common in each rectification current path formed in one half cycle of the commercial AC power supply, and the voltage doubler When the voltage rectification operation and the equal voltage rectification operation are performed, the second diode element inserted in common in each rectification current path formed in the other half cycle of the commercial AC power supply is a fast recovery type With a bridge rectifier circuit
The power factor improving means is
In the rectifying and smoothing means, an inductor element connected in series with respect to a connection point between the first diode element and the second diode element;
A power factor improving series resonance capacitor connected in parallel to each of the first diode element and the second diode element to form a series resonance circuit with the inductor element;
A filter capacitor connected in parallel to the bridge rectifier circuit, the primary side with respect to a connection point of the first diode element and the second diode element and a connection point of the inductor element It is formed by connecting a series resonant circuit,
Furthermore, according to the level of the commercial AC power supply, switching means configured to switch the inductance of the inductor element is provided.
The switching power supply circuit according to claim 1.

Images