JP2001197738A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a switching power supply circuit which can receive an input voltage over a wide range and can meet actual use conditions corresponding to an AC input voltage fluctuation. SOLUTION: A power factor improving circuit, in which a power supply circuit, a so-called push-pull switching frequency control composite resonance converter, is provided, has a magnetic coupling transformer MCT. A switching output voltage obtained in a primary resonance circuit is fed back by a magnetic coupling method, and the changeover between a push-pull operation and a single end operation is practiced by a switching operation changeover means according to an AC input voltage.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit having a power factor correction circuit.

[0002]

2. Description of the Related Art Conventionally, an AC input voltage of 100 V system and 2
A wide range power factor improving soft switching power supply commonly used for the 00V system has various circuits configured by combining a current resonance type converter and an electrostatic coupling or magnetic coupling type power factor improving circuit. First, the configurations and problems of various switching power supply circuits will be described with reference to FIGS.

FIGS. 10 and 11 show a power factor improving soft switching power supply in which a conventional current resonance type converter and a power feedback type power factor improving circuit are combined, respectively.
0 is a system employing an electrostatic coupling type power factor improving circuit 20,
FIG. 11 shows a system using a power factor improving circuit 21 of a magnetic coupling type.

The power supply circuits shown in FIGS. 10 and 11 are provided with a bridge rectifier circuit Di for performing full-wave rectification of a commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectifier circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 20 or 21, and both ends of the smoothing capacitor Ci have a level of one time of the AC input voltage VAC. A corresponding rectified smoothed voltage Ei is obtained. Also, in the rectifying / smoothing circuit (Di, Ci), an inrush current limiting resistor Ri is inserted in the rectification current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. ing.

Further, the switching power supply circuits shown in FIGS. 10 and 11 are provided with a self-excited current resonance type converter using a rectified smoothed voltage Ei, which is a voltage across a smoothing capacitor Ci, as an operating power supply. In this current resonance type converter, the switching elements Q100 and Q200 of two bipolar transistors are half-bridge-coupled as shown in the figure, and then inserted between the connection point on the positive side of the smoothing capacitor Ci and the ground. Connected. Note that clamp diodes DD100 and DD200 are connected in parallel between the emitter and collector of switching elements Q100 and Q200, respectively. The switching elements Q100 and Q200 perform a switching operation at a required switching frequency based on a signal from the control circuit 1, respectively.

[0006] Insulated converter transformer PIT (Power Is
olation Transformer) is the switching element Q100, Q2
The switching output of 00 is transmitted to the secondary side. That is,
The switching elements Q100 and Q200 alternately open and close with the terminal voltage of the smoothing capacitor Ci as the operating power supply, thereby forming the primary winding N1 of the insulating converter transformer PIT.
To supply a drive current close to the resonance current waveform to obtain an alternating output to the secondary winding N2. This insulation converter transformer P
One end of the primary winding N1 of the IT is connected to the switching element Q100.
Is connected to the contact point (switching output point) between the emitter of the switching element Q200 and the collector of the switching element Q200, so that a switching output can be obtained.

The other end of the primary winding N1 is connected via a series resonance capacitor C100 to a power factor improving circuit 20 (or 2).
1). In this case, the series resonance capacitor C100 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C100 and the leakage inductance of the insulating converter transformer PIT including the primary winding N1 (series resonance winding) A leakage inductance component forms a primary side series resonance circuit for making the operation of the switching converter a current resonance type.

On the secondary side of the insulated converter transformer PIT shown in FIGS. 10 and 11, a center tap is provided for the secondary winding N2, and a rectifying diode D2 is provided.
By connecting O1, DO2 and the smoothing capacitor CO as shown in the figure, a full-wave rectifier circuit is formed, and a DC output voltage EO is generated. The control circuit 1 performs constant voltage control by controlling the switching frequency of the switching elements Q100 and Q200 such that the level is varied according to the level of the DC voltage output EO on the secondary side, for example.

The power factor improving circuit 20 in the case of FIG.
The filter choke coil LN and the high speed recovery type diode D1 are inserted in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. Here, the filter capacitor CN is composed of the filter choke coil LN and the fast recovery type diode D
It is provided in parallel with one series connection circuit. And
With such a connection form, the filter capacitor C
N forms a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3 is provided in parallel with the high-speed recovery type diode D1. Although a detailed description is omitted here, for example, the resonance capacitor C3 forms a parallel resonance circuit together with, for example, the filter choke coil LN and the like, so that its resonance frequency is substantially equal to the resonance frequency of the series resonance circuit described later. Is set. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced.

The power factor improving circuit 20 includes a filter choke coil LN and a high speed recovery type diode D1.
The primary side series resonance circuit (N
1, C100) are connected.

In such a connection form, the switching output obtained from the primary winding N1 is connected to the series resonance capacitor C10.
The switching output is fed back to the rectified current path via the zero capacitive coupling. In this case, the filter choke coil LN and the fast recovery type diode D1
Is fed back so that the resonance current obtained through the primary winding N1 flows, and a switching output is applied.

When the switching output is fed back as described above, the alternating voltage of the switching cycle is superimposed on the rectified current path. In the diode D1, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN also increases due to the interrupting action. Further, a voltage is generated at both ends of the resonance capacitor C3 when a current of a switching period flows through the resonance capacitor C3. However, the level of the rectified smoothed voltage Ei is reduced by the voltage across the resonance capacitor C3. Accordingly, even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci, the smoothing capacitor C
The charging current to i flows. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, the conduction angle of the AC input current is enlarged, and the power factor is improved.

A power factor improving circuit 2 in the case of FIG.
Reference numeral 1 denotes a filter choke coil LN, a high-speed recovery type diode D1, and a choke coil LS which are connected in series between a positive output terminal of the bridge rectifier circuit Di and a positive terminal of the smoothing capacitor Ci. The filter capacitor CN is inserted between the anode side of the high-speed recovery diode D1 and the positive terminal of the smoothing capacitor Ci to form a normal mode low-pass filter together with the filter choke coil LN. The resonance capacitor C3 is provided in series with the high-speed recovery type diode D1.

For the power factor correction circuit 21, the cathode of the high-speed recovery type diode D1 and the choke coil L
The primary side series resonance circuit (N1, C10
0) is connected. That is, the series resonance circuit (N
1, C100) is fed back to the rectified current path via an inductive reactance (magnetic coupling) assumed to be possessed by the choke coil LS itself.

The switching output fed back as described above causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. The superimposed voltage of the switching cycle causes the high speed recovery type diode D1 to be superimposed. In this case, an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the filter choke coil LN and choke coil LS also increases due to the intermittent action. This allows the charging current to flow to the smoothing capacitor Ci even during a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

FIG. 12 shows an AC input voltage VAC = 10 in the switching power supply circuits of FIGS. 10 and 11.
Operating waveforms of the AC input current at 0 V and VAC = 230 V are shown, and FIG. 13 shows a power factor PF-AC input voltage VAC characteristic. As can be seen from these figures, even when the power factor PF is set to 0.85 when the AC input voltage VAC is 100 V, the power factor PF is changed when the AC input voltage VAC is 230 V.
= 0.65. This means that
It also means that a high power factor cannot be stably obtained over a V system to a 200 V system. Therefore, the switching power supply circuits of FIGS. 10 and 11 are not suitable as a wide range power factor improving soft switching power supply commonly used for the 100 V system and the 200 V system.

FIG. 14 shows a series resonance circuit (L
30 and C30), so that the series resonance current I01 of the choke coil L30 and the resonance capacitor C30 is fed back to the parallel connection point of the high-speed recovery type diode D1 and the capacitor C3 of the power factor correction circuit 20. is there. In the following description of each drawing, the same portions as those of the power supply circuit in the drawings already described are denoted by the same reference numerals, and description thereof is omitted.

In this power supply circuit, the switching element Q10
1. A series resonance circuit (N1, C100) provided to make the switching operation of Q201 a current resonance type has one end of a primary winding N1 connected to the source-drain of switching elements Q101, Q201 via a resonance capacitor C100. A switching output is supplied by being connected to a connection point (switching output point) and being grounded to the primary side ground at the other end. Further, the series resonance circuit (L30, C30) is added as described above, and the series resonance current I01
Is fed back to the power factor correction circuit 20.

The switching elements Q101 and Q201 are
A drive pulse having a required switching frequency is supplied from the drive / oscillation circuit 2 to the gate based on a trigger from the start circuit 3 to alternately turn on / off.
Performs off switching. Further, the control circuit 1 changes the pulse frequency (switching frequency) supplied from the drive / oscillation circuit 2 to the switching elements Q101 and Q201 so that the level is varied according to the level of the secondary-side DC voltage output EO. By controlling, constant voltage control is performed.

The power factor PF of the power supply circuit shown in FIG.
-AC input voltage VAC characteristics and power factor PF-Load power Po
The characteristics are shown in FIGS. FIGS. 15A and 15B
As can be seen from the figure, in the case of this power supply circuit, the load power Po
= AC input voltage VAC for fluctuations of 113W to 47W =
In both the case of 100 V and the case of AC input voltage VAC = 230 V, the power factor PF is approximately 0.7 or more, which is sufficient. However, in this power supply circuit, since the primary side series resonance current I02 and the power feedback series resonance current I01 overlap and flow through the switching elements Q101 and Q201, the switching loss of the switching elements Q101 and Q201 increases and the power conversion efficiency decreases. I do. Therefore, the load power P
It can be said that this is a method that can be applied only to light loads of about 100 W or less.

When the load power Po is a heavy load of 200 W or more, a power feedback type power factor correction circuit is inserted into the AC line as shown in FIG. 16 to provide four sets of rectifier diodes (D101 to D1).
04) and two sets of electrolytic capacitors for smoothing (Ci1, Ci2)
And the double voltage / full-wave rectifier circuit switching method using the electromagnetic power relay RY, the AC input voltage VAC is 100 V
A wide range of 00V system can be supported.

That is, in the power supply circuit of FIG. 16, in the power factor correction circuit 22, a filter choke coil LN is inserted in series with the positive input line of the commercial AC power supply AC, and is connected in parallel to the commercial AC power supply AC. A normal mode low-pass filter is formed together with the filter capacitor CN to prevent a harmonic current from flowing to the commercial AC power supply AC. In this case, two parallel resonance capacitors C31,
C32 is provided, the parallel resonance capacitor C31 is inserted in parallel with the rectifier diode D101, and the parallel resonance capacitor C32 is inserted in parallel with the rectifier diode D102.
The parallel resonance capacitors C31 and C32 form a parallel resonance circuit together with, for example, the inductance of the filter choke coil LN. The resonance frequency of the parallel resonance circuit is substantially the same as the resonance frequency of a series resonance circuit described later, for example. It is set as follows.

In this case, four rectifying diodes D101 to
As a bridge rectifier circuit using D104, a high-speed recovery type is used in response to a high-frequency current of a switching cycle flowing through a rectified current path along with a power factor improving operation as described later. The primary winding N1 of the insulated converter transformer PIT is connected to the connection point of the rectifier diodes D101 and D102 via the series resonance capacitor C100, and the switching of the switching converter (described later) obtained on the primary winding N1 is performed. Output is a series resonance capacitor C10
The signal is fed back to the rectified current path via the zero capacitive coupling.

In this power supply circuit, two smoothing capacitors Ci1 and Ci2 are connected in series and provided so as to be inserted between the positive output terminal of the bridge rectifier circuit and the primary side ground. The connection point of the smoothing capacitors Ci1 and Ci2 is connected via a switch S to the negative input terminal of the bridge rectifier circuit. The rectified and smoothed voltage obtained at both ends of the smoothing capacitors Ci1 and Ci2 connected in series is input to a later-stage separately excited current resonance type converter. The current resonance converter performs a switching operation based on the input rectified and smoothed voltage, and finally outputs a stabilized secondary-side DC output voltage EO.

The switch S is provided for switching between a voltage doubler rectifying / smoothing operation and a normal rectifying / smoothing operation, and is controlled on / off by an electromagnetic relay RY. Electromagnetic relay RY
Are driven by the relay drive circuit 40. The relay drive circuit 40 includes a half-wave rectifier circuit including a rectifier diode D105 for half-wave rectifying the commercial AC power supply AC and a smoothing capacitor C33. A resistance is provided between the output of the half-wave rectifier circuit and the primary side ground. R1 and R2 are connected in series. A zener diode ZD is inserted between the voltage dividing points of the resistors R1 and R2 and the base of the transistor Q300. In this case, when the AC input voltage VAC supplied to the commercial AC power supply AC is 150 VAC or more, the resistors R1 and R2
It is assumed that the above components are selected so that the Zener diode ZD is turned on by the voltage value divided by the above. That is, a voltage detection circuit for detecting whether or not the AC input voltage level is equal to or higher than AC 150 V is formed by each of the above components. Transistor Q300 drives electromagnetic relay RY. A resistor R3 and a capacitor C34 are connected between the base of the transistor Q300 and the primary side ground. The collector of the transistor Q300 is grounded to the primary side ground. The emitter is connected via an electromagnetic relay RY to a tertiary winding N3, a rectifier diode D300 and a smoothing capacitor C10 of an insulation converter transformer PIT, which will be described later.
1 is connected to the line of the low-voltage DC voltage obtained. A protection diode D5 for flowing a reverse current is connected in parallel to the electromagnetic relay RY.

For example, as an AC100V system, AC150
When the AC input voltage VAC equal to or lower than V is supplied, the Zener diode ZD does not conduct, so that the base current of the transistor Q300 flows through the resistor R3, and the transistor Q300 is turned on. Thereby, the emitter current is conducted to the electromagnetic relay RY. And the electromagnetic relay RY
The switch S is turned on by the exciting action of. Thus, the connection point between the smoothing capacitors Ci1 and Ci2 and the negative input terminal of the bridge rectifier circuit are connected via the switch S. In such a connection form, when the AC input voltage VAC is positive, a rectification current path for charging the smoothing capacitor Ci1 with the commercial AC power supply AC rectified by the rectifier diode D102 is formed.
A voltage across the level corresponding to the AC input voltage of the AC 100 V system is generated in the smoothing capacitor Ci1. On the other hand, when the AC input voltage VAC is negative, the rectifier diode D
The commercial AC power supply AC rectified at 101 is converted to a smoothing capacitor Ci.
By forming a rectified current path for charging the capacitor 2, a voltage across the level corresponding to the AC input voltage of the AC 100 V system is also generated in the smoothing capacitor Ci 2. Therefore, 1 is connected to both ends of the smoothing capacitors Ci1-Ci2 connected in series.
Rectified smoothing voltage E at a level equivalent to almost twice that of the 00V system
The double voltage rectification smoothing operation in which i occurs is performed.

When an AC input voltage VAC of 150 V AC or more is supplied as an AC 200 V system, the Zener diode ZD is turned on, and the transistor Q3 is turned on.
The base potential of 00 is raised above a predetermined value so that the base current does not flow, and the transistor Q300 is turned off. Therefore, the emitter current of the transistor Q300 does not flow through the electromagnetic relay RY, and the switch S is turned off. In this case, the commercial AC power supply AC is connected to the bridge rectifier circuit (rectifier diodes D101 to D104).
And a full-wave rectification smoothing operation of charging the series connection of the smoothing capacitors Ci1-Ci2,
The rectified smoothed voltage Ei of the AC 200 V system corresponding to the AC input voltage VAC is obtained.

The switching converter shown in this figure is a separately excited current resonance type converter. In this current resonance type converter, for example, two switching elements Q101 and Q201 formed by MOS-FETs are provided in a half-bridge connection. These switching elements Q10
1, Q201 is alternately turned on / off by the oscillation drive circuit 2.
The switching drive is performed so that the OFF operation is repeated, and the rectified smoothed voltage Ei is intermittently turned into a switching output. Note that damper diodes D101 and D201 are provided in parallel with the direction shown in the drawing between the drain and source of each of the switching elements Q101 and Q201.

Sources of the switching elements Q101 and Q201
For the drain connection point (switching output point),
One end of a primary winding N1 of the insulated converter transformer PIT is connected to supply a switching output to the primary winding N1. The primary winding N1 of the insulating converter transformer PIT is connected in series with the series resonance capacitor C100.
And the leakage inductance component of the insulated converter transformer PIT including the primary winding N1,
A series resonance circuit for forming the switching power supply circuit into a current resonance type is formed.

The insulating converter transformer PIT transfers the alternating voltage obtained by the switching output supplied to the primary winding N1 to the secondary side. On the secondary side of the insulated converter transformer PIT, a DC output voltage EO is obtained as in the cases of FIGS.

The starting circuit 3 is provided for detecting the voltage or current obtained on the rectifying / smoothing line immediately after the power is turned on, and for starting the oscillation drive circuit 2. The starting circuit 3 has an operating power supply. Tertiary winding N3 and rectifier diode D30 provided in the insulated converter transformer PIT
0 and the low-voltage DC voltage supplied by the smoothing capacitor C101.

As described above, the power factor improving circuit 22 returns the switching output obtained from the primary winding N1 of the insulating converter transformer PIT to the rectified current path via the capacitive coupling of the series resonance capacitor C100. Has been. The switching output thus fed back acts so as to superimpose an alternating voltage (switching voltage) of a switching cycle on the rectified output voltage via the inductance of the filter choke coil LN. Rectifier diode D
101 and D102 operate so that the rectified current is interrupted in the switching cycle. Since the rectifier diodes D101 and D102 are in the path of the rectified current in both the case of voltage doubler rectification and the case of full wave rectification, the above-described operation is performed in both the case of voltage doubler rectification and the case of full wave rectification. Will be

By this operation, for example, during the voltage doubler rectification operation, the rectified output voltage is charged to the smoothing capacitors Ci1 and Ci2 in a state where the switching voltage is superimposed. The voltages at both ends of Ci1 and Ci2 are reduced in the switching cycle. For this reason, the smoothing capacitors Ci1 and C2 also operate during the period in which the rectified output voltage level is lower than the voltage across each of the smoothing capacitors Ci1 and Ci2.
The charging current to i2 is caused to flow. In the full-wave rectification operation, the rectified output voltage charges the smoothing capacitors Ci1 and Ci2 connected in series by the rectified output voltage on which the switching voltage is superimposed. The voltage (rectified smoothed voltage) across the smoothed capacitors Ci1-Ci2 is reduced in the switching cycle. For this reason,
Smoothing capacitor C with rectified output voltage level connected in series
The charging current is allowed to flow also in a period that is lower than the voltage between i1 and Ci2.

As a result, in either case of the voltage doubler rectification operation or the full-wave rectification operation, the average waveform of the AC input current IAC is made to approach the waveform of the AC input voltage VAC, and the AC input current IAC is reduced. The conduction angle will be enlarged. Thus, in the power supply circuit shown in this figure, the power factor is improved both in the voltage doubler rectification operation and in the full-wave rectification operation.

FIGS. 17A and 17B show a power factor PF-AC input voltage VAC characteristic and a power factor PF-load power Po characteristic of the power supply circuit of FIG. FIG. 17 (a)
As can be seen from (b), in the case of this power supply circuit, the AC input voltage VAC is 230 V and the AC input voltage VAC is 230 V when the load power Po varies from 250 W to 150 W.
In both cases, the power factor PF is sufficient. However, in the circuit shown in FIG. 16, since the component elements for improving the power factor, such as the normal mode low-pass filters (LN, CN) and the parallel resonance capacitors C31 and C32, are provided in the AC input line, for example, worldwide It is necessary to adopt a safety certified product, which is disadvantageous in terms of cost. Also, the detection circuit of the AC input voltage VAC may malfunction due to an instantaneous power failure or disturbance noise, so that measures must be taken to prevent malfunction.

FIG. 18 shows four switching elements (Q41 to Q41).
44) Combining a full-bridge coupled current resonance type converter and a magnetic coupling type power feedback system power factor correction circuit, a series resonance circuit of the primary inductance L30 and C100 of the magnetic coupling transformer (MCT) is added, and the secondary inductance LR
The power is fed back via the.

In this power supply circuit, a separately-excited current resonance type converter in which four switching elements Q41 to Q44 are full-bridge-coupled is provided as a switching converter. Such a separately excited current resonance type converter by full bridge coupling is configured such that a set of switching elements Q41 and Q42 and a set of switching elements Q43 and Q44 are driven by push-pull. At this time, damper diodes DD41 to DD44 are provided also between the respective drains and sources of the switching elements Q41 to Q44.

In this case, drive circuit 2B outputs a switching drive signal to each gate of switching elements Q41 and Q42, and drive circuit 2C outputs a switching drive signal to each gate of switching elements Q43 and Q44. It is configured as follows. In this case, the drive circuits 2B and 2C alternately switch the set of [switching elements Q41, Q44] and [switching elements Q42, Q43] at a required switching frequency based on the oscillation outputs of the opposite phases from the oscillation circuit 2A. A switching drive signal is output so as to perform an on / off switching operation.

When the AC input voltage VAC is detected to be 150 V or more by the VAC detection circuit 2D for detecting the charging voltage of the smoothing capacitor Ci, the VAC detection circuit 2D
If the switching element Q43 is turned off and Q44 is turned on under the control of D, the half-bridge coupled current resonance type converter operates. That is, a full-bridge / half-bridge coupling switching system is used to enable a wide range.

The power factor improving circuit 23 includes a magnetic coupling transformer MCT, and the switching output supplied to the series resonance circuit is fed back to the rectified current path by the magnetic coupling operation. In the power factor correction circuit 23, the positive output terminal of the bridge rectifier circuit Di and the smoothing capacitor C
A filter choke coil LN, a high-speed recovery type diode D1, and a winding NR are connected in series between the positive terminals of i and inserted. The filter capacitor CN is provided in parallel with the series connection circuit of the filter choke coil LN and the winding NR, but also forms a low-pass filter together with the filter choke coil LN by such a connection form.

The magnetic coupling transformer MCT is formed by winding the winding NR and the winding N30 so as to be magnetically tightly coupled to the core. One end of the winding N30 is connected to a connection point between the switching elements Q41 and Q42, and the other end is connected to the switching elements Q43 and Q4 via a series resonance capacitor C100.
Connected to connection point 4. This connection point is connected to one end of the primary winding N1 via the series resonance capacitor C1.
With this connection configuration, the switching outputs of the switching elements Q41 to Q44 are supplied to the series resonance circuit on the primary winding N1 side, and are connected to the winding N30 connected in series to the series resonance circuit. In contrast, a switching output can be obtained.

The power factor improving circuit 2 thus formed
In 3, the switching output obtained at the winding N30 is transmitted to the winding LR through the magnetic coupling action of the magnetic coupling transformer MCT. As a result, a voltage of the switching period (switching voltage) is generated in the winding NR. However, since the winding NR is inserted in the rectification current path, the operation is performed such that the switching voltage is superimposed on the rectification output voltage. I do. Then, the superimposed portion of the switching output voltage causes the high-speed recovery type diode D1 to operate such that the rectified current is intermittently switched in the switching cycle, thereby increasing the conduction angle of the AC input current and improving the power factor. .

FIG. 19 shows the power factor PF-AC input voltage VAC characteristic of the power supply circuit of FIG. This FIG.
As can be seen from the figure, in the case of this power supply circuit, the load power Po
= 192 W to 84 W, a high power factor PF can be obtained. However, there are disadvantages that the circuit configuration becomes complicated, the number of components increases, and the printed circuit board mounting area increases.

[0044]

Problems to be Solved by the Invention FIGS.
Various power supply circuit examples have been given, but the problems of these conventional power supply circuits can be summarized as follows.

When an LC series resonance circuit is provided in parallel with the series resonance circuit of the half-bridge coupled current resonance type converter and combined with the power feedback type power factor correction circuit, the switching current of the two switching elements increases and the power conversion Efficiency decreases.・ In the case of the AC input voltage rectification switching method of the half bridge coupling current resonance type converter, the maximum load power is 250W
Although components are connected to the AC line, it is necessary to select a product approved by safety standards.
It is necessary to take measures against malfunctions of the rectification method switching circuit due to momentary power failure or disturbance noise during system operation.・ AC100 full bridge coupled current resonance type converter
AC half-bridge coupled current resonance type converter
In the case of a full-bridge or half-bridge switching system operating on a 200 V system, the drive stages of the switch elements are two sets, and the circuit configuration is complicated. Therefore, the number of components is large and the board area is increased.

[0046]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a gap is formed between a rectifying and smoothing unit that receives a commercial AC power supply to generate a rectified smoothed voltage and outputs the rectified smoothed voltage as a DC input voltage, and a required coupling coefficient that is loosely coupled is obtained. And a first and a second switching unit for intermittently outputting the DC input voltage by a push-pull operation and outputting the DC input voltage to a primary winding of the insulating converter transformer. Means, and at least a leakage inductance component including a primary winding of the insulated converter transformer and a capacitance of a primary side parallel resonance capacitor, the operation of the first and second switching means being of a voltage resonance type. First and second primary side resonance circuits. Further, as the power factor improving means, a magnetic coupling transformer for magnetically coupling a first winding connected to the primary winding, a second winding inserted to the rectification current path, and a magnetic coupling transformer inserted to the rectification current path. At least a switching element. Then, the operation of the second switching means is stopped according to the level of the commercial AC power supply, so that the DC input voltage is intermittently operated by a single-ended operation by only the first switching means, and the primary of the insulating converter transformer is A switching operation switching means capable of outputting to a winding, a leakage inductance component of a secondary winding of the insulating converter transformer, and a secondary formed on a secondary side by a capacitance of a secondary side resonance capacitor. A side resonance circuit is formed including the secondary side resonance circuit, and an alternating voltage obtained in a secondary winding of the insulating converter transformer is input, and a rectification operation is performed to generate a secondary side DC output voltage. DC output voltage generating means configured as described above and the secondary DC output voltage in accordance with the level of the secondary DC output voltage. Configuring the switching power supply circuit and a constant voltage control means arranged to perform constant voltage control for the voltage.

According to the above configuration, a primary-side resonance circuit is obtained by providing a magnetic coupling transformer with respect to a power factor improving circuit provided in a power supply circuit called a push-pull type switching frequency control type composite resonance type converter. The switching output voltage is fed back by the magnetic coupling method, and the switching operation switching means switches to a single-ended operation when the AC voltage is 150 V or more, for example, so that a wide range of input voltage can be supported and the AC 100 V system And the power factor can be kept constant by the AC200V system.

[0048]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention.
On the primary side of the power supply circuit shown in this figure, a voltage resonance type switching converter (voltage resonance type converter) is provided. A power factor improving circuit 10 is provided for the voltage resonance type converter.

The power supply circuit shown in this figure is provided with a bridge rectifier circuit Di for full-wave rectification of a commercial AC power supply AC. In this case, the rectified output rectified by the bridge rectifier circuit Di is charged to the smoothing capacitor Ci via the power factor correction circuit 10, and both ends of the smoothing capacitor Ci correspond to the level of one time of the AC input voltage VAC. The rectified smoothed voltage Ei is obtained. Further, in the rectifying / smoothing circuit (Di, Ci), an inrush current limiting resistor Ri is inserted into the rectification current path so as to suppress the inrush current flowing into the smoothing capacitor Ci when the power is turned on. I have to.

The power supply circuit includes two sets of switching elements Q1 and Q2. When the AC input voltage is, for example, a 100 V system, the switching element Q1 is switched by a single-ended operation.
When the AC input voltage VAC is, for example, a 200 V system,
The switching elements Q1 and Q2 can be switched by a push-pull operation.
That is, one of the switching operations of the single-ended operation and the push-pull operation is selected and performed based on the AC input voltage VAC.

An orthogonal drive transformer PRT is provided for variably controlling the switching frequency of the switching elements Q1 and Q2. As the structure of the orthogonal drive transformer PRT, for example, a structure having two magnetic legs 2
A three-dimensional core is formed by joining the ends of the magnetic legs of two double U-shaped cores. Then, the detection winding ND and the drive windings NB1 and NB2 are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the detection winding. By winding in the direction orthogonal to ND and the drive windings NB1 and NB2, a saturable reactor is formed.

The drive windings NB1 and NB2 are formed in two parts by providing a center tap for one winding. In this case, the center tap terminal is connected to the primary side ground. An end of the drive winding NB1 is connected to a base current limiting resistor RB1 and a resonance capacitor CB1.
Are connected to the base of the switching element Q1 through the series connection of The end of the drive winding NB2 is connected to the base of the switching element Q2 via the resonance capacitor CB2, the base current limiting resistor RB2, and the switch S. That is, the drive winding NB1, the base current limiting resistor RB1, and the resonance capacitor CB1 form a self-excited oscillation driving circuit for the switching element Q1, and the drive winding NB2 and the resonance capacitor C
Switching element Q by B2 and base current limiting resistor RB2
Form a self-excited oscillation drive circuit for 2.

In the detection winding ND, an alternating voltage corresponding to a switching output is detected by a switching operation described later. In the drive windings NB1 and NB2, alternating voltages having opposite polarities different from each other by 180 ° can be obtained in accordance with the switching output detected by the detection winding ND.

The switching elements Q1 and Q2 provided for single-end operation or push-pull operation include:
High breakdown voltage bipolar transistors (BJTs: junction transistors) are employed.

The drive circuit (drive winding NB1-base current limiting resistor RB1-resonance capacitor CB1), clamp diode DD1, and parallel resonance capacitor Cr1 are connected to the switching element Q1 as shown in FIG. Is connected to a drive circuit (drive winding NB2-base current limiting resistor RB2-resonance capacitor CB2), clamp diode DD2, and parallel resonance capacitor Cr2 as shown in the figure.

Here, the clamp diodes DD1, DD2
Are connected in parallel with each other between the base and collector of the switching elements Q1 and Q2. The parallel resonance capacitor Cr1 is connected between the collector and the emitter of the switching element Q1. Also the parallel resonance capacitor Cr
2 is connected between the collector and the emitter of the switching element Q2. Further, the emitters of the switching elements Q1 and Q2 are respectively connected to the primary side ground.

The push-pull operation and the single-end operation are selected based on the voltage excited in the tertiary winding N5 formed in the insulating converter transformer PIT. As will be described in detail later, the electromagnetic relay RY is driven by the alternating voltage excited by the tertiary winding N5, and whether the drive current is supplied to the switching element Q2 by switching the switch S by the electromagnetic relay RY is determined. Is selectively switched. For example, when the switch S is off, the drive circuit (drive winding NB2-resonant capacitor CB2-base current limiting resistor RB2) does not supply the drive current to the base of the switching element Q2, and the switching element Q2 operates. The switching drive is performed in a single-ended operation in which only the switching element Q1 is operated. When the switch S is on, a drive current is supplied from the drive circuit to the base of the switching element Q2, and the switching drive is performed by a push-pull operation in which both the switching element Q1 and the switching element Q2 operate. Done.

The starting resistor Rs is connected between the positive electrode of the smoothing capacitor Ci and the base of the switching element Q1. That is, the drive circuit (drive winding NB1-resonant capacitor CB1) is connected to the switching element Q1.
Connected in parallel with the base current limiting resistor RB1). In this case, the startup resistor Rs is inserted between the positive terminal of the smoothing capacitor Ci and the base of the switching element Q1, so that at startup, the current supplied from the voltage Ei that is the voltage across the smoothing capacitor Ci is used as the startup current. , To the base of the switching element Q2.

The insulating converter transformer PIT for transmitting the switching output of the switching elements Q1 and Q2 to the secondary side is, as shown in FIG. 2, for example, E-type cores CR1 and CR2 made of a ferrite material.
An EE-type core is provided so that the magnetic legs of the EE-type core are opposed to each other. A primary winding N1 (N1A, N1B) and a tertiary winding The wire N5 and the secondary winding N2 are wound separately. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is the center magnetic leg of E-shaped cores CR1 and CR2.
It can be formed by forming it shorter than the outer magnetic leg of the book. Further, as the coupling coefficient k, for example, k５0.85
And a saturated state is hardly obtained.

In this case, the insulation converter transformer PIT
Is divided into primary windings N1A and N1B, and one end of the primary winding N1A is connected to a smoothing capacitor C1 through a series connection of an inductance winding Lc of a choke coil CH.
i, and the other end is connected to the collector of the switching element Q1 via the detection winding ND and the magnetic coupling transformer MCT of the power factor improvement circuit 10. Primary winding N1
One end of B is connected to the collector of the switching element Q2, and the other end is connected to the positive electrode of the smoothing capacitor Ci via the series connection of the inductance winding Lc of the choke coil CH.

In this case, the above-described parallel resonance capacitor C
r1 is the leakage inductance component of the primary winding N1A (L1
A) and the combined inductance (L1A + Lc) of the inductance winding Lc form a parallel resonance circuit for causing the switching element Q1 to operate in a voltage resonance type. Similarly, the parallel resonance capacitor Cr2 is connected to the primary winding N1B.
And the combined inductance (L1B + Lc) of the inductance winding Lc and the leakage inductance component (L1B), form a parallel resonance circuit for causing the switching element Q2 to operate in a voltage resonance type. Although not described in detail here, when each of the switching elements Q1 and Q2 is turned off, the voltage across the resonance capacitors Cr1 and Cr2 actually becomes a sinusoidal pulse waveform due to the action of these parallel resonance circuits. A resonance type operation is obtained.

Here, the configuration of the relay drive circuit 40 and the operation related to the relay drive circuit 40 will be described. The switch S is on / off controlled by the electromagnetic relay RY. The electromagnetic relay RY is driven by the relay drive circuit 4. The relay drive circuit 4 is configured by connecting resistors R1 to R4, a Zener diode ZD, capacitors C5 and C6, and a diode D4 as shown in the figure. In this relay drive circuit 4, resistors R1 and R2 are connected in series between the positive electrode of the smoothing capacitor Ci and the primary side ground. The voltage dividing point of the resistors R1 and R2 and the transistor Q
A Zener diode ZD is inserted between the bases of the third.
In this case, when the rectified and smoothed voltage appearing in the smoothing capacitor Ci is 150 V AC or more, the above components are selected so that the Zener diode ZD conducts by the voltage value divided by the resistors R1 and R2. You. That is, the AC input voltage level is AC1
A voltage detection circuit for detecting whether the voltage is 50 V or more is formed. The transistor Q3 drives the electromagnetic relay RY. A protection diode D4 for flowing a reverse current is connected in parallel to the electromagnetic relay RY.

As an operating power supply for operating the relay drive circuit 4, a tertiary winding N5 is wound around the insulating converter transformer PIT, and the alternating voltage excited by the tertiary winding N5 is supplied to a rectifier diode D3. And a DC voltage obtained by rectification by a half-wave rectifier circuit comprising a capacitor C6.

When an AC input voltage VAC of 150 V AC or less is supplied as a 100 V AC system by such a relay driving circuit 4, for example, the Zener diode Z
Since D does not conduct, the transistor Q3 is turned on by allowing the base current to flow through the resistor R3. Thereby, the emitter current is conducted to the electromagnetic relay RY. Then, the switch S is turned on by the exciting action of the electromagnetic relay RY. This allows
As described above, the switching element Q2 is in the operating state, and the push-pull operation by the switching operation of both the switching element Q1 and the switching element Q2 is performed.

On the other hand, as an AC200V system, AC150V
When the AC input voltage VAC is supplied, the Zener diode ZD is turned on and the transistor Q3 is turned off. Therefore, the emitter current of the transistor Q3 does not flow through the electromagnetic relay RY, and the switch S is turned off. That is, the switching element Q2 is in a non-operation state. When the switching element Q2 is in a non-operating state, the switching power supply circuit performs a single-ended operation of only the switching operation by the switching element Q1.

In such a configuration on the primary side, for example, 20
In the switching operation based on the single-ended operation, which is an operation of the 0 V system, a drive current (base current) as an alternating current is supplied to each base of the switching element Q1 in the self-excited oscillation drive circuit including the drive winding NB1. by this,
The switching element Q1 performs an operation of turning on / off at a switching frequency determined by a constant of the self-excited oscillation drive circuit. That is, a switching operation by a voltage resonance type and a single end operation can be obtained.
Then, depending on the switching operation of the switching element Q1, the collector of the switching element Q1 → the emitter of the switching element Q1 → the smoothing capacitor Ci → the choke coil CH → the primary winding N1A → the detection winding ND → the primary winding of the magnetic coupling transformer MCT. The switching output current flows through the path of Np.

Further, for example, the switching operation by the push-pull operation which is the operation of the 100 V system is performed by the driving winding N
Since alternating voltages of opposite polarities are obtained in B1 and the driving winding NB2, the self-excited oscillation driving circuit including the driving winding NB1 and the self-excited oscillation driving circuit including the driving winding NB2 respectively have opposite polarities. , A drive current (base current) as an alternating current flows through each base of the switching elements Q1 and Q2. Thereby, the switching element Q1,
As for Q2, an operation of alternately turning on / off at a switching frequency determined by a constant of the self-excited oscillation driving circuit is obtained. That is, a switching operation by a voltage resonance type and push-pull is obtained.

Then, depending on the switching operation on the switching element Q1 side, a switching output current flows through the same path as in the single-ended operation described above, and depending on the switching operation on the switching element Q2 side, the collector of the switching element Q2 → the switching element Q2 , A smoothing capacitor Ci, a choke coil CH, and a primary winding N1B.

On the secondary side of the insulated converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. . In the present specification, a switching converter having a configuration in which a resonance circuit is provided for the primary side and the secondary side and operates as described above is described below.
It is also referred to as a “composite resonance type switching converter”.

In this case, for the secondary parallel resonance circuit formed as described above, a tap is provided for the secondary winding N2, and then the rectifier diodes DO1, DO2, DO are provided.
3, DO4 and the smoothing capacitors CO1, CO2 are connected as shown in the figure, so that a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies, as a control current, a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT, as will be described later. The constant voltage control is performed as follows.

By the way, the insulation converter transformer PIT
, The polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2, DO3, DO4)
, The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 may be + M or -M. For example, when the connection configuration shown in FIG. 3A is adopted, the mutual inductance is + M (additive polarity: forward method), and when the connection configuration shown in FIG. 3B is employed, the mutual inductance is −M (depolarization: Flyback method). If this is made to correspond to the operation of the secondary side of the power supply circuit shown in FIG. 1, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the rectifier diode DO1 (DO3)
Can be regarded as an operation mode of + M (forward mode). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectification diode DO2 (D
The operation in which the rectified current flows through O4) can be regarded as the -M operation mode (flyback method). That is, in this power supply circuit, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance becomes + M / -M
Mode.

The control circuit 1 changes the level of the control current (DC current) flowing through the control winding NC in accordance with the change in the secondary side DC output voltage level (EO1), thereby winding the control coil PRT on the orthogonal control transformer PRT. The inductances LB1 and LB2 of the mounted drive windings NB1 and NB2 are variably controlled. As a result, in the case of the single-ended operation, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB1 of the drive winding NB changes, In the case of operation, the drive winding NB1,
The resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching elements Q1 and Q2 formed including the inductances LB1 and LB2 of NB2 changes. This is an operation of changing the switching frequency of the switching elements Q1 and Q2. This operation has an effect of stabilizing the secondary-side DC output voltage.

Next, the configuration of the power factor correction circuit 10 will be described. The power factor correction circuit 10 shown in FIG. 1 includes a filter capacitor CN, a high speed recovery type diode D1, and a magnetic coupling transformer MCT. Here, the magnetic coupling transformer MCT is configured by winding a primary winding (first winding) Np and a secondary winding (second winding) Ns in a tightly coupled state, for example. Further, in the present embodiment, assuming that the inductance Ls of the primary winding Np and the inductance Ls of the secondary winding Ns take a larger value than the inductance Lp, A predetermined inductance value is selected. For this purpose, for example, the number of turns (turn ratio) of the primary winding Np and the secondary winding Ns is set according to the inductance value to be actually selected.

The winding start end of the primary winding Np of the magnetic coupling transformer MCT is connected to the primary winding N1A of the insulating converter transformer PIT via the detection winding ND, and the winding end is connected to the collector of the switching element Q1. Connected to That is, it is connected in series with the primary winding N1A. As a result, the switching output of the switching element Q1 is transmitted to the primary winding Np of the magnetic coupling transformer MCT via the detection winding ND and the primary winding N1A.

The winding start end of the secondary winding Ns of the magnetic coupling transformer MCT is connected to the cathode of the high-speed recovery type diode D1. Here, the anode of the fast recovery diode D1 is connected to the positive output terminal of the bridge rectifier circuit Di. The winding end of the secondary winding Ns is connected to the positive terminal of the smoothing capacitor Ci. That is, a series connection circuit of the high-speed recovery type diode D1 and the secondary winding Ns is inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci (that is, the rectified current path). The primary winding Np and the secondary winding Ns of the magnetic coupling transformer MCT shown in FIG.
3A, the operation is the same as that of FIG. 3A, and the operation is in the polarity (+ M) mode.

Further, in this case, the filter capacitor CN
Is inserted between the anode side of the high-speed recovery type diode D1 and the primary side ground to form, for example, a normal mode low-pass filter together with the secondary winding Ns of the magnetic coupling transformer MCT.

The power factor improving operation by the power factor improving circuit 10 having such a configuration is basically as follows. In the power factor correction circuit 10, first, the isolation converter transformer P
The switching output of switching element Q1 obtained in IT primary winding N1A is transmitted to primary winding Np of magnetically coupled transformer MCT. Then, via the magnetic coupling in the magnetic coupling transformer MCT, the switching output obtained in the primary winding Np is excited with respect to the secondary winding Ns. That is, the switching output is fed back to the rectified current path by the magnetic coupling of the magnetic coupling transformer MCT.

With the switching output thus fed back, the high-speed recovery type diode D1 operates so that the rectified current is intermittently switched in the switching cycle. Due to this intermittent operation, the inductance LS of the secondary winding Ns functioning as a choke coil also increases, and the charging current to the smoothing capacitor Ci also increases during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci. Flows. Therefore, also in this case, as a result, the conduction angle of the AC input current is enlarged, and the power factor is improved.

FIG. 4 shows operation waveforms of the power supply circuit shown in FIG. Here, FIGS. 4A to 4E show the operation of each unit in the commercial power supply cycle, and FIGS. 4F to 4J show the operation of each unit in the switching cycle. This operation is performed under the condition that the AC input voltage VAC = 100 V and the maximum load power 2
The operation is a push-pull operation at 90 W.

Here, the frequency of the commercial power supply is 50 Hz, and the AC input voltage VAC has a sinusoidal waveform having a half cycle of 10 ms as shown in FIG. 4A.
As the rectified current flowing through the bridge rectifier circuit Di, when the AC input current IAC flows as shown in FIG. 4B, the high-speed recovery type diode D1 switches so as to intermittently switch the current. The switching current ID having the waveform shown in FIG.

Here, the operation of the switching cycle during the period when the AC input voltage VAC is high and the AC input current IAC flows is as follows. For example, assuming that the switching element Q1 performs a switching operation during this period, both ends of the parallel resonance capacitor Cr are connected to each other as shown in FIG.
As shown in (f), in the period Toff (3 μs) during which the switching element Q1 is turned off, a parallel resonance voltage Vcp that becomes a sine wave pulse is generated. Then, during a period Ton (7 μs) during which the switching element Q1 is turned on,
4 between the collector and the emitter of the switching element Q1.
As shown in (g), the switching output current Icp flows, and this switching output current Icp is smoothed from the filter capacitor CN in the power factor correction circuit 10 via the high-speed recovery type diode D1 → secondary winding Ns. It flows to the capacitor Ci. At this time, the switching current ID flowing through the high-speed recovery type diode D1 has a waveform having a level of 11 Ap on a substantially sine wave as shown in FIG.

Further, as shown in FIG. 4 (i), the voltage VL between both ends of the secondary winding Ns in the switching cycle becomes a sine waveform having a positive direction of 100 Vp in the period Toff, and in the period Ton in the period Toff. In the start period, a waveform of 60 Vp in the negative direction is obtained. FIG. 4 (j) shows the secondary winding Ns in the switching cycle.
-The voltage V1 across the series connection circuit of the smoothing capacitor Ci is shown, and this voltage V1 is
A sinusoidal waveform of 260 Vp is obtained, and in a period Ton, a sinusoidal waveform of 100 Vp in the reverse direction is obtained in the start period, and thereafter, 150 V is maintained.

Then, as an operation based on the commercial power supply cycle,
In a period in which the AC input voltage VAC is low and the AC input current IAC does not flow (a period in which the high-speed recovery type diode D1 does not perform a switching operation), the isolation converter transformer is connected to the primary winding Np of the magnetic coupling transformer. A primary-side switching current I1 flowing through the primary winding N1 of the PIT flows, thereby generating an excitation voltage in the secondary winding Ns of the magnetic coupling transformer MCT. Thereby, the voltage VL across the secondary winding Ns as viewed in the commercial power supply cycle is
FIG. 4D shows that the voltage level is almost constantly obtained. And, this secondary winding Ns
Of the rectified smoothed voltage Ei (smoothing capacitor C
(e), the voltage V1 across the series connection circuit of the secondary winding Ns and the smoothing capacitor Ci has the waveform shown in FIG. 4E in the commercial power supply cycle.
As can be seen from the waveform shown in FIG. 4 (e), in the present embodiment, the level of the rectified smoothed voltage Ei (DC input voltage) is increased by the voltage excited in the secondary winding Ns of the magnetic coupling transformer MCT. It is working to make it. This operation is performed, as described above, by the inductance Lp of the primary winding Np of the magnetic coupling transformer MCT.
And the inductance Ls of the secondary winding Ns is obtained by setting the inductance Ls to be larger than the inductance Lp.

As described above, by increasing the level of the DC input voltage, for example, the cathode potential V1 of the high-speed recovery type diode D1 changes to the input voltage (the high-speed recovery type diode D1) of the power factor correction circuit. The high-speed recovery diode D1 operates so as to continue the switching operation even during a period that is lower than the anode side potential). In other words, the switching period of the high-speed recovery type diode D1 is extended every half cycle of the commercial power supply cycle. By such an operation, for example, for a half cycle of the commercial power supply of 10 ms, FIG.
As shown in (b), the conduction angle of the AC input current IAC is 6 ms.
To a higher degree, so that a higher power factor is obtained.

Here, FIGS. 5, 6 and 7 show experimental results of the power supply circuit shown in FIG. In addition,
To obtain the experimental results shown in these figures, an EI type core of EI-25 was used for the magnetic coupling transformer MCT, and the inductance Lp = 1 for the primary winding Np.
3 μH, the inductance Lp = 1 for the secondary winding Ns.
05 μH. And the filter capacitor CN
= 1 μF, the parallel resonance capacitor Cr = 2700 pF, and the switching elements Q1 and Q2 are selected so that VCBO> 1500V. The operating condition is that the load power Po = 29
0 W to 0 W and the AC input voltage VAC = 80 V to 288 V.

First, FIG. 5 shows that the AC input voltage VAC = 100
4 shows the relationship between the load and the power factor under the condition that V and the AC input voltage VAC = 230 V are constant. As shown in this figure, in the present embodiment, the load power Po = 1
From about 00 W or more, the power factor PF is maintained at about 0.75 or more. Then, the characteristic that the power factor gradually increases until the power factor PF becomes approximately 0.8 at the load power Po = 300 W is obtained. Thus, in the present embodiment, a high power factor can be obtained over a wide range of load conditions.

FIG. 6 shows that the load power Po = 290 W
2 shows the relationship between the AC input voltage VAC and the power factor under the condition that is constant. As can be seen from this figure, as the power factor PF, a high power factor of about PF = 0.9 or more has already been obtained near the AC input voltage VAC = 100 V.
As the AC input voltage VAC fluctuates so as to increase, the power factor PF decreases, but even when the AC input voltage VAC = 230 V, PF =
A value close to 0.8 is maintained.

As described above, in the power supply circuit of the present embodiment, in the single-ended operation and the push-pull operation,
The switching output fed back to the rectified current path by the magnetic coupling of the magnetic coupling transformer MCT becomes substantially the same. Therefore, as shown in FIG. 5 and FIG.
Power factor PF even when is 100V system or 200V system
Have almost the same characteristics. Further, in the power supply circuit of the present embodiment, a high power factor can be maintained even when the AC input voltage and the load fluctuate. For this reason, the power supply circuit according to the present embodiment is not limited to a television receiver or the like in which the AC input voltage or the load condition is specified, and is applied to office equipment such as an office equipment or a personal computer in which the load condition fluctuates. It is practically possible to mount it.

Further, according to the power supply circuit of the present embodiment, the power factor PF becomes very stable with respect to changes in the load power Po and the AC input voltage VAC, as compared with the conventional power feedback system. Further, the fluctuation range of the load power Po is greatly expanded. As described above, the AC input voltage and the fact that the high power factor can be maintained even when the load fluctuates are not limited to the television receiver or the like in which the AC input voltage and the load condition are specified.
For example, it is practically sufficiently possible to mount the power supply circuit according to the present embodiment on office equipment such as office equipment and personal computers in which load conditions fluctuate.
Further, as a configuration corresponding to a wide range, an AC input voltage V for switching between push-pull operation / single-end operation is provided.
The detection of AC is performed from the DC input voltage Ei appearing on the positive electrode of the smoothing capacitor Ci. Therefore, even if an erroneous detection of the commercial AC current level occurs due to, for example, an instantaneous power failure or noise from disturbance, only switching between push-pull operation and single-ended operation is performed, and DC power as an operation power supply for switching is obtained. The switching operation can be continued without causing a level change of the input voltage Ei. That is, in the components such as the switching elements Q1 and Q2, a withstand voltage product that can cope with the AC input voltage VAC due to, for example, a momentary power failure or disturbance noise is selected, and the switching elements Q1 and Q2 are protected from erroneous detection. There is no need to take measures such as adopting a configuration. Therefore, it is not necessary to take a countermeasure against malfunction, so that it is possible to prevent an increase in circuit scale.

Further, in the power supply circuit of the present embodiment, the power conversion efficiency increases when the AC input voltage is VAC = 100 V and VAC = 230 V, for example, 1.5 times higher than when the power factor improvement circuit 10 is not provided. %, And as AC input power,
The power is reduced by about 4.4 W as compared with the case where the power factor improving circuit 10 is not provided. This is because, as a configuration of the power factor correction circuit 10 of the present embodiment, the magnetic coupling transformer MCT
By having. For example, in the configuration of the power factor improvement circuit 20 illustrated in FIG. 14, the switching output branches and flows to the primary side series resonance circuit and the LC resonance circuit for feeding back power to the power factor improvement circuit 20. Therefore, the switching current increases, and the power loss increases accordingly. On the other hand, in the present embodiment, the switching output of the switching element Q1 is simply
If it is transmitted to the primary winding NP of T, the magnetic coupling transformer M
The feedback of the switching output to the rectified current path is performed by the magnetic coupling action of the CT. This allows
In achieving the same power factor improvement, the amount of switching current can be made smaller than in the case of FIG.

As an experimental result, feedback of the switching current I1 from the primary winding Np to the secondary winding Ns by changing the ratio of the inductance values between the windings (Np, Ns) of the magnetic coupling transformer MCT. It has been confirmed that the power factor characteristics can be changed by changing the amount. That is, in the present embodiment, the winding (N
By changing the inductance value of (p, Ns), the adjustment can be performed so that the power factor characteristics under conditions suitable for actual use can be obtained. In order to change the ratio of the inductance value of the primary winding Np / secondary winding Ns of the magnetic coupling transformer MCT as described above, for example, the winding ratio of the primary winding Np / secondary winding Ns may be changed. It is.

Here, as the inductance value of the primary winding Np / secondary winding Ns in the magnetic coupling transformer MCT,
For the primary winding Np, the inductance Lp = 13 μH,
FIG. 7 shows operation waveforms of the respective parts obtained when the turns ratio of the secondary winding Ns is set such that the inductance Lp = 90 μH.

As shown in FIG. 7A, for example, 50 Hz
Assuming that the AC input voltage VAC is input in the commercial power supply cycle of FIG.
The switching operation is performed as indicated by the switching current ID in (c). The voltage VL across the secondary winding Ns of the magnetic coupling transformer MCT at this time is as shown in FIG.
As shown in, the AC input voltage VAC is assumed to be low.
The alternating waveform corresponding to the switching cycle is obtained substantially corresponding to the period other than the period, and the voltage (DC input voltage) V1 obtained in the series connection circuit of the secondary winding Ns and the smoothing capacitor Ci is the AC voltage. A waveform substantially the same as the voltage VL is superimposed on a voltage level of 130 Vp during the τ period when the input voltage VAC is assumed to be high, and 40 V during periods other than the τ period.

When the winding ratio of the magnetic coupling transformer MCT is changed as described above, the AC input voltage is changed to VAC.
FIG. 8 shows the relationship between the load and the power factor under the condition that the constant was set at 100 V and VAC = 230 V. For example, in comparison with FIG. 5, in the case of FIG. 8, in the case of the load power Po = 100 W or more, although the value of the power factor PF becomes lower, the power factor PF becomes constant at about PF = 0.75 regardless of the fluctuation of the load power. When the load power is maintained at about Po = 100 W or less, the power factor increases.

FIG. 9 shows that the load power is Po = 10 in the configuration in which the winding ratio of the magnetic coupling transformer MCT is also changed.
The relationship between the AC input voltage VAC and the power factor under the condition that the power is constant at 0 W and Po = 290 W is shown. Also in this case, as for the power factor PF, when the load power Po = 100 W, the power factor P
The characteristic that the power factor PF becomes substantially constant when F = 0.75 or less and the power factor PF becomes high when VAC = about 100 V or less is obtained. When the load power Po = 290 W, the AC input voltage VAC
When the power factor PF is about 200 V or more, the power factor PF becomes substantially constant at about 0.75, and when the VAC is about 200 V or less, the power factor PF becomes high.

In the configuration of the switching power supply circuit shown in FIG. 1, primary winding N of magnetically coupled transformer MCT
Regarding the relationship between p and the winding direction of the secondary winding Ns, the relationship shown in FIG. 3B, that is, even when the magnetic coupling transformer MCT is configured to operate with a reduced polarity (−M: flyback method), The characteristics shown in FIGS. 5 and 6 can be obtained. Further, by changing the turns ratio between the primary winding and the secondary winding of the magnetic coupling transformer MCT, for example, the characteristics shown in FIGS. 8 and 9 can be obtained. In this case, when the switching element Q1 shown in FIG.
In (f), the switching current ID flows through the high-speed recovery type diode D1, and the phases of the high-frequency waveforms of the terminal voltage VL and the terminal voltage V1 are reversed.

Although the embodiments have been described above, the present invention may have various other modifications. For example, the present applicant has already proposed a configuration provided with a quadruple voltage rectifier circuit using a secondary-side series resonance circuit as a composite resonance type switching converter, but such a configuration is also a modification of the present embodiment. It can be established as an example. That is, the present embodiment is not particularly limited as the configuration of the secondary-side resonance circuit and the rectifier circuit.

Further, in the present embodiment, an example has been described in which the switching output of switching element Q1 is fed back to power factor correction circuit 10 when the push-pull operation is performed, but the switching output of switching element Q2 is performed. Is fed back to the power factor correction circuit 10. Furthermore, in the present embodiment, an orthogonal control transformer is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the control transformer, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, for example, as in the case of the orthogonal control transformer, a three-dimensional structure is obtained by combining two sets of double U-shaped cores having four magnetic legs. Form a mold core. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, the control winding N
C and one of the drive windings NB are wound around two magnetic legs adjacent to each other among the four magnetic legs, and the other winding is wound diagonally. It is assumed that there is a positional relationship 2
It is wound around the magnetic legs of the book. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. Thereby, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

[0101]

As described above, the present invention is provided with a power factor improving circuit having a magnetic coupling transformer in a rectified current path of a power supply circuit called a composite frequency converter of a push-pull type switching frequency control type. The switching output obtained in the primary side resonance circuit is fed back via magnetic coupling by a magnetic coupling transformer, and the switching operation switching means switches between push-pull operation and single-ended operation according to the AC input voltage. . As a result, there is an effect that a wide range of input voltage can be supported and a sufficient power factor can be maintained over a wide range with respect to fluctuations of the AC input voltage and the load power. For example, A
When a heavy load with a load power of 200 W or more is used for both the C100 V system and the 200 V system, a high power factor can be stably obtained with respect to a wide range of load fluctuation of 290 W to 0 W. Therefore, it is suitable as a power factor improving soft switching power supply corresponding to a wide range shared by 100 V system and 200 V system, and suitable as a power factor improving power supply circuit for office equipment and information equipment having a large load variation.

Further, since the power factor improving means includes a magnetic coupling transformer, for example, if the first winding of the magnetic coupling transformer is inserted into a system to supply the switching output to the primary winding, It is possible to realize feedback of the switching output to the rectified current path. this is,
For example, it means that it is not necessary to add a resonance circuit system for feeding back the switching output to the rectification current path, and accordingly, the switching current flowing through the switching element can be reduced and the power conversion efficiency can be improved. Can be planned. Further, since the power factor improving means is not inserted in the commercial power supply line and is inserted between the rectified output point and the smoothing capacitor, it is not necessary to select a safety standard approved product, so that it can be configured at a low cost. become. Further, according to the present invention, switching between push-pull operation and single-ended operation is performed as a configuration corresponding to a wide range. This is because, for example, in the detection circuit system for detecting the level of the commercial AC power supply, even if an erroneous detection occurs due to an instantaneous power failure or noise of disturbance, only switching between push-pull operation / single-end operation is performed. It does not cause a change in the DC input voltage level which is an operation power supply for switching due to the erroneous detection. Therefore, for example, since it is not necessary to take a protective measure corresponding to, for example, an erroneous detection, the circuit scale can be prevented from being increased.

Further, by configuring the primary side resonance circuit as a voltage resonance type, a single-ended operation using the first switching element in accordance with the AC input voltage and a push-type operation using the first and second switching elements can be performed. By switching the pull operation, it is possible to support a wide range.
This makes it possible to reduce the number of components compared to a power supply circuit configured to perform switching operation by switching between half-bridge coupling and full-bridge coupling, for example, by using the primary-side resonance circuit as a current resonance type. Therefore, the switching current can be reduced, and the power conversion efficiency can be improved. Further, by reducing the number of components, the circuit configuration can be simplified, and the board area can be reduced.

In the magnetic coupling transformer, the first
By setting the inductance of the second winding to be larger than the inductance of the winding, the DC input voltage increases. As a result, as a result, the AC input voltage and the load power are reduced. Even with fluctuations, a power factor sufficient for practical use is maintained. Therefore, for example, it becomes suitable as a power factor improving power supply circuit for office equipment and information equipment having a large load variation.
In addition, the power factor characteristics can be made substantially constant with respect to fluctuations in the AC input voltage and the load power by adjusting, for example, the ratio of the inductance of the first winding to the inductance of the second winding. Therefore, the ripple component can be suppressed. In addition, the power conversion efficiency is improved by increasing the DC input voltage.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention.

FIG. 2 is a side sectional view showing a structure of an insulating converter transformer employed in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing each operation when the mutual inductance is + M / -M.

4 is a waveform chart showing a switching operation by a push-pull operation of the switching power supply circuit shown in FIG.

FIG. 5 is a characteristic diagram showing a relationship between load power and power factor for the switching power supply circuit shown in FIG.

6 is a characteristic diagram showing a relationship between an AC input voltage and a power factor for the switching power supply circuit shown in FIG.

FIG. 7 is a waveform chart showing an operation of the switching power supply circuit shown in FIG. 1 when the winding ratio of the magnetic coupling transformer is changed.

FIG. 8 is a characteristic diagram showing a relationship between load power and power factor of the switching power supply circuit shown in FIG. 1 when the winding ratio of the magnetic coupling transformer is changed.

9 is a characteristic diagram showing a relationship between an AC input voltage and a power factor of the switching power supply circuit shown in FIG. 1 when a winding ratio of a magnetic coupling transformer is changed.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 11 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 12 is a waveform chart showing an operation of the power supply circuit of FIGS. 10 and 11;

FIG. 13 is a characteristic diagram showing a relationship between an AC input voltage and a power factor of the power supply circuits of FIGS. 10 and 11;

FIG. 14 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

15 is a characteristic diagram showing a relationship between load power and a power factor and a relationship between an AC input voltage and a power factor for the power supply circuit of FIG. 14;

FIG. 16 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

17 is a characteristic diagram showing the relationship between load power and power factor and the relationship between AC input voltage and power factor for the power supply circuit of FIG.

FIG. 18 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 19 is a characteristic diagram illustrating a relationship between an AC input voltage and a power factor for the power supply circuit of FIG. 18;

[Explanation of symbols]

1 control circuit, 4 relay drive circuit, 10 power factor improvement circuit, Di bridge rectifier circuit, Ci smoothing capacitor,
D1 High-speed recovery type diode, MCT magnetic coupling transformer, Cr1, Cr2 parallel resonance capacitor, C2
Secondary parallel resonance capacitor, PRT orthogonal control transformer, PIT isolation converter transformer, Q1, Q2 switching element, Q3 transistor, RY electromagnetic relay, S switch

Claims (3)

[Claims] A gap is formed between a rectifying / smoothing means for inputting a commercial AC power supply to generate a rectified / smoothed voltage and outputting the rectified / smoothed voltage as a DC input voltage, and a required coupling coefficient to be loosely coupled. An insulating converter transformer provided for transmitting a primary side output to a secondary side; and first and second insulating converter transformers configured to intermittently output the DC input voltage by a push-pull operation to output to a primary winding of the insulating converter transformer. 2 switching means, and at least a leakage inductance component including a primary winding of the insulated converter transformer and a capacitance of a primary side parallel resonance capacitor, and the operation of the first and second switching means is a voltage resonance type. First and second primary side resonance circuits, a first winding connected to the primary winding, and a rectifying current path A power factor improving means for performing a power factor improving operation by at least including a magnetic coupling transformer for magnetically coupling the second winding and a switching element inserted into a rectified current path; By stopping the operation of the second switching means, the DC input voltage is intermittently output by a single-ended operation by only the first switching means and output to the primary winding of the insulating converter transformer. Switching operation switching means, a secondary resonance circuit formed on the secondary side by a leakage inductance component of a secondary winding of the insulating converter transformer, and a capacitance of a secondary resonance capacitor; Input the alternating voltage that is formed including the circuit and obtained in the secondary winding of the above-mentioned insulated converter transformer. A DC output voltage generating means configured to generate a secondary DC output voltage by performing a rectification operation; and a constant voltage for the secondary DC output voltage according to the level of the secondary DC output voltage. A switching power supply circuit, comprising: a constant voltage control unit configured to perform control. 2. The switching power supply circuit according to claim 1, wherein in the magnetic coupling transformer, the second winding has a predetermined inductance larger than that of the first winding. . 3. A turns ratio between the first winding and the second winding so that a required value is obtained for each of the inductance values of the first winding and the second winding of the magnetic coupling transformer. The switching power supply circuit according to claim 1, wherein the switching power supply can be changed and set.