JP2005006028A

JP2005006028A - Temperature compensated piezoelectric oscillator

Info

Publication number: JP2005006028A
Application number: JP2003167102A
Authority: JP
Inventors: Tomio Sato; 富雄佐藤
Original assignee: Toyo Communication Equipment Co Ltd
Current assignee: Toyo Communication Equipment Co Ltd
Priority date: 2003-06-11
Filing date: 2003-06-11
Publication date: 2005-01-06

Abstract

<P>PROBLEM TO BE SOLVED: To provide a temperature compensation system which has simple circuit configuration and high temperature compensation accuracy and consequently reduces noise and is made low-cost and is suitable for a portable telephone or the like. <P>SOLUTION: The temperature compensation system includes: a crystal oscillation circuit 12 provided with a crystal vibrator 11; a varactor diode 14 whose capacitance varies with a voltage across the varactor diode 14; high resistance resistors 16 to 22 for isolating each circuit; capacitors 1, 2, 4, 5; a temperature sensor section 101 for sensing ambient temperature; compensation voltage generating circuits 102, 103 for generating a temperature compensation voltage on the basis of a voltage of the temperature sensor section 101; and a reference voltage generating section 104 for applying a reference voltage of the compensation voltage generating circuits 102, 103 to them. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

【０００１】
【発明の属する技術分野】
本発明は、水晶振動子等の圧電振動子を使用した圧電発振器に関し、特に簡単な回路構成によってＡＴカット水晶振動子の発振周波数の温度補償が可能な温度補償型圧電発振器に関するものである。
【０００２】
【従来の技術】
携帯電話に代表される陸上移動体通信器の使用エリアは拡大の一途を辿っている。それと同時に、携帯電話の普及もすさまじく技術開発競争は激化している。携帯電話に使用される水晶発振器も小型化、ローコスト化、更に高性能化が要求されている。特にＧＰＳシステムとの共存を要求されるシステムでは温度特性が優れているだけでなく、低ノイズ化が強く要求されている。
図９に携帯電話に使用されている水晶振動子（Ａｔ−Ｃｕｔ）の切断角度の違いによる温度特性を示す。図に示す様に振動子の温度特性は３次関数に近い特性を示すが、これだけでは特性上十分ではなくこの特性以上の高い周波数安定度が得られるよう温度補償を行っていた。
従来の温度補償方式は大きく直接温度補償方式と間接温度補償方式に分けることができる。図１０は直接温度補償方式の補償回路の１例を示す図である。この方式は発振回路の発振ループ内にサーミスタと抵抗及びコンデンサにより補償回路を構成し、サーミスタの温度による抵抗変化をリアクタンスに変換し温度補償するもので、回路構成は非常にシンプルである。
また間接温度補償方式は、温度補償特性を得るための温度補償電圧を発生させる関数発生部を発振回路ループ外に設け、温度補償電圧を発振ループ内に設けた可変容量ダイオードに印加し温度補償を行うものである。そして温度補償電圧を発生する方法には、ＩＣ化技術を駆使し半導体の接合電位の温度特性を使用し多次関数を発生させる方式、補償電圧を予めメモリに保存し、温度変化に基づき電圧を印加するデジタル温度補償方式、演算増幅器を用いた分割線近似方式等がある。いずれにしても、補正電圧発生回路が非常に複雑になり、そのため回路からの雑音発生が大きく低ノイズ化が難しいといった問題がある。
【０００３】
図１１は従来技術として特許第３２５３２０７号に開示されている温度補償水晶発振器のブロック図である。本方式では温度センサ部１０１より温度に応じて変化する電圧出力を電圧関数発生回路１０２〜１０６へ入力する。５つの電圧関数発生回路を設けたので、電圧加算器１０７は各電圧関数発生回路からの計５本の入力電圧を合成し得られた図１２に示す様な折れ線関数の電圧を可変容量ダイオード１４へ入力している。図１３に２つの演算増幅器１１０、１１１で構成された１つの電圧関数発生回路を示す。
図１４は従来技術として実開昭６１−９５１０４号公報に開示されている温度補償水晶発振器のブロック図である。図１５は動作説明図を示す。図１５に示すように温度センサ２０１からの入力信号により電圧関数発生回路２０２の出力電圧は水晶振動子の３次曲線変化を呈する周波数温度特性の低温側頂点温度まで温度上昇と共に飽和電圧に達するまで単調上昇し、頂点温度以上の温度範囲では飽和電圧（一定電圧）となる。同様に電圧関数発生回路２０３の出力電圧は水晶振動子の周波数温度特性の高温側頂点温度まで一定でその頂点温度以上で温度上昇と共に単調上昇する。それぞれの出力電圧をダイオードＤ１４、１５のアノードに印加するので低温側では温度上昇に伴い容量が低下し、その結果、発振周波数を高くする。電圧関数発生回路２０２、２０３の出力電圧が共に一定である温度範囲内では、容量変化が無いので周波数の変化制御は行なわれず、高温側では温度上昇に伴い容量が増大し、その結果発振周波数を低下させる。また温度補償用コンデンサＣ１６、１７、１８、１９としては温度上昇に対し容量が単調に減少するような特性のものを用いて周波数が単調に上昇するよう制御する。即ち、先ほどの電圧関数発生回路２０２、２０３の出力電圧が共に一定である温度範囲内では、発振周波数は温度補償用コンデンサの影響により上昇する。尚、低温側、高温側ともこのコンデンサの影響を受けて周波数は上昇しているがコンデンサの影響を加味した変化量を補償していることは言うまでもない。
このように総合的には振動子の３次曲線を補償することができるが、本方式では頂点間温度の補正を容量、温度特性共にばらつきが大きい温度補償用コンデンサに頼った補償方式であるため、当然補償精度のばらつきが大きく量産化に適さないといった問題がある。
【特許文献１】特許第３２５３２０７号
【特許文献２】実開昭６１−９５１０４号公報
【０００４】
【発明が解決しようとする課題】
しかしながら上記の直接温度補償方式では回路は非常にシンプルであり、低ノイズという利点はあるが、補償曲線が頂点を持たない温度上昇に対し一方向増加の近３次曲線となるため、特に低温側、高温側の両サイドで極端に周波数補償量が増加するため優れた温度特性を得られ難いといった問題がある。
また特許文献１については、発振周波数温度特性に対し３次補償曲線をもとめ、曲線上に交点をもつ最適な直線補償方式を求め、全ての直線を任意に設定するという複雑な方法を取り入れているため、振動子の温度特性に非常に柔軟な対応が可能であり、優れた温度特性を得ることができるが、個々の直線電圧に対応した電圧関数発生回路が必要であるので回路構成が非常に複雑となり必然的にコストアップ、及び回路から発生するノイズが大きく位相雑音特性を悪化させる問題がある。
また特許文献２では簡易な直線補償方式を取り入れており、低ノイズ化は図られているが、温度補償コンデンサという、個体間の電気的特性のばらつきが大きいデバイスを使用しているため量産化に適さないと言う問題点がある。
本発明は、かかる課題に鑑み、簡単な回路構成でしかも温度補償精度が高く、それにより低ノイズ化が図れて低コストが可能な携帯電話等に適した温度補償方式を提供することを目的する。
【０００５】
【課題を解決するための手段】
本発明はかかる課題を解決するために、請求項１は、所定の周波数で励振される圧電素子を備えた圧電振動子と、該圧電素子に電流を流して励振させる発振用増幅器と、温度変化による発振周波数の変化を補償する周波数温度補償回路と、を備えた圧電発振器であって、前記周波数温度補償回路は、周囲温度によりパラメータが変化する温度検出部と、該温度検出部により変化したパラメータに基づいて２種類の異なる電圧を発生する温度補償用電圧発生部と、所定の基準電圧を発生する基準電圧発生部と、前記温度補償用電圧発生部の出力電圧に基づいて容量が変化する可変容量素子と、を備え、前記温度補償用電圧発生部は、前記圧電素子が有する温度特性の低温側の起点より前記低温側頂点を越えた点まで飽和電圧の略１／２の電圧を保持し、前記低温側頂点を越えた点より高温側頂点の手前まで単調増加し、前記高温側頂点の手前より前記飽和電圧により一定値となる電圧を連続的に発生する第１の制御電圧発生部と、前記圧電素子が有する温度特性の低温側の起点より前記低温側頂点の手前まで最低電圧から単調増加し、前記低温側頂点の手前より前記高温側頂点を越えた点まで前記飽和電圧の略１／２の電圧を保持し、前記高温側頂点を越えた点より前記高温側の終点まで単調増加して前記飽和電圧に達する電圧を連続的に発生する第２の制御電圧発生部とを備え、前記第１の制御電圧発生部及び第２の制御電圧発生部の出力端を前記可変容量素子の両端に印加することにより、前記圧電発振器の負荷容量を変化させて前記圧電振動子の発振周波数の温度特性を補償することを特徴とする。
ＡＴカットによる水晶振動子の温度特性は略３次関数的に変化する。この温度特性に対して、理想的には相反する特性をもつように、発振ループ内の負荷容量を変化させて補正することである。しかし、この特性を持つ回路を実現することは不可能ではないが、現実的に回路構成が非常に複雑となり、その分コストアップを避けることができず、さらに回路の複雑さからくる位相特性の劣化が問題となる。従って、回路構成は可能な限り簡単な程良いのは言うまでもない。しかし、従来の回路では低温と高温の端部での特性が実際より異なった特性となり、正確に補正することができなかった。そこで本発明では、合成した時に近似３次関数的に変化する２種類の電圧を発生させ、その電圧を可変容量素子の両端に印加することにより、可能な限りＡＴカットによる水晶振動子の温度特性に近づけるものである。
かかる発明によれば、合成した時に近似３次関数的に変化する２種類の制御電圧を発生させ、その電圧を可変容量素子の両端に印加するので、可変容量素子の容量変化によりＡＴカットによる水晶振動子の温度特性を正確に補正することができる。
【０００６】
請求項２は、前記第１の制御電圧発生部及び第２の制御電圧発生部の出力端のノイズを同レベルで且つ同相、若しくは異なるレベルで且つ同相になるようにしてノイズによる位相雑音を低減することを特徴とする。
前記第１の制御電圧発生部及び第２の制御電圧発生部の出力端の電圧が同じ発生源から発生されれば、その信号線に重畳されるノイズの位相は同相となる。また、ノイズレベルも大きく変化しないで略同レベルとなる。従って、これらの制御電圧を可変容量素子の両端に印加すると、ノイズによる電位差が発生しないので、相対的なノイズレベルを低下することができる。
かかる発明によれば、温度補償制御電圧を可変容量素子の両端に印加するので、ノイズの位相が同相になってノイズによる電位差がゼロとなり、発振回路の位相雑音を低減することができる。
請求項３は、前記第１の制御電圧発生部及び第２の制御電圧発生部の出力端を前記可変容量素子の両端に印加することにより、前記可変容量素子の両端にはＡＴカット水晶振動子の発振周波数偏差の近似３次関数に対し、温度変化に対して平行な２種類の温度補償電圧を含む５種類の直線温度補償電圧を発生することを特徴とする。
第１の制御電圧発生部からは、１種類の折れ線電圧と飽和電源電圧とグランド電圧の２種類の電圧を含めた３種類の電圧を発生し、第２の制御電圧発生部からは、２種類の折れ線電圧と飽和電源電圧の１／２の電圧を含めた３種類の電圧を発生し、全体として５種類の直線温度補償電圧を発生する。これらの電圧を可変容量素子の両端に印加すると、近似３次関数に近い電位差を発生する。それにより、可変容量素子の容量が変化して発振回路の負荷容量を変化させる。
かかる発明によれば、５種類の直線温度補償電圧を可変容量素子の両端に印加することで、近似３次関数に近い電位差を発生することができるので、簡易な回路構成により温度補償を実現することができ、コスト的にも安価に実現することができる。
請求項４は、前記可変容量素子はバラクタ、可変容量ダイオード若しくは、印加電圧により容量が可変する半導体デバイスを用いたことを特徴とする。
容量が外部の印加電圧により変化すれば可変容量素子として、可変容量ダイオード、接合型ＦＥＴのゲート、ソース又はゲート、ドレイン容量、ＭＯＳ型ＦＥＴのゲート、ソース又はゲート、ドレイン容量、バイポーラトランジスタのベース、エミッタ容量、又はベース、コレクタ容量を用いても本発明の発振器を構成することができる。
かかる発明によれば、可変容量素子として可変容量ダイオードや印加電圧により容量が可変する半導体デバイスを用いることもできるので、回路構成に幅が拡がり、それに伴って回路特性のバリエーションが広くなる。
【０００７】
【発明の実施の形態】
以下、本発明を図に示した実施形態を用いて詳細に説明する。但し、この実施形態に記載される構成要素、種類、組み合わせ、形状、その相対配置などは特定的な記載がない限り、この発明の範囲をそれのみに限定する主旨ではなく単なる説明例に過ぎない。
図１は本発明の実施形態に係る温度補償方式のブロック図である。この温度補償方式は、水晶振動子１１を備えた水晶発振回路１２と、両端の電位差により容量が変化する可変容量ダイオード１４と、各回路をアイソレーションする高抵抗１６〜２２と、コンデンサ１、２、４、５と、周囲温度を検出する温度センサ部１０１と、温度センサ部１０１の電圧に基づいて温度補償電圧を発生する補償電圧発生回路１０２、１０３と、補償電圧発生回路１０２、１０３の基準となる電圧を供給する基準電圧発生部１０４とを備えて構成される。
図２は図１の温度補償方式の動作を説明する説明図である。本実施形態では、温度センサ出力のほぼ常温、即ち振動子温度特性の変曲点温度に設定する。図の▲２▼に示す補償電圧発生回路１０２は低温側（ａ点）から水晶振動子の周波数温度特性の低温側頂点温度を越えた温度（ｃ点）まで約１／２Ｖｃｃの一定電圧であり温度（ｃ点）より高温側頂点の手前（ｄ点）まで温度上昇と共に単調増加し、温度点ｄにて飽和電圧Ｖｃｃに達する。補償電圧発生回路１０２の出力電圧を抵抗１６、１７で分圧され且つ抵抗１８を介して可変容量ダイオード１４のカソードに入力する。また図の▲３▼に示す補償電圧発生回路１０３は低温側（ａ点）より水晶振動子の周波数温度特性の低温側頂点温度の手前（ｂ点）まで単調増加し、温度（ｂ点）にて飽和電圧の約１／２Ｖｃｃに達する。その後、温度（ｂ点）から、高温側頂点を越えた（ｅ点）まで継続、再度高温側（ｆ点）まで単調増加し温度（ｆ点）にて飽和電圧Ｖｃｃに達する。補償電圧発生回路１０３の出力電圧は抵抗１９、２０、２１から成る分圧回路にて分圧され、且つ抵抗２２を介して可変容量ダイオード１４のアノードに入力する。その結果図の▲４▼に示す可変容量ダイオード１４のカソード、アノードの両端の電位差は、補償電圧発生回路１０２の出力電圧と補償電圧発生回路１０３の出力電圧の差に基づくものであるから、（ａ点）から（ｂ点）まで単調減少して（ｂ点）から（ｃ点）の間で一定となり（ｃ点）から（ｄ点）まで増加して（ｄ点）から（ｅ点）の間で再度一定であり、（ｅ点）から（ｆ点）まで単調減少する。そして、このような制御電圧に基づく可変容量ダイオード１４の端子間容量の制御によって、特性▲５▼のごとく周波数を変化させることができるので、この周波数変化と図の▲１▼に示す補償前の振動子１１の周波数温度特性とが相殺し、図の▲６▼に示す補償操作後の発振周波数の温度特性が得られる。
【０００８】
図３は本発明の温度補償方式の実施形態の一例を示す回路図である。図４は、図３の温度補償方式の実施回路定数を記した図である。前記図１のブロック図に示される温度センサ部１０１は、抵抗Ｒ１、Ｒ２、Ｒ３、Ｒ４、Ｒ５、ダイオードＤ１、演算増幅器ＩＣ１で構成され、温度変化に対するＤ１の順方向電位の変化（約−２ｍＶ／℃）を演算増幅器ＩＣ１により同相アンプして、出力端子（ａ）から温度検出信号を出力する。また補償電圧発生部１０２は抵抗Ｒ１０、Ｒ１１、演算増幅器ＩＣ２で構成され、可変容量ダイオード１４のカソードに印加する温度補償電圧を発生する。補償電圧発生部１０３は抵抗Ｒ１２、Ｒ１３、演算増幅器ＩＣ３、及び抵抗Ｒ１４、Ｒ１５、演算増幅器ＩＣ４で構成され、可変容量ダイオード１４のアノード入力の温度補償電圧を発生する。基準電圧発生部１０４は抵抗Ｒ６、Ｒ７、Ｒ８、Ｒ９で構成し、ＩＣ２、ＩＣ３、ＩＣ４の（＋）入力の基準電圧を発生する。Ｒ７、Ｒ８の接続点電位は出力端子（ａ）に発生した電源電圧Ｖｃｃのほぼ１／２とし、演算増幅器ＩＣ２の非反転入力端子（＋）に入力する。温度センサ部１０１の出力信号はＩＣ２の反転入力端子（−）に入力する。尚ほぼ常温時に、即ち振動子温度特性の変曲点温度で非反転入力端子（＋）の電位と反転入力端子（−）の電位とが同電位となるように設定する。ＩＣ２の利得はＲ１１／Ｒ１０で与えられ、利得の設定は補償電圧発生回路１０２の出力が図２の低温側頂点から高温側シフトした点まで最低飽和電圧を維持、単調増加後高温側頂点より低温側へシフトした点で最大飽和電圧になり一定電圧を維持する傾斜となるように行う。ＩＣ２の出力端子（ｂ）に発生した出力電圧と電源電圧Ｖｃｃとの電圧差をＲ１６とＲ１７で分圧して得られた。分圧出力（ｅ）は抵抗Ｒ１８を介して可変容量ダイオードＤ２のカソードへ接続される。Ｃ４はパスコンデンサであり、Ｃ４とＲ１６、Ｒ１７でローパスフィルタを構成して高周波成分（ノイズ成分）をカットする。またＲ６とＲ７の接続点の電圧は、ＩＣ３の非反転入力端子（＋）に入力する。反転入力端子（−）は出力端子（ａ）点に接続する。利得はＲ１３／Ｒ１２で与えられ、ＩＣ３の動作範囲で温度補償の最適な近似を与える傾斜に設定する。Ｒ８とＲ９の接続点の電圧は、ＩＣ４の（＋）に入力する。（−）入力は（ａ）点に接続、利得はＲ１５／Ｒ１４で与えられ、ＩＣ４の動作範囲で温度補償の最適な近似を与える傾斜に設定する。ＩＣ３の出力とＩＣ４の出力は抵抗Ｒ１９とＲ２０を経由して接続し、抵抗Ｒ２２を介してＧＮＤに接続分圧される。分圧出力（ｆ）は抵抗Ｒ２２を介して可変容量ダイオードＤ２のアノードに接続される。Ｃ５はパスコンデンサでありＲ１９、Ｒ２０、Ｒ２１でローパスフィルタを構成しており高周波成分をカットする。また、Ｄ２のカソード、アノードはそれぞれＣ１、Ｃ２を介してＸｔａｌを含む発振回路の発振ループに入力される。
【０００９】
図４に実施例回路例の１つとしてシミュレーション時の定数と、図１のブロック図との関係を示す。
補償電圧発生部１０１：Ｒ１、３、４＝１０ｋΩ、Ｒ２＝６．５Ω、Ｒ５＝可変調整、Ｄ１＝１Ｓ９５３、ＩＣ１＝ＴＣ７５Ｓ５１ＦＵ、
補償電圧発生部１０２：Ｒ１０＝１０ｋΩ、Ｒ１１＝３０ｋΩ、ＩＣ２＝ＴＣ７５Ｓ５１ＦＵ、
補償電圧発生部１０３：Ｒ１２＝１０ｋΩ、Ｒ１３＝１４０ｋΩ、ＩＣ３＝ＴＣ７５Ｓ５１ＦＵ、Ｒ１４＝１０ｋΩ、Ｒ１５＝１４０ｋΩ、ＩＣ４＝ＴＣ７５Ｓ５１ＦＵ、
基準電圧発生部１０４：Ｒ６＝９．６ｋΩ、Ｒ７＝１１．１ｋΩ、Ｒ８＝１１．３ｋΩ、Ｒ９＝８ｋΩ、
Ｒ１６、１７、１８、２１、２２＝１００ｋΩ、Ｒ１９、２０＝２００ｋΩ、Ｄ２＝ＭＡ２Ｓ３０４、Ｃ３、４、５＝０．１μＦ、Ｘｔａｌ＝１３ＭＨｚ、γ＝２４０、Ｃ０＝１．３５ｐＦ、Ｃｐ＝４０ｐＦ、Ｃｓ＝３５ｐＦ、Ｖｃｃ＝３．０Ｖ
図５は、図４の回路定数でのシミュレーション結果を示す図である。また本発明の温度補償方式の補償電圧発生部の各部の電圧変化を示す。温度センサ部１０１の出力（ａ）点の電圧変化、補償電圧発生部１０２（ｂ）点の電圧変化、補償電圧発生部１０３（ｃ）点（ｄ）点の電圧変化を示す。Δｔ［℃］の０℃は振動子の変曲点温度で基準化しており、本実施例では変局点温度はΔｔ＝２８℃である。
この図から明らかなように、（ａ）点は０．０２Ｖ／℃で単調減少、Δｔ＝０℃で１．４５Ｖを示す。この電圧はＩＣ２の（＋）入力である。（ｂ）点はΔｔ＝−２５℃より、０Ｖから０．０６Ｖ／℃で単調増加、Δｔ＝２６℃で飽和電圧３Ｖとなる。（ｃ）点はΔｔ＝−５９℃より、０Ｖから０．１４Ｖ／℃で単調増加、Δｔ＝−３６℃で飽和電圧３Ｖとなる。（ｄ）点はΔｔ＝３７℃より、０Ｖから０．１４Ｖ／℃で単調増加、Δｔ＝５９℃で飽和電圧３Ｖとなる。
【００１０】
図６は本発明の変曲点温度を基準とする温度補償シミュレーション結果を示す図である。この図から明らかなように、（ｅ）点は１．５Ｖ継続、Δｔ＝−２５℃より、０．０３Ｖ／℃で単調増加、Δｔ＝２６℃で飽和電圧３Ｖとなる。（ｆ）点はΔｔ＝−５９℃より、０Ｖから０．０７Ｖ／℃で単調増加、Δｔ＝−３６℃で飽和電圧１．５ＶとなりΔｔ＝３７℃まで継続、再度０．０７Ｖ／℃で単調増加、Δｔ＝５９℃で飽和電圧３Ｖとなる。Ｄ２のカソード、アノード間電位ＶＤ２は（ｅ）点と（ｆ）点の電位差で示され、Δｔ＝−５９℃で１．５Ｖより０．０７Ｖ／℃で単調減少、Δｔ＝−３６℃で０Ｖ、Δｔ＝−２５℃より０．０３Ｖ／℃で単調増加、Δｔ＝２６℃で１．５Ｖ、Δｔ＝３７℃より−０．０７Ｖ／℃単調減少、Δｔ＝５９℃で０Ｖとなる。
図７は本発明の変曲点温度を基準とする温度補償特性と３次近似式シミュレーション結果を示す図である。可変容量ダイオードＤ２のカソード、アノード間の温度補償電圧による容量変化、並列容量Ｃｐ及び発振回路容量を含む直列容量Ｃｓ、振動子のパラメータγ、Ｃ０より発振周波数の温度補償量をシミュレーションした特性、さらに３次式近似シミュレーション結果を示す。この図から明らかなように、かなり良い３次式近似を得ている。
図８は温度補償のシミュレーション結果を示す図である。この図は振動子の温度特性を示し、更に上記温度補償特性と温度補償による周波数偏差を示す。図より周囲温度−３０℃から＋８０℃の変化において発振周波数偏差±２ｐｐｍを得ることができる。これにより現在市場より要求されている一般的温度特性±２．５ｐｐｍ＠−２０〜７０℃を十分満足できる。
【００１１】
【発明の効果】
以上記載のごとく請求項１の発明によれば、合成した時に近似３次関数的に変化する２種類の制御電圧を発生させ、その電圧を可変容量素子の両端に印加するので、可変容量素子の容量変化によりＡＴカットによる水晶振動子の温度特性を正確に補正することができる。
また請求項２では、温度補償制御電圧を可変容量素子の両端に印加するので、ノイズの位相が同相になってノイズによる電位差がゼロとなり、発振回路の位相雑音を低減することができる。
また請求項３では、５種類の直線温度補償電圧を可変容量素子の両端に印加することで、近似３次関数に近い電位差を発生することができるので、簡易な回路構成により温度補償を実現することができ、コスト的にも安価に実現することができる。
また請求項４では、可変容量素子として可変容量ダイオードや印加電圧により容量が可変する半導体デバイスを用いることもできるので、回路構成に幅が拡がり、それに伴って回路特性のバリエーションが広くなる。
【図面の簡単な説明】
【図１】本発明の実施形態に係る温度補償方式のブロック図である。
【図２】本発明の図１の温度補償方式の動作を説明する説明図である。
【図３】本発明の温度補償方式の実施形態の一例を示す回路図である。
【図４】本発明の図３の温度補償方式の実施回路定数を表す図である。
【図５】本発明の図４の回路定数でのシミュレーション結果を示す図である。
【図６】本発明の変曲点温度を基準とする温度補償シミュレーション結果を示す図である。
【図７】本発明の変曲点温度を基準とする温度補償特性と３次近似式シミュレーション結果を示す図である。
【図８】本発明の温度補償のシミュレーション結果を示す図である。
【図９】携帯電話に使用されている水晶振動子（Ａｔ−Ｃｕｔ）の切断角度の違いによる温度特性を示す図である。
【図１０】従来の直接温度補償方式の補償回路の１例を示す図である。
【図１１】従来技術として特許第３２５３２０７号に開示されている温度補償水晶発振器のブロック図である。
【図１２】従来の温度補償方式を説明するための図である。
【図１３】従来の温度補償方式の電圧関数発生回路図である。
【図１４】従来技術として実開昭６１−９５１０４号公報に開示されている温度補償水晶発振器のブロック図である。
【図１５】従来の温度補償方式を説明するための図である。
【符号の説明】
１、２、４、５コンデンサ、１１水晶振動子、１２水晶発振回路、１４可変容量ダイオード、１６〜２２高抵抗、１０１温度センサ部、１０２、１０３補償電圧発生回路、１０４基準電圧発生部[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a piezoelectric oscillator using a piezoelectric resonator such as a crystal resonator, and more particularly to a temperature compensated piezoelectric oscillator capable of temperature compensation of an oscillation frequency of an AT cut crystal resonator with a simple circuit configuration.
[0002]
[Prior art]
The area of use of land mobile communication devices represented by mobile phones is steadily expanding. At the same time, the competition for technological development is intensifying due to the proliferation of mobile phones. Crystal oscillators used in mobile phones are also required to be smaller, lower cost, and higher performance. In particular, a system that requires coexistence with a GPS system not only has excellent temperature characteristics, but also strongly requires low noise.
FIG. 9 shows temperature characteristics depending on a difference in cutting angle of a crystal resonator (At-Cut) used in a mobile phone. As shown in the figure, the temperature characteristic of the vibrator shows a characteristic close to a cubic function, but this alone is not sufficient in the characteristic, and temperature compensation is performed so as to obtain a higher frequency stability than this characteristic.
Conventional temperature compensation methods can be broadly divided into direct temperature compensation methods and indirect temperature compensation methods. FIG. 10 is a diagram showing an example of a direct temperature compensation type compensation circuit. In this method, a thermistor, a resistor, and a capacitor are formed in the oscillation loop of the oscillation circuit, and the resistance change due to the temperature of the thermistor is converted into reactance to compensate the temperature, and the circuit configuration is very simple.
In the indirect temperature compensation method, a function generator for generating a temperature compensation voltage for obtaining temperature compensation characteristics is provided outside the oscillation circuit loop, and the temperature compensation voltage is applied to a variable capacitance diode provided in the oscillation loop to compensate the temperature. Is what you do. As a method for generating the temperature compensation voltage, a method of generating a multi-order function using the temperature characteristics of the junction potential of the semiconductor by utilizing IC technology, the compensation voltage is stored in a memory in advance, and the voltage is calculated based on the temperature change. There are a digital temperature compensation method to be applied, a dividing line approximation method using an operational amplifier, and the like. In any case, there is a problem that the correction voltage generation circuit becomes very complicated, and therefore noise generation from the circuit is large and it is difficult to reduce the noise.
[0003]
FIG. 11 is a block diagram of a temperature compensated crystal oscillator disclosed in Japanese Patent No. 3253207 as a prior art. In this method, a voltage output that changes according to temperature is input from the temperature sensor unit 101 to the voltage function generation circuits 102 to 106. Since five voltage function generation circuits are provided, the voltage adder 107 synthesizes a total of five input voltages from each voltage function generation circuit, and converts the voltage of the polygonal line function as shown in FIG. Is input. FIG. 13 shows one voltage function generation circuit composed of two operational amplifiers 110 and 111.
FIG. 14 is a block diagram of a temperature compensated crystal oscillator disclosed in Japanese Utility Model Laid-Open No. 61-95104 as a prior art. FIG. 15 shows an operation explanatory diagram. As shown in FIG. 15, the output voltage of the voltage function generation circuit 202 is increased by the input signal from the temperature sensor 201 until the saturation voltage is reached as the temperature rises to the low temperature side peak temperature of the frequency temperature characteristic exhibiting the cubic curve change of the crystal resonator. It rises monotonously and reaches saturation voltage (constant voltage) in the temperature range above the peak temperature. Similarly, the output voltage of the voltage function generating circuit 203 is constant up to the high temperature side vertex temperature of the frequency temperature characteristic of the crystal resonator, and monotonously increases with the temperature rise above that temperature. Since the respective output voltages are applied to the anodes of the diodes D14 and 15, the capacitance decreases with increasing temperature on the low temperature side, and as a result, the oscillation frequency is increased. In the temperature range where the output voltages of the voltage function generators 202 and 203 are both constant, there is no change in capacitance, so frequency change control is not performed. On the high temperature side, the capacitance increases as the temperature rises. Reduce. Further, the temperature compensating capacitors C16, 17, 18, and 19 are controlled so that the frequency monotonously increases by using a capacitor whose capacitance monotonously decreases with an increase in temperature. That is, within the temperature range where the output voltages of the voltage function generation circuits 202 and 203 are both constant, the oscillation frequency rises due to the influence of the temperature compensation capacitor. Needless to say, both the low-temperature side and the high-temperature side are affected by this capacitor, but the frequency is increased.
In this way, the third-order curve of the vibrator can be compensated comprehensively. However, in this method, compensation for the temperature between vertices is based on a temperature compensation capacitor that has large variations in both capacitance and temperature characteristics. Of course, there is a problem that variation in compensation accuracy is not suitable for mass production.
[Patent Document 1] Japanese Patent No. 3253207 [Patent Document 2] Japanese Utility Model Publication No. 61-95104
[Problems to be solved by the invention]
However, in the above direct temperature compensation system, the circuit is very simple and has the advantage of low noise. However, the compensation curve becomes a near cubic curve that increases in one direction with respect to the temperature rise without a vertex, and therefore, particularly on the low temperature side. There is a problem that it is difficult to obtain excellent temperature characteristics because the frequency compensation amount is extremely increased on both sides of the high temperature side.
Patent Document 1 adopts a complicated method in which a third-order compensation curve is obtained for the oscillation frequency temperature characteristic, an optimum linear compensation method having an intersection on the curve is obtained, and all straight lines are arbitrarily set. Therefore, the temperature characteristics of the vibrator can be handled very flexibly, and excellent temperature characteristics can be obtained.However, since a voltage function generation circuit corresponding to each linear voltage is required, the circuit configuration is very high. There are problems that it becomes complicated and inevitably increases the cost, and the noise generated from the circuit is large and the phase noise characteristic is deteriorated.
In addition, Patent Document 2 adopts a simple linear compensation method to reduce noise. However, since a device called a temperature compensation capacitor, which has a large variation in electrical characteristics between individuals, is used for mass production. There is a problem that it is not suitable.
SUMMARY OF THE INVENTION The present invention has been made in view of the above problems, and an object of the present invention is to provide a temperature compensation system suitable for a mobile phone or the like that has a simple circuit configuration and high temperature compensation accuracy, thereby enabling low noise and low cost. .
[0005]
[Means for Solving the Problems]
In order to solve the above problems, the present invention provides a piezoelectric vibrator including a piezoelectric element excited at a predetermined frequency, an oscillation amplifier that excites the piezoelectric element by passing a current, and a temperature change. A frequency oscillator circuit comprising a frequency temperature compensation circuit that compensates for a change in oscillation frequency due to a temperature detection unit, wherein the frequency temperature compensation circuit includes a temperature detection unit whose parameter changes according to an ambient temperature, and a parameter that is changed by the temperature detection unit A voltage generator for temperature compensation that generates two different voltages based on the reference voltage generator, a reference voltage generator for generating a predetermined reference voltage, and a variable whose capacitance changes based on the output voltage of the voltage generator for temperature compensation The temperature compensation voltage generator generates a voltage that is approximately half of the saturation voltage from the starting point on the low temperature side of the temperature characteristics of the piezoelectric element to the point beyond the low temperature side vertex. A first control voltage that is monotonously increased from a point beyond the low temperature side vertex to a point before the high temperature side vertex, and continuously generating a voltage that is constant by the saturation voltage from a point before the high temperature side vertex. And a monotonically increasing voltage from the low temperature side starting point of the temperature characteristics of the piezoelectric element to the point before the low temperature side apex, and the saturation voltage from the point before the low temperature side apex to the point exceeding the high temperature side apex. A second control voltage generator that maintains a voltage of approximately ½ and continuously generates a voltage that reaches the saturation voltage by monotonically increasing from the point beyond the high temperature side vertex to the high temperature side end point; And applying the output ends of the first control voltage generation unit and the second control voltage generation unit to both ends of the variable capacitance element, thereby changing the load capacitance of the piezoelectric oscillator to oscillate the piezoelectric vibrator. Compensates for temperature characteristics of frequency It is characterized in.
The temperature characteristics of the crystal unit due to the AT cut change approximately in a cubic function. This temperature characteristic is corrected by changing the load capacity in the oscillation loop so that it has ideally opposite characteristics. However, it is not impossible to realize a circuit having this characteristic, but in reality, the circuit configuration becomes very complicated, and the cost increase cannot be avoided, and the phase characteristic resulting from the complexity of the circuit. Deterioration becomes a problem. Therefore, it goes without saying that the circuit configuration is as simple as possible. However, in the conventional circuit, the characteristics at the ends of the low temperature and the high temperature are different from those of the actual circuit, and cannot be corrected accurately. Therefore, in the present invention, two types of voltages that change in an approximate cubic function when synthesized are generated, and the voltages are applied to both ends of the variable capacitance element, so that the temperature characteristics of the quartz resonator by AT cut as much as possible. It is close to.
According to this invention, two types of control voltages that change in an approximate cubic function when they are synthesized are generated and applied to both ends of the variable capacitance element. The temperature characteristics of the vibrator can be accurately corrected.
[0006]
According to a second aspect of the present invention, the noise at the output terminals of the first control voltage generator and the second control voltage generator is set to the same level and in phase, or at different levels and in phase, thereby reducing phase noise due to noise. It is characterized by doing.
If the voltages at the output terminals of the first control voltage generation unit and the second control voltage generation unit are generated from the same generation source, the phase of noise superimposed on the signal line is in phase. Further, the noise level does not change greatly and becomes substantially the same level. Therefore, when these control voltages are applied to both ends of the variable capacitance element, a potential difference due to noise does not occur, so that the relative noise level can be lowered.
According to this invention, since the temperature compensation control voltage is applied to both ends of the variable capacitance element, the phase of the noise becomes the same phase, the potential difference due to the noise becomes zero, and the phase noise of the oscillation circuit can be reduced.
According to a third aspect of the present invention, the output terminals of the first control voltage generation unit and the second control voltage generation unit are applied to both ends of the variable capacitance element, so that AT-cut crystal resonators are provided at both ends of the variable capacitance element. 5 types of linear temperature compensation voltages including two types of temperature compensation voltages parallel to the temperature change are generated with respect to the approximate cubic function of the oscillation frequency deviation.
The first control voltage generator generates three types of voltages including two types of voltages, one broken line voltage, a saturated power supply voltage, and a ground voltage, and the second control voltage generator generates two types of voltages. Three types of voltages including a broken line voltage and a half of the saturation power supply voltage are generated, and five types of linear temperature compensation voltages are generated as a whole. When these voltages are applied to both ends of the variable capacitance element, a potential difference close to an approximate cubic function is generated. As a result, the capacitance of the variable capacitance element changes to change the load capacitance of the oscillation circuit.
According to this invention, since a potential difference close to an approximate cubic function can be generated by applying five types of linear temperature compensation voltages to both ends of the variable capacitance element, temperature compensation is realized with a simple circuit configuration. Can be realized at low cost.
According to a fourth aspect of the present invention, the variable capacitance element is a varactor, a variable capacitance diode, or a semiconductor device whose capacitance is changed by an applied voltage.
If the capacitance is changed by an external applied voltage, the variable capacitance diode, junction FET gate, source or gate, drain capacitance, MOS FET gate, source or gate, drain capacitance, bipolar transistor base, The oscillator of the present invention can also be configured using an emitter capacitor, a base, or a collector capacitor.
According to such an invention, a variable capacitance diode or a semiconductor device whose capacitance can be changed by an applied voltage can be used as the variable capacitance element. Therefore, the circuit configuration is widened, and accordingly, variations in circuit characteristics are widened.
[0007]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, the present invention will be described in detail with reference to embodiments shown in the drawings. However, the components, types, combinations, shapes, relative arrangements, and the like described in this embodiment are merely illustrative examples and not intended to limit the scope of the present invention only unless otherwise specified. .
FIG. 1 is a block diagram of a temperature compensation system according to an embodiment of the present invention. In this temperature compensation system, a crystal oscillation circuit 12 including a crystal resonator 11, a variable capacitance diode 14 whose capacitance is changed by a potential difference between both ends, high resistances 16 to 22 for isolating each circuit, capacitors 1, 2 4, 5, a temperature sensor unit 101 that detects the ambient temperature, compensation voltage generation circuits 102 and 103 that generate a temperature compensation voltage based on the voltage of the temperature sensor unit 101, and a reference for the compensation voltage generation circuits 102 and 103 And a reference voltage generator 104 that supplies a voltage to be
FIG. 2 is an explanatory diagram for explaining the operation of the temperature compensation method of FIG. In the present embodiment, the temperature sensor output is set to approximately normal temperature, that is, the inflection point temperature of the transducer temperature characteristic. The compensation voltage generating circuit 102 shown in (2) in the figure is a constant voltage of about ½ Vcc from the low temperature side (point a) to the temperature (point c) exceeding the low temperature side peak temperature of the frequency temperature characteristic of the crystal resonator. The temperature monotonously increases as the temperature rises from the temperature (point c) to just before the apex on the high temperature side (point d), and reaches the saturation voltage Vcc at the temperature point d. The output voltage of the compensation voltage generation circuit 102 is divided by resistors 16 and 17 and input to the cathode of the variable capacitance diode 14 via the resistor 18. In addition, the compensation voltage generation circuit 103 shown in (3) in the figure monotonously increases from the low temperature side (point a) to the temperature temperature characteristic of the crystal resonator before the low temperature side peak temperature (point b), and reaches the temperature (point b). And reaches about 1/2 Vcc of the saturation voltage. Thereafter, the temperature continues from the temperature (b point) until it exceeds the apex of the high temperature side (point e), increases monotonically again to the high temperature side (point f), and reaches the saturation voltage Vcc at the temperature (point f). The output voltage of the compensation voltage generation circuit 103 is divided by a voltage dividing circuit including resistors 19, 20, and 21, and is input to the anode of the variable capacitance diode 14 via the resistor 22. As a result, the potential difference across the cathode and anode of the variable capacitance diode 14 shown in (4) in the figure is based on the difference between the output voltage of the compensation voltage generation circuit 102 and the output voltage of the compensation voltage generation circuit 103. It decreases monotonically from (point a) to (point b), becomes constant between (point b) and (point c), increases from (point c) to (point d), and increases from (point d) to (point e). It is constant again, and decreases monotonically from (e point) to (f point). Then, by controlling the capacitance between the terminals of the variable capacitance diode 14 based on such a control voltage, the frequency can be changed as indicated by the characteristic (5). Therefore, the frequency change and the pre-compensation shown in (1) in the figure can be changed. The frequency temperature characteristic of the vibrator 11 cancels out, and the temperature characteristic of the oscillation frequency after the compensation operation shown in (6) in the figure is obtained.
[0008]
FIG. 3 is a circuit diagram showing an example of an embodiment of the temperature compensation system of the present invention. FIG. 4 is a diagram showing implementation circuit constants of the temperature compensation method of FIG. The temperature sensor unit 101 shown in the block diagram of FIG. 1 includes resistors R1, R2, R3, R4, R5, a diode D1, and an operational amplifier IC1, and changes in the forward potential of D1 with respect to temperature changes (about −2 mV). / ° C) is in-phase amplified by the operational amplifier IC1, and a temperature detection signal is output from the output terminal (a). The compensation voltage generator 102 includes resistors R10 and R11 and an operational amplifier IC2, and generates a temperature compensation voltage to be applied to the cathode of the variable capacitance diode 14. The compensation voltage generation unit 103 includes resistors R12 and R13, an operational amplifier IC3, resistors R14 and R15, and an operational amplifier IC4, and generates a temperature compensation voltage for the anode input of the variable capacitance diode 14. The reference voltage generation unit 104 includes resistors R6, R7, R8, and R9, and generates a reference voltage for (+) inputs of IC2, IC3, and IC4. The connection point potential of R7 and R8 is approximately ½ of the power supply voltage Vcc generated at the output terminal (a) and is input to the non-inverting input terminal (+) of the operational amplifier IC2. The output signal of the temperature sensor unit 101 is input to the inverting input terminal (−) of IC2. It should be noted that the non-inverting input terminal (+) and the inverting input terminal (-) are set to have the same potential at substantially normal temperature, that is, at the inflection point temperature of the vibrator temperature characteristic. The gain of IC2 is given by R11 / R10, and the gain setting is such that the minimum saturation voltage is maintained until the output of the compensation voltage generation circuit 102 is shifted from the low temperature side peak to the high temperature side in FIG. The maximum saturation voltage is reached at the point shifted to the side, and the slope is maintained to maintain a constant voltage. It was obtained by dividing the voltage difference between the output voltage generated at the output terminal (b) of IC2 and the power supply voltage Vcc by R16 and R17. The divided voltage output (e) is connected to the cathode of the variable capacitance diode D2 via the resistor R18. C4 is a pass capacitor, and C4, R16, and R17 form a low-pass filter to cut a high frequency component (noise component). The voltage at the connection point between R6 and R7 is input to the non-inverting input terminal (+) of IC3. The inverting input terminal (−) is connected to the output terminal (a) point. The gain is given by R13 / R12 and is set to a slope that gives an optimal approximation of temperature compensation in the operating range of IC3. The voltage at the connection point between R8 and R9 is input to (+) of IC4. The (−) input is connected to point (a), the gain is given by R15 / R14, and the slope is set to give the optimum approximation of temperature compensation in the operating range of IC4. The output of IC3 and the output of IC4 are connected via resistors R19 and R20, and are divided by GND through a resistor R22. The divided voltage output (f) is connected to the anode of the variable capacitance diode D2 through the resistor R22. C5 is a pass capacitor, and R19, R20, and R21 form a low-pass filter that cuts high-frequency components. The cathode and anode of D2 are input to the oscillation loop of the oscillation circuit including Xtal via C1 and C2, respectively.
[0009]
FIG. 4 shows the relationship between the constants at the time of simulation and the block diagram of FIG. 1 as one of the circuit examples of the embodiment.
Compensation voltage generator 101: R1, 3, 4 = 10 kΩ, R2 = 6.5Ω, R5 = variable adjustment, D1 = 1S953, IC1 = TC75S51FU,
Compensation voltage generator 102: R10 = 10 kΩ, R11 = 30 kΩ, IC2 = TC75S51FU,
Compensation voltage generator 103: R12 = 10 kΩ, R13 = 140 kΩ, IC3 = TC75S51FU, R14 = 10 kΩ, R15 = 140 kΩ, IC4 = TC75S51FU,
Reference voltage generator 104: R6 = 9.6 kΩ, R7 = 11.1 kΩ, R8 = 11.3 kΩ, R9 = 8 kΩ,
R16, 17, 18, 21, 22 = 100 kΩ, R19, 20 = 200 kΩ, D2 = MA2S304, C3, 4, 5 = 0.1 μF, Xtal = 13 MHz, γ = 240, C0 = 1.35 pF, Cp = 40 pF, Cs = 35pF, Vcc = 3.0V
FIG. 5 is a diagram showing a simulation result with the circuit constants of FIG. The voltage change of each part of the compensation voltage generator of the temperature compensation system of the present invention is also shown. The voltage change at the output (a) point of the temperature sensor unit 101, the voltage change at the compensation voltage generation unit 102 (b), and the voltage change at the compensation voltage generation unit 103 (c) point (d) are shown. 0 ° C. of Δt [° C.] is normalized by the inflection point temperature of the vibrator, and in this embodiment, the inflection point temperature is Δt = 28 ° C.
As is apparent from this figure, the point (a) shows a monotonic decrease at 0.02 V / ° C. and 1.45 V at Δt = 0 ° C. This voltage is the (+) input of IC2. The point (b) monotonically increases from 0 V to 0.06 V / ° C. from Δt = −25 ° C., and reaches a saturation voltage of 3 V at Δt = 26 ° C. The point (c) monotonically increases from 0 V to 0.14 V / ° C. from Δt = −59 ° C., and reaches a saturation voltage of 3 V at Δt = −36 ° C. The point (d) monotonically increases from 0 V to 0.14 V / ° C. from Δt = 37 ° C., and reaches a saturation voltage of 3 V at Δt = 59 ° C.
[0010]
FIG. 6 is a diagram showing a temperature compensation simulation result based on the inflection point temperature of the present invention. As is apparent from this figure, the point (e) continues for 1.5 V, Δt = −25 ° C., monotonically increasing at 0.03 V / ° C., and saturation voltage 3 V at Δt = 26 ° C. The point (f) is monotonically increasing from 0 V to 0.07 V / ° C. from Δt = −59 ° C., reaches a saturation voltage of 1.5 V at Δt = −36 ° C., and continues to Δt = 37 ° C., then monotonous again at 0.07 V / ° C. The saturation voltage becomes 3 V at an increase of Δt = 59 ° C. The cathode-anode potential VD2 of D2 is indicated by the potential difference between point (e) and point (f), and monotonically decreases from 1.5V at 0.07V / ° C. at Δt = −59 ° C., and at 0V at Δt = −36 ° C. Δt = −25 ° C., monotonically increasing at 0.03 V / ° C., Δt = 26 ° C., 1.5 V, Δt = 37 ° C., monotonic decreasing at −0.07 V / ° C., Δt = 59 ° C., 0 V.
FIG. 7 is a diagram showing the temperature compensation characteristic based on the inflection point temperature of the present invention and the result of the third-order approximate expression simulation. Capacitance change due to temperature compensation voltage between cathode and anode of variable capacitance diode D2, series capacitance Cs including parallel capacitance Cp and oscillation circuit capacitance, characteristics simulating temperature compensation amount of oscillation frequency from vibrator parameters γ, C0, and A cubic approximate simulation result is shown. As is clear from this figure, a fairly good cubic approximation is obtained.
FIG. 8 is a diagram showing a simulation result of temperature compensation. This figure shows the temperature characteristics of the vibrator, and further shows the temperature compensation characteristics and the frequency deviation due to temperature compensation. From the figure, an oscillation frequency deviation of ± 2 ppm can be obtained when the ambient temperature is changed from −30 ° C. to + 80 ° C. As a result, the general temperature characteristics ± 2.5 ppm @ -20 to 70 ° C., which are currently required by the market, can be sufficiently satisfied.
[0011]
【The invention's effect】
As described above, according to the first aspect of the present invention, two types of control voltages that change in an approximate cubic function when they are combined are generated and applied to both ends of the variable capacitance element. The temperature characteristics of the crystal resonator due to the AT cut can be accurately corrected by the capacitance change.
According to the second aspect of the present invention, since the temperature compensation control voltage is applied to both ends of the variable capacitance element, the phase of the noise is in phase, the potential difference due to the noise becomes zero, and the phase noise of the oscillation circuit can be reduced.
According to the third aspect of the present invention, a potential difference close to an approximate cubic function can be generated by applying five types of linear temperature compensation voltages to both ends of the variable capacitance element, so that temperature compensation is realized with a simple circuit configuration. Can be realized at low cost.
According to the fourth aspect of the present invention, a variable capacitance diode or a semiconductor device whose capacitance can be changed by an applied voltage can be used as the variable capacitance element. Therefore, the circuit configuration is widened, and accordingly, variations in circuit characteristics are widened.
[Brief description of the drawings]
FIG. 1 is a block diagram of a temperature compensation method according to an embodiment of the present invention.
FIG. 2 is an explanatory diagram for explaining the operation of the temperature compensation method of FIG. 1 according to the present invention.
FIG. 3 is a circuit diagram showing an example of an embodiment of a temperature compensation system of the present invention.
4 is a diagram illustrating an implementation circuit constant of the temperature compensation method of FIG. 3 according to the present invention.
FIG. 5 is a diagram showing a simulation result with the circuit constants of FIG. 4 of the present invention.
FIG. 6 is a diagram showing a temperature compensation simulation result based on the inflection point temperature of the present invention.
FIG. 7 is a diagram showing a temperature compensation characteristic based on an inflection point temperature of the present invention and a third-order approximate expression simulation result.
FIG. 8 is a diagram showing a simulation result of temperature compensation according to the present invention.
FIG. 9 is a diagram illustrating temperature characteristics depending on a cutting angle of a crystal resonator (At-Cut) used in a mobile phone.
FIG. 10 is a diagram illustrating an example of a compensation circuit of a conventional direct temperature compensation method.
FIG. 11 is a block diagram of a temperature compensated crystal oscillator disclosed in Japanese Patent No. 3253207 as a prior art.
FIG. 12 is a diagram for explaining a conventional temperature compensation method;
FIG. 13 is a voltage function generation circuit diagram of a conventional temperature compensation method.
FIG. 14 is a block diagram of a temperature compensated crystal oscillator disclosed in Japanese Utility Model Laid-Open No. 61-95104 as a prior art.
FIG. 15 is a diagram for explaining a conventional temperature compensation method;
[Explanation of symbols]
1, 2, 4, 5 Capacitor, 11 Crystal resonator, 12 Crystal oscillation circuit, 14 Variable capacitance diode, 16-22 High resistance, 101 Temperature sensor unit, 102, 103 Compensation voltage generation circuit, 104 Reference voltage generation unit

Claims

A piezoelectric vibrator having a piezoelectric element excited at a predetermined frequency, an oscillation amplifier that excites the piezoelectric element by passing a current, and a frequency temperature compensation circuit that compensates for a change in oscillation frequency due to a temperature change. A piezoelectric oscillator,
The frequency temperature compensation circuit includes: a temperature detection unit whose parameter changes according to an ambient temperature; a temperature compensation voltage generation unit that generates two different voltages based on the parameter changed by the temperature detection unit; and a predetermined reference voltage A reference voltage generator that generates a variable capacitance element whose capacitance changes based on the output voltage of the temperature compensation voltage generator,
The temperature-compensating voltage generator holds a voltage approximately half of the saturation voltage from the starting point on the low-temperature side of the temperature characteristics of the piezoelectric element to the point beyond the low-temperature side vertex, and exceeds the low-temperature side vertex. A temperature that the piezoelectric element has, and a first control voltage generator that monotonously increases from the point to the point on the high temperature side and continuously generates a constant voltage from the point on the high temperature side by the saturation voltage. From the starting point on the low temperature side of the characteristic, monotonously increases from the lowest voltage to the point just before the low temperature side vertex, and holds about half the saturation voltage from the point before the low temperature side vertex to the point beyond the high temperature side vertex. A second control voltage generation unit that continuously generates a voltage that monotonously increases from the point beyond the high temperature side vertex to the end point on the high temperature side to reach the saturation voltage;
By applying the output ends of the first control voltage generation unit and the second control voltage generation unit to both ends of the variable capacitance element, the load capacitance of the piezoelectric oscillator is changed to change the oscillation frequency of the piezoelectric vibrator. A temperature compensated piezoelectric oscillator characterized by compensating temperature characteristics.

The noise at the output terminals of the first control voltage generation unit and the second control voltage generation unit is reduced to the same level and in phase, or at different levels and in phase, thereby reducing phase noise due to noise. The temperature compensated piezoelectric oscillator according to claim 1.

By applying the output terminals of the first control voltage generation unit and the second control voltage generation unit to both ends of the variable capacitance element, the oscillation frequency deviation of the AT-cut crystal resonator is applied to both ends of the variable capacitance element. 3. The temperature compensated piezoelectric oscillator according to claim 1, wherein five types of linear temperature compensation voltages including two kinds of temperature compensation voltages parallel to the temperature change are generated for the approximate cubic function. .

4. The temperature compensated piezoelectric oscillator according to claim 1, wherein the variable capacitance element is a varactor, a variable capacitance diode, or a semiconductor device whose capacitance is changed by an applied voltage. 5.