JP2004166185A - High frequency component, high frequency module and communication device using the same - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide high frequency components, etc. of which the insertion loss in a necessary fundamental frequency band is minimized and harmonic attenuation quantity in an unnecessary frequency band is maximized. <P>SOLUTION: In the high frequency component, it has a branching circuit which branches a signal to a plurality of transmission/reception systems with different pass bands, a switch circuit which switches a transmission system and a reception system to each of the transmission/reception systems, a switch circuit having a low-pass filter and a high frequency amplifier circuit having a semiconductor element, a power supply circuit and a matching circuit in the respective transmission systems of the switch circuit, a phase adjustment circuit is provided between an output matching circuit of the high frequency amplifier circuit and a transmission terminal of the switching circuit, matching between the switch circuit and the high frequency amplifier circuit is in relation of phases equivalent to conjugative matching in the necessary fundamental frequency band, however, the phase is adjusted as relation equivalent to non-conjugative matching in unnecessary two or three times of frequency bands. <P>COPYRIGHT: (C)2004,JPO

Description

【００３３】
また実施態様によっては、EGSM系をさらにGSM850(送信周波数：824〜849MHz、受信周波数：869〜894MHz)とEGSMに分けて、クワッドバンド対応とすることもできる。この場合、送信系は共通端子を用いることができ、受信系は前記トリプルバンド対応アンテナスイッチのEGSM受信端子部にGSM850とEGSMを切り替えるスイッチを接続することにより構成できる。また、前記スイッチの代わりにGSM850、EGSM帯のλ/4共振器である伝送線路を用いて、両者間の周波数を分けることでも実現できる。
【００３４】
図７は本実施例の高周波部品を構成する高周波増幅器側の回路図を示す。高周波増幅器側の整合回路端の出力端子Ｐ0を図６のアンテナスイッチモジュールの例えばEGSM Txの送信端子Ｐ1に接続し、増幅した送信信号をアンテナスイッチ側に送る役割を果たす。出力端子P0には、直流カットコンデンサCa2を介して、伝送線路ASL1の一端が接続される。伝送線路ASL1には一端を接地されたコンデンサCa3、Ca4が接続されて出力整合回路を構成する。伝送線路ASL1の他端は、半導体素子の一種である電界効果スイッチングトランジスタ（ＦＥＴ）Ｑ１のドレインに接続される。また、ＦＥＴＱ１のソースは接地され、ゲートはバイポーラスイッチング素子（Ｂ−Ｔｒ）Ｑ２のコレクタに接続される。
【００３５】
他方、伝送線路ASL1の他端と電界効果スイッチングトランジスタFET Q1のドレインDとの接続点は、λ／４ストリップライン等からなるインダクタSL1とコンデンサCa5との直列回路を介して接地され、インダクタSL1とコンデンサCa5との接続点はドレイン電圧端子Vdd1に接続されている。また、電界効果スイッチングトランジスタFET Q1のゲートとバイポーラスイッチング素子Q2のコレクタとの接続点は、コンデンサCa6を介して接地されると共にゲート電圧端子Vgにも接続される。
【００３６】
更に、バイポーラスイッチング素子Q2のエミッタは接地され、ベースは伝送線路SL3の一端に接続される。バイポーラスイッチング素子Q2のコレクタは、ストリップライン等からなるインダクタSL2とコンデンサCa7との直列回路を介して接地され、インダクタSL2とコンデンサCa7との接続点は、コレクタ電圧端子Vcに接続される。また、インダクタSL2とコンデンサCa7との接続点は、バイポーラスイッチング素子Q2のベースと伝送線路SL3の一端との接続点にも接続される。伝送線路SL3の他端は、コンデンサCa8を介して接地されると共に入力端子Pinに接続される。
尚、図６及び図７の等価回路において伝送線路及びインダクタはストリップラインで構成されることが多いものの、マイクロストリップライン、コプレーナガイドライン等で構成されていてもよい。また、増幅器回路側は、半導体素子Q3と電源供給回路を同様に付加して増幅回路を3段、またそれ以上の多段となしハイパワーアンプとして構成することも出来る。尚、トランジスタはQ1をFET、Q2をB-Trとしたが、それぞれ他の種類のトランジスタでも良い。例えば、Si-MOSFET、GaAsFET、Siバイポーラトランジスタ、GaAsHBT（ヘテロ接合バイポーラトランジスタ）、ＨＥＭＴ(高電子移動度トランジスタ)等があげられる。もちろん、いくつものトランジスタを集積化したMMIC(モノリシックマイクロ波集積回路)を用いても良い。また、本実施例では伝送線路SL3とトランジスタQ2の間を直接繋いでいるが、抵抗を介して接続しても良い。
【００３７】
図５に位相調整回路の周辺を図示している。位相調整回路は伝送線路またはハイパスフィルタやローパスフィルタ等のＬＣ回路を挿入することで実施でき、その設定指針と調整方法については上述した通りである。ここではEGSM Tx端子Ｐ1−Ｐ0間に図４▲２▼のハイパスフィルタを挿入した実施態様について説明を加える。図８は高周波増幅器側を固定しアンテナスイッチモジュール側を調整したときの基本波（f）、2倍高調波（2f）、3倍高調波（3f）の動向をスミスチャート上にプロットしたものである。パワーアンプ側を見た位相は2倍高調波（2f）、3倍高調波（3f）共にθの領域（左上）にある。出力整合回路の伝送線路ASL1の長さとコンデンサ付加を調整して共役整合の関係となし、アンテナスイッチモジュール側の基本波（f）はほぼ中心の略50Ωで共役整合関係に調整されている（略中央の点線丸内の■印）。2倍高調波（2f）、3倍高調波（3f）の現在位置は共役整合のθ1の領域（左下）にあるが、これを例えば18nHのインダクタと6pFのコンデンサで構成したハイパスフィルタを挿入したことにより上述した位相θ2の領域（右下）に移動しており、本例では２倍波（2f）の方が目標とするθ0の領域（右上）により近いθ0−90度の領域に、３倍波（3f）はθ0−120度の領域に調整することが出来ている。尚、調整する位相や方向は場合によって種々異なるが、不要高調波の主要成分である２倍波（2f）を優先させてθ0に近づけることを基本とし、場合によっては３倍波（3f）は犠牲にしてθ2の許容範外でも良いと見なせる。
【００３８】
EGSM（880〜960MHz）での2倍波（2f）、3倍波（3f）の位相と減衰量の関係は図９に示すようであった。即ち図９（Ａ）は2倍波の位相―減衰量特性を、図９（Ｂ）は3倍波の位相―減衰量特性を示し、縦軸に減衰量（dB）を、横軸にアンテナスイッチモジュール側の入力位相（deg）を表している。（Ａ）によれば最大減衰量を示す位相θ0は-73(dB)の30度付近にある。この位相まで調整できれば申し分無いが、実際の設計上は許容範囲を設けることが相当である。例えば3dB程度の減衰量の劣化を許容できる範囲は180度（θ0±90度）あり、5dB程度の劣化を許容できる範囲は240度（θ0±120度）であることが分かる。これは（Ｂ）の3倍波の減衰量特性においても同様の結果であった。実際3dB劣化でも-70（dB）を、5dB劣化でも-65（dB）以上を確保しているのでベストのθ0位相に対し±120度以内に位相調整すれば十分効果があることが分かった。そして、好ましくは±90度、更に好ましくは±45度程度であると言える。また、DCS／PCS系についても同様の範囲で効果が得られることが分かった。
【００３９】
次に、図１０は図６の等価回路で示されるアンテナスイッチモジュールと図７の等価回路で示される高周波増幅器を一つの積層体内に収めた積層モジュールのうち上部2層と下部2層の誘電体グリーンシートを抜き出した展開図である。グリーンシートは▲１▼が最上層で以下順に15層で構成され、最後のシート(15)は積層体の裏面を示している。本実施例で使用したグリーンシートは９５０℃以下の低温同時焼成が可能なＬＴＣＣ材料からなる。例えば、Ａｌ_２Ｏ_３換算で１０〜６０質量％、ＳｉＯ_２換算で２５〜６０質量％、ＳｒＯ換算で７．５〜５０質量％、ＴｉＯ_２換算で２０質量％以下のＡｌ，Ｓｉ，Ｓｒ，Ｔｉと、Ｂｉ_２Ｏ_３換算で０．１〜１０質量％、Ｎａ_２Ｏ換算で０．１〜５質量％、Ｋ_２Ｏ換算で０．１〜５質量％、ＣｕＯ換算で０．０１〜５質量％、ＭｎＯ_２換算で０．０１〜５質量％のＢｉ、Ｎａ、Ｋ、Ｃｕ、Ｍｎをそれぞれ含有した誘電体組成物が用いられる。
グリーンシートは伝送線路や容量を形成しやすいようにシート厚みは４０〜２００μｍのものを使用した。電極材は銀系のものを用いた。このセラミックグリーンシートの各層に伝送線路やコンデンサ容量を電極パターンにより形成し、適宜スルーホールを設けて回路を構成した。このセラミックグリーンシートを順次積層圧着し、９５０℃で焼成することにより高周波部品が複合化された積層体モジュールが得られる。積層体の大きさは横13.75mm×縦8mm×高さ0.75mm程度であり、積層体の上面にはダイオードやトランジスタ及びチップインダクタ、チップコンデンサを搭載し、その上に金属ケースを被せて完成品とする。完成後の全高は1.8mm程度である。ただし、金属ケースの代わりに、樹脂封止パッケージとしても良く、この場合の全高は1.5mm程度である。また、他の積層体の大きさとしては横10mm×縦8mm×高さ0.75mmのものも実施した。この場合も完成品としての全高は、金属ケースの場合で1.8mm程度、樹脂封止パッケージの場合で1.5mm程度である。
【００４０】
積層体内の概略構成は、アンテナスイッチモジュール側は、上部層に分波器回路及びローパスフィルタを構成する伝送線路を、中部層に分波器回路、スイッチ回路及びローパスフィルタを構成するコンデンサ容量を、下部層にスイッチ回路を構成する伝送線路が主に形成されている。一方高周波増幅器側は、上部層に初段整合回路の主に伝送線路を、中部層に初段、後段整合回路の主にコンデンサ容量を、下部層にサーマルビアや後段整合回路の主に伝送線路、電源供給用ラインが主に形成されている。高周波増幅器とアンテナスイッチモジュールの接続は上層にあり、相互干渉を避けるためにグリーンシート▲１▼の伝送線路ASL1とグリーンシート▲２▼の伝送線路ASL2とは層を変えて且つ上下に重ならないような位置に形成している。両者の間に介在させるハイパスフィルタは、ＬＣ回路をチップインダクタとチップコンデンサで構成し積層体の上面に搭載する。これにより積層体モジュールを作成した後でも調整が出来るので、試作調整に時間がかからず望ましい。
【００４１】
この積層体モジュールは、高周波増幅器HPAを構成する電極パターンは左側領域に、他方アンテナスイッチモジュールASMを構成する電極パターンは右側領域に形成し、グリーンシートは積層方向全てに渡って２つの領域に区分して構成している。これにより機能毎の回路をまとめて設けたので特性劣化を招くことなく小型化ができる。さらに左右の領域の間に帯状のグランド電極Ｇ0を設け、以上により各電極パターンの配置設計が比較的簡易にできるし、高周波部品間の相互干渉が抑制できる。寸法配置的に余裕がある場合は全部のグリーンシートにこのようなグランド電極Ｇ0を設けることが望ましいが、小型化により多くの場合それが出来ないのでグリーンシート▲２▼に示すようにグランド電極G0に繋がるスルーホール電極HGを間欠的に設けることでグランド電極G0の作用をなすことができる。ここでスルーホール電極HGは簾状であるので間隔を考慮する必要があり、スルーホール電極HGの間隔ｇを干渉を防ぎたい最も高い周波数の波長(λ)の１／１０以下とすることが望ましい。好ましくはλ／10〜λ／50程度で遮蔽効果が高まることが分かった。この実施例では間隔gは不等間隔であるが、おおよそDCS帯の3倍波のλ/20（略1mm）〜λ/25程度とした。
以上により、グランド電極G0及びスルーホール電極HGによるグランドによって両者間の干渉が無くなり、高周波増幅器の発振等の不安定動作を防止できる。また必要信号(送信信号)と不要信号とのスプリアス発生を抑えることができ、通過特性の悪化を防止できるものである。さらに、高周波部品を一つの積層体内に集約したので携帯電話内のプリント配線基板上の実装面積は、従来のパワーアンプとアンテナスイッチを別々に基板に実装した場合に比べて25%〜50%の小型化が出来た。よって、携帯電話やPDA等の小型情報端末などの通信機に搭載することで小型軽量化のニーズに答えることが出来る。
【００４２】
また、実施例のように高周波増幅器HPAとアンテナスイッチモジュールASMとを積層体内に一体化することによって、夫々の回路間を接続する線路の長さが短縮され、積層体内でも線路損失を低減でき、マッチングずれも発生しない。また、プリント配線基板上に従来両者を接続するために必要だった配線が必要でなくなり、直接高周波モジュールをプリント配線基板上に実装するため線路損失が低減でき、マッチングずれも発生しない。また、高周波増幅器HPAとアンテナスイッチモジュールASMとを一体設計することができるため、相互のインピーダンス整合を最適化した設定が行え、従来のようにHPAとASMの各々に設けられていた整合回路を簡略化できる。このような最適設計により、小型かつ低挿入損失、高出力電力、高出力効率である高周波モジュールが完成した。
特性について言えば、従来はGSM送信時において、効率35%、2倍高調波-25dBm、3倍高調波-25dBm 程度であったが、本実施例によれば効率43%以上、2倍高調波-38dBm以下、3倍高調波-36dBm以下と特性向上が図られた。このとき、DCS/PCS送信時の特性は効率34%以上、2倍高調波-38dBm以下、3倍高調波-36dBm以下を達成できた。これにより、GSM帯域のみならず、DCS/PCS帯域においても本発明の有効性を確認できた。
以上の特性向上により、この高周波モジュールを携帯電話に用いた場合、従来のように別々に部品を実装した場合と比べて、5〜10%程度の高効率化が可能となった。これにより、送信時の消費電力が少なくなり、バッテリーの持ちが良くなり、一充電あたりの通話時間として10〜20%程度の長期通話が可能となった。
また、上記のようにこれまでは携帯端末メーカが所望の特性を得るために複数種の高周波増幅器とアンテナスイッチモジュールを組合せて評価をしたり、また、それらを使いこなすために、プリント基板上に整合回路や高調波フィルタ等を形成するといった様々な工夫を行っていたが、これらを全て省略できる。
【００４３】
また、図６に示したアンテナスイッチの等価回路、図７に示した高周波増幅器の等価回路は一例である。例えば、スイッチ回路はpinダイオードを用いた例を示したが、SPDT（Single Pole Dual Throw）、SP3T等のSPnT型のGaAsスイッチを用いてスイッチ回路を構成することも出来る。この場合、単純にpinダイオードスイッチをSPDTのGaAsスイッチに置き換えた場合、pinダイオードスイッチで必要なλ/4ラインが不要となるため、積層体内にその分の余裕ができる。このため、このスペースを削減したり、また新たに機能素子を形成するなどして更なる小型化、高集積化に有利である。また、分波器まで含めた全体を置き換えることもできる。この場合、トリプルバンドアンテナスイッチを例にとると、SP5TのGaAsスイッチで置き換えれば全経路のスイッチングが可能となる。その上、上記したようにλ/4ラインが数本不要となり更なる小型化、高集積化に有利である。ただし、送信側のローパスフィルタや特性を満足させるために挿入する各種フィルタ類は積層体および／または搭載部品で構成することになる。また、この場合、アンテナと直接接続することになるため、GaAsスイッチは静電気サージ対策が施されていることが望ましい。スイッチ自体にサージ未対策のものを使用する場合は、アンテナとGaAsスイッチの間にLC回路等からなるフィルタなどのサージ対策回路を挿入する必要がある。
【００４４】
以下にGaAsスイッチを用いてトリプルバンドのアンテナスイッチ回路を構成した実施例を図１２〜図１４を用いて説明する。高周波増幅器HPAとの一体化は図1７のブロック図の左ブロックを図１２〜図１４のブロック図で置き換えることにより実施できる。図１２ではSPDT(SP2T)スイッチSW1とSP3TスイッチSW2を用いている。まず、ダイプレクサDipにより、例えば低周波側のEGSM帯と高周波側のDCS/PCS帯を分波し、スイッチSW1でEGSM帯の送信（PA1に接続）と受信（RX1）を切り替え、スイッチSW2でDCS/PCS帯のDCS/PCS送信（PA2に接続）、DCS受信（RX2）またはPCS受信（RX3）の３経路を切り替えるようにしている。図１３はSP2Tスイッチのみを3個使用した場合の例である。この場合も上記と同様に、ダイプレクサDipにより通信システムのうち低周波側であるEGSM帯と高周波側であるDCS/PCS帯を分波し、スイッチSW1でEGSM帯の送信（PA1に接続）と受信（RX1）を切り替え、スイッチSW2でDCS/PCS帯のDCS/PCS送信（PA2に接続）、DCS/PCS受信を切り替え、更に、スイッチSW3でDCS受信（RX2）とPCS受信（RX3）を切り替えている。図１４はSP5Tスイッチを使用した場合の例である。この場合、ダイプレクサDipは使用せず、全ての経路をスイッチのみで切り替えるようにする。尚、図１２、図１３において、GaAsSPDTスイッチの回路ブロックを適宜、PINダイオードを使用したSPDTスイッチに置き換えた構成とすることも可能である。このようにGaAsスイッチを用いてアンテナスイッチモジュールを構成したとしても、上記したPINダイオードを用いた実施例と同様に、高周波増幅器HPAとアンテナスイッチモジュールASMとの間の整合(位相)関係を上述した位相調整回路により、通過帯域では共役整合関係となし、不要な高調波帯域では非共役整合関係へと位相調整することによって高周波モジュール全体の特性を向上することができる。更に、上記の出力検出器の検出値に基づいて出力トランジスタのゲート電圧を制御するAPC回路をも含んだ一体化モジュールとして構成することもできる。
【００４５】
本発明の高周波モジュールは、アンテナスイッチモジュールと高周波増幅器の間にカプラ回路やアイソレータ回路を備えても良く、受信系経路にはSAWフィルタを挿入しても良い。上記実施例では高周波増幅器の半導体チップに出力電力をモニタする機能を複合した場合で説明してきた。一方でカプラ回路により電力を検出する手法を用いた構成もありうる。この場合の実施例を図１５及び図１６を用いて説明する。図１５に示すようにカプラCPLはアンテナスイッチモジュールのローパスフィルタLPFと高周波増幅器PAとの間に挿入して使用される。図１６に示す高周波増幅器HPAからの高周波出力の伝送経路CSL1を主線路、出力電力の一部を取り出すための伝送線路CSL2を結合線路と一般に呼ぶ。また、結合線路の一端は出力モニタとして検波器へ送られ、高周波増幅器HPAの出力電力制御に用いられる。一方、他端は抵抗Rにて終端される。一般に50Ωで終端されるが、結合度、アイソレーションの調整をするため、適宜、変更することも可能である。このように構成した場合、高周波増幅器HPAとアンテナスイッチモジュールASMとの間にカプラが挿入されるため、この場合は高周波増幅器とアンテナスイッチモジュールとの相対位相の調整をするのではなく、高周波増幅器HPAとカプラCPLとの相対位相を上述した実施例のように位相調整回路を設け、通過帯域では共役整合関係となし、不要高調波帯域では非共役整合関係へと調整することを行う。この場合、カプラCPLとアンテナスイッチモジュールASMは所望特性が得られるように一体設計することが望ましい。更に望ましい改良手段としては、カプラCPLとアンテナスイッチモジュールASMとの相対位相も上述のように、通過帯域では共役整合関係となし、不要高調波帯域では非共役整合関係へ調整すると言う本発明を実施することで全体特性は格段に向上する。
【００４６】
通常、カプラCPLを用いた場合は、カプラ回路による損失が0.2〜0.3dB程度発生するのが一般的である。しかしながら、高周波増幅器HPAとカプラ回路CPL及びアンテナスイッチモジュールASMを一つの積層体内に最適化設計することにより、従来の各回路を別々にプリント配線基板上に実装する場合と比較すると、25〜50％の小型化が達成でき、かつ、2〜7%程度の高効率化が達成できる。これにより、送信時の消費電力が少なくなり、バッテリーの持ちが良くなる。例えば一充電あたりの通話時間としては5〜15%程度の長期通話が可能となる。更に、上記カプラの検出値に基づいて出力トランジスタのゲート電圧を制御するAPC回路をも含んだ一体化モジュールとして構成することもできる。
【００４７】
尚、本発明で用いられる送受信系システムとしては、上記した以外にもPDC800帯域(810〜960MHz)、GPS帯域(1575.42MHz)、PHS帯域(1895〜1920MHz)、Bluetooth帯域(2400〜2484MHz)や、米国で普及が見込まれるCDMA2000、中国で普及が見込まれるTD-SCDMAなどを組み合わせたマルチバンド対応としたアンテナスイッチ回路の場合も同様の効果が期待できる。これらの場合の回路を用いてデュアルバンド、３バンド、４バンド、５バンド等のマルチモードマルチバンドのアンテナスイッチ回路が得られる。
【００４８】
【発明の効果】
本発明によれば、必要な基本周波数帯域での挿入損失を最小に抑えると共に、不用周波数帯域での高調波減衰量を最大にした、損失が無く電力付加効率の高いアンテナ出力特性の優れたマルチバンド用高周波複合部品が得られた。さらに一つの積層体内に２つ或いは２つ以上の機能を一体化したマルチバンド用の高周波モジュールとなり安価で一層の軽量小型が達成できた。そしてこれらを用いた高性能な通信機を提供することができた。
【図面の簡単な説明】
【図１】本発明のマルチバンド用高周波複合部品の位相調整を説明するスミスチャート上の説明図である。
【図２】同じく位相調整の手順を説明する簡略的なスミスチャートの説明図である。
【図３】位相調整の具体的手段の一つであるローパスフィルタを示す回路図である。
【図４】位相調整の具体的手段の一つであるハイパスフィルタを示す回路図である。
【図５】位相調整の具体的手段の一つである伝送線路の場合を示す高周波増幅器の出力整合回路端子とアンテナスイッチモジュール側の送信端子付近の回路図である。
【図６】本発明の一実施例のトリプルバンド用アンテナスイッチモジュールの等価回路図である。
【図７】本発明の一実施例の高周波増幅器の等価回路図である。
【図８】本発明の一実施例であってハイパスフィルタを用いた場合の位相調整の状況を示すスミスチャート図である。
【図９】本発明による位相調整と減衰量の関係を示すもので（Ａ）は位相−２倍波減衰量、（Ｂ）は位相−３倍波減衰量の特性線図である。
【図１０】本発明のマルチバンド用複合積層モジュールの一実施例を示す、積層体のグリーンシート展開図である。
【図１１】本発明の位相調整を検討したモデルのブロック図である。
【図１２】GaAsスイッチを用いてトリプルバンド用アンテナスイッチモジュールを構成した一実施例のブロック図である。
【図１３】同じくGaAsスイッチを用いた実施例のブロック図である。
【図１４】更に他のGaAsスイッチを用いた実施例のブロック図である。
【図１５】アンテナスイッチモジュールと高周波増幅器の間にカプラ回路を挿入して構成した場合の実施例を示すブロック図である。
【図１６】カプラ回路の一実施例を示す図である。
【図１７】本発明のマルチバンド用高周波部品、高周波モジュールの形態を説明するブロック図である。
【符号の説明】
ASM：アンテナスイッチモジュール
HPA：ハイパワーアンプ
Dip：ダイプレクサ（分波器、分波回路）
SW：スイッチ回路
LPF：ローパスフィルタ回路
SAW：弾性表面波フィルタ
L、SL、ASL：インダクタ、伝送線路
C、Ca：コンデンサ
Ｑ１、Ｑ２：半導体スイッチング素子
G0：グランド電極
HG：スルーホールによるグランド電極[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a radio communication system for transmitting and receiving signals of two or more different frequencies using one antenna in common, a multi-band high-frequency component in which a switch circuit and a high-frequency amplifier circuit are combined, and configuring these components in one laminated body And a communication device using the same.
[0002]
[Prior art]
For portable wireless systems, for example, EGSM (Extended Global System for Mobile Communications) and DCS (Digital Cellular System) systems, which are popular in Europe, PCS (Personal Communication Service) systems, which are popular in the United States, are adopted in Japan. There are various systems using time division multiple access (TDMA) such as a PDC (Personal Digital Cellular) system. With the rapid spread of mobile phones in recent years, especially in major metropolitan areas of developed countries, the frequency band allocated to each system cannot cover system users, making it difficult to connect or connecting during a call There is a problem such as disconnection. Therefore, it has been proposed that a user can use a plurality of systems to substantially increase usable frequencies, further expand a service area, and effectively use a communication infrastructure of each system.
2. Description of the Related Art Conventionally, as a small and lightweight high-frequency circuit component corresponding to a plurality of systems, for example, a dual-band high-frequency switch module used in a portable communication device corresponding to two systems of EGSM and DCS is disclosed in Patent Document 1. Patent Document 2 proposes a triple-band high-frequency switch module used in a portable communication device compatible with three systems of EGSM, DCS, and PCS.
[0003]
FIG. 17 shows an example of a block configuration of a triple band high frequency switch module. A signal in the EGSM frequency band and a signal in the DCS / PCS frequency band by a demultiplexing circuit Dip (hereinafter sometimes referred to as a demultiplexer or diplexer, which is equivalent) connected to the common antenna ANT terminal (Combined in the reverse direction, but will be described in the present specification as demultiplexing), the first high-frequency switch SW1 switches between the EGSM transmission terminal Tx and the EGSM reception terminal Rx, and the second high-frequency switch SW2 Switches between the DCS / PCS transmission terminal Tx, the DCS reception terminal Rx, and the PCS reception terminal Rx. The low-pass filters LPF1 and LPF2 are inserted in the transmission path to reduce the amount of harmonic distortion generated by the high power amplifier. The band-pass filters SAW1, SAW2, and SAW3 remove unnecessary frequency components from the received signal from the antenna ANT and send only the necessary components to the low-noise amplifier. Therefore, high-power amplifiers HPA1 and HPA2 are provided before the EGSM transmission terminal Tx and the DCS / PCS transmission terminal Tx, and low-noise amplifiers LNA1 and LNA2 are provided after the EGSM reception terminal Rx, the DCS reception terminal Rx, and the PCS reception terminal Rx. , LNA3 are provided.
[0004]
There is still a strong demand for smaller and lighter portable communication devices, and modularization that integrates components and integrates functions is being promoted. For example, a circuit component surrounded by a dotted line in FIG. 17 is formed by forming a transmission line and a capacitor by an electrode pattern in a multilayer body in which dielectric sheets such as LTCC (Low Temperature Co-fired Ceramics) are stacked in multiple layers, and a diode or the like. This is realized as a multi-band antenna switch module ASM mounted on a laminate (see the above-mentioned patent publication). Also, modularization of the area enclosed by the dashed line is realized by mounting a discrete SAW filter on the laminate.
[0005]
On the other hand, on the transmitting side of the portable communication device, a high-power amplifier of about several W is used to output a signal of relatively high power (in the present invention, a high-frequency amplifier is not distinguished from a power amplifier or the like. Is sometimes used.) Is used. Since mobile phones and the like need to be small and have low power consumption, a high-power amplifier that consumes most of DC power has high DC-RF power conversion efficiency (also referred to as power added efficiency) and is small. Is required. Especially for mobile phones, etc., the small size of the device and the length of talk time per charge of the battery are important selling points of the product, so the miniaturization and high efficiency of high power amplifiers Is required. Although modularization of circuit components, including high-power amplifiers, is an issue for study, it has not been realized to actually form a laminated module using LTCC or the like.
[0006]
As a related art related to the amplifier and the antenna or the antenna switch module, Patent Document 3 discloses an antenna device in which a reception-only antenna and an amplifier are mounted on a stacked body and a phase adjustment circuit is provided between the two. However, this is for adjusting the phase shift of the closed loop when the reception-only antenna (patch antenna) itself receives the electromagnetic wave leaked from the amplifier. That is, it is not a composite of the high-frequency switch function.
Patent Document 4 discloses that a multi-layer substrate formed by laminating a plurality of dielectric layers incorporates a transmission line and a capacitor constituting a high-frequency switch and an amplifier, and a transistor and the like are mounted on the multi-layer substrate to form a module. Have been. However, in this case, there is no disclosure of any practical problem or solution when the two are integrated only by showing a concept, and it is difficult to realize.
Further, Patent Document 5 discloses a high-frequency transmission module in which a high-power amplifier and a coupler for monitoring the output power are integrated, and matching between the two is set to non-conjugate matching at a spurious frequency. However, this high-frequency transmission module is intended for modularization of a high-power amplifier and a coupler, and specific solutions for reducing insertion loss between high-frequency components and deteriorating harmonic attenuation characteristics are presented. Not.
[0007]
[Patent Document 1]
JP-A-11-225088
[Patent Document 2]
JP 2000-165288 A
[Patent Document 3]
JP 2000-183612 A
[Patent Document 4]
JP-A-10-126307
[Patent Document 5]
JP 2002-171137 A
[0008]
[Problems to be solved by the invention]
As described above, heretofore, there is no example in which a multi-band antenna switch circuit and a high-frequency amplifier circuit are realized as a composite module in one laminate. Therefore, assembling manufacturers of portable communication equipment, various combinations of existing high-frequency amplifier circuits and existing antenna switch modules, and monitoring of transmission characteristics such as conversion efficiency and harmonic generation amount or antenna output characteristics, have resulted in The fact is that the design method for selecting combinations is limited.
Therefore, it is conceivable to combine the high-frequency amplifier HPA and the antenna switch module ASM into a composite module in one laminated body (the range of ASM + HPA in FIG. 17). However, even if the size is reduced by using a composite module, the same problem as in the case of the discrete combination cannot be solved. For example, the output terminal of the high-frequency amplifier and the transmission terminal of the antenna switch module are each designed for approximately 50Ω matching, but each is not exactly 50Ω, and is set to around 50Ω as much as possible. Is common. For example, even if the high-frequency amplifier HPA side is at 45Ω and at a phase of 120 degrees, and the antenna switch module ASM side is at 52Ω and at a phase of 80 degrees, it is considered that they match each other by 50Ω. In the end, we aim at the vicinity of 50 Ω in the center of the Smith chart in the transmission band, but do not consider the impedance relationship including the reactance of each other. Therefore, without specific design guidelines, the effectiveness cannot be improved, and as a result, there is a problem that the loss in the required frequency band increases and the attenuation in the unnecessary frequency band is insufficient. . This problem also occurs when a high-frequency amplifier circuit and an antenna switch module are individually mounted on a circuit board and a matching circuit is provided between the two.
[0009]
The present invention has been made in view of such a problem, and has a specific solution capable of minimizing insertion loss in a necessary fundamental frequency band and maximizing the amount of harmonic attenuation in an unnecessary n-fold frequency band. To provide. First, a high-frequency component that combines a switch circuit and a high-frequency amplifier circuit and provides a phase adjustment circuit between the two is provided. Finally, a communication device using these is provided.
[0010]
[Means for Solving the Problems]
The present invention relates to a high-frequency component for handling a plurality of transmission / reception systems having different pass bands, a demultiplexing circuit for demultiplexing a high-frequency signal and a low-frequency signal, and a transmission system connected to the demultiplexing circuit. Or a switch circuit for switching connection with a reception system, a high-frequency amplifier circuit having at least a semiconductor element, a power supply circuit, and a matching circuit, and a phase adjustment circuit between the switch circuit and the high-frequency amplifier circuit, wherein the switch The matching between the circuit and the high-frequency amplifier circuit via the phase adjustment circuit has a phase relationship corresponding to conjugate matching in a necessary fundamental frequency band, but an unnecessary frequency of n times (n is a natural number of 2 or more) is required. In the band, it is a high-frequency component having a phase relationship corresponding to non-conjugate matching.
The high-frequency component has a relationship between an impedance Z1 when viewing the high-frequency amplifier side from a connection reference plane between the switch circuit and the high-frequency amplifier circuit, and an impedance Z2 when viewing the switch circuit side from the connection reference plane. However, when a phase having a conjugate matching relationship with the phase θ of the Z1 is defined as θ1, it is desirable to adjust the phase θ2 of the Z2 to a non-conjugate matching region of 180 ° ± 120 ° with respect to θ1. More preferably, the phase is adjusted to a range of 180 ° ± 90 °, more preferably 180 ° ± 45 °.
[0011]
The present invention is a high-frequency component that handles a plurality of transmission / reception systems having different passbands, a demultiplexing circuit for demultiplexing a high-frequency signal and a low-frequency signal, and a transmission system connected to the demultiplexing circuit. Or a switch circuit for switching connection with a reception system, wherein the branching circuit is configured by an LC circuit, and the switch circuit has a main configuration including a switching element and a transmission line, and an LC circuit of the branching circuit and the switch. At least a part of the transmission line of the circuit is constituted by an electrode pattern in a laminate of an electrode pattern and a dielectric layer, and the switching element or the chip element constituting a part of the LC circuit is a switch arranged on the laminate. A module unit, at least a semiconductor element, a power supply circuit, and a matching circuit, and at least a part of a transmission line and an LC circuit forming the power supply circuit and the matching circuit, A chip element that is formed by an electrode pattern in the laminate of the pole pattern and the dielectric layer, and that constitutes a part of the semiconductor element or the LC circuit; a high-frequency amplifier module unit disposed on the laminate; A phase adjustment circuit between a module unit and the switch module unit is constituted by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit is provided in the laminate of the electrode pattern and the dielectric layer, This is a high-frequency module configured by an electrode pattern, wherein the switch module section, the high-frequency amplifier module section, and the phase adjustment circuit are formed in the single laminate.
[0012]
The matching between the switch module unit and the high-frequency amplifier module unit via the phase adjusting circuit by the phase adjusting circuit constituted by the transmission line or the LC circuit corresponds to conjugate matching in a necessary basic frequency band. This is a high-frequency module that has a phase relationship, but has a phase relationship corresponding to non-conjugate matching in an unnecessary n-fold (n is a natural number of 2 or more) frequency band.
In addition, the transmission line or the phase adjustment circuit constituted by the LC circuit allows the impedance Z1 when the high-frequency amplifier side is viewed from the connection reference plane between the switch module section and the high-frequency amplifier module section, and the connection reference plane. When the relationship with the impedance Z2 when the switch side is viewed from the above is defined as θ1 when the phase in the relationship of conjugate matching of the phase θ of the Z1 is θ1, the phase θ2 of the Z2 is 180 ° ± 120 ° with respect to θ1. It is desirable to adjust the phase in the conjugate matching region. More preferably, the phase is adjusted to a non-conjugate matching region of 180 ° ± 90 °, more preferably 180 ° ± 45 °.
[0013]
In the high-frequency module of the present invention, the impedance Z1 when viewing the high-frequency amplifier side from the connection reference plane, the impedance Z2 when viewing the switch side from the connection reference plane, and the switch side from the switch side of the phase adjustment circuit. When the relationship with the impedance Z3 when viewed is that the phase in the relationship of the conjugate matching of the phase θ of Z1 is θ1 and the 180 ° opposite phase to θ1 is θ0, the phase θ3 of Z3 is counterclockwise with respect to θ0. When in the direction, it is desirable that a low-pass filter including an LC circuit is used as the phase adjustment circuit, and the phase θ2 of Z2 is adjusted so as to be closer to θ0 than θ3.
Incidentally, at this time, when a transmission line is used as the phase adjustment circuit, the phase of the Z2 is adjusted so as to be longer than the current transmission line length so that the phase θ2 is closer to θ0 than θ3. desirable.
[0014]
In the high-frequency module of the present invention, the impedance Z1 when viewing the high-frequency amplifier side from the connection reference plane, the impedance Z2 when viewing the switch side from the connection reference plane, and the switch side from the switch side of the phase adjustment circuit. When the relationship with the impedance Z3 when viewed is that the phase that is in the conjugate matching relationship with the phase θ of Z1 is θ1, and the 180 ° opposite phase to θ1 is θ0, the phase θ3 of Z3 is clockwise with respect to θ0. , It is preferable that a high-pass filter composed of an LC circuit is used as the phase adjustment circuit, and the phase θ2 of Z2 is adjusted so as to be closer to θ0 than θ3.
In this case, when a transmission line is used as the phase adjustment circuit, the phase θ2 of the Z2 is adjusted to be shorter than the current length of the transmission line so that the phase θ2 is closer to θ0 than θ3. desirable.
[0015]
Here, when the high-pass filter is formed of an LC circuit, it is desirable that one end of the inductor L, the other end of which is grounded, can be connected to the switch module side without using a capacitor.
In addition, when a low-pass filter or a high-pass filter is configured with a chip inductor and a chip capacitor, the chip inductor and the chip capacitor are mounted on the laminate, or are external components mounted on a circuit board. It is desirable that the adjustment can be made appropriately.
The present invention can be realized by interposing a coupler circuit between the high-frequency amplifier circuit and the switch circuit, or by configuring a coupler circuit instead of the switch circuit.
Further, according to the present invention, by mounting the above-described high-frequency component or high-frequency module, a communication device that is small and lightweight and has high power addition efficiency can be provided.
[0016]
BEST MODE FOR CARRYING OUT THE INVENTION
The inventor of the present application first studied the effect of the phase relationship between the high-frequency amplifier circuit and the antenna switch module circuit on the antenna output characteristics using the model shown in FIG. Specifically, the phase relationship between the output matching circuit end of the high-frequency amplifier circuit and the transmitting terminal of the antenna switch module is determined by placing the phase shifter PS in front of the antenna switch module ASM and setting the impedance on the high-frequency amplifier side as viewed from the connection reference plane. Z1 was fixed, the impedance Z2 on the antenna switch module side was changed, and the phase change, insertion loss, attenuation of the fundamental frequency, the second harmonic frequency, and the third harmonic frequency and their tendency were plotted on a Smith chart.
Note that, for convenience, the present invention will be described based on an example in which the impedance Z1 on the high-frequency amplifier side is fixed and the impedance Z2 on the antenna switch module side is changed and adjusted. The present invention can also be implemented by changing and adjusting the impedance Z1 on the amplifier side. Further, the present invention can be implemented by appropriately adjusting both sides (high-frequency amplifier HPA, antenna switch module ASM). These are also true of the claims. In this specification, the term "antenna switch module" is used as a name when the high-frequency module of the present invention is connected to a common antenna, and is substantially the same as the high-frequency component and the high-frequency module of the present invention. Are equivalent.
[0017]
As a result of the above studies, there is a close relationship between the amount of phase change and the antenna output characteristics (such as insertion loss and harmonic attenuation), and the mutual impedance characteristics in the nth harmonic band also strongly affect the overall characteristics. I learned to give. Summarizing with reference to FIG.
(1) When the phase on the antenna switch module side as viewed from the reference plane is the phase θ1 (−θ) obtained by inverting the sign of the phase θ on the high frequency amplifier side, the insertion loss and the attenuation are minimized. That is, the insertion loss is minimized at the phase θ1 corresponding to the conjugate matching of the impedance Z1. However, at this time, the amount of attenuation is also minimized.
(2) When the phase on the antenna switch module side as viewed from the reference plane is opposite to θ1, θ0 (θ1 ± 180 degrees), the insertion loss and the attenuation are maximized. That is, the attenuation becomes maximum at the phase θ0 opposite to the phase θ1 corresponding to the conjugate impedance. However, at this time, the insertion loss also becomes maximum.
Note that the conjugate impedance is an impedance having a complex conjugate relationship with a certain impedance. It refers to the relationship between impedances having equal resistance components and reactance components having equal magnitudes and opposite signs, and the impedance of θ (R + jX) and the impedance of θ1 (R-jX) are complex conjugate. Such a relationship is called conjugate matching, and impedance matching is realized. Although the conjugate matching by (R + jX) and (R-jX) is ideal and optimal here, it has been found from the present study that the effect can be obtained only by adjusting only the reactance portion of (R'-jX). . Further, it has been found that it is difficult to match all the impedances (R + jX), but it has a sufficient effect at least in consideration of the phase and is a practical adjustment means.
[0018]
In the present invention, it is possible to capture the relationship of the conjugate matching by the phase, satisfy the condition (1) in the necessary fundamental frequency band, minimize the insertion loss, and set the phase relationship corresponding to the conjugate matching of the impedance Z1. The first guideline. That is, LC is adjusted by a phase adjustment circuit that connects the high-frequency amplifier and the antenna switch module, and between them, approximately 50Ω and no conjugate matching are performed, and insertion loss is minimized.
Next, in the unnecessary second and third harmonic n-th frequency bands, the impedance Z2 is adjusted to the phase θ2 corresponding to the non-conjugate matching so as to satisfy (2) in order to maximize the harmonic attenuation. I do. Non-conjugate matching is a state other than conjugate matching. It was found that θ0, which is in the opposite phase to θ1, is the most desirable position, and that there was an allowable range. This point is one of the features of the present invention, which is defined as a region of a phase θ2 of θ0 ± 120 degrees that allows 5 dB deterioration with respect to the maximum attenuation. Preferably, it is θ0 ± 90 degrees allowing 3 dB degradation, and more preferably θ0 ± 45 degrees. The region was taken as a ± range because the conditions differed in each case and the moving direction was different. Therefore, a second guideline is to first shift the phase in the direction opposite to the phase θ1 corresponding to the conjugate matching, and more specifically, to adjust the phase within the range of θ2.
[0019]
Here, the phase adjustment circuit for performing the phase adjustment can be implemented by a transmission line or an LC circuit. Here, as an example, FIG. 5 shows an embodiment using a transmission line. A semiconductor element FET (field effect transistor) Q1 is provided on the amplifier output side, and power is supplied to a drain terminal D of the transistor Q1 from a power supply terminal Vdd1 via a transmission line or an inductor SL1. The drain terminal D is connected to the output terminal P0 via the transmission line ASL1 and the DC cut capacitor Ca2. On the transmitting side of the antenna switch, a DC cut capacitor Ca1 and a transmission line ASL2 are connected to a transmission terminal P1, and a transmission line L5, which is a part of a transmission system circuit, and a low-pass filter LPF including an LC circuit are connected behind the DC terminal. When connecting the output terminal P0 and the transmission terminal P1, one of the capacitors Ca1 and Ca2 can be omitted. Then, an LC circuit is inserted here and / or the transmission lines ASL1 and ASL2 are used as phase adjustment circuits. Since the transmission line ASL2 is required to connect the transmission terminal P1 and the low-pass filter LPF within the laminate, the transmission line ASL2 is used at the same time. In the actual setting, first, while adjusting the length or width of the transmission line ASL1 on the amplifier output side, insert capacitors Ca3 and Ca4 of appropriate capacitance at appropriate positions and adjust the input impedance of the antenna switch to approximately 50Ω matching. Aim. Next, the length or width of the transmission line ASL2 is adjusted as necessary. Further, the length and width of L5, the capacitance value of the parallel C on the LPF transmission terminal side, and the like may be adjusted as necessary.
[0020]
In addition, the following has been made clear regarding the adjusting means. As shown in FIG.
(A) When the phase θ3 of the impedance Z3 when the antenna switch is viewed from the antenna switch side of the phase adjustment circuit is counterclockwise with respect to θ0 on the Smith chart, the target phase region θ2 is the best than θ3. Adjust so as to be longer than the current transmission line length so as to approach the phase θ0, and adjust the length and width of L5, the capacitance value of the parallel C on the LPF transmission terminal side, etc. Move clockwise to adjust. The width of the transmission line may be reduced here, but qualitatively, when the position at this time is in the upper half circle of the Smith chart, it moves clockwise, but when it is in the lower half circle, Make a movement close to counterclockwise. Therefore, the means for reducing the width needs to consider the current position and the rotation direction.
(B) Similarly, when the phase θ3 is in the clockwise direction with respect to θ0, the direction is adjusted to be shorter than the current transmission line length so that the target phase region θ2 approaches the best phase θ0 than θ3, If necessary, adjust the length and width of L5, the capacitance value of the parallel C on the LPF transmission terminal side, and the like, and move θ3 in the counterclockwise direction. It is preferable that the transmission line ASL1 be adjusted to be thick and short. However, when the width of the transmission line is widened, it moves counterclockwise when you are in the upper half circle of the Smith chart, but it is almost clockwise when you are in the lower half circle. In this case, it is necessary to consider the current position and the rotation direction.
[0021]
If these adjustments are too late or you want to make fine adjustments after modularization,
(C) When the phase θ3 is in a counterclockwise direction with respect to θ0, a low-pass composed of the LC circuits shown in FIGS. 3A to 3C so that the target phase region θ2 is closer to the best phase θ0 than θ3. Insert a filter. Here, either one of the DC cut capacitors Ca1 and Ca2 can be omitted, so here, Ca1 is omitted and indicated by a dotted line. If the capacitor Ca1 on the antenna switch side is omitted in (2) and (4), a parallel resonance can be created with the transmission line L5 falling to the ground. In this case, the length of the transmission line L5 may be short, which is convenient for layout adjustment at the time of lamination.
(D) When the phase θ3 is in the clockwise direction with respect to θ0, a high-pass filter including the LC circuits of FIGS. 4A to 4C so that the target phase region θ2 approaches the best phase θ0 more than θ3. Insert Also in this case, one of the DC cut capacitors Ca1 and Ca2 can be omitted, and the other C operates even without the capacitor.Therefore, only the capacitor Ca2 or C on the high frequency amplifier side is left, and the other end is grounded. What is necessary is just to connect one end of the inductor L or the transmission line SL to the antenna switch side. Therefore, the phase adjustment when a high-pass filter is inserted is desirable because the circuit is simplified. The above adjusting means is also a feature of the present invention.
[0022]
From the above, when designing a circuit of a high-frequency composite component including a high-frequency amplifier, not only the fundamental component frequency but also the harmonic components from the semiconductor elements included in the component, for example, from the output matching circuit of the transistor to the connection portion with the antenna switch section are reduced. Also, it was confirmed that the transmission output characteristic from the antenna was significantly improved when the design was also made in consideration of the conventional method in which only the fundamental frequency was considered.
[0023]
Hereinafter, an embodiment (antenna switch module) of the high-frequency component of the present invention will be described with reference to the drawings, taking a mobile phone system as an example.
Generally, in a mobile phone system, a control signal (power control signal) is transmitted from a base station to a mobile phone so that a transmission output has a minimum power required for communication in order to avoid interference with surrounding mobile phones. Have been. An APC (Automatic Power Control) circuit that operates based on this control signal controls the gate voltage of the high-frequency amplifier at the transmission-side output stage so that the transmission output becomes an output necessary for a call. This is controlled by comparing a detection signal obtained by monitoring the power actually output from the high-frequency amplifier with a power control signal from the base station. In this way, the mobile phone communication system is constructed to make it difficult to cause interference with other mobile phones and maintain stable communication quality by making calls with variable output to adapt to the surrounding environment. Have been. In a digital mobile phone system in Europe, an output detection circuit of a high-frequency amplifier is roughly classified into two types. One is to attach a coupler circuit to the output terminal of the high-frequency amplifier and detect the output power.The other is to attach a resistor of about 1 to 10Ω to the high-frequency amplifier and calculate the power consumption from the voltage drop. There are two methods of converting to. Generally, the former is realized by forming a circuit on a laminate, and the latter has various kinds of deriving means. For example, it is realized by circuit integration on a mounted component or a semiconductor chip. The following embodiment is a case where a semiconductor chip having a function of monitoring output power in the latter method is used in advance.
FIG. 6 of the embodiment shows an equivalent circuit diagram of an antenna switch module for EGSM, DCS, and PCS triple bands. FIG. 7 shows an equivalent circuit diagram of the high-frequency amplifier. FIG. 5 shows the periphery of a phase matching circuit connecting the two. Further, in this embodiment, a high-frequency module in which the range shown by the solid line in FIG. 17 is combined into one laminate is shown, and FIG. 10 is a partially developed view of the laminate sheet.
[0024]
In FIG. 6, the diplexer Dip includes transmission lines L1 to L4 and capacitors C1 to C4. The transmission line L2 and the capacitor C1 form a series resonance circuit, and are used in the DCS band (transmission frequency: 1710-1785MHz, reception frequency: 1805-1880MHz) and PCS band (transmission frequency: 1850-1910MHz, reception frequency: 1930-1990MHz) Design to have a resonance frequency. In this example, the attenuation pole was adjusted to 1.8 GHz. The transmission line L4 and the capacitor C3 form a series resonance circuit and are designed to have a resonance frequency in the EGSM band (transmission frequency: 880 to 915 MHz, reception frequency: 925 to 960 MHz). In this example, the attenuation pole was adjusted to 0.9 GHz. With this circuit, it is possible to separate and combine the EGSM-based signal and the DCS / PCS-based signal. The transmission lines L1 and L3 are preferably set to a certain length so as to have a high impedance with respect to the frequency of the DCS / PCS signal. This makes it difficult for DCS / PCS signals to be transmitted to EGSM paths. Note that the transmission line L3 may be omitted. Conversely, the capacitances C2 and C4 are preferably set to relatively small capacitance values so as to have a high impedance at the frequency of the EGSM signal. This makes it difficult for EGSM signals to be transmitted to DCS / PCS paths.
[0025]
The first switch circuit SW1 includes capacitors C5 and C6, transmission lines L5 and L6, PIN diodes D1 and D2, and a resistor R1. The lengths of the transmission lines L5 and L6 are set so as to be λ / 4 resonators in the transmission frequency band of EGSM. However, the transmission line L5 can be replaced by a choke coil whose ground level looks open (high impedance state) at the transmission frequency of the EGSM. In this case, the inductance value is desirably about 10 to 100 nH. The resistor R1 determines a current flowing through the first and second diodes D1 and D2 when the control power supply VC1 is in a high state. In this example, 100Ω to 200Ω was used. Capacitors C5 and C6 are necessary for DC cut of the control power supply. When the control power supply VC1 is high, since the PIN diode D2 has a parasitic inductance such as a connection wire, the PIN diode D2 is series-resonated with the capacitor C6 so as to cancel the parasitic inductance. The capacitance value of the capacitor C6 is appropriately set.
[0026]
As described above, when the control power supply VC1 is High, the first and second diodes D1 and D2 are both turned on, the connection point of the second diode D2 and the transmission line L6 is at the ground level, and the transmission is a λ / 4 resonator. The impedance on the opposite side of the line L6 becomes infinite. Therefore, when the control power supply VC1 is high, a signal cannot pass through a path between the diplexer Dip and the EGSM Rx, and a signal easily passes through a path between the diplexer Dip and the EGSM Tx. On the other hand, when the control power supply VC1 is low, the first diode D1 is also turned off, and the signal cannot pass through the path between the diplexer Dip and the EGSM Tx, and the second diode D2 is also off. In the path, the signal easily passes. With the above configuration, it is possible to switch between transmission and reception of the EGSM signal.
[0027]
The second switch circuit SW2 includes capacitors C7 to C10, transmission lines L7 to L10, PIN diodes D3 to D6, and resistors R2 and R3. The lengths of the transmission lines L7 to L10 are set so as to be λ / 4 resonators at the frequency of the DCS / PCS signal. However, each of the transmission lines L7 and L9 can be replaced by a choke coil whose ground level looks open (high impedance state) at the transmission frequency of the PCS at the transmission frequency of the DCS. In this case, the inductance value is desirably about 5 to 60 nH. The resistor R2 determines the current flowing through the third and fourth diodes D3 and D4 when the control power supply VC2 is in the high state. In this example, 100Ω to 200Ω was used. The resistor R3 determines the current flowing through the fifth and sixth diodes D5 and D6 when the control power supply VC3 is in the high state. In this embodiment, 100Ω to 2kΩ is used. Capacitors C7, C8 and C10 are necessary for DC cut of the control power supply. When the control power supply VC2 is high, since the PIN diode D4 has a parasitic inductance such as a connection wire, the capacitance value of the capacitor C7 is set so as to resonate in series with the capacitor C7.
[0028]
As described above, when the control power supply VC2 is High, the third and fourth diodes D3 and D4 are both turned on, the connection point of the fourth diode D4 and the transmission line L8 is at the ground level, and the transmission is a λ / 4 resonator. The impedance on the opposite side of the line L8 becomes infinite. Therefore, when the control power supply VC2 is High, signals cannot pass through the path between the diplexers Dip to PCS Rx and between the diplexers Dip and DCS Rx, and signals easily pass through the path between the diplexers Dip and DCS / PCS Tx. On the other hand, when the control terminal VC2 is low, the third diode D3 is also turned off, so that no signal can pass through the path between the diplexer Dip and DCS / PCSTx, and since the fourth diode D4 is also off, the diplexers Dip to PCS Rx and In the path between the diplexer Dip and the DCS Rx, a signal can easily pass.
[0029]
Further, when the control terminal VC3 is High, since the PIN diode D6 has a parasitic inductance such as a connection wire, the capacitance value of the capacitance C10 is set so as to resonate in series with the capacitance C10. As a result, when the control terminal VC3 is high, the fifth and sixth diodes D5 and D6 are both turned on, the connection point between the sixth diode D6 and the transmission line L10 is at the ground level, and the transmission is a λ / 4 resonator. The impedance on the opposite side of the line L10 becomes infinite. Therefore, when the control terminal VC3 is high, no signal can pass through the path between DCS Rx, and the signal easily passes through the path between PCS Rx. Conversely, when the control terminal VC3 is low, the fifth diode D5 is also turned off, so that the signal cannot pass through the path between PCS Rx and the sixth diode D6 is also off, so that the signal is not transmitted through the path between DCS Rx. Easy to pass. With the above configuration, switching to DCS / PCS Tx when the control terminal VC2 is High, switching to PCS Rx when the control terminals VC2 and VC3 are Low and High, and switching to DCS Rx when the control terminals VC2 and VC3 are Low Becomes possible.
[0030]
The first low-pass filter LPF1 is a π-type low-pass filter including a transmission line L11 and capacitors C11 to C13. Here, L11 and C11 form a parallel resonance circuit, and the resonance frequency is set to twice or three times the transmission frequency of EGSM. In this embodiment, the frequency is set to 2.7 GHz which is three times as large as that of the first embodiment. With the above configuration, it is possible to remove the harmonic distortion included in the transmission signal on the EGSM side input from the power amplifier. Accordingly, it is preferable in terms of characteristics to provide the low-pass filter LPF1 in the transmission path, but it is not always essential. Further, the first low-pass filter LPF1 is disposed between the first diode D1 of the first high-frequency switch SW1 and the transmission line L5, but is disposed between the diplexer Dip and the first high-frequency switch SW1. Alternatively, it may be arranged between the transmission line L5 and EGSMTx. If the capacitance connected to the ground of the first low-pass filter LPF1 is arranged in parallel with the transmission line L5, a parallel resonance circuit is formed, and the line length of the transmission line L5 can be made shorter than λ / 4. The inductance value of the choke coil can be reduced.
[0031]
The second low-pass filter LPF2 is a π-type low-pass filter including the transmission line L12 and the capacitors C14 to C16. Here, the transmission line L12 and the capacitor C14 form a parallel resonance circuit, and the resonance frequency is set to twice or three times the DCS / PCS transmission frequency. In this embodiment, it is set to 3.6 GHz, which is twice as large. With the above configuration, it is possible to remove harmonic distortion included in the DCS / PCS side transmission signal input from the power amplifier. Therefore, it is preferable in terms of characteristics to provide the low-pass filter LPF2 in the transmission path, but it is not always essential. Also, the second low-pass filter LPF2 may be disposed between the diplexer Dip and the second high-frequency switch SW2, similarly to the first low-pass filter LPF1, or the transmission line L7 and the DCS transmission terminal DCS Tx may be disposed. It may be arranged between them. The first and second low-pass filters LPF1, LPF2 are configured between the diode D1 and the transmission line L5 and between the diode D3 and the transmission line L7, and are provided in the switch circuit. This is preferable in circuit design but is not essential. The low-pass filter may be provided at any position on the transmission path between the diplexer through which the transmission signal passes and the transmission terminal.
Table 1 summarizes the control logic of the above high-frequency components (antenna switch module).
[0032]
[Table 1]

[0033]
In some embodiments, the EGSM system can be further divided into GSM850 (transmitting frequency: 824 to 849 MHz, receiving frequency: 869 to 894 MHz) and EGSM to support quad bands. In this case, a common terminal can be used for the transmission system, and the reception system can be configured by connecting a switch for switching between GSM850 and EGSM to the EGSM reception terminal of the triple-band compatible antenna switch. In addition, it can also be realized by using a transmission line that is a λ / 4 resonator in the GSM850 or EGSM band instead of the switch and dividing the frequency between the two.
[0034]
FIG. 7 is a circuit diagram of the high-frequency amplifier constituting the high-frequency component of this embodiment. The output terminal P0 at the end of the matching circuit on the high-frequency amplifier side is connected to the transmission terminal P1 of, for example, EGSM Tx of the antenna switch module of FIG. 6, and plays a role of transmitting the amplified transmission signal to the antenna switch side. One end of the transmission line ASL1 is connected to the output terminal P0 via the DC cut capacitor Ca2. The transmission line ASL1 is connected to capacitors Ca3 and Ca4 whose one ends are grounded to form an output matching circuit. The other end of the transmission line ASL1 is connected to the drain of a field effect switching transistor (FET) Q1, which is a type of semiconductor device. The source of the FET Q1 is grounded, and the gate is connected to the collector of the bipolar switching element (B-Tr) Q2.
[0035]
On the other hand, the connection point between the other end of the transmission line ASL1 and the drain D of the field effect switching transistor FET Q1 is grounded via a series circuit of an inductor SL1 and a capacitor Ca5 formed of a λ / 4 strip line or the like. The connection point with the capacitor Ca5 is connected to the drain voltage terminal Vdd1. The connection point between the gate of the field effect switching transistor FET Q1 and the collector of the bipolar switching element Q2 is grounded via a capacitor Ca6 and is also connected to a gate voltage terminal Vg.
[0036]
Further, the emitter of the bipolar switching element Q2 is grounded, and the base is connected to one end of the transmission line SL3. The collector of the bipolar switching element Q2 is grounded via a series circuit of an inductor SL2 and a capacitor Ca7, such as a strip line, and a connection point between the inductor SL2 and the capacitor Ca7 is connected to a collector voltage terminal Vc. The connection point between the inductor SL2 and the capacitor Ca7 is also connected to a connection point between the base of the bipolar switching element Q2 and one end of the transmission line SL3. The other end of the transmission line SL3 is grounded via the capacitor Ca8 and is connected to the input terminal Pin.
Although the transmission line and the inductor in the equivalent circuits of FIGS. 6 and 7 are often constituted by strip lines, they may be constituted by microstrip lines, coplanar guidelines, or the like. On the amplifier circuit side, the semiconductor element Q3 and the power supply circuit can be similarly added, and the amplifier circuit can be configured as a high power amplifier having three stages or more stages. Although Q1 is an FET and Q2 is a B-Tr transistor, other types of transistors may be used. For example, Si-MOSFET, GaAsFET, Si bipolar transistor, GaAsHBT (heterojunction bipolar transistor), HEMT (high electron mobility transistor) and the like can be mentioned. Of course, an MMIC (monolithic microwave integrated circuit) in which a number of transistors are integrated may be used. In this embodiment, the transmission line SL3 and the transistor Q2 are directly connected, but may be connected via a resistor.
[0037]
FIG. 5 shows the periphery of the phase adjustment circuit. The phase adjustment circuit can be implemented by inserting an LC circuit such as a transmission line or a high-pass filter or a low-pass filter, and the setting guidelines and the adjustment method are as described above. Here, an embodiment in which the high-pass filter shown in FIG. 4B is inserted between the EGSM Tx terminals P1 and P0 will be described. FIG. 8 is a plot of the trends of the fundamental wave (f), the second harmonic (2f), and the third harmonic (3f) when the high-frequency amplifier is fixed and the antenna switch module is adjusted, on a Smith chart. is there. The phase seen on the power amplifier side is in the region of θ (upper left) for both the second harmonic (2f) and the third harmonic (3f). The length of the transmission line ASL1 of the output matching circuit and the addition of a capacitor are adjusted to establish a conjugate matching relationship, and the fundamental wave (f) on the antenna switch module side is adjusted to a conjugate matching relationship at approximately 50Ω at the center (approximately). ■ in the center dotted circle). The current positions of the 2nd harmonic (2f) and 3rd harmonic (3f) are in the conjugate matching θ1 region (lower left). For this, a high-pass filter consisting of an inductor of 18nH and a capacitor of 6pF was inserted. As a result, the phase shifts to the above-described region of the phase θ2 (lower right). In this example, the second harmonic (2f) is shifted to a region of θ0−90 degrees closer to the target θ0 region (upper right). The harmonic (3f) can be adjusted to the range of θ0-120 degrees. Although the phase and direction to be adjusted vary depending on the case, the fundamental is to give priority to the second harmonic (2f), which is the main component of the unnecessary harmonic, to approach θ0. In some cases, the third harmonic (3f) is It can be considered at the cost of being outside the allowable range of θ2.
[0038]
FIG. 9 shows the relationship between the phase of the second harmonic (2f) and the third harmonic (3f) and the amount of attenuation in the EGSM (880 to 960 MHz). That is, FIG. 9A shows the phase-attenuation characteristic of the second harmonic, FIG. 9B shows the phase-attenuation characteristic of the third harmonic, the attenuation (dB) is plotted on the vertical axis, and the antenna is plotted on the horizontal axis. Indicates the input phase (deg) on the switch module side. According to (A), the phase θ0 showing the maximum attenuation is around -73 (dB) of 30 degrees. It is satisfactory if the phase can be adjusted up to this phase, but it is considerable to provide an allowable range in actual design. For example, it can be seen that the range in which the attenuation of about 3 dB can be tolerated is 180 degrees (θ0 ± 90 degrees), and the range in which the degradation of about 5 dB can be tolerated is 240 degrees (θ0 ± 120 degrees). This is the same result in the attenuation characteristic of the third harmonic of (B). Actually, since -70 (dB) is secured even with 3dB degradation and -65 (dB) or more with 5dB degradation, it has been found that adjusting the phase within ± 120 degrees with respect to the best θ0 phase is sufficiently effective. Then, it can be said that it is preferably about ± 90 degrees, and more preferably about ± 45 degrees. It was also found that the same effect can be obtained for the DCS / PCS system in the same range.
[0039]
Next, FIG. 10 shows a dielectric module of two upper layers and two lower layers of a laminated module in which the antenna switch module shown by the equivalent circuit of FIG. 6 and the high-frequency amplifier shown by the equivalent circuit of FIG. It is a development view which extracted a green sheet. The green sheet (1) is composed of the uppermost layer and 15 layers in the following order, and the last sheet (15) shows the back surface of the laminate. The green sheet used in this embodiment is made of an LTCC material that can be simultaneously fired at a low temperature of 950 ° C. or lower. For example, Al ₂ O ₃ 10-60 mass% in conversion, SiO ₂ 25 to 60% by mass in conversion, 7.5 to 50% by mass in SrO conversion, TiO ₂ Al, Si, Sr, Ti of not more than 20% by mass in conversion and Bi ₂ O ₃ 0.1 to 10% by mass in conversion, Na ₂ 0.1 to 5% by mass in terms of O, K ₂ 0.1 to 5% by mass in terms of O, 0.01 to 5% by mass in terms of CuO, MnO ₂ A dielectric composition containing 0.01 to 5% by mass of Bi, Na, K, Cu, and Mn in terms of conversion is used.
The green sheet used had a sheet thickness of 40 to 200 μm so that a transmission line and a capacity could be easily formed. A silver-based electrode material was used. A transmission line and a capacitor were formed in each layer of the ceramic green sheet by an electrode pattern, and through holes were appropriately provided to form a circuit. The ceramic green sheets are sequentially laminated and pressed and fired at 950 ° C. to obtain a laminated module in which high frequency components are combined. The size of the laminated body is about 13.75 mm wide × 8 mm long × 0.75 mm high, and a diode, transistor, chip inductor, chip capacitor is mounted on the top of the laminated body, and a metal case is placed on top of it to complete the product And The overall height after completion is about 1.8mm. However, instead of the metal case, a resin-sealed package may be used, and the total height in this case is about 1.5 mm. In addition, another laminate having a size of 10 mm wide × 8 mm long × 0.75 mm high was also implemented. Also in this case, the total height of the finished product is about 1.8 mm in the case of a metal case, and about 1.5 mm in the case of a resin-sealed package.
[0040]
The schematic configuration inside the laminate, the antenna switch module side, the transmission line constituting the duplexer circuit and the low-pass filter in the upper layer, the duplexer circuit, the capacitor capacitance constituting the switch circuit and the low-pass filter in the middle layer, A transmission line constituting a switch circuit is mainly formed in a lower layer. On the other hand, on the high-frequency amplifier side, the transmission line is mainly used for the first-stage matching circuit in the upper layer, the capacitor is mainly used for the first-stage and the second-stage matching circuit in the middle layer, and the transmission line and power supply are mainly used for the thermal via and the second-stage matching circuit in the lower layer A supply line is mainly formed. The connection between the high-frequency amplifier and the antenna switch module is in the upper layer. To avoid mutual interference, the transmission line ASL1 of the green sheet (1) and the transmission line ASL2 of the green sheet (2) are changed layers and do not overlap each other. It is formed in a suitable position. In the high-pass filter interposed between the two, the LC circuit is composed of a chip inductor and a chip capacitor, and is mounted on the upper surface of the laminate. As a result, the adjustment can be performed even after the laminate module is created, and thus it is desirable that the adjustment of the prototype does not take much time.
[0041]
In this laminated module, the electrode pattern forming the high-frequency amplifier HPA is formed in the left region, while the electrode pattern forming the antenna switch module ASM is formed in the right region, and the green sheet is divided into two regions in all the laminating directions. It is composed. As a result, since the circuits for each function are provided together, the size can be reduced without deteriorating the characteristics. Further, a strip-shaped ground electrode G0 is provided between the left and right regions, whereby the layout design of each electrode pattern can be made relatively simple, and mutual interference between high-frequency components can be suppressed. It is desirable to provide such a ground electrode G0 on all the green sheets if there is room in the dimensional arrangement. However, since it is not possible in many cases due to miniaturization, as shown in the green sheet {circle around (2)}, the ground electrode G0 is not provided. Intermittently provided with a through-hole electrode HG that leads to the function of the ground electrode G0. Here, since the through-hole electrode HG is in the shape of a blind, it is necessary to consider the interval, and the interval g of the through-hole electrode HG is desirably set to 1/10 or less of the wavelength (λ) of the highest frequency at which interference is to be prevented. . It has been found that the shielding effect is preferably enhanced at about λ / 10 to λ / 50. In this embodiment, the interval g is unequal, but is set to approximately λ / 20 (approximately 1 mm) to λ / 25 of the third harmonic of the DCS band.
As described above, the ground between the ground electrode G0 and the through-hole electrode HG eliminates interference between the two, and can prevent unstable operation such as oscillation of the high-frequency amplifier. Further, it is possible to suppress the occurrence of spurious signals between the necessary signal (transmission signal) and the unnecessary signal, and prevent deterioration of the transmission characteristics. Furthermore, since the high-frequency components are integrated into one laminate, the mounting area on the printed wiring board in the mobile phone is 25% to 50% compared to the case where the conventional power amplifier and antenna switch are separately mounted on the board. The size can be reduced. Therefore, by mounting it on a communication device such as a small information terminal such as a mobile phone or a PDA, it is possible to meet the need for a smaller and lighter device.
[0042]
Further, by integrating the high-frequency amplifier HPA and the antenna switch module ASM in the laminate as in the embodiment, the length of the line connecting the respective circuits is shortened, and the line loss can be reduced even in the laminate. No matching deviation occurs. In addition, the wiring that was conventionally required to connect the two is not required on the printed wiring board. Since the high-frequency module is directly mounted on the printed wiring board, the line loss can be reduced, and no misalignment occurs. In addition, since the high-frequency amplifier HPA and the antenna switch module ASM can be integrally designed, settings can be made to optimize mutual impedance matching, and the matching circuits provided in each of the conventional HPA and ASM can be simplified. Can be With such an optimal design, a compact, high-frequency module with low insertion loss, high output power, and high output efficiency has been completed.
Speaking of characteristics, conventionally, when transmitting GSM, the efficiency was about 35%, the second harmonic -25 dBm, and the third harmonic -25 dBm, but according to the present embodiment, the efficiency is 43% or more and the second harmonic The characteristics were improved to -38 dBm or less and the third harmonic -36 dBm or less. At this time, the DCS / PCS transmission characteristics achieved an efficiency of 34% or more, a second harmonic of -38 dBm or less, and a third harmonic of -36 dBm or less. As a result, the effectiveness of the present invention could be confirmed not only in the GSM band but also in the DCS / PCS band.
Due to the improvement of the characteristics described above, when this high-frequency module is used for a mobile phone, it is possible to increase the efficiency by about 5 to 10% as compared with a case where components are separately mounted as in the conventional case. As a result, the power consumption during transmission was reduced, the battery life was improved, and long-term calls with a talk time of about 10 to 20% per charge were made possible.
In addition, as described above, mobile terminal manufacturers have been evaluating combinations of multiple types of high-frequency amplifiers and antenna switch modules in order to obtain desired characteristics, and matching them on printed circuit boards in order to make full use of them. Various measures have been taken, such as forming circuits and harmonic filters, but these can all be omitted.
[0043]
The equivalent circuit of the antenna switch shown in FIG. 6 and the equivalent circuit of the high-frequency amplifier shown in FIG. 7 are examples. For example, although an example in which a pin diode is used as the switch circuit has been described, a switch circuit can be configured using a SPnT type GaAs switch such as a single pole dual throw (SPDT) or SP3T. In this case, if the pin diode switch is simply replaced with a GaAs switch of SPDT, the lambda / 4 line required for the pin diode switch is not required, so that the laminated body can have a sufficient amount. For this reason, it is advantageous for further miniaturization and higher integration by reducing this space or forming a new functional element. Further, the whole including the duplexer can be replaced. In this case, if a triple band antenna switch is taken as an example, all paths can be switched by replacing with a SP5T GaAs switch. In addition, several λ / 4 lines are not required as described above, which is advantageous for further miniaturization and high integration. However, the low-pass filter on the transmitting side and various filters to be inserted to satisfy the characteristics are constituted by a laminate and / or mounted components. In this case, since the GaAs switch is directly connected to the antenna, it is preferable that the GaAs switch is provided with a countermeasure against electrostatic surge. In the case of using a switch that is not designed for surge suppression, it is necessary to insert a surge suppression circuit such as a filter including an LC circuit between the antenna and the GaAs switch.
[0044]
An embodiment in which a triple band antenna switch circuit is formed using a GaAs switch will be described below with reference to FIGS. Integration with the high-frequency amplifier HPA can be implemented by replacing the left block in the block diagram of FIG. 17 with the block diagrams of FIGS. In FIG. 12, an SPDT (SP2T) switch SW1 and an SP3T switch SW2 are used. First, the diplexer Dip separates, for example, the EGSM band on the low frequency side and the DCS / PCS band on the high frequency side, and switches EGSM band transmission (connected to PA1) and reception (RX1) with switch SW1, and DCS with switch SW2. DCS / PCS transmission in the / PCS band (connected to PA2), DCS reception (RX2) or PCS reception (RX3) are switched. FIG. 13 shows an example in which only three SP2T switches are used. In this case, as in the above case, the diplexer Dip separates the EGSM band on the low frequency side and the DCS / PCS band on the high frequency side of the communication system, and the switch SW1 transmits (connects to the PA1) and receives the EGSM band. Switch (RX1), switch DCS / PCS transmission (connected to PA2) and DCS / PCS reception with switch SW2, and switch DCS reception (RX2) and PCS reception (RX3) with switch SW3. I have. FIG. 14 shows an example in which an SP5T switch is used. In this case, the diplexer Dip is not used, and all routes are switched only by switches. In FIGS. 12 and 13, the circuit block of the GaAs SPDT switch may be appropriately replaced with an SPDT switch using a PIN diode. Even when the antenna switch module is configured using the GaAs switch in this manner, the matching (phase) relationship between the high-frequency amplifier HPA and the antenna switch module ASM is described above, similarly to the above-described embodiment using the PIN diode. The phase adjustment circuit makes the conjugate matching relationship in the pass band and adjusts the phase to the non-conjugate matching relationship in the unnecessary harmonic band, thereby improving the characteristics of the entire high-frequency module. Further, it can be configured as an integrated module including an APC circuit for controlling the gate voltage of the output transistor based on the detection value of the output detector.
[0045]
The high-frequency module of the present invention may include a coupler circuit or an isolator circuit between the antenna switch module and the high-frequency amplifier, and may insert a SAW filter in the receiving system path. In the above embodiment, the case where the function of monitoring the output power is combined with the semiconductor chip of the high-frequency amplifier has been described. On the other hand, a configuration using a technique of detecting power by a coupler circuit is also possible. An embodiment in this case will be described with reference to FIGS. As shown in FIG. 15, the coupler CPL is used by being inserted between the low-pass filter LPF of the antenna switch module and the high-frequency amplifier PA. A transmission path CSL1 for high-frequency output from the high-frequency amplifier HPA shown in FIG. 16 is generally called a main line, and a transmission line CSL2 for extracting a part of output power is generally called a coupling line. One end of the coupling line is sent to a detector as an output monitor, and is used for output power control of the high-frequency amplifier HPA. On the other hand, the other end is terminated by a resistor R. Generally, termination is performed at 50Ω, but it can be changed as appropriate to adjust the degree of coupling and isolation. In such a configuration, a coupler is inserted between the high-frequency amplifier HPA and the antenna switch module ASM. In this case, instead of adjusting the relative phase between the high-frequency amplifier and the antenna switch module, the high-frequency amplifier HPA is not adjusted. A phase adjustment circuit is provided as in the above-described embodiment to adjust the relative phase between the coupler and the coupler CPL so that the pass band has a conjugate matching relationship and the unnecessary harmonic band has a non-conjugate matching relationship. In this case, it is desirable that the coupler CPL and the antenna switch module ASM be integrally designed so as to obtain desired characteristics. As a more desirable improvement, the present invention is implemented in which the relative phase between the coupler CPL and the antenna switch module ASM is adjusted to a conjugate matching relationship in the pass band and to a non-conjugate matching relationship in the unnecessary harmonic band as described above. By doing so, the overall characteristics are significantly improved.
[0046]
Usually, when a coupler CPL is used, a loss of about 0.2 to 0.3 dB due to the coupler circuit generally occurs. However, by optimizing the design of the high-frequency amplifier HPA, coupler circuit CPL, and antenna switch module ASM in a single laminate, 25 to 50% compared to the conventional case where each circuit is separately mounted on a printed wiring board. And a high efficiency of about 2 to 7% can be achieved. This reduces power consumption during transmission and improves battery life. For example, a long-term call with a talk time of about 5 to 15% per charge is possible. Furthermore, it can be configured as an integrated module including an APC circuit that controls the gate voltage of the output transistor based on the detection value of the coupler.
[0047]
Incidentally, as the transmission and reception system used in the present invention, in addition to the above, PDC800 band (810 to 960 MHz), GPS band (1575.42 MHz), PHS band (1895 to 1920 MHz), Bluetooth band (2400 to 2484 MHz), Similar effects can be expected in the case of a multi-band compatible antenna switch circuit that combines CDMA2000, which is expected to spread in the United States, and TD-SCDMA, which is expected to spread in China. By using the circuits in these cases, a multi-mode multi-band antenna switch circuit such as a dual-band, three-band, four-band, or five-band antenna can be obtained.
[0048]
【The invention's effect】
ADVANTAGE OF THE INVENTION According to this invention, while minimizing the insertion loss in the required fundamental frequency band, and maximizing the amount of harmonic attenuation in the unnecessary frequency band, the antenna with excellent antenna output characteristics with no loss and high power addition efficiency is provided. A high frequency composite part for a band was obtained. Furthermore, a multi-band high-frequency module in which two or two or more functions are integrated in one laminated body is inexpensive and lighter and smaller. And a high-performance communication device using these could be provided.
[Brief description of the drawings]
FIG. 1 is an explanatory diagram on a Smith chart for explaining phase adjustment of a multiband high-frequency composite component of the present invention.
FIG. 2 is an explanatory diagram of a simple Smith chart similarly illustrating a procedure of phase adjustment.
FIG. 3 is a circuit diagram showing a low-pass filter which is one of specific means for phase adjustment.
FIG. 4 is a circuit diagram showing a high-pass filter which is one of specific means for phase adjustment.
FIG. 5 is a circuit diagram showing an output matching circuit terminal of a high-frequency amplifier and a vicinity of a transmission terminal on the antenna switch module side, showing a transmission line as one of specific means for phase adjustment.
FIG. 6 is an equivalent circuit diagram of the triple-band antenna switch module according to one embodiment of the present invention.
FIG. 7 is an equivalent circuit diagram of the high-frequency amplifier according to one embodiment of the present invention.
FIG. 8 is a Smith chart showing a state of phase adjustment when a high-pass filter is used according to an embodiment of the present invention.
9A and 9B are graphs showing the relationship between the phase adjustment and the attenuation according to the present invention, wherein FIG. 9A is a characteristic diagram of the phase-two harmonic attenuation, and FIG. 9B is a characteristic diagram of the phase-third harmonic attenuation.
FIG. 10 is an exploded view of a green sheet of a laminate showing one embodiment of the multi-band composite laminate module of the present invention.
FIG. 11 is a block diagram of a model in which phase adjustment of the present invention is considered.
FIG. 12 is a block diagram of one embodiment in which a triple band antenna switch module is configured using a GaAs switch.
FIG. 13 is a block diagram of an embodiment using a GaAs switch.
FIG. 14 is a block diagram of an embodiment using still another GaAs switch.
FIG. 15 is a block diagram showing an embodiment when a coupler circuit is inserted between the antenna switch module and the high-frequency amplifier.
FIG. 16 is a diagram showing one embodiment of a coupler circuit.
FIG. 17 is a block diagram illustrating an embodiment of a multi-band high-frequency component and a high-frequency module according to the present invention.
[Explanation of symbols]
ASM: Antenna switch module
HPA: High power amplifier
Dip: Diplexer (demultiplexer, demultiplexer)
SW: Switch circuit
LPF: Low-pass filter circuit
SAW: Surface acoustic wave filter
L, SL, ASL: Inductor, transmission line
C, Ca: Capacitor
Q1, Q2: semiconductor switching element
G0: Ground electrode
HG: Ground electrode with through hole

Claims (13)

A high-frequency component that handles a plurality of transmission / reception systems having different pass bands,
A demultiplexing circuit for demultiplexing a high-frequency signal and a low-frequency signal,
A switch circuit that is connected to the demultiplexing circuit and switches connection with a transmission system or a reception system,
A high-frequency amplifier circuit having at least a semiconductor element, a power supply circuit, and a matching circuit,
Having a phase adjustment circuit between the switch circuit and the high-frequency amplifier circuit,
The matching between the switch circuit and the high-frequency amplifier circuit via the phase adjustment circuit has a phase relationship corresponding to conjugate matching in a necessary basic frequency band, but is unnecessary n times (n is a natural number of 2 or more). A high-frequency component having a phase relationship corresponding to non-conjugate matching in the frequency band described above. The relationship between the impedance Z1 when looking at the high-frequency amplifier side from the connection reference plane between the switch circuit and the high-frequency amplifier circuit, and the impedance Z2 when looking at the switch circuit side from the connection reference plane is represented by Z1. 2. The high-frequency wave according to claim 1, wherein, when a phase having a conjugate matching relationship with the phase θ is defined as θ1, the phase θ2 of the Z2 is adjusted to a non-conjugate matching region of 180 ° ± 120 ° with respect to θ1. parts. A high-frequency component that handles a plurality of transmission / reception systems having different pass bands,
A demultiplexing circuit for demultiplexing a high-frequency signal and a low-frequency signal,
A switch circuit that is connected to the demultiplexing circuit and that switches a connection with a transmission system or a reception system,
The demultiplexing circuit is composed of an LC circuit, the switch circuit is mainly composed of a switching element and a transmission line, and the LC circuit of the demultiplexing circuit and at least a part of the transmission line of the switch circuit are formed of an electrode pattern and a dielectric. A switch module portion configured by an electrode pattern in a layered body and a chip element constituting a part of the switching element or the LC circuit;
At least a part of a transmission line and an LC circuit that constitute the matching circuit and the power supply circuit and the matching circuit have at least a semiconductor element, a power supply circuit, and a matching circuit. A high-frequency amplifier module unit configured by a pattern, wherein the semiconductor element or a chip element configuring a part of the LC circuit is disposed on the laminate.
A phase adjustment circuit between the high-frequency amplifier module section and the switch module section is constituted by a transmission line or an LC circuit, and at least a part of the transmission line or the LC circuit is formed by stacking an electrode pattern and a dielectric layer. In the body, constituted by the electrode pattern,
The high-frequency module, wherein the switch module unit, the high-frequency amplifier module unit, and the phase adjustment circuit are formed in the one laminate. By the phase adjustment circuit constituted by the transmission line or the LC circuit, the matching of the switch module unit and the high-frequency amplifier module unit via the phase adjustment circuit is performed in a required basic frequency band. 4. The high-frequency module according to claim 3, wherein in a frequency band of n times (n is a natural number of 2 or more) unnecessary, a phase relationship corresponding to non-conjugate matching exists. By the transmission line or the phase adjustment circuit formed of the LC circuit, the impedance Z1 when the high-frequency amplifier side is viewed from the connection reference plane between the switch module section and the high-frequency amplifier module section, and the switch from the connection reference plane. When the relationship with the impedance Z2 when looking at the side is a phase that is in a conjugate matching relationship with the phase θ of the Z1, the phase θ2 of the Z2 is 180 ° ± 120 ° non-conjugate matching with respect to θ1. 5. The high-frequency module according to claim 3, wherein the phase is adjusted in the region. Impedance Z1, when viewing the high-frequency amplifier side from the connection reference plane,
Impedance Z2 when viewing the switch side from the connection reference plane,
The relationship with the impedance Z3 when viewing the switch side from the switch side of the phase adjustment circuit,
The phase in the conjugate matching relationship of the phase θ of Z1 is θ1,
When 180 ° opposite phase to θ1 is θ0,
When the phase θ3 of Z3 is counterclockwise with respect to θ0,
6. The high-frequency module according to claim 3, wherein a low-pass filter including an LC circuit is used as the phase adjustment circuit, and the phase θ2 of Z2 is adjusted so as to be closer to θ0 than θ3. 7. When the transmission line is used as the phase adjusting circuit, the Z2 phase is adjusted so as to be longer than the current length of the transmission line so that the phase θ2 is closer to θ0 than θ3. The high-frequency module as described. Impedance Z1, when viewing the high-frequency amplifier side from the connection reference plane,
Impedance Z2 when viewing the switch side from the connection reference plane,
The relationship with the impedance Z3 when viewing the switch side from the switch side of the phase adjustment circuit,
The phase in the conjugate matching relationship of the phase θ of Z1 is θ1,
When 180 ° opposite phase to θ1 is θ0,
When the phase θ3 of Z3 is clockwise with respect to θ0,
The high-frequency module according to any one of claims 3 to 5, wherein a high-pass filter including an LC circuit is used as the phase adjustment circuit, and the phase θ2 of Z2 is adjusted so as to be closer to θ0 than θ3. 9. The method according to claim 8, wherein when a transmission line is used as the phase adjustment circuit, the Z2 phase is adjusted to be shorter than the current length of the transmission line so that the phase θ2 is closer to θ0 than θ3. The high-frequency module as described. 9. The high-frequency module according to claim 8, wherein one end of the inductor L of the high-pass filter is connected to the switch module unit side without a capacitor, and the other end is grounded. The low-pass filter or the high-pass filter is configured by a chip inductor and a chip capacitor, and the chip inductor and the chip capacitor are mounted on the laminated body or are external components mounted on a circuit board. The high-frequency module according to any one of 6, 8, and 10. The high-frequency module according to any one of claims 3 to 11, wherein a coupler circuit is interposed between the high-frequency amplifier circuit and the switch circuit, or a coupler circuit is configured instead of the switch circuit. A communication device comprising the high-frequency composite component according to claim 1 or 2 or the high-frequency module according to claim 3.

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