JP2004159383A - Motor controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor controller of rectangular waveform switching control capable of conducting quick switching in response to changes in rotational speed with a simple structure, preventing generation of an imbalanced phase state with a current detector, and operating power factor with both operations of driving and braking. <P>SOLUTION: This controller comprises an inverter 2 for mutually converting DC power and AC power and driving or controlling a motor 4b connected to AC side, a current detector 6b for detecting DC current of the inverter 2, and a control arithmetic operation unit 3 for switching the inverter into a rectangular waveform for control. The controller is structured so that switching the control arithmetic operation unit 3 is conducted. <P>COPYRIGHT: (C)2004,JPO

Description

【００２９】
上式より、スイッチング位相区分（６０度）毎にｉｄｃｎの位相は６０度ずつシフトする関係にあるため、ｉｄｃｎは各スイッチング位相区分毎に相似の波形となる。
この時、力率角ψを変えて力率を変化させた場合の各相線電流と直流側電流ｉｄｃｎの波形は図５のように示される。この図において、（ａ）は力率＝−１．０の場合、（ｂ）は同じく力率＝−０．９の場合、（ｃ）は力率＝−０．５の場合、（ｄ）は力率＝０．０の場合、（ｅ）は力率＝＋０．５の場合、（ｆ）は力率＝＋０．９の場合、（ｇ）は力率＝＋１．０の場合をそれぞれ示す。
スイッチング位相一周期で見ればｉｄｃｎは線電流基本波の６倍周波数でリップルを有する波形であり、その直流成分は力率＝−１．０の場合に最小、力率＝＋１．０の場合に最大、力率＝０．０の場合に０となる。また、リップル成分は力率＝−１．０、＋１．０の場合に最小、力率＝０．０の場合に最大となる。
【００３０】
ここでｉｄｃｎの直流成分ｉｄｃｎ_−ＤＣは、１つのスイッチング位相区分について平均して流れる電流を求める形として、次式のように算出される。ただしθ、ψの単位はラジアンを用いており、位相区分▲２▼を対象としている。
【００３１】
【数１】

【００３２】
また、リップル成分ｉｄｃｎ_−ｒｐｌはｉｄｃｎから直流成分ｉｄｃｎ_−ＤＣを減算したものとなる。図５に示されるようにリップル成分は力率によって大きく変化する。そこで、力率とリップル成分を関係付ける量として、スイッチング位相区分の半周期（３０度）の積分（Ａｒｐｌ）について考えると、その算出式は次式のようになる。ただしθ、ψの単位はラジアンを用いており、位相区分▲２▼を対象としている。
【００３３】
【数２】

【００３４】
図６は、力率＝−１．０と力率＝＋０．５の場合のリップル成分ｉｄｃｎ_−ｒｐｌとリップル３０度積分量Ａｒｐｌについて模式的に表した図である。力率＝ｃｏｓ（ψ）であるため、（６）式が示すように力率＝−１．０の場合はψ＝０度でありＡｒｐｌ＝０となる。
また、力率＝＋０．５の場合はψ＝−６０度であり、Ａｒｐｌ＝−（２−√３）／４×Ｉａ＝−０．０６６９９×Ｉａとなる。
【００３５】
次に、リップル３０度積分量Ａｒｐｌと直流成分ｉｄｃｎ_−ＤＣの比は次式のようになる。
【００３６】
【数３】

【００３７】
以上から、力率角ψはリップル３０度積分量Ａｒｐｌと直流成分ｉｄｃ ｎ_−ＤＣを用いて、次式により算出することができる。
【００３８】
【数４】

【００３９】
力率角ψに対する直流成分ｉｄｃｎ_−ＤＣ、リップル３０度積分量Ａｒｐｌ、リップル３０度積分量Ａｒｐｌと直流成分ｉｄｃｎ_−ＤＣの比の特性は、図７のように示される。
直流成分ｉｄｃｎ_−ＤＣはψの余弦（ＣＯＳ）関数、リップル３０度積分量Ａｒｐｌはψの正弦（ＳＩＮ）関数、リップル３０度積分量Ａｒｐｌと直流成分ｉｄｃｎ_−ＤＣの比はψに対する正接（ＴＡＮ）関数の特性を持つ。
また、線電流については、ｉｄｃｎを検出して（８）式により力率角ψを算出すれば、各スイッチング位相区分の範囲（６０度）内は、（４ａ）〜（４ｆ）式に応じた電流を観測できる。さらに線電流の振幅Ｉａは、（５）式の関係から力率角ψとｉｄｃｎの直流成分ｉｄｃｎ_−ＤＣにもとづき次式で求めることができる。
【００４０】
【数５】

【００４１】
この他、スイッチング位相区分の切り替り直後の直流側電流ｉｄｃｎをサンプリングするとｉｄｃｎ＝−Ｉａ・ｓｉｎ（ψ−１２０）であるため、線電流の振幅Ｉａは力率角ψとｉｄｃｎにもとづき次式で求めることもできる。
【００４２】
【数６】

【００４３】
さらには、リップル３０度積分の積分終了時点、すなわちスイッチング位相区分の中点の３０度時点で直流側電流ｉｄｃｎをサンプリングするとｉｄｃｎ＝−Ｉａ・ｓｉｎ（ψ−９０）＝Ｉａ・ｃｏｓψであるため、線電流の振幅Ｉａは力率角ψとｉｄｃｎにもとづき次式で求めることもできる。
【００４４】
【数７】

【００４５】
以上の基本原理にもとづき、力率角ψ、線電流振幅Ｉａが算出される。この原理を用いた図１の具体的な動作は次のようになる。
まず、図４の１８０度矩形波スイッチングにより電動機４ｂが回転し、線電流ｉｕ、ｉｖ、ｉｗが流れているとする。この時、直流側電流ｉｄｃｎが直流側電流検出器６ｂにより検出され、制御演算装置３へ取り込まれる。
【００４６】
図２は、制御演算装置３の詳細な構成を示すブロック図である。この図において、１０は低域通過フィルタ、１１は減算器、１２は微分手段、１３は不感帯演算器、１４は力率角算出手段、１５は線電流振幅算出手段、１６は力率角制御器、１７は過電流検知手段、１８はゲート信号生成手段である。
続いて、制御演算装置３の動作について説明する。直流側電流ｉｄｃｎは低域通過フィルタ１０へ入力され高周波成分を遮断して直流成分ｉｄｃｎ_−ＤＣが出力される。低域通過フィルタ１０の遮断周波数は電動機４ｂの駆動周波数帯の６倍周波数成分を遮断するような値に設定される。
【００４７】
次に、減算器１１にて直流側電流ｉｄｃｎから直流成分ｉｄｃｎ_−ＤＣが減算されてリップル成分ｉｄｃｎ_−ｒｐｌが出力される。また、微分手段１２では直流側電流ｉｄｃｎの２階の微分演算が行われる。直流側電流ｉｄｃｎはスイッチング位相区分毎に切り替る位相角６０度毎の相似波形であることから、スイッチング位相区分の切り替りに際してｉｄｃｎの変化率は大きく変化し、図８に示すように、２階の微分演算をするとスパイク状のパルスが発生する。
微分手段１２の出力は不感帯演算器１３により、直流側電流ｉｄｃｎの変化率の小さな部分は取り除かれて、スイッチング位相区分の切り替りで発生するスパイク状のパルスのみ通過し、位相区分切り替りパルスとなす。このスパイク状のパルスにより、スイッチング位相区分の切り替りを認識することができる。
さらに、力率角算出手段１４に直流成分ｉｄｃｎ_−ＤＣ、リップル成分ｉｄｃｎ_−ｒｐｌ、不感帯演算器１３からの位相区分切り替りパルス（ｉｄｃｎの２階微分信号）を入力すると、（５）〜（８）式にもとづく演算により力率角ψが算出される。
【００４８】
図９は、力率角算出手段１４の詳細な構成を示すブロック図である。この図において、２１は位相３０度タイミング生成手段、２２は切り替えスイッチ、２３は積分演算器、２４は除算器、２５は逆正接算出手段である。
【００４９】
続いて、力率角算出手段１４の動作について説明する。まず、位相区分切り替りパルスを位相３ ０度タイミング生成手段２１に入力すると、図１０のＴ３０−ｓｍｐで示されるような位相区分切り替りパルスに同期して立ち上がると共に、位相角３０度毎に立ち下りと立ち上がりを繰り返す矩形信号を出力する。切り替えスイッチ２２ は信号Ｔ３０−ｓｍｐが立ち上がるとスイッチをリップル成分ｉｄｃｎ_−ｒｐｌ側に切り替え、立ち下がると逆（信号レベルゼロ）側に切り替える。
積分演算器２３は切り替えスイッチ２２の出力信号を積分演算する。位相区分切り替り時点から位相角３０度の間はリップル成分ｉｄｃｎ_−ｒｐｌの値を積分し、信号Ｔ３０− ｓｍｐが立ち下がってからは信号レベルゼロを積分することとなるから、すなわち、（６）式のリップル３０度積分を行ってＡｒｐｌを算出することに相当する。
【００５０】
さらに、除算器２４により（７）式のリップル３０度積分量Ａｒｐｌ／直流成分ｉｄｃｎ_−ＤＣの演算が行なわれ、その出力は直流成分ｉｄｃｎ_−ＤＣと共に逆正接算出手段２５に入力される。逆正接算出手段２５では（８）式にもとづく逆正接演算により力率角ψが算出される。逆正接演算にはｔａｎの数値データと力率角ψとを対応付けた変換マップを用いると演算量が少なく適切である。また、力率角ψは直流成分ｉｄｃｎ_−ＤＣが正（プラス）の場合に２７０〜０〜９０度の範囲にあり、負（マイナス）の場合に９０〜１８０〜２７０度の範囲にある。また、リップル３０度積分量Ａｒｐｌ／直流成分ｉｄｃｎ_−ＤＣが正（プラス）の場合に０〜９０度あるいは１８０〜２７０度の範囲にあり、負（マイナス）の場合に９０〜１８０度あるいは２７０〜３６０度の範囲にあることから、この関係より、直流成分ｉｄｃｎ_−ＤＣとリップル３０度積分量Ａｒｐｌ／直流成分ｉｄｃｎ_−ＤＣとから力率角ψを算出することができる。また、逆正接算出手段２５は力率角ψの算出が完了すると、図１０のＴｒｓｔで示されるような形で矩形信号を立ち下げる。すなわちＴｒｓｔは各位相区分毎に、位相区分切り替りパルスに同期して立ち上がると共に力率角ψの算出完了時点で立ち下がる矩形信号である。
【００５１】
次に、力率角ψと力率角指令ψ^＊を力率角制御器１６に入力すると、公知の比例積分（ＰＩ）演算、あるいは比例（Ｐ）演算などによりψ^＊とψを一致させるようなスイッチング位相の補正量θ_{ｃｏｍｐ（ｋ）}を出力する。
また、線電流振幅算出手段１５に直流成分ｉｄｃｎ_−ＤＣ、力率角ψ、信号Ｔｒｓｔを入力すると、（９）式にもとづく演算により線電流振幅Ｉａが算出される。
【００５２】
ここで、図１１は線電流振幅算出手段１５の詳細な構成を示すブロック図である。この図において、３１はサンプリング指示発生器、３２はサンプラＡ、３３はサンプラＢ、３４は余弦算出手段、３５は除算器、３６は係数器である。
【００５３】
続いて、線電流振幅算出手段１５の動作について説明する。まず、信号Ｔｒｓｔをサンプリング指示発生器３１に入力するとＴｒｓｔが立ち下がるタイミング、すなわち、力率角ψの算出完了のタイミングでサンプラＡ３２、サンプラＢ３３に対してサンプリング指示信号を出力する。サンプラＡ３２はサンプリング指示信号に応じて直流成分ｉｄｃｎ_−ＤＣをサンプリングしてｉｄｃｎ_{−ＤＣ（ｋ）}となす。また、サンプラＢ３３はサンプリング指示信号に応じて力率角ψをサンプリングしてψ_（ｋ） となす。ψ_（ｋ） を余弦算出手段３４へ入力するとψ_（ｋ） の余弦関数値ｃｏｓ（ψ_（ｋ））を出力する。余弦演算には上述の逆正接演算の場合と同様に、力率角ψとｃｏｓψとを対応付けた変換マップを用いると演算量が少なく適切である。
【００５４】
次に、除算器３５、係数器３６により、（９）式にもとづく演算が行われる。すなわち、除算器３５にｉｄｃｎ_{−ＤＣ（ｋ）}とｃｏｓ（ψ_（ｋ））を入力するとｉｄｃｎ_{−ＤＣ（ｋ）}／ｃｏｓ（ψ_（ｋ））の演算が行なわれ、係数器３６にてπ／３倍して出力される。係数器３６の出力は線電流振幅Ｉａである。さらに、線電流振幅Ｉａは図２の過電流検知手段１７に入力される。過電流検知手段１７は予め設定される過電流振幅レベルＩａ_−ｏｃと線電流振幅Ｉａを突き合わせて比較し、過電流状態であると判断されると、電動機４ｂやパワーＭＯＳ−ＦＥＴ７ａ〜７ｆの損傷を予防するためゲート信号生成手段１８へ過電流検知信号を出力し、電流量を抑制するよう促す。
【００５５】
続いて、ゲート信号生成手段１８では、図４に示される１８０度矩形波スイッチングにより、ゲート信号Ｇｕｐ〜Ｇｗｎを生成して出力する。ゲート信号生成手段１８には不感帯演算器１３が出力する位相区分切り替りパルスと力率角制御器１６が出力するスイッチング位相の補正量θ_{ｃｏｍｐ（ｋ）}と線電流振幅算出手段１５が出力する線電流振幅Ｉａと低域通過フィルタ１０が出力する直流成分ｉｄｃ ｎ_−ＤＣと過電流検知手段１７が出力する過電流検知信号が入力される。
【００５６】
ここで、制御演算装置３が力率角指令ψ^＊に従って電動機を制御しようとする場合、そのスイッチングは図４のスイッチング位相角θに補正量θ_{ｃｏｍｐ（ｋ）}を加算した位相角に対して各スイッチング位相区分を算出し、ゲート信号を生成することとなる。この動作の一例を図１２にもとづき説明する。
【００５７】
図１２は、力率角指令ψ^＊を−１５４．２度（力率＝−０．９）から−１８０度（力率＝−１．０）へ変化させていった場合の線電流、スイッチング位相の補正量θ_{ｃｏｍｐ（ｋ）}を時間量に換算したＴ_{ｃｏｍｐ（ｋ）}、スイッチング位相区分の切り替えタイミングとスイッチング位相区分、及び、直流側電流ｉｄｃｎの変化の模様を示した図である。まず、波形の左端の区分▲３▼にて力率角算出手段１４により力率角ψが算出され、これを力率角指令ψ^＊と一致させるべく力率角制御器１６で補正量θ_{ｃｏｍｐ（ｋ）}が算出される。
その後、区分▲３▼から区分▲４▼へスイッチングが切り替るとともに前回区分での切り替りから切り替りまでの時間、すなわち、区分▲３▼の実所要時間ｐａ（ｋ）が測定される。この実所要時間ｐａ（ｋ）は区分▲４▼の標準予想所要時間ｐｅ（ｋ＋１）となる。
力率角算出手段１４内の位相３０度タイミング生成手段２１は、このｐａ（ｋ）に対して信号Ｔ３０−ｓｍｐを生成し、力率角算出のためのリップル３０度積分演算が行われる。
【００５８】
次に標準予想所要時間ｐｅ（ｋ＋１）に対して補正量θ_{ｃｏｍｐ（ｋ）}を時間量に換算したＴ_{ｃｏｍｐ（ｋ）}が加算され、実所要時間ｐａ（ｋ＋１）となる。すなわち、区分▲３▼から区分▲４▼への切り替りからｐａ（ｋ＋１）経過した時点で区分▲５▼のスイッチングへ切り替える。
図では補正量θ_{ｃｏｍｐ（ｋ）}は負（マイナス）の量、すなわち、θ_{ｃｏｍｐ（ｋ）}の時間換算量Ｔ_{ｃｏｍｐ（ｋ）}、は負（マイナス）の量であるためｐｅ（ｋ＋１）＞ｐａ（ｋ＋１）であり、補正量θ_{ｃｏｍｐ（ｋ）}に相当する時間分だけ早めにスイッチングを切り替えることとなる。また、同時に力率角ψがその時点での力率角指令ψ^＊に一致するように力率角制御器１６で補正量θ_{ｃｏｍｐ（ｋ＋１）}が算出される。この補正量θ_{ｃｏｍｐ（ｋ＋１）}は、次の区分▲５▼に切り替った際にスイッチングタイミングの補正量として反映される。以上のスイッチング位相区分毎の動作の繰り返しにより、力率角指令ψ^＊に従った電動機の制御が行なわれる。また、過電流検知手段１７からの過電流検知信号を認識した場合には、電流量を抑制するようスイッチングを行なう。
【００５９】
この他、制御演算装置３が直流成分の指令ｉｄｃｎ_−ＤＣ ^＊に従って電動機４ｂを制御しようとする場合、力率角制御器１６と同様な様態で、公知の比例積分（ＰＩ）演算、あるいは比例（Ｐ）演算などによりｉｄｃｎ_−ＤＣ ^＊とｉｄｃｎ_−ＤＣを一致させるようなスイッチング位相の補正量θ_{ｃｏｍｐ（ｋ）}を算出し、この補正量θ_{ｃｏｍｐ（ｋ）}を用いてスイッチング位相区分の切り替りタイミングを操作する形態とすることも可能である。
【００６０】
また、制御演算装置３が所定の線電流振幅指定値Ｉａ^＊となるよう電動機を制御しようとする場合、力率角制御器１６と同様な様態で、公知の比例積分（ＰＩ）演算や比例（Ｐ）演算によりスイッチング位相の補正量θ_{ｃｏｍｐ（ｋ）}を算出し、この補正量θ_{ｃｏｍｐ（ｋ）}を用いてスイッチング位相区分の切り替りタイミングを操作する形態とすることも可能である。あるいは、力率角指令ψ^＊や直流成分の指令ｉｄｃｎ_−ＤＣ ^＊に従って電動機４ｂを制御している際に、線電流振幅指定値Ｉａ^＊を超えないように電動機を制御しようとする場合、線電流振幅値Ｉａが線電流振幅指定値Ｉａ^＊を超えたか否かで補正量θ_{ｃｏｍｐ（ｋ）}を制限、解除して、スイッチング位相区分の切り替りタイミングを操作する形態とすることも可能である。
【００６１】
続いて、ゲート信号生成手段１８から出力されるゲート信号Ｇｕｐ、Ｇｕｎ、Ｇｖｐ、Ｇｖｎ、Ｇｗｐ、Ｇｗｎはゲート駆動回路８を経てパワーＭＯＳ−ＦＥＴ７ａ〜７ｆをスイッチングすることになる。パワーＭＯＳ−ＦＥＴ７ａ〜７ｆのスイッチングにより電動機４ｂの三相端子間に制御演算に応じた電圧が印加され、ひいては電動機４ｂが制御されることとなる。
【００６２】
以上の構成によれば、電流検出器６ｂを１つだけ用いる簡易な構成にて位相角６０度毎にスイッチングのタイミングを制御できることから、従来装置の位相角３６０度毎のスイッチングタイミングの制御に比較して、位相角（電気角）の変化率の変動、すなわち、回転速度の変動を素早くスイッチングに反映させることができる。なお、従来装置においても電流検出器を３つ用いて各線電流の極性反転を検出する構成とすれば、位相角６０度毎にスイッチングのタイミングを制御可能であるが大きなコストアップとなる。
【００６３】
さらに、電流検出器としてシャント抵抗を１つだけ用いる構成において、従来装置では電動機４ｂのうち、１つの相にのみ抵抗成分が挿入されることから、相不平衡状態となる恐れがあるが、この実施の形態による構成では直流側に電流検出器６ｂを設けるため、相不平衡状態とはならない。
【００６４】
また、力率が＋１．０あるいは−１．０となるよう電動機４ｂを制御する場合、この実施の形態の構成を簡略化することができる。すなわち、力率が＋１．０あるいは−１．０の場合は力率角が０度あるいは１８０度のため、（６）式よりリップル３０度積分量Ａｒｐｌはゼロとなる。このため、力率角算出手段１４内の除算器２４、逆正接算出手段２５を削除して、リップル３０度積分量Ａｒｐｌを出力する形態とすることができる。また、この時、信号Ｔ３０−ｓｍｐと信号Ｔｒｓｔを等しくＴ３０−ｓｍｐ＝Ｔｒｓｔとできる。力率角制御器１６はリップル３０度積分量Ａｒｐｌをゼロとすべく演算を行ない、スイッチング位相の補正量θ_{ｃｏｍｐ（ｋ）}を出力する。
【００６５】
ここで、上述のように力率を＋１．０あるいは−１．０となるよう電動機４ｂを制御すると、リップル成分ｉｄｃｎ_−ｒｐｌの振幅は最小となる。このリップル成分ｉｄｃｎ_−ｒｐｌは、大部分が平滑コンデンサ９に吸収され、また、吸収されるリップル電流に起因して平滑コンデンサ９は発熱する。発熱量はコンデンサの耐久性と相関があることから、所定の耐久性を確保するため、一般的に平滑コンデンサ９の個体当りに吸収されるリップル電流が所定値以下に収まるよう、平滑コンデンサ９の本数、容量が選択される。しかし、インバータ２を低容積、廉価に製造するために平滑コンデンサ９の本数、容量を減らすことが強く望まれる。このため、力率を＋１．０あるいは−１．０となるよう制御すればリップル成分ｉｄｃｎ_−ｒｐｌ自体を低減することができ、インバータ２を低容積、廉価に製造できるという効果がある。
【００６６】
なお、この実施の形態では図２、図９、図１１で示されるブロック構成にもとづいてその動作を説明したが、この発明はこれらブロック構成に何ら限定されるものではなく、この発明の根幹である力率の算出、並びに、線電流振幅の算出の基本原理にもとづく様態であればどのような構成でも良い。
【００６７】
【発明の効果】
この発明に係る電動機制御装置は、直流電力と交流電力を相互に変換し、交流側に接続された電動機を駆動または制動するインバータと、上記インバータの直流側電流を検出する電流検出器と、上記インバータを矩形波状にスイッチングして制御する制御演算装置を備え、上記電流検出器によって検出された上記インバータの直流側電流にもとづいて上記制御演算装置のスイッチングを行なうものであるため、簡易な構成で廉価な電動機制御装置を提供することができる。また、電動機の線電流ではなくインバータの直流側電流を検出することから、電流検出器としてシャント抵抗を用いる場合でも電動機の相インピーダンスが電流検出相と非検出相とで異なる相不平衡状態を回避することができる。
【００６８】
この発明に係る電動機制御装置は、また、直流電力と交流電力を相互に変換し、交流側に接続された電動機を駆動または制動するインバータと、上記インバータの直流側電流を検出する電流検出器と、上記インバータを矩形波状にスイッチングして制御する制御演算装置を備え、上記電流検出器によって検出された上記インバータの直流側電流にもとづいて力率を算出し、この力率にもとづいて上記制御演算装置のスイッチングを行なうものであるため、位相角６０度毎に力率角を算出してスイッチングのタイミングを制御できることから、位相角（電気角）の変化率の変動、すなわち、回転速度の変動を素早くスイッチングに反映させることができる。また、駆動と制動の両動作において任意の力率角で制御可能な矩形波状スイッチング制御方式の電動機制御装置を提供することができる。
【００６９】
この発明に係る電動機制御装置は、また、上記力率が＋１．０または−１．０となるように上記制御演算装置のスイッチングを行なうものであるため、インバータの直流側に流れる電流の変動成分（リップル）を最小に抑制することができる結果、平滑コンデンサの流入、流出電流（直流側電流のリップル成分）を抑制することができ、平滑コンデンサの耐久性の向上、平滑コンデンサの本数、容量の低減などを図ることができる。従って、さらに低容積で廉価なインバータを用いた電動機制御装置を提供することができる。
【００７０】
この発明に係る電動機制御装置は、また、直流電力と交流電力を相互に変換し、交流側に接続された電動機を駆動または制動するインバータと、上記インバータの直流側電流を検出する電流検出器と、上記インバータを矩形波状にスイッチングして制御する制御演算装置を備え、上記電流検出器によって検出された上記インバータの直流側電流にもとづいて上記電動機の線電流を算出し、この線電流にもとづいて上記制御演算装置のスイッチングを行なうものであるため、電動機の線電流量に応じた、例えば過電流検出、保護などを行なうことにより、信頼性の向上を図ることができる。
【図面の簡単な説明】
【図１】この発明の実施の形態１の全体構成を示すブロック図である。
【図２】実施の形態１における制御演算装置の具体的な構成を示すブロック図である。
【図３】実施の形態１の直流側電流検出に関する動作原理の説明図である。
【図４】実施の形態１の１８０度矩形波スイッチングによるゲートＯＮタイミングの説明図である。
【図５】実施の形態１において力率が変化した場合の線電流波形と直流側電流波形を示す特性図である。
【図６】実施の形態１におけるリップル成分ｉｄｃｎ_−ｒｐｌとリップル３０度積分量Ａｒｐｌを例示する説明図である。
【図７】実施の形態１における直流成分ｉｄｃｎ_−ＤＣとリップル３０度積分量Ａｒｐｌと比Ａｒｐｌ／ｉｄｃｎ_−ＤＣの特性を示す説明図である。
【図８】実施の形態１における直流側電流ｉｄｃｎとその１階微分、２階微分を例示する説明図である。
【図９】実施の形態１における力率角算出手段の具体的な構成を示すブロック図である。
【図１０】実施の形態１におけるリップル成分ｉｄｃｎ_−ｒｐｌと位相区分切り替りパルスとタイミング信号（Ｔ_{３０−ｓｍｐ}、Ｔ_ｒｓｔ）の動作説明図である。
【図１１】実施の形態１における線電流振幅算出手段の具体的な構成を示すブロック図である。
【図１２】実施の形態１における力率角制御の動作を例示する説明図である。
【図１３】従来装置の構成を示すブロック図である。
【図１４】従来装置におけるゲート信号動作の説明図である。
【符号の説明】
１ 電動機制御装置、 ２ インバータ、 ３ 制御演算装置、
４ｂ 電動機、 ５ 直流電源、 ６ｂ 直流側電流検出器、
７ａ〜７ｆ パワーＭＯＳ−ＦＥＴ、 ８ ゲート駆動回路、
９ 平滑コンデンサ、 １０ 低域通過フィルタ、
１１ 減算器、 １２ 微分手段、 １３ 不感帯演算器、
１４ 力率角算出手段、 １５ 線電流振幅算出手段、
１６ 力率角制御器、 １７ 過電流検知手段、
１８ ゲート信号生成手段。[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a motor control device, and more particularly to a motor control device having an inverter that switches in a rectangular wave shape.
[0002]
[Prior art]
FIG. 13 is a circuit diagram showing an example of a conventional motor control device for controlling a motor by switching a power semiconductor in a rectangular wave shape, which is applied to a rectifier of a vehicle charging generator functioning as a motor or a charging generator. It is.
In the figure, a motor control device 1 includes an inverter 2 for mutually converting DC power and AC power, a control operation device 3 for controlling the same, and a power M constituting the inverter based on a command from the control operation device. A gate drive circuit 8 that supplies a gate drive signal to an OS-FET (detailed later). A DC power supply 5 is connected to the DC input terminals P and N of the inverter 2, and a vehicle charging generator 4 a is connected to the AC output terminals U, V and W.
[0003]
The inverter 2 has a U-phase arm formed by connecting a pair of power MOS-FETs 7a and 7b in series, a V-phase arm formed by connecting power MOS-FETs 7c and 7d in series, and power MOS-FETs 7e and 7f connected in series. The smoothing capacitor 9 and the bridge circuit in which the W-phase arms are connected in parallel are connected to the DC input terminals P and N.
Further, of each phase arm, a shunt resistor 6a is connected between the connection point of the power MOS-FETs 7a and 7b of the U-phase arm and the AC output terminal U for detecting a line current. The connection points of the power MOS-FETs of the arm and the AC output terminals V and W are directly connected. The U-phase line current is detected as a potential difference between both ends of the line current detector 6a, and is taken into the control arithmetic unit 3.
[0004]
FIGS. 14A and 14B are explanatory diagrams of the gate signal operation of the conventional motor control device shown in FIG. 13. FIG. 14A shows the U-phase line current, and FIG. 14B shows the power supply to the U-phase upper arm power MOSFET. (C) is a gate signal also applied to the U-phase lower arm, (d) is a gate signal applied to the V-phase upper arm, (e) is a gate signal applied to the V-phase lower arm, (f) Indicates a gate signal applied to the W-phase upper arm, and (g) indicates a gate signal applied to the W-phase lower arm.
In this conventional device, the U-phase line current in the armature winding of the vehicle charging generator 4a is detected by the line current detector 6a, the polarity of this line current is determined, and the polarity is reversed (so-called zero crossing). If the line current polarity is negative between the next time and when the polarity is reversed, among the power MOS-FETs constituting each phase arm, the upper arm MOS-FET is switched on and the lower arm MOS-FET is turned on. -Turn off the FET.
When the line current polarity is positive, the upper arm MOS-FET is turned off and the lower arm MOS-FET is turned on.
[0005]
Further, for the V and W phases in which the line current is not detected, the time from when the polarity of the line current of the detected phase is inverted to the next time is measured, and this time is divided by the division ratio according to the number of phases. In addition to the above, the MOS-FETs of the upper and lower arms are switched at a timing appropriately delayed from the switching timing of the detection phase using the divided time, similarly to the line current detection phase.
The control arithmetic unit 3 determines the polarity of the U-phase line current, measures the time from when the polarity is inverted to when the polarity is next inverted, and divides this by a dividing ratio of 2/3 and 4/3. Generate delay timing. In FIG. 14, a time interval TA from the polarity inversion of the U-phase line current waveform from minus to plus to the next polarity inversion from plus to minus is measured, and TA × 2/3 and TA × 4/3 are calculated. Generate delay timing. Further, a time interval TB from the polarity inversion from plus to minus to the next polarity inversion from minus to plus is measured, and delay timings of TB × 2/3 and TB × 4/3 are generated.
[0006]
Further, as shown in FIG. 14B, the control arithmetic unit 3 switches the U-phase upper arm MOS-FET 7a at a timing after a delay time t1 from the time when the time interval TA is measured, to switch ON the gate signal Gup. Turn ON. At the time when the time interval TB is measured, the gate signal Gup is turned off so as to switch off the U-phase upper arm MOS-FET 7a. Further, as shown in FIG. 14C, when the time interval TA is measured, the gate signal Gun is turned off so as to switch off the U-phase lower arm MOS-FET 7b. Further, the gate signal Gun is turned ON so as to switch ON the U-phase lower arm MOS-FET 7b at a timing after a delay time t2 from the time when the time interval TB is measured.
[0007]
As shown in FIG. 14 (d), the gate signal Gvp of the V-phase upper arm is switch ON with a time delay of TA × ２ from the gate signal Gup of the U-phase upper arm, and a time delay of TB × 2/3. Then the switch is turned off. As shown in FIG. 14E, the gate signal Gvn of the V-phase lower arm is switched off with a time delay of TA × ２ from the gate signal Gun of the U-phase lower arm, and the time delay of TB × 2/3. Then switch on.
[0008]
Similarly, as shown in FIG. 14 (f), the gate signal Gwp of the upper arm of the W-phase is switched on with a time delay of TA × 4/3 from the gate signal Gup of the upper arm of the U-phase, and the switch is set to TB × 4/3. The switch is turned off with a time delay of. As shown in FIG. 14 (g), the gate signal Gwn of the lower arm of the W-phase is switched off with a time delay of TA × 4/3 from the gate signal Gun of the lower arm of the U-phase, and the time delay of TB × 4/3. Then switch on. The gate signals Gup, Gun, Gvp, Gvn, Gwp, Gwn are respectively supplied to the power MOS-FETs 7a to 7f via the gate drive circuit 8, and are switched as described above.
[0009]
This switching operation is generally called 180-degree rectangular wave switching. In the conventional apparatus, in the case of the minimum configuration, the switching of the power MOS-FET of the other phase is controlled by detecting the line current of one phase, so that the configuration is simple and inexpensive.
Further, since a power MOS-FET is used instead of a diode as a rectifying element, there is a characteristic that heat loss can be suppressed. (See, for example, Patent Document 1).
[0010]
[Patent Document 1]
JP-A-2002-171678 (paragraphs 0011, 0013, 0015, FIG. 1)
[0011]
[Problems to be solved by the invention]
However, in the conventional device, the switching of the power MOS-FET of the other phase is controlled based on the time interval of the inversion of the polarity of the line current of a certain phase (electrical angle of 180 degrees). When the interval fluctuates, there is a problem that the fluctuation of the time interval is delayed with respect to the switching of the phase in which the line current is not detected. Further, when the current detector is constituted by a shunt resistor, the impedance of the detection phase of the line current and the impedance of the non-detection phase are different, so that there is a problem that a phase imbalance occurs.
[0012]
SUMMARY OF THE INVENTION The present invention has been made to solve the above problems, and the change in the rate of change of the electrical angle, that is, the change in the rotation speed is quickly reflected in the switching with a simple configuration, and the current detection To provide a motor control device of a rectangular-wave switching control system capable of preventing a phase imbalance even when a shunt resistor is used and controlling a power factor in both driving and braking operations. Aim.
[0013]
[Means for Solving the Problems]
The motor control device according to the present invention includes an inverter that converts DC power and AC power to each other and drives or brakes a motor connected to an AC side, a current detector that detects a DC side current of the inverter, A control operation device for switching and controlling the inverter in a rectangular waveform is provided, and the control operation device is switched based on the DC current of the inverter detected by the current detector. By doing so, a phase imbalance state can be prevented.
[0014]
The motor control device according to the present invention also includes an inverter that converts DC power and AC power to each other and drives or brakes a motor connected to the AC side, and a current detector that detects a DC side current of the inverter. A control operation device that controls the inverter by switching the inverter in a rectangular waveform, calculates a power factor based on the DC current of the inverter detected by the current detector, and performs the control operation based on the power factor. This is for switching the device. Since the calculation of the power factor angle can be performed for each phase angle of 60 degrees, even when the change rate of the electric angle fluctuates, it can be promptly reflected in the switching.
[0015]
The electric motor control device according to the present invention switches the control arithmetic unit so that the power factor becomes +1.0 or -1.0.
By controlling the switching so that the power factor is +1.0 when the motor is driven and -1.0 when braking, the fluctuation component (ripple) of the current flowing on the DC side of the inverter is minimized. Can be suppressed. As a result, it is possible to suppress the inflow and outflow current (the fluctuation component of the DC side current) flowing into and out of the smoothing capacitor arranged for the purpose of suppressing the fluctuation of the DC side voltage of the inverter.
[0016]
The motor control device according to the present invention also includes an inverter that converts DC power and AC power to each other and drives or brakes a motor connected to the AC side, and a current detector that detects a DC side current of the inverter. A control operation device that controls the inverter by switching the inverter in a rectangular waveform, and calculates a line current of the motor based on a DC side current of the inverter detected by the current detector, and calculates a line current based on the line current. The switching of the control arithmetic unit is performed. For this reason, for example, overcurrent detection and protection can be performed according to the line current amount of the motor.
[0017]
BEST MODE FOR CARRYING OUT THE INVENTION
Embodiment 1 FIG.
Hereinafter, a first embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram illustrating the overall configuration of the motor control device according to the first embodiment.
In the figure, a motor control device 1 includes an inverter 2 for mutually converting DC power and AC power, a control operation device 3 for controlling the same, and a power MOS constituting the inverter based on a command from the control operation device. And a gate drive circuit 8 that supplies a gate drive signal to the FET. A DC power supply 5 is connected to DC input terminals P and N of the inverter 2, and a motor 4 b is connected to AC output terminals U, V and W.
[0018]
The inverter 2 has a U-phase arm formed by connecting a pair of power MOS-FETs 7a and 7b in series, a V-phase arm formed by connecting power MOS-FETs 7c and 7d in series, and power MOS-FETs 7e and 7f connected in series. The bridge circuit in which the W-phase arms are connected in parallel and the smoothing capacitor 9 are connected to DC input terminals P and N, and each phase arm is connected to AC output terminals U, V and W. Further, a DC side current detector 6b is connected to the DC side of the bridge circuit.
[0019]
In FIG. 1, an inverter 2 switches DC power and AC power mutually by switching a total of six power MOS-FETs, which are semiconductor power conversion elements constituting each phase arm, to drive or brake the electric motor 4b. . The power MOS-FETs 7a to 7f of the inverter 2 perform a switching operation in a rectangular waveform based on a gate signal output from the gate drive circuit 8 in response to a command from the control operation device 3.
[0020]
Prior to the description of the specific operation in FIG. 1, the basic principle of the calculation of the power factor angle and the calculation of the line current amplitude, which are the basis of the present invention, will be described.
FIG. 3 shows a current idcn detected by the DC current detector 6b arranged at the N-side potential of the DC power supply 5, a current idcn-u flowing through the power MOS-FET 7b, and a current idcn-v flowing through the power MOS-FET 7d. FIG. 9 is a diagram schematically illustrating a relationship between a current idcn-w flowing through a power MOS-FET 7f and a line current flowing through a motor 4b. In the figure, idcn-u, idcn-v, and idcn-w have a positive (positive) direction flowing into the electric motor 4b from the N potential side.
The direction of the DC side current idcn flowing into the smoothing capacitor 9 from the power MOS-FET is defined as plus (positive). That is, the relationship between idcn and idcn-u, idcn-v, idcn-w is as follows.
[0021]
idcn =-(idcn_−u+ Idcn_-V+ Idcn_-W(1)
[0022]
FIG. 4 is a diagram showing the gate ON timing of each of the power MOS-FETs 7a to 7f by 180-degree rectangular wave switching. In the 180-degree rectangular wave switching, the upper arm power MOS-FET is turned on over a section of 180 degrees in one electrical angle cycle, and the lower arm power MOS-FET is turned on over the remaining 180 degrees. The U-phase, V-phase, and W-phase switching are performed with a phase difference of 120 degrees from each other. Therefore, switching of the switching occurs every 60 degrees.
One cycle of the switching phase angle is (1) 0 to 60 degrees, (2) 60 to 120 degrees, (3) 120 to 180 degrees, (4) 180 to 240 degrees, (5) 240 to 300 degrees, (6) 300 When the lower arm is divided at intervals of 60 degrees, for example, the lower arm is switched on in each of the following phases: the division (1) is the V phase, the division (2) is the V phase and the W phase, and the division (3) is the W phase. Phase, section (4) is U phase and W phase, section (5) is U phase, and section (6) is U phase and V phase.
[0023]
Here, since the lower-arm power MOS-FET and the upper-arm power MOS-FET are switched on in a complementary relationship, what is detected as the DC side current idcn in each switching phase section is the lower-arm power MOS-FET. This is the reverse sign value of the sum of the line currents of the phases of the FETs that are switched ON. That is, the DC side current idcn in each switching phase section is represented by the following equation. However, the line currents flowing through the motor 4b are represented by iu, iv, iw, respectively, in the U, V, and W phases.
[0024]
Classification (1) idcn = -idcn-v = -iv (2a)
Classification {circle around (2)} idcn = − (idcn−v + idcn−w) = − iv−iw (2b)
Classification (3) idcn = -idcn-w = -iw (2c)
Classification (4) idcn =-(idcn-w + idcn-u) =-iw-iu (2d)
Classification (5) idcn = -idcn-u = -iu (2e)
Classification (6) idcn =-(idcn-u + idcn-v) = -iu-iv (2f)
[0025]
Here, the line current is defined as the following equation, where the phase difference between the switching phase and the three-phase current, that is, the power factor angle is ψ, and the amplitude is Ia.
[0026]
iu = Ia · sin (θ + ψ) (3a)
iv = Ia · sin (θ + ψ−120) (3b)
iw = Ia · sin (θ + ψ−240) (3c)
[0027]
At this time, the DC side current idcn in each switching phase section is as follows.
[0028]

[0029]
From the above equation, since the phase of idcn shifts by 60 degrees for each switching phase section (60 degrees), idcn has a similar waveform for each switching phase section.
At this time, the waveforms of each phase line current and the DC side current idcn when the power factor is changed by changing the power factor angle ψ are shown in FIG. In this figure, (a) is a case where the power factor is -1.0, (b) is a case where the power factor is -0.9, (c) is a case where the power factor is -0.5, (d) Is the case where the power factor is 0.0, (e) is the case where the power factor is +0.5, (f) is the case where the power factor is +0.9, and (g) is the case where the power factor is +1.0. Show.
When viewed in one cycle of the switching phase, idcn is a waveform having a ripple at six times the frequency of the line current fundamental wave, and its DC component is minimum when the power factor is -1.0, and is small when the power factor is +1.0. The maximum is 0 when the power factor is 0.0. The ripple component is minimum when the power factor is -1.0 and +1.0, and is maximum when the power factor is 0.0.
[0030]
Where the DC component idcn of idcn_−DCIs calculated as the following equation as a form for obtaining the current flowing on average for one switching phase section. However, the units of θ and ψ are in radians, and are intended for phase division (2).
[0031]
(Equation 1)

[0032]
Also, the ripple component idcn_-RplIs the DC component idcn from idcn_−DCIs subtracted. As shown in FIG. 5, the ripple component changes greatly depending on the power factor. Considering the integration (Arpl) of the half cycle (30 degrees) of the switching phase division as an amount relating the power factor and the ripple component, the calculation formula is as follows. However, the units of θ and ψ are in radians, and are intended for phase division (2).
[0033]
(Equation 2)

[0034]
FIG. 6 shows the ripple component idcn when the power factor = −1.0 and the power factor = + 0.5._-RplFIG. 7 is a diagram schematically illustrating a ripple 30-degree integration amount Arpl. Since the power factor = cos (ψ), as shown by the equation (6), when the power factor = −1.0, ψ = 0 degree and Arpl = 0.
When the power factor is +0.5, ψ = −60 degrees, and Arpl = − (2−3) /4×Ia=−0.06699×Ia.
[0035]
Next, the ripple 30-degree integration amount Arpl and the DC component idcn_−DCIs given by the following equation.
[0036]
(Equation 3)

[0037]
From the above, the power factor angle ψ is calculated by integrating the ripple 30 degree integration amount Arpl and the DC component idcn._−DCAnd can be calculated by the following equation.
[0038]
(Equation 4)

[0039]
DC component idcn for power factor angle ψ_−DC, Ripple 30-degree integration amount Arpl, ripple 30-degree integration amount Arpl, and DC component idcn_−DCThe characteristic of the ratio is shown in FIG.
DC component idcn_−DCIs the cosine (COS) function of ψ, the ripple 30-degree integration amount Arpl is the sine (SIN) function of ψ, the ripple 30-degree integration amount Arpl and the DC component idcn_−DCHas the property of a tangent to T (関 数) function.
For the line current, if idcn is detected and the power factor angle ψ is calculated by the equation (8), the range (60 degrees) of each switching phase section corresponds to the equations (4a) to (4f). Current can be observed. Further, the amplitude Ia of the line current is represented by the power factor angle か ら and the DC component_−DCIt can be obtained by the following equation based on
[0040]
(Equation 5)

[0041]
In addition, when the DC side current idcn immediately after switching of the switching phase section is sampled, idcn = −Ia · sin (ψ−120). Therefore, the amplitude Ia of the line current is expressed by the following equation based on the power factor angles ψ and idcn. You can also ask.
[0042]
(Equation 6)

[0043]
Furthermore, when the DC side current idcn is sampled at the end of the integration of the ripple 30-degree integration, that is, at the 30-degree midpoint of the switching phase division, idcn = −Ia · sin ({− 90) = Ia · cos}. The amplitude Ia of the line current can also be obtained by the following equation based on the power factor angle ψ and idcn.
[0044]
(Equation 7)

[0045]
The power factor angle 原理 and the line current amplitude Ia are calculated based on the above basic principles. The specific operation of FIG. 1 using this principle is as follows.
First, it is assumed that the electric motor 4b is rotated by the 180-degree rectangular wave switching shown in FIG. 4 and line currents iu, iv, and iw are flowing. At this time, the DC side current idcn is detected by the DC side current detector 6b and is taken into the control arithmetic unit 3.
[0046]
FIG. 2 is a block diagram showing a detailed configuration of the control arithmetic unit 3. In this figure, 10 is a low-pass filter, 11 is a subtractor, 12 is a differentiator, 13 is a dead zone calculator, 14 is a power factor angle calculator, 15 is a line current amplitude calculator, 16 is a power factor angle controller. , 17 are overcurrent detecting means, and 18 is a gate signal generating means.
Next, the operation of the control arithmetic unit 3 will be described. The DC side current idcn is input to the low-pass filter 10 to cut off the high frequency component and to cut off the DC component idcn._−DCIs output. The cutoff frequency of the low-pass filter 10 is set to a value that cuts off a frequency component six times the driving frequency band of the electric motor 4b.
[0047]
Next, the DC component idcn is subtracted from the DC side current idcn by the subtractor 11._−DCIs subtracted to obtain a ripple component idcn._-RplIs output. Further, the differentiating means 12 performs a second-order differential operation of the DC side current idcn. Since the DC side current idcn has a similar waveform at every 60 degrees of the phase angle which switches for each switching phase section, the rate of change of idcn greatly changes when the switching phase section switches, as shown in FIG. , A spike-like pulse is generated.
The output of the differentiating means 12 has a small change rate of the DC side current idcn removed by the dead zone calculator 13 and only the spike-like pulse generated by switching of the switching phase section passes, and the phase section switching pulse is output. Eggplant The switching of the switching phase division can be recognized by the spike-like pulse.
Further, the DC factor idcn is supplied to the power factor angle calculating means 14._−DC, The ripple component idcn_-RplWhen the phase division switching pulse (second differential signal of idcn) from the dead zone calculator 13 is input, the power factor angle ψ is calculated by the calculation based on the equations (5) to (8).
[0048]
FIG. 9 is a block diagram showing a detailed configuration of the power factor angle calculating means 14. In this figure, 21 is a phase 30-degree timing generation means, 22 is a changeover switch, 23 is an integration calculator, 24 is a divider, and 25 is an arc tangent calculation means.
[0049]
Next, the operation of the power factor angle calculating means 14 will be described. First, when the phase division switching pulse is input to the phase 30-degree timing generation means 21, it rises in synchronization with the phase division switching pulse as shown by T30-smp in FIG. A rectangular signal that repeats falling and rising is output. The changeover switch 22 sets the switch to the ripple component idcn when the signal T30-smp rises._-RplSide, and when it falls, it switches to the opposite side (signal level zero).
The integration calculator 23 performs an integration operation on the output signal of the changeover switch 22. A ripple component idcn during a phase angle of 30 degrees from the phase section switching time._-RplIs integrated and the signal level zero is integrated after the signal T30-smp falls, that is, it is equivalent to calculating Rpl by performing the ripple 30-degree integration of Expression (6).
[0050]
Further, the divider 24 calculates the ripple 30-degree integral amount Arpl / DC component idcn of the equation (7)._−DCIs calculated, and the output is a DC component idcn_−DCIs input to the arc tangent calculating means 25 together with the tangent. The arc tangent calculating means 25 calculates the power factor angle に よ り by arc tangent calculation based on the equation (8). It is appropriate to use a conversion map in which the numerical data of tan and the power factor angle 対 応 付 け are associated with each other for the arc tangent calculation because the calculation amount is small. The power factor angle ψ is a DC component idcn_−DCIs in the range of 270 to 90 degrees when positive (plus), and in the range of 90 to 180 to 270 degrees when negative (minus). Further, the ripple 30-degree integration amount Arpl / DC component idcn_−DCIs positive (plus) in the range of 0 to 90 degrees or 180 to 270 degrees, and negative (minus) in the range of 90 to 180 degrees or 270 to 360 degrees. Component idcn_−DCAnd ripple 30 degree integration amount Arpl / DC component idcn_−DCFrom this, the power factor angle ψ can be calculated. When the calculation of the power factor angle ψ is completed, the arc tangent calculating means 25 causes the rectangular signal to fall in the form as shown by Trst in FIG. That is, Trst is a rectangular signal that rises in synchronization with the phase section switching pulse and falls at the time of completion of calculation of the power factor angle ψ for each phase section.
[0051]
Next, the power factor angle ψ and the power factor angle command ψ^*Is input to the power factor angle controller 16 and a known proportional integration (PI) calculation or a proportional (P) calculation is performed.^*The amount of switching phase correction θ to make_{comp (k)}Is output.
Also, the DC component idcn is supplied to the line current amplitude calculating means 15._−DC, The power factor angle ψ, and the signal Trst, the line current amplitude Ia is calculated by calculation based on the equation (9).
[0052]
Here, FIG. 11 is a block diagram showing a detailed configuration of the line current amplitude calculating means 15. In this figure, 31 is a sampling instruction generator, 32 is a sampler A, 33 is a sampler B, 34 is a cosine calculating means, 35 is a divider, and 36 is a coefficient unit.
[0053]
Subsequently, the operation of the line current amplitude calculation means 15 will be described. First, when the signal Trst is input to the sampling instruction generator 31, a sampling instruction signal is output to the samplers A32 and B33 at the timing when Trst falls, that is, when the calculation of the power factor angle 率 is completed. The sampler A32 receives the DC component idcn in response to the sampling instruction signal._−DCIs sampled and idcn_{−DC (k)}And The sampler B33 samples the power factor angle ψ according to the sampling instruction signal, and ψ_(K) And ψ_(K) Is input to the cosine calculation means 34._(K) Cosine function value cos (ψ_(K)) Is output. As in the case of the arc tangent operation described above, it is appropriate to use a conversion map in which the power factor angles ψ and cos 対 応 付 け are associated with each other, since the calculation amount is small and appropriate.
[0054]
Next, the divider 35 and the coefficient unit 36 perform an operation based on the equation (9). That is, idcn is added to the divider 35._{−DC (k)}And cos (ψ_(K)) And idcn_{−DC (k)}/ Cos (ψ_(K)) Is performed, and is multiplied by π / 3 in the coefficient unit 36 and output. The output of the coefficient unit 36 is the line current amplitude Ia. Further, the line current amplitude Ia is input to the overcurrent detecting means 17 in FIG. The overcurrent detecting means 17 has a preset overcurrent amplitude level Ia._-OcAnd the line current amplitude Ia is compared with each other, and when it is determined that the current is in an overcurrent state, an overcurrent detection signal is output to the gate signal generation means 18 to prevent damage to the motor 4b and the power MOS-FETs 7a to 7f. Urge the user to reduce the amount of current.
[0055]
Subsequently, the gate signal generation means 18 generates and outputs gate signals Gup to Gwn by 180-degree rectangular wave switching shown in FIG. The gate signal generating means 18 has a phase section switching pulse output from the dead zone calculator 13 and a correction amount θ of the switching phase output from the power factor angle controller 16._{comp (k)}And the line current amplitude Ia output by the line current amplitude calculation means 15 and the DC component idcn output by the low-pass filter 10_−DCAnd an overcurrent detection signal output from the overcurrent detection means 17 are input.
[0056]
Here, the control arithmetic unit 3 outputs the power factor angle command ψ^*When the motor is to be controlled in accordance with the following equation, the switching is performed by adding the correction amount θ to the switching phase angle θ in FIG._{comp (k)}Is calculated for each switching phase section with respect to the phase angle obtained by adding the above, and a gate signal is generated. An example of this operation will be described with reference to FIG.
[0057]
FIG. 12 shows a power factor angle command ψ^*Is changed from −154.2 degrees (power factor = −0.9) to −180 degrees (power factor = −1.0), the correction amount θ of the line current and the switching phase._{comp (k)}T converted to the amount of time_{comp (k)}FIG. 9 is a diagram showing switching timings of switching phase sections, switching phase sections, and patterns of changes in DC side current idcn. First, a power factor angle ψ is calculated by the power factor angle calculation means 14 in the section (3) at the left end of the waveform, and the power factor angle command ψ^*Correction amount θ by the power factor angle controller 16 so that_{comp (k)}Is calculated.
Thereafter, switching is performed from the section (3) to the section (4), and the time from the previous section to the next section, that is, the actual required time pa (k) of the section (3) is measured. The actual required time pa (k) is the standard expected required time pe (k + 1) of the category (4).
The phase 30-degree timing generation unit 21 in the power factor angle calculation unit 14 generates a signal T30-smp for this pa (k), and performs a ripple 30-degree integration calculation for calculating the power factor angle.
[0058]
Next, the correction amount θ with respect to the standard expected required time pe (k + 1)_{comp (k)}T converted to the amount of time_{comp (k)}Is added to the actual required time pa (k + 1). That is, when pa (k + 1) elapses after the switching from the section (3) to the section (4), the switching to the section (5) is performed.
In the figure, the correction amount θ_{comp (k)}Is a negative (minus) quantity, that is, θ_{comp (k)}Time conversion amount T_{comp (k)}, Is a negative (minus) amount, so that pe (k + 1)> pa (k + 1), and the correction amount θ_{comp (k)}The switching is to be switched earlier by the time corresponding to. At the same time, the power factor angle ψ is^*Is corrected by the power factor angle controller 16 so that_{comp (k + 1)}Is calculated. This correction amount θ_{comp (k + 1)}Is reflected as the correction amount of the switching timing when switching to the next section (5). By repeating the above operation for each switching phase section, the power factor angle command ψ^*Control of the electric motor according to the above. When the overcurrent detection signal from the overcurrent detection means 17 is recognized, switching is performed so as to suppress the amount of current.
[0059]
In addition, the control operation device 3 executes the DC component command idcn._−DC ^*When the motor 4b is controlled in accordance with the following equation, idcn is calculated by a known proportional integration (PI) calculation or a proportional (P) calculation in the same manner as the power factor angle controller 16._−DC ^*And idcn_−DCThe correction amount θ of the switching phase so that_{comp (k)}And the correction amount θ_{comp (k)}It is also possible to adopt a mode in which the switching timing of the switching phase section is operated by using.
[0060]
Further, the control arithmetic unit 3 determines that the predetermined line current amplitude designated value Ia^*When the motor is controlled so as to satisfy the following condition, in the same manner as the power factor angle controller 16, the correction amount θ of the switching phase is calculated by a known proportional integration (PI) calculation or proportional (P) calculation._{comp (k)}And the correction amount θ_{comp (k)}It is also possible to adopt a mode in which the switching timing of the switching phase section is operated by using. Alternatively, the power factor angle command ψ^*Or DC component command idcn_−DC ^*When the motor 4b is controlled in accordance with^*When the motor is controlled so as not to exceed the line current amplitude value Ia, the line current amplitude value Ia^*Correction amount θ depending on whether or not_{comp (k)}May be limited or released to operate the switching timing of the switching phase division.
[0061]
Subsequently, the gate signals Gup, Gun, Gvp, Gvn, Gwp, Gwn output from the gate signal generating means 18 switch the power MOS-FETs 7a to 7f via the gate drive circuit 8. By the switching of the power MOS-FETs 7a to 7f, a voltage corresponding to the control calculation is applied between the three-phase terminals of the electric motor 4b, and the electric motor 4b is controlled.
[0062]
According to the above configuration, the switching timing can be controlled for each phase angle of 60 degrees with a simple configuration using only one current detector 6b. Thus, the change in the change rate of the phase angle (electrical angle), that is, the change in the rotation speed can be promptly reflected in the switching. In the conventional device, if the polarity inversion of each line current is detected by using three current detectors, the switching timing can be controlled at every 60 degrees of the phase angle, but the cost is greatly increased.
[0063]
Furthermore, in a configuration using only one shunt resistor as a current detector, in the conventional device, a resistance component is inserted into only one phase of the electric motor 4b, so that there is a possibility that a phase unbalanced state occurs. In the configuration according to the embodiment, since the current detector 6b is provided on the DC side, no phase imbalance occurs.
[0064]
When the electric motor 4b is controlled so that the power factor becomes +1.0 or -1.0, the configuration of the present embodiment can be simplified. That is, when the power factor is +1.0 or -1.0, the power factor angle is 0 or 180 degrees, and the ripple 30-degree integration amount Arpl is zero according to the equation (6). For this reason, the divider 24 and the arctangent calculating means 25 in the power factor angle calculating means 14 can be omitted to output the ripple 30-degree integral amount Arpl. Also, at this time, the signal T30-smp and the signal Trst can be made equal, and T30-smp = Trst. The power factor angle controller 16 performs an operation to reduce the ripple 30-degree integration amount Arpl to zero, and corrects the switching phase θ_{comp (k)}Is output.
[0065]
Here, when the electric motor 4b is controlled so that the power factor becomes +1.0 or -1.0 as described above, the ripple component idcn_-RplHas the minimum amplitude. This ripple component idcn_-RplAre mostly absorbed by the smoothing capacitor 9 and the smoothing capacitor 9 generates heat due to the absorbed ripple current. Since the calorific value correlates with the durability of the capacitor, in order to secure a predetermined durability, generally, the smoothing capacitor 9 is controlled so that the ripple current absorbed per individual smoothing capacitor 9 falls below a predetermined value. The number and capacity are selected. However, it is strongly desired to reduce the number and capacity of the smoothing capacitors 9 in order to manufacture the inverter 2 with low volume and low cost. Therefore, if the power factor is controlled to be +1.0 or -1.0, the ripple component idcn_-RplTherefore, the inverter 2 can be manufactured at a low volume and at a low cost.
[0066]
In this embodiment, the operation has been described based on the block configurations shown in FIG. 2, FIG. 9, and FIG. 11, but the present invention is not limited to these block configurations at all, and is the basis of the present invention. Any configuration may be used as long as it is based on the basic principle of calculating a certain power factor and calculating the line current amplitude.
[0067]
【The invention's effect】
The motor control device according to the present invention includes an inverter that converts DC power and AC power to each other and drives or brakes a motor connected to the AC side, a current detector that detects a DC side current of the inverter, A control operation device that controls the inverter by switching it in a rectangular waveform is provided, and the control operation device is switched based on the DC current of the inverter detected by the current detector. An inexpensive motor control device can be provided. In addition, since the DC side current of the inverter is detected instead of the line current of the motor, even when a shunt resistor is used as the current detector, a phase imbalance state where the phase impedance of the motor differs between the current detection phase and the non-detection phase is avoided. can do.
[0068]
The motor control device according to the present invention also includes an inverter that converts DC power and AC power to each other, drives or brakes a motor connected to the AC side, and a current detector that detects a DC side current of the inverter. A control operation device that controls the inverter by switching the inverter in a rectangular wave shape, calculates a power factor based on the DC current of the inverter detected by the current detector, and performs the control operation based on the power factor. Since the switching of the device is performed, the power factor angle can be calculated for each phase angle of 60 degrees and the switching timing can be controlled. It can be reflected in switching quickly. Further, it is possible to provide a motor control device of a rectangular wave switching control system which can be controlled at an arbitrary power factor angle in both driving and braking operations.
[0069]
Further, the motor control device according to the present invention switches the control operation device so that the power factor becomes +1.0 or -1.0. (Ripple) can be suppressed to a minimum, so that the inflow and outflow current (ripple component of the DC current) of the smoothing capacitor can be suppressed, the durability of the smoothing capacitor can be improved, and the number and capacity of the smoothing capacitor can be reduced. Reduction can be achieved. Therefore, it is possible to provide a motor control device using an inexpensive inverter with a smaller volume.
[0070]
The motor control device according to the present invention also includes an inverter that converts DC power and AC power to each other, drives or brakes a motor connected to the AC side, and a current detector that detects a DC side current of the inverter. A control operation device that controls the inverter by switching the inverter in a rectangular wave shape, and calculates a line current of the electric motor based on a DC side current of the inverter detected by the current detector, based on the line current. Since the switching of the control arithmetic unit is performed, reliability can be improved by performing, for example, overcurrent detection and protection in accordance with the line current amount of the electric motor.
[Brief description of the drawings]
FIG. 1 is a block diagram showing an overall configuration of a first embodiment of the present invention.
FIG. 2 is a block diagram illustrating a specific configuration of a control operation device according to the first embodiment.
FIG. 3 is an explanatory diagram of an operation principle relating to DC side current detection according to the first embodiment.
FIG. 4 is an explanatory diagram of gate ON timing by 180-degree rectangular wave switching according to the first embodiment.
FIG. 5 is a characteristic diagram showing a line current waveform and a DC current waveform when the power factor changes in the first embodiment.
FIG. 6 shows a ripple component idcn in the first embodiment._-RplFIG. 5 is an explanatory diagram illustrating a ripple 30-degree integration amount Arpl.
FIG. 7 shows a DC component idcn in the first embodiment._−DCAnd ripple 30 degree integration amount Arpl and ratio Arpl / idcn_−DCFIG. 4 is an explanatory diagram showing characteristics of the present invention.
FIG. 8 is an explanatory diagram exemplifying a DC side current idcn and its first and second derivatives in the first embodiment;
FIG. 9 is a block diagram illustrating a specific configuration of a power factor angle calculating unit according to the first embodiment.
FIG. 10 shows a ripple component idcn in the first embodiment._-Rpl, Phase division switching pulse and timing signal (T_30-smp, T_rstFIG.
FIG. 11 is a block diagram showing a specific configuration of a line current amplitude calculation unit according to the first embodiment.
FIG. 12 is an explanatory diagram illustrating an operation of power factor angle control according to the first embodiment;
FIG. 13 is a block diagram showing a configuration of a conventional device.
FIG. 14 is an explanatory diagram of a gate signal operation in a conventional device.
[Explanation of symbols]
1 motor control device, 2 inverter, 3 control operation device,
4b motor, 5 DC power supply, 6b DC side current detector,
7a-7f power MOS-FET, 8 gate drive circuit,
9 smoothing capacitor, 10 low-pass filter,
11 subtractor, 12 differentiator, 13 dead zone calculator,
14 power factor angle calculating means, 15 line current amplitude calculating means,
16 power factor angle controller, 17 overcurrent detection means,
18 Gate signal generation means.

Claims (4)

An inverter that converts DC power and AC power to each other and drives or brakes a motor connected to the AC side, a current detector that detects the DC side current of the inverter, and controls the inverter by switching the inverter into a rectangular wave. An electric motor control device comprising: a control operation device that performs switching of the control operation device based on a DC current of the inverter detected by the current detector. An inverter that converts DC power and AC power to each other and drives or brakes a motor connected to the AC side, a current detector that detects the DC side current of the inverter, and controls the inverter by switching the inverter into a rectangular wave. An electric motor which calculates a power factor based on a DC side current of the inverter detected by the current detector, and performs switching of the control arithmetic device based on the power factor. Control device. The motor control device according to claim 2, wherein the switching of the control operation device is performed so that the power factor becomes +1.0 or -1.0. An inverter that converts DC power and AC power to each other and drives or brakes a motor connected to the AC side, a current detector that detects the DC side current of the inverter, and controls the inverter by switching the inverter into a rectangular wave. And calculating a line current of the electric motor based on the DC side current of the inverter detected by the current detector, and performing switching of the control operation device based on the line current. Motor control device.

Images