JP4663684B2 - AC motor control device and control method - Google Patents

Description

  The present invention relates to an AC motor control device and control method.

  Conventionally, as a method that does not use a rotational speed sensor or a position sensor of an AC motor, a control device that detects a phase current of the motor and performs an estimation calculation of a magnetic pole position is known (see, for example, Patent Document 1).

  Further, as a method not using a current sensor, a current reproduction method has been proposed in which a direct current of an inverter that drives an electric motor is detected, and an alternating current of the electric motor is reproduced from the instantaneous value and the switching state of the inverter (for example, cited) Patent Document 2). This system uses a gate pulse signal for driving an inverter, samples and holds the motor current that appears instantaneously in the DC current of the inverter, and indirectly detects the motor current.

  However, since the current reproduction method reproduces the motor current from the DC current of the inverter and the gate pulse signal, it becomes difficult to capture the motor current component when the gate pulse is extremely short. In particular, the higher the average switching frequency (carrier frequency) of the inverter, the shorter the gate pulse, and the more difficult it is to reproduce the current. If measures are taken to reduce the carrier frequency of the inverter, the efficiency may decrease due to an increase in current harmonics and electromagnetic noise may be generated. Further, it is necessary to perform current sampling at least twice within the carrier period of the inverter, and a special circuit is required. Moreover, two analog input terminals are indispensable to realize with a one-chip microcomputer, and two sets of AD converters are prepared in the microcomputer, or one high-speed AD converter is provided and current is continuously read. There is a need.

JP 2001-251889 A Japanese Patent Laid-Open No. 2-197295

  The present invention provides a control device and a control method that realize a high-performance motor drive with a simple control configuration and a high carrier frequency.

In order to solve the above-mentioned problem, for example, it may be configured as described in the claims.

  According to the present invention, a high-performance AC motor with a simple control configuration and a high carrier frequency can be used without using a position sensor that detects the rotor position of the AC motor and a current sensor that detects current. A control device can be realized.

  Next, an embodiment of a control device for controlling an AC motor according to the present invention will be described with reference to FIGS. In the following embodiments, a permanent magnet synchronous motor will be described as an AC motor. However, other AC motors such as an induction motor and a reluctance motor can be similarly realized.

Example 1
FIG. 1 is a block diagram showing a system configuration of Embodiment 1 of an AC motor control apparatus according to the present invention. The control device according to the present embodiment includes a rotation speed command generator 1 that gives a rotation speed command ω _r ^* to the electric motor, and a control that calculates an AC applied voltage of the electric motor, converts it to a pulse width modulation signal (PWM signal), and outputs the pulse width modulation signal 2, an inverter 3 driven by this PWM signal, a DC power supply 4 for supplying power to the inverter 3, a permanent magnet type electric motor 5 to be controlled, and a current I supplied to the inverter 3 by the DC power supply 4. _The current detector 6 detects ₀ .

The controller 2 calculates the conversion gain 7 for converting the rotational speed command ω _r ^* into the electrical angular frequency command ω ₁ ^* of the electric motor 5 using the number of poles P of the electric motor 5 and the AC phase θ _dc inside the control device. An integrator 8; a current sampler 9 that samples a detected value of the current I ₀ ; an I _0s ^* generator 10 that gives a command to the sampled current value I _0s ; an adder 11 that adds a signal; A current controller 12 that calculates the applied voltage command to the motor 5 so that the sampled current I _0s matches I _0s ^* , and dq inverse that calculates the AC voltage to the motor 5 based on the applied voltage command. It consists of a converter 13 and a PWM generator 14 that creates a gate pulse for driving the inverter 3 based on an AC voltage command.

  The inverter 3 includes a main circuit unit 31 of the inverter and a gate driver 32 that generates a gate signal to the main circuit. The DC power supply 4 that supplies power to the inverter 3 includes an AC power supply 41 and a diode that rectifies the AC. -It is comprised with the bridge | bridging 42 and the smoothing capacitor 43 which suppresses the pulsation component contained in DC power supply.

Next, the operation principle of the first embodiment will be described with reference to FIG. The conversion gain 7 calculates and outputs the electrical angular frequency ω ₁ ^* of the electric motor 5 based on the rotational speed command ω _r ^* from the rotational speed command generator 1. Further, the integrator 8 is used to integrate ω ₁ ^* to calculate the AC phase θ _dc . The current sampler 9 samples and holds the DC current I ₀ of the inverter 3 and takes in a value as I _0s . I _0s is controlled by the current controller 12 so as to coincide with the current command I _0s ^* output from the I _0s ^* generator 10. The dq inverse converter 13 calculates AC voltage commands vu ^{* to} vw ^* based on the applied voltage command V _qc ^* and V _dc ^* which are outputs of the current controller 12. In this embodiment, V _dc ^* is 0. The arithmetic expression of the dq inverse converter 13 is as follows.

Next, the PWM generator 14 converts the AC voltage command into a PWM signal. The gate driver 32 drives the switching element based on the PWM signal, and applies an AC voltage corresponding to V _dc ^* and V _qc ^* to the electric motor 5.

2 and 3 are waveforms showing how the PWM generator 14 creates a gate pulse from an AC voltage command. As shown in the figure, the gate pulse is created by comparing the triangular wave carrier of the carrier wave signal with the magnitude of the AC voltage command. Here, as shown in FIG. 2A, the AC voltage command is a condition of vu ^* > vv ^* > vw ^* and | vw ^* |> | vu ^* |> | vv ^* |. The gate pulses GP _{u to} GP _w at this time are as shown in FIG. In the figure, when the values of GP _{u to} GP _w are “1”, the upper elements (S _up , S _vp , S _wp ) in the main circuit section 31 of the inverter are turned on, and “0” In this case, it means that the lower elements (S _un , S _vn , S _wn ) are turned on. Under the conditions of FIG. 2, the DC current I _{0 of the} inverter that appears in the current detector 6 has a waveform as shown in FIG. That is, I ₀ has an intermittent pulse-like current waveform, and two phase currents appear instantaneously in each, and in FIG. 2, currents in the W phase and the U phase can be observed. These two phase currents that can be observed are the current in the maximum voltage phase and the current in the minimum voltage phase. Furthermore, the energization period with the larger absolute value of the voltage command among these two phase currents becomes longer. Here, the maximum voltage phase is the U phase, the minimum voltage phase is the W phase, and the phase having a large absolute value is the W phase.

That is, if the current is sampled within the conduction period of I ₀ , which is the intermittent current, and in the vicinity of the midpoint of the energization period, the current of the phase having the largest absolute value of the voltage command can be detected. In the three-phase embodiment, the same effect as the sampling in the vicinity of the intermediate point can be obtained as long as it is within the range of 33 to 67% of the energization period from the rise time of the inverter current.

FIG. 3 is different from FIG. 2 in that vu ^* > vv ^* > vw ^* and | vu ^* |> | vw ^* |> | vv ^* |. In this case, the U-phase current value can be detected by sampling the current near the midpoint of the I ₀ energization period.

  Here, the AC voltage command is defined as in Equation 2.

In this case, the voltage command waveform is as shown in FIG. When sampling is performed near the midpoint of the I ₀ energization period, the detectable phase current changes every 60 degrees as shown in FIG. 4B in accordance with the voltage phase θ _v . The current waveform I _0s after sampling has a waveform like the thick line in FIG.

  Since the AC motor has an inductance component, the current is delayed in phase with respect to the voltage. Therefore, the relationship shown in FIGS. 4A and 4C is obtained. Although the current phase varies depending on the motor constant, the load condition, etc., a waveform including almost the maximum current can be observed.

In this embodiment, the I _{0 s,} performs the current control to match the current command I _{0 s} ^*. As a result, a predetermined amount of alternating current flows through the electric motor 5. By sufficiently flowing the current, it is possible to ensure a torque at the time of starting.

  In the conventional “current reproduction method”, when the motor is started, the higher the frequency of the triangular wave carrier, the narrower the gate pulse width, and the current detection itself becomes difficult. At the time of start-up, it is difficult to flow a predetermined amount of current without feedback due to the influence of dead time (short-circuit prevention period of the upper and lower switching elements of the inverter) and the on-voltage drop of the switching elements. However, in this embodiment, the current is sampled in the vicinity of the intermediate time point of the conduction pulse width and the value is controlled, so that the current necessary for starting can be surely passed. Also, current sample timing is simple and does not require complex current detection algorithms. Therefore, in order to realize the present embodiment using a microcomputer, the output of the current detector 6 may be connected to one analog input terminal, and one AD converter (not shown) may be provided. .

Next, a method of generating timing for sampling and holding I ₀ will be described.

In order to sample the current in the vicinity of the middle point of the energization period of I ₀ , for example, the pulse width is measured using the rising edge and falling edge of I ₀ as a trigger, and the middle of the pulse from the rising edge of the next pulse. This can be realized by estimating the time point and generating a sample signal. However, with this method, the hardware is complicated, and there is a concern about malfunction due to the influence of noise or the like.

A method for generating a sample signal will be described with reference to FIG. In the PWM generation method shown in FIGS. 2 and 3, the intermediate time of the energization period coincides with the intermediate time of the upper and lower peaks of the triangular wave carrier. That is, if a sample signal of I ₀ is generated using a zero cross of a triangular wave carrier as a trigger, I _0s can be easily captured.

Depending on the PWM method, a voltage command waveform as shown in FIG. 6 may be used. These are PWM modulation methods called “two-phase switching method”, and one of the three phases does not perform a switching operation. For example, in FIG. 6, in the period of 60 degrees <θ _v <120 degrees, the U-phase switching element keeps the upper side (S _{up in} FIG. 1) turned on and the lower side (S _{un in} FIG. 1) shows the off state. keep. This voltage command can be realized by adding a common voltage component (zero phase component) to all three phases to the original sinusoidal voltage command.

FIG. 7 shows the voltage command, gate pulse, and I ₀ under these conditions.

  As can be seen from FIG. 7, when the U-phase voltage is larger than the upper peak of the triangular wave carrier, the upper peak time of the triangular wave carrier is the intermediate time of the current passing period. Therefore, the current sampling may be performed at this timing. Note that when the voltage command of one phase is saturated to the negative side, for example, in the period of 0 to 60 degrees in FIG. 7, the current may be sampled at the lower peak time of the triangular wave carrier.

(Example 2)
Next, Embodiment 2 according to the present invention will be described with reference to FIG.

  In the first embodiment, the current flowing through the electric motor is caused to flow to a predetermined value, and the current necessary for driving is ensured. On the other hand, the second embodiment aims to detect the “effective current” flowing through the motor and control the motor with high performance.

FIG. 8 shows a block configuration of the controller 2A, and the second embodiment can be realized by using the controller 2A in FIG. 8 instead of the controller 2 in FIG. In FIG. 8, the filter 15 for I _0s and the ω ₁ compensator 16 for adding Δω ₁ to the electrical angular frequency command ω ₁ ^* of the motor based on the output of the filter 15 and the application from the ω ₁ ^* to the motor. The voltage command calculator 17 for determining the voltage is a different block from the controller 2 of FIG.

The controller 2A does not perform current control, and uses the voltage command calculator 17 to calculate the voltage command V _qc ^* directly from ω ₁ ^* . Thus, the motor control method is based on V / F constant control, but there is no problem even if current control is used as in the controller of FIG.

Next, the filter 15 and the ω ₁ compensator 16 which are characteristic parts of the present embodiment will be described. In the controller 2A of FIG. 8, a filter 15 is provided for I _0s , and this filter output is an effective current I _a . The principle of obtaining the effective current I _a by _passing the filter 15 through I _0s will be described below.

  The applied voltage V to the motor and the current I are defined as in Equation 3. The voltage V is a u-phase voltage and the current I is a u-phase current.

In _Equation 3, V ₀ is a voltage amplitude, and in this embodiment, coincides with V _qc ^* , I ₀ is a current amplitude, θ _v is a voltage phase, and δ is a current phase (power factor angle). Here, I in Equation 3 can be expressed as in Equation 4.

From Equation 4, the magnitudes I _a and I _r of the effective current and reactive current are expressed as in Equation 5.

Since Equation 4 is a u-phase current, it is observed as I _{0s in} a period of 60 degrees <θ _v <120 degrees. The filter 15, it is assumed that the current of the period is averaged, try seeking during this time mean value I _m of the number 4. Because it is the average value of 60 degrees-120 degrees period,

It becomes. That is, the effective current component I _a, by using the average value I _m,

Can be calculated as Therefore, the effective current I _a can be detected by averaging I _0s through the filter 15.

Active current I _a, since it indicates the magnitude of the load of the motor directly, by effectively used to control the control device of a more stable motor can be realized. In the controller 2A shown in FIG. 8, Δω ₁ that is a compensation amount to ω ₁ ^* is calculated based on I _a . The ω ₁ compensator 16 performs incomplete differentiation on I _a , lowers the electrical angular frequency when the load increases and the effective current increases, and conversely increases the electrical angular frequency when the load decreases. Is working. As a result, transient vibration due to load fluctuation can be greatly reduced, and a more stable AC motor control device can be realized.

The filter 15 needs to remove a harmonic component that is six times the frequency for driving the electric motor. In this case, the pulsation component can be easily removed by using a moving average rather than through a first-order lag filter. FIG. 9 is a block diagram when the filter 15 is a moving average filter 15B. In FIG. 9, the moving average filter 15 </ b> B includes a signal delay unit 151 that is a delay element for one calculation cycle, an adder 11, and a filter gain 152. The period for taking this moving average may be set so as to correspond to the electrical angle of 60 degrees. As a result, the pulsating component included in I _0s is ideally deleted, and the effective current I _a can be detected more accurately.

(Example 3)
Next, Embodiment 3 according to the present invention will be described with reference to FIGS. 10 and 11. FIGS. 10 and 11 are examples in which a reactive current detection method and control using the reactive current are embodied.

FIG. 10 shows a block configuration of the controller 2C, and the third embodiment can be realized by using FIG. 10 instead of the controller 2 in FIG. In Figure 10, the I _a · I _r calculator 18 for calculating at least one of the active current component I _a and the reactive current component I _r flowing from I _{0 s} to the motor, operation start of I _a · I _r calculator 18 An interrupt generator 19 that generates an interrupt and an I _r ^* generator 20 that gives a current command I _r ^* to a reactive current I _r are different from the controllers 2 and 2A in the above embodiments. It is.

Next, the operation principle of this embodiment will be described. The I _a · I _r calculator 18 calculates an effective current and a reactive current flowing through the motor. The interrupt generator 19 generates an interrupt signal every 60 degrees of θ _v = 0, 60 degrees, 120 degrees, 180 degrees, 240 degrees, and 300 degrees from the AC voltage command, and the I _a · I _r computing unit Is triggering.

When the current sample shown in FIG. 4 is performed, the current phase detected as I _0s changes every 60 degrees of θ _v . Here, as shown in FIG. 11, the current value immediately after the change of the observable current phase is defined as I _1, and the value immediately before the change is defined as I ₂ . Now, θ _v is assumed to be 60 to 120 degrees period. Since U-phase current can be observed during this period, I ₁ and I ₂ are

It can be expressed as. From Equation 8,

It becomes. From Equation 5, I _a and I _r are

It becomes. Therefore, if the current sample values before and after the detectable current phase is switched are used, the effective current and reactive current of the motor can be observed. Further, in the steady state, as shown in FIG. 11, since I ₂ ′ = I ₂ , the calculation can be performed using I ₁ and I ₂ ′. According to this method, the effective current and the reactive current can be calculated without any calculation delay.

Thus, if the effective current and the reactive current can be observed, the phase information of the motor current can be obtained, and more advanced motor control can be realized. In Figure 10, relative to the reactive current I _r, giving a current command I _r ^*, reactive current control is performed to a predetermined value. By controlling the reactive current, it becomes possible to realize an efficiency optimization operation of the electric motor, a field weakening control, and the like, and a higher-performance AC motor control device can be provided.

Example 4
Next, Embodiment 4 according to the present invention will be described with reference to FIGS. 12 and 13.

  In Example 3 relating to the detection method of the effective current flowing through the electric motor and the reactive current, the timing that can be calculated was only once at 60 °. Therefore, it may be affected by noise and the like.

The fourth embodiment provides a method for calculating the effective current and the reactive current by integrating the current sample value I _0s in order to make it less susceptible to noise.

FIG. 12 shows a block configuration of the controller 2D. In FIG. 12, an active current component I _a that flows from I _0s to the electric motor, an I _a · I _r calculator 18D that calculates a reactive current component I _r , and a periodic function F _c that is used by the I _a · I _r calculator 18D. , And a function generator 21 that generates F _s and an I _r ^* generator 20 that gives a command current I _r ^* to a reactive current I _r , are the controllers 2, 2A, It is a part different from 2C.

Next, the operation principle of this embodiment will be described. The operation of the controller 2D shown in FIG. 12 is basically the same as that shown in FIG. 10, but the method for obtaining I _a and I _r is greatly different.

The function generator 21 generates the waveforms (F _c (θ _v ), F _s (θ _v )) shown in FIG. Function F _s is a function of repeatedly outputting the waveform of 60 to 120 degrees duration of sin [theta _v, the function F _c is one repeated waveform of 60-120 degree period of cos [theta] _v.

The I _a / I _r calculator 18D performs the following integral calculation.

In the above equation, θ _v0 is an arbitrary voltage command phase.

Now, the case theta _v is 60 to 120 degrees, illustrating the operation principle. In this period, F _s = sin θ _v and F _c = cos θ _v , and I _0s can also be expressed by _Equation 4. When Expression 13 is expanded, it becomes as follows.

  Therefore, from Equation 15,

Thus, the effective current I _a can be calculated from I _am .

  Similarly, when the number 14 is expanded,

It becomes.

  Therefore, from Equation 15,

Thus, the reactive current I _r can be calculated from I _rm .

In the fourth embodiment, the effective current I _a and the reactive current I _r can be calculated within 60 degrees by integral calculation. Because of the integral calculation, it is difficult to be influenced by external factors such as noise, and it is possible to realize a control device for an AC motor with higher accuracy and increased stability.
(Example 5)
Next, Example 5 will be described with reference to FIGS. 14 and 15.

  In actual motor control, “vector control” is often used to control the motor current by separating the motor current into a magnetic flux axis component (d-axis component) and a component (q-axis component) orthogonal thereto. . In the present embodiment, this vector control is realized.

FIG. 14 shows a block configuration of the controller 2E. In FIG. 14, an ar-dq converter 22 for calculating currents of I _dc and I _qc axes with reference to the magnetic flux axis of the motor from the active current component I _a and the reactive current component I _r , and on the d axis An I _d ^* generator 23 that generates a current command I _d ^*, and a speed controller 24 that calculates a difference between the rotational speed command ω _r ^* and the estimated speed value ω _r and outputs a current command I _q ^* on the q-axis. , An axis error estimator 25 for calculating an axis error Δθ between the d-axis position (phase) of the motor and a control phase (θ _dc ), and a PLL controller for correcting the rotational speed so that the axis error Δθ becomes zero. 26 is a portion different from the controllers 2, 2A, 2C, and 2D in the above embodiments. In addition, the current controller 12 is provided so that I _dc and I _qc coincide with I _d ^* and I _q ^* , respectively.

  Next, the operation of this embodiment will be described.

I _a · I _r I _a obtained in the computing unit 18D, and based on the I _r, the ar-dq converter 22, I _dc, I _qc is calculated. Here, I _dc and I _qc are _{obtained according} to _Equation 19.

  In Equation 19, ψ is the phase difference angle between the voltage phase and the q axis,

It can be asked. The phase relationship between the voltage and current of these motors is shown in FIG.

I _dc and I _qc are components corresponding to the excitation current and torque current of the motor, respectively, and are controlled by the current controller 12 so as to coincide with the command values I _d ^* and I _q ^* , respectively.

The axis error calculator 25 estimates and calculates an error angle Δθ between the d-axis phase (θ _dc ) assumed in the control and the d-axis phase in the actual motor. Δθ can be calculated by using a voltage command and a current detection value. The PLL controller 26 outputs the motor speed ω _r so that the axis error Δθ becomes zero. In a steady state, Δθ becomes zero, and the dq axis of the motor can be matched with the control axis without directly detecting the magnetic pole axis. Further, ω _r is an estimated rotational speed value of the electric motor, and the torque current command I _q ^* is calculated in the speed controller 24 so that the deviation from the rotational speed command ω _r ^* becomes zero. I _q ^* is compared with I _qc, and is controlled via the current control 12 so that they match. In addition, current control is also performed for the d-axis current so that I _dc becomes a predetermined value. In a non-salient magnetic motor, I _d ^* = 0 is usually set.

  As described above, according to the present embodiment, individual control of the torque current and excitation current of the electric motor is possible, and vector control can be realized.

(Example 6)
Next, Embodiment 6 of the present invention will be described with reference to FIG.

  In this embodiment, a control device capable of high-speed response at medium and high speeds is provided.

FIG. 16 shows a block configuration of the controller 2F. In FIG. 16, two current samplers 9 for sampling I ₀ at two locations within a half cycle of the triangular wave carrier are added, and the phase current reproducer 27 reproduces the three-phase current of the motor. For this phase current reproduction method, the conventional technique described in JP-A-2-197295 may be used. The three-phase current is coordinate-converted by the dq converter 28, and the calculated values of I _dc and I _qc are switched by the switch 29.

When driving an electric motor with an inverter, the lower the speed and the higher the carrier frequency, the narrower the gate pulse signal of the inverter becomes, and the operation of the phase current reproducer 27 becomes difficult. However, in such a condition, the two switches 29 are respectively switched to the upper side to switch to current detection calculated from I _a · I _r . On the other hand, when the pulse width is sufficiently large, the motor current is detected by the current reproducer 27 to realize current control with high response.

  As described above, according to the present embodiment, a higher-performance motor control device can be realized by switching the current detection method depending on conditions.

(Example 7)
FIG. 17 is a schematic diagram of an AC motor control device according to the present invention. The part numbers 1 to 3, 5, 6, 41, 42, and 43 shown in the figure are the same as those having the same numbers in FIG.

  This embodiment is characterized in that the controller 2, the inverter 3, the current detector 6, and the diode bridge 42 are integrated and modularized. The module is provided with a rotational speed command signal from the rotational speed command generator 1, an input terminal of the power supply 41, a connection terminal of the smoothing capacitor 43, and a connection terminal of the AC motor 5, all other components being in the module. It is contained in. In this embodiment, the rotational speed command generator 1 uses a microcomputer. In the module, a controller 2 using a microcomputer, an inverter 3 constituted by a switching device 3, a current detector 6 comprising a shunt resistor 6, and a diode bridge 42 are housed.

  According to the embodiments described so far, a high-performance control device for an AC motor without a position sensor and a current sensor can be realized by an inexpensive microcomputer, and thus such a control device can be modularized.

  As a result, the power module can be handled like a single component, and assembling becomes easy, and at the same time, the entire apparatus can be miniaturized.

It is a block diagram which shows the motor control apparatus which is one Example of this invention. It is an example of a waveform which shows the principle of the PWM modulation | alteration in the Example of this invention, and the relationship of a current sample. It is an example of a waveform which shows the principle of the PWM modulation | alteration in the Example of this invention, and the relationship of a current sample. It is a wave form diagram which shows the relationship between the voltage command in the Example of this invention, a phase current, and an electric current sample value. It is a wave form diagram which shows the timing of the current sample in the Example of this invention. It is a wave form diagram which shows the voltage command at the time of the two-phase modulation of a prior art, and the relationship of a triangular wave carrier. In the Example of this invention, it is a wave form diagram which shows the timing of the current sample at the time of using a two-phase modulation system. It is a block diagram which shows the motor control apparatus by the other Example of this invention. It is a block diagram which shows the moving average filter in the other Example of this invention. It is a block diagram which shows the electric motor control apparatus in the other Example of this invention. It is a wave form diagram explaining operation | movement of the electric motor control apparatus in the other Example of this invention. It is a block diagram which shows the electric motor control apparatus in the other Example of this invention. It is a wave form diagram explaining operation | movement of the electric motor control apparatus in the other Example of this invention. It is a block diagram which shows the electric motor control apparatus in the other Example of this invention. It is a vector diagram which shows the dq coordinate axis on the basis of the magnetic flux axis of an electric motor, and the relationship between a voltage and an electric current. It is a block diagram which shows the electric motor control apparatus in the other Example of this invention. It is a schematic diagram of the electric motor control apparatus of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Speed command generator, 2 ... Controller, 3 ... Inverter, 4 ... DC power supply, 5 ... AC motor, 6 ... Current detector, 7 ... Conversion gain, 8 ... Integrator, 9 ... Current sampler, 10 ... I _0s ^* Generator, 11 ... Adder, 12 ... Current controller, 13 ... dq inverse converter, 14 ... PWM generator, 31 ... Inverter main circuit part, 32 ... Gate driver, 41 ... AC power supply, 42 ... Diode bridge, 43 ... smoothing capacitor.

Claims (2)

In the control method of the AC motor that controls the inverter that drives the AC motor using a sinusoidal continuous current,
Sample means for sampling the inverter current detected by the current detection means for detecting the current supplied to the inverter by the DC power supply once in a half cycle of the PWM carrier signal ,
Triggered by an intermediate time between the positive peak value and the negative peak value of the carrier signal,
The AC motor control method according to claim 1, wherein the sampling means samples the inverter current value using the trigger. In an AC motor control device that controls an inverter that drives an AC motor using a sinusoidal continuous current,
Sample means for sampling the inverter current detected by the current detection means for detecting the current supplied to the inverter by the DC power supply once in a half cycle of the PWM carrier signal ,
Triggered by an intermediate time between the positive peak value and the negative peak value of the carrier signal,
The said sample means samples the said inverter electric current value using the said trigger, The control apparatus of the AC motor characterized by the above-mentioned.

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