JP5278723B2 - Motor control device and motor control method - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To suppress disorder in the current response at switching of the control system, in a device for controlling a motor in which PWM control and rectangular wave control can be switched. <P>SOLUTION: The device for controlling the motor is constituted of a first control part 4, in which the applied voltage to the motor 1 is controlled by the sinusoidal wave pulse width modulation for outputting a first drive signal that drives an inverter 2; a second control part 5, in which the applied voltage to the motor 1 is controlled by the phase adjustment of a rectangular wave to output a second drive signal that drives the inverter 2; a selection part that selects the drive signal of the inverter 2 from the first and second drive signals, according to a driving state of the motor 1; an effective value calculating part 52, in which an effective value of one cycle of an electric angle of an AC voltage supplied to the motor 1 is calculated by being separated into a positive side and a negative side, with the voltage at a neutral point N of the motor 1 set as a reference; and an adjustment part 53 in which the pulse width of the rectangular wave generated in the second control part 5 is adjusted, based on the difference between the positive-side effective value and the negative-side effective value. <P>COPYRIGHT: (C)2009,JPO&amp;INPIT

Description

  The present invention relates to a motor driving device that drives an AC motor.

  2. Description of the Related Art A motor driving device that drives a three-phase AC motor by converting a DC voltage into an AC voltage using an inverter is known. In such a motor drive device, the motor current is generally controlled by sinusoidal pulse width modulation (PWM). Further, in a general permanent magnet type three-phase AC motor, the obtained torque varies depending on the current phase even if the current value applied to the motor is the same. Therefore, maximum torque control is also performed to change the current phase so that the maximum torque can be obtained.

  Further, in such a motor, there is a phenomenon in which the torque starts to rapidly decrease when the rotation speed exceeds a certain threshold value. Since the motor has a coil component inside, the induced voltage increases as the rotational speed increases. If the induced voltage exceeds the voltage applied to the motor at a certain rotation speed, the motor cannot be controlled, so it is necessary to lower the induced voltage. Therefore, the induced voltage is lowered by advancing the phase rather than the current phase by the maximum torque control by utilizing the characteristic of the motor that “the induced voltage can be reduced as the phase of the current waveform is advanced”. This control is called field weakening control. However, when this control is performed, the torque obtained by the advance of the current phase decreases. That is, the torque in the middle rotation region and high rotation region of the motor is insufficient.

Therefore, in order to obtain a large output in the middle and high rotation range of the motor, control called rectangular wave control or overmodulation PWM control may be performed.
When a DC power source is used as the power source of the inverter, the output voltage waveform from the inverter is basically a set of pulses in which the positive side of the power source voltage of the inverter is high level and the negative side is low level. Usually, the duty ratio of each pulse is controlled so that the fundamental wave component in the set of pulses becomes a sine wave in a certain period. This control method is referred to as sine wave PWM control. However, in the sine wave PWM control, the effective value of the fundamental wave component can be increased only up to 0.61 times the power supply voltage of the inverter. This magnification is called a modulation rate, which means a limit to increasing the amplitude of the fundamental wave component.

  Here, in order to further increase the amplitude of the fundamental wave component beyond the above limit, the fundamental wave component is distorted. Specifically, the duty ratio of each pulse is made larger than the sine wave PWM control on the peak side of the fundamental wave component, and smaller on the valley side. The most distorted state is a rectangular wave of one pulse, which is 100% on the mountain side and 0% on the valley side, and this is called a rectangular wave control method. In the case of rectangular wave control, the modulation factor is 0.78, which is higher than the maximum value 0.61 of sine wave PWM control, and the effective value of the fundamental wave component is high. In general, the rectangular wave control is preferably applied in a high rotation range where field-weakening control is required. By such control, the amount of current phase that must be advanced for field-weakening control can be reduced, and a larger torque can be obtained accordingly.

  The sine wave PWM control and the rectangular wave control method have different suitable application areas depending on the rotation speed of the motor, so that the control method is switched during the rotation of the motor. However, in the case of a sine wave and a rectangular wave having the same phase, the rectangular wave generates a larger torque. Therefore, if the control method is simply switched, a large torque fluctuation occurs at the time of switching. Thus, for example, Patent Document 1 shown below describes an invention of a motor control device capable of suppressing torque fluctuation at the time of switching between sine wave PWM control and rectangular wave control. This motor control device controls the phase and amplitude of the sine wave to be changed simultaneously and continuously during the switching, unlike the conventional case where the phase adjustment of the rectangular wave is performed during the switching. Specifically, PWM control is performed based on a modified sine wave signal having an intermediate phase and amplitude between voltage waveforms before and after switching. This is referred to as overmodulation PWM control. Overmodulation PWM control uses an intermediate voltage waveform between sinusoidal PWM control and rectangular wave control, and the modulation factor is 0.61 to 0.78. In general, overmodulation PWM control is a suitable application range in the middle rotation range or higher.

  In such a motor, current feedback control is performed so that the current command is equal to the feedback value in which the current actually flowing in the motor is detected. However, the current actually flowing through the motor may include a direct current offset. This offset is a DC noise component detected even when the motor is stopped, but may change due to a temperature change or the like. In addition, since the current detection circuit has non-linearity, even if the measurement is performed when the motor is stopped, the amount of offset during operation of the motor may not coincide. Therefore, in Patent Document 2 shown below, the offset of the circuit of the current detection system for each of the three phases is estimated, and current feedback is performed by subtracting the offset from the detected current value. The offset is estimated based on an integral value obtained by integrating a product of a torque command for performing constant speed control and a value corresponding to the electrical angle for one cycle with the electrical angle.

Japanese Patent Laid-Open No. 11-285288 (see paragraphs 3 to 28, etc.) JP-A-8-9671 (see paragraphs 2-8)

  The influence of the offset of the DC component of the current detection system on the feedback control differs between the sine wave PWM control and the rectangular wave control. Therefore, when the control method is switched from sine wave PWM control to rectangular wave control, the current response becomes discontinuous and the motor current is disturbed.

  The present invention was devised in view of such problems, and in a motor control device capable of switching between PWM control and rectangular wave control, it suppresses disturbance of current response that occurs when switching the control method. With the goal.

In order to achieve the above object, the characteristic configuration of the motor control device according to the present invention is as follows:
An inverter that converts a DC voltage into an AC voltage for driving the motor based on the inverter drive signal;
A current detector for detecting a motor current flowing through the motor;
Based on a voltage command signal on which an offset current calculated using the motor current is superimposed, a voltage applied to the motor is controlled by sinusoidal pulse width modulation, and a first drive signal for driving the inverter is output. A first control unit;
Based on the voltage command signal offset current that will be computed is superimposed using the motor current, the voltage applied to the motor is controlled by the phase adjustment of the rectangular wave, and outputs a second driving signal for driving the inverter A second control unit;
A selection unit for selecting the inverter drive signal for driving the inverter according to the operation state of the motor from the first and second drive signals;
An effective value calculation unit for calculating an effective value for one period of an electrical angle of an AC voltage applied to the motor, by dividing the positive value and the negative value on the basis of the voltage at the neutral point of the motor;
When the inverter drive signal is switched from the first drive signal to the second drive signal by the selection unit, the positive-side effective value and the negative-side effective value of the first drive signal in the immediately preceding cycle The pulse adjustment width obtained by dividing the difference by the peak value of the inverter drive signal to the inverter is added to the pulse with the larger effective value of the square wave pulses on the positive side and the negative side. An adjustment unit for adjusting a pulse width of the rectangular wave generated by the second control unit by subtracting from a pulse having a smaller effective value ;
It is in the point provided with.

  According to this configuration, the pulse width of the rectangular wave is adjusted based on the effective value for one electrical angle cycle of the AC voltage applied to the motor. Therefore, the positive side and the negative side of the effective values of the inverter output by the pulse width modulation control method before switching to the rectangular wave control method and the inverter output after switching can be made substantially equal. As a result, it is possible to suppress the mismatch between the positive and negative effective values at the time of switching the control method, and it is possible to suppress the disturbance of the current response that occurs at the time of the switching.

Further, the effective value of the inverter output by the rectangular wave control method is a constant value, and can be easily obtained based on the amplitude (or peak value) of the inverter drive signal and the pulse adjustment width. Therefore, the pulse adjustment width which becomes the adjustment width of the pulse width can be obtained from the difference between the effective values and the crest value calculated by dividing the voltage at the neutral point of the motor into the positive side and the negative side. Since the pulse adjustment width for adjusting the pulse width can be obtained by a simple calculation, it is possible to suppress disturbance of the current response at the time of switching the control method with a simple configuration.

Further, in the motor control device according to the present invention, the current detection unit includes:
Obtain the value of the electrical angle from the rotation detection unit provided in the motor for each predetermined sampling time,
When the obtained electrical angle value is smaller than the electrical angle at the previous sampling, determine one cycle of the electrical angle,
The motor current value detected over the determined electrical angle of one cycle is integrated to detect the motor current and the DC offset current of the motor current.
It is characterized by that.

  According to this configuration, the electrical angle is acquired from the rotation detection unit every predetermined sampling time, and the acquired electrical angle is compared with the electrical angle at the previous sampling. When the electrical angle acquired this time is larger than the electrical angle acquired last time, the electrical angle has increased, so it can be determined that one cycle has not been reached. If the electrical angle obtained this time is smaller than the electrical angle obtained last time, the electrical angle has exceeded 2π (= 360 °). Therefore, it can be determined that the electrical angle has reached one cycle. Corresponding to the electrical angle of one cycle, the motor current and the offset current for one electrical angle cycle are detected. According to this configuration, it is possible to reliably know one cycle of the electrical angle based on the difference between the electrical angles acquired at each sampling timing. In other words, an electrical angle period that changes at the sampling interval can be used as an error range, and one cycle of the electrical angle can be determined with sufficient accuracy. Therefore, the motor current and the offset current can be detected with high accuracy. As a result, control of the inverter drive signal and calculation of the effective value can be performed smoothly.

The characteristic configuration of the motor control method according to the present invention is as follows.
A voltage conversion step of applying an inverter drive signal to the inverter and converting the DC voltage into an AC voltage for driving the motor;
A current detection step of detecting a motor current flowing through the motor;
Based on a voltage command signal on which an offset current calculated using the motor current is superimposed, a voltage applied to the motor is controlled by sinusoidal pulse width modulation, and a first drive signal for driving the inverter is output. 1 control process;
Based on the voltage command signal offset current that will be computed is superimposed using the motor current, the voltage applied to the motor is controlled by the phase adjustment of the rectangular wave, and outputs a second driving signal for driving the inverter A second control step;
A selection step of selecting, from the first and second drive signals, the inverter drive signal for driving the inverter according to the operating state of the motor;
An effective value calculation step of calculating an effective value for one period of an electrical angle of an AC voltage applied to the motor by dividing the effective value into a positive side and a negative side based on a voltage at a neutral point of the motor;
When the step of generating the inverter drive signal is changed from the first control step to the second control step, the positive-side effective value and the negative-side effective value of the first drive signal in the immediately preceding cycle are changed. The pulse adjustment width obtained by dividing the difference from the value by the peak value of the inverter drive signal to the inverter is the pulse having the larger effective value among the square wave pulses on the positive side and the negative side. In addition, an adjustment step of adjusting a pulse width of the rectangular wave generated in the second control step by subtracting from a pulse having a smaller effective value ,
It is in the point provided with.

  According to this motor control method, the same operational effects as the operational effects of the motor control device described above and the operational effects thereof can be obtained.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram schematically showing a configuration example of a motor control device according to the present invention. The motor 1 is a three-phase AC motor. The motor 1 includes, for example, a permanent magnet type rotor and a stator having a multi-phase stator coil that generates a magnetic field for applying a rotational force to the rotor. The stator includes U-phase, V-phase, and W-phase three-phase stator coils 1u, 1v, and 1w. One end of each of the stator coils 1u, 1v, and 1w is commonly connected at an electrical neutral point C and Y (star) connected. The other end is connected to an inverter 2 described later.

  The inverter 2 converts the DC voltage into an AC voltage for driving the motor 1 based on the inverter drive signal. The inverter 2 is configured as a bridge circuit in which two switching elements are connected in series between the power source VDC and the ground, that is, between the positive side and the negative side of the DC power source, and three lines are connected in parallel. For example, a power MOSFET (hereinafter simply abbreviated as FET) can be used as the switching element. A p-channel FET is used as a high-side switch on the power supply side, and an n-channel FET is used as a low-side switch on the ground side. Each FET has a built-in flywheel diode (also referred to as a freewheel diode) or is connected in parallel. Of course, other elements such as a power transistor may be used, or a booster circuit and the like may be provided and all may be configured using an n-channel type. The other ends of the stator coils 1u, 1v, and 1w described above are connected to a connection point between a p-channel FET and an n-channel FET connected in series.

  Two of the lines connected from the inverter 2 to the other ends of the stator coils 1u, 1v, 1w are provided with current sensors 3v, 3w. The current sensors 3v and 3w detect the current flowing through the motor 1 and output the detected current to the calculation unit 10. Since the sum of instantaneous values of the three-phase stator coils 1u, 1v, and 1w is zero, it is sufficient to detect the current for two phases of the current sensors 3v and 3w as shown in FIG. Since high-frequency noise is superimposed on the current flowing through the motor 1, the calculation unit 10 performs calculation using the current value after passing through the low-pass filters 31 to 33. In this way, the motor current flowing through the motor 1 and the DC offset current of the motor current are detected. Therefore, the current sensors 3v and 3w, the low-pass filters 31 to 33, and the calculation unit 10 correspond to the current detection unit 3 of the present invention.

  Further, the motor 1 is provided with a rotation angle sensor (resolver) 11 (rotation detection unit), and detects the rotation angle (mechanical angle) of the rotor of the motor 1. The resolver 11 is set according to the number of poles of the rotor of the motor 1, converts the rotation angle to the electrical angle θ, and outputs a signal corresponding to the electrical angle θ to the calculation unit 10. Based on the output of the resolver 11, the arithmetic unit 10 acquires the rotational speed of the motor 1, the rotational position of the rotor, and the electrical angle θ. The calculation unit 10 performs known vector control and torque control calculations using information input from the current detection unit 3 and the resolver 11. The control command output unit 9 of the calculation unit 10 generates a control command that is a source of a voltage command for driving the motor 1. The calculation unit 10 further performs PI control (proportional integration control) that performs proportional gain in the P operation and phase compensation in the I operation, and outputs a voltage command to the PWM signal output unit 41 and the rectangular wave signal output unit 51. In this example, the PWM signal output unit 41 outputs a first voltage command signal for PWM control, and the rectangular wave signal output unit 51 outputs a second voltage command signal for rectangular wave control.

  In the motor control apparatus of the present embodiment, the inverter drive signal given to the inverter 2 is configured to be switchable by the selection unit 6. The selection unit 6 selects an inverter drive signal from a first drive signal and a second drive signal, which will be described later, based on an instruction from the calculation unit 10 according to the operating state of the motor 1. The first drive signal is a signal that has been subjected to sinusoidal pulse width modulation based on the first voltage command signal, and is output from the PWM signal output unit 41. The second drive signal is a rectangular wave signal generated based on the second voltage command signal, and is output from the rectangular wave signal output unit 51. The operating state is, for example, the rotational speed of the motor 1, and when the rotational speed increases, the sine wave PWM control is switched to the rectangular wave control to increase the torque as described above.

The PWM output unit 41 and the calculation unit 10 control the voltage applied to the motor 1 by sinusoidal pulse width modulation based on a voltage command (first voltage command signal) calculated using the motor current. Therefore, the PWM output unit 41 and the calculation unit 10 correspond to the first control unit 4 of the present invention. Further, the rectangular wave signal output unit 51 and the arithmetic unit 10, based on a command voltage computed using the motor current (second voltage command signal), controlling the voltage applied to the motor 1 by the phase adjustment of the rectangular wave To do. Therefore, the rectangular wave signal output unit 51 and the calculation unit 10 correspond to the second control unit 5 of the present invention.

  The rectangular wave signal output unit 51 is provided with an effective value calculation unit 52 and an adjustment unit 53. The effective value calculation unit 52 calculates the effective value of one cycle of the AC voltage applied to the motor 1 by dividing it into a positive side and a negative side with reference to the neutral point voltage of the motor 1. The adjustment unit 53 adjusts the duty of the rectangular wave generated by the second control unit 5 based on the difference between the positive effective value and the negative effective value. Specifically, the pulse width is changed by adjusting the phase of the changing point of the rectangular wave having a duty of 50%. In the present embodiment, for the sake of convenience, the configuration in which the square wave signal output unit 51 is provided with the effective value calculation unit 52 and the adjustment unit 53 is illustrated, but of course, other configurations such as the calculation unit 10 may be used. . In the above configuration example, the rectangular wave signal output unit 51 in FIG. 1 is also preferably illustrated so that the first voltage command signal is input, but is omitted to simplify the drawing.

  FIG. 2 is a waveform diagram schematically showing an inverter drive signal (first drive signal D1) by sine wave PWM control and a fundamental wave component B1 at the time of sine wave PWM control. As is well known, an inverter drive signal by PWM control is modulated using a carrier wave such as a triangular wave. In the sine wave PWM control, the amplitude of the fundamental wave component B1 varies depending on the modulation rate, and is 0.68 at the maximum. Therefore, the waveforms of the first drive signal D1 and the fundamental wave component B1 shown in FIG. 2 are examples. Ideally, the amplitude center of the waveform of the fundamental wave component B1 of the first drive signal D1 is ideally the neutral point N (= 1/2 of the power supply voltage VDC) as shown in FIG. However, as described above, the current detection unit 3 that detects the motor current includes a DC offset current. Thereby, the first voltage command signal also includes a direct current offset, and the direct current offset may also be superimposed on the inverter drive signal modulated based on the first voltage command signal.

  FIG. 3 is a waveform diagram schematically showing a fundamental wave component B1 having a direct current offset. In the case of having a DC offset a, the center of amplitude of the fundamental wave component B1 deviates from the neutral point N as shown in the figure. As a result, a difference occurs between the positive voltage effective value S1 and the negative voltage effective value S2 with the neutral point N (= VDC / 2) as a boundary. For example, as shown in FIG. 3, the positive voltage effective value S1 is larger than the negative voltage effective value S2. In addition, since the power supply voltage has an upper limit (for example, VDC) and a lower limit, the portion exceeding the upper limit and the lower limit is conceptually clipped.

  On the other hand, the inverter drive signal by the rectangular wave control is not modulated using a carrier wave, and the modulation rate is fixed (= 0.78). Therefore, the voltage command (second voltage command signal) is not affected by the DC offset a as in the sine wave PWM control. FIG. 4 is a waveform diagram schematically showing an inverter drive signal (second drive signal D2) by the rectangular wave control and its fundamental wave component B2. Since the modulation factor by the rectangular wave control is larger than the modulation factor by the sine wave PWM control, the fundamental wave component B2 of the second drive signal D2 has a larger amplitude than the fundamental wave component B1 of the first drive signal D1. Further, the duty of the rectangular wave is 50%, and the signal changes at T / 2 which is half of the period T. Therefore, as shown in FIG. 4, the effective value U1 on the positive side and the effective value U2 on the negative side are equal in principle.

  Here, a case where the inverter drive signal is switched from the first drive signal D1 by the sine wave PWM control to the second drive signal D2 by the rectangular wave control by the selection unit 6 in a state where there is an offset a for the direct current. Think. When the inverter 2 was driven by the first drive signal D1, there was a difference between the positive-side effective value S1 and the negative-side effective value S2. However, when the inverter 2 is driven by the second drive signal D2, a drive signal in which there is no difference in effective value is given to the inverter. For this reason, mismatching occurs at the time of switching, the current flowing through the motor 1 is disturbed, and the rotation of the motor 1 is affected.

  Therefore, the rectangular wave signal output unit 51 corrects the duty of the second drive signal D2, and suppresses disturbance of the current flowing through the motor 1. First, the effective value calculator 52 calculates the positive effective value S1 and the negative effective value S2 of the first drive signal D1. This calculation can be performed based on the first voltage command signal on which the DC component offset a is superimposed. Alternatively, the calculation may be performed based on the fundamental wave component B1 derived from the first voltage command signal. The calculation may be performed based on the modulated first drive signal D1. That is, when the calculation is performed based on the fundamental wave component B1 as shown in FIG. 3, the calculation can be performed by integrating the following equation (1), where A is the amplitude of the fundamental wave component. When integrating, the effective value corresponding to clipped may be subtracted.

  B1 = A · sin θ + a (1)

  Next, the adjustment unit 53 obtains the difference ΔS between the effective values S1 and S2, and adjusts the pulse width of the rectangular wave so that the effective values U1 and U2 of the second drive signal D2 have the same difference ΔS. . A specific example will be described with reference to FIG. FIG. 5 is a waveform diagram schematically showing an example of adjusting the pulse width of the inverter drive signal by rectangular wave control. The pulse width of the rectangular wave when the duty is 50% is π. If the positive pulse width after adjustment is PP and the negative pulse width is NP, the respective pulse widths are expressed by the following equations (2) and (3). Further, the positive side effective value u1 and the negative side effective value u2 at this time are expressed by the following equations (4) and (5).

PP = π + 2 · Δφ (2)
NP = π-2 · Δφ (3)
u1 = U1 + 2 · Δu (4)
u2 = U2-2−Δu (5)

  Here, since U1 = U2, the expression (5) can be transformed as the following expression (6), and the difference between the effective values u1 and u2 can be changed from the following expressions (4) and (6) ( It can be shown as 7).

u2 = U1-2−Δu (6)
u1-u2 = (U1 + 2 · Δu) − (U1-2 · Δu) = 4 · Δu (7)

  Since the amplitude of the rectangular wave is VDC / 2 and the difference between the effective values u1 and u2 is the difference between the effective values S1 and S2, the pulse adjustment width Δφ is obtained by the following equations (8) and (9). be able to.

Δu = (S1-S2) / 4 (8)
Δφ = Δu / (VDC / 2) = (S1-S2) / (2 · VDC) (9)

In the example shown in FIG. 5, the adjustment based on the pulse adjustment width Δφ is performed before and after the pulse width. Therefore, the pulse adjustment width for one cycle of electrical angle is 2 · Δφ. The adjustment unit 53 adds the pulse adjustment width 2 · Δφ to the pulse with the smaller effective value in addition to the pulse with the larger effective value among the positive (u1) and negative (u2) rectangular wave pulses. The pulse width is adjusted by subtracting from. From equation (9), the pulse adjustment width 2 · Δφ is obtained by dividing the effective value difference (S1-S2) by the peak value (peak-to-peak) (= VDC) of the inverter drive signal. It is.

A specific example will be described with reference to FIG. The sine wave waveform in the figure shows the fundamental wave component B1 in the sine wave PWM control. This fundamental wave component B1 has an offset a. The rectangular wave in the figure indicates the inverter drive signal D2 in the rectangular wave control. In this example, sine wave PWM control is performed in the period T _n−1 , and control is switched from sine wave PWM control to rectangular wave control in the middle of the period T _n . This switching timing is determined as described above. The period T _n−1 , the period T _n , and the period T _{n + 1} are target periods and may have different lengths. However, the electrical angle of each cycle is 2π. The phase θ in the figure is a value that makes the start of each period zero.

The determination of the switching timing is performed in all cycles. Thus, for example, when switched in the section of the period T _n, on the basis of the operation timing of the operation pulse adjustment width in the previous period T _n-1 interval, operation timing of the rectangular wave is determined. The switching instruction is given in an arbitrary section, but since the timing of the rectangular wave is always calculated based on the immediately preceding effective value, stable control with good followability is possible.

Hereinafter, a case where the control is switched to the rectangular wave control in the section of the cycle T _n will be described as an example. First, in the section of the cycle T _n−1, the positive effective value S1 _n−1 and the negative effective value S2 _n−1 by the sine wave PWM control are calculated. Then, the phase adjustment width Δθ is calculated based on the following equation (10) based on the above equation (9). In this example, the phase adjustment width Δθ is provided on one side of the pulse of the rectangular wave, which is different from the example shown in FIG. 5 and equation (9) in which the pulse width adjustment width Δφ is provided on both sides of the pulse. Yes. Accordingly, the divisor “2” is not included in the following formula (10). Thus, when applying the present invention, those skilled in the art can easily understand that the definition of the adjustment range can be changed as appropriate.

Δθ _n-1 = (S1 _n-1 -S2 _n-1 ) / VDC (10)

In the case of rectangular wave control with a duty of 50%, the center T _n-1 / 2 of the cycle T _n-1 is the switching timing. Accordingly, if switching to the rectangular wave control is performed in the section of the cycle T _n−1 , the switching timing θb _n−1 is obtained by the following equation (11).

θb _n-1 = (T _n-1 / 2) + Δθ _n-1 (11)

  Actually, the phase adjustment width Δθn−1 calculated in the section of the cycle Tn−1 is used when switching to the rectangular wave control in the section of the cycle T. The switching timing in this case is obtained by the following equation (12).

θb _n = {(T _n−1 / 2) + Δθ _n−1 } · (T _n / T _n−1 ) (12)

That is, for the switching timing determined in the previous period T _n-1 of the section, by multiplying the ratio of the previous period T _n-1 and the current period T _n, switching timing in the interval of the current cycle Tn To decide. If the equation (12) is modified, the following equation (13) is obtained, and it can be easily understood that the same shape as the above equation (11) is obtained.

θb _n = (T _n / 2) + Δθ _n (13)

Hereinafter, the motor control method by the motor control apparatus according to the present invention will be described with reference to the flowcharts shown in FIGS.
The arithmetic unit 10 first acquires the current of the three-phase stator coil (# 1). Filtering processing calculation is performed on the acquired motor current of the three-phase stator coil (# 2). Then, the current of the three-phase stator coil for one electrical angle cycle is calculated (# 3). In this manner, the motor current flowing through the stator core of the motor 1 and the motor current are detected (current detection step).

The motor current is acquired by a procedure as shown in FIG. The computing unit 10 acquires the electrical angle λ _i output from the resolver 11 at every predetermined sampling time (# 11). The obtained electrical angle λ _i is compared with the electrical angle λ _i-1 at the time of the previous sampling (# 12). If the electrical angle λ _i is larger, the electrical angle is increased, so one cycle It is determined that it has not reached. Therefore, the acquired motor current values are integrated (# 13). When the electrical angle λ _i is compared with the electrical angle λ _i-1 (# 12), and the electrical angle λ _i is smaller, the electrical angle exceeds 2π (= 360 °). Therefore, although the motor current for one electrical angle cycle is calculated based on the current value of the motor current accumulated so far, the offset current can be calculated together (# 14). The current value accumulated so far is reset.

  Again referring to FIG. When the motor current is acquired in this way, the calculation unit 10 performs a PI control calculation and calculates a voltage command to the inverter 2 (# 4). The arithmetic unit 10 obtains the number of rotations of the motor 1 based on the input from the resolver 11 and determines the control method. In other words, depending on the operating state of the motor 1, it is selected which of the first control process and the second control process to be described later to output the inverter drive signal generated (selection process).

  When the control method is sinusoidal PWM control (# 5), the PWM signal output unit 41 outputs the first drive signal D1 as the inverter drive signal (# 6). Process # 6 controls the voltage applied to the motor 1 by sinusoidal pulse width modulation based on the first voltage command signal calculated using the motor current and the offset current, and sends the first drive signal D1 to the inverter 2 This corresponds to the output first control step.

  When the control method is rectangular wave control (# 5), particularly when the control method is changed from sine wave PWM control to rectangular wave control, the effective value at the time of sine wave PWM control is calculated as described above. (# 7). That is, the effective value of one cycle of the AC voltage applied to the motor 1 is calculated separately on the positive side and the negative side with reference to the voltage at the neutral point N of the motor 1 (effective value calculating step). The effective value is calculated based on the first voltage command signal output from the calculation unit 10 to the PWM signal output unit 41 or the rectangular wave signal output unit 51 without directly observing the AC voltage output from the inverter 2. Can do.

  When the effective value is calculated, the pulse width of the rectangular wave is adjusted as described above based on the effective value (# 8). That is, when the process of generating the inverter drive signal is changed from the first control process to the second control process, the pulse width of the rectangular wave is adjusted (adjustment process). And the rectangular wave signal output part 51 outputs the 2nd drive signal D2 as an inverter drive signal (# 9). Processes # 7 to # 9 control the applied voltage to the motor 1 by phase adjustment of a rectangular wave based on the second voltage command signal calculated using the motor current and the offset current, and the second drive signal D2 This corresponds to the output second control step.

  As described above, when the control method is switched from sine wave PWM control to rectangular wave control, control switching is performed by adjusting the pulse width of rectangular wave control based on the effective voltage value during sine wave PWM control. The disturbance of the motor current at the time can be suppressed. FIG. 9 shows a simulation result of the motor current when the present invention is applied. iu, iv, and iw indicate motor currents of the three-phase stator coils of the motor 1, respectively. Vu, Vv, and Vw indicate drive voltages for the three-phase stator coil. As can be seen from the simulation results, no disturbance is observed in the motor current when switching from PWM control to rectangular wave control.

The block diagram which shows typically the structural example of the motor control apparatus which concerns on this invention Waveform diagram schematically showing an example of an inverter drive signal and its fundamental wave component by sine wave PWM control Waveform diagram schematically showing the fundamental wave component with DC offset Waveform diagram schematically showing an example of an inverter drive signal and its fundamental wave component by rectangular wave control Waveform diagram schematically showing an example of adjusting the pulse width of an inverter drive signal by rectangular wave control Waveform diagram showing an example of pulse width adjustment of an inverter drive signal by rectangular wave control when switching from sine wave PWM control to rectangular wave control The flowchart which shows the procedure of the motor control method which concerns on this invention The flowchart which shows the procedure of the motor control method which concerns on this invention Waveform diagram showing simulation results of motor current when the present invention is applied

1: Motor 2: Inverter 3, 3v, 3w: Current detection unit 4: First control unit 41: PWM signal output unit 5: Second control unit 51: Rectangular wave output unit 52: Effective value calculation unit 53: Adjustment unit 6 : Selection unit 11: Resolver (rotation detection unit)

Claims (2)

An inverter that converts a DC voltage into an AC voltage for driving the motor based on the inverter drive signal;
A current detector for detecting a motor current flowing through the motor;
Based on a voltage command signal on which an offset current calculated using the motor current is superimposed, a voltage applied to the motor is controlled by sinusoidal pulse width modulation, and a first drive signal for driving the inverter is output. A first control unit;
Based on the voltage command signal offset current that will be computed is superimposed using the motor current, the voltage applied to the motor is controlled by the phase adjustment of the rectangular wave, and outputs a second driving signal for driving the inverter A second control unit;
A selection unit for selecting the inverter drive signal for driving the inverter according to the operation state of the motor from the first and second drive signals;
An effective value calculation unit for calculating an effective value for one period of an electrical angle of an AC voltage applied to the motor, by dividing the positive value and the negative value on the basis of the voltage at the neutral point of the motor;
When the inverter drive signal is switched from the first drive signal to the second drive signal by the selection unit, the positive-side effective value and the negative-side effective value of the first drive signal in the immediately preceding cycle The pulse adjustment width obtained by dividing the difference by the peak value of the inverter drive signal to the inverter is added to the pulse with the larger effective value of the square wave pulses on the positive side and the negative side. An adjustment unit for adjusting a pulse width of the rectangular wave generated by the second control unit by subtracting from a pulse having a smaller effective value ;
A motor control device comprising: A voltage conversion step of applying an inverter drive signal to the inverter and converting the DC voltage into an AC voltage for driving the motor;
A current detection step of detecting a motor current flowing through the motor;
Based on a voltage command signal on which an offset current calculated using the motor current is superimposed, a voltage applied to the motor is controlled by sinusoidal pulse width modulation, and a first drive signal for driving the inverter is output. 1 control process;
Based on the voltage command signal offset current that will be computed is superimposed using the motor current, the voltage applied to the motor is controlled by the phase adjustment of the rectangular wave, and outputs a second driving signal for driving the inverter A second control step;
A selection step of selecting, from the first and second drive signals, the inverter drive signal for driving the inverter according to the operating state of the motor;
An effective value calculation step of calculating an effective value for one period of an electrical angle of an AC voltage applied to the motor by dividing the effective value into a positive side and a negative side based on a voltage at a neutral point of the motor;
When the step of generating the inverter drive signal is changed from the first control step to the second control step, the positive-side effective value and the negative-side effective value of the first drive signal in the immediately preceding cycle are changed. The pulse adjustment width obtained by dividing the difference from the value by the peak value of the inverter drive signal to the inverter is the pulse having the larger effective value among the square wave pulses on the positive side and the negative side. In addition, an adjustment step of adjusting a pulse width of the rectangular wave generated in the second control step by subtracting from a pulse having a smaller effective value ,
A method for controlling a motor comprising:

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