JP5263067B2 - Inverter fault detection system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To accurately detect open failure of a semiconductor switching element for power which constitutes an inverter for driving an AC motor that is controlled by switching two or more control modes, even when its control mode is frequently switched, within a short interval. <P>SOLUTION: A controller gets a filter current value regarding each phase which is obtained by low-pass filtering of phase currents, separately for each control mode (S120). When the absolute value of the filter current value exceeds a determined value, open failure of the switching element of a relevant phase is detected (S200). With respect to switching of the control mode, the filter current value is kept, when a mode other than the target control mode is selected (S160); and at switching to the target control mode, offset modification is carried out on the filter current value so as to prevent misdetection of the open failure (S140). <P>COPYRIGHT: (C)2011,JPO&amp;INPIT

Description

This invention relates to a fault detection system of the inverter, and more particularly, relates to the detection of an open fault of the power semiconductor switching elements constituting the inverter.

Power semiconductor switching elements constituting the inverter that drives and controls the AC motor (hereinafter, simply referred to as "switching element") of the fault detection has been proposed. For example, JP-2005-094873 Patent Publication (Patent Document 1), it integrates the current for each phase of the inverter, open-circuit failure in accordance with the accumulated value during the motor rotating two periods (failure to remain off) It describes a configuration for detecting a. Specifically, the integrated value, for positive (greater than 0) the failure of the upper switching element is described to be judged to be normal if a failure, 0 of the lower side switching element in the case of a negative ing.

Another example of a method for detecting open-circuit failure of the inverter, Japanese 2002-136147 (Patent Document 2), the diagnosis of the open fault by measuring the current flowing to the shunt resistor connected to an inverter it has been described. Further, Japanese Patent 2006-320176 (Patent Document 3), established a resistor divider between the inverter DC terminals, forming a diagnostic point connection point between the voltage dividing resistors connected with any AC terminal as well as it has been described to detect short circuit faults (leave becomes a failure on) and open-circuit failure by comparing the potential of this diagnostic point with a reference value made from the voltage of the smoothing capacitor.

JP 2005-094873 JP JP 2002-136147 JP JP 2006-320176 JP

In the control of the AC motor, a pulse width modulation control (PWM) control or, such as the rectangular wave voltage control, being carried out be switched in accordance with the operation state of the plurality of control modes AC motor. In such a case, the current behavior each control mode is changed, the integral value of the phase currents, such as described above, or the detection of an open fault based on filter current value processed by the low-pass filter, the current integral it is preferable to determine the value or filter current value to the control mode for each.

However, when the control mode is frequently switched in a short time, electrical cycle of the AC motor, i.e. the control mode within the one period of the phase current is not passed could switched. In this case, in the configuration of Patent Document 1 that performs open-circuit failure based on the phase current integral value of one period becomes 0 on success, there is a possibility that it becomes impossible to detect the accurate open fault.

As for the Patent Documents 2 and 3, for the method for the control mode is to detect the open-circuit failure in response to the case where frequently switched within a short time, not described.

The present invention was made to solve the above problems, the object is a semiconductor power constituting an inverter for driving an AC motor in which a plurality of control modes are switched by control of the present invention the open-circuit failure of the switching element, is also possible to detect accurately when the control mode is frequently switched in a short time.

The present invention Inverter failure detecting apparatus according to, a failure detection device of a plurality of phases of the inverter to control the AC motor, a current detecting means for detecting a current of each phase of the inverter, the operating state of the AC motor a plurality of mode selection means for selecting one of the control modes, by low-pass filtering the detected phase current by the current detecting means, separately phase filter current values ​​for each control mode in accordance with and filtering means for calculating a, if the filter processing filter current value calculated by means exceeds a predetermined judgment value, the failure detection means for detecting the open-circuit failure of the corresponding phase of the switching elements of the inverter If, when the control mode is switched, the phase of the filter current value corresponding to the control mode after the switching, the absolute value of the predetermined range And a correction means for correcting to decrease within.

Preferably, the correction means decreases the absolute value of the filter current value is greater than the maximum offset value corresponds to the integral value of the half cycle of the maximum rated current of each phase current, the absolute value by the maximum offset value while modifying the filter current to, when the absolute value of the filter current is less than or equal to the maximum offset value sets the filter current value to zero.

In this way, when the control mode is switched before the electrical cycle of AC motor has elapsed, the open failure filter current value even when no occurred (current integral substantially the equivalent) is addressed to offset not zero occurs, the initial value of the current filter value can be corrected offset for each control mode switching. Therefore, when the control mode is frequently switched in the short time, can prevent erroneous detection of an open fault due to the filter current is continuously increased. In particular, compared to the case of zero-cleared filter current value for each switching of the control modes, you can miss the open failure is low.

More preferably, the maximum offset value, as a small value in accordance with increase in the rotational speed of the AC motor is variably set according to the rotational speed of the AC motor.

In this manner, in that the in correspondence seeking filter current value by the low-pass filtering, it becomes possible to properly detect the open-circuit failure of the switching element.

Also preferably, the current detection circuit includes a current sensor arranged in each phase, except for predetermined one phase of a plurality of phases, and a current calculation means for calculating a current having a predetermined single-phase from the output of the current sensor . Then, the filter processing unit, while calculating a filter current value based on the calculated value by the current calculation unit is in a predetermined single-phase, the filter current value based on a value detected by the current sensor in each phase, except for certain 1-phase It is calculated. Further, the determination value at a given one phase is set larger than the determination value in each phase except one predetermined phase.

Thus, without providing a current sensor in each phase, for a given 1-phase in the structure is omitted placement of the current sensor, it is possible to accurately detect the open-circuit failure of the power switching element.

According to the present invention, even when the open failure of the power semiconductor switching devices constituting an inverter for driving an AC motor in which a plurality of control modes are switched by control, the control mode is frequently switched in a short time it can be accurately detected.

Configuration example of a motor drive system fault detection device of the inverter according to the present embodiment is applied is a block diagram showing the. Is a diagram illustrating schematically the control mode of AC motor in the motor drive system shown in FIG. Is a flowchart illustrating the selection method of the control mode shown in FIG. It is a waveform diagram illustrating the behavior between the phase current and the filter current at the time of normal. Behavior of the phase current and the filter current at the open failure is a waveform diagram showing a. It is a waveform diagram for explaining the problems of the open-circuit failure detection in case of setting the filter current value (current integral) in each control mode. It is a waveform diagram illustrating the behavior between the phase current and the filter current at the inverter fault detection system according to this embodiment. It is a schematic block diagram showing a control configuration of an inverter failure detection according to the present embodiment. Is a table illustrating the setting of the phase current and the filter current of each control mode. It is a flowchart illustrating a series of control processes in the inverter fault detection according to the present embodiment. It is a flowchart illustrating the details of the offset correction process shown in FIG. 10.

It will be described in detail with reference to the drawings, embodiments of the present invention. Incidentally, the same or corresponding portions in the following figures are denoted by the same reference numerals description thereof will not be repeated in principle.
(The entire configuration of an electric motor control)
Figure 1 is a block diagram showing a configuration example of a motor drive system fault detection device of the inverter according to the present embodiment is applied.

Figure 1 is an overall configuration diagram of a motor drive system according to the embodiment of the present invention.
1, a motor drive system 100 includes a DC voltage generating unit 10 #, a smoothing capacitor C0, an inverter 14, an AC motor M1, and a control device 30.

AC motor M1 is, for example, electric vehicles (hybrid vehicles, electric vehicles and shall refer to an automobile for generating a vehicle driving force by electric energy such as a fuel cell vehicle) to driving wheels for generating a torque for driving the which is the driving electric motor. Alternatively, the alternating-current motor M1 may be configured to have a function of a generator driven by an engine may be configured to have both the functions of the electric motor and the electric generator. Furthermore, AC motor M1 may operate as a motor for the engine, for example, may be incorporated in a hybrid vehicle as such may start the engine.

DC voltage generating unit 10♯ includes a DC power supply B, system relays SR1, SR2, a smoothing capacitor C1, and a converter 12.

DC power supply B is typically constituted by a power storage device such as a secondary battery or an electric double layer capacitor, such as a nickel hydride or lithium ion. DC current Ib to be the DC voltage Vb and output DC power supply B outputs are respectively detected by the voltage sensor 10 and current sensor 11.

System relay SR1 is connected between a positive terminal and a power line 6 of the DC power supply B, system relays SR2 is connected between the negative terminal and the ground wires 5 of the DC power supply B. System relays SR1, SR2 are turned on / off by a signal SE from control device 30.

Converter 12 includes a reactor L1, the semiconductor switching elements Q1, Q2 for power, and diodes D1, D2. Semiconductor switching elements Q1 and Q2 for power are connected in series between power line 7 and earth line 5. On and off of power semiconductor switching elements Q1 and Q2 is controlled by switching control signals S1 and S2 from control device 30.

In the embodiment of the present invention, the power semiconductor switching elements (hereinafter, simply referred to as "switching element") as a, IGBT (Insulated Gate Bipolar Transistor),
MOS power (Metal Oxide Semiconductor) transistor or can be used a power bipolar transistor or the like. For switching elements Q1, Q2, antiparallel diodes D1, D2 are arranged. Reactor L1 is connected between the switching elements Q1 and Q2 of the connection node and the power line 6. Further, smoothing capacitor C0 is connected between power line 7 and earth line 5.

Inverter 14 are provided in parallel between power line 7 and earth line 5, the U-phase upper and lower arms 15, V-phase upper and lower arms 16, and W-phase upper and lower arms 17. Phase upper and lower arms is composed of switching devices connected in series between power line 7 and earth line 5. For example, U-phase upper and lower arms 15 includes switching elements Q3, Q4, V-phase upper and lower arms 16 includes switching elements Q5, Q6, W-phase upper and lower arms 17 includes switching elements Q7, Q8. Further, the switching elements Q 3 -Q 8, anti-parallel diodes D3~D8 are connected. On and off of the switching element Q3~Q8 is controlled by switching control signals S3~S8 from control device 30.

Typically, AC motor M1 is a permanent magnet synchronous motor of three phases, U, V, one of the three coils of the W-phases each connected in common to a neutral point. The other end of each phase coil is connected to an intermediate point of the switching elements of each phase upper and lower arms 15 to 17.

Converter 12, the voltage step-up operation, (this DC voltage equivalent to the input voltage to the inverter 14, hereinafter referred to as "system voltage") a DC voltage VH obtained by boosting the DC voltage Vb supplied from DC power supply B to inverter 14 supplied to. More specifically, in response to switching control signals S1, S2 from control device 30, Q2 of the on period of the ON period and the switching element of the switching element Q1 (or a period during which both switching elements Q1, Q2 is turned off ) are provided alternately, the step-up ratio becomes one corresponding to the ratio of these oN periods. Alternatively, if the fixed switching elements Q1 and Q2 on and off, can also be a VH = Vb (voltage step-up ratio = 1.0).

Further, the converter 12, during step-down operation to charge the DC power source B by lowering the DC voltage supplied from inverter 14 via smoothing capacitor C0 VH (system voltage). More specifically, in response to switching control signals S1, S2 from control device 30, a period in which only switching element Q1 is turned on, a period in which both switching elements Q1, Q2 is turned off (or, Q2 of the switching element the oN period) and are alternately provided, the step-down ratio is in accordance with the duty ratio of the oN period.

Smoothing capacitor C0 smoothes the DC voltage from the converter 12, and supplies the DC voltage smoothed to inverter 14. Voltage sensor 13 detects the voltage across smoothing capacitor C0, i.e., detects a system voltage VH, and outputs the detected value to control unit 30.

Inverter 14, torque command value of AC motor M1 is in the case of positive (Trqcom> 0), the DC voltage from the smoothing capacitor C0 in response to switching control signals S3~S8 from control device 30 to be supplied, the switching by the switching operation of the device Q3~Q8 converts the DC voltage into an AC voltage for driving AC motor M1 to output a positive torque. Further, inverter 14, if the torque command value of AC motor M1 is 0 (Trqcom = 0), the switching operation in response to switching control signals S 3 to S 8, torque converts the DC voltage into an AC voltage 0 drives AC motor M1 so as to. Thus, AC motor M1 is driven to generate zero or positive torque designated by torque command value Trqcom.

Further, during regenerative braking of the electric vehicle motor driving system 100 mounted thereon, torque command value Trqcom of AC motor M1 is set to a negative (Trqcom <0). In this case, the inverter 14, the switching control signal by the switching operation in response to S 3 to S 8, the AC voltage generated by AC motor M1 into a DC voltage, the converted DC voltage (system voltage) to the smoothing capacitor C0 supplied to the converter 12 through the. The regenerative braking here refers to and includes braking accompanied by regenerative power generation when a foot brake operation by a driver of the electric vehicle, but does not operate the foot brake, regenerative, by releasing the accelerator pedal during running while the power generation includes decelerating the vehicle (or stop of acceleration).

Current sensor 24 detects a motor current flowing through AC motor M1 (phase current), and outputs the detected value to control unit 30. Incidentally, three-phase currents Iu, Iv, since the sum of instantaneous values ​​of Iw is 0, the current sensor 24 as shown in Figure 1 is of two phases motor currents (e.g., V-phase current Iv and W-phase current Iw) It suffices arranged to detect. Alternatively, it is also possible to arrange the current sensor 24 in each phase.

Rotation angle sensor (resolver) 25 detects a rotor rotation angle θ of AC motor M1, and sends the rotation angle θ obtained by the detection to the controller 30. In the control device 30, the rotational speed of the AC motor M1 based on rotational angle θ (rotation speed) Nmt and angular velocity ω (rad / s) can be calculated. Incidentally, the rotation angle sensor 25, by calculating directly from the motor voltage and current at a rotation angle θ of the controller 30, may be omitted placement.

Controller 30 is constituted by an electronic control unit containing a CPU (not shown) (Central Processing Unit) and a memory, based on a map and a program stored in the memory, it performs calculation processing using the value detected by each sensor configured. Alternatively, at least a portion of the ECU80 may be configured to execute predetermined numerical and logic operation processing by hardware such as an electronic circuit.
(Description of control mode)
It will be described in more detail control of AC motor M1 by control device 30.

Figure 2 is a diagram illustrating schematically the control mode of AC motor M1 in the motor drive system 100 according to an embodiment of the present invention.

As shown in FIG. 2, in the motor drive system 100 according to an embodiment of the present invention, control of AC motor M1, that is, the power conversion in inverter 14, using switch three control modes.

Sine wave PWM control is intended to be used as a general PWM control, ON and OFF of each phase upper and lower arm elements are controlled in accordance with voltage comparison between sinusoidal voltage command and a carrier wave (typically, triangular wave) . As a result, a high-level period corresponding to the ON period of the upper arm device, for a set of low-level period corresponding to the ON period of the lower arm element, the duty so that the fundamental wave component within a certain period is a sine wave There is controlled. As is well known, in the sinusoidal wave PWM control the amplitude of the voltage command sinusoidal is limited to the scope of the following carrier amplitude, fundamental of the voltage applied to the AC motor M1 (hereinafter, simply referred to as "motor applied voltage") It can not be increased only a component to about 0.61 times that of the inverter of the DC link voltage. Hereinafter, in this specification, the DC link voltage of inverter 14 (i.e., system voltage VH) the ratio of the fundamental wave component of the motor voltage applied to (the line voltage) (effective value) is also referred to as "modulation factor".

On the other hand, in the rectangular wave voltage control, within the fixed period, the ratio of the high-level period and low level period is 1: apply 1 rectangular wave for one pulse to the AC motor M1. Thus, the modulation rate is increased to 0.78.

Overmodulation PWM control is to the amplitude of the voltage command (sine wave component) perform PWM control similar to the above sinusoidal PWM control with greater range than the carrier wave amplitude. In particular, it is possible to increase the fundamental wave component by distorting the voltage command from the original sinusoidal wave (amplitude correction), to increase the modulation factor to the range from the highest modulation factor of 0.78 for a sine-wave PWM control mode can. In overmodulation PWM control, the amplitude of the voltage command (sine wave component) is larger than the carrier wave amplitude, the line voltage applied to AC motor M1 is a voltage distorted rather than a sine wave.

In AC motor M1, the rotation speed or output torque increases the induced voltage with increasing the driving voltage (motor required voltage) becomes higher required. Boosted voltage or by the converter 12, the system voltage VH needs to be set higher than this motor required voltage. On the other hand, the boosted voltage i.e. by the converter 12, the system voltage VH exists a limit value (VH maximum voltage).

Thus, in accordance with the operation state of AC motor M1, it controls the amplitude and phase of the motor applied voltage (AC) by the feedback of the motor current, the PWM control mode by sine wave PWM control or overmodulation PWM control, and rectangular wave voltage any of the control modes is selectively applied. In the rectangular wave voltage control, the amplitude of the motor applied voltage is fixed, based on a deviation between the actual torque value and the torque command value, the torque control is executed by phase control of the rectangular wave voltage pulse.

Figure 3 is a flowchart illustrating the selection method of the control mode.
As shown in the flowchart of FIG. 3, the high-level ECU, not shown, receives the torque command value Trqcom of AC motor M1 based on the vehicle required output according to the accelerator opening degree is calculated (step S100), control device 30 based on the preset map or the like, the torque command value Trqcom and motor required voltage from the rotational speed of the AC motor M1 (induced voltage) is calculated (step S110), further, the maximum required motor voltage and the system voltage VH according to the relationship between the value (VH maximum value), and apply one of the rectangular wave voltage control mode and the PWM control mode to determine whether to perform the motor control (step S120). Basically, the modulation factor corresponding to the voltage command value control mode is selected can be realized. In PWM control mode, in accordance with the modulation rate, one is selected sine wave PWM control and the overmodulation PWM control. According to the control flow according to the operating conditions of AC motor M1, the proper control mode is applied from among a plurality of control modes shown in FIG.

Next, each phase of the motor current supplied from inverter 14 to AC motor M1 (hereinafter, simply phase current also referred to) detection technique of open-circuit failure of the switching element Q3~Q8 based on will be explained.

Hereinafter, in the W-phase is described as an example an open failure detection of the switching element Q7 (upper arm) and Q8 (lower arm).

4 shows a phase current of W-phase in the normal (W-phase current) Iw of the waveform is shown.
During normal, W-phase current Iw, a sinusoidal current cycle corresponding to the electrical cycle of AC motor M1. Thus, the filter current Iwf obtained by low-pass filtering the W-phase current Iw, returns to 0 during one period of the motor current. As a result, not a filter current IWF continues to rise, against the decision value IJD for open fault detection, | Iwf | <Ijd is maintained.

In contrast, in FIG. 5, the waveforms in the case where open-circuit failure in the switching element Q8 of the lower arm has occurred is shown.

Referring to FIG. 5, by open-circuit failure in the lower arm is generated, W-phase current Iw, only the positive direction of current flow, nature, and Iw = 0 in the period in which the negative current flows.

As a result, the filter current Iwf which is low-pass filtering, without returning to zero even after current one period, gradually increases. As a result, when the plurality of cycles has elapsed, by filtering current Iwf reaches the determination value IJD, open-circuit failure in the W-phase is detected to have occurred. Furthermore, since it is IWF> 0, it can also be specified that the open-circuit failure occurs in the lower arm element.

Although not shown, when the open-circuit failure in the upper arm device occurs, becomes opposite polarity from that of FIG. 5, W-phase current Iw does not flow only in the negative direction. As a result, the filter current IWF is gradually absolute value increases in the negative direction, | IWF |> they become a Ijd, it is possible to detect that the open-circuit failure occurs in the same manner W phase. Furthermore, since it is IWF <0, it can also identify the open-circuit failure occurs in the upper arm device.

Here, in the motor drive system shown in FIG. 1, as described in FIG. 2 and FIG. 3, a plurality of control modes are selectively applied. Then, between control modes, to come different degree of overlapping high-frequency component of the current behavior, in particular the phase currents, as described in FIG. 4 and FIG. 5, corresponds to the integration process, based on the filter value of the phase current for even open-circuit failure detection is preferably performed separately for each control mode. That is, the filter current and the determination value also, more to the individual control mode for each, it can be more accurately detected open-circuit failure of the switching element.

In this embodiment, when the sine wave PWM control (hereinafter, PWM mode) and the time of the rectangular wave voltage control (hereinafter, a rectangular wave mode) for each, and the filter current value and the determination value separately set, the switching element and it detects the open-circuit failure. Note that the overmodulation PWM control, basically, because it is intended to be temporarily selected when the control mode transition between the sinusoidal wave PWM control and the rectangular wave voltage control, run for open-circuit failure of the switching element and those that do not.

Figure 6 is a problem with open-circuit failure detection in case of setting the filter current value (current integral) in each control mode is shown.

In particular, in Figure 6, as a case where the offset amount left in the filter current value is maximized, if the electrical half cycle every control mode of AC motor M1 is switched and the like. This offset, without short-circuit fault has occurred, which is generated by the control mode switching in one cycle way, as described below, it causes the open failure misdetection.

In Figure 6, W-phase current Iw is a normal waveform, open-circuit failure in the W-phase is to be understood that not occurred. Therefore, when the PWM mode is continued throughout the period of the phase current, by filtering current Iwf returns to zero, will not be open-circuit failure is detected.

However, after time t1, the control mode for each half cycle of the phase current is switched between the PWM mode and the rectangular wave, the switched time t1 control mode to the rectangular-wave mode, until the time t2 to return again to the PWM during the filter current Iwf is held in the switching immediately prior to the value (value at time t1). Then from the time t2, it is updated again filter current IWF. That is equivalent to that the integral of the current value is resumed.

As shown in FIG. 6, assuming a worst case erroneous detection, for certain polarity period of the phase current (the period in FIG. 6 the phase current is negative), the control mode is the rectangular-wave mode, the opposite polarity in the period (the period in FIG. 6 phase current positive), the control mode is a PWM mode. In this case, the filter current Iwf of PWM is that it does not be updated in the negative direction, depending on the time, so that the continuously increasing. As a result, actually even though the open-circuit failure does not occur, | IWF | is by reaching the IJD, open-circuit failure of the W-phase (lower arm) from being erroneously detected.

As in FIG. 6, in the control mode is mode changes case synchronously with every half cycle the polarity of the phase current, the offset to be left in the filter current value to be a 0 is maximized with respect to the normal phase current. That is, the maximum value of the offset (hereinafter referred to as the maximum offset amount) corresponds to the filter current value at half cycle elapse when the phase current is maximum rated current (or current integral of a half cycle) It will be.

The inverter fault detection according to the present embodiment, by modifying the filter current value within the range of the maximum offset amount for each switching of the control mode, prevent erroneous detection of an open fault is achieved.

Referring to FIG. 7, PWM mode until the time t1 is selected, between times t1 to t2, as in the case of FIG. 6, period square wave mode of a half cycle of the phase current is selected. Then, at time t2 and later, it is selected again the PWM control mode.

Thus, between times t1~t2, similar to FIG. 6, the filter current Iwf will be held at the value of time point t1. Then, at time t2, when the updating of the filter current value in the PWM mode is resumed, in the range of offset correction amount Icr which is determined corresponding to the maximum offset amount, the filter current value Iwf is corrected.

As a result, the normal short circuit fault has not occurred, at time point t1, even if occurred phase current half-cycle of the offset by modifying the offset when the PWM mode is resumed from the time t2, the filter current Iwf is returned to near zero again. If this result, the normal phase currents after, never filter current Iwf reaches continuously increased to the judgment value IJD. That is, it is possible to prevent erroneous detection. If, as in FIG. 6, as the rectangular wave mode is selected during times t3 to t4, at time t4 the PWM mode is resumed, the same offset correction and the time t2, the filter current Iwf 0 because back in the vicinity, erroneous detection is understood that it is not generated.

On the other hand, when an abnormality that open failure has occurred, be modified similarly offset and normal at time t2, by continuously filtering the current value Iwf rises again, open-circuit failure eventually it is possible to detect the.

Figure 8 is a schematic block diagram showing a control configuration of an inverter failure detection according to the present embodiment. Note that the respective functional blocks shown in FIG. 8, may be constituted by a circuit (hardware) having the function corresponding to the block, the controller 30 executes a software processing in accordance with a preset program it may be realized by. Further, in FIG. 8, but illustrating the configuration for the open-circuit failure detection of one phase (W-phase), each phase, i.e., the U-phase and V-phase, confirmed to that same structure is provided described.

Referring to FIG 8, the inverter fault detection apparatus 101 according to an embodiment of the present invention includes a filter processing unit 110, an offset correction unit 120, a determination value setting unit 130, a detection unit 140.

Filter processing unit 110, the W-phase current Iw detected by the current sensor 24 is filtered according to the following equation (1). That filtering section 110 by low-pass filtering the current detection value Iw, calculates the filter current IWF.

Iwf = {Iw-Iwf (0)} · fa + Iwf (0) ... (1)
(1) In the formula, IWF (0) indicates the previous value of the filter current IWF. Then, smoothing coefficient fa has a value ranging from 0 to 1.0, the time constant of the filter as fa is close to 0 is increased, the time constant of the filter closer to fa 1.0 is small. That is, smoothing coefficient fa is determined according to the time constant should be given to the filter processing unit 110. The time constant of the low pass filter process is basically is preferably set to be equivalent to determining the current integral of the W-phase current Iw by the filter unit 110.

Incidentally, as shown in FIG. 1, the current sensor 24 is not necessarily provided for each phase. That is, the current sensor 24 only to each of the two phases other than the predetermined one of the three phases are arranged, for the remaining one phase, since the sum of each phase current instantaneous value is zero, determined by the calculation it is also possible. For example, in the arrangement of FIG. 1, for the U-phase current Iu, it can also be obtained in accordance with the following equation (2).

Iu = - (Iv + Iw) ... (2)
The U-phase, (2) based on the phase currents Iu calculated by equation, the filter processing unit 110 calculates the filter current value (IUF). Thus, the phase current sensor 24 is disposed, and for any of the non-placement of the phases can be applied to the common open fault detection shown in FIG. However, the erroneous detection due to the influence of the error is a concern when seeking phase current through calculation. Therefore, for each control mode, as compared with the determination value in the phase current sensor 24 is disposed, it is preferable to increase the determination value for the phase of the non-placement.

The filter processing unit 110, a control signal CMD from the control mode selection section 35 are input. Control mode selection section 35 in accordance with the control processing shown in the flowchart of FIG. 3, in accordance with the operation state of AC motor M1, to select one of the plurality of control modes shown in FIG. Control signal CMD indicates the control mode selected by the control mode selection section 35.

Determination value setting unit 130 in accordance with a control signal CMD, and outputs the detection unit 140 sets a determination value corresponding to the control mode selected.

Referring to FIG. 10, the filter processing unit 110, for each phase of the filter current value is set to each control mode. For example, in response to the PWM mode, the filter current IUF (1) for each of the U-phase ~W phase, Ivf (1), Iws (1) is set. Further, in response to a square wave mode, for each of the U-phase ~W phase, filter current Iuf (1), Ivf (1), separately from the Iws (1), the filter current Iuf (2), Ivf (2), Iws (2) has been set.

As mentioned above, during PWM mode selection, filtering section 110, (1) filter current IUF (1) according to equation, IVF (1), while sequentially updating the IWF (1), the filter current Iuf (2), Ivf (2), Iwf (2) holds. On the other hand, in the rectangular wave mode selection, filtering section 110, (1) filter current IUF (2) according to equation, IVF (2), while sequentially updating the IWF (2), the filter current IUF (1), Ivf (1), Iwf (1) holds.

Further, the determination value is preferably set to each control mode also IJD. During PWM mode application example, the filter current IUF (1), IVF (1), IWF while (1) is compared with the determination value IJD (1), at the time of the rectangular wave mode selection, the filter current IUF (2) , Ivf (2), Iwf (2) is compared with the determination value IJD (2).

Referring again to FIG. 8, the offset correction unit 120 based on the control signal CMD, and detects the switching of the control mode, at the time of switching of the control mode, the initial filter current value corresponding to the control mode after the switching values ​​and offset correction as shown in FIG.

Offset correction amount Icr in this case is determined according to the following equation (3).
Icr = Ifmax · Nn♯ / Nm ... (3)
(3) In the formula, Ifmax is when the rotational speed of the AC motor M1 is the reference speed Nn♯, equivalent to the maximum offset amount. The maximum amount of offset, the phase current is equivalent to the filter current Iwf at half period elapse of time is the maximum rated current.

In view of the low-pass filtering is performed by the filter processing unit 110, when the rotational speed of the AC motor M1 is high, the offset amount with respect to the phase current of the same amplitude since also small, the offset correction amount Icr also needs to be reduced. Therefore, (3) As shown in equation, the offset correction amount Icr, as will become smaller when the motor rotational speed increases, is variably set according to the motor rotation speed. That is, the offset correction amount Icr was set in consideration of the motor rotational speed, which corresponds to the "maximum offset value" which occurs in the filter current when switched control mode during the current cycle.

Thus, the offset correction unit 120, when switching of the control mode, the filter current Iwf control mode to be resumed is corrected by the offset correction amount Icr. And, other timings, i.e. at a time other than during the control mode switching, the filter current Iwf calculated by the filter processing unit 110 is output to the detection section 140 as it is.

Detection unit 140, in this way by comparing the judgment value IJD set by the filter current Iwf a determination value setting unit 130 obtained, opened and the absolute value of the filter current Iwf is greater than the determination value IJD detects a fault, it turns on the detection flag FDG. When the detection flag FDG is turned on, so that the protection processing and the like of the inverter corresponding to the open-circuit failure is performed.

Figure 10 is a flowchart illustrating a series of control processes in the inverter fault detection according to the present embodiment. Each step of the flowchart shown in later, is basically realized by software processing by the control unit 30 may be realized by hardware processing.

Incidentally, in FIG. 10, the control process for the open-circuit failure detection based on the calculated filter current Iwf in PWM mode W-phase are shown, the same control process corresponds to the phase currents in each control mode It is provided Te.

Referring to FIG. 10, the control device 30, the step S100, obtains the phase current Iw. Phase current Iw as described above may be a detected value of current sensor 24 may be a calculated value obtained from the value detected by the current sensor 24 of the other phases.

Furthermore, the control device 30, in step S110, whether or not the current control mode is the control mode of the target, in the example of FIG. 10, whether a PWM mode based on the control signal CMD (Fig. 9) It is determined.

By the time the current control mode is other than PWM mode (YES in S110), the control unit 30, in step S160, to hold the filter current Iwf the (1) to the current value.

On the other hand, the control device 30, when the current control mode is the control mode of the target (YES in S110), in step S120, the control mode switching time, i.e., whether the resume timing of the control mode judge. Then, when a control mode switch timing (YES in S120), control device 30 proceeds to step S140, to offset correction filter current IWF (1) which is held so far.

Figure 11 is a detail of the offset correction process in step S140 of FIG. 10 is shown.
Referring to FIG. 11, the control device 30, in step S141, it obtains the current motor rotational speed Nm of AC motor M1. Then, the control device 30, in step S142, in accordance with the motor rotational speed Nm of the income, to determine the offset correction amount Icr according to the above (3).

Furthermore, the control device 30, in step S143, determines whether the absolute value of the control mode changeover time filter current values ​​held by IWF (1) is the offset correction amount Icr hereinafter set in step S142 . Then, the control device 30, | Iwf (1) | The time of ≦ Icr (YES in S143), it advances the process to step S145, and Iwf (1) = 0. Thus, a small current offset than the offset correction amount Icr determined to correspond to the maximum offset amount would control mode of the target is cleared when to resume.

Controller 30, | Iwf (1) | in> when Icr (determination of NO at S143) determines, in step S144, the polarity of the filter current IWF (1) being held. Then, when the filter current IWF (1) is positive (YES in S144), control device 30 proceeds to step S147, subtracts the offset correction amount Icr from the filter current IWF (1) by executes offset correction. On the other hand, by the time the filter current IWF (1) is negative (determination of NO at S144), control device 30, in step S146, for adding the offset correction amount Icr Filter current IWF (1), carry out the offset correction.

In this way, without inverting the polarity of the filter current IWF (1), within the range of the absolute value of the offset correction amount Icr, control mode switching filter current IWF that has been held before (1) is modified It is is will be. Incidentally, instead of a simple zero clear, failures in case by modifying only the offset amount derived from the control mode changeover (Icr), the occurrence of open failure absolute value of the filter current is increased toward the judgment value it is possible to prevent omission of detection.

Referring again to Figure 10, controller 30, in addition the switching timing of the control mode (determination of NO at S120), it advances the process to step S130, the low-pass filtering in accordance with equation (1), to update the filter current IWF (1).

Thus, by the processing in step S130 or S140, the filter current value of the PWM mode IWF (1) is determined, the control device 30, the step S150, the absolute value and the determination value of the filter current IWF (1) comparing the IJD (1). Then, the control device 30, | IWF (1) | in> when IJD of (1) (YES in S150), the relevant phase (here, W-phase) in step S200 to detect an open failure of the. On the other hand, | IWF (1) | The time ≦ IJD of (1) (determination of NO at S150), since the step S200 is skipped, open-circuit failure is not detected.

According to the failure detection device of the inverter according to the present embodiment as described above, in a normal (non-occurrence of an open fault), continually be switched before the control mode is elapses one period of the phase current It never is generated, for detecting the open-circuit failure erroneous. Further, as compared with the case of simply reset to zero filter current value for each switching of the control mode (current integral value), the possibility of miss the open failure is low, it is possible to improve detection accuracy.

Thus, the open-circuit failure of the power semiconductor switching devices constituting an inverter for driving an AC motor in which a plurality of control modes are switched by control, also accurately detected when the control mode is frequently switched in a short time be able to.

In the present embodiment, as a preferred configuration example, the input voltage to the inverter 14 (system voltage VH) to variably controllable, DC voltage generating unit 10♯ of the motor control system shows a structure including a converter 12 It was, but the input voltage to the inverter 14 variably controlled if, not limited to the illustrated configuration to a DC voltage generating unit 10♯ this embodiment. Moreover, not necessarily essential that the inverter input voltage is variable, configuration in which the output voltage of the DC power supply B is directly input to the inverter 14 (e.g., configuration omitting the arrangement of the converter 12) also present invention to it is applicable.

As for the AC motor as a load of the motor control system, in this embodiment, the electric vehicle (a hybrid vehicle, an electric vehicle, etc.) it is assumed a permanent magnet motor mounted as a vehicle driven, the other device construction for a load any AC motor used for can also be applied to the present invention.

The embodiments disclosed herein are to be considered as not restrictive but illustrative in all respects. The scope of the invention is defined by the appended claims rather than by the foregoing description, and is intended to include all modifications within the meaning and range of equivalency of the claims.

This invention can be used in open-circuit failure detection of power semiconductor switching elements constituting the inverter.

5 ground wire, 6,7 power lines, 10 and 13 voltage sensor, 10 # DC voltage generating unit, 11 current sensor, 12 converter, 14 inverter, 15 U-phase upper and lower arms, 16 V-phase upper and lower arms, 17 W-phase upper and lower arms, 24 current sensor, 25 rotation angle sensor, 30 a control unit (ECU), 35 control mode selecting section, 100 a motor drive system, 101 the failure detection device, 110 filter unit, 120 an offset correction unit, 130 determination value setting unit 140 detects part, B DC power supply, C0, C1 smoothing capacitor, CMD control signal (control mode), D1 to D8 antiparallel diodes, FDG detection flag (open failure), Ib DC current, Icr offset correction amount, IJD determination value, Iu, iv, Iw phase current, Iuf, Ivf, Iwf filter current, L1 reactor, M1 Flow motor, Nm motor rotation speed, the semiconductor switching element for Q1~Q8 power, S1 to S8 switching control signal, SR1, SR2 system relay, Trqcom torque command value, Vb, VH DC voltage, theta rotor rotation angle.

Claims (4)

1. A failure detection device of a plurality of phases of the inverter to control the AC motor,
   A current detecting means for detecting a current of each phase of the inverter,
   Depending on the operating state of the AC motor, and mode selection means for selecting one of a plurality of control modes,
   By low-pass filtering the phase currents detected by said current detecting means, and filtering means for calculating a phase of the filter current values ​​separately for each said control mode,
   When the filter current value calculated by the filtering means exceeds a predetermined judgment value, the failure detection means for detecting the open-circuit failure of the switching element of the corresponding phase of the inverter,
   When the control mode is switched, the filter current value of each phase corresponding to the control mode after the switching, comprising absolute value of a correction means for correcting to decrease within a predetermined range, the inverter failure detection device.
2. Said correction means, the absolute value of the filter current value is greater than the maximum offset value corresponds to the integral value of the half cycle of the maximum rated current of each phase current, the absolute value is only the maximum offset value while modifying the filter current value as decrease, the when the absolute value of the filter current is less than the maximum offset value sets the filter current value to zero, fault detection device of the inverter according to claim 1 .
3. The maximum offset value, as a small value in accordance with increase in the rotational speed of the AC motor is variably set according to the rotational speed of the AC motor, an inverter failure detecting apparatus according to claim 1 or 2 .
4. It said current detecting means,
   A current sensor arranged in each phase, except for predetermined one phase among the plurality of phases,
   And a current calculating means for calculating the current of the predetermined one-phase from the output of said current sensor,
   Said filtering means, said at predetermined one phase while calculating the filter current value based on the calculated value by the current calculation unit, in each phase, except for the predetermined single-phase based on the value detected by said current sensor It calculates the filter current value each,
   Wherein said judgment value of a given one phase, said is larger than the judgment value of each phase except one predetermined phase, inverter fault detection according to any one of claims 1 to 3 apparatus.

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