JP2002078338A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce power loss in a composite resonance type switching converter, secure stable ZVS under heavy load conditions, and improve manufacturing efficiency. SOLUTION: The primary winding and secondary winding of an insulating converter transformer, provided in a composite resonance type switching power supply circuit, are wound respectively in directions opposite to each other and are connected to each other in an additive polarity manner. With such a constitution, since a primary flux and a secondary flux cancel each other, the core of the insulating converter transformer is not saturated, even if a gap is not formed in the core.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 7 is a circuit diagram showing an example of a prior art switching power supply circuit which can be constructed based on the invention proposed by the present applicant. As a basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in FIG. 1, a rectified and smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC is generated from a commercial AC power supply (AC input voltage VAC) by a bridge rectifier circuit Di and a smoothing capacitor Ci. .

[0005] A single-ended single-end system is adopted as a voltage resonance type converter that receives and inputs the rectified smoothed voltage Ei (DC input voltage) and is intermittent. The drive system employs a self-excited configuration. In this case, the switching element Q1 forming the voltage resonance type converter includes:
A high breakdown voltage bipolar transistor (BJT; junction transistor) is selected. A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. A clamp diode DD is connected between the base and the emitter. Here, the parallel resonance capacitor Cr forms a primary side parallel resonance circuit together with the leakage inductance L1 obtained in the primary winding N1 of the insulating converter transformer PIT, so that an operation as a voltage resonance type converter can be obtained. It has become. A self-excited oscillation drive circuit including a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB is connected to the base of the switching element Q1. The switching element Q1 is switched by being supplied with a base current based on an oscillation signal generated by the self-excited oscillation drive circuit.
In addition, at the time of startup, it is started by a startup current flowing from the line of the rectified smoothed voltage Ei to the base via the startup resistor Rs and the base current limiting resistor RB.

The orthogonal control transformer PRT is configured by winding a control winding Nc so that the winding direction of the drive winding NB and the current detection winding ND is orthogonal to the winding direction. It is provided for controlling the switching frequency of the primary side voltage resonance type converter as described later.

[0007] The insulating converter transformer PIT is provided for transmitting the switching output of the switching converter obtained on the primary side to the secondary side. As the structure of the insulating converter transformer PIT, for example, as shown in FIG. 8, an EE-type core including ferrite E-type cores CR1 and CR2 is provided. Then, as shown in the figure, using the divided bobbin B, the primary winding N1 and the secondary winding N2, both of which are litz wires, are wound around the divided regions. Here, the primary winding N1 and the secondary winding N2 are both wound in the same winding direction. A gap G is formed in the center magnetic leg of the EE type core as shown in the figure. The leakage inductance in the insulating converter transformer PIT is determined by the gap length of the gap G, and loose coupling by a required coupling coefficient can be obtained. Here, as the coupling coefficient k, for example, a loose coupling state of k ≒ 0.85 is obtained, and accordingly, a saturated state is hardly obtained. This gap G is an E-shaped core C
The center magnetic legs of R1 and CR2 can be formed by making them shorter than the two outer magnetic legs. In this case, the gap length is about 1 mm.

By the way, the insulation converter transformer PIT
In the operation in, depending on the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection relationship of the rectifier diode DO,
Regarding the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2, +
There is a case where the operation mode is M (addition polarity mode: forward operation) and a case where the operation mode is -M (depolarization mode: flyback operation). For example, assuming that the polarity (winding direction) of the primary winding N1 and the secondary winding N2 are the same, the mutual inductance is + M when the circuit is equivalent to the circuit shown in FIG. Mutual inductance is -M when equivalent to the circuit shown.

The winding start end of the primary winding N1 of the insulating converter transformer PIT having the above-mentioned structure is connected to the collector of the switching element Q1, and the winding end is
It is connected to the line of the rectified smoothed voltage Ei via the current detection winding ND. The winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end is connected to the positive terminal of the smoothing capacitor CO via the rectifier diode DO. In such a connection form, the primary winding N1 and the secondary winding N2 of the insulating converter transformer PIT are connected by additional polarity, which is equivalent to the equivalent circuit shown in FIG. It corresponds to.

The switching output of the switching element Q1 is transmitted to the primary winding N1 of the insulating converter transformer PIT having the above-described structure, and further transmitted to the secondary winding N2 so as to be excited.

In this case, the insulation converter transformer PIT
On the secondary side, a secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2 as shown in the figure, so that the secondary parallel resonance capacitor C2 is connected together with the leakage inductance L2 of the secondary winding N2. Form a circuit. Furthermore, a secondary-side series resonance capacitor Cs is connected in series to the winding end end of the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the secondary-side series resonance capacitor Cs are connected. Depending on the capacitance of the resonance capacitor Cs, a secondary-side series resonance circuit is formed. That is, the power supply circuit shown in this figure employs a configuration in which the secondary side parallel resonance circuit and the secondary side series resonance circuit are combined and combined on the secondary side. In this specification, such a configuration on the secondary side is also referred to as a “secondary-side composite resonance circuit”.

In this case, a bridge rectifier circuit DBR is connected to the secondary side parallel resonant circuit as shown in FIG.
And a rectifying / smoothing circuit including a smoothing capacitor Co, so that a secondary-side DC output voltage EO is obtained at both ends of the smoothing capacitor CO.

Here, the invention as the secondary-side composite resonance circuit as described above has been previously filed by the present application as Japanese Patent Application No. 11-331027. For example, when only the secondary parallel resonance circuit is used, the leakage inductance component L2 of the secondary winding N2 and the junction capacitance of the rectifier diode act to turn on the rectifier diode when the secondary rectifier diode is turned on. A high-frequency ringing current (oscillating current) is superimposed on the current, and this is the power supply noise (EMI;
It will be radiated as Electromagnetic Interference. When only the secondary-side series resonance circuit is used, ringing of the secondary-side rectified current does not occur, but the load power is, for example, 50 W to 120 W.
In a region where the load is intermediate, that is, the range of W, the switching element Q1 is connected to a so-called ZVS (Zero Voltage Switch).
An abnormal operation that deviates from the (ching) operation occurs, resulting in a reduction in power loss. Therefore, if a secondary-side composite resonance circuit is formed by combining the secondary-side parallel resonance circuit and the secondary-side series resonance circuit, it is possible to eliminate both ringing of the secondary-side rectified current and stable ZVS. It can be realized.

When such a power supply circuit is viewed as an entire configuration, a parallel resonance circuit is provided on the primary side to make the switching operation a voltage resonance type, and a secondary resonance circuit is provided on the secondary side. This means that a side composite resonance circuit is provided. Therefore, in the present specification, such a switching converter of a power supply circuit is also referred to as a “composite resonance type switching converter” in the sense that a resonance circuit is provided on the primary side and the secondary side in a complex manner. To
this

The control circuit 1 allows a variable DC current to flow through the control winding Nc of the orthogonal control transformer PRT as a control current in accordance with the level of the secondary DC output voltage EO. In this manner, the level of the control current flowing through the control winding Nc is varied, so that the orthogonal control transformer PRT is used.
In, the control is performed so that the inductance LB of the drive winding NB is variable. As a result, the resonance frequency of the resonance circuit including the drive winding NB and the resonance capacitor CB in the self-excited oscillation drive circuit changes, and the switching frequency of the switching element Q1 is variably controlled. By changing the switching frequency of the switching element Q1 in this manner, the secondary side DC output voltage EO
Is controlled to be constant. That is, the power supply is stabilized.

[0016]

In the power supply circuit shown in FIG. 7, as shown by the structure of the insulated converter transformer PIT in FIG. 8, the primary winding N
The winding directions of 1 and the secondary winding N2 are the same. Therefore, the primary winding current I1 flowing through the primary winding N1 generates a magnetomotive force in the primary winding N1, and similarly, the secondary winding N2 depends on the secondary winding current I2 flowing through the secondary winding N2. Generates a magnetomotive force. As a result, a primary magnetic flux φ1 is generated on the primary side and a secondary magnetic flux φ2 is generated on the secondary side as shown in FIG.
Occurs. Here, as described above, since the primary winding N1 and the secondary winding N2 in the circuit of FIG. 7 are connected with additional polarity, the above-described primary magnetic flux φ1 and secondary magnetic flux φ
2 are operations to be added to each other, so that a magnetic flux represented by φ1 + φ2 is generated in the center magnetic leg of the insulating converter transformer PIT.

That is, since the primary winding N1 and the secondary winding N2 are in the same winding direction and are connected in a polarized manner, the primary magnetic flux φ1 and the secondary magnetic flux φ2 are applied to the center magnetic leg. A relatively large magnetic flux will be generated. Here, assuming that no gap is formed in the center magnetic leg of the core of the insulating converter transformer PIT (gap length = 0),
For example, when the load power Po is about 100 W or more, the ferrite core enters a saturation region of a so-called magnetization curve. In this specification, “saturation” refers to a state in which the magnetization curve enters a saturation region. As a result, the inductance of the core is sharply reduced, and the possibility that the switching element Q1 of the BJT is broken is increased. Therefore, the insulation converter transformer PI
As for T, the gap G is formed as shown in FIG. 8 so that a loose coupling state with a required coupling coefficient can be obtained, thereby preventing saturation.

In order to avoid the above-mentioned phenomenon and satisfy the regulation range in the case of the power supply circuit having the configuration shown in FIG. 7, the gap length of the gap G formed in the insulating converter transformer PIT should be 1 mm ±
It is necessary to perform management with an accuracy of 0.1 mm. In order to satisfy the gap length accuracy described above, the E-type cores CR1 and CR2 need to be manufactured and polished at the end of each central magnetic leg and controlled with an accuracy of 0.5 mm ± 0.05 mm. Come. Therefore, the process for polishing the center magnetic leg of the E-shaped core with high precision is required, so that the manufacturing time becomes longer. In addition, it corresponds to the case where the same E-shaped core is manufactured with a different gap length from the insulated converter transformer. It is also difficult to manage products. In other words, the necessity of forming a gap causes a reduction in manufacturing efficiency.

When a gap G is formed in the insulating converter transformer PIT, a leakage flux called fringe flux is generated in the vicinity of the gap G, so that a litz wire primary winding N1 and a secondary winding N2 are formed. In this case, eddy current loss occurs and local heat generation occurs. This heat is conducted to the winding having a lower temperature, thereby increasing the temperature of the winding itself. As a result, it has been found that power loss called copper loss increases and power conversion efficiency decreases. In particular, in the circuit shown in FIG.
Since the amount of high-frequency current flowing through the primary winding current I1 flowing through the secondary winding N2 and the amount of high-frequency current flowing through the secondary winding current I2 flowing through the secondary winding N2 is large, the DC current as a litz wire in the primary winding current I1 and the secondary winding N2 is large. Heat generation due to the resistance and the eddy current loss described above is remarkable.

Further, in the circuit shown in FIG. 7, the level of the AC input voltage VAC is 7
When the voltage drops to about 5V to 85V, the operation of the primary-side switching element Q1 is set to ZVS (Zero Volta).
There is also a problem that a period of abnormal operation that does not result in ge switching) operation occurs. If such a phenomenon continues, the switching element Q1 may generate heat and be destroyed in a short time.

[0021]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. A switching unit formed with a switching element for switching and outputting an input DC input voltage; and a primary side parallel resonance circuit having an operation of the switching unit as a voltage resonance type. And a structure in which a required coupling coefficient that is loosely coupled between the primary side and the secondary side is obtained, and the output of the switching means obtained on the primary side is transmitted to the secondary side. An insulating converter transformer. Further, for the secondary side, a secondary side parallel resonance circuit formed by connecting a secondary side parallel resonance capacitor in parallel to the secondary winding of the insulating converter transformer,
A secondary composite resonance circuit is provided which is combined with a secondary series resonance circuit formed by connecting a secondary series resonance capacitor in series to a secondary winding of the insulating converter transformer. The secondary converter is formed with a secondary composite resonance circuit, and is configured to obtain a secondary DC output voltage by inputting an alternating voltage obtained in a secondary winding of the insulating converter transformer and performing a rectification operation. DC output voltage generating means, and constant voltage control means for performing constant voltage control by variably controlling the switching frequency of the switching element according to the level of the secondary DC output voltage. The insulating converter transformer has a core in which no gap is formed, and the primary winding and the secondary winding are wound around the core in opposite winding directions, and the primary winding and the secondary winding are wound. Polarity connection was made to the wire.

According to the above configuration, the primary side includes the primary side parallel resonance circuit for forming the voltage resonance type converter, and the secondary side includes the secondary winding, the secondary side parallel resonance capacitor, and the secondary side parallel resonance circuit. A so-called composite resonance type switching converter including a secondary side composite resonance circuit formed by the side series resonance capacitor is obtained. Further, the constant voltage control is performed by changing the switching frequency of the primary-side voltage resonance type converter. Based on this configuration, for the insulated converter transformer, the primary winding and the secondary winding are wound in mutually opposite winding directions, and then the primary winding and the secondary winding are connected in polarities. As a result, the magnetic fluxes obtained in the primary winding and the secondary winding act so as to cancel each other out, so that the magnetic flux generated in the core can be small and the saturation state can be suppressed accordingly. . For this reason, no gap is provided for the purpose of suppressing saturation in the core of the insulating converter transformer in the switching power supply circuit of the present invention.

[0023]

FIG. 1 shows a configuration example of a switching power supply circuit according to a first embodiment of the present invention. The power supply circuit shown in FIG. 1 employs a configuration as a composite resonance type switching converter including a voltage resonance type converter on the primary side and a secondary side composite resonance circuit on the secondary side.

In the power supply circuit shown in this figure, a bridge rectification circuit Di is used as a rectification smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) and obtaining a DC input voltage.
And a full-wave rectifier circuit including a smoothing capacitor Ci, and generates a rectified smoothed voltage Ei corresponding to a level equal to one-half of the AC input voltage VAC.

The voltage resonance type switching converter provided in the power supply circuit has a single switching element Q1.
It has a self-excited configuration with In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1.

Between the base of the switching element Q1 and the ground on the primary side, a series resonance circuit for driving self-oscillation composed of a series connection circuit of a drive winding NB, a resonance capacitor CB and a base current limiting resistor RB is connected. The base of the switching element Q1 is a base current limiting resistor RB−a starting resistor R
It is also connected to the positive side of the smoothing capacitor Ci (rectified and smoothed voltage Ei) via S, so that the base current at the time of startup is obtained from the rectified and smoothed line.

The switching element Q1 is connected to a clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci.
A path of a clamp current flowing when the power supply is off is formed. The collector of the switching element Q1 is connected to one end of the primary winding N1 of the insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Forms a primary parallel resonance circuit of the voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT described later. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage VQ1 across the parallel resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit due to the action of the parallel resonance circuit. Operation is obtained.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a resonance current detection winding ND, a drive winding NB, and a control winding NC are wound. This orthogonal control transformer PRT drives the switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, although not shown, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. . Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is constructed by winding in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detecting winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching element Q1 Is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is applied to the drive winding NB. appear. This drive voltage is output from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

The insulation converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. The present embodiment is characterized by the structure of the insulation converter transformer PIT, which will be described later.

The winding end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the winding start is connected to the resonance current detecting winding ND in series with the smoothing capacitor Ci. It is connected to the positive electrode (rectified smoothed voltage Ei).

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a secondary-side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, and a secondary-side series resonance capacitor C2 is connected to one end (winding start end) of the secondary winding N2. The resonance capacitor Cs is connected in series.
That is, on the secondary side, the secondary side parallel resonance capacitor C
2 and the leakage inductance L2 of the secondary winding N2 form a secondary parallel resonance circuit (voltage resonance circuit), the capacitance of the secondary series resonance capacitor Cs and the leakage inductance of the secondary winding N2. L2
Thus, a secondary series resonance circuit (current resonance circuit) is formed. In other words, the secondary side employs a configuration in which a voltage resonance circuit and a current resonance circuit having the secondary winding N2 in common as an inductance are combined in a complex manner. In other words, on the secondary side, there is provided a “secondary-side composite resonance circuit” that is a combination of a voltage resonance circuit that is a secondary-side parallel resonance circuit and a current resonance circuit that is a secondary-side series resonance circuit. Things.

When such a configuration is considered as a whole power supply circuit, a primary side parallel resonance circuit is provided on the primary side to make the switching operation a voltage resonance type, and a secondary side composite resonance circuit is provided on the secondary side. The circuit has a configuration as a “composite resonance type switching converter” provided with a circuit.

In the case of this power supply circuit, a rectification circuit including a bridge rectification circuit DBR and a smoothing capacitor CO is provided for the secondary side formed as described above, as shown in the figure. Here, the positive input terminal of the bridge rectifier circuit DBR is connected to the secondary side series resonance capacitor Cs.
Is connected to the winding start end of the secondary winding N2.
The bridge rectifier circuit DBR inputs the resonance output of the composite resonance circuit composed of [secondary winding N2, secondary-side parallel resonance capacitor, and secondary-side series resonance capacitor], performs rectification, and charges the smoothing capacitor CO. Is performed to obtain the secondary-side DC output voltage EO as the voltage across the smoothing capacitor CO. The DC output voltage EO is also branched and input to the control circuit 1.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is varied according to the change in the secondary side DC output voltage level EO, so that the control current is wound around the orthogonal control transformer PRT. The variable inductance LB of the driving winding NB is controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This changes the switching frequency of the switching element Q1, thereby stabilizing the DC output voltage on the secondary side.

FIG. 2 shows the structure of the insulating converter transformer PIT provided in the power supply circuit shown in FIG. As shown in the figure, the insulating converter transformer PIT forms an EE-type core with two E-type cores CR1 and CR2. The EE-type core is provided with a divided bobbin B. As shown in the drawing, for example, a primary winding N1 is wound around a winding region of the divided bobbin B on the E-type core CR1 side, and the E-type core is wound. The secondary winding N for the winding area on the CR2 side
Wrap 2 In the case of the present embodiment, the winding directions of the primary winding N1 and the secondary winding N2 are opposite to each other as shown by arrows on the left and right sides of the core in the drawing. , A so-called reverse winding structure.
Further, in the case of the present embodiment, E-type cores CR1, CR
No gap is formed at the opposing portion of each of the center magnetic legs.

Referring again to FIG. 1, the connection between the primary winding N1 and the secondary winding N2 of the insulating converter transformer PIT will be described. As shown in FIG. 1, for example, the connection between the winding start end and the winding end end of the primary winding N1 is opposite to that of the circuit shown in FIG. 7 as the prior art. That is,
In the circuit shown in FIG. 1, the winding start end of the primary winding N1 is connected to the positive terminal of the smoothing capacitor Ci via the current detection winding ND, and the winding end is connected to the switching element Q1.
Connected to one collector. In the secondary winding N2, the winding end is connected to the positive terminal of the smoothing capacitor CO via the rectifier diode DO, and the winding end is connected to the secondary side ground. . That is, even if the power supply circuit shown in FIG. 1 includes the insulating converter transformer PIT in which the primary winding N1 and the secondary winding N2 have a reverse winding structure as shown in FIG. The primary winding N1 and the secondary winding N2 are connected so as to have the additional polarity shown in the equivalent circuit of FIG.

According to such a configuration, the primary magnetic flux φ1 generated by the primary winding current I1 flowing through the primary winding N1.
And the polarity of the secondary magnetic flux φ2 generated by the secondary winding current I2 flowing through the secondary winding N2 are as shown by arrows in the core in FIG. This is, for example, that the polarity of the primary magnetic flux φ1 is reversed with respect to the case of the insulating converter transformer PIT of FIG. 8 shown as the prior art. The polarity of the secondary magnetic flux φ2 is the same as that of the insulating converter transformer PIT in FIG. In the present embodiment, the polarity relationship between the primary magnetic flux φ1 and the secondary magnetic flux φ2 as shown in FIG. 2 is obtained. As a result, the primary magnetic flux φ1 and the secondary magnetic flux φ2 are Act to cancel each other. That is, the magnetic flux (Δφ) obtained at the center magnetic leg of the insulating converter transformer PIT is expressed as | φ1−φ2 | = Δφ. This indicates that, as described above, the primary magnetic flux φ1 and the secondary magnetic flux φ2 cancel each other, and do not add together, for example, as in the case of the circuit of FIG. Therefore, in the present embodiment, the magnetic flux obtained in the center magnetic leg of the insulating converter transformer PIT can be weaker than before. As a result, as the coupling coefficient k between the primary side and the secondary side, a loose coupling state of, for example, k = about 0.8 to 0.9 can be obtained. Thereby, the insulating converter transformer PIT of the present embodiment can prevent the core from being saturated without intentionally forming a gap with respect to the center magnetic leg, and as a result, as shown in FIG. And no gap is provided. In practice, for example, a so-called core squeal of audible sound may occur at the joint surface of the center magnetic leg with the gap length set to 0, so that, for example, a Mylar film is applied to the joint surface of the center magnetic leg. Therefore, a gap having a gap length of 0.1 mm or less may be formed.

By configuring the insulating converter transformer PIT in this way, the magnetic flux obtained at the center magnetic leg is much weaker than before, and therefore, for example, as shown in FIG. The temperature rise of the winding due to the fringe magnetic flux generated around the gap and the reduction of the power conversion efficiency due to the temperature rise are also eliminated.

Further, as the insulating converter transformer PIT of the present embodiment, the magnetic flux (Δφ) obtained in the center magnetic leg is used.
Is weak, the primary winding N1 and the secondary winding N2
Will also be reduced. And
As a result, even under a heavy load condition of, for example, a load power Po of about 200 W, the switching element Q1 can realize a stable ZVS operation.

FIG. 3 is a waveform diagram showing an operation of a main part in the power supply circuit of FIG. 1 having the above-described configuration. here,
AC input voltage VAC = 100V, load power Po = 200W
The operation under the condition of is shown. FIG. 3A shows a parallel resonance voltage V obtained at both ends of a primary-side parallel resonance capacitor Cr.
Q1 is shown. As shown in this figure, a pulse of, for example, 600 Vp is obtained for the parallel resonance voltage VQ1 during the period TOFF when the switching element Q1 is off, and a waveform that becomes 0 level during the period TON when the switching element Q1 is on.

As shown in FIG. 3B, the primary winding voltage V1 obtained at both ends of the primary winding N1 has a positive polarity pulse corresponding to the period TOFF and corresponds to the period TON. As a result, a waveform which is clamped at the absolute value level of the rectified smoothed voltage Ei is obtained in the negative polarity region. The primary winding current I1 flowing through the same primary winding N1 is shown in FIG.
Although an alternating waveform as shown in (c) is obtained, for example, a level of 3.7 Ap is obtained in a positive polarity region. On the other hand, in the circuit shown in FIG. 7, a level of 4 Ap, which is higher than that in the present embodiment, is obtained as shown by a broken line, for example.

As shown in FIG. 3D, the secondary winding voltage V2 obtained at both ends of the secondary winding N2 has a pulse shape in the negative polarity region and the secondary winding voltage V2 in the positive polarity region. An alternating waveform clamped at the level of the side DC output voltage EO is obtained. As shown in FIG. 3 (e), the secondary winding current I2 flowing through the secondary winding N2 has a pulse shape in the negative polarity region, and has a level of 4 Ap in the positive polarity region. It becomes an alternating waveform. On the other hand, in the circuit shown in FIG. 7, a higher level of 4.5 Ap was obtained as shown by the broken line in the figure.

Here, the primary winding current I
The peak level lower than that of the circuit of FIG. 7 for the primary winding current I2 and the secondary winding current I2 is due to, for example, a reduction in the leakage inductance in the insulated converter transformer PIT. In this way, the levels of the primary winding current I1 and the secondary winding current I2 are reduced, and the current amounts are reduced, so that the primary winding current I1 and the secondary winding current I2 are reduced.
In this case, power loss and heat generation due to the DC resistance component and eddy current of the litz wire are reduced.

FIG. 4 shows the primary side parallel resonance voltage V1.
And a switching output current IQ1 flowing through the switching element Q1. The conditions at this time are as follows: load power Po = 200 W, AC input voltage V
It is assumed that the pressure is reduced at about AC = 75V to 85V. FIG. 4 (a)
The primary side parallel resonance voltage VQ1 and the switching output current IQ1 show the operation of the circuit of the present embodiment shown in FIG. 1, and the primary side parallel resonance voltage VQ1 and the switching state shown in FIG. The output current IQ1 indicates the operation of the circuit shown in FIG. As can be seen from the waveforms of FIGS.
At the timing when the switching output current IQ1 is inverted from the negative polarity level to the positive polarity level when ON, that is, at the timing when the clamp current flowing through the clamp diode DD ends and the collector current starts to flow through the switching element Q1, the primary side parallel resonance voltage A phenomenon occurs in which VQ1 and the switching output current IQ1 appear as pulses at a positive level. In other words, the abnormal operation is not the ZVS operation. On the other hand, in the circuit shown in FIG. 1, as shown in FIGS.
, The pulse of the primary side parallel resonance voltage VQ1 has disappeared, and at the same time, the waveform of the switching output current IQ1 has a normal waveform in which no pulse appears. That is, this embodiment shows that the ZVS operation is performed normally even under the condition of a heavy load and a low AC input voltage.

Here, the specifications of the main parts of the power supply circuit shown in FIG. 1 will be described. First, the gap length Gap of the insulating converter transformer PIT is set to zero. The number of turns of the primary winding N1 and the secondary winding N2 is N
1 = 50T and N2 = 43T. Further, the primary-side parallel resonance capacitor Cr = 2700 pF, the secondary-side parallel resonance capacitor C2 = 8200 pF, and the secondary-side series resonance capacitor Cs = 0.022 μF. Further, in the power supply circuit shown in FIG.
EE-40 core is adopted, and Gap = 1 mm. Also, the primary winding N1 = 45T, the secondary winding N2 = 38
T, primary-side parallel resonance capacitor Cr = 3300 pF, secondary-side parallel resonance capacitor C2 = 0.01 μF, and secondary-side series resonance capacitor Cs = 0.027 μF.

Then, load power Po = 200 W, VAC =
As a result of an experiment under the condition of 100 V, the circuit shown in FIG. 7 has an AC input power of 222.0 W and a power conversion efficiency of 90.1%. In contrast, the circuit shown in FIG. 1 has an AC input power of 219.3 W and a power conversion efficiency of 91.2%, both of which are improved. The circuit shown in FIG. 7 has a temperature rise value of 4 in the primary winding N1.
7 ° C. and 54.5 ° C. inside the secondary winding N2.
On the other hand, in the circuit shown in FIG. 1, a result of 42 ° C. inside the primary winding N1 and 49.0 ° C. inside the secondary winding N2 was obtained, and a large value of about 5 ° C. was obtained for each. Significant decline has been seen.

FIG. 5 shows a configuration example of a switching power supply circuit according to a second embodiment of the present invention. In addition,
In this figure, the same parts as those in FIG.

The primary-side voltage resonance type converter of the power supply circuit shown in FIG. 1 employs a single-ended configuration of a separately excited type. In this case, a MOS-FET is employed as the switching element Q1. This M
The drain of the switching element Q1 as the OS-FET is connected to the winding end of the primary winding N1, and the source is connected to the primary side ground. The parallel resonance capacitor Cr is connected in parallel between the drain and the source of the switching element Q1. Further, a clamp diode DD is also connected in parallel between the drain and source of the switching element Q1.

The switching drive section 2 is provided for driving the switching element Q1 in a separately-excited manner and for controlling the switching frequency, and can be configured as, for example, a single IC. The switching drive unit 2 includes an oscillation circuit 3 and a drive circuit 4. At the time of startup, the switching drive unit 2 is configured to obtain power for startup via the startup resistor Rs from the line of the rectified smoothed voltage Ei.

The oscillation circuit 3 generates an oscillation signal and outputs it to the drive circuit 4. In the drive circuit 4, the input oscillation signal is converted into a drive voltage capable of driving the switching element Q1, which is a MOS-FET,
Output to the gate of switching element Q1. As a result, the switching element Q1 is driven for switching. The oscillation circuit 3 is configured to vary the frequency of the oscillation signal in accordance with the error detection output of the secondary DC output voltage EO output from the control circuit 1. In this way, the switching frequency of the switching element Q1 is varied by driving the switching element Q1 based on the oscillation signal whose frequency is varied, thereby stabilizing the secondary DC output voltage EO. Will be planned.

On the secondary side of the circuit shown in FIG. 5, two rectifier diodes D01 and DO2 and a smoothing capacitor CO are connected to the secondary-side composite resonance circuit in a connection form as shown. Thus, a voltage doubler half-wave rectifier circuit is formed. The voltage doubler rectifier circuit charges the secondary side series resonance capacitor Cs by the current rectified by the rectifier diode DO2 in a half cycle of the alternating voltage obtained in the secondary winding N2. Then, in the next half cycle, the secondary-side series resonance capacitor C
The operation in which the rectifier diode D01 conducts while the obtained potential is applied to s to charge the smoothing capacitor CO is repeated. By such an operation, a level corresponding to twice the alternating voltage level obtained in the secondary winding N2 is obtained as the secondary DC output voltage EO which is a voltage across the smoothing capacitor CO. . Therefore,
In the case where the voltage doubler half-wave rectifier circuit is provided on the secondary side in this way, if the secondary side DC output voltage EO can be a level equivalent to the level obtained by the equal-voltage rectifier circuit, The number of turns of the next winding N2 can be reduced to about 1/2 of the normal number. Even with such a configuration, as the isolated converter transformer PIT, FIG.
Having a primary winding N1 and a secondary winding N2.
The effect similar to that of the power supply circuit shown in FIG.

FIG. 6 shows a configuration example of a switching power supply circuit according to the third embodiment. In this figure, the same parts as those in FIGS. 1 and 5 are denoted by the same reference numerals, and description thereof will be omitted. In the circuit shown in this figure, an IGBT (insulated gate bipolar transistor) is employed as the switching element Q1 forming the primary side voltage resonance type converter. IGBTs are known for having high performance switching characteristics. The switching drive unit 2 is provided as in the case of FIG.
T is driven separately. For example, SIT (static induction thyristor) other than IGBT may be adopted.

In the circuit shown in this figure, as a circuit for obtaining a rectified smoothed voltage Ei from the AC input voltage VAC, two rectifier diodes Di1, Di2 and two smoothing capacitors Ci1-- connected in series. With the provision of Ci2, a rectified smoothed voltage Ei corresponding to twice the AC input voltage VAC is obtained at both ends of the series connection circuit of the smoothing capacitors Ci1-Ci2. That is, a voltage doubler rectifier circuit is provided.

On the secondary side, two secondary-side series resonance capacitors Cs1 and Cs2 are provided as shown. Also, four rectifying diodes DO1, DO2,
It comprises a series connection circuit of DO3, DO4 and two smoothing capacitors COA-COB. Then, by connecting these elements as shown in the figure, a quadruple voltage rectifier circuit is formed.

This quadruple voltage rectifier circuit has a [secondary winding N
2, secondary side series resonance capacitor Cs1, rectifier diode D
O1, DO2, smoothing capacitor COA] and a voltage doubler rectifier circuit composed of [secondary winding N2, secondary side series resonance capacitor Cs2, rectifier diodes DO3, DO4, and smoothing capacitor COB]. Are formed so as to be stacked in series. In other words, this 4
In the voltage doubler rectifier circuit, a rectified smoothed voltage corresponding to twice the level of the alternating voltage of the secondary winding N2 is obtained at both ends of each of the smoothing capacitors COA and COB by the two voltage doubler rectifier circuits. Therefore, a secondary side DC output voltage EO corresponding to a level four times the alternating voltage of the secondary winding N2 is obtained at both ends of the series connection circuit of the smoothing capacitors COA-COB. In this case, the number of turns of the secondary winding N2 can be reduced to about 1/4 of the usual number.
In the third embodiment, the input side is provided with the voltage doubler rectifier circuit as described above, and the output side is provided with the quadruple voltage rectifier circuit. It is something that is possible.

Even in such a configuration, FIG.
Is provided, and the connection between the primary winding N1 and the secondary winding N2 is made to be a polar connection, so that the same effects as those of the power supply circuits of FIGS. 1 and 5 can be obtained. Things.

In this embodiment, an orthogonal control transformer is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the orthogonal control transformer, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, for example, as in the case of the orthogonal control transformer, a three-dimensional structure is obtained by combining two sets of double U-shaped cores having four magnetic legs. Form a mold core. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, the control winding N
C and one of the drive windings NB are wound around two magnetic legs adjacent to each other among the four magnetic legs, and the other winding is wound diagonally. It is assumed that there is a positional relationship 2
It is wound around the magnetic legs of the book. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. As a result, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced. Further, as the power supply circuit of the present invention, for example, various combinations of a primary side voltage resonance type converter and a secondary side rectifier circuit can be considered in addition to those shown in the drawings.

[0060]

As described above, according to the present invention, an insulation converter transformer provided in a switching power supply circuit as a composite resonance type switching converter is provided with a so-called reverse winding for a primary winding and a secondary winding. The primary winding and the secondary winding are connected in polarity. In this structure, the primary magnetic flux and the secondary magnetic flux act so as to cancel each other out.Therefore, according to the present invention, it is not necessary to provide a gap for suppressing saturation in the core of the insulating converter transformer. Become. By eliminating the need for providing a gap to the core of the insulating converter transformer as described above, in the present invention, a step for forming a gap is omitted in the manufacture of the insulating converter transformer, and Management becomes easier. That is, the manufacturing efficiency of the power supply circuit including the insulating converter transformer is improved.

Further, since no gap is formed, the generation of fringe magnetic flux near this gap is also eliminated, so that significant heat generation and power loss in the primary winding and the secondary winding are achieved. In particular, depending on the structure of the insulated converter transformer of the present invention, the amount of current flowing through the primary winding and the secondary winding can be made smaller than before, which also suppresses the above-described heat generation and power loss. Can be further promoted.

Further, in the structure of the insulated converter transformer PIT according to the present invention, since the leakage inductance is reduced, the ZVS operation is ensured even under the condition of a low AC input voltage at the load, and the reliability as the power supply circuit is improved. Is done.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a cross-sectional view illustrating a structure of an insulating converter transformer provided in the switching power supply circuit of the present embodiment.

FIG. 3 is a waveform chart showing an operation of a main part in the switching power supply circuit of the present embodiment.

FIG. 4 is a waveform diagram for comparing ZVS operations of the present embodiment and a switching power supply circuit of the prior art.

FIG. 5 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a second embodiment of the present invention.

FIG. 6 is a circuit diagram illustrating a configuration example of a switching power supply circuit according to a third embodiment of the present invention.

FIG. 7 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

8 is a cross-sectional view showing a structure of an insulating converter transformer provided in the switching power supply circuit shown in FIG.

FIG. 9 is an equivalent circuit diagram showing each operation of the insulating converter transformer when mutual inductance is in a positive polarity mode and a negative polarity mode.

[Explanation of symbols]

1 control circuit, 2 switching driver, 3 oscillation circuit, 4 drive circuit, Q1 switching element, Cr
Primary parallel resonance capacitor, DD clamp diode, C2 secondary parallel resonance capacitor, Cs secondary series resonance capacitor, PIT isolation converter transformer,
PRT orthogonal control transformer

Claims (1)

[Claims] 1. A switching means formed with a switching element for switching and outputting an input DC input voltage, and a primary-side parallel resonance circuit having a voltage resonance type operation of the switching means is formed. And a primary side parallel resonant capacitor provided in such a manner as to obtain a required coupling coefficient that is loosely coupled between the primary side and the secondary side. An insulation converter transformer for transmitting to the secondary side, a secondary side parallel resonance circuit formed by connecting a secondary side parallel resonance capacitor in parallel to a secondary winding of the insulation converter transformer, and the insulation converter transformer The secondary side which is combined with the secondary side series resonance circuit formed by connecting the secondary side series resonance capacitor in series with the secondary winding of A secondary side DC output voltage is obtained by inputting an alternating voltage obtained in a secondary winding of the insulating converter transformer and performing a rectifying operation by forming a composite resonance circuit and the secondary side composite resonance circuit. DC output voltage generation means configured as described above, and constant voltage control means for performing constant voltage control by variably controlling the switching frequency of the switching element according to the level of the secondary DC output voltage The insulating converter transformer includes a core in which no gap is formed, and the primary winding and the secondary winding are wound around the core in winding directions opposite to each other, and A switching power supply circuit, wherein the winding and the secondary winding are connected in polarities.