JP2002027745A - Switching power supply - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce load power. SOLUTION: A secondary winding N2 for obtaining a DC output voltage E01 for a horizontal deflection output circuit, and a tertiary winding N3 for obtaining a DC output voltage E03 (200 V) for an image output circuit, are wound around the secondary side of a first insulation converter transformer PIT-1 constituted by a compound resonance type excellent in voltage conversion efficiency. In order to obtain the DC output voltage E03 for the image output circuit, an AC voltage generated in the tertiary winding N3 is subjected to half-wave rectification, and a DC voltage (65 V) generated on both ends of a filter capacitor C03 is applied to a DC output voltage (135 V) generated on both ends of a filter capacitor C01 by half-wave rectifying an AC voltage generated in the secondary winding N2 with a rectifier diode D01. As a result, the DC output voltage E03 for the image output circuit is obtained from the secondary side of the insulation converter transformer PIT-1. As a result, a switching power supply circuit can be constituted by using a compound resonance switching converter whose voltage conversion efficiency is superior to the conventional one, and power loss in a power supply can be reduced.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit suitable as, for example, a power supply for a television receiver.

[0002]

2. Description of the Related Art For example, in a color television receiver provided with a cathode ray tube (CRT: Cathode-ray Tube) for displaying an image, an electron beam emitted from an electron gun inside the CRT is horizontally (horizontally) directed. The power supply circuit mainly includes a horizontal deflection system circuit that deflects the current and a switching power supply of a soft switching type using a current resonance type converter.

FIG. 7 is a diagram showing the configuration of a horizontal deflection system circuit provided in a television receiver and its peripheral circuits. The switching power supply 10 shown in FIG. 7 is a DC-DC converter that performs switching of an input DC voltage, converts the DC voltage to a predetermined voltage level, and outputs the converted DC voltage. It is constituted by a circuit.

In the preceding stage of the switching power supply 10, a full-wave rectification type bridge rectifier circuit Di and a smoothing capacitor Ci are provided.
And a rectifying / smoothing circuit is provided. The rectifying / smoothing circuit rectifies and smoothes a commercial AC power supply (AC input voltage VAC) to obtain a DC voltage Ei, and the DC voltage Ei is input to the switching power supply 10. I have. The switching power supply 10 outputs a DC output voltage EO (EO1, EO2, EO4, EO5) converted to a predetermined voltage level.
In this case, for example, the DC output voltage EO1 output from the switching power supply 10 is a voltage for driving a horizontal deflection circuit of the television receiver, the DC output voltage EO2 is a voltage for driving a signal system circuit, and the DC output voltages EO4 and EO5 are audio. Each of the DC output voltages EO1, EO2, EO4,
The actual voltage levels of EO5 are, for example, 135V DC output voltage EO1, 15V DC output voltage EO2, and DC output voltage E0.
O4 and EO5 are set to ± 20V.

The horizontal output circuit 20 generates a horizontal deflection current IDY for scanning the electron beam emitted from the electron gun of the CRT in the horizontal direction, and generates a high voltage in a high voltage generation circuit 40 described later. It is configured to generate a flyback pulse.

For this reason, a horizontal synchronizing signal fH (15.73) of a video signal is supplied from a horizontal drive circuit (not shown) to the base of the horizontal output transistor Q11 of the horizontal output circuit 20.
45 kHz). The collector is a flyback transformer FB to be described later.
The switching power supply 10 is connected to a secondary output terminal (secondary DC output voltage EO1) of the switching power supply 10 through a primary winding N11 of T, and its emitter is grounded. Horizontal output transistor Q
A damper diode D1
1, horizontal retrace capacitor Cr1, and [horizontal deflection yoke H
DY, horizontal linear correction coil HLC, S-shaped correction capacitor CS
1] are connected in parallel.

The horizontal output circuit 20 having such a configuration is
Then, the capacitance of the horizontal retrace capacitor Cr1 and
A voltage resonance type converter is formed by the leakage inductance component of the primary winding N11 of the flyback transformer FBT. Then, the horizontal output transistor Q11 performs a switching operation by a pulse voltage input from a horizontal drive circuit (not shown), so that the horizontal deflection yoke H
A horizontal deflection current IDY having a sawtooth waveform flows through DY, and during a period in which the horizontal output transistor Q11 is turned off, the resonance operation of the inductance LDY of the horizontal deflection yoke HDY and the capacitance of the horizontal retrace capacitor Cr1 and the damper diode By the action of D11, the horizontal retrace capacitor Cr
A pulse voltage (flyback pulse voltage) V11 which is a relatively high voltage is generated at both ends of 1. Although not described in detail, the horizontal straight line correction coil HLC and the S-shaped correction capacitor CS1 correct, for example, the horizontal deflection current IDY to correct the distortion of the image displayed on the screen of the CRT.

A high-voltage generating circuit 40 shown by a dashed line.
Is a flyback transformer FBT (Fly Back Tra
nsformer) and a high-voltage rectifying / smoothing circuit, and boosts the flyback pulse voltage V11 generated by the horizontal output circuit 20 to generate a high voltage corresponding to the anode voltage level of the CRT.

A primary winding N11 is wound on the primary side of the flyback transformer FBT, and five sets of high-voltage windings NVH1, NVH2, NVH3, NHV4, NHV are wound on the secondary side thereof.
5 is divided and wound by slit winding or interlayer winding. On the primary side of the flyback transformer FBT, the tertiary winding N is tightly coupled to the primary winding N11.
12, N13 are wound. At this time, the high-voltage windings NHV1 to NHV5 are wound around the primary winding N11 so that the polarities (winding directions) thereof are opposite to each other, and the tertiary windings N12 and N13 have the same polarity with respect to the primary winding N11. It is wound so as to be polar.

In this case, the winding start end of the primary winding N11 is connected to the secondary output terminal (DC output voltage E) of the switching power supply 10.
O1), and the winding end is connected to the collector of the horizontal output transistor Q11. In addition, each high-voltage winding NHV1
To high voltage rectifier diodes DHV1 and DHV5
The anodes of HV2, DHV3, DHV4, and DHV5 are connected. Then, the cathode of the high-voltage rectifier diode DHV1 is connected to the positive terminal of the high-voltage capacitor CHV, and the cathodes of the high-voltage rectifier diodes DHV2 to DHV5 are connected to the winding start ends of the high-voltage windings NHV1 to NHV4, respectively.

[0011] In such a connection form, [high-voltage winding NVH
1, high-voltage rectifier diode DHV1, high-voltage winding NHV2, high-voltage rectifier diode DHV2, high-voltage winding NHV3, high-voltage rectifier diode DHV3, high-voltage winding NHV4, high-voltage rectifier diode DHV4, high-voltage winding NHV5, High voltage rectifier diode D
HV5] are connected in series to form a so-called multisingle-type half-wave rectifier circuit.

Therefore, on the secondary side of the flyback transformer FBT, these five sets of half-wave rectifier circuits
~ NHV5 is rectified to smoothing capacitor CH
By performing an operation of charging V, a DC voltage having a level corresponding to five times the voltage induced in each of the high-voltage windings NHV1 to NHV5 is obtained at both ends of the smoothing capacitor CHV. The DC high voltage EHV obtained at both ends of the CHV is used, for example, as the anode voltage of the CRT. The high-voltage windings NHV1 to NHV5 have 6
An induced voltage boosted to KV is obtained, and an anode voltage of 30 KV is obtained as the DC high voltage EHV.

The primary winding N11 of the flyback transformer FBT is provided with a tap, and a positive pulse voltage obtained from the tap is rectified and smoothed by a half-wave rectifying and smoothing circuit including a rectifying diode DO3 and a smoothing capacitor CO3. By doing so, a DC output voltage EO3 is obtained. The voltage level of the DC output voltage EO3 is, for example, 200
V and supplied to a cathode electrode of a cathode ray tube via a video signal amplifier (not shown).

The negative pulse voltage obtained from the tertiary winding N12 wound on the primary side of the flyback transformer FBT is supplied to the rectifier diode DO6 and the smoothing capacitor CO6.
The DC output voltages EO6 and EO7 are obtained by performing rectification and smoothing by a rectifying and smoothing circuit comprising a rectifying diode DO7 and a rectifying and smoothing circuit comprising a smoothing capacitor CO7. The DC output voltages EO6 and EO7 have voltage levels of +15 V and -15 V, respectively, and are output as voltages for a vertical deflection circuit (not shown). further,
The negative pulse voltage obtained from the tertiary winding N13 is rectified and smoothed by a rectifying and smoothing circuit including a rectifying diode DO8 and a smoothing capacitor CO8, thereby obtaining a DC output voltage EO8. The DC output voltage EO8 is, for example, 6.3 V
And output as a heater voltage for the CRT.

Here, FIG. 8 shows a configuration of a switching converter circuit which is currently frequently used as the switching power supply 10 shown in FIG. The switching power supply circuit shown in FIG. 8 employs a configuration including a current resonance type switching converter in which two switching elements (bipolar transistors) Q21 and Q22 are half-bridge-coupled.
The DC input voltage Ei from the full-wave rectifier circuit shown in FIG. 7 is input.

The gates of the switching elements Q21 and Q22 are connected to the oscillation / drive circuit 21. The drain of the switching element Q21 is connected to the DC input voltage line, and the source is grounded to the primary side ground via the series resonance capacitor C1 and the primary winding N21 of the insulating converter transformer PIT. In addition, each switching element Q2
1, clamp diodes DD11 and DD12 are connected in parallel between the drain and source of Q22, respectively. Here, since the series resonance capacitor C1 is connected in series to the primary winding N21 of the insulating converter transformer PIT as shown in the drawing for the switching elements Q21 and Q22, the capacitance of the series resonance capacitor C1 and the primary winding The leakage inductance component of the insulating converter transformer PIT including the line N21 forms a primary side series resonance circuit for making the operation of the switching converter a current resonance type.

The switching elements Q21 and Q22 are driven by the oscillation / drive circuit 21 so that a switching operation is obtained. Therefore, the control circuit 1 supplies a current or voltage of a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation / drive circuit 21. Then, in the oscillation / drive circuit 21, the switching drive signal (voltage) whose cycle is varied according to the output level from the control circuit 1 is switched to the switching elements Q21 and Q21 so that the DC output voltage EO1 is stabilized. Output is made to the gate of Q22. As a result, the switching frequency of the switching elements Q21 and Q22 is varied, and the DC output voltage EO1 is stabilized. Starting resistance RS
Is provided to supply a starting current obtained in the rectifying / smoothing line to the oscillation / drive circuit 21 when the commercial AC power is turned on.

The insulating converter transformer PIT transmits the switching output of the switching elements Q21 and Q22 to the secondary side. Although a detailed description is omitted here, a loose coupling state can be obtained by forming a gap with respect to the core in the insulating converter transformer PIT. On the secondary side of the insulation converter transformer PIT,
As shown, three sets of secondary windings N22, N23, N24 are wound independently of each other. Then, two rectifier diodes DO11 and DO12 and a smoothing capacitor CO1 are connected to the secondary winding N22 in a connection form as shown in the drawing, so that the DC output voltage EO1 (1
35V), and also connect the two rectifier diodes DO21, DO22 and the smoothing capacitor CO2 to the secondary winding N23 in the same connection form to obtain the DC output voltage EO2 (15V) of the signal system circuit. I have to. Further, the secondary winding N24 is provided with two sets of rectifying / smoothing circuits including rectifying / smoothing circuits including rectifying diodes DO41 and DO42 and a smoothing capacitor CO4, and rectifying / smoothing circuits including rectifying diodes DO51 and DO52 and a smoothing capacitor CO5. By connecting in the connection form as shown, DC output voltages EO4 (+ 20V) and EO5 (-20V) for the audio output circuit are obtained.

FIG. 9 shows the operation waveform of each part of the circuit shown in FIG. In the circuit shown in FIG. 7, a pulse voltage synchronized with the horizontal synchronizing signal fH (15.7345 kHz) of the video signal is input to the base of the horizontal output transistor Q11.
The switching frequency of 11 corresponds to the horizontal synchronizing signal fH, the on period (horizontal scanning period) Tt of the horizontal output transistor Q11 is 51.5 μs, and the off period (horizontal retrace period) Tr is 12 μs. Period T
t and the horizontal scanning period Tr (TH = 63.5 μm)
S) corresponds to the cycle of the horizontal synchronization signal fH.

In this case, due to the switching operation of the horizontal output transistor Q11, a primary current I11 having a waveform shown in FIG. 9B flows through the primary winding N11 of the flyback transformer FBT, and flows through the horizontal deflection yoke HDY. Fig. 9 (c)
A horizontal deflection current IDY having a waveform as shown in FIG. Also, a rectified current I3 having a waveform as shown in FIG. 9 (e) flows through the rectifier diode DO3 via a tap provided in the primary winding N11.

At this time, the voltage V11 across the horizontal retrace capacitor Cr1 connected between the collector and the emitter of the horizontal output transistor Q11 is, as shown in FIG. 9A, a period during which the horizontal output transistor Q11 is turned on. At Tt, the level becomes 0 level, and during the period Tr in which the horizontal output transistor Q11 is turned off, the inductance component L of the horizontal deflection yoke HDY is obtained.
The flyback pulse voltage V11 of, for example, about 1200 Vp is generated by the resonance operation of DY and the capacitance of the horizontal retrace capacitor Cr1. In the high-voltage generating circuit 40 shown in FIG. 7, the positive pulse voltage applied to the primary side of the flyback transformer FBT is boosted by the flyback pulse voltage V11 as described above, and the high-voltage winding on the secondary side is increased. Various DC output voltages of a predetermined voltage level are obtained from the lines NHV1 to NHV5 and the tertiary windings N12 and N13.

Further, as shown in FIG. 9D, a pulse voltage V3 of, for example, about 200 Vp is generated at both ends of the smoothing capacitor CO3 during the period Tr in which the horizontal output transistor Q11 is turned off. By rectifying and smoothing V3 with a rectifying diode DO3 and a smoothing capacitor CO3, a DC output voltage EO3 of, for example, 200 Vp is obtained.

[0023]

In the circuit shown in FIG. 7, in the high-voltage generating circuit 40, the DC output voltage EO3 (200 V) for the video output circuit from the primary side of the flyback transformer FBT and the vertical deflection circuit , And a DC output voltage EO8 (6.3 V) for a CRT heater. However, the flyback transformer FBT has a power conversion efficiency of about 85% when converting a DC voltage input from the switching power supply 10 into DC output voltages EO3, EO6, EO7, and EO8, and has a loss of about 15%. Electric power is generated. In addition, the load power of the DC output voltage EO1 (135 V) output from the switching power supply 10 increases, and the AC input power input to the switching power supply 10 increases.

In the high-voltage generating circuit 40, the DC output voltage E is obtained from the positive pulse voltage (flyback pulse voltage) input to the primary winding N11 of the flyback transformer FBT.
I try to get O3. However, in this case, since the conduction angle of the rectifier diode DO3 becomes narrower, for example, at the timing when the horizontal deflection current IDY becomes 0 level, as shown in FIG.
As shown in (e), since the steep peak current I3 flows through the rectifier diode DO3, the current I3 contains noise generated from the rectifier diode DO3. For this reason, when actually configuring the circuit, it was necessary to connect a ferrite bead inductor in series with the rectifier diode DO3 and further connect a ceramic capacitor or the like in parallel with the ferrite bead inductor as a measure against unnecessary radiation of noise. .

[0025]

Therefore, a switching power supply circuit according to the present invention is configured as follows in consideration of the above-mentioned problems. In other words, a switching unit formed with a switching element for intermittently outputting the input DC input voltage, and a primary-side parallel resonance circuit that makes the operation of the switching unit a voltage resonance type are formed. A primary side parallel resonance capacitor is provided, and is provided for transmitting the output of the primary side to the secondary side. A primary winding is wound on the primary side, and at least a first secondary winding is wound on the secondary side. The wire and the second secondary winding are wound, and the primary winding and the first
And a first insulating converter transformer capable of obtaining a required degree of coupling, which is loosely coupled. Then, a secondary parallel resonance capacitor is connected in parallel to the first secondary winding, and / or a secondary parallel resonance capacitor is connected to the winding composed of the first and second secondary windings. A secondary side parallel resonance circuit formed by connecting a capacitor in parallel, and formed including a secondary side parallel resonance circuit,
A first DC output voltage generating means configured to obtain a first DC output voltage by inputting an alternating voltage obtained to a first secondary winding and performing a rectification operation; A DC output voltage obtained by inputting an alternating voltage obtained in the next winding and performing a rectification operation is stacked on a first DC output voltage to obtain a second DC output voltage. A second DC output voltage generating means for variably controlling a switching frequency of the switching element according to a level of the first DC output voltage, and switching the switching element so as to vary an ON period in a switching cycle. Switching driving means for performing constant voltage control.

That is, according to the present invention, a second output for obtaining a DC output voltage for a horizontal deflection output circuit is provided on the secondary side of a first insulating converter transformer constituting a switching power supply circuit of a complex resonance type. One secondary winding (N2) and a second secondary winding (N3) for obtaining a DC output voltage for a video output circuit. When obtaining the DC output voltage for the video output circuit, the DC voltage obtained from the second secondary winding is stacked on the first DC output voltage so that the second DC output voltage for the video output circuit is obtained. By obtaining the output voltage, the load power is reduced.

[0027]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure is provided with a voltage resonance type converter having a single switching element Q1 and performing a switching operation by a so-called single-end system by a self-excited system. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1. Then, the commercial AC power supply (AC input voltage VAC) generates a rectified and smoothed voltage Ei corresponding to a level that is one time higher than the AC input voltage VAC by the bridge rectifier circuit Di and the smoothing capacitor Ci.

The base of the switching element Q1 is connected to a smoothing capacitor Ci (rectified smoothing voltage E) via a starting resistor RS.
It is connected to the positive electrode side of i) so that the base current at the time of startup is obtained from the rectification smoothing line. Between the base of the switching element Q1 and the ground on the primary side, a series resonance circuit for self-excited oscillation driving, which is composed of a series connection circuit of a drive winding NB, a resonance capacitor CB and a base current limiting resistor RB, is connected.
A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. The collector of the switching element Q1 is connected to one end of a primary winding N1 formed in the insulation converter transformer (first insulation converter transformer) PIT-1, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr is connected to its own capacitance and the insulation converter transformer PIT-
The primary parallel resonance circuit of the voltage resonance type converter is formed by the leakage inductance L1 on the primary winding N1 side. Although not described in detail here, when the switching element Q1 is turned off, the voltage V1 across the resonance capacitor Cr is obtained by the operation of the primary side parallel resonance circuit.
Is actually a sinusoidal pulse waveform so that a voltage resonance type operation can be obtained.

The orthogonal control transformer PRT is a saturable reactor on which a resonance current detecting winding ND, a driving winding NB, and a control winding NC are wound. This orthogonal control transformer P
RT drives switching element Q1 and is provided for constant voltage control. This orthogonal control transformer P
As a structure of the RT, although not shown, a three-dimensional core is formed by joining ends of two magnetic legs of two double U-shaped cores having four magnetic legs. Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is constructed by winding in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detection winding ND of the orthogonal control transformer PRT is connected to the positive electrode of the smoothing capacitor Ci,
By being inserted in series with the primary winding N1, the switching output of the switching element Q1 is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is applied to the drive winding NB. appear. This drive voltage is output from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

Insulation converter transformer (Power Isolation
Transformer) PIT-1 transmits the switching output of switching element Q1 to the secondary side. As shown in FIG. 5, the insulating converter transformer PIT-1 is provided with an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. The primary winding N1 and the secondary winding N2 are wound around the magnetic legs in a state of being divided using the divided bobbin B. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. The gap G can be formed by making the central magnetic legs of the E-shaped cores CR1 and CR2 shorter than the two outer magnetic legs. The coupling coefficient k is, for example, k ≒ 0.
A loose coupling state of 85 is obtained, so that a saturated state is hardly obtained.

Incidentally, the insulation converter transformer PIT
-1, the secondary winding N1 and the secondary winding N2
(Winding direction) and the connection relationship of the rectifier diode DO, and the polarity change of the alternating voltage excited in the secondary winding,
Regarding the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2, +
There are a case where the operation mode is M (additional polarity mode; forward operation) and a case where the operation mode is -M (depolarization mode; flyback operation). For example, when the circuit is equivalent to the circuit shown in FIG. 6A, the mutual inductance is + M, and when the circuit is equivalent to the circuit shown in FIG. 6B, the mutual inductance is -M. In the power supply circuit shown in FIG. 1, the primary winding N of the insulation converter transformer PIT-1 is provided.
1 and the polarity of the secondary winding N2 are in the operation mode of + M, the smoothing capacitor C is connected via the rectifier diode DO1.
It is assumed that the charging operation of O1 is performed.

The winding start end of the primary winding N1 of the insulated converter transformer PIT-1 is connected to the collector of the switching element Q1 as shown in FIG. 1, and the winding end is connected in series with the resonance current detection winding ND. Connected to the positive electrode (rectified smoothed voltage Ei) of the smoothing capacitor Ci. On the secondary side, a secondary winding N2 (first secondary winding), a tertiary winding N3 (second secondary winding) and a tertiary winding N4 are provided in an independent state. It is wound.

The winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end is connected to a rectifier diode D2.
O1 is connected to the anode of this rectifier diode D
A half-wave rectifying / smoothing circuit composed of O1 and a smoothing capacitor CO1 obtains a DC output voltage EO1 (for example, 135 V) for horizontal deflection of 100V to 140V. That is, in this case, the DC output voltage EO1 is generated by inputting the resonance voltage supplied from the secondary parallel resonance circuit to the rectifier diode DO1. Note that FIG.
Although this is not shown, this DC output voltage EO1 is supplied to a high-voltage generation circuit, where, for example, a CRT
To generate a DC high voltage (30 KV) which is used as an anode voltage of the DC power supply.

A tap is provided at a required position of the secondary winding N2, and a rectifier diode DO2
Anodes are connected. Then, a DC output voltage EO2 (15 V) for a signal circuit is obtained by a half-wave rectifying / smoothing circuit including a rectifier diode DO2 and a smoothing capacitor CO2.

In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2,
A parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2, and the alternating voltage induced in the secondary winding N2 becomes a resonance voltage, and the isolated converter transformer PIT-
A voltage resonance operation is obtained on the secondary side of the power supply.

That is, in the power supply circuit shown in FIG. 1, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is provided with a parallel resonance circuit for obtaining the voltage resonance operation. Be provided. In the present specification, a switching converter having a configuration in which a resonance circuit is provided for the primary side and the secondary side and operates as described above is described below.
It is also referred to as a “composite resonance type switching converter”. The configuration as such a composite resonance type switching converter is realized by forming a gap G with respect to the insulated converter transformer PIT and performing loose coupling by a required coupling coefficient as described above with reference to FIG. ,
This is further realized by obtaining a state that is unlikely to be saturated. For example, when the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to hope that this is done properly.

Further, the insulation converter transformer PIT-1
, The winding start end of the primary winding N1 is connected to the collector of the switching element Q1, and the winding end is connected to the rectified smoothing voltage line of the smoothing capacitor Ci. Further, the connection form is such that the winding start end of the secondary winding N2 is connected to the secondary-side ground, and the winding end is connected to the anode of the rectifier diode DO1. In the case of such a connection form, the magnetic flux of the primary winding N1 and the secondary winding N2
For example, to connect the starting end of the primary winding N1 to a rectified smoothing voltage line,
The magnetic flux density of the insulating converter transformer PIT-1 can be reduced as compared with the case where the winding start end of the secondary winding N2 is connected to the anode of the rectifier diode DO1. Therefore, in the case of such a connection form, a state in which the insulating converter transformer PIT is less likely to be saturated can be obtained, and as a result, the interval of the gap G provided in the center magnetic leg of the EE type core can be reduced. become.

The above-described DC output voltage EO1 is also branched and input to the control circuit 1. In the control circuit 1,
The DC output voltage EO1 is used as a detection voltage. The control circuit 1 varies the level of the control current (DC current) flowing through the control winding NC in accordance with the change in the DC output voltage level EO1 on the secondary side, whereby the drive wound around the orthogonal control transformer PRT is changed. The inductance LB of the winding NB is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This changes the switching frequency of the switching element Q1, thereby stabilizing the DC output voltage on the secondary side. Note that the DC output voltage EO3 is supplied to the control circuit 1.
To input and stabilize the DC output voltage on the secondary side.

In the power supply circuit shown in FIG. 1, when an orthogonal control transformer PRT for variably controlling the inductance LB of the drive winding NB is provided, a period during which the switching element Q1 is turned off for changing the switching frequency is used. With the TOFF constant, the ON period TON
Is variably controlled. In other words, in this power supply circuit, as the constant voltage control operation, the switching frequency is variably controlled to perform the resonance impedance control for the switching output, and at the same time, the conduction angle control of the switching element Q1 in the switching cycle ( PWM control). This complex control operation is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Further, the insulation converter transformer PIT-1
On the secondary side, together with the secondary winding N2, the tertiary windings N3, N4
Are wound, these tertiary windings N3, N4
Also, an induced voltage induced by the primary winding N1 is generated. In the power supply circuit shown in FIG. 1, a DC output voltage EO3 for a video output circuit is generated from an induced voltage induced in the tertiary winding N3. Tertiary winding N
3, a half-wave rectifying / smoothing circuit including a rectifying diode DO3 and a smoothing capacitor CO3 is formed. From this rectifying / smoothing circuit, a DC output voltage EO3 (200
In the power supply circuit shown in FIG. 1, by connecting the negative side of the smoothing capacitor CO3 to the positive side of the smoothing capacitor CO1, the video output from both ends of the series connection circuit of the smoothing capacitors CO1 to CO3 is obtained. A DC output voltage EO3 of the circuit is obtained.

That is, in the power supply circuit shown in FIG. 1, in order to obtain the DC output voltage EO3 of the video output circuit, the DC output voltage EO1 generated across the smoothing capacitor CO1 is added to the DC output voltage EO1 generated across the smoothing capacitor CO3. Stack up voltage,
That is, the DC output voltage EO1 obtained from the secondary winding N2,
The DC output voltage EO3 is obtained by adding the DC output voltages obtained from the tertiary winding N3. For this reason,
As a configuration of the rectifying / smoothing circuit including the tertiary winding N3, the rectifying diode DO3, and the smoothing capacitor CO3, a DC output of 90V to 60V obtained by subtracting a DC output voltage EO1 of 110V to 140V from a DC output voltage EO3 (200V). It is sufficient that a voltage can be obtained.

In the tertiary winding N4, the center tap provided on the tertiary winding N4 is connected to the connection point of the smoothing capacitors CO4-CO5 connected in series (the negative side of the smoothing capacitor CO4 and the positive side of the smoothing capacitor CO5). Connection point). The cathode of the rectifier diode DO4 connected to the winding end of the tertiary winding N3 is connected to the positive electrode of the smoothing capacitor CO4, and the anode of the rectifier diode DO5 connected to the winding start is connected to the negative electrode of the smoothing capacitor CO5. Side, DC output voltages EO4 and EO5 (± 20 V) for audio output are obtained from the anode.

Further, the power supply circuit shown in FIG.
An insulation converter transformer PIT-2 is provided as an insulation converter transformer. The primary winding N5 of the insulation converter transformer PIT-2 is connected in parallel with the secondary winding N2 of the insulation converter transformer PIT-1, and transmits the alternating voltage VN2 input to the primary winding N5 to the secondary side.
In this case, the winding end of the insulating converter transformer PIT-2 is connected to the winding end of the insulating converter transformer PIT-1, and the winding start is grounded to the secondary ground.

On the secondary side of the insulated converter transformer PIT-2, a secondary winding N6 tightly coupled to the primary winding N5 is provided.
(Third secondary winding) and a secondary winding N7 (fourth secondary winding) are wound independently of each other, and the primary windings are wound around these secondary windings N6 and N7. An induced voltage induced by the line N5 is generated. In the secondary winding N6, a center tap provided on the secondary winding N6 is grounded, and this center tap is connected to a smoothing capacitor CO6 connected in series.
-Connect to the connection point of CO7. The anode of the rectifier diode O6 is connected to the winding end of the secondary winding N6, and the cathode of the rectifier diode DO7 is connected to the winding start of the secondary winding N6, so that the anode of the rectifier diode DO6 and the anode of the rectifier diode DO7 are connected. DC output voltages EO6 and EO7 (± 15 V) for the vertical deflection circuit are obtained. In the secondary winding N7, a rectifier diode DO8
A DC output voltage EO8 (6.3 V) for the heater is obtained by a half-wave rectifying / smoothing circuit including a smoothing capacitor CO8 and a smoothing capacitor CO8.

As described above, in the power supply circuit shown in FIG. 1, the primary side is a voltage resonance type converter, and the secondary side is a composite resonance type switching converter having a half-wave rectification type voltage resonance circuit. At the same time, on the secondary side of the insulating converter transformer PIT-1, a secondary winding N2 for obtaining a voltage of, for example, 135 V as a DC output voltage EO1 for a horizontal deflection output circuit, and a DC output voltage EO3 for a video output circuit. (200 V) is wound around the tertiary winding N3. When the DC output voltage EO3 for the video output circuit is obtained, the alternating voltage generated in the tertiary winding N3 is half-wave rectified by the rectifying diode DO3, thereby obtaining the smoothing capacitor C03.
The DC output voltage (65 V) obtained at both ends of O3 is converted into a DC output voltage (135 V) obtained at both ends of the smoothing capacitor CO1 by half-wave rectifying the alternating voltage generated in the secondary winding N2 by the rectifier diode DO1. By stacking, the DC output voltage EO3 (200 V) for the video output circuit is obtained from the secondary side of the insulating converter transformer PIT-1.

Further, in the power supply circuit shown in FIG. 1, a second insulating converter transformer PIT-2 is provided, and the DC output voltage EO1 obtained from the first insulating converter transformer PIT-1 is supplied to the second insulating converter transformer PIT-2. -2 primary windings N5 and N6 wound on the secondary side as primary inputs
Output voltage EO6, EO7 for vertical deflection circuit directly from
(± 15V) and DC output voltage EO8 for driving the heater
(6.3 V). That is, in the power supply circuit shown in FIG. 1, the DC output voltages EO3, EO6 to EO8 which are obtained from the flyback transformer FBT in the conventional circuit shown in FIG. 1, PIT-2.

FIG. 2 shows operation waveforms of main circuit portions of the power supply circuit shown in FIG. In the power supply circuit shown in FIG. 1, the switching element Q1 performs a switching operation by a series resonance circuit (NB, CB) as a self-excited oscillation driving circuit.
Both ends of switching element Q1 (between collector and emitter)
The primary side parallel resonance voltage V1 as shown in FIG. 2A is obtained by the operation of the parallel resonance circuit. As shown in the figure, the voltage V1 between both ends of the switching element Q1 (collector-emitter voltage) is, for example, a peak voltage level of 600 Vp during a period TON when the switching element Q1 is on and zero during a period TOF when the switching element Q1 is off. Is obtained.

The insulation converter transformer PIT-1
The waveform of the alternating voltage VN2 generated in the secondary winding N2 of FIG.
FIG. 2B shows the waveform of the alternating voltage VN3 generated in the tertiary winding N3 as shown in FIG. In this case, since the conduction angle of the rectifier diode DO3 constituting the rectifying and smoothing circuit of the tertiary winding N3 increases, the current IN3 flowing through the rectifier diode DO3 becomes smooth as shown in FIG. A sinusoidal waveform is obtained, and noise included in the current IN3 does not occur.

According to the experiment, the size of the CRT was 34
The maximum load power of each part of the television receiver, which is assumed to be inches, is DC high voltage EHV (30 KV) / 60 W, DC output voltage EO3 (200 V) / 10 W for video output circuit, DC output voltage EO6 for vertical deflection circuit. , EO7 (± 15V) / 7
W, the DC output voltage EO8 (6.3 V) / 4 W for the heater, and the total maximum load power Pomax of the entire 34-inch television receiver is 81 W. Therefore, the maximum load power of the DC output voltages EO3, EO6, EO7, and EO8 excluding the DC high voltage EHV is set to 21 W.

The voltage conversion efficiency η of the conventional circuit shown in FIG.
Since DC-DC is about 85%, when such a power supply circuit is adopted in a 34-inch television receiver, 21W ÷ 0.85 = 24.7W is output from the switching power supply 10. Is the load power of the DC output voltage EO1.

On the other hand, in the power supply circuit shown in FIG. 1, since the voltage conversion efficiency η DC-DC of the composite resonance type switching converter is about 95%, the power supply circuit shown in FIG. In this case, 21W ÷ 0.95 = 22.1W is the load power of the power supply circuit shown in FIG. Therefore, it was found that if the power supply circuit shown in FIG. 1 was adopted for a 34-inch television receiver, a power loss of about 2.6 W could be reduced as compared with the case where the conventional circuit shown in FIG. 7 was applied. .

The current resonance type converter constituting the switching power supply 10 shown in FIG. 7 has an AC-DC power conversion efficiency ηAC-DC of about 90% and a DC output voltage EO.
1, EO2, EO4, EO5 total load power (Pom
ax) is 200 W, the AC input power is 222.2 W
Becomes In contrast, the power supply circuit shown in FIG.
-DC power conversion efficiency ηAC-DC is about 92%, and the maximum load power (Pomax) is reduced by about 2.6W from 200W to about 197.4W, so that the AC input power is about 214.6W. Become. Therefore, when the power supply circuit shown in FIG. 1 is compared with the conventional circuit shown in FIG. 7, the power supply circuit shown in FIG. 1 can reduce the AC input power by about 7.6 W, thereby saving energy. It becomes possible to plan.

Further, in the power supply circuit shown in FIG.
As shown in (d), the current IN3 flowing through the rectifier diode DO3 has a smooth sinusoidal waveform, and the DC output voltage EO3 is conventionally obtained from the flyback pulse voltage input to the primary winding N11 of the flyback transformer FBT. At this time, noise generated in the rectifier diode DO3 can be prevented. Therefore, there is an advantage that a ferrite bead inductor, a ceramic capacitor, and the like, which have been required to suppress unnecessary radiation of noise in the related art, become unnecessary.

Further, the power supply circuit of the present invention is not limited to the circuit configuration shown in FIG. FIG. 3 is a circuit diagram showing a configuration of a secondary side of a power supply circuit according to a second embodiment of the present invention. The configuration of the primary side of the power supply circuit shown in FIG. 3 is the same as the configuration of FIG. Also, the same parts as those in FIG.

The insulating converter transformer P shown in FIG.
On the secondary side of the IT, a tertiary winding N3 is exposed by winding up the secondary winding N2. The secondary winding N2 and the tertiary winding N3 (N2 + N3) are connected to the secondary side in parallel. The resonance capacitor C2 is connected in parallel. Also in this case, a DC output voltage EO1 (135 V) for a horizontal deflection circuit is obtained by a rectifying and smoothing circuit including a secondary winding N2, a rectifying diode DO1, and a smoothing capacitor CO1, and a tertiary winding is applied to the DC output voltage EO1. The DC output voltage (65 V) obtained by the rectifying and smoothing circuit including the line N3, the rectifying diode DO3, and the smoothing capacitor CO3 is stacked.
That is, by adding the DC output voltage E for the video output circuit,
O3 (200V) is obtained.

The secondary winding N2 is provided with two taps T1 and T2. The output obtained from the tap T1 is rectified and smoothed by a rectifier diode DO6 and a smoothing capacitor CO6, thereby providing a DC output voltage for a vertical deflection circuit. EO6 (+15 V) is obtained, and the DC output voltage EO6 is made constant by a three-terminal regulator Q3 or the like, thereby obtaining a DC output voltage EO2 (12 V) for a signal system circuit.

A tap T2 provided on the secondary winding N2
Is rectified and smoothed by a rectifying diode DO8 and a smoothing capacitor CO8, and the rectified and smoothed output is made constant by a three-terminal regulator Q2 to obtain a DC output voltage EO8 (6.3 V) for a heater. ing.

Further, on the secondary side of the insulating converter transformer PIT, a tertiary winding N8 (fifth secondary winding) is formed so as to wind up the winding start end of the secondary winding N2. The output of the winding N8 is rectified and smoothed by the rectifier diode D07 and the smoothing capacitor C07 to obtain a DC output voltage EO7 (-15 V) for the vertical deflection circuit.

When the secondary side of the power supply circuit is constructed as described above, the capacitance of the secondary-side parallel resonance capacitor C2 is determined as shown in FIG.
In the power supply circuit shown in FIG. 3, 0.01 μF is selected, whereas in the power supply circuit shown in FIG. 3, only 4700 pF is required. In this case, the number of turns of the secondary winding N2 is 45.
T (turn), the number of turns of the tertiary winding N3 is 22 turns.

The secondary winding N2 and the tertiary winding N3 (N2 +
N3), a secondary parallel resonance capacitor C2 is connected in parallel, and a secondary parallel resonance capacitor C2A as shown by a broken line is connected in parallel to both ends of the secondary winding N2 to form a secondary parallel resonance capacitor. In this case, the secondary side parallel resonance capacitor C2 is 2200p
F, 4700 pF is selected for the secondary side parallel resonance capacitor C2A.

Operation waveforms of the power supply circuit shown in FIG. 3 are as shown in FIG. Here, the secondary parallel resonance capacitor C2A is connected in parallel with the secondary winding N2-tertiary winding N3 without connecting the secondary parallel resonance capacitor C2A in parallel to both ends of the secondary winding N2 of the insulating converter transformer PIT. In this case, the alternating voltage VN3 generated between the secondary winding N2 and the tertiary winding N3 is shown in FIG. 4A, and its peak voltage is 700 Vp. At that time, the current I2 flowing through the secondary winding N2 is shown in FIG. 4B, and its peak current is 6 App.

On the other hand, by adding a parallel connection of a secondary parallel resonance capacitor C2A to both ends of the secondary winding N2, and dividing the secondary parallel resonance capacitor by C2 and C2A, , The alternating voltage VN3 is shown as in FIG. 4 (c), the peak voltage of which is reduced to 600 Vp, and the peak value of the current I2 flowing through the secondary winding N2 is as shown in FIG.
As shown in (d), it is reduced to 5 App. Therefore, when the secondary parallel resonance capacitor is divided by C2 and C2A, the secondary parallel resonance capacitor C2
In comparison with the case of only the configuration, the heat generation of the secondary winding N2 and the tertiary winding N3 can be reduced by about 10 ° C., whereby the AC input power can be reduced by 0.7 W.

Therefore, when comparing the power supply circuit shown in FIG. 3 with the conventional circuit shown in FIG. 7, the power supply circuit shown in FIG. 3 shows the power loss of the AC input power described above with reference to FIG. 7.6W reduction, secondary winding N2 and tertiary winding N3
Is added by the power loss reduction of 0.7 W due to the decrease in the heat generation of the AC power supply, and the AC input power can be reduced by 8.3 W (7.6 W + 0.7 W).

In the present embodiment, the DC output voltage EO3 for the video output circuit, the DC output voltage EO6 for the vertical deflection circuit,
EO7 and a DC output voltage EO8 for driving the heater are obtained. However, the present invention provides at least a DC output voltage EO3 for a video output circuit which consumes a large amount of power and generates noise. The effect can be expected simply by constructing it from the trans PIT.

Further, in the present embodiment, an orthogonal control transformer is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the orthogonal control transformer, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, for example, as in the case of the orthogonal control transformer, a three-dimensional structure is obtained by combining two sets of double U-shaped cores having four magnetic legs. Form a mold core. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, the control winding N
C and one of the drive windings NB are wound around two magnetic legs adjacent to each other among the four magnetic legs, and the other winding is wound diagonally. It is assumed that there is a positional relationship 2
It is wound around the magnetic legs of the book. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. As a result, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

[0068]

As described above, the switching power supply circuit of the present invention is provided on the secondary side of the first insulating converter transformer constituted by the composite resonance type for obtaining the DC output voltage for the horizontal deflection output circuit. One secondary winding and a second secondary winding for obtaining a DC output voltage for a video output circuit are wound. When obtaining the DC output voltage for the video output circuit, the DC output voltage obtained from the second secondary winding is stacked on the first DC output voltage, so that the second DC output voltage for the video output circuit is obtained. To obtain a DC output voltage. Since such a switching power supply circuit of the present invention is constituted by a composite resonance type switching converter which is considered to have a higher voltage conversion efficiency than the conventional one, it is possible to apply the switching power supply circuit of the present invention to a television receiver. If
Power loss in the power supply circuit can be reduced. Further, as the power loss in the switching power supply circuit is reduced, the AC input power can also be reduced, so that energy can be saved. Furthermore, if the secondary side parallel resonance capacitors are divided and connected in parallel between the first secondary winding and the first secondary winding and the second secondary winding, the first Further, since the heat generated by the second secondary winding can be reduced, it is possible to further reduce the AC input power.

Further, by obtaining the second DC output voltage from the second secondary winding wound on the secondary side of the first insulating converter transformer, the second DC voltage generating means can be used. Since the conduction angle of the rectifier diode increases, the rectifier current flowing through the rectifier diode has a smooth sinusoidal waveform and no noise is generated. As a result, there is also an advantage that a ferrite bead inductor, a ceramic capacitor, and the like, which are conventionally required to suppress unnecessary radiation of noise, are not required.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 3 is a diagram illustrating a circuit configuration on a secondary side of a power supply circuit according to a second embodiment;

FIG. 4 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 5 is a cross-sectional view showing a structure of the insulating converter transformer according to the present embodiment.

FIG. 6 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 7 is a diagram showing a configuration of a conventional television power supply circuit.

8 is a circuit diagram showing a configuration of a switching power supply provided in the television power supply shown in FIG.

9 is a waveform chart showing an operation of a main part of the television power supply circuit shown in FIG.

[Explanation of symbols]

1 control circuit, AC commercial AC power supply, Ci smoothing capacitor, Cr primary side parallel resonance capacitor, C2 C2A
Secondary side parallel resonance capacitor, CB resonance capacitor, CO
1 ~ CO8 Smoothing capacitor, Di bridge rectifier circuit,
DD clamp diode, DO1 to DO8 rectifier diode, EO1 to EO8 DC output voltage, N1 N5 primary winding, N
2 N6 N7 N8 secondary winding, N3 N4 N5 tertiary winding, NB drive winding, NC control winding, PIT-1 PIT
-2 Insulated converter transformer, PRT orthogonal control transformer, Q1 switching element, Q2 Q3 three-terminal regulator, RS starting resistor, RB base current limiting resistor, T1 T2 tap

 ──────────────────────────────────────────────────続 き Continued on front page F term (reference) 5C068 AA06 CA03 CA07 CB03 5H730 AA02 AA14 AA16 AS01 BB26 BB43 BB52 BB65 BB66 BB67 BB82 BB88 BB94 CC01 DD02 DD04 DD26 DD35 EE02 EE07 EE65 EE73 EE74 FD07

Claims (9)

[Claims] 1. A switching means formed with a switching element for intermittently outputting an input DC input voltage, and a primary-side parallel resonance circuit having a voltage resonance type operation of the switching means. A primary-side parallel resonant capacitor provided in such a manner that the primary-side output is transmitted to the secondary side, a primary winding is wound on the primary side, and the secondary side has at least The first secondary winding and the second secondary winding are wound, and a required degree of loose coupling is obtained for the primary winding and the first secondary winding. And a secondary parallel resonance capacitor connected in parallel to the first secondary winding, and / or the first and second secondary windings Secondary parallel resonance capacitor for the winding consisting of And a secondary-side parallel resonance circuit formed by connecting them in parallel; and a rectifying operation formed by inputting an alternating voltage obtained in the first secondary winding and including the secondary-side parallel resonance circuit. Performing a rectifying operation by inputting a first DC output voltage generating means configured to obtain a first DC output voltage, and an alternating voltage obtained in the second secondary winding. A second DC output voltage generating means configured to obtain a second DC output voltage by stacking the DC output voltage obtained by the first DC output voltage on the first DC output voltage; The switching frequency of the switching element is variably controlled in accordance with the voltage level, and the constant voltage control is performed by switchingly driving the switching element such that the ON period in the switching cycle is varied. Switching power supply circuit, characterized in that it is configured with a constant voltage control means that, the. 2. The first DC output voltage is transmitted to a secondary side as a primary side input, and a third secondary winding and a fourth secondary winding are wound around the secondary side. A third DC output voltage is obtained by inputting an alternating voltage obtained to the third secondary winding and performing a rectification operation by inputting the alternating voltage obtained in the third secondary winding. And a fourth DC output configured to obtain a fourth DC output voltage by performing a rectifying operation by inputting the alternating voltage obtained in the fourth secondary winding. The switching power supply circuit according to claim 1, further comprising: a voltage generation unit. 3. A fifth secondary winding is wound on the secondary side of the first insulating converter transformer together with the first and second secondary windings. The alternating voltage obtained by the first tap provided on the winding and the fifth secondary winding is input to perform a rectification operation, thereby obtaining a third DC output voltage. A third DC output voltage generating means, and an alternating voltage obtained from a second tap provided on the first secondary winding is input to perform a rectification operation, thereby converting the fourth DC output voltage. The switching power supply circuit according to claim 1, further comprising: a fourth DC output voltage generation means configured to obtain the DC output voltage. 4. The switching power supply circuit according to claim 1, wherein said first secondary winding and said second secondary winding are formed independently of each other. 5. The switching power supply circuit according to claim 1, wherein said second secondary winding is formed so as to wind up said first secondary winding. 6. The switching power supply circuit according to claim 1, wherein the first DC output voltage is a voltage for a horizontal deflection circuit of a television receiver. 7. The switching power supply circuit according to claim 1, wherein the second DC output voltage is a voltage for a video output circuit of a television receiver. 8. The switching power supply circuit according to claim 2, wherein the third DC output voltage is a voltage for a vertical deflection circuit of a television receiver. 9. The switching power supply circuit according to claim 2, wherein the fourth DC output voltage is a heater voltage of a television receiver.