JP2001218458A - High-voltage stabilized circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve power conversion efficiency, and realize downsizing, and cost reduction for a high-voltage stabilized circuit. SOLUTION: A pair of voltage resonance converter is provided on the primary of an isolation converter transformer to obtain high DC voltage (CRT anode voltage) through a multiplying-rectifier circuit on the secondary side thereof. By providing a parallel resonance circuit on the secondary, a composite resonance converter is formed for the whole high-voltage stabilized circuit. The stabilization of the high DC voltage is performed by composite control of simultaneously controlling the switching frequency and continuity angle of the primary side voltage resonance converter according to the level of the high DC voltage. It is thus possible to reduce the number of the switching converters to be attached on the primary to only one pair.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a method for obtaining an anode voltage to be applied to, for example, an anode electrode of a cathode ray tube.
The present invention relates to a high-voltage stabilization circuit that generates a stabilized high voltage from a commercial AC power supply.

[0002]

2. Description of the Related Art In television receivers, projectors, and display devices for personal computers, those employing a cathode ray tube (hereinafter referred to as a CRT (Cathode Ray Tube)) as a display device have become widespread. .

As is well known, in various display devices having such a CRT, it is necessary to stably supply a high voltage (anode voltage) at a required level to an anode electrode of the CRT. That is, the anode voltage level is generally set to about 25 KV to 35 KV, but this anode voltage fluctuates according to, for example, an AC input voltage or a load fluctuation. The vertical and horizontal screen sizes of the image displayed on the CRT also change due to the change in the anode voltage. Further, when a high-luminance white peak image is displayed on a screen, a phenomenon occurs that the image is distorted. For example, as shown in FIG. 11, if a rectangular white peak image originally indicated by a solid line is displayed, the image is distorted so as to have a trapezoidal shape as indicated by a broken line. Therefore, a so-called high voltage stabilizing circuit configured to stabilize and output the anode voltage is provided as a circuit for supplying the anode voltage.

FIG. 8 shows an example of a high-voltage stabilizing circuit. The high-voltage stabilizing circuit shown in FIG. 1 is provided in a display device compatible with so-called multi-scan as a display device for a personal computer.

FIG. 8 shows a rectifying and smoothing circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. This rectifying and smoothing circuit performs rectifying and smoothing on the commercial AC power supply AC, and obtains a rectified and smoothed voltage Ei corresponding to the same level as the commercial AC power supply AC level. This rectified smoothed voltage Ei is supplied to the switching power supply circuit unit 10 as a DC input voltage.

The switching power supply circuit section 10 receives a rectified and smoothed voltage Ei to perform switching and stabilization, thereby obtaining a DC output voltage Eo.
The converter is a C converter, which outputs a DC output voltage Eo stabilized at 240V.

[0007] The DC output voltage Eo is input to the step-down converter 20. This step-down converter 20
Are connected between the line of the DC output voltage Eo and the positive terminal of the smoothing capacitor COA, for example, between the drain-source of the MOS-FET switching element Q11 and the choke coil CH.
1 are connected in series in order, and a diode DD1 is inserted between a connection point between the source of the switching element Q11 and the choke coil CH1 and the primary side ground. The switching element Q11 is driven in a separately excited manner by a drive voltage from a first drive circuit 14, which will be described later, to perform switching with respect to the DC output voltage Eo.
The smoothing capacitor COA is charged through the choke coil CH1 and the diode DD1. Then, as the output of the step-down converter 20, a step-down DC voltage EOA which is a voltage across the smoothing capacitor COA is obtained.

The step-down DC voltage EOA is supplied to the voltage resonance type converter 30. The voltage resonance type converter 30 shown in this figure includes a single MOS-FET switching element Q12 and performs single-ended operation by separately-excited. In the voltage resonance type converter 30, the drain of the switching element Q12 is connected to the positive terminal of the step-down DC voltage EOA via the choke coil CH2, and the source is connected to the primary side ground. The gate is supplied with a drive voltage output from a second drive circuit 16 described later. The switching element Q12 is switched by this drive voltage.

The drain of the switching element Q12 is connected to a primary winding N of a flyback transformer FBT described later.
Connected to the start end of winding 1. In this case, the winding end end of the primary winding N1 is grounded to the primary side ground via the DC blocking capacitor C11. As a result, the switching output of switching element Q12 is connected to flyback transformer F
The power is transmitted to the primary winding N1 of the BT, and an alternating voltage corresponding to the switching frequency is obtained in the primary winding N1.

The drain of the switching element Q12
A parallel resonance capacitor Cr is connected in parallel between the sources. The parallel resonance capacitor Cr is connected to its own capacitance and the inductance L12 of the choke coil CH2.
The primary parallel resonance circuit of the voltage resonance type converter is formed by the leakage inductance L1 of the primary winding N1 side of the flyback transformer FBT. Although the detailed description is omitted here, when the switching element Q12 is turned off, the voltage V2 across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit, and the voltage resonance type Operation is obtained. Further, a clamp diode DD2 is connected in parallel between the drain and source of the switching element Q12, thereby forming a path for a clamp current flowing when the switching element Q12 is turned off.

Here, the configuration for switching driving of the switching elements Q11 and Q12 on the primary side is as follows. In the synchronous circuit 11,
Based on the horizontal synchronization signal frequency fH corresponding to the currently set resolution, a horizontal synchronization signal having this frequency fH is generated and output. In this case, since it is a multi-scan, the horizontal synchronizing signal frequency fH can be varied in a range of, for example, 30 KHz to 120 KHz. In this case, the horizontal synchronizing signal generated by the synchronizing circuit 11 is input to the oscillating circuit 12, where it is converted into an oscillating frequency signal used to drive the switching elements Q11 and Q12, and is subjected to PWM control. Output to the circuit 13 and the second drive circuit 16.

The second drive circuit 16 generates a drive voltage for driving the switching element Q12 from the input oscillation frequency signal, and outputs it to the gate of the switching element Q12. Therefore, the switching frequency of the switching element Q12 matches the currently set horizontal synchronizing signal frequency fH.

The PWM control circuit 13 performs PWM control on the input oscillation frequency signal in accordance with the detection output from the error amplifier circuit 15. That is, the duty of the oscillation frequency signal in the ON / OFF period within one cycle is variably controlled and output to the first drive circuit 14. The first drive circuit 14 generates a drive voltage using the PWM oscillation frequency signal output from the PWM control circuit 13 and outputs the drive voltage to the switching element Q11.
Therefore, the switching frequency of the switching element Q11 also matches the currently set horizontal synchronizing signal frequency fH. That is, both the switching element Q11 and the switching element Q12 perform a switching operation at a switching frequency synchronized with the horizontal synchronization signal frequency fH. However, the duty of the switching element Q12 in the ON / OFF period within one switching cycle follows the waveform (one cycle duty) of the oscillation frequency signal changed by the PWM control circuit 13. This involves stabilizing the level of the DC high voltage EHV, that is, relating to constant voltage control, which will be described later.

The switching output obtained from the switching element Q12 of the voltage resonance type converter 30 is supplied to a high voltage generation circuit 40. High voltage generation circuit 40
Is formed with a flyback transformer FBT and a high-voltage rectifier circuit provided on the secondary side thereof, and the switching output of the switching element Q12 is transmitted to the primary winding N1 of the flyback transformer FBT. .

As shown in the drawing, the flyback transformer FBT has a primary winding N1 wound on the primary side. Also,
On the secondary side, five sets of boost windings NVH1,
NHV2, NHV3, NHV4, and NHV5 are wound. These boost windings NHV1 to NHV5 are actually divided and wound around the core in an independent state. These boost windings N
HV1 to HV5 are wound so as to have the opposite polarity with respect to the primary winding N1, so that a flyback operation can be obtained.

These boost windings NVH1, NVH2, NHV3, NH
As shown in the figure, V4 and NHV5 are connected in series with each of the high-voltage rectifier diodes DHV1, DHV2, DHV3, DHV4, and DHV5 to form a total of five sets of half-wave rectifier circuits. A half-wave rectifier circuit is further connected in series. Then, a smoothing capacitor COHV is connected in parallel with the series connection of these five half-wave rectifier circuits.

Therefore, on the secondary side of the flyback transformer FBT, five sets of half-wave rectifier circuits are provided with the step-up windings NVH1 to NHV1.
Rectify the voltage induced in NVH5 and smoothing capacitor COHV
Is charged. As a result, a DC voltage having a level corresponding to five times the voltage induced in each of the boost windings NVH1 to NHV5 is obtained at both ends of the smoothing capacitor COHV. The DC voltage obtained between both ends of the smoothing capacitor COHV is used as a DC high voltage EHV, and is used as, for example, an anode voltage of a CRT.

Next, the constant voltage operation of the circuit shown in FIG. 1 will be described. A voltage dividing resistor R1 is connected to both ends of the smoothing capacitor COHV provided on the secondary side of the flyback transformer FBT.
A series connection circuit of -R2 is connected in parallel. Therefore, a voltage level obtained by dividing the DC high voltage EHV is obtained at the connection point of the voltage dividing resistors R1 and R2 according to the voltage dividing ratio. The connection point of the voltage dividing resistors R1 and R2 is connected to the input of the error amplifier 15, and the error amplifier 15 compares the level of the divided DC high voltage EHV with a predetermined reference level. , The error is detected. That is, an error of the DC high voltage EHV level with respect to the predetermined reference level is detected. Then, for example, a DC current or a DC voltage of a level that is varied according to the error amount is output.

The detection output of the error amplifier 15 is supplied to a PWM control circuit 13. The PWM control circuit 13 performs PWM on the input oscillation frequency signal based on the detection output of the error amplifier 15 and outputs the signal to the first drive circuit 14. Accordingly, the switching element Q11 driven by the first drive circuit 14 is fixed at a switching frequency synchronized with the horizontal synchronizing signal frequency, and is then turned on / off by PWM control according to the DC high voltage EHV level fluctuation. The switching operation is performed according to the duty ratio in the off period. Here, the level of the step-down DC voltage EOA is the same as the level of the DC output voltage Eo and the switching element Qo.
The switching frequency of the switching element 11 is determined by the duty of the on / off period within one switching cycle.
If the ON period in one cycle is TON1, then EOA = Eo · TO
It is represented by N1 / Ts. Therefore, as described above, the switching element Q11
If the duty ratio of the on / off period within one switching cycle of the above is varied by PWM control, the step-down DC voltage EOA
Can be variably controlled.

FIG. 9 shows operation waveforms of the switching elements Q11 and Q12 in the circuit shown in FIG. FIGS. 9A to 9D show the horizontal synchronization signal frequency fH = 30.
9 (e) to 9 (h) show the operation when the horizontal synchronizing signal frequency fH = 120KHz for the same parts as those in FIGS. 9 (a) to 9 (d). It is shown. FIGS. 9A and 9E show the voltage V1 between the drain and source of the switching element Q11, and FIGS. 9B and 9F show the switching current I1 flowing through the drain of the switching element Q11. FIG.
(C) and (g) show the voltage (parallel resonance voltage) V2 across the parallel resonance capacitor Cr connected in parallel to the switching element Q12, and FIGS. 9 (d) and (h) show the drain of the switching element Q11. Is shown.

Here, the horizontal synchronizing signal frequency fH = 30
As an operation at KHz, as shown in FIGS. 9A and 9B, one switching cycle Ts of the switching element Q11 is performed.
Is 3.3 μs, and the PWM control is performed such that the period TON1 during which the switching element Q11 is on is 6.9 μs. At this time, the level of the step-down DC voltage EOA is about 50V.

The horizontal synchronizing signal frequency fH = 120K
As an operation at Hz, as shown in FIGS. 9E and 9F, one switching cycle Ts of the switching element Q11 is set to 8.3 μs, and a period TON1 during which the switching element Q11 is turned on is set to 0.4 μs. 2 shows a state when the PWM control is performed. At this time, the step-down DC voltage E
The OA level is about 220V.

The voltage resonance type converter 30 which operates using the stepped-down DC voltage EOA as an operation power supply is as follows.
As shown in the operation of the switching element Q12 as shown in FIGS. 9C, 9D and 9G, 9H, the period TOFF2 during which the switching element Q12 is turned off regardless of the change of the horizontal synchronizing signal frequency fH. Is constant at 3 μs. This can be set by selecting the capacitance of the parallel resonance capacitor Cr.

As described above, for the switching operation of the switching element Q11, the PWM control is performed according to the fluctuation of the DC high voltage EHV level, and the off period TOFF2 of the switching element Q12 is fixed at 3 μs. , The parallel resonance voltage V2 is
As shown in (c), control can be performed so as to be constant with respect to changes in the horizontal synchronizing signal frequency fH. The parallel resonance voltage V2 is transmitted as a switching output to the primary winding N1 of the flyback transformer FBT. Therefore, when the parallel resonance voltage V2 is controlled to be constant, the alternating voltage level obtained in the primary winding N1 of the flyback transformer FBT is also maintained to be constant. And the flyback transformer F
The DC high voltage EHV level generated on the secondary side of the BT is also controlled to be constant. In this way,
In the high voltage stabilizing circuit shown in FIG. 8, the DC high voltage EHV level is stabilized.

As a summary of the operations described above, FIG. 10 shows the fluctuation characteristics of the DC high voltage EHV, the DC output voltage Eo, and the step-down DC voltage EOA with respect to the horizontal synchronizing signal frequency fH. According to this figure, first, the DC output voltage Eo is stably obtained at a level of 240 V by the stabilizing operation of the switching power supply circuit unit 10.

The step-down DC voltage EOA is controlled so as to linearly change in the range of about 50 V to 220 V with respect to the fluctuation range of the horizontal synchronizing signal frequency fH = 30 KHz to 120 KHz. Here, the step-down DC voltage EOA
3 shows a case where the load current IHV supplied to the load by the DC high voltage EHV is 1 mA and a case where the load current IHV is 0 mA. Although the step-down DC voltage EOA tends to be slightly lower when the load current IHV = 1 mA than when the load current IHV = 0 mA, substantially the same voltage value is obtained.
Then, the level of the step-down DC voltage EOA is controlled as described above, and as described above, the parallel resonance voltage V2 generated by the switching operation of the switching element Q12 changes (changes) the horizontal synchronizing signal frequency fH. Is set to be constant with respect to the DC high voltage E
HV is constant at about 27 KV, for example.

In the high voltage stabilizing circuit shown in FIG. 8, by stabilizing the DC high voltage EHV in this way, the CR
The vertical and horizontal screen sizes of the image displayed on T are prevented from changing. In the case of the configuration of the high-voltage stabilizing circuit shown in FIG. 8, the trapezoidal distortion of the high-luminance rectangular white image is caused by the increase in the high-voltage load current IHV that occurs at the white peak. This is caused by the fluctuation component ΔEHV.
Countermeasures are being taken by selecting capacitors with O, COA, and COHV capacitances increased as required.

[0028]

However, the high voltage stabilizing circuit having the structure shown in FIG. 8 has the following problems. First, the high-voltage stabilization circuit having the configuration shown in FIG. 8 performs a two-stage power conversion by the step-down converter 20 and the voltage resonance type converter 30 in the process from input of the commercial AC power to generation of the DC high voltage EHV. Is performed. For this reason, one reason is that the overall power conversion efficiency as a whole of the high-voltage stabilization circuit is low. For example, actually, the power when the high-voltage load power PHV of the DC high voltage EHV = 27 W (= EHV × IHV) The conversion efficiency is only about 72%, and about 10.5 W is reactive power. The input power is correspondingly large.

Further, in such a configuration, the circuit to be actually assembled has a complicated structure, and accordingly, the number of parts increases. For example, specifically, two sets of ferrite transformers (FBT and an insulated converter transformer in the switching power supply circuit unit 10), two sets of ferrite choke coils (CH1, CH2), and a minimum of three sets of switching elements ( Q11 and Q12, switching elements in the switching power supply circuit section 10, etc. are required. For this reason, for example, the mounting area of the printed circuit board is increased, the circuit size itself is increased, and the cost is correspondingly high. In addition, as described above, in order to suppress the trapezoidal distortion of a high-brightness rectangular white image, the capacitance of each of the smoothing capacitors CO, COA, and COHV is increased. This has led to increased costs and increased costs.

[0030]

SUMMARY OF THE INVENTION In view of the above problems, the present invention is configured as a high voltage stabilizing circuit as follows. That is, rectifying and smoothing means for obtaining a DC input voltage by rectifying and smoothing a commercial AC power supply, switching means having a switching element for intermittently outputting a DC input voltage, and transmitting an output on a primary side to a secondary side. Provided in
The primary winding, the first secondary winding, and the second secondary winding are wound, and a required degree of loose coupling is established between the primary winding and the first secondary winding. To be obtained, the first
An isolated converter transformer configured to obtain a tightly coupled state with respect to the secondary winding and the second secondary winding,
A primary-side parallel resonance circuit formed so that the operation of the switching means is of a voltage resonance type. Also, the first
A secondary parallel resonance circuit formed with a secondary winding of
DC output voltage generating means for obtaining a secondary side DC output voltage by inputting an alternating voltage obtained in the first secondary winding and performing a rectification operation, and obtaining a DC output voltage in the second secondary winding. DC high voltage generating means which is adapted to obtain a DC high voltage which is a predetermined high level by inputting an alternating voltage and performing a rectifying operation, and a switching frequency of a switching element according to the level of the DC high voltage. And a constant voltage control means for performing constant voltage control on the DC high voltage by making it variable.

As described above, the high-voltage stabilization circuit of the present invention includes a parallel resonance circuit for making the switching converter a voltage resonance type on the primary side and a first secondary winding on the secondary side. A circuit configuration as a complex resonance type converter including a parallel resonance circuit formed including a wire is employed. For the DC high voltage, a second secondary winding (boost winding) is wound around the secondary side of the insulating converter transformer,
The alternating voltage obtained in the second secondary winding is obtained by inputting it to a DC high voltage generating means. The switching frequency of the primary-side voltage resonance type converter is variably controlled in accordance with the level of the DC high voltage, thereby stably stabilizing the DC high voltage. In such a configuration, only one set of the voltage resonance type converter is used as the switching converter that performs power conversion on the DC input voltage obtained from the commercial AC power supply on the primary side.

[0032]

FIG. 1 is a circuit diagram showing the configuration of a high-voltage stabilizing circuit according to an embodiment of the present invention. The circuit shown in this figure is provided with a voltage resonance type converter having a single switching element Q1 on its primary side and performing a switching operation by a self-excited method in a so-called single-ended system. In this case, the switching element Q1 includes a high withstand voltage bipolar transistor (BJ).
T: junction type transistor).

In this case, a rectifying / smoothing circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci is connected to the commercial AC power supply AC, and generates a rectified / smoothed voltage corresponding to a level approximately equal to the AC input voltage VAC. Then, the DC voltage is supplied to the voltage resonance type converter as a DC input voltage.

The switching element Q1 has a base connected to a smoothing capacitor Ci (rectified smoothing voltage E) via a starting resistor RS.
It is connected to the positive electrode side of i) so that the base current at the time of startup is obtained from the rectification smoothing line. A drive winding N is provided between the base of the switching element Q1 and the primary side ground.
B, a series resonance circuit for driving self-excited oscillation, comprising a series connection circuit of a resonance capacitor CB and a base current limiting resistor RB is connected. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Forms a primary parallel resonance circuit of the voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT described later. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage V1 across the resonance capacitor Cr becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit. Is obtained.

The orthogonal control transformer PRT is a saturable reactor on which a resonance current detecting winding ND, a driving winding NB, and a control winding NC are wound. This orthogonal control transformer P
RT drives switching element Q1 and is provided for constant voltage control. This orthogonal control transformer P
As a structure of the RT, although not shown, a three-dimensional core is formed by joining ends of two magnetic legs of two double U-shaped cores having four magnetic legs. Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is constructed by winding in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detecting winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching element Q1 Is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is applied to the drive winding NB. appear. This drive voltage is output from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit (NB, CB).

The insulation converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. On the secondary side of this insulation converter transformer PIT,
A secondary winding N2 as a first secondary winding and a boost winding NHV as a second secondary winding are wound.

Here, the insulation converter transformer PI
FIG. 7 shows the structure of T. Insulation converter transformer PIT
As shown in FIG. 7A, an EE-type core CR in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other is provided.
The primary winding N1 and the secondary winding N2 are wound around the center magnetic leg of the EE type core CR using the divided bobbin LB1. In this case, a litz wire formed by bundling a plurality of single wires is used as the wire material of the primary winding N1 and the secondary winding N2.

A gap G is formed in the center magnetic leg of the EE-type core CR as shown in FIG. As a result, a loose coupling state with a required coupling coefficient can be obtained for the primary winding N1 and the secondary winding N2. The gap G is defined by each E-shaped core CR1,
The CR2 can be formed by making the central magnetic leg shorter than the two outer magnetic legs. In addition, as a coupling coefficient k between the primary winding N1 and the secondary winding N2, for example, a loosely coupled state of k ≒ 0.85 is obtained, so that a saturated state is hardly obtained. I have.

Further, in the insulating converter transformer PIT, a high-voltage winding bobbin HB1 is provided on the divided bobbin LB1 around which the secondary winding N2 is wound. Then, the step-up winding NVH is wound using the high-voltage winding bobbin HB1. In this case, a tightly coupled state is obtained between the boost winding NHV and the secondary winding N2. Further, as the wire material of the boost winding NHV, for example, while the above-described primary winding N1 and secondary winding N2 are litz wires, for example, the wire diameter is 30 μm or more.
A thin line (single line) of about 60 μm is used.

Incidentally, in the above-described isolated converter transformer PIT, as described later, the voltage level of the induced voltage VHV induced in the boost winding NVH is changed to the secondary winding N2.
Is much higher than the voltage level of the induced voltage V2. For this reason, when winding the step-up winding NVH around the high-voltage winding bobbin HB1, for example, as shown in FIGS.
The winding method as shown in FIG.

FIG. 7B shows a so-called split winding (slit winding) in which the boost winding NHV is divided and wound on a high-voltage winding bobbin HB1 divided into a plurality of regions. When winding the step-up winding NVH by split winding,
As shown in the drawing, a plurality of slits S, which are winding areas, are formed by a partition plate DV provided integrally with the inside of the high-voltage winding bobbin HB1. By winding the step-up winding NVH around each slit S, insulation between the step-up windings NHV is obtained. In this case, the step-up winding NHV wound around the high-voltage winding bobbin HB1 is used.
Is molded with an insulating resin such as an epoxy resin EP. Further, FIG. 7 (c) shows that when the boosting winding NVH is wound around the high-voltage winding bobbin HB1, the winding layers formed by winding the boosting winding NHV are:
A form of interlayer winding in which insulation between the step-up windings NHV is obtained by inserting an interlayer film F is shown.

The end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1 as shown in FIG. 1, and the start of the winding is connected in series with the resonance current detecting winding ND. Via the smoothing capacitor C
i (the rectified smoothed voltage Ei). Then, an induced voltage is generated in the secondary winding N2 by the switching current flowing through the primary winding N1. In this case, since the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, the parallel resonance is caused by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2. A circuit is formed. The alternating voltage induced in the secondary winding N2 by this parallel resonance circuit becomes the resonance voltage V2. That is, a voltage resonance operation is obtained on the secondary side of the insulating converter transformer PIT. Things.

As described above, in the high-voltage stabilizing circuit shown in FIG. 1, the primary side is provided with the parallel resonance circuit for performing the voltage resonance type switching operation, and the secondary side is provided with the voltage resonance operation. Are provided. In the present specification, a switching converter having a configuration in which a resonance circuit is provided for the primary side and the secondary side to operate as described above is referred to as a “composite resonance type switching converter”.
I will call it.

In this case, the insulation converter transformer P
First, for the secondary side parallel resonance circuit composed of the IT secondary winding N2 and the secondary side parallel resonance capacitor C2, the secondary winding N
After a center tap is applied to 2 and a full-wave rectifier circuit composed of two rectifier diodes DO1 and DO2 and a smoothing capacitor CO is connected as shown in the figure, a DC output voltage EO of a required level is generated. I am trying to do it. That is, in this case, [Rectifier diodes DO1, D0
A rectifier circuit composed of O2 and a smoothing capacitor CO] rectifies and smoothes the parallel resonance voltage V2 obtained by the secondary side parallel resonance circuit, thereby obtaining a DC output voltage EO. This DC output voltage EO can be supplied as power to a required functional circuit section in a CRT display including the high-voltage stabilization circuit of the present embodiment, for example. Further, in this case, the control circuit 1 is also branched and inputted as its operating power. The control circuit 1 is provided for constant voltage control described later.

Further, an induced voltage induced by the primary winding N1 is generated in the boost winding NHV of the insulating converter transformer PIT, similarly to the secondary winding N2. As described above, since the boost winding NHV is wound in a tightly coupled state with the secondary winding N2, the voltage level of the induced voltage induced in the boost winding NHV is N2, the voltage level of the resonance voltage V2, the secondary winding N2 and the boost winding N
It is determined by the winding ratio with HV. That is, the secondary winding N2
Assuming that the number of turns (number of turns) is N2, the voltage level of the resonance voltage V2 is V2, and the number of turns of the boost winding NVH is NHV, the voltage level VH of the alternating voltage VHV induced in the boost winding NVH
V is represented by VHV = V2 × NHV / N2.
That is, V2 / N2 = VHV / NHV between the resonance voltage V2 and the alternating voltage VHV and between the secondary winding N2 and the boost winding NHV.
Holds. In practice, the number of windings of the boost winding NHV is determined based on this relationship, so that the boost winding NHV is determined.
The alternating voltage excited by the HV is boosted to a required level.

The multiplying voltage rectifier circuit 2 is connected to the boosting winding NHV. The multiplying voltage rectifying circuit 2 receives the boosted alternating voltage VHV generated in the boosting winding NHV. By performing the multiple voltage rectification operation using the alternating voltage VHV in the multiple voltage rectifier circuit 2, a DC high voltage EHV of a predetermined high level (for example, 27 KV) is applied across the smoothing capacitor COHV. can get. This DC high voltage EHV is used as the anode voltage of the CRT.

FIG. 4 shows a Jones & Waters circuit known as a symmetric cascade rectifier circuit as a specific configuration example of the multiple voltage rectifier circuit 2 provided in the circuit shown in FIG. I have. As shown in this figure, the Jones & Waters circuit firstly applies a high voltage capacitor CHV to the winding start end of the boost winding NVH.
A first capacitor series circuit composed of a series connection of A1, CHVA2,... CHVAn is connected, and an end of this capacitor series circuit on the high voltage capacitor CHVAn side is a smoothing capacitor via a high voltage rectifier diode DHVA (n + 1). Connected to the positive terminal of COHV (DC high voltage output terminal). A second high-voltage capacitor CHVB1, CHVB2,... CHVBn is connected in series to the winding end end of the step-up winding NVH.
Capacitor series circuit is connected to the high voltage capacitor CHV
The end of Bn is connected to the positive terminal of the smoothing capacitor COHV via the high voltage rectifier diode DHVB (n + 1).

A high voltage rectifier diode DHVA0, DHVA is connected between the ground and the positive terminal of the smoothing capacitor COHV.
.., DHVB2,..., DHVBn, DHVA (n + 1) are connected in series to a first diode series circuit. As shown in the figure, the circuit and the high-voltage capacitors of the second capacitor series circuit are sequentially connected to each other and connected to the connection points. Similarly, high voltage rectifier diodes DHVB0, DHVB1, DHVA2,... Are connected between the secondary side ground and the positive terminal of the smoothing capacitor COHV.
.., a second diode series circuit consisting of a series connection of DHVAn and DHVB (n + 1) is connected, and each connection point of these high-voltage rectifier diodes is connected to the first capacitor series circuit and the second capacitor. The connection points are sequentially connected to the connection points between the high-voltage capacitors of the series circuit.

In such a connection form, [DHVA1, DHV
B1, CHVA1, CHVB1] ... [DHVAn, DHVBn, CHVA
n, CHVBn] are connected to form an entire rectifier circuit. According to the symmetric cascade rectifier circuit having such a configuration, both ends of the smoothing capacitor COHV are 2 (n + 1) times the induced voltage VHV induced in the boost winding NHV (where n is the rectifier circuit). , Which indicates the number of stages of the partial rectifier circuit, and is an integer equal to or greater than 1), thereby obtaining a high-level DC high voltage EHV.

Here, for example, the number of stages of the symmetric cascade rectifier circuit is as follows.
What is necessary is just to determine appropriately according to the relationship between the level of the DC high voltage EHV actually required and the step-up alternating voltage VHV induced in the step-up winding NVH. As an example, assuming that the step-up alternating voltage VHV induced in the step-up winding NHV is about 3.4 KV and it is necessary to obtain 27 KV as the DC high voltage EHV, the symmetric cascade rectifier circuit shown in FIG. In this case, an eight-fold voltage rectifier circuit having three stages may be used.

In the case of the high voltage stabilizing circuit of the present embodiment,
As shown in FIG. 1, a series connection circuit of voltage dividing resistors R1 and R2 is provided in parallel with respect to a smoothing capacitor COHV from which a high DC voltage EHV is obtained. The voltage dividing resistor R1−
The voltage dividing point of R2 is connected to the control circuit 1. That is, in the present embodiment, a voltage level obtained by dividing the DC high voltage EHV by the voltage dividing resistors R1 and R2 is input to the control circuit 1 as the detection voltage. This means that the control circuit 1 detects the level fluctuation of the DC high voltage EHV.

The control circuit 1 varies the level of the control current (DC current) supplied to the control winding NC in accordance with the change in the DC high voltage EHV, so that the drive winding wound on the orthogonal control transformer PRT is changed. The inductance LB of NB is variably controlled.
Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. this is,
The operation changes the switching frequency of the switching element Q1, and this operation changes the energy transmitted from the primary side to the secondary side in the insulating converter transformer PIT. As a result, control is performed such that the required constant level is maintained for the DC high voltage EHV.

Further, in this embodiment, when the operation of changing the switching frequency is performed as described above, the period TOFF during which the switching element Q1 is off is fixed, and the period TON during which the switching element Q1 is on is reduced. It is variably controlled. That is, in this circuit, as a constant voltage control operation, the switching frequency is variably controlled to perform resonance impedance control on the switching output, and at the same time, the conduction angle control (PWM) of the switching element Q1 in the switching cycle Control). This complex control operation is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”. In this embodiment, a configuration for stabilizing the DC high voltage EHV is employed in this manner.

FIG. 2 shows operation waveforms of main parts of the high-voltage stabilization circuit having the configuration shown in FIG. In addition,
In order to obtain the measurement results shown in FIG. 2, the primary winding N1 = 60T and the secondary winding N2 = 5
T, the step-up winding NHV = 1125T, and the high-voltage capacitors CHVB1 to CHVBn are respectively 1000p
F has been selected. In addition, regarding the multiple voltage rectification circuit 2,
The Jones & Waters circuit shown in FIG. 4 forms an eight-fold voltage rectifier circuit. Further, assuming that the AC input voltage VAC = 100 V, the switching element Q1
The measurement was performed under the operating conditions in which the switching frequency was within the control range of 100 KHz to 167 KHz.

FIGS. 2A to 2D show the waveforms shown in FIG. 2, respectively.
27W (DC high voltage EHV = 27KV, high voltage load current IHV
= 1 mA), the primary side parallel resonance voltage V1, the switching output current (collector current of the switching element Q1) I1, and the secondary side parallel resonance voltage V of the secondary side parallel resonance circuit.
2 and the waveform of the step-up alternating voltage VHV generated in the step-up winding NHV. In each of FIGS. 2 (e) to 2 (h),
High voltage load power PHV = 0W (DC high voltage EHV = 27K
V, high voltage load current IHV = 0 mA).
The waveform of the same part as (d) is shown.

In this case, when the high-voltage load power PHV is 27 W, the switching frequency of the switching element Q1 is controlled so as to be 100 KHz, which is the lower limit of the control range. The primary side parallel resonance voltage V1 at this time is, as shown in FIG.
A pulse on a sine wave is obtained during the period TOFF when the switch is off, which corresponds to the voltage resonance type operation. Further, the switching output current I1 of the switching element Q1 at this time depends on the period T during which the switching element Q1 is turned on.
At ON, the current flows according to the waveform shown in FIG.

Further, as the secondary side parallel resonance voltage V2,
Due to the resonance operation of the secondary parallel resonance circuit, a sinusoidal waveform is obtained as shown in FIG. Also,
As the cycle, the switching frequency fs at this time
= 100 KHz, and the peak level is about 15 V both positively and negatively.

As the boosted alternating voltage VHV obtained in the boosted winding NHV, the boosted winding NHV is tightly coupled to the secondary winding N2.
2 is obtained as a waveform that increases the amplitude. Here, for example, a sine waveform having a peak level of about 3375 V is obtained.

On the other hand, when the high-voltage load power PHV = 0 W, the switching frequency of the switching element Q1 is 167 KHz, which is the upper limit of the control range. The primary parallel resonance voltage V1 at this time is as shown in FIG. In this case as well, the primary side parallel resonance voltage V1 has a waveform in which a pulse on a sine wave is obtained during the period TOFF when the switching element Q1 is off.
As can be seen by comparing with the waveform of the primary side parallel resonance voltage V1 shown in FIG.
OFF is constant at 3 μs, and the period TON during which the switching element Q1 is on changes. That is,
These waveforms show that the above-described composite control is being performed. The switching output current I1 when the high-voltage load power PHV is 0 W is shown in FIG.
A saw-tooth wave as shown in (f) flows in the period TON.

At this time, the secondary side parallel resonance voltage V2
Is the switching frequency fs as shown in FIG.
= Sine wave having a period corresponding to 167 KHz,
The peak level is 12V. The boosted alternating voltage VHV corresponds to the secondary side parallel resonance voltage V2 shown in FIG. 2 (g), and similarly, has a period corresponding to the switching frequency fs = 167 KHz, and is 3375 V
It becomes a sine waveform having a peak level of the order. Supplementally, the DC high voltage EHV at this time is 27 KV ±
50V, and the ripple voltage δV of the DC high voltage EHV is 3
It is suppressed to 2V.

FIG. 3 shows, as characteristics of the high voltage stabilizing circuit shown in FIG. 1, the relationship between the boosted alternating voltage VHV and the secondary DC voltage Eo with respect to the horizontal synchronizing signal frequency fH. In this figure, the case where the horizontal synchronizing signal frequency fH (horizontal resolution) is changed in the range of 30 KHz to 120 KHz is shown. As understood from this figure, in the present embodiment, the boosted alternating voltage VHV is controlled to be constant at 27 KV regardless of the horizontal synchronizing signal frequency fH. The secondary DC voltage Eo is also 15 V when the high-voltage load power PHV = 27 W (DC high voltage EHV = 27 KV, high-voltage load current IHV = 1 mA), and high-voltage load power PHV
= 0W (DC high voltage EHV = 27KV, high voltage load current IHV
= 0 mA), it can be seen that the voltage is controlled to be constant at 12 V.

Here, comparing the high voltage stabilizing circuit shown in FIG. 8 as a conventional example with the circuit shown in FIG. 1 as the present embodiment, the following can be said. The circuit shown in FIG. 8 is configured such that the switching frequency of the switching converter for performing power conversion on the primary side is synchronized with the horizontal synchronization signal frequency fH. Therefore, in order to stabilize the DC high voltage EHV, it is necessary to variably control (PWM control) the ON period within one switching cycle under the condition that the switching frequency is fixed according to the horizontal synchronization signal frequency. Therefore,
On the primary side, it is necessary to configure so that power conversion is performed by two sets of switching converters of the step-down converter 20 and the voltage resonance converter 30.

On the other hand, the switching frequency of the voltage resonance type converter provided on the primary side of the high voltage stabilizing circuit shown in FIG. 1 is not related to the horizontal synchronizing signal frequency fH and is asynchronous. In addition, it is possible to stabilize the DC high voltage EHV by the above-described combined control method. That is, in the high voltage stabilization circuit of the present embodiment, stabilization of the DC high voltage EHV is realized by providing only one set of the voltage resonance type converter on the primary side.

Thus, in the circuit shown in FIG. 1, the power conversion efficiency is greatly improved. For example, as the power conversion efficiency under the condition of high voltage load power PHV = 27 W,
The circuit shown in FIG. 8 is 72%, whereas the circuit shown in FIG. 1 is up to 90%. When the high-voltage load power PHV is 27 W, the AC input power is 37.5 W in the circuit shown in FIG. 8, but is reduced to about 30 W in the circuit shown in FIG. =
At 0 W (no load), the circuit shown in FIG. 8 has 3.5 W, whereas the circuit shown in FIG. 1 has 0.6 W.

Further, in the circuit shown in FIG. 1, since the primary side is provided with one set of the voltage resonance type converter as described above, the number of parts constituting the circuit is reduced accordingly. Therefore, it is possible to promote downsizing of the circuit board size and cost reduction.

For example, in the circuit shown in FIG. 1, in order to suppress the distortion of the white peak image described with reference to FIG. 11, the control range for stabilization is expanded so that the high voltage load current IHV = 3 mA. The circuit can be easily realized if the circuit design is designed to cope with a certain load condition. Therefore, it is not necessary to increase the capacitance of various capacitor elements, and this also makes it possible to reduce the size and cost of the circuit.

Further, in the present invention, as the multiple voltage rectifier circuit 2 provided in the high voltage stabilizing circuit of the present embodiment, a configuration other than the symmetric cascade rectifier circuit shown in FIG. 2 is adopted. No problem. Hereinafter, another configuration example of the multiple voltage rectification circuit 2 will be described with reference to FIGS.

FIG. 5 shows a Cockcroft & Walton circuit known as a basic cascade rectifier circuit as another example of the circuit configuration of the multiple voltage rectifier circuit 2. In this case, the winding end end of the boost winding NVH (secondary ground)
Are the high voltage capacitors CHVA1, CHVA2, CHVA3, ... C
A first capacitor series circuit composed of a series connection of HVAn is connected. The winding start end is connected to a second capacitor series circuit composed of a series connection of high voltage capacitors CHVB1, CHVB2, CHVB3,... CHVBn. The winding end end of the boost winding NVH and the smoothing capacitor C
High-voltage rectifier diodes DHVB1, DHVA1, DHVB2, DHVA2, D
HVB3, DHVA3,..., DHVBn and DHVAn are connected in series with a diode series circuit.

Then, as shown, each of the high-voltage capacitors CHVA1, CHVA forming the first capacitor series circuit
2, CHVA3,..., CHVAn include series circuits [DHVB1-DHVA1], [DHVB2] composed of two sets of high-voltage rectifier diodes among the high-voltage rectifier diodes forming a diode series circuit.
-DHVA2], [DHVB3-DHVA3] ... [DHVBn-DHV
An] are connected in parallel. On the other hand, the high-voltage capacitor CHVB1 forming the second capacitor series circuit is
Start winding end of boost winding NVH and high voltage rectifier diode DHV
The high voltage capacitors CHVB2, CHVB3... CHVBn, which are inserted between the cathodes of B1 and B1, are connected in series with two sets of high voltage rectifier diodes [DHVA1-DHVB2], [DHVA2 −DHVB3], ・
[DHVA (n-1) -DHVBn] are connected in parallel.
In such a connection form, [DHVA1, DHVB1, CHVA1,
CHVB1]... [DHVAn, DHVBn, CHVAn, CHVBn] are connected to an n-stage partial rectifier circuit to form the entire rectifier circuit.

In the rectifying operation of the basic cascade rectifier circuit, first, during a period in which a negative current flows through the boost winding NHV, the high-voltage rectifier diode DHVB1 is turned on, and the voltage from the high-voltage rectifier diode DHVB1 is reduced. The charging operation for the high-voltage capacitor CHVB1 is obtained by the rectified current. Next, during a period in which a positive current flows through the boost winding NHV, the high-voltage rectifier diode D is generated by the induced voltage VHV induced in the boost winding NHV and the voltage across the high-voltage capacitor CHVB1.
HVA1 is turned on, and a charging operation for the high-voltage capacitor CHVA1 is obtained by the rectified current rectified by the high-voltage rectifier diode DHVA1. Next, during a period in which a negative current flows through the boost winding NHV, the induced voltage V
The high voltage rectifier diode DHVB2 is turned on by the voltage across HV and the high voltage capacitor CHVA1, and a charging operation for the high voltage capacitor CHVB2 is obtained.

Thereafter, the high-voltage capacitors CHVA2, CHVA3,..., CHVAn,
And the high voltage capacitors CHVB3... CHVBn forming the second capacitor series circuit are charged. Then, each of the high-voltage capacitors CHVA1 to CHVAn and CHVB1 to CHVBn, which are charged in this manner,
Is charged to the smoothing capacitor COHV by each of the potentials, a DC high voltage EHV corresponding to 2n times the induced voltage VHV is obtained at both ends of the smoothing capacitor COHV.

FIG. 6 shows a Mitchell circuit known as a modified cascade rectification circuit as another example of the circuit configuration of the multiple voltage rectification circuit 2. The connection form of the circuit shown in this figure is substantially the same as that of the basic cascade rectifier circuit shown in FIG.
Is connected to the midpoint of a second capacitor series circuit composed of a series connection of high voltage capacitors CHVB1 to CHVBn. The winding end of the boost winding NHV is connected to the middle point of the first capacitor series circuit composed of the series connection of the high-voltage capacitors CHVA1 to CHVAn and the high-voltage rectifier diode DHV.
B1, DHVA1, DHVB2, DHVA2,..., DHVBn, and DHVAn are respectively connected to the midpoints of diode series circuits formed in series. Again, [DHVA1, DHV
B1, CHVA1, CHVB1] ··· [DHVAn, DHVBn, CHVA
n, CHVBn] are connected to form an entire rectifier circuit. As a result, as a Mitchell circuit configured by such a connection form,
A DC high voltage EHV corresponding to 2n times the induced voltage VHV is obtained across the smoothing capacitor COHV.

As described above, in the high-voltage stabilizing circuit of the present embodiment shown in FIG. 1, the symmetric cascade rectifier circuit shown in FIG. 2 or the basic cascade rectifier circuit shown in FIGS. The rectifier circuit and the modified cascade rectifier circuit can be provided as the multiple voltage rectifier circuit 2. However, in the case where the rectifier circuit is formed by a multi-stage rectifier circuit including a large number of high-voltage capacitors and high-voltage rectifier diodes, the component of the ripple voltage δV is superimposed on the DC high voltage EHV output from the multiple voltage rectifier circuit 2. At the same time, a voltage drop ΔEHV (regulation: voltage fluctuation) occurs. For example, ripple voltage δ superimposed on DC high voltage EHV
From the viewpoint of V, the symmetric cascade rectifier circuit shown in FIG. Also,
From the viewpoint of the voltage drop ΔEHV generated in the multiple voltage rectifier circuit 2, the modified cascade rectifier circuit shown in FIG. In addition,
When the modified cascade rectifier circuit shown in FIG. 6 is a multiple voltage rectifier circuit 2, a high voltage of about 1/2 of the DC high voltage EHV is applied to the boost winding NHV. It is preferable to improve the insulation by molding with an epoxy resin or the like.

In this embodiment, an orthogonal control transformer is used for performing constant voltage control under a configuration having a self-excited resonance converter on the primary side. Alternatively, the oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, for example, as in the case of the orthogonal control transformer, a three-dimensional structure is obtained by combining two sets of double U-shaped cores having four magnetic legs. Form a mold core. The control winding NC and the driving winding NB are applied to the three-dimensional core.
At this time, the winding direction of the control winding and the driving winding is obliquely crossed. Specifically, one of the control winding NC and the driving winding NB is wound around two magnetic legs in a positional relationship adjacent to each other among the four magnetic legs, The other winding is wound around two magnetic legs which are considered to be in a diagonal positional relationship. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained.
Thereby, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

In the circuit of the present embodiment, the switching converter to be provided on the primary side is
Although the case in which the self-excited voltage resonance type converter is provided has been described as an example, a configuration having a separately excited type voltage resonance type converter may be adopted.

Further, the description has been given on the assumption that the high-voltage stabilizing circuit according to the present embodiment is applied to a display device for a personal computer. For example, a CRT including a television receiver may be used.
Can be applied to various types of display devices having a display device.

[0079]

As described above, the present invention provides a high-voltage stabilizing circuit in which the primary side of the isolated converter transformer is connected to the primary side.
A set of voltage-resonant converters is provided, on the secondary side of which a multiplying voltage rectifier circuit is adapted to obtain a high DC voltage, for example as the anode voltage of a CRT. In addition, by providing a parallel resonance circuit on the secondary side, a so-called composite resonance converter is formed as the whole high voltage stabilization circuit. In such a configuration, in order to stabilize the DC high voltage, the switching frequency and the conduction angle of the primary-side voltage resonance type converter are simultaneously controlled according to the level of the DC high voltage, so that the composite control is performed. You. In this case, the switching frequency of the primary-side voltage resonance type converter is free-run with respect to the horizontal synchronization signal frequency, for example.

With such a configuration, only one set of switching converters to be provided on the primary side as a high voltage stabilizing circuit is required. Therefore, the overall power conversion efficiency of the high-voltage stabilization circuit is greatly improved, and the AC input power is also reduced.

In addition, since the number of switching converters provided on the primary side is one, the number of components is greatly reduced, and it is possible to promote downsizing of the circuit board size and cost reduction. Furthermore, since the distortion of the white peak image can be sufficiently suppressed by setting so as to expand the control range of the switching frequency, the capacitance of various capacitors in the circuit is also increased. There is no necessity, and this can also reduce the size and cost of the circuit.

Further, the high voltage stabilizing circuit of the present invention can be constituted by a multiple voltage rectifying circuit employing a Jones & Waters circuit, a Cockcroft & Walton circuit or a Mitchell circuit as a DC high voltage generating means. Is done. Thus, even if the induced voltage induced in the second secondary winding of the insulating converter transformer is not generated at a sufficiently high level, the DC high voltage generating means can reduce the anode voltage of the CRT, for example. It is possible to obtain a DC high voltage of a level sufficient for practical use in accordance with a required purpose such as obtaining. In addition, if the Jones & Waters circuit is applied to the DC high voltage generating means, the ripple voltage superimposed on the DC high voltage can be reduced, and if the Mitchell circuit is applied, the DC high voltage generating means can be reduced. Since it is possible to effectively suppress the voltage drop, it is also possible to realize an optimum configuration for each device to which the high-voltage stabilization circuit of the present invention is applied.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a high-voltage stabilization circuit as an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part in the high voltage stabilization circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing a DC high voltage level and a secondary DC output voltage level with respect to a horizontal synchronization signal frequency in the high voltage stabilization circuit of the present embodiment.

FIG. 4 is a circuit diagram showing a configuration of a multiple voltage rectification circuit provided in the high voltage stabilization circuit of the present embodiment.

FIG. 5 is a circuit diagram showing another configuration of the multiple voltage rectifier circuit.

FIG. 6 is a circuit diagram showing another configuration of the multiple voltage rectifier circuit.

FIG. 7 is a cross-sectional view illustrating a structural example of the insulating converter transformer according to the present embodiment.

FIG. 8 is a circuit diagram showing a configuration example of a conventional high-voltage stabilization circuit.

9 is a waveform chart showing an operation of a main part of the high voltage stabilization circuit shown in FIG.

10 is an explanatory diagram showing a DC high voltage level and a secondary DC output voltage level with respect to a horizontal synchronization signal frequency in the high voltage stabilization circuit shown in FIG. 8;

FIG. 11 is an explanatory diagram showing how a white peak image displayed on a CRT is distorted.

[Explanation of symbols]

1 control circuit, 2 multiple voltage rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, PRT orthogonal control transformer, Cr primary side parallel resonance capacitor, C2 secondary side parallel resonance capacitor, N1 primary winding, N2 secondary winding, NHV boost winding,
NC control winding, NB drive winding, ND resonance current detection winding, CB resonance capacitor, DO1, DO2 rectifier diode, DHVA0 to DHVA (n + 1) DHVB0 to DHVB (n + 1) High voltage rectifier diode, CHVA1 to CHVAAn CHVB1 to CHVBn High voltage capacitors, CO, COHV smoothing capacitors

Claims (4)

[Claims] 1. A rectifying / smoothing means for rectifying and smoothing a commercial AC power supply to obtain a DC input voltage; a switching means comprising a switching element for intermittently outputting the DC input voltage; And a primary winding, a first secondary winding, and a second secondary winding, and the primary winding and the first secondary winding are The required degree of loose coupling is obtained,
The first and second secondary windings are insulated converter transformers configured to obtain a tightly coupled state, and the switching means is formed to operate in a voltage resonance type. A primary-side parallel resonance circuit, a secondary-side parallel resonance circuit formed with the first secondary winding, and rectification by inputting an alternating voltage obtained in the first secondary winding. A DC output voltage generating means for obtaining a secondary-side DC output voltage by performing an operation; and a rectifying operation by inputting an alternating voltage obtained to the second secondary winding and performing a rectification operation. DC high voltage generating means configured to obtain a DC high voltage that is a high voltage level of the DC high voltage, and by changing a switching frequency of the switching element according to the level of the DC high voltage, a constant with respect to the DC high voltage. Voltage control High voltage stabilization circuit, characterized in that it comprises a voltage control means. 2. The high voltage stabilizing circuit according to claim 1, wherein said DC high voltage generating means includes a Jones and Waters circuit. 3. The high-voltage stabilizing circuit according to claim 1, wherein said DC high-voltage generating means includes a Cockcroft and Walton circuit. 4. The high voltage stabilizing circuit according to claim 1, wherein said DC high voltage generating means is provided with a Mitchell circuit.