JP2002112545A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To make a DC high-voltage outputted from a secondary side constant. SOLUTION: After a composite resonance type switching power supply circuit is constituted to stabilize a DC output voltage E01 outputted from the secondary side of an insulating converter transformer PIT, an active clamping circuit 3 is connected in parallel with the primary winding N of the insulating converter transformer PIT to stabilize a DC high-voltage EHV outputted from a high- voltage generating circuit 4. With such a constitution, a power loss in a switching power supply circuit, which is produce by the fluctuation of the DC high voltage EHV, can be reduced in comparison with a conventional constitution.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit suitable for use in, for example, a large-sized color television receiver having a high resolution, a projector device, and the like.

[0002]

2. Description of the Related Art Cathode ray tubes (CRTs)
For example, among the video equipment provided with a high-definition television receiver, a high-definition television receiver, a high-definition television broadcast (HDTV;
Definition Television (hereinafter referred to as “HD television”), projector devices, and the like, have a horizontal synchronization frequency twice as high as that of a normal television receiver (31) in order to realize high resolution. .5
KHz). Further, for example, a cathode ray tube (hereinafter, “C
RT ”) is 30K.
V or more, and the beam current is 2 mA or more.

Meanwhile, in a television receiver equipped with a CRT, a high voltage to be supplied to the anode electrode of the CRT is usually generated using a flyback pulse voltage generated during a horizontal flyback period. However, in a high-resolution television receiver as described above, the pulse width of the flyback pulse voltage is narrower than the pulse width of the flyback pulse generated in a normal television receiver. Therefore, when a high voltage to be supplied to the anode electrode is generated by using the flyback pulse voltage, for example, when the beam current of the CRT becomes zero, C
There is a drawback that the high pressure supplied to the RT anode fluctuates.

Therefore, in the above-mentioned television receiver, a high-voltage regulator circuit is inserted before the flyback transformer, and the input voltage level input to the primary side of the flyback transformer is output from the secondary side. There is known a device in which a high voltage level is made constant by performing variable control according to a high voltage level to be applied.

FIG. 7 is a diagram showing a configuration of a conventional power supply circuit provided in a high-resolution television receiver or the like as described above. In FIG. 7, the switching power supply 10 switches the input DC voltage,
This is a DC-DC converter that converts the output to a predetermined DC voltage level and outputs the converted DC-DC level.

In the preceding stage of the switching power supply 10, a bridge rectifier circuit Di of full-wave rectification and a smoothing capacitor Ci are provided.
A rectifying / smoothing circuit is provided. The rectifying / smoothing circuit rectifies and smoothes a commercial AC power supply (AC input voltage VAC) to obtain a DC voltage Ei. And
The DC voltage Ei is input to the switching power supply 10, and the switching power supply 10 outputs at least a DC output voltage EO1 at a predetermined voltage level. The DC output voltage EO1 is a drive voltage for driving the horizontal deflection circuit of the television receiver.

On the secondary side of the switching power supply 10, a smoothing capacitor CO1 and a high-voltage regulator circuit 20 indicated by a broken line are provided. The smoothing capacitor CO1 is
The output of the switching power supply 10 is smoothed, and a DC output voltage EO1 is obtained from both ends of the smoothing capacitor CO1. The high voltage regulator circuit 20 includes a PWM (Pu
The step-down chopper circuit is a step-down type chopper circuit of an lse Width Modulation (control) type and includes, for example, a switching element Q11 composed of a MOS-FET, a flywheel diode D11, a choke coil CH, and a smoothing capacitor CO11. In this case, the gate of the switching element Q11 is connected to the drive circuit 14 of the switching drive unit 30, which will be described later, and its drain is connected to the secondary output terminal (DC output voltage EO1) of the switching power supply 10. The source is a high voltage generating circuit 4 via a choke coil CH.
0 is connected to the winding start end of the primary winding N11 of the flyback transformer FBT provided in the zero. The flywheel diode D11 is connected between the source of the switching element Q11 and the secondary side ground in the illustrated direction, and the smoothing capacitor CO11 is connected to the connection line between the choke coil CH and the primary winding N11 of the flyback transformer FBT. Is done. High voltage regulator circuit 2 having such a configuration
At 0, the input DC output voltage EO1 is stepped down to a predetermined level DC output voltage EO11 and output in accordance with the switching operation of the switching element Q11.

[0008] The flyback transformer FBT boosts the input on the primary side and transmits it to the secondary side. The primary winding N11 provided on the primary side of the flyback transformer FBT is
The winding start end is connected to the positive electrode of the smoothing capacitor CO11 of the high-voltage regulator circuit 20, and the winding end is connected to the collector of the switching element Q12. The switching element Q12 is, for example, an IGBT (Insulated Gate B
The base is connected to the drive circuit 15 of the switching drive unit 30. A damper diode D12 and a parallel resonance capacitor Cr11 are connected in parallel between the collector and the emitter.

That is, in the power supply circuit shown in FIG. 7, a high voltage regulator circuit 20 is connected to the output line of the DC output voltage EO1 of the switching power supply 10, and a flyback transformer F
A voltage resonance type converter comprising a BT primary winding N11, a switching element Q12, a damper diode D12 and a parallel resonance capacitor Cr11 is provided. A high voltage generation circuit 40 including the flyback transformer FBT is provided on the secondary side of the flyback transformer FBT.

[0010] Switching drive unit 3 shown by a broken line
0 of the horizontal synchronization signal fH (3
1.5KHz). Oscillation circuit 1
2 performs an oscillation operation based on a synchronization signal output from the synchronization circuit 11 and outputs an oscillation signal generated by the oscillation operation to the PWM circuit 13 and the drive circuit 15. PW
The M circuit 13 outputs a pulse width modulated signal to the drive circuit 14 based on an oscillation signal from the oscillation circuit 12 and a control signal from the control circuit 1 described later. Drive circuit 14 converts the modulated signal from PWM circuit 13 into a drive voltage and outputs the drive voltage to the gate of switching element Q11. The drive circuit 15 receives an oscillation signal from the oscillation circuit 12, and outputs a drive current obtained by converting the oscillation signal to the switching element Q12. As a result, the switching element Q11 performs a switching operation based on the modulation signal generated by the PWM circuit 13, and the switching element Q12 performs a switching operation at a fixed switching frequency based on the synchronization signal fH.

On the secondary side of the flyback transformer FBT, five sets of boost windings NVH11, NVH12, NVH13, NHV14,
NHV15 is divided and wound. In this case, the polarity (winding direction) of each of the boost windings NHV11 to NHV15 with respect to the primary winding N11 is reversed. Each boost winding NVH11 ~ NHV
At the end of 15 windings, each high voltage rectifier diode DHV11, DHV
The anodes of 12, DHV13, DHV14, and DHV15 are connected. And the cathode of the high voltage rectifier diode DHV11 is
The cathodes of the high-voltage rectifier diodes DHV12 to DHV15 are connected to the positive terminal of the smoothing capacitor COHV, and the cathodes of the high-voltage rectifier diodes DHV12 to DHV15 are connected to the winding start ends of the step-up windings NVH11 to NHV14, respectively. That is, on the secondary side of the flyback transformer FBT shown in this figure, a so-called multisingle-type half-wave rectification circuit in which five sets of half-wave rectification circuits are connected in series is formed.

Therefore, on the secondary side of the flyback transformer FBT, these five sets of half-wave rectifiers are connected to the boost winding NVH11.
To rectify the current induced in NHV15 to smoothing capacitor C
The operation of charging the OHV is performed, and a DC voltage having a level corresponding to five times the induced voltage induced in each of the boost windings NVH11 to NHV15 is obtained at both ends of the smoothing capacitor COHV. And this smoothing capacitor COH
The DC high voltage EHV obtained at both ends of V is output as an anode voltage to, for example, an anode electrode of a CRT. In addition,
In this case, an induced voltage boosted to 6 KV is generated in each of the boost windings NHV11 to NHV15, and the DC high voltage EHV is 30V.
It is assumed that a high voltage of KV can be obtained.

A smoothing capacitor COFV is inserted between the cathode of the high-voltage rectifier diode DHV14 and the secondary-side ground. A DC output voltage EFV obtained at both ends of the smoothing capacitor COFV is used as a focus voltage.
For example, it is output to the fourth grid (focus electrode) of T.

Further, a series circuit in which resistors R1 and R2 are connected in series is provided between the cathode of the high-voltage rectifier diode DHV11 and the winding start end (secondary ground) of the boost winding NHV15. The voltage divided by the series circuit is input to the control circuit 1 as a detection voltage of the DC high voltage EHV.

The control circuit 1 uses the control current or control voltage corresponding to the level change of the DC high voltage EHV output from the secondary side of the high voltage generation circuit 40 as a control signal as a control signal.
3 is output. As described above, the switching drive unit 30 controls the DC high voltage E output from the high voltage generation circuit 40.
A drive voltage whose duty ratio is changed in one switching cycle according to the voltage level of HV is supplied to the switching element Q11 of the high-voltage regulator circuit 20. As a result, the on / off period in one switching cycle of the switching element Q11 is controlled, and the high voltage regulator circuit 20 outputs the DC output voltage EO11 according to the voltage level of the DC high voltage EHV. Then, the DC output voltage EO11 is used as an input voltage of a voltage resonance type converter formed on the primary side of the flyback transformer FBT.
To stabilize the DC high voltage EHV output from the secondary side.

Here, for example, 30
When the power supply circuit shown in FIG. 7 is actually configured so that a DC high voltage EHV of KV can be obtained, the primary winding N11 of the flyback transformer FBT = 70T (turn), and the boost windings NHV11 to NHV15 = 462T, parallel resonance capacitor Cr
11 = 0.01μF, smoothing capacitor COHV = 2000P
F, a smoothing capacitor COFV = 220 PF is selected.

FIG. 8 shows an example of operation waveforms of the power supply circuit shown in FIG. FIGS. 8A to 8D show that, for example, the AC input voltage VAC is 100 V and the high-voltage load of the high-voltage generation circuit 40 is the maximum load power Pomax = 60 W (IHV = 2 m
Operation waveforms under the condition A) are shown in FIGS.
FIG. 8H shows that the AC input voltage VAC is 100 V,
The high voltage load of the high voltage generation circuit 40 is the minimum load power Pomin
The operation waveform under the condition of = 0 W (IHV = 0 mA) is shown.

When the high-voltage load of the high-voltage generation circuit 40 is set to the maximum load power, the ON / OFF period TON1 / TOFF1 of the switching element Q11 is controlled to 28.5 μs / 3.3 μs, and the drain-source of the switching element Q11 is controlled. Voltage V
FIG. 11 is shown as in FIG. The drain current I11 flowing through the switching element Q11 is shown as in FIG. At this time, the DC output voltage EO11 generated in the smoothing capacitor CO11 of the high-voltage regulator circuit 20 is
As shown in FIG. 8C, for example, when the voltage level is 1
21V.

On the other hand, the ON / OFF period TON2 / TOFF2 of the switching element Q12 is controlled to 26.25 μs / 5.5 μs, and the switching element Q12 has a collector current I12 having a waveform as shown in FIG. Flows. Then, during the period TOFF2 in which the switching element Q12 is turned off, both ends of the parallel resonance capacitor Cr11 are connected as shown in FIG.
The flyback pulse voltage (resonance pulse voltage) shown in FIG.

On the other hand, when the high-voltage load of the high-voltage generation circuit 40 is set to the minimum load power (no load), the ON / OFF period TON1 / TOFF1 of the switching element Q11 is set to 26 μs.
/5.8 μs, and the drain-source voltage V11 of the switching element Q11 becomes as shown in FIG. The drain current I11 flowing through the switching element Q11 is as shown in FIG. At this time, the DC output voltage EO11 generated in the smoothing capacitor CO11 of the high-voltage regulator circuit 20 has, for example, a voltage level of 111 V as shown in FIG.
DC output voltage EO11 at maximum load power shown in (c)
(121 V). That is, in the high-voltage regulator circuit 2, the on / off period TON1 / TOFF1 of the switching element Q11 is changed by the high-voltage load of the high-voltage generation circuit 40, and the voltage level of the DC output voltage EO11 that is output accordingly is changed. It is supposed to change.

On the other hand, the ON / OFF period TON2 / TOFF2 of the switching element Q12 is kept at 26.25 μs / 5.5 μs, and the flyback pulse voltage V12 generated at both ends of the parallel resonance capacitor Cr11 is shown in FIG. ), And a voltage substantially equivalent to the flyback transformer voltage waveform at the time of the maximum load power (FIG. 8C) is obtained. The collector current I12 flowing through the switching element Q12 is as shown in FIG.

FIG. 9A shows the DC high voltage EHV and the fluctuation characteristics of the DC output voltages EO1 and EO11 with respect to the change of the beam current IHV output from the high voltage generating circuit 40 of the power supply circuit shown in FIG. Show. As can be seen from FIG. 9A, the beam current IHV output from the high voltage generation circuit 40 is output.
Fluctuates in the range of 0 mA to 2 mA, only the voltage level of the DC output voltage EO11, which is the output of the high-voltage regulator circuit 20, fluctuates in accordance with the change in the beam current IHV, and the voltage level of the DC output voltage EO1 and the DC voltage The voltage level of voltage EHV is maintained at a substantially constant level.

[0023]

In the power supply circuit shown in FIG. 7, the switching power supply 10, the high-voltage regulator circuit 20, and the high-voltage generation circuit 40
Since each of them performs power conversion, there is a disadvantage that overall power conversion efficiency is reduced. For example, the AC / DC conversion efficiency (ηAC-DC) of the switching power supply 10 is about 90
%, The voltage conversion efficiency of the high-voltage regulator circuit 20 (ηDC−
DC) is about 95%, and their power conversion efficiency is about 8%.
5.5%. In the high-voltage generation circuit 40, the power loss in the resistor R1 for obtaining the detection voltage corresponding to the DC high voltage EHV, the power loss in the focus circuit for obtaining the focus voltage EFV, the high-voltage rectifier diode DHV
Power loss in 11 to DHV15 and switching element Q
The power conversion efficiency characteristic is shown in FIG. 9B due to the switching loss and the like of FIG. 12, for example, when the high effective load power PHV of the high voltage generation circuit 40 is 60 W (EHV = 30 K
V, IHV = 2mA), power conversion efficiency (ηDC-DC)
Is about 82%. Accordingly, the total power conversion efficiency η of the power supply circuit shown in FIG. 7 is about 70.1%, and for example, about 85.6 to obtain a high-voltage load power PHV of 60 W.
W input power Pin is required. That is, about 25.6 W of power loss occurs in the power supply circuit shown in FIG.

In the power supply circuit shown in FIG. 7, it is necessary to provide the high-voltage regulator circuit 20, which increases the number of parts and complicates the circuit configuration.
The area for mounting components increases. In addition, since two sets of relatively expensive switching elements and diode elements are required, there is a disadvantage that the cost of component materials is significantly increased.

Further, since the switching operation waveform of the high-voltage regulator circuit 20 is a rectangular waveform as shown in FIGS. 8A and 8E, switching noise occurs in the switching element Q11. Therefore, when the power supply circuit shown in FIG. 7 is provided in an actual television receiver, it is necessary to take some measures to suppress noise.

[0026]

Therefore, a switching power supply circuit according to the present invention is configured as follows in consideration of the above-mentioned problems. That is, a switching means formed with a main switching element for intermittently outputting the input DC input voltage and a primary-side parallel resonance circuit that makes the operation of the switching means a voltage resonance type are formed. A primary-side parallel resonance capacitor provided for transmitting the output of the primary side to the secondary side, a primary-side winding is wound on the primary side, and a secondary-side winding is wound on the secondary side. An insulation converter transformer that is wound so that a required degree of coupling can be obtained in which the primary winding and the secondary winding are loosely coupled. A secondary parallel resonance circuit formed by connecting the side parallel resonance capacitors in parallel, and a voltage in the positive period of the alternating voltage obtained from the secondary winding formed including the secondary side parallel resonance circuit About half-wave rectification A DC output voltage generating means configured to obtain a DC output voltage, and a switching frequency of the main switching element is variably controlled according to the DC output voltage level, and an OFF period in the switching cycle is kept constant. A first constant voltage control means for performing constant voltage control by switching the main switching element so as to vary the ON period. By transmitting the voltage input to the primary side to the secondary side, a step-up transformer formed so as to obtain a boosted voltage boosted to a predetermined level from the secondary side, and the primary-side operation of the step-up transformer is resonated. In order to operate, at least a series resonance circuit formed by connecting a series resonance capacitor in series with the primary winding of the step-up transformer, and supplied by branching the switching output of the switching means, and a step-up transformer A rectification operation is performed on the boosted voltage obtained on the next side, thereby comprising a DC high voltage generating means configured to obtain a DC high voltage, and at least a series connection circuit of a clamp capacitor and an auxiliary switching element, and a series connection circuit Is provided with active clamp means connected in parallel to the primary winding of the insulated converter transformer. By performing the conduction angle control of the switching elements, and the like and a second constant voltage control means is configured to perform constant voltage control.

According to the above configuration, a voltage resonance type converter is formed on the primary side of the insulating converter transformer, and a DC output voltage generating means for obtaining a DC output voltage and a DC high voltage are provided on the secondary side. A DC high voltage generating means for obtaining the DC voltage is formed. Further, by providing an active clamp means on the primary side of the insulating converter transformer, the DC output voltage and the DC high voltage are stabilized. In this case, in order to finally generate a DC high voltage from the DC input voltage, it is only necessary to provide a two-stage power conversion section,
It is possible to reduce power loss due to power conversion.

[0028]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure has a configuration as a complex resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in FIG. 1, first, a full-wave rectifier comprising a bridge rectifier circuit Di and a smoothing capacitor Ci as a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage. A smoothing circuit is provided to generate a rectified smoothed voltage (DC input voltage) Ei corresponding to a level that is one time higher than the AC input voltage VAC.

The switching converter which inputs and outputs the DC input voltage Ei is provided with a single main switching element Q1 and a so-called single-ended type self-excited voltage resonance type converter which performs a switching operation. . In this case, a high withstand voltage bipolar transistor (BJ) is connected to the main switching element Q1.
T: junction type transistor).

The base of the main switching element Q1 is
It is connected to the positive electrode of the smoothing capacitor Ci via the current limiting resistor RB and the starting resistor RS, and its emitter is grounded to the primary side ground. Further, a series resonance circuit for self-excited oscillation driving composed of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB is connected between the base of the main switching element Q1 and the primary side ground. Also,
The clamp diode DD1 inserted between the base of the main switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the main switching element Q1 is turned off. The collector of the main switching element Q1 is connected to one end of a primary winding N1 formed on the primary side of the insulating converter transformer PIT, and its emitter is grounded.

A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the main switching element Q1. The primary-side parallel resonance capacitor Cr has its own capacitance and the primary-side winding N.
The primary side parallel resonance circuit of the voltage resonance type converter is formed by the leakage inductance L1 on the one side. And
Although detailed description is omitted here, when the main switching element Q1 is turned off, the voltage V1 generated across the primary-side parallel resonance capacitor Cr by the operation of the primary-side parallel resonance circuit is actually a sinusoidal pulse waveform. Thus, a voltage resonance type operation is obtained.

The orthogonal control transformer PRT is a saturable reactor on which a resonance current detecting winding ND, a driving winding NB, and a control winding NC are wound. This orthogonal control transformer P
The RT drives the main switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, although not shown, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. . Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is configured to be wound in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detecting winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1, thereby providing the switching output of the main switching element Q1. Is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the resonance current detection winding N
The switching output obtained at D is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is generated in the drive winding NB. This drive voltage is applied to a series resonance circuit (N
B, CB) is output as a drive current to the base of the main switching element Q1 via the base current limiting resistor RB. Thereby, the main switching element Q1
Performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

Insulation converter transformer (Power Isolation
Transformer) PIT transmits the switching output of main switching element Q1 to the secondary side. As shown in FIG. 5, the insulating converter transformer PIT has an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. The primary winding N1 and the secondary winding N2 are wound around the center magnetic leg in a divided state using the divided bobbin B, respectively. A gap G is formed for the center magnetic leg as shown in the figure. Thereby, loose coupling with a required coupling coefficient is obtained. The gap G can be formed by making the central magnetic legs of the E-shaped cores CR1 and CR2 shorter than the two outer magnetic legs. The coupling coefficient k is, for example, k ≒ 0.
A loose coupling state of 85 is obtained, so that a saturated state is hardly obtained.

By the way, the insulation converter transformer PIT
Of the secondary winding includes a primary winding N1 and a secondary winding N2.
Of the primary winding N1 and the secondary winding N1 by the polarity (winding direction) of the rectifier diode D0 and the polarity of the alternating voltage excited in the secondary winding.
2 with respect to the mutual inductance M with the inductance L2, the operation mode of + M (polarization mode; forward operation) and the operation mode of -M (depolarization mode;
Flyback operation). For example, FIG.
Mutual inductance is + M when equivalent to the circuit shown in FIG. 6A, and −M when equivalent to the circuit shown in FIG. 6B. Note that, in the power supply circuit shown in FIG.
During the period when the polarity of the primary winding N1 and the secondary windings N2 and N3 are in the + M operation mode, the rectifier diodes DO1,
Charging operation of the smoothing capacitors CO1 and CO3 is performed via DO3.

As shown in FIG. 1, the winding start end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the main switching element Q1, and the winding end thereof is connected in series with the resonance current detection winding ND. Connected to the positive electrode of the smoothing capacitor Ci. Also, on the secondary side,
As a secondary winding, a secondary winding N2 and a tertiary winding N3 formed by winding up a winding end portion of the secondary winding N2.
Is provided. The secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2.

In this case, the winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end is connected to the rectifier diode D.
Connected to the anode of O1. Then, a half-wave rectifying / smoothing circuit including the rectifying diode DO1 and the smoothing capacitor CO1 obtains a DC output voltage EO1 (for example, 135V) for horizontal deflection whose voltage level is 110V to 140V.

Further, on the secondary side of the insulated converter transformer PIT, the ending of the tertiary winding N3 wound around the secondary winding N2 is connected to the anode of the rectifying diode DO3 so that the rectifying diode DO3 is connected to the rectifying diode DO3. For example, a DC output voltage EO3 (200 V) for a video output circuit is obtained by a half-wave rectifier circuit including a smoothing capacitor C03. In the power supply circuit shown in FIG.
By connecting the negative side of O3 to the positive side of the smoothing capacitor CO1, a DC output voltage EO3 for the video output circuit is obtained from both ends of the series connection circuit of the smoothing capacitors CO1 to CO3. That is, in the power supply circuit shown in FIG. 1, in order to obtain the DC output voltage EO3, the DC output voltage EO1 generated across the smoothing capacitor CO1 is stacked with the DC output voltage generated across the smoothing capacitor CO3. Winding N2
And the DC output voltage obtained from the tertiary winding N3 are superimposed to obtain the DC output voltage EO3. Therefore, the rectifying and smoothing circuit including the tertiary winding N3, the rectifying diode DO3, and the smoothing capacitor CO3 provides a DC output voltage of 90V to 60V obtained by subtracting the DC output voltage E01 (110V to 140V) from the DC output voltage EO3 (200V). Is required to be obtained.
Although not shown, the DC output voltage (± 15 V) for the vertical deflection circuit and the DC output voltage (15
V), a DC output voltage (6.3 V) for the heater, and the like can be obtained from the secondary side of the insulated converter transformer PIT.

A secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2. In this case, a secondary parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2, and an alternating voltage induced on the secondary side of the insulating converter transformer PIT. Becomes a resonance voltage, and a voltage resonance operation is obtained on the secondary side of the insulating converter transformer PIT.

That is, in the power supply circuit shown in FIG. 1, the primary side of the insulated converter transformer PIT is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a voltage resonance operation for obtaining the voltage resonance operation. Are provided. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”. It should be noted that such a configuration as a composite resonance type switching converter has an insulation converter transformer P as described with reference to FIG.
This is realized by forming a gap G with respect to IT and performing loose coupling by a required coupling coefficient, thereby obtaining a state that is more difficult to be saturated. For example, when the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to hope that this is done properly.

The DC output voltage EO1 is also branched and input to the control circuit 1. The control circuit 1 is constituted by, for example, an error amplifier or the like, and flows to the control winding NC of the orthogonal control transformer PRT according to a change in the DC output voltage level EO1 output from the secondary side of the insulating converter transformer PIT. By varying the control current (DC current) level, the inductance LB of the drive winding NB wound around the orthogonal control transformer PRT is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the main switching element Q1 formed including the inductance LB of the drive winding NB changes, and the switching frequency of the main switching element Q1 is varied. Operation. This operation allows the isolated converter transformer PIT
Of the DC output voltage output from the secondary side of the power supply is stabilized. Note that the DC output voltage EO3 may be branched and input to the control circuit 1 to make the DC output voltage constant.

When the orthogonal control transformer PRT for variably controlling the inductance LB of the drive winding NB is provided as in the power supply circuit shown in FIG. 1, the main switching element Q1 is turned off to change the switching frequency. After the constant period TOFF, the ON period TON is variably controlled. In other words, in the power supply circuit shown in FIG. 1, as a constant voltage control operation, the switching frequency is variably controlled to perform resonance impedance control on the switching output. PWM control). This complex control operation is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Further, in the power supply circuit of the present embodiment shown in FIG. 1, an active clamp circuit 3 is provided in parallel with the primary winding N1 of the isolated converter transformer PIT, and the main switching element Q1 At both ends, a series resonance capacitor C3 and a primary winding N4 of a step-up transformer HVT provided in a high-voltage generation circuit 4 described later.
Are connected in parallel. For this reason, the resonance voltage V1 generated at both ends of the primary side parallel resonance capacitor Cr is input to the primary side of the step-up transformer HVT via the series resonance circuit. By operation, the step-up transformer HV
Both the current I3 flowing through the primary winding N4 of T and the voltage V4 across the primary winding N4 have a substantially sinusoidal resonance waveform.

The active clamp circuit 3 includes an auxiliary switching element Q2, a clamp capacitor CCL, and a clamp diode DD2. As the clamp diode DD2, a so-called body diode incorporated as a component in the switching element Q2 which is a MOS-FET is used. Further, as a drive circuit system for driving the auxiliary switching element Q2, a drive winding N
g, a capacitor Cg, and a resistor Rg.

In this case, the clamp diode DD2 is provided between the drain and the source of the auxiliary switching element Q2.
Are connected in parallel. Here, the clamp diode D
The anode of D2 is connected to the source and the cathode is connected to the drain. The drain of the auxiliary switching element Q2 is connected to a connection point between the line of the rectified and smoothed voltage Ei and the start end of the primary winding N1 via the clamp capacitor CCL. The source of the auxiliary switching element Q2 is connected to the winding terminal of the primary winding N1. That is, the active clamp circuit 3 of the present embodiment is configured by connecting a clamp capacitor CCL in series to the parallel connection circuit of the auxiliary switching element Q2 // clamp diode DD2. The circuit thus formed is connected in parallel with the primary winding N1 of the insulating converter transformer PIT.

As a driving circuit system for the auxiliary switching element Q2, a series connection circuit of a resistor Rg, a capacitor Cg, and a driving winding Ng is connected to the gate of the auxiliary switching element Q2, as shown in the figure. This series connection circuit forms a self-excited oscillation drive circuit for the auxiliary switching element Q2. Here, the drive winding Ng is formed so as to wind up the winding end portion side of the primary winding N1 in the insulating converter transformer PIT, and the number of turns in this case is, for example, 1T (turn). As a result, a voltage excited by the alternating voltage obtained in the primary winding N1 is generated in the driving winding Ng. In practice, if the number of turns of the drive winding Ng is 1T, the operation is guaranteed, but the present invention is not limited to this. Further, the gate of the auxiliary switching element Q2 is connected to a photo coupler PC.
The control circuit 2 is also connected via (PC1, PC2) to a control circuit 2 described later, and a control voltage corresponding to a level change of the DC high voltage EHV output from the high-voltage generation circuit 4 described later is input from the control circuit 2 to the control circuit 2. .

High-voltage generating circuit 4 enclosed by a dashed line
Is composed of a step-up transformer HVT and a high-voltage rectifier circuit, and boosts a resonance voltage V4 input to the primary winding N4 of the step-up transformer HVT to generate a high voltage corresponding to, for example, an anode voltage level of a CRT. . For this reason,
On the secondary side of the step-up transformer HVT, four to five sets of step-up windings NHV are wound separately by slit winding or interlayer winding. In this case, the primary winding N4 and the boost winding NHV are wound so as to be tightly coupled, and furthermore, are wound so that their polarities (winding directions) are opposite. Therefore, on the secondary side of the step-up transformer HVT, the primary winding N4
, The negative resonance voltage of the resonance voltage V4 is inverted, and a boosted voltage boosted by the turn ratio (NHV / N4) between the boost winding NHV and the primary winding N4 is obtained.

Here, FIG. 3 is a cross-sectional view of the step-up transformer HVT, and the structure of the step-up transformer HVT will be described with reference to FIG. The step-up transformer HVT shown in FIG.
For example, a square core CR30 is formed by combining the magnetic legs of two U-shaped cores CR1 and CR2 so as to face each other. A gap G is provided at a portion where the end of the U-shaped core CR1 and the end of the U-shaped core CR2 face each other. Further, as shown,
By attaching the low-voltage winding bobbin LB and the high-voltage winding bobbin HB to one of the magnetic legs of the rectangular core CR30, the primary winding N4 is connected to each of the low-voltage winding bobbin LB and the high-voltage winding bobbin HB. And the boost winding NHV is divided and wound. In this case, the primary winding N4 is wound around the low-voltage winding bobbin LB, and the high-voltage winding bobbin H
A plurality of step-up windings NHV are wound around B by interlayer winding in which the interlayer film F is inserted and wound.

In the power supply circuit shown in FIG. 1, the resonance voltage V4 input to the primary winding N4 of the step-up transformer HVT is used.
From the secondary side of the isolated converter transformer PIT, the frequency thereof corresponds to the switching frequency of the main switching element Q1, and the switching frequency is, for example, several tens kHz to 2 kHz.
The frequency is in the range of about 00 kHz. In this case, an eddy current may be generated in the primary winding N4 of the step-up transformer HVT. Therefore, in the present embodiment, an eddy current is generated in the primary winding N4 using a litz wire for the primary winding N4. I try to prevent it.

In the present embodiment, the case where the boost winding NHV of the boost transformer HVT is wound by interlayer winding is shown, but the winding of the boost winding NHV is limited to the interlayer winding. For example, although not shown, for example, the high-pressure bobbin HB may be divided into a plurality of regions, and the boost winding NHV may be wound around each of the divided regions, that is, winding may be performed by so-called divided winding. In other words, the structure of the step-up transformer HVT may be such that a plurality of step-up windings NHV wound around the high-voltage bobbin HB are wound in a state of being insulated from each other.

In the power supply circuit shown in FIG.
On the secondary side of the VT, five sets of boost windings NVH1, NVH2, NVH
3, NHV4 and NHV5 are wound independently of each other, and high-voltage rectifier diodes DHV1, DHV2, DHV3, DHV4,
The anode side of DHV5 is connected. Then, the cathode of the high voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV, and the remaining high voltage rectifier diodes DHV2 to
The respective cathodes of DHV5 are respectively boosted windings NVH1 to NHV4
Is connected to the winding start end.

That is, on the secondary side of the step-up transformer HVT,
[Step-up winding NHV1, high-voltage rectifier diode DHV1], [Step-up winding NHV2, high-voltage rectifier diode DHV2], [Step-up winding N
HV3, high-voltage rectifier diode DHV3], [boost winding NHV4,
A so-called multisingle-type half-wave rectification circuit in which five sets of half-wave rectification circuits of a high-voltage rectifier diode DHV4 and a booster winding NHV5 and a high-voltage rectifier diode DHV5 are connected in series is formed.

Therefore, on the secondary side of the step-up transformer HVT, an operation is performed in which five sets of half-wave rectifier circuits rectify the current induced in the step-up windings NHV1 to NHV5 and charge the smoothing capacitor COHV. The smoothing capacitor COHV
, A DC high voltage (anode voltage) EHV having a level corresponding to about five times the induced voltage induced in each of the boost windings NHV1 to NHV5 is obtained.

A series circuit composed of a resistor R1 and a resistor R2 is connected between the cathode of the high voltage rectifier diode DHV1 and the secondary side ground, and a voltage divided by the resistors R1 and R2 is used as a control circuit. 2 is input. Control circuit 2
Is also configured by an error amplifier or the like, and outputs, for example, a control voltage corresponding to a change in the voltage level of the DC high voltage EHV as a control signal. The control signal from the control circuit 2 is applied to the gate of the auxiliary switching element Q2 of the active clamp circuit 3 in a state where the primary side and the secondary side are DC-insulated through the photocoupler PC. Thus, the auxiliary switching element Q2 performs a switching operation at a required ON / OFF timing according to the DC high voltage level. That is, the switching operation is performed by PWM control for variably controlling the ON period (conduction angle) of the auxiliary switching element Q2.

When the auxiliary switching element Q2 performs the switching operation, the resonance voltage V1 generated in the primary side parallel resonance capacitor Cr when the main switching element Q1 is turned off is clamped, and the voltage level changes with the fluctuation of the DC high voltage level. It will be variably controlled accordingly. Accordingly, the current level of the current I3 input to the primary winding N4 of the step-up transformer HVT also changes from the resonance voltage V1 via the series resonance circuit including the series resonance capacitor C3 and the primary winding N4. The voltage level of the resonance voltage V4 generated at both ends of the winding N4 is variably controlled. As a result, the level of the induced voltage induced on the secondary side of the step-up transformer HVT is varied, and the DC high voltage EHV output from the high voltage generation circuit 4 is stabilized.

As described above, in the power supply circuit according to the present embodiment shown in FIG. 1, the switching frequency of main switching element Q1 and the conduction angle thereof are simultaneously controlled in accordance with the voltage level of DC output voltage EO1. The combined control system makes the DC output voltage EO1 a constant voltage, and controls the conduction angle of the auxiliary switching element Q2 of the active clamp circuit 3 in accordance with the voltage level of the DC high voltage EHV. The voltage EHV is made constant.

When the power supply circuit shown in FIG. 1 is actually constructed, the primary winding N1 of the isolated converter transformer PIT =
45T, secondary winding N2 = 45T, tertiary winding N3 = 22T,
Primary parallel resonance capacitor Cr = 2200 PF, secondary parallel resonance capacitor C2 = 0.015 μF, series resonance capacitor C3 = 0.015 μF, clamp capacitor CC
L = 0.047 μF, the primary winding N4 of the step-up transformer HVT = 35T, the step-up windings NHV1 to NHV5 = 460T, the smoothing capacitor COHV = 2000PF, and the resistor R1 = 1GΩ.

FIG. 2 shows an example of the operation waveform of each part of the power supply circuit shown in FIG. In FIG. 2, for example, the AC input voltage VAC = 100 V, the secondary side DC output voltage EO1
The load power of the DC output voltage is 135 W (135 V × 1 A), the load power of the DC output voltage EO3 is 10 W (200 V × 50 mA), and the DC high voltage EHV is 30 KV.
The operation waveform under the condition that the high-voltage load power is the maximum load power and the operation waveform under the condition that the high-load power is the minimum load power are shown. FIGS. 2A to 2G show a high voltage generation circuit 4.
Is the maximum load power Pomax = 60 W (I
FIG. 2 shows operation waveforms under the condition of HV = 2 mA).
(H) to FIG. 2 (n), for example, the high-voltage load power of the high-voltage generation circuit 4 is the minimum load power Pomin = 0W (IHV = 0m
The operation waveform under the condition A) is shown.

When the high voltage load of the high voltage generating circuit 4 is set to the maximum load power, the switching cycle of the main switching element Q1 is, for example, 12 μS (switching frequency = 8).
(3.3 kHz). in this case,
As shown in FIG. 2A, a substantially sinusoidal wave is formed at both ends of the primary-side parallel resonance capacitor Cr during a period TOFF when the switching main switching element Q1 is turned off, as shown in FIG. A resonance voltage V1 having a pulse waveform is obtained. A collector current I1 as shown in FIG. 2B flows through the main switching element Q1.

In the power supply circuit of the present embodiment, the active clamp circuit 3 is connected in parallel with the primary winding N1 of the isolated converter transformer PIT, and the active clamp circuit 3 conducts, thereby causing the clamp diode DD2 → Current flows through the path of the clamp capacitor CCL. In this case, as shown in FIG. 2C, the clamp current I4 becomes a sawtooth wave that flows in the positive direction as time elapses from the negative direction. When the active clamp circuit 3 is conducting, most of the current flows to the clamp capacitor CCL as the clamp current I4, and the primary side parallel resonance capacitor C
It is assumed that the flow hardly flows for r. Therefore, while the active clamp circuit 3 is conducting, the resonance voltage V1 generated in the primary side parallel resonance capacitor Cr is clamped, and as a result, as shown in FIG. It is suppressed to about 550 Vp.

By performing the switching operation as described above, the secondary parallel resonance capacitor C2 provided on the secondary side of the insulating converter transformer PIT is placed in FIG.
In the positive period in which the resonance voltage V2 as shown in (d) is generated and the rectifier diode DO1 operates, a voltage clamped at 135 Vp is obtained.

As shown in FIG. 2E, a sine-wave current I is applied to the series resonance capacitor C3 connected to the primary winding N1 of the insulating converter transformer PIT.
3, the primary winding N4 of the step-up transformer HVT
2 (f), a sinusoidal resonance voltage V4 having a level of 400 Vp is generated at both ends.

On the other hand, when the high-voltage load of the high-voltage generation circuit 4 is set to the minimum load power (no load), control is performed so that the switching cycle of the main switching element Q1 is, for example, 11 μS (switching frequency = 90.9 kHz). You. In this case as well, both ends of the primary-side parallel resonance capacitor Cr are connected by the resonance operation of the primary-side parallel resonance circuit, as shown in FIG.
A resonance voltage V1 having a substantially sinusoidal pulse waveform as shown in FIG. 2G is obtained, and a collector current I1 as shown in FIG. 2H flows. In this case, by controlling the conduction period of the active clamp circuit 3 to be longer than the conduction period at the time of the maximum load power, as shown in FIG. The period during which the clamp current I3 flows becomes longer.
As a result, the resonance voltage V1 generated in the primary side parallel resonance capacitor Cr is clamped, and as a result, as shown in FIG.
Vp is suppressed.

At this time, the main switching element Q1
2 (h), the collector current I1 flows as shown in FIG. 2 (h), and the series resonant capacitor C3 flows as shown in FIG. 2 (k), the resonance current I3 suppressed to about 2 Ap. Thereby, the primary winding N of the insulating converter transformer PIT
2 is connected to the series resonance capacitor C3 connected to
The current I3 flows as shown in FIG.
A resonance voltage V4 whose voltage level is suppressed to about 380 Vp is generated at both ends of the VT primary winding N4 as shown in FIG. In this case as well, a resonance voltage V2 as shown in FIG. 2 (j) is generated in the secondary side parallel resonance capacitor C2, and its peak voltage level becomes 135 Vp during the positive period when the rectifier diode DO1 operates. It is assumed that a constant voltage is obtained.

FIGS. 2A to 2F and FIG. 2G
2 (l), when the high-voltage load power of the high-voltage generation circuit 4 fluctuates from the maximum load power Pomax to the minimum load power Pomin, the active clamp circuit 3 changes the high-voltage load power. It can be seen that the conduction angle control for variably controlling the conduction angle of the auxiliary switching element Q2 is performed in accordance with the following. That is, the active clamp circuit 3 variably controls the voltage level generated in the primary-side series resonance capacitor Cr in accordance with the DC high voltage level, thereby changing the voltage level of the resonance voltage V4 generated in the primary winding N4 of the step-up transformer HVT. Is variably controlled.

In this case, the main switching element Q
Since the switching cycle of 1 changes from 11 μs to 10 μs, it is understood that the switching frequency of the primary-side switching converter is variably controlled according to the high-voltage load fluctuation. This is because when the conduction period of the active clamp circuit 3 is variably controlled in accordance with the high-voltage load power fluctuation, the voltage level of the resonance voltage V1 generated at both ends of the primary-side series resonance capacitor Cr changes. This is because the voltage level of the DC output voltage EO1 obtained from the voltage V1 fluctuates.

In the power supply circuit of this embodiment having such a configuration, the high-voltage effective load power PHV is 6
The power conversion efficiency (ηDC−DC) of the high-voltage generating circuit 4 when 0 W (EHV = 30 KV, IHV = 2 mA) is about 92%. Therefore, the power conversion efficiency (ηDC-DC) can be improved by about 10% as compared with the high-voltage generation circuit 40 of the conventional power supply circuit shown in FIG.

Further, in the power supply circuit of the present embodiment, it is not necessary to provide the high-voltage regulator circuit provided in the conventional power supply circuit shown in FIG. 7, so that the power conversion efficiency can be improved accordingly. The overall power conversion efficiency η (AC → DC) is about 89%. As a result, 60
The AC input power Pin required to obtain the high-voltage load power of W is only about 67.4 W, and the AC input power Pin can be reduced by about 18 W as compared with the power supply circuit shown in FIG. .

In the conventional power supply circuit shown in FIG.
While the peak power that can be supplied by the switching power supply 10 and the high-voltage regulator circuit 20 is restricted, the power supply circuit according to the present embodiment shown in FIG. 1 does not require a high-voltage regulator circuit and can be supplied. Since the peak power is limited only by the switching power supply, the supply capacity can be increased. As a result,
Although not shown, for example, when a white peak window screen is displayed on a CRT screen displaying a black screen, the conventional power supply circuit causes the white window screen to bend, whereas FIG. In the power supply circuit shown in
It is possible to eliminate such image curving of the window screen of the white peak.

Further, in the power supply circuit shown in FIG. 1, the number of parts can be reduced because the high voltage regulator circuit 20 provided in the conventional power supply circuit shown in FIG. Since simplification can be achieved, the mounting area of components can be reduced. This makes it possible to reduce the size of the switching power supply. Further, in the conventional power supply circuit, two sets of relatively expensive switching elements and diode elements are required, so that the component material cost can be reduced.

Further, since the operation waveforms of the respective parts are all resonance waveforms, it is possible to suppress the switching noise generated due to the switching operation, and to suppress the switching noise required in the conventional power supply circuit. There is also an advantage that the countermeasure of the above becomes unnecessary.

Further, the power supply circuit of the present invention is not limited to the circuit configuration shown in FIG. FIG. 4 is a diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. The same parts as those of the power supply circuit shown in FIG. In the power supply circuit shown in FIG. 4, the voltage resonance type converter provided on the primary side adopts a separately-excited configuration, and is provided with a main switching element Q3 of, for example, one MOS-FET. The drain of the main switching element Q3 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the source is connected to the primary side ground. Here, the primary side parallel resonance capacitor Cr
Are connected in parallel between the drain and the source. Further, a clamp diode DD3 is connected in parallel between the drain and the source. The starting resistance RS
Is provided to supply a starting current obtained in the rectifying / smoothing line to the primary-side switching drive unit 6 when the commercial AC power is turned on.

The primary-side switching drive section 6 according to the present embodiment includes a drive circuit 7 and an oscillation circuit 8 as shown in FIG. The oscillation circuit 8, for example, as an oscillation signal fixed at 100 kHz,
An oscillation signal having a waveform corresponding to the on / off timing of 3 is output to the drive circuit 7. The drive circuit 7 converts a signal input from the oscillation circuit 8 into a voltage signal to generate a switching drive signal for driving the main switching element Q3, which is a MOS-FET, and applies the switching driving signal to the gate terminal of the main switching element Q3. I do. In response to this switching drive signal, the main switching element Q3 performs a switching operation. Note that the primary-side switching drive unit 6 according to the present embodiment having the above configuration is configured as one IC.

On the secondary side of the insulating converter transformer PIT formed as described above, similarly to the power supply circuit shown in FIG. 1, a secondary winding N2 and a tertiary winding N3 are wound. A DC output voltage EO1 is obtained by a half-wave rectifier circuit comprising a rectifier diode DO1 connected to N2 and a smoothing capacitor CO3, and a half-wave rectifier comprising a rectifier diode DO3 connected to a tertiary winding N3 and a smoothing capacitor CO3. A DC output voltage EO3 is obtained by a circuit.

Also in this case, the DC output voltage EO1 is controlled by the control circuit 1
, The inductance LB of the drive winding NB of the quadrature control transformer PRT is variably controlled based on the control current output from the control circuit 1 so that the secondary The DC output voltage EO1 output from the side is stabilized.

In the power supply circuit shown in FIG. 4, the circuit configuration of the high voltage generation circuit 5 for obtaining the DC high voltage EHV is different from the power supply circuit shown in FIG.
The high-voltage generation circuit 5 surrounded by a dashed line is composed of a step-up transformer HVT and a high-voltage rectifier circuit.
On the secondary side of the step-up transformer HVT, for example, a set of step-up windings NHV1 is divided and wound by slit winding or interlayer winding. In this case as well, the primary winding N4 and the boost winding NHV1 are wound so that the polarities (winding directions) are opposite to each other, and the boost winding NVT is wound on the secondary side of the boost transformer HVT.
A boosted voltage is obtained by the turn ratio between the HV1 and the primary winding N4 (NHV1 / N4). Boost winding NVH
The winding end portion 1 is connected to a connection point between the anode of the high-voltage rectifier diode DHV1 and the cathode of the rectifier diode DHV2 via a high-voltage capacitor CHV1 composed of, for example, a film capacitor or a ceramic capacitor. Anode of high voltage rectifier diode DHV3 and high voltage rectifier diode DHV4 through connection
Is connected to the connection point of the cathode. On the other hand, the winding start end of the boost winding NHV1 is connected to a connection point between the negative electrode of the smoothing capacitor COHV1 and the positive electrode of the smoothing capacitor COHV2. The anode of the high-voltage rectifier diode DHV2 and the cathode of the high-voltage rectifier diode DHV3 are connected to a connection point between the negative electrode of the smoothing capacitor COHV1 and the positive electrode of the smoothing capacitor COHV2. The smoothing capacitors COHV1 and COHV2 are connected in series by connecting the negative electrode of the smoothing capacitor COHV1 and the positive electrode of the smoothing capacitor COHV2.
The positive electrode of the smoothing capacitor COHV1 is connected to the high voltage rectifier diode DHV
1 and connected so that the negative electrode of the smoothing capacitor COHV2 is connected to the secondary side ground.

In such a connection form, as a result,
[High voltage capacitor CHV1, high voltage rectifier diode DHV1,
DHV2, a smoothing capacitor COHV1], and a second voltage doubler rectifying circuit including a set of [high voltage capacitor CHV2, rectifying diodes DHV3, DHV4, smoothing capacitor COHV2]. The outputs of these first and second voltage doubler rectifier circuits (smoothing capacitors COHV1,
COHV2) are provided connected in series. As a whole rectifier circuit combining the first and second voltage doubler rectifier circuits, four ends of the alternating voltage obtained in the boost winding NHV1 are provided at both ends of the series-connected smoothing capacitor COHV1 and smoothing capacitor COHV2. A secondary output voltage corresponding to the double is obtained. That is, a quadruple voltage full-wave rectifier circuit is formed as an entire rectifier circuit combining the first and second voltage doubler rectifier circuits.

As an operation of such a four-fold voltage full-wave rectifier circuit, when a switching output is obtained on the primary winding N4 by the switching operation of the primary side of the step-up transformer HVT, this switching output is excited by the step-up winding NHV1. Is done. The quadruple voltage rectifier circuit performs the rectification operation by inputting the alternating voltage obtained to the boost winding NHV1. At this time, the [high-voltage capacitor CHV1, high-voltage rectifier diodes DHV1, DHV1
The operation of the first voltage doubler rectifier circuit comprising HV2 and smoothing capacitor COHV1] will be described below. First, rectifier diode DHV
During the period when 1 is off and the rectifier diode DHV2 is on, an operation of charging the series resonant capacitor CHV1 with the rectified current rectified by the high-voltage rectifier diode DHV2 is obtained. During the period in which the rectifier diode DHV2 is turned off and the rectifier diode DHV1 is turned on and the rectifying operation is performed, an operation in which the potential of the high-voltage capacitor CHV1 is added to the voltage induced in the boost winding NHV1 is obtained. At both ends of COHV1, a boost winding N
A DC voltage (rectified smoothed voltage) corresponding to almost twice the induced voltage of HV1 is obtained. In addition, [High voltage condenser CHV2,
High voltage rectifier diodes DHV3, DHV4, smoothing capacitor COH
V2], the same operation is performed in the second voltage doubler rectifier circuit, which includes a pair of smoothing capacitors COHV2.
A DC voltage corresponding to approximately twice the induced voltage of the boost winding NHV1 is obtained.

As described above, the voltage doubler rectification operation is performed by each of the first and second voltage doubler rectifier circuits. As a result, a step-up winding is provided across both ends of the series-connected smoothing capacitor COHV1 and smoothing capacitor COHV2. The secondary side DC output voltage EHV corresponding to almost four times the induced voltage induced on the line NVV1
The DC high voltage EHV obtained at both ends of the smoothing capacitors COHV1 to COHV2 is used as the anode voltage of the CRT.

Also in this case, a series circuit consisting of a resistor R1 and a resistor R2 is connected in parallel to the smoothing capacitor COHV2, and a voltage divided by the resistors R1 and R2 is input to the control circuit 2. The control circuit 2 also includes an error amplifier and the like. A control signal corresponding to a voltage divided by the resistors R1 and R2, that is, a control signal corresponding to a level change of the DC high voltage EHV is transmitted to the primary side and the secondary side via the photocoupler PC. With the DC side insulated on the side, the primary side switching drive unit 6
Is input to the oscillation circuit 8 in the inside.

Therefore, in the power supply circuit shown in FIG. 4, the same effects as those of the power supply circuit of FIG. 1 described above can be obtained. In addition, since only one set of the boost winding NHV1 wound around the secondary side of the boost transformer HVT is required, the size of the boost transformer HVT can be reduced as compared to the power supply circuit shown in FIG.

In this embodiment, the orthogonal control transformer PRT is used as the control transformer for performing the constant voltage control under the configuration having the self-excited resonance converter for the primary side. Instead of the orthogonal control transformer PRT, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, as in the case of the orthogonal control transformer, for example, 4
A three-dimensional core is formed by combining two sets of double U-shaped cores having two magnetic legs. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, one of the control winding NC and the drive winding NB is
Of the four magnetic legs, winding is performed on two magnetic legs adjacent to each other, and the other winding is wound on two magnetic legs that are considered to be in a diagonal positional relationship. It is to be wound.
When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. This allows
The current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is shortened, and the switching element is turned off. The power loss can be further reduced.

[0083]

As described above, in the switching power supply circuit according to the present invention, a voltage resonance type converter is formed on the primary side of the insulating converter transformer, and a DC output voltage generator for obtaining a DC output voltage is formed on the secondary side. Means and DC high voltage generating means for obtaining a DC high voltage. By providing active clamping means on the primary side of the insulating converter transformer, the output DC output voltage and the output DC high voltage are stabilized. In this case, in order to finally generate a high DC voltage from the DC input voltage obtained by rectifying the commercial AC power supply, two power conversion parts, which conventionally required three stages, can be used. It becomes possible to reduce.

When such a switching power supply circuit of the present invention is applied to a video device such as a large color television receiver or a projector device having a high resolution, for example, the power conversion efficiency of the DC high voltage generating means is reduced. As a result, the overall power conversion efficiency can be improved, and the power loss in the switching power supply circuit can be significantly reduced.

Further, the switching power supply circuit of the present invention
Since the peak power supply capability can be increased as compared with the conventional power supply circuit, it is possible to eliminate image bending or the like when a white peak window screen is displayed on a CRT screen, for example. Further, since the circuit configuration can be simplified as compared with the conventional power supply circuit, the cost of parts can be significantly reduced, and the switching power supply circuit can be downsized. Furthermore, since the operation waveforms of the respective units are all resonance waveforms, there is an advantage that noise generated due to the switching operation is suppressed and a measure for suppressing the noise is not required.

Further, if the DC high voltage generating means is constituted by a quadruple voltage rectifier circuit, only one boost winding needs to be wound on the secondary side of the boost transformer, so that the size of the boost transformer can be reduced. It becomes possible.

[Brief description of the drawings]

FIG. 1 is a diagram showing a configuration of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 3 is a sectional view showing a configuration of a boosting transformer provided in the power supply circuit shown in FIG.

FIG. 4 is a diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention.

FIG. 5 is a cross-sectional view illustrating a structure of an insulating converter transformer.

FIG. 6 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 7 is a diagram showing a configuration of a power supply circuit provided in a conventional high-resolution television receiver.

8 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

9 is a diagram illustrating a variation characteristic of each unit with respect to a change in a beam current of the power supply circuit illustrated in FIG. 7, and a diagram illustrating a voltage conversion efficiency characteristic of a high-voltage generation circuit.

[Explanation of symbols]

1 2 control circuit, 3 active clamp circuit, 4
5 High voltage generation circuit, 6 Primary side switching drive, 7
Drive circuit, 8 oscillation circuit, AC commercial AC power supply, C
i Smoothing capacitor, Cr primary parallel resonance capacitor, C2 secondary parallel resonance capacitor, C3 series resonance capacitor, CB resonance capacitor, CCL clamp capacitor, COHV COHV1 COHV2 smoothing capacitor, Di
Bridge rectifier circuit, DD1 to DD3 Clamp diode, DHV1 to DHV5 High voltage rectifier diode, DO1 DO3
Rectifier diode, HVT step-up transformer, NHV1 to NHV5
Boost winding, N1 N4 primary winding, N2 secondary winding, N3
Tertiary winding, NB Ng drive winding, NC control winding, NR
Controlled winding, PC photocoupler, PIT isolation converter transformer, PRT orthogonal control transformer, Q1 Q3
Main switching element, Q2 auxiliary switching element, R1 R2 Rg RS RB resistance

Claims (4)

[Claims] 1. A switching means comprising a main switching element for intermittently outputting an input DC input voltage, and a primary-side parallel resonance circuit which makes the operation of the switching means a voltage resonance type. And a primary side parallel resonance capacitor provided to transmit the output of the primary side to the secondary side, a primary side winding is wound on the primary side, and a secondary side is wound on the secondary side. An insulation converter transformer, on which the side winding is wound, and a required degree of loose coupling between the primary winding and the secondary winding is obtained; and A secondary-side parallel resonance circuit formed by connecting a secondary-side parallel resonance capacitor to the winding in parallel; and a secondary-side resonance circuit formed including the secondary-side parallel resonance circuit, from the secondary-side winding. Of the resulting alternating voltage in the positive period By performing half-wave rectification operation for the pressure, and the DC output voltage generation means configured to obtain a DC output voltage, depending on the DC output voltage level, thereby variably controlling the switching frequency of the main switching element,
First constant voltage control means for performing constant voltage control by switching the main switching element so as to make the on period variable while keeping the off period within the switching cycle constant; A booster transformer formed to obtain a boosted voltage boosted to a predetermined level from the secondary side by transmitting a voltage input to the secondary side to the secondary side, and a primary operation of the booster transformer as a resonance operation. A serial resonance capacitor formed in series with at least the primary winding of the step-up transformer, and a switching output of the switching means is branched and supplied. DC high voltage generation means configured to obtain a DC high voltage by performing a rectification operation on the boosted voltage obtained on the secondary side; A series connection circuit of a pump capacitor and an auxiliary switching element, wherein the series connection circuit includes active clamp means connected in parallel to a primary winding of the insulating converter transformer, and according to the DC high voltage level. And a second constant voltage control means for performing a constant voltage control by performing a conduction angle control of the auxiliary switching element. 2. The step-up high-voltage generating means includes: a plurality of step-up windings wound independently on a secondary side of the step-up transformer; and the step-up winding obtained by each of the step-up windings. A multiple rectifier circuit formed by performing a half-wave rectification operation on a voltage and connecting a plurality of rectifier circuits provided in series so as to obtain an output voltage of a level corresponding to substantially equal to the boosted voltage, The switching power supply circuit according to claim 1, comprising: 3. The DC high voltage generating means performs a rectifying operation on one boost winding wound on the secondary side of the boost transformer and a boost voltage obtained on the boost winding. The switching power supply circuit according to claim 1, further comprising: a quadruple voltage rectifier circuit configured to obtain a DC high voltage having a level corresponding to approximately four times the boosted voltage. 4. The switching power supply according to claim 1, wherein the second constant voltage control means is provided with a photocoupler for DC-insulating the primary side and the secondary side in a DC manner. circuit.