JP2001218460A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve various types of performance such as power conversion efficiency as a power circuit outputting high DC voltage used for CRT anode voltage. SOLUTION: A booster wiring NHV is wound on the secondary of an isolation converter transformer PIT forming a switching power circuit, and alternating voltage VHV is inputted into a multiplying-rectifier circuit 2 to obtain the high DC voltage to be kept at a prescribed high voltage level. It is thus possible to provide a structure of obtaining high DC voltage, for example, necessary for the horizontal deflection of a TV receiving set without using a horizontal deflection circuit system.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices, and more particularly to a switching power supply circuit for outputting a high voltage used for a required application.

[0002]

2. Description of the Related Art Conventionally, some electronic devices such as a television receiver and a projector device have a cathode ray tube (CRT) for displaying images. In a television receiver equipped with a cathode ray tube (hereinafter referred to as "CRT"), as is well known, an electron beam output from an electron gun provided inside a CRT is horizontally (horizontally) directed. And a vertical deflection system block for deflecting in the vertical direction (vertical direction). Further, the horizontal deflection circuit block is provided with a high voltage generation circuit for supplying a high voltage of, for example, about 20 kV to 35 kV to the anode electrode of the CRT.

FIG. 8 is a diagram showing a configuration of a horizontal deflection circuit system block provided in a television receiver and a peripheral circuit thereof. Switching power supply circuit 1 shown in FIG.
Reference numeral 0 denotes a DC-DC converter that performs switching with respect to the input DC voltage, and finally converts the DC voltage to a predetermined voltage level and outputs the DC voltage. A rectifying and smoothing circuit including a full-wave rectifying bridge rectifying circuit Di and a smoothing capacitor Ci is provided at a stage preceding the switching power supply circuit 10. The rectifying and smoothing circuit rectifies and smoothes a commercial AC power supply (AC input voltage VAC). To obtain a DC voltage Ei. The DC voltage Ei is input to the switching power supply circuit 10. In this case, the switching power supply circuit 10 outputs a DC output voltage EO (EO1, EO2, EO) converted to a predetermined voltage level.
3) is output. The actual voltage levels of the respective DC output voltages EO1, EO2, EO3 are, for example, DC output voltage EO1 = 135V, DC output voltage EO2 = 1.
5V and the DC output voltage EO3 = 7V.

The horizontal oscillation circuit 20 receives a horizontal synchronizing signal fH included in a video signal or the like. The oscillation frequency (15.75K) corresponding to the horizontal synchronizing signal fH
Hz), and outputs a pulse voltage synchronized with the horizontal synchronization signal fH.

[0005] A horizontal drive circuit 30 surrounded by an alternate long and short dash line amplifies a pulse output from the horizontal oscillation circuit 20,
A sufficiently large drive current (drive current) is supplied to a horizontal output circuit 40 described later. In this case, the configuration of the horizontal drive circuit 30 includes a horizontal output circuit 40 serving as a load.
In order to prevent the period of the pulse output supplied from the horizontal oscillation circuit 20 from fluctuating, a transformer-coupled amplifier circuit having a common emitter is usually used.

In the horizontal drive circuit 30, as shown, for example, the base of a transistor Q11 is connected to the horizontal oscillation circuit 20 via a capacitor C11, so that a pulse voltage from the horizontal oscillation circuit 20 is applied to the transistor Q11. Is entered in the base. Also, its base-
A bias resistor R11 is inserted between the emitters, and a predetermined bias voltage is applied to the base. The collector of the switching element Q11 is a horizontal drive transformer H
The DT is connected to a secondary output terminal (DC output voltage EO1) of the switching power supply circuit 10 via a primary winding N11 and a collector resistor R13, and the emitter is grounded. Further, a damping circuit comprising a series connection circuit of a capacitor C12 and a resistor R12 is provided between the collector and the emitter. The damping circuit composed of the series connection of the capacitor C12 and the resistor R12 prevents the surge current and the oscillating current (ringing current) from being superimposed on the current flowing through the primary winding N11 of the horizontal drive transformer HDT. Preventing. Further, a capacitor C13 provided between the winding start end of the primary winding N11 of the horizontal drive transformer HDT and the secondary side ground is a capacitor for removing noise.

The winding start end of the primary winding N11 of the horizontal drive transformer HDT is connected to the DC output terminal (DC output voltage EO1) of the switching power supply circuit 10 via the collector resistor R13, and the winding end end thereof. The part is connected to the collector of the transistor Q11. The winding end of the secondary winding N21 is connected to the base of an output transistor Q12 of a horizontal output circuit 40, which will be described later, and the winding start is grounded to ground.

In the transistor Q11, the horizontal oscillation circuit 20
Output that is obtained by amplifying the horizontal oscillation output output from
This amplified output is transmitted to the primary winding N11 of the horizontal drive transformer HDT. And the horizontal drive transformer H
In the DT, the output obtained from the primary winding N11 is transmitted to the secondary winding N21. In this case, the horizontal drive transformer HD
The primary winding N11 and the secondary winding N21 of T are wound so that the polarities (winding directions) thereof are opposite to each other.

A horizontal output circuit 40 surrounded by a dashed line.
Amplifies the output obtained from the secondary side of the horizontal drive transformer HDT, thereby scanning the electron beam output from the electron gun of the CRT in the horizontal direction.
Generate. At the same time, a high-voltage generation circuit 50 described later
Is configured to generate a flyback pulse for generating a high voltage.

In the horizontal output circuit 40, the base of the output transistor Q12 is connected to the horizontal drive transformer HDT.
Of the switching power supply circuit 10 via a primary low-voltage winding NLV of a flyback transformer FBT to be described later (secondary output voltage (secondary output voltage)). EO1). The emitter is grounded. A damper diode D11 and a horizontal retrace capacitor Cr1 are connected in parallel between the collector and the emitter of the output transistor Q12. Further, a series connection circuit including [horizontal deflection yoke HDY, horizontal linear correction coil HLC, and S-shaped correction capacitor CS1] is connected between the collector and the emitter.

The horizontal output circuit 40 constructed as described above
Then, the capacitance of the horizontal retrace capacitor Cr1 and
The leakage inductance component L LV of the primary low voltage winding N LV of the flyback transformer FBT forms a voltage resonance type converter. The output transistor Q12 is output by the secondary output of the horizontal drive transformer HDT.
Are turned on / off, a horizontal deflection current IDY having a sawtooth waveform flows through the horizontal deflection yoke HDY. Further, during the period when the output transistor Q12 is off, the horizontal retrace capacitor C is set due to the resonance operation of the inductance LDY of the horizontal deflection yoke HDY and the capacitance of the horizontal retrace capacitor Cr1 and the action of the damper diode D11.
At both ends of r1, a pulse voltage (flyback pulse voltage) V11 which is a relatively high voltage is generated. Although the operations of the horizontal straight line correction coil HLC and the S-shaped correction capacitor CS1 are omitted, for example, the operation is performed so as to correct the horizontal deflection current IDY to correct the distortion of the image displayed on the screen of the CRT. I have.

A high-voltage generating circuit 50 surrounded by a dashed line.
Is a flyback transformer FBT (Fly Back Tra
nsformer) and a high-voltage rectifier circuit, and boosts the flyback pulse voltage V11 generated by the horizontal output circuit 40 to generate a high voltage corresponding to, for example, an anode voltage level of a CRT.

Here, the flyback transformer FBT
Will be described with reference to the sectional view of FIG. In the flyback transformer FBT shown in this figure, the K-shaped core CR30 is formed by combining the magnetic legs of the two U-shaped cores CR10 and CR20 so as to face each other. A gap G is provided at a portion where the magnetic leg end of the U-shaped core CR10 and the magnetic leg end of the U-shaped core CR20 face each other. And, as shown in the figure, with respect to one magnetic leg of the K-shaped core CR30,
By attaching the low-voltage winding bobbin LB and the high-voltage winding bobbin HB, the primary-side low-voltage winding NLV and the step-up winding NHV are divided and wound on the low-voltage winding bobbin LB and the high-voltage winding bobbin HB, respectively. I wear it. in this case,
The low voltage winding bobbin LB is made of a single wire and the primary low voltage winding N
LV is wound, and the high-voltage winding bobbin HB is similarly wound with the boost winding NHV using a single wire. At this time, for example, a plurality of boost windings NVH need to be wound on the high-voltage winding bobbin HB in an insulated state.
It is a so-called interlayer winding in which an interlayer film F is inserted and wound up for each of two winding layers obtained by winding each boost winding NHV a predetermined number of times.

The description returns to FIG. As a practical example of the flyback transformer FBT having the structure shown in FIG. 12, for example, as shown in FIG. 8, five sets of boost windings NVH1, NVH2, NHV3, NHV4, and NHV5 are divided as the boost winding NHV. Each is wound independently. It should be noted that only one winding is wound as the primary low-voltage winding NLV.
Here, each boost winding NHV1 with respect to the primary side low voltage winding NLV1
HVNHV5 are wound so that the polarity (winding direction) is opposite. The winding start end of the primary side low voltage winding NLV is connected to the secondary side output terminal (DC output voltage E
O1), and the winding end is connected to the output transistor Q12.
Connected to the collector. In addition, the boost winding N
For each of HV1 to NHV5, the high voltage rectifier diodes DHV1, DHV2, DHV3, DHV
4, the anode side of DHV5 is connected. The cathode of the high-voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV via the resistor RHV. The cathodes of the high-voltage rectifier diodes DHV1, DHV2, DHV3, DHV4, and DHV5 are connected to the boost windings NHV1, NHV2, NHV
3, Connected to the winding start end of NHV4.

[0015] In such a connection form, the step-up winding NVH
1, high-voltage rectifier diode DHV1, [boost winding NVH2, high-voltage rectifier diode DHV2], [boost winding NHV3, high-voltage rectifier diode DHV3], [boost winding NVH4, high-voltage rectifier diode DHV4], [boost winding NVH5, High voltage rectifier diode D
HV5], and five sets of half-wave rectifier circuits are formed.
These five half-wave rectifier circuits are connected in series.

Therefore, on the secondary side of the flyback transformer FBT, five sets of half-wave rectifier circuits
~ NHV5 is rectified to smooth the capacitor CO
By performing the operation of charging the HV, a DC voltage having a level corresponding to five times the voltage induced in each of the boost windings NHV1 to NHV5 is obtained at both ends of the smoothing capacitor COHV. That is, a five-fold voltage half-wave rectifier circuit is formed. The DC voltage obtained at both ends of the smoothing capacitor COHV is converted into a DC high voltage EHV, which is used, for example, as an anode voltage of a CRT.

It is to be noted that the resistor R1, which is inserted between the cathode of the high voltage rectifier diode DHV3 and the secondary side ground,
A series connection circuit composed of a variable resistor R2 and a resistor R3] is provided to obtain a voltage level lower than the DC high voltage EHV, and outputs a DC output voltage EFV used as a focus voltage of a CRT, for example.

FIG. 9 shows the operation waveform of each part of the circuit shown in FIG. In the circuit shown in FIG. 8, since the pulse voltage is input to the base of the output transistor Q12 from the horizontal oscillation circuit 20 amplified by the horizontal drive circuit 30, the switching frequency of the output transistor Q12 is
This corresponds to the synchronization frequency (15.75 KHz) of the horizontal synchronization signal fH. For example, as shown in the figure, the ON period (horizontal scanning period) TON of the output transistor Q12 is 52.
7μs, OFF period (horizontal retrace period) TOFF is 10.8μ
s, and a period (63.5 μS) of the total of the period TON and the period TOFF corresponds to the period of the horizontal synchronizing signal fH.

In this case, a collector current IC having a waveform as shown in FIG. 9D flows through the collector of the output transistor Q12 due to the on / off operation of the switching element Q12. As a result, the primary current I11 having a waveform as shown in FIG. 9C flows through the primary low-voltage winding NLV of the flyback transformer FBT, and the primary current I11 shown in FIG.
The horizontal deflection current IDY having the waveform shown in FIG.

At this time, the voltage V11 across the horizontal retrace capacitor Cr1 connected in parallel between the collector and the emitter of the output transistor Q12 is turned on when the output transistor Q12 is turned on, as shown in FIG. During the period TON, the level becomes 0. In the period TOFF during which the output transistor Q12 is turned off, for example, 1000 Vp to 12 V, due to the resonance operation of the inductance component LDY of the horizontal deflection yoke HDY and the capacitance of the horizontal retrace capacitor Cr1.
A flyback pulse voltage V11 of about 00 Vp is generated.

In the high-voltage generating circuit 50 shown in FIG. 8, the flyback pulse voltage V11
The positive pulse voltage applied to the primary side of the flyback transformer FBT is boosted, and a predetermined DC high voltage E is applied from the secondary side.
I try to get HV. For example, horizontal retrace capacitor C
When a flyback pulse voltage V11 of 1000 Vp to 1200 Vp is generated at both ends of r1, as shown in FIG. 10, the primary low-voltage winding NLV of the flyback transformer FBT, as shown in FIG.
Is applied with a positive pulse voltage of about 900 Vp.
As a result, an induced voltage is generated in each of the boost windings NHV1 to NHV5 by boosting the positive pulse voltage to about 6.5 kV. The high voltage generating circuit 50 has five sets of boost windings NVH
Since 1 to NHV5 are wound and a five-fold voltage half-wave rectifier circuit is provided, the high voltage generation circuit 50 outputs a DC high voltage EHV of about 32 kV.

Incidentally, such a flyback transformer F
The number of windings of the primary side low-voltage winding NLV and the boost windings NHV1 to NHV5 of the BT is, for example, after winding about 500T (turn) around the high-voltage winding bobbin HB as each of the boost windings NHV1 to NHV5. The primary low-voltage winding NLV is wound around the low-voltage winding bobbin LB for a predetermined turn so that a predetermined DC high voltage EHV is obtained.

[0023]

The circuit shown in FIG. 8 uses a flyback pulse voltage V11 obtained by the horizontal output circuit 40 to obtain a DC high voltage EHV from a high voltage generating circuit 50. I have. Therefore, the power conversion efficiency when converting the input power to the high-voltage load power is about 70%, and the reactive power when obtaining the high-voltage load power is relatively large.

In the high-voltage generating circuit 50, a positive pulse voltage (flyback pulse voltage) input to the primary low-voltage winding NLV of the flyback transformer FBT is used to generate an induced voltage induced in each boost winding NHV. The peak value of the induced voltage is half-wave rectified by each of the high-voltage rectifier diodes DHV to obtain a DC high voltage EHV.
However, in this case, the conduction angle of the high-voltage rectifier diode DHV is narrow, and equivalently the power supply impedance is high, so that the voltage level of the DC high voltage EHV is easily affected by the fluctuation of the high-voltage load.

For example, when the circuit shown in FIG. 8 is applied to a television having a CRT screen size of 29 inches or more, the high-voltage generation circuit 50 outputs an anode electrode from the high-voltage generation circuit 50 in order to secure the CRT screen brightness. Must be supplied with a beam current IHV of 2 mA or more. That is, CR
Assuming that the voltage level of the DC high voltage EHV supplied to the anode electrode of T is 32 kV, for example,
The high-voltage load power from 0 is 64 W (32 kV × 2 m
A) Yes. Therefore, from the high voltage generation circuit 50,
At least 0W (IHV = 0mA) as high voltage load power
It is conceivable that it fluctuates up to 64 W (IHV = 2 mA).

As an example, the high voltage load power is set to 0 W (IHV = 0).
mA) to 64 W (IHV = 2 mA),
FIG. 11 shows how the DC high voltage EHV output from the high voltage generation circuit 50 changes. In this case, the high-voltage load power is 0 W
When (IHV = 0 mA), the voltage level of the DC high voltage EHV is 32 kV. On the other hand, when the high-voltage load power increases to 64 W (IHV = 2 mA), the DC high voltage drops to about 30.5 kV due to the voltage drop due to the high-voltage rectifier diode DHV and the inrush current limiting resistor RHV. That is, when the circuit shown in FIG. 8 is applied to an actual television receiver or the like, the voltage level width ΔEHV of the DC high voltage EHV in the actual use range (0 W to 64 W) of the high voltage load power is about 1.5 kV. become.

When the voltage level of the DC high voltage EHV fluctuates in this way, for example, when the current value of the horizontal deflection current IDY is constant, the horizontal amplitude of the electron beam output from the CRT changes. For this reason, in an actual television receiver, a zooming correction circuit or the like for correcting the current value of the horizontal deflection current IDY is provided in a horizontal output circuit so that the horizontal amplitude of the electron beam does not change due to the fluctuation of the DC high voltage EHV. 40 had to be provided.

Further, as described above, since the flyback transformer FBT is wound on only one magnetic leg, the flyback transformer FBT receives a signal from the gap G of the other magnetic leg that is not wound. Ringing (vibration) may occur due to the leakage magnetic flux or the distributed capacitance of the leakage inductance of the boost winding NVH. For example, when ringing (vibration) occurs at the timing when the induced voltage induced in the boost winding NHV becomes a negative level as shown in FIG. 10 due to the leakage inductance of the boost winding NHV, as shown in FIG. Primary current I11 flowing through the primary side of the flyback transformer FBT
Is superimposed with a ringing component. In this case, FIG.
Since the ringing current component is also superimposed on the horizontal deflection current IDY shown in (1), raster ringing occurs at the left end of the screen of the CRT, for example. For this reason, in an actual television receiver, some measures are required to prevent raster ringing.

In the flyback transformer FBT, since a DC component is superimposed on the primary current (flyback current) I11 flowing through the primary low voltage winding NLV, the core shape is set so that the flyback transformer FBT is not saturated. Or the primary low-voltage winding N through which the primary current I11 flows.
It is necessary to increase the winding diameter of LV. As a result, there is also a disadvantage that the shape of the flyback transformer FBT becomes large.

When the DC current component is superimposed on the primary current I11 flowing through the primary side of the flyback transformer FBT, the peak level of the collector current IC flowing through the output transistor Q12 increases accordingly. For this reason, it is necessary to take measures such as forming the output transistor Q12 by a high withstand voltage transistor that can withstand a large amount of power or attaching a heat sink or the like for suppressing heat generation of the output transistor Q12.

[0031]

In view of the above-mentioned problems, the present invention provides a switching power supply circuit as follows. A switching means for intermittently outputting the input DC input voltage with a switching element, and a switching means provided for transmitting the output of the primary side to the secondary side, and having a primary winding wound on the primary side, On the side the first secondary winding and the second
And a required degree of loose coupling between the primary winding and the first secondary winding is obtained. (2) an insulated converter transformer capable of obtaining a tightly coupled state with respect to the secondary winding, a primary-side series resonance circuit inserted so that the operation of the switching means is of a current resonance type, and A primary-side partial resonance circuit formed so as to obtain a partial resonance operation during a switching operation. A DC output voltage generating means configured to obtain a secondary DC output voltage by inputting an alternating voltage obtained in the first secondary winding and performing a rectifying operation; Constant voltage control means for performing constant voltage control by varying the switching frequency of the switching element according to the voltage level, and rectifying operation by inputting an alternating voltage obtained to the second secondary winding DC high voltage generating means configured to obtain a DC high voltage that is set to a predetermined high voltage level by performing the operation.

According to the above-described structure, the present invention provides a switching system including a series resonance circuit for forming a current resonance type converter on the primary side and a primary side partial resonance circuit for obtaining a so-called partial resonance operation. A power supply circuit is formed. Then, a first secondary winding and a second secondary winding (boost winding) are provided to the secondary side of the insulating converter transformer for transmitting the output of the primary side current resonance type converter to the secondary side. Wrap. The alternating voltage obtained in the first secondary winding is subjected to rectification and smoothing to obtain a secondary DC output voltage, and the switching of the primary current resonance type converter is performed based on the secondary DC output voltage. By changing the frequency, low voltage control is achieved. Also, the second
The alternating voltage obtained in the secondary winding is input to a DC high voltage generating means to obtain a DC high voltage at a predetermined high voltage level. That is, in the present invention, for example, a horizontal deflection circuit is not interposed to obtain a DC high voltage required for performing horizontal deflection of a television receiver.

[0033]

FIG. 1 is a circuit diagram showing the configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure employs a configuration including a self-excited current resonance type switching converter in which two switching elements (bipolar transistors) are half-bridge-coupled. In this current resonance type switching converter, two switching elements Q1,
After Q2 is half-bridge-coupled, it is connected so as to be inserted between the connection point on the positive electrode side of the smoothing capacitor Ci and ground. In this case, the switching element Q
A bipolar transistor (BJT: junction type transistor) is used for 1 and Q2.

In this case, a rectifying and smoothing circuit composed of a bridge rectifying circuit Di and a smoothing capacitor Ci is connected to the commercial AC power supply AC, and generates a rectified and smoothed voltage corresponding to a level almost equal to the AC input voltage VAC. The current resonance type switching converter is supplied as a DC input voltage.

Each collector of the switching elements Q1 and Q2
Starting resistors RS1 and RS2 are inserted between the bases. Clamp diodes DD1 and DD2 are inserted between the bases and the emitters of the switching elements Q1 and Q2, respectively. A resonance capacitor CB1, a base current limiting resistor RB1, a drive winding NB1 is provided between the base of the switching element Q1 and the collector of the switching element Q2.
Is inserted. The resonance capacitor CB1 forms a series resonance circuit for self-excited oscillation driving together with its own capacitance and the inductance LB1 of the drive winding NB1, and thereby determines the switching frequency of the switching element Q1. Similarly, a series connection circuit composed of a resonance capacitor CB2, a base current limiting resistor RB2, and a drive winding NB2 is inserted between the base of the switching element Q2 and the primary side ground.
A series resonance circuit for self-excited oscillation is formed together with B2 and the inductance LB2 of the drive winding NB2 to determine the switching frequency of the switching element Q2.

A partial resonance capacitor Cc is connected between the contact (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 (switching output point) and the primary side ground. This partial resonance capacitor Cc
Has a capacitance sufficiently smaller than a series resonance capacitor C1 described later, and has an action of absorbing switching noise of the switching elements Q1 and Q2. The partial resonance capacitor Cc has collector currents ICI and ICI of the switching elements Q1 and Q2 due to its own capacitance and the leakage inductance component of the insulating converter transformer PIT.
A partial resonance circuit for forming C2 into a substantially sinusoidal resonance current is formed. Accordingly, at least the switching element Q corresponds to the switching frequency variably controlled by the constant voltage control operation performed as described later.
Zero voltage switching (ZV
It also has a resonance function for obtaining S: Zero Volt Switching operation and zero current switching operation (ZCS: Zero Current Switching). That is, a so-called partial resonance operation is obtained as the operation of the primary-side current resonance type converter. Thereby, the switching element Q
1, the switching loss in Q2 is reduced.

Drive transformer PRT (Power Regulatin)
g Transformer) is provided to drive the switching elements Q1 and Q2 and to perform constant voltage control by variably controlling the switching frequency. In this case, the drive windings NB1 and NB2 and the resonance current detection are performed. Winding ND
Are wound, and a control winding NC is provided for each of these windings.
Are orthogonally wound saturable reactors (orthogonal transformers) wound in the direction orthogonal to each other. One end of the drive winding NB1 of the drive transformer PRT is connected to a resonance capacitor C
The switching element Q1 is connected through a series connection of B1-resistor RB1.
And the other end is connected to the emitter of the switching element Q1. One end of the drive winding NB2 is grounded and the other end is connected to the resonance capacitor CB2−.
The resistor RB2 is connected to the base of the switching element Q2 via a series connection. The drive winding NB1 and the drive winding NB2 are wound so that voltages of opposite polarities are generated.

As the switching operation of the power supply circuit having the above configuration, when a commercial AC power supply is first turned on, for example, the switching elements Q1, Q2 are connected via the starting resistors RS1, RS2.
A starting current is supplied to the base of the switching element Q. For example, if the switching element Q1 is turned on first, the switching element Q2 is controlled to be turned off. As an output of the switching element Q1, a resonance current flows through the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1, and when the resonance current becomes zero, the switching element Q2 is turned on and the switching element Q1 is turned on. It is controlled to be off. Then, a resonance current flows in the opposite direction through the switching element Q2. Thereafter, a self-excited switching operation in which the switching elements Q1 and Q2 are turned on alternately is started. As described above, the switching element Q is used with the terminal voltage of the smoothing capacitor Ci as the operating power supply.
A primary current (drive current) I1 close to the resonance current waveform is supplied to the primary winding N1 of the insulated converter transformer PIT by alternately opening and closing 1 and Q2 to obtain an alternating output on the secondary side.

An insulation converter transformer PIT (Power Iso)
lation Transformer) transmits the switching output of the switching elements Q1, Q2 to the secondary side. Here, the structure of the insulating converter transformer PIT is shown in FIG. As shown in FIG. 4A, the insulating converter transformer PIT includes, for example, an E-shaped core CR1 made of a ferrite material,
EE combining CR2 with their magnetic legs facing each other
A shaped core CR is provided. And this EE type core CR
The primary winding N1 and the secondary winding N2 are wound around the center magnetic leg in a divided state using the divided bobbin LB1. In this case, a litz wire formed by bundling a plurality of single wires is used as the wire material of the primary winding N1 and the secondary winding N2.

A gap G is formed in the center magnetic leg of the EE-type core CR as shown in FIG. As a result, the primary winding N1 and the secondary winding N2 can be in a loosely coupled state with a required coupling coefficient. Note that the gap G is determined by each E-shaped core CR.
1 and CR2 can be formed by making the center magnetic leg shorter than the two outer magnetic legs. The coupling coefficient k between the primary winding N1 and the secondary winding N2 is, for example, k ≒ 0.8
I try to get a loosely coupled state of 5,
A saturated state is hardly obtained.

Further, in the insulating converter transformer PIT, a high-voltage winding bobbin HB1 is provided on the divided bobbin LB1 around which the secondary winding N2 is wound. Then, the step-up winding NVH is wound using the high-voltage winding bobbin HB1. In this case, a tightly coupled state is obtained between the boost winding NHV and the secondary winding N2. Further, as the wire material of the boost winding NHV, for example, while the above-described primary winding N1 and secondary winding N2 are litz wires, for example, the wire diameter is 30 μm or more.
A thin line (single line) of about 60 μm is used.

Incidentally, in the above-described isolated converter transformer PIT, as described later, the voltage level of the induced voltage VHV induced in the boost winding NVH is changed to the secondary winding N2.
Is much higher than the voltage level of the induced voltage V2. For this reason, when winding the boost winding NHV around the high-voltage winding bobbin HB1, for example, in order to ensure sufficient insulation of the boost winding NHV, for example, FIGS.
The winding method as shown in FIG.

FIG. 4B shows a so-called split winding (slit winding) in which the boost winding NHV is divided and wound on a high-voltage winding bobbin HB1 divided into a plurality of regions. When winding the step-up winding NVH by split winding,
As shown in the drawing, a plurality of slits S, which are winding areas, are formed by a partition plate DV provided integrally with the inside of the high-voltage winding bobbin HB1. By winding the step-up winding NVH around each slit S, insulation between the step-up windings NHV is obtained. In this case, the step-up winding NHV wound around the high-voltage winding bobbin HB1 is used.
Is molded with an insulating resin such as an epoxy resin EP. FIG. 4C shows that when the boost winding NHV is wound around the high-voltage winding bobbin HB1, the winding layers formed by winding the boost winding NHV are:
A form of interlayer winding in which insulation between the step-up windings NHV is obtained by inserting an interlayer film F is shown.

The description returns to FIG. In this case, one end of the primary winding N1 of the insulating converter transformer PIT is connected to the contact point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, and the other end. By being grounded to the primary side ground via the series resonance capacitor C1, a switching output is obtained.

In the above connection mode, since the series resonance capacitor C1 and the primary winding N1 are connected in series, the insulation converter transformer including the capacitance of the series resonance capacitor C1 and the primary winding N1 (series resonance winding). The leakage inductance (leakage inductance) component of the PIT forms a primary side series resonance circuit for making the operation of the switching converter a current resonance type. That is, a current resonance type converter is provided on the primary side of the power supply circuit of the present embodiment shown in FIG.

Therefore, in the circuit shown in FIG. 1, the primary-side series resonant circuit for making the operation of the primary-side switching converter a current resonance type, and the switching operation of the switching elements Q1 and Q2 as described above. In the above, a configuration in which a partial resonance circuit for obtaining a partial resonance operation is provided in a complex manner is adopted. In this specification, a switching converter configured to operate with two sets of resonance circuits, that is, a primary-side series resonance circuit and a partial resonance circuit with respect to the primary side, is described below.
It is also referred to as "complex current resonance type switching converter".

On the secondary side of the insulating converter transformer PIT, a secondary winding N2 serving as a first secondary winding and a boost winding NHV serving as a second secondary winding are wound. ing. First, the configuration on the secondary winding N2 side will be described. An alternating voltage is induced in the secondary winding N2 by the switching output supplied to the primary winding N1. Secondary winding N2
Is provided with a rectifying / smoothing circuit composed of a bridge rectifying circuit DBR and a smoothing capacitor CO1, thereby providing a DC output voltage E having a predetermined voltage level (for example, 135 V).
I try to get O1. Note that this DC output voltage EO1
Is also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage.

The control circuit 1 supplies a DC current whose level varies in accordance with the level of the DC voltage output EO1 on the secondary side to the control winding NC of the drive transformer PRT as a control current, for example. The constant voltage control is performed as described above.

For example, if the secondary output voltage EO1 fluctuates due to fluctuations in the AC input voltage and load power, the control circuit 1 controls the control winding N in accordance with the fluctuation in the secondary output voltage EO1.
The level of the control current flowing through C is variably controlled. Due to the influence of the magnetic flux generated in the drive transformer PRT by this control current, the state of the saturation tendency in the drive transformer PRT changes, which acts to change the inductance of the drive windings NB1, NB2. Control is performed so that the switching frequency changes due to a change in circuit conditions. In the power supply circuit shown in this figure, the switching frequency is set in a frequency region higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1, but when the switching frequency increases, for example, The switching frequency is set apart from the resonance frequency. Thereby, the resonance impedance of the primary side series resonance circuit with respect to the switching output increases. By increasing the resonance impedance in this way, the drive current supplied to the primary winding N1 of the primary-side series resonance circuit is suppressed, so that the secondary-side output voltage is suppressed, and the constant voltage is reduced. Control will be achieved. Hereinafter, such a constant voltage control method is referred to as a “switching frequency control method”.

Next, as a configuration on the secondary side of the insulating converter transformer PIT, a configuration on the boost winding NHV side will be described. An induced voltage induced by the primary winding N1 is generated in the boost winding NHV, similarly to the secondary winding N2. As described above, since the boost winding NHV is wound in a tightly coupled state with the secondary winding N2, the voltage level of the induced voltage induced in the boost winding NHV is It is determined by the voltage level of the resonance voltage V2 obtained at N2 and the turn ratio between the secondary winding N2 and the boost winding NHV.
That is, if the number of turns (number of turns) of the secondary winding N2 is N2, the voltage level of the resonance voltage V2 is V2, and the number of turns of the boost winding NHV is NHV, the alternating current induced in the boost winding NHV is obtained. Voltage V
The voltage level VHV of HV is VHV = V2 × NHV / N2
Indicated by That is, the resonance voltage V2 and the alternating voltage VH
V, and between the secondary winding N2 and the boost winding NVH, V2 /
The relationship of N2 = VHV / NHV holds. In practice, by determining the number of windings of the boost winding NVH based on this relationship, the alternating voltage excited by the boost winding NVH is boosted to a required level.

The multiplying voltage rectifier circuit 2 is connected to the boosting winding NHV. The multiplying voltage rectifying circuit 2 receives the boosted alternating voltage VHV generated in the boosting winding NHV. Then, the multiple voltage rectification operation using the alternating voltage VHV is performed in the multiple voltage rectification circuit 2, so that a predetermined high level (for example, 32 KV) DC high voltage EHV is applied across the smoothing capacitor COHV. can get. This DC high voltage EHV is used as the anode voltage of the CRT.

FIG. 2 shows a Jones & Waters circuit known as a symmetric cascade rectifier circuit as a specific circuit configuration example of the multiple voltage rectifier circuit 2 provided in the switching power supply circuit shown in FIG. It is shown.
As shown in this figure, as the Jones & Waters circuit, first, a first capacitor series consisting of a series connection of high-voltage capacitors CHVA1, CHVA2,. The circuit is connected,
The end on the high-voltage capacitor CHVAn side of this capacitor series circuit is connected to the positive terminal (output terminal of DC high voltage) of the smoothing capacitor COHV via the high-voltage rectifier diode DHVA (n + 1). Further, a second capacitor series circuit composed of a series connection of high-voltage capacitors CHVB1, CHVB2,... CHVBn is connected to the winding end end of the step-up winding NHV, and the end of the high-voltage capacitor CHVBn is high-voltage rectified. It is connected to the positive terminal of the smoothing capacitor COHV via a diode DHVB (n + 1).

High voltage rectifier diodes DHVA0 and DHVA are connected between the ground and the positive terminal of the smoothing capacitor COHV.
.., DHVB2,..., DHVBn, DHVA (n + 1) are connected in series to a first diode series circuit. As shown in the figure, the circuit and the high-voltage capacitors of the second capacitor series circuit are sequentially connected to each other and connected to the connection points. Similarly, high voltage rectifier diodes DHVB0, DHVB1, DHVA2,... Are connected between the secondary side ground and the positive terminal of the smoothing capacitor COHV.
.., a second diode series circuit consisting of a series connection of DHVAn and DHVB (n + 1) is connected, and each connection point of these high-voltage rectifier diodes is connected to the first capacitor series circuit and the second capacitor. The connection points are sequentially connected to the connection points between the high-voltage capacitors of the series circuit.

In such a connection form, [DHVA1, DHV
B1, CHVA1, CHVB1] ... [DHVAn, DHVBn, CHVA
n, CHVBn] are connected to form an entire rectifier circuit. According to the symmetric cascade rectifier circuit having such a configuration, both ends of the smoothing capacitor COHV are 2 (n + 1) times the induced voltage VHV induced in the boost winding NHV (where n is the rectifier circuit). , Which indicates the number of stages of the partial rectifier circuit, and is an integer equal to or greater than 1), thereby obtaining a high-level DC high voltage EHV.

Here, for example, the number of stages of the symmetric cascade rectifier circuit is as follows.
What is necessary is just to determine appropriately according to the relationship between the level of the DC high voltage EHV actually required and the step-up alternating voltage VHV induced in the step-up winding NVH. As an example, assuming that the step-up alternating voltage VHV induced in the step-up winding NHV is about 4 KV and it is necessary to obtain 32 KV as the DC high voltage EHV, the symmetric cascade rectifier circuit shown in FIG. It suffices to form an eight-fold voltage rectifier circuit with three stages.

FIG. 3 shows operation waveforms of main parts of the switching power supply circuit shown in FIG. 1 when the symmetric cascade rectifier circuit shown in FIG.
Shown in Here, FIGS. 3A to 3F show that the AC input voltage VAC = 100 V, and the high voltage load power in the multiple voltage rectification circuit 2 is 64 W (DC high voltage EHV = 32 KV, load current IHV =
FIG. 3 shows the operation waveforms of the respective parts when the current value is 2 mA).
(G) to (l) show that the AC input voltage VAC = 100V,
When the high-voltage load power in the multiplying voltage rectifier circuit 2 is 0 W (DC high voltage EHV = 32 KV, load current IHV = 0 mA), the operation waveforms of the same portions as those in FIGS. It is shown. In this case, the insulation converter transformer P
The number of windings of the IT primary winding N1 is 40T (turn), the number of windings of the boost winding NHV is 1300T, and each of the high voltage capacitors CHVA1, CHVA2,.
000 pF is selected. Also, the secondary side DC output voltage EO1 at this time is controlled at a constant voltage so that EO1 = 135 V, and the output current IOn level at this time is IO = 0.9A.

First, the operation when the high-voltage load power is 64 W will be described with reference to FIGS. When the high-voltage load power is 64 W, the collector-emitter voltage V1 of the switching element Q2 in FIG. 3A is 0 level during the period TON when the switching element Q2 is on and rectangular during the period TOFF when the switching element Q2 is off. A wavy pulse is obtained. As can be seen from the fact that TON / TOFF, which is the actual ON / OFF period of the switching elements Q1 and Q2, is 5 μs, the switching frequency of the switching elements Q1 and Q2 is controlled to be, for example, 100 kHz. You.

The collector current IC2 flowing through the collector of the switching element Q2 is equal to the partial resonance capacitor Cc.
3d, the waveform becomes a substantially sinusoidal waveform as shown by the solid line in FIG. Thus, the current waveform of the collector current IC2 flowing when the switching element Q2 is turned off is, for example, the partial resonance capacitor C2 shown by a broken line in FIG.
The current level becomes smaller than the current level of the collector current IC2 when c is absent, and when the switching elements Q1 and Q2 are turned off, an operation close to ZCS is obtained.

The operation waveform of switching Q1 is shown as a waveform whose phase is shifted by 180 degrees from the operation waveform of switching element Q2 described above. That is,
The switching element Q1 has a waveform that turns on during a period TOFF when the switching element Q2 turns off. Then, a collector current IC1 having a waveform as shown in FIG. 3C flows through the collector due to the resonance action of the partial resonance capacitor Cc. Thus, the switching element Q
By performing a switching operation in which 1 and Q2 alternately turn on / off alternately, a primary side resonance current I1 having a waveform as shown in FIG. 3B flows through the primary winding N1 of the insulating converter transformer PIT.

The switching output is transmitted to the secondary side of the insulated converter transformer PIT by the on / off operation of the switching elements Q1 and Q2 described above. In this case, the secondary winding N2 of the insulated converter transformer PIT is connected to FIG.
As shown in (e), a rectangular wave alternating voltage V2 of 135 Vp is obtained. Also, the boost winding NVH is
As shown in (g), a high alternating voltage VHV of 4 KVp is obtained. The cycles of the alternating voltages V2 and VHV obtained on the secondary side follow the switching frequency on the primary side, as can be seen from comparison with the waveform shown in FIG.

When the high-voltage load power is 0 W, the switching frequency of the switching elements Q 1 and Q 2 is, for example, 1
It is controlled to be about 85 KHz, and the actual ON / OFF period TON / TOFF of the switching elements Q1 and Q2 is 2.7 μs. And this high voltage load power is 0
At the time of W, each waveform when the high-voltage load power described with reference to FIGS. 3A to 3F is 64 W is shown in FIG.
(G) to change as shown in FIG. That is, as the switching frequency increases to 185 KHz, the period is changed so as to shorten the period while maintaining substantially the same waveform.

FIGS. 3A to 3F and FIG. 3G
As can be seen by comparing the operation waveforms shown in (1) to (1), the load is 64 W (EHV = 32 KV, IHV = 2 mA) to 0 W
(EHV = 32 KV, IHV = 0 mA), the switching frequency of the switching elements Q1 and Q2 changes in the control range of 100 kHz to 185 kHz.

Comparing the circuit of the present embodiment having such a configuration shown in FIG. 1 with the conventional circuit shown in FIG. 8, the circuit shown in FIG.
The flyback pulse voltage V11 obtained by the horizontal output circuit 40 is boosted by the secondary DC voltage EO1 of 0 to obtain the DC high voltage EHV from the high voltage generation circuit 50.
On the other hand, in the power supply circuit according to the present embodiment shown in FIG. 1, the alternating voltage VHV output from the secondary side of the insulating converter transformer PIT is input to the multiple voltage rectification circuit 2 so that the DC high voltage EHV I'm trying to get That is, in the circuit of the present embodiment shown in FIG. 1, the alternating voltage EHV obtained from the secondary side of the insulating converter transformer PIT is directly input to the multiple voltage rectification circuit 2, so that the circuit shown in FIG. DC output voltage EO1 of the switching power supply circuit 10 like a circuit
The DC high voltage EHV is obtained by the multiple voltage rectification circuit 2 without the interposition of the horizontal output circuit 40 for converting the voltage to the flyback pulse voltage.

As a result, the conventional circuit shown in FIG. 8 has a power conversion efficiency of about 70% when obtaining the high DC voltage EHV from the input voltage, whereas the conventional circuit shown in FIG. In the power supply circuit of the embodiment, the power conversion efficiency is 90.5%
It is possible to improve up to. Accordingly, the input power can be reduced by about 18.5 W.

When the power supply circuit according to the present embodiment is applied to a television receiver or the like, for example, when the high-voltage load power is 0 W (EHV = 32 KV, IHV = 0 mA), the voltage level of the DC high voltage EHV is reduced. If it is 32KV, even if the high-voltage load power increases to 64W (EHV = 32KV, IHV = 2mA), the fluctuation width ΔEHV of the voltage level of the DC high voltage EHV output from the circuit remains at about 0.9KV. was gotten. On the other hand, in the circuit shown in FIG.
High-voltage load power is no-load (IHV = 0mA) to 64W (IHV
= 2mA), the fluctuation width Δ of the DC high voltage EHV
The EHV was 1.5 KV. Therefore, if the circuit according to the present embodiment shown in FIG. 1 is applied to, for example, a television receiver, and a high DC voltage EHV is supplied to the anode electrode of the CRT, the C
Since the horizontal amplitude fluctuation of the electron beam output from the RT can be suppressed, it is not necessary to provide a zooming correction circuit or the like in the horizontal output circuit of the television receiver.

Further, in the circuit of the present embodiment shown in FIG. 1, by winding the boost winding NVH on the secondary side of the insulating converter transformer PIT, the DC high voltage EHV is supplied from the boost winding NHV. An alternating voltage VHV for obtaining the same is obtained. As described above with reference to FIG. 4, the structure of the insulating converter transformer PIT according to the present embodiment is different from that of the divided bobbin LB1 for the EE-type core CR in which the gap G is formed in the center magnetic leg. Is used to wind the primary winding N1 and the secondary winding N2, and the high-voltage winding bobbin HB1
Is used to wind the step-up winding NVH. In this case, the gap G formed in the center magnetic leg of the EE-type core CR is covered by the winding, so that the leakage magnetic flux from the gap G does not leak to the outside. As a result, the step-up winding NVH is caused by leakage magnetic flux and leakage inductance from the insulating converter transformer PIT.
Ringing does not occur in the induced voltage.

Therefore, even when the circuit of the present embodiment is applied to a television receiver, for example, raster ringing does not occur on the screen of a CRT, and even if ringing does occur, the present embodiment does not. In the circuit,
Since the multiplying voltage rectifier circuit 2 is formed independently of the horizontal deflection circuit, no ringing current component is superimposed on the horizontal deflection current IDY, so that raster ringing does not occur on the screen of the CRT.

The switching frequency of the insulating converter transformer PIT corresponds to the switching frequency of the switching element Q1, and is asynchronous with, for example, the frequency of the horizontal synchronizing signal fH of the video signal. In this case, if the level of the ripple voltage δV superimposed on the DC high voltage EHV is large, a beat is generated on the screen due to the interference between the horizontal synchronization signal fH and the switching frequency of the switching converter. If a cascade rectifier circuit is provided as a multiple voltage rectifier circuit 2,
The ripple voltage δV superimposed on the DC high voltage EHV is relatively small, and the beat caused by interference between the horizontal synchronization signal fH and the switching frequency fs is inconspicuous.

Since the DC component is not superimposed on the primary current flowing through the primary winding N1, which is the primary side of the boost winding NHV, the core of the insulating converter transformer PIT can be reduced in size. Since it is not necessary to increase the diameter of the primary winding N1, the size of the primary winding N1 can be reduced. Further, since the peak current value of the current flowing through the switching element Q1 is also reduced, the heat generation of the switching element Q1 is suppressed, and it is not necessary to take measures such as attaching a heat sink to the switching element Q1.

Here, in the present invention, the multiple voltage rectification circuit 2 provided in the switching power supply circuit shown in FIG.
Alternatively, a configuration other than the symmetric cascade rectifier circuit shown in FIG. 2 may be adopted. Hereinafter, the multiple voltage rectification circuit 2
Another configuration example will be described with reference to FIGS.

FIG. 5 shows a Cockcroft & Walton circuit known as a basic cascade rectifier circuit as another example of the circuit configuration of the multiple voltage rectifier circuit 2. In this case, the winding end end of the boost winding NVH (secondary ground)
Are the high voltage capacitors CHVA1, CHVA2, CHVA3, ... C
A first capacitor series circuit composed of a series connection of HVAn is connected. The winding start end is connected to a second capacitor series circuit composed of a series connection of high voltage capacitors CHVB1, CHVB2, CHVB3,... CHVBn. The winding end end of the boost winding NVH and the smoothing capacitor C
High-voltage rectifier diodes DHVB1, DHVA1, DHVB2, DHVA2, D
HVB3, DHVA3,..., DHVBn and DHVAn are connected in series with a diode series circuit.

Then, as shown, each of the high-voltage capacitors CHVA1, CHVA forming the first capacitor series circuit
2, CHVA3,..., CHVAn include series circuits [DHVB1-DHVA1], [DHVB2] composed of two sets of high-voltage rectifier diodes among the high-voltage rectifier diodes forming a diode series circuit.
-DHVA2], [DHVB3-DHVA3] ... [DHVBn-DHV
An] are connected in parallel. On the other hand, the high-voltage capacitor CHVB1 forming the second capacitor series circuit is
Start winding end of boost winding NVH and high voltage rectifier diode DHV
The high voltage capacitors CHVB2, CHVB3... CHVBn, which are inserted between the cathodes of B1 and B1, are connected in series with two sets of high voltage rectifier diodes [DHVA1-DHVB2], [DHVA2 −DHVB3], ・
[DHVA (n-1) -DHVBn] are connected in parallel.
In such a connection form, [DHVA1, DHVB1, CHVA1,
CHVB1]... [DHVAn, DHVBn, CHVAn, CHVBn] are connected to an n-stage partial rectifier circuit to form the entire rectifier circuit.

The rectifying operation of such a basic cascade rectifier circuit is as follows. First, during a period in which a negative current flows through the boost winding NHV, the high-voltage rectifier diode DHVB1 is turned on, and The charging operation for the high-voltage capacitor CHVB1 is obtained by the rectified current. Next, during a period in which a positive current flows through the boost winding NHV, the high-voltage rectifier diode D is generated by the induced voltage VHV induced in the boost winding NHV and the voltage across the high-voltage capacitor CHVB1.
HVA1 is turned on, and a charging operation for the high-voltage capacitor CHVA1 is obtained by the rectified current rectified by the high-voltage rectifier diode DHVA1. Next, during a period in which a negative current flows through the boost winding NHV, the induced voltage V
The high voltage rectifier diode DHVB2 is turned on by the voltage across HV and the high voltage capacitor CHVA1, and a charging operation for the high voltage capacitor CHVB2 is obtained.

Thereafter, the high-voltage capacitors CHVA2, CHVA3,..., CHVAn,
And the high voltage capacitors CHVB3... CHVBn forming the second capacitor series circuit are charged. Then, each of the high-voltage capacitors CHVA1 to CHVAn and CHVB1 to CHVBn, which are charged in this manner,
Is charged to the smoothing capacitor COHV by each of the potentials, a DC high voltage EHV corresponding to 2n times the induced voltage VHV is obtained at both ends of the smoothing capacitor COHV.

FIG. 6 shows a Mitchell circuit known as a modified cascade rectification circuit as another example of the circuit configuration of the multiple voltage rectification circuit 2. The connection form of the circuit shown in this figure is substantially the same as that of the basic cascade rectifier circuit shown in FIG.
Is connected to the midpoint of a second capacitor series circuit composed of a series connection of high voltage capacitors CHVB1 to CHVBn. The winding end of the boost winding NHV is connected to the middle point of the first capacitor series circuit composed of the series connection of the high-voltage capacitors CHVA1 to CHVAn and the high-voltage rectifier diode DHV.
B1, DHVA1, DHVB2, DHVA2,..., DHVBn, and DHVAn are respectively connected to the midpoints of diode series circuits formed in series. Again, [DHVA1, DHV
B1, CHVA1, CHVB1] ··· [DHVAn, DHVBn, CHVA
n, CHVBn] are connected to form an entire rectifier circuit. As a result, as a Mitchell circuit configured by such a connection form,
A DC high voltage EHV corresponding to 2n times the induced voltage VHV is obtained across the smoothing capacitor COHV.

As described above, in the switching power supply circuit of the present embodiment shown in FIG. 1, the symmetric cascade rectifier circuit shown in FIG. 2 or the basic cascade rectifier circuit shown in FIGS. In addition, the modified cascade rectification circuit can be provided as the multiple voltage rectification circuit 2. However, in the case where the rectifier circuit is formed by a multi-stage rectifier circuit including a large number of high-voltage capacitors and high-voltage rectifier diodes, the component of the ripple voltage δV is superimposed on the DC high voltage EHV output from the multiple voltage rectifier circuit 2. At the same time, a voltage drop ΔEHV (regulation: voltage fluctuation) occurs. For example, ripple voltage δ superimposed on DC high voltage EHV
From the viewpoint of V, the symmetric cascade rectifier circuit shown in FIG. Also,
From the viewpoint of the voltage drop ΔEHV generated in the multiple voltage rectifier circuit 2, the modified cascade rectifier circuit shown in FIG. In addition,
When a switching power supply circuit is configured by using the modified cascade rectifier circuit shown in FIG. 6 as the multiple voltage rectifier circuit 2, a high voltage of about 1/2 of the DC high voltage EHV is applied to the boost winding NHV. It is preferable to improve the insulation by molding the step-up winding NVH with an epoxy resin or the like.

Further, the circuit configuration of the power supply circuit of the present invention is not limited to the circuit configuration shown in FIG. FIG. 7 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG. This power supply circuit is also provided with a current resonance type converter in which two switching elements are half-bridge-coupled, but the drive system is separately excited. Therefore, the circuit shown in FIG. 7 includes the self-excited oscillation driving circuit [NB1 shown in FIG.
-RB1-CB1, NB2-RB2-CB2], a separately excited oscillation / drive circuit 3 is provided. Further, MOS-FETs are used as the switching elements Q21 and Q22. The gates of the switching elements Q21 and Q22 are connected to the oscillation / drive circuit 3. The drain of the switching element Q21 is connected to the positive electrode of the smoothing capacitor Ci, and the source is connected to the primary side ground via the series resonance capacitor C1 and the primary winding N1. The drain of the switching element Q22 is connected to the source of the switching element Q21, and the source is connected to the primary side ground. Here, the partial resonance capacitor Cc is connected between the contact (switching output point) between the source of the switching element Q21 and the drain of the switching element Q22 (switching output point) and the primary side ground. Therefore, the partial resonance operation by the partial resonance capacitor Cc can be obtained here. Further, a clamp diode D is provided between the drain and source of each of the switching elements Q21 and Q22.
D1 and DD2 are connected in parallel.

The switching elements Q21 and Q22 are switched by the oscillation / drive circuit 3 so as to obtain the same switching operation as described above with reference to FIG. That is, in the control circuit 1, the DC output voltage EO1
Oscillates a current or voltage at a level that fluctuates in accordance with
It is supplied to the drive circuit 3. In the oscillation / drive circuit 3, the stabilization of the DC output voltage EO1 is achieved.
A switching drive signal (voltage) whose cycle is varied according to the output level from the control circuit 1 is output to the gates of the switching elements Q21 and Q22. As a result, the switching frequency of the switching elements Q21 and Q22 is varied.

In this case, the starting circuit 4 is supplied with the rectified smoothed voltage Ei obtained from the smoothing capacitor Ci as the operating power. Further, the starting circuit 4 has a voltage obtained by converting the alternating voltage obtained at the time of starting the winding N3 additionally wound around the insulating converter transformer PIT into a direct current by a half-wave rectifying circuit including a diode D3 and a capacitor C3. Although supplied, the starting circuit 4 executes an operation for starting the oscillation / drive circuit 3 in response to the input of the DC voltage.

On the secondary side of the insulating converter transformer PIT in this case, the secondary winding N2 is provided with a center tap grounded to the secondary side ground, and the secondary winding N2 Are wound up by the required number of turns to form a set of boost windings NVV.

A full-wave rectifier circuit composed of rectifier diodes DO1 and DO2 and a smoothing capacitor CO1 is connected to the secondary winding N2 as shown in FIG.
The secondary side DC output voltage EO1 is obtained.

The alternating voltage obtained in the boost winding NHV formed by winding up the secondary winding N2 is supplied to the multiple voltage rectification circuit 2 as shown in the figure. Things. It should be noted that any one of the circuit configurations shown in FIGS. 2, 5, and 6 may be employed as the multiple voltage rectifier circuit 2 shown in FIG. With such a configuration, the same operation and effect as those of the power supply circuit shown in FIG. 1 can be obtained.

In this embodiment, an orthogonal control transformer is used for performing constant voltage control under a configuration having a self-excited resonance converter on the primary side. Alternatively, the oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, for example, as in the case of the orthogonal control transformer, a three-dimensional structure is obtained by combining two sets of double U-shaped cores having four magnetic legs. Form a mold core. The control winding NC and the driving winding NB are applied to the three-dimensional core.
At this time, the winding direction of the control winding and the driving winding is obliquely crossed. Specifically, one of the control winding NC and the driving winding NB is wound around two magnetic legs in a positional relationship adjacent to each other among the four magnetic legs, The other winding is wound around two magnetic legs which are considered to be in a diagonal positional relationship. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained.
Thereby, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

Further, the insulation converter transformer PIT
Although a rectifier circuit of a full-wave rectification method is provided as an example of a rectifier circuit for obtaining a secondary output voltage EO1 on the secondary side of the present invention, the rectifier circuit is limited to such a rectifier circuit. Insulated converter transformer P according to the present invention
Various configurations are possible for the secondary rectifier circuit of the IT.

Further, in the power supply circuit of the present embodiment, the case where the primary side is provided with the primary-side composite resonance type switching converter has been described as an example. However, the circuit configuration of the primary side of the present invention is not necessarily required. It is not necessary to adopt the configuration of the complex resonance type switching converter. For example, the primary side circuit configuration may be a current resonance type switching converter.

[0086]

As described above, the present invention comprises a current resonance type converter and a composite current resonance type converter provided on the primary side with a partial resonance circuit for performing the so-called partial resonance operation of the current resonance type converter. As a switching power supply circuit, a second secondary winding (boost winding) is wound around a secondary side of an insulating converter transformer constituting the switching power supply circuit, and the second secondary winding is wound around the second secondary winding. The alternating voltage obtained as described above is input to the DC high voltage generating means. Then, a DC high voltage which is set to a predetermined high voltage level by the DC high voltage generating means is obtained. Therefore,
If the switching power supply circuit of the present invention is applied to a television receiver, for example, when obtaining a high DC voltage to be supplied to the anode of a cathode ray tube, the secondary side DC output voltage is converted to a flyback pulse voltage in a horizontal deflection circuit. There is no need to convert the horizontal deflection circuit, and the configuration can be such that the horizontal deflection circuit is omitted. As a result, the power conversion efficiency when a DC high voltage is obtained from the input voltage can be improved.

Further, according to the present invention, even when the high-voltage load fluctuates, the DC high voltage output by the DC high voltage generating means can have a smaller range of voltage fluctuation than in the prior art. Therefore, if the present invention is applied to, for example, a high voltage supply unit of a television receiver, it becomes possible to suppress, for example, horizontal amplitude fluctuations of an electron beam output from a cathode ray tube.

Further, since the first secondary winding (secondary winding) and the second secondary winding (step-up winding) are wound around the secondary side of the insulating converter transformer, the conventional method is used. It is not necessary to provide a high-voltage transformer for obtaining a high DC voltage as in (1), and the peak value of the current flowing through the switching element is reduced, and the amount of heat generated by the switching element is reduced. It is not necessary to take measures such as mounting a plate.

Further, the high voltage stabilizing circuit of the present invention can be constituted by a multiple voltage rectifying circuit employing a Jones & Waters circuit, a Cockcroft & Walton circuit or a Mitchell circuit as a DC high voltage generating means. Is done. Thus, even if the induced voltage induced in the second secondary winding of the insulating converter transformer is not generated at a sufficiently high level, the DC high voltage generating means can reduce the anode voltage of the CRT, for example. It is possible to obtain a DC high voltage of a level sufficient for practical use in accordance with a required purpose such as obtaining. In addition, if the Jones & Waters circuit is applied to the DC high voltage generating means, the ripple voltage superimposed on the DC high voltage can be reduced, and if the Mitchell circuit is applied, the DC high voltage generating means can be reduced. Since it is possible to effectively suppress the voltage drop, it is also possible to realize an optimum configuration for each device to which the switching power supply circuit of the present invention is applied.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a power supply circuit according to an embodiment of the present invention.

FIG. 2 is a circuit diagram showing one configuration example of a multiple voltage rectifier circuit as the present embodiment.

FIG. 3 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 4 is an insulation converter transformer PIT according to the present embodiment.
FIG. 3 is a cross-sectional view showing the structure of FIG.

FIG. 5 is a circuit diagram showing another configuration example of the multiple voltage rectifier circuit.

FIG. 6 is a circuit diagram showing another configuration example of the multiple voltage rectifier circuit.

FIG. 7 is a circuit diagram illustrating a configuration example of a power supply circuit according to another embodiment.

FIG. 8 is a circuit diagram showing a configuration of a conventional high voltage generating circuit and its peripheral circuits.

9 is a waveform chart showing an operation of a main part of the circuit shown in FIG.

10 is an explanatory diagram conceptually showing an operation of a flyback transformer provided in the power supply circuit shown in FIG.

11 is an explanatory diagram showing a relationship between a DC high voltage EHV and a high-voltage load current IHV as characteristics of the power supply circuit shown in FIG. 8;

FIG. 12 is a cross-sectional view showing a structure of a flyback transformer provided in the power supply circuit shown in FIG.

[Explanation of symbols]

1 control circuit, 2 multiple voltage rectification circuit, Ci smoothing capacitor, Q1, Q2, Q11, Q12 switching element, PI
T isolation converter transformer, PRT drive transformer, C1 primary side series resonance capacitor, N1 primary winding,
N2 secondary winding, NHV boost winding, NC control winding, NB
Drive winding, ND resonance current detection winding, CB resonance capacitor, DBR bridge rectifier circuit, DO1, DO2 rectifier diode, DHVA0 to DHVA (n + 1) DHVB0 to DHVB (n + 1) High voltage rectifier diode, CHVA1 to CHVAn CHVB1 ~ CHVBn
High voltage capacitor, CO1, COHV smoothing capacitor

Claims (4)

[Claims] A switching device for intermittently outputting an input DC input voltage; a switching device provided for transmitting an output of a primary side to a secondary side; a primary winding; A primary winding and a second secondary winding are wound, and a required degree of loose coupling is obtained for the primary winding and the first secondary winding.
The first secondary winding and the second secondary winding are insulated converter transformers capable of obtaining a tightly coupled state, and are inserted so that the operation of the switching means is a current resonance type. A primary-side series resonance circuit, a primary-side partial resonance circuit formed so as to obtain a partial resonance operation at the time of the switching operation of the switching means, and an alternating voltage obtained in the first secondary winding. DC output voltage generating means configured to obtain a secondary side DC output voltage by performing a rectification operation, and changing a switching frequency of the switching element according to a level of the secondary side DC output voltage. And a constant voltage control unit configured to perform constant voltage control by inputting an alternating voltage obtained to the second secondary winding and performing a rectification operation to achieve a predetermined high voltage level. Switching power supply circuit, characterized in that and a high DC voltage generating means arranged to obtain a high DC voltage. 2. The switching power supply circuit according to claim 1, wherein said DC high voltage generation means includes a Jones and Waters circuit. 3. The switching power supply circuit according to claim 1, wherein said DC high-voltage generating means includes a Cockcroft and Walton circuit. 4. The switching power supply circuit according to claim 1, wherein said DC high voltage generation means includes a Mitchell circuit.