JP2001178125A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a switching power supply circuit capable of outputting a stable DC high voltage. SOLUTION: A primary winding N4 of a step-up transformer HVT is connected in parallel with a secondary winding N2 of an isolation converter transformer PIT, for constituting the switching power supply circuit. Thus, a DC high voltage EHV is obtained, based on a resonance voltage V3 output from the secondary side of the transformer PIT in a high voltage generator 4. In this case, since the DC high voltage EHV can be obtained from the resonance voltage V3 output from the transformer PIT, the power conversion efficiency when the voltage EHV is obtained from the primary side DC voltage Ei is improved.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices, and more particularly to a switching power supply circuit for outputting a stable high voltage.

[0002]

2. Description of the Related Art Conventionally, some electronic devices such as a television receiver and a projector device have a cathode ray tube (CRT) for displaying images. In a television receiver equipped with a cathode ray tube (hereinafter referred to as "CRT"), as is well known, an electron beam output from an electron gun provided inside a CRT is horizontally (horizontally) directed. And a vertical deflection system block for deflecting in the vertical direction (vertical direction). Further, the horizontal deflection circuit block is provided with a high voltage generation circuit for supplying a high voltage of, for example, about 20 kV to 35 kV to the anode electrode of the CRT.

FIG. 11 is a diagram showing the configuration of a horizontal deflection circuit system block provided in a television receiver and its peripheral circuits. The switching power supply circuit 10 shown in this figure is a DC-DC converter that performs switching on an input DC voltage, and finally converts the DC voltage into a DC voltage of a predetermined voltage level and outputs the DC voltage. A rectifying and smoothing circuit including a full-wave rectifying bridge rectifying circuit Di and a smoothing capacitor Ci is provided at a stage preceding the switching power supply circuit 10. The rectifying and smoothing circuit rectifies and smoothes a commercial AC power supply (AC input voltage VAC). To obtain a DC voltage Ei. The DC voltage Ei is input to the switching power supply circuit 10. And in this case,
The switching power supply circuit 10 outputs a DC output voltage EO (EO1, EO2, EO3) converted to a predetermined voltage level. Each of the above DC output voltages EO1,
The actual voltage levels of EO2 and EO3 are, for example, DC output voltage EO1 = 135V, DC output voltage EO2 = 15V, and DC output voltage EO3 = 7V.

The horizontal oscillation circuit 20 receives a horizontal synchronizing signal fH included in a video signal or the like. The oscillation frequency (15.75 k) corresponding to the horizontal synchronization signal fH
Hz), and outputs a pulse voltage synchronized with the horizontal synchronization signal fH.

[0005] A horizontal drive circuit 30 surrounded by a dashed line amplifies a pulse voltage from the horizontal oscillation circuit 20,
A sufficiently large drive current (drive current) is supplied to a horizontal output circuit 40 described later. In this case, the configuration of the horizontal drive circuit 30 includes a horizontal output circuit 40 serving as a load.
In order to prevent the period of the pulse voltage supplied from the horizontal oscillation circuit 20 from fluctuating, a transformer-coupled amplifier circuit having a common emitter is usually used.

In the horizontal drive circuit 30, as shown, for example, the base of a transistor Q11 is connected to the horizontal oscillation circuit 20 via a capacitor C11, and a pulse voltage from the horizontal oscillation circuit 20 is applied to the base of the transistor Q11. Has been entered. A bias resistor R11 is inserted between the base and the emitter, and a predetermined bias voltage is applied to the base. The collector of the switching element Q11 is connected to the secondary output terminal (DC output voltage EO) of the switching power supply circuit 10 via the primary winding N11 of the horizontal drive transformer HDT and the collector resistor R13.
1) and its emitter is grounded. Further, a damping circuit comprising a series connection circuit of a capacitor C12 and a resistor R12 is provided between the collector and the emitter. The damping circuit composed of the series connection of the capacitor C12 and the resistor R12 prevents the surge current and the oscillating current (ringing current) from being superimposed on the current flowing through the primary winding N11 of the horizontal drive transformer HDT. Preventing. The horizontal drive transformer H
The capacitor C13 provided between the winding start end of the primary winding N11 of the DT and the secondary side ground is a capacitor for removing noise.

[0007] The horizontal drive transformer HDT is an insulating transformer for transmitting the output of the primary winding N11 to the secondary winding N21. In this case, the winding (winding direction) of the primary winding N11 and the secondary winding N21 of the horizontal drive transformer HDT is reversed. The winding start end of the primary winding N11 is connected to the DC output terminal (DC output voltage EO1) of the switching power supply circuit 10 via the collector resistor R13, and the winding end is connected to the collector of the transistor Q11. Have been. The winding end of the secondary winding N21 is connected to an output transistor Q of a horizontal output circuit 40 described later.
It is connected to 12 bases, and its winding start end is grounded to earth.

A horizontal output circuit 40 surrounded by a dashed line.
Amplifies the output obtained from the secondary side of the horizontal drive transformer HDT, thereby scanning the electron beam output from the electron gun of the CRT in the horizontal direction.
Generate. At the same time, a high-voltage generation circuit 50 described later
Is configured to generate a flyback pulse for generating a high voltage.

In the horizontal output circuit 40, the base of the output transistor Q12 is connected to the horizontal drive transformer HDT.
Of the switching power supply circuit 10 via a primary low-voltage winding NLV of a flyback transformer FBT to be described later (secondary output voltage (secondary output voltage)). EO1). The emitter is grounded. A damper diode D11 and a horizontal retrace capacitor Cr1 are connected in parallel between the collector and the emitter of the output transistor Q12. Further, a series connection circuit including [horizontal deflection yoke HDY, horizontal linear correction coil HLC, and S-shaped correction capacitor CS1] is connected between the collector and the emitter.

The horizontal output circuit 40 having such a configuration
Then, the capacitance of the horizontal retrace capacitor Cr1 and
The leakage inductance component L LV of the primary low voltage winding N LV of the flyback transformer FBT forms a voltage resonance type converter. The output transistor Q12 is output by the secondary output of the horizontal drive transformer HDT.
Are turned on / off, a horizontal deflection current IDY having a sawtooth waveform flows through the horizontal deflection yoke HDY. Further, during the period when the output transistor Q12 is off, the horizontal retrace capacitor C is set due to the resonance operation of the inductance LDY of the horizontal deflection yoke HDY and the capacitance of the horizontal retrace capacitor Cr1 and the action of the damper diode D11.
At both ends of r1, a pulse voltage (flyback pulse voltage) V1, which is a relatively high voltage, is generated. Although the operations of the horizontal straight line correction coil HLC and the S-shaped correction capacitor CS1 are omitted, for example, the operation is performed so as to correct the horizontal deflection current IDY to correct the distortion of the image displayed on the screen of the CRT. I have.

A high-voltage generating circuit 50 surrounded by a dashed line.
Is a flyback transformer FBT (Fly Back Tra
nsformer) and a high-voltage rectifier circuit, and boosts the flyback pulse voltage V1 generated by the horizontal output circuit 40 to generate a high voltage corresponding to, for example, an anode voltage level of a CRT.

FIG. 15 is a sectional view of the flyback transformer FBT. The structure of the flyback transformer FBT will be described with reference to FIG. In the flyback transformer FBT shown in this figure, a square core 30 is formed by combining the magnetic legs of two U-shaped cores CR10 and CR20 so as to face each other. Then, a gap G is provided at a portion where the end of the U-shaped core CR10 and the end of the U-shaped core CR20 face each other. As shown in the figure, the low-voltage winding bobbin LB and the high-voltage winding bobbin H
By attaching B, the primary low-voltage winding NLV and the secondary high-voltage winding NHV are separately wound around these low-voltage winding bobbin LB and high-voltage winding bobbin HB. In this case, the low voltage winding bobbin LB is wound with the primary side low voltage winding NLV using a single wire, and the high voltage winding bobbin H
A secondary high-voltage winding NVH is wound around B using a single wire. At this time, for example, a plurality of secondary high-voltage windings NHV need to be wound on the high-voltage winding bobbin HB in an insulated state. Insert the interlayer film F between the high-voltage windings NHV and wind up.
It is a so-called interlayer winding.

As a practical example of the flyback transformer FBT having the structure shown in FIG.
As HV, five sets of secondary high voltage windings NHV1, NHV2, NHV3,
NHV4 and NHV5 are divided and wound independently. It should be noted that only one winding is wound as the primary low-voltage winding NLV. Here, the polarity (winding direction) of each of the secondary high voltage windings NHV1 to NVV5 with respect to the primary low voltage winding NLV.
Are wound so as to have opposite polarities. The winding start end of the primary low-voltage winding NLV is connected to the secondary output terminal (DC output voltage EO1) of the switching power supply circuit 10, and the winding end is connected to the collector of the output transistor Q12. . The anodes of the high-voltage rectifier diodes DHV1, DHV2, DHV3, DHV4, and DHV5 are connected to the winding end ends of the secondary high-voltage windings NHV1 to NHV5. And the high voltage rectifier diode DHV
The cathode of 1 is a smoothing capacitor CHVO1 via a resistor RHV.
And the high-voltage rectifier diode DHV
The respective cathodes of DHV1, DHV2, DHV3, DHV4, and DHV5 are respectively connected to the winding start ends of the secondary high-voltage windings NHV1, NHV2, NHV3, and NHV4.

In such a connection form, [secondary high voltage winding NHV1, high voltage rectifier diode DHV1], [secondary high voltage winding NHV2, high voltage rectifier diode DHV2], [secondary high voltage winding NHV3, high voltage Five sets of half-wave rectifier circuits of rectifier diode DHV3, [secondary high voltage winding NHV4, high voltage rectifier diode DHV4] and [secondary high voltage winding NVH5, high voltage rectifier diode DHV5] are formed. This means that a set of half-wave rectifier circuits are connected in series.

Therefore, on the secondary side of the flyback transformer FBT, five sets of half-wave rectifier circuits rectify the voltage induced in the secondary high voltage windings NHV1 to NHV5 and charge the smoothing capacitor CHVO1. By performing such an operation, both secondary high-voltage windings NHV are connected to both ends of the smoothing capacitor CHVO1.
A DC voltage at a level corresponding to five times the voltage induced at 1 to NHV5 is obtained. That is, a five-fold voltage half-wave rectifier circuit is formed. The DC voltage obtained between both ends of the smoothing capacitor CHVO1 is converted into a DC high voltage EHV, and is used, for example, as an anode voltage of a CRT.

The resistor R1, R1 is inserted between the cathode of the high voltage rectifier diode DHV3 and the secondary side ground.
A series connection circuit composed of a variable resistor R2 and a resistor R3] is provided to obtain a voltage level lower than the DC high voltage EHV, and outputs a DC output voltage EFV used as a focus voltage of a CRT, for example.

FIG. 12 shows the operation waveform of each part of the circuit shown in FIG. In the circuit shown in FIG. 11, since the pulse voltage is input to the base of the output transistor Q12 from the horizontal oscillation circuit 20 amplified by the horizontal drive circuit 30, the switching frequency of the output transistor Q12 is changed to the horizontal synchronization signal fH Synchronization frequency (15.75 kHz
z). For example, as shown, the ON period (horizontal scanning period) TON of the output transistor Q12 is 52.7 μs, and the OFF period (horizontal retrace period) TOFF is 1
This period is 0.8 μs, and a period of one cycle (63.5 μS) including the period TON and the period TOFF corresponds to the period of the horizontal synchronization signal fH.

In this case, the collector current IC having a waveform as shown in FIG. 12D is applied to the collector of the output transistor Q12 by the on / off operation of the switching element Q12.
Flows. As a result, a primary current I1 having a waveform as shown in FIG. 12C flows through the primary low-voltage winding NLV of the flyback transformer FBT, and a horizontal deflection yoke HDY shown in FIG.
A horizontal deflection current IDY having a waveform as shown in FIG.

At this time, the voltage V1 across the horizontal retrace capacitor Cr1 connected in parallel between the collector and the emitter of the output transistor Q12 is turned on when the output transistor Q12 is turned on, as shown in FIG. During the period TON, the level becomes 0. In the period TOFF during which the output transistor Q12 is turned off, for example, 1000 Vp to 12 V, due to the resonance operation of the inductance component LDY of the horizontal deflection yoke HDY and the capacitance of the horizontal retrace capacitor Cr1.
A flyback pulse voltage V1 of about 00 Vp is generated.

The high voltage generation circuit 50 shown in FIG.
In the above, the positive pulse voltage applied to the primary side of the flyback transformer FBT is boosted by the flyback pulse voltage V1 as described above, and a predetermined high DC voltage EHV is obtained from the secondary side. For example, when a flyback pulse voltage V1 of 1000 Vp to 1200 Vp is generated between both ends of the horizontal retrace capacitor Cr1, as shown in FIG. 13, a positive voltage of about 900 Vp is applied to the primary low voltage winding NLV of the flyback transformer FBT. A pulse voltage is applied. As a result, the secondary high-voltage windings NHV1 to NHV5 have:
An induced voltage is generated by boosting the positive pulse voltage to about 6.5 kV. The high voltage generation circuit 50 is wound with five sets of secondary high voltage windings NHV1 to NHV5 and provided with a five-fold voltage half-wave rectifier circuit. EHV will be output.

Incidentally, such a flyback transformer F
BT primary low voltage winding NLV and secondary high voltage winding NHV1 ~
The number of turns of the NHV5 is, for example, each of the secondary high voltage windings NHV1 to NHV.
5, after winding a winding of about 500T (turn) around the high-voltage winding bobbin HB, the primary-side low-voltage winding NLV is applied to the low-voltage winding bobbin LB for a predetermined turn so that a predetermined DC high voltage EHV is obtained. It is configured by winding.

[0022]

The circuit shown in FIG. 11 uses a flyback pulse voltage V1 obtained by the horizontal output circuit 40 to obtain a DC high voltage EHV from a high voltage generation circuit 50. I have. Therefore, the power conversion efficiency when converting the input power into the high-voltage load power is about 70%.
And the reactive power when obtaining high-voltage load power is relatively large.

In the high voltage generating circuit 50, a positive pulse voltage (flyback pulse voltage) input to the primary low voltage winding NLV of the flyback transformer FBT is induced in each secondary high voltage winding NHV. An induced voltage is obtained, and the peak value of the induced voltage is half-wave rectified by each high-voltage rectifier diode DHV to obtain a DC high voltage EHV. However, in this case, the high-voltage rectifier diode D
Since the conduction angle of the HV is narrow and the power source impedance is equivalently increased, the voltage level of the DC high voltage EHV has a disadvantage that it is easily affected by fluctuations of the high-voltage load.

For example, when the circuit shown in FIG. 11 is applied to a television receiver having a CRT screen size of 29 inches or more, the high-voltage generation circuit 50 outputs an anode electrode from the high-voltage generation circuit 50 to secure the CRT screen brightness. Must be supplied with a beam current IHV of 2 mA or more. That is, C
DC high voltage EHV supplied to the anode electrode of RT
Is 32 kV, for example, the high voltage load power from the high voltage generation circuit 50 is 64 W (32 kV × 2).
mA). Therefore, the high voltage generation circuit 50 outputs at least 0 W (IHV = 0 m) as the high voltage load power.
A) It is conceivable that it fluctuates up to 64 W (IHV = 2 mA).

As an example, the high voltage load power is set to 0 W (IHV = 0
mA) to 64 W (IHV = 2 mA),
FIG. 14 shows how the DC high voltage EHV output from the high voltage generation circuit 50 changes. In this case, the high-voltage load power is 0 W
When (IHV = 0 mA), the voltage level of the DC high voltage EHV is 32 kV. On the other hand, when the high-voltage load power increases to 64 W (IHV = 2 mA), the DC high voltage drops to about 30.5 kV due to the voltage drop due to the high-voltage rectifier diode DHV and the inrush current limiting resistor RHV. That is, when the circuit shown in FIG. 11 is applied to an actual television receiver or the like, the voltage level width ΔEHV of the DC high voltage EHV in the actual use range (0 W to 64 W) of the high voltage load power is about 1.5 kV. become.

When the voltage level of the DC high voltage EHV fluctuates in this way, for example, when the current value of the horizontal deflection current IDY is constant, the horizontal amplitude of the electron beam output from the CRT changes. For this reason, in an actual television receiver, a zooming correction circuit or the like for correcting the current value of the horizontal deflection current IDY is provided in a horizontal output circuit so that the horizontal amplitude of the electron beam does not change due to the fluctuation of the DC high voltage EHV. 40 had to be provided.

As described above, since the flyback transformer FBT is wound on only one of the magnetic legs, the flyback transformer FBT receives the signal from the gap G of the other magnetic leg that is not wound. Ringing (vibration) due to the leakage flux and the distributed capacitance of the leakage inductance of the secondary high-voltage winding NVH
May occur. For example, if ringing (vibration) occurs at the timing when the induced voltage induced in the secondary high-voltage winding NHV becomes a negative level due to the leakage inductance of the secondary high-voltage winding NVH as shown in FIG.
The ringing component is superimposed on the primary current I1 flowing through the primary side of the flyback transformer FBT shown in FIG.
As a result, the ringing current component is also superimposed on the horizontal deflection current IDY shown in FIG.
Raster ringing occurs at the left edge of the screen. For this reason,
In an actual television receiver, some measures are required to prevent raster ringing.

In the flyback transformer FBT, since a DC component is superimposed on the primary current (flyback current) I1 flowing through the primary low voltage winding NLV, the core shape is set so that the flyback transformer FBT is not saturated. And the primary low-voltage winding NL through which the primary current I1 flows
It is necessary to increase the winding diameter of V. As a result, there is also a disadvantage that the shape of the flyback transformer FBT becomes large.

When the DC current component is superimposed on the primary current I1 flowing through the primary side of the flyback transformer FBT, the peak level of the collector current IC flowing through the output transistor Q12 increases accordingly. For this reason, it is necessary to take measures such as forming the output transistor Q12 by a high withstand voltage transistor that can withstand a large amount of power or attaching a heat sink or the like for suppressing heat generation of the output transistor Q12.

[0030]

Therefore, a switching power supply circuit according to the present invention is configured as follows in consideration of the above-mentioned problems. In other words, the switching means includes a switching element and outputs the input DC input voltage intermittently, an insulating converter transformer that transmits the output of the switching means to the secondary side, and the operation of the switching means is a voltage resonance type. A primary-side voltage resonance circuit, a secondary-side series resonance circuit formed by connecting a secondary-side series resonance capacitor to the secondary winding of the insulation converter transformer in series, and an insulation converter transformer DC output voltage generating means configured to obtain a secondary DC output voltage by inputting an alternating voltage obtained in the secondary winding and performing a rectifying operation, and to a level of the secondary DC output voltage. And a constant voltage control means for performing constant voltage control by varying the switching frequency of the switching element. Then, a step-up transformer configured to step up the alternating voltage of the insulating converter transformer input to the primary side and transmit the stepped-up voltage to the secondary side, and input a step-up voltage obtained at the secondary side of the step-up transformer to perform rectification operation By doing so, a DC high voltage generating means configured to obtain a DC high voltage at a predetermined high voltage level is provided.

That is, according to the present invention, the resonance voltage obtained from the secondary side of the insulated converter transformer constituting the switching power supply circuit of the complex resonance type is directly input to the primary side of the step-up transformer. I have to. Then, after boosting the resonance voltage in the boosting transformer, the DC high voltage generating means obtains a DC high voltage that is set to a predetermined high voltage level. That is, in the present invention,
For example, in order to obtain a high DC voltage required to perform horizontal deflection of a television receiver, a system for obtaining an alternating voltage using a flyback pulse obtained in a horizontal deflection circuit system should be omitted. Can be.

[0032]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure is provided with a voltage resonance type converter having a single switching element Q1 and performing a switching operation by a so-called single-end system by a self-excited system. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1. Then, the commercial AC power supply (AC input voltage VAC) generates a rectified and smoothed voltage Ei corresponding to a level that is one time higher than the AC input voltage VAC by the bridge rectifier circuit Di and the smoothing capacitor Ci.

The base of the switching element Q1 is connected to a smoothing capacitor Ci (rectified smoothing voltage E) via a starting resistor RS.
It is connected to the positive electrode side of i) so that the base current at the time of startup is obtained from the rectification smoothing line. A drive winding N is provided between the base of the switching element Q1 and the primary side ground.
B, a series resonance circuit for driving self-excited oscillation, comprising a series connection circuit of a resonance capacitor CB and a base current limiting resistor RB is connected. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Forms a primary parallel resonance circuit of the voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT described later. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vcp across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit due to the action of the voltage resonance type operation. Is obtained.

The orthogonal control transformer PRT is a saturable reactor on which a resonance current detection winding ND, a driving winding NB, and a control winding NC are wound. This orthogonal control transformer P
RT drives switching element Q1 and is provided for constant voltage control. This orthogonal control transformer P
As a structure of the RT, although not shown, a three-dimensional core is formed by joining ends of two magnetic legs of two double U-shaped cores having four magnetic legs. Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is constructed by winding in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detecting winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching element Q1 Is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is applied to the drive winding NB. appear. This drive voltage is output from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. As shown in FIG. 7, the insulating converter transformer PIT includes, for example, E-shaped cores CR1 and CR2 made of a ferrite material.
An EE-type core is provided so that the magnetic legs of the EE-type core are opposed to each other. A primary winding N1 and a secondary winding N2 are attached to the center magnetic leg of the EE-type core by using a divided bobbin B. Each is wound in a divided state. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. The gap G is the E type core CR1,
The CR2 can be formed by making the central magnetic leg shorter than the two outer magnetic legs. Also, as the coupling coefficient k,
For example, a loosely coupled state of k と い う 0.85 is obtained, and accordingly, a saturated state is hardly obtained.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1 as shown in FIG. 1, and the other end is connected to a smoothing capacitor through a series connection of the resonance current detecting winding ND. It is connected to the positive electrode of Ci (rectified smoothed voltage Ei).

As a configuration on the secondary side shown in this figure, a center tap is provided for the secondary winding N2. The winding portion assumed to be between the winding end end of the secondary winding N2 and the center tap is defined as the first secondary winding N2A, and the winding start end of the secondary winding N2. A winding portion located between the center tap and the center tap is a second secondary winding N3. That is, as the secondary winding N2 of the insulating converter transformer PIT, the second secondary winding N3 is wound around one end of the first secondary winding N2A for obtaining the secondary DC output voltage EO1. It is formed.

In this case, the end of the first secondary winding N2A on the center tap side is connected to the connection point between the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via the series connection of the series resonance capacitor Cs. Connected. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.

As a result, in such a connection form, a voltage doubler half-wave rectifier circuit composed of a set of [series resonance capacitor Cs, rectifier diodes DO1, DO2, smoothing capacitor CO1] is provided. Here, the series resonance capacitor Cs
Is a rectifier diode DO1, which is caused by its own capacitance and the leakage inductance component of the first secondary winding N2A.
A series resonance circuit corresponding to the ON / OFF operation of DO2 is formed. That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is provided with a series resonance circuit for obtaining a current resonance operation. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

Here, the [series resonance capacitor Cs,
The double voltage half-wave rectification operation by the combination of the rectifier diodes DO1, DO2 and the smoothing capacitor CO1] is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2 (N2A). In the period during which the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the polarity (mutual inductance M) between the primary winding N1 and the secondary winding N2 (N2A) is -M. In operation, the operation of charging the series resonance capacitor Cs with the rectified current I2 rectified by the rectifier diode DO2 is obtained by the leakage inductance of the secondary winding N2 (N2A) and the series resonance capacitor Cs. Then, during the period when the rectifying diode DO2 is turned off and the rectifying diode DO1 is turned on to perform the rectifying operation, the primary winding N
The polarity mode (mutual inductance M) between 1 and the secondary winding N2 (N2A) is an additional polarity mode in which the polarity is + M, and the voltage induced in the secondary winding N2A is applied with the potential of the series resonance capacitor Cs to smooth the state. The operation is such that the capacitor CO1 is charged.

By the way, the insulation converter transformer PIT
, The inductance L1 of the primary winding N1 and the secondary winding are determined by the relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 (N2A) and the connection of the rectifier diode DO (DO1, DO2). The mutual inductance M of the inductance L2A of N2A may be + M or -M. For example, when the connection configuration shown in FIG. 8A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 8B is employed, the mutual inductance is -M.
If this is made to correspond to the operation on the secondary side of the power supply circuit of the present embodiment shown in FIG. 1, for example, when the alternating voltage obtained in the secondary winding N2 (N2A) has a positive polarity, The operation in which the rectification current flows in the rectification can be regarded as the + M operation mode (forward mode), and conversely, the secondary winding N2 (N2A)
The operation in which the rectified current flows through the doubled voltage half-wave rectifier circuit when the obtained alternating voltage has the negative polarity can be regarded as the -M operation mode (flyback method). That is, in the power supply circuit shown in FIG. 1, each time the alternating voltage obtained in the secondary winding N2 (N2A) becomes positive / negative, it operates in the mode of mutual inductance of + M / -M. As described above, the rectifying operation is performed by using both the positive polarity mode (+ M; forward operation) and the depolarization mode (-M; flyback operation), so that the smoothing capacitor CO1 has: DC output voltage (for example, 135 V) EO1 corresponding to almost twice the induced voltage of the first secondary winding N2A
Is obtained. The DC output voltage EO1 is controlled by the control circuit 1
Is also branched and input. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is changed in accordance with the change in the DC output voltage level EO1 on the secondary side, so that the control current is wound around the orthogonal control transformer PRT. The inductance LB of the driven winding NB is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This changes the switching frequency of the switching element Q1, thereby stabilizing the secondary-side DC output voltage EO1.

In the power supply circuit of the present embodiment shown in FIG. 1, when an orthogonal control transformer PRT for variably controlling the inductance LB of the drive winding NB is provided, the switching element Q1 is used to change the switching frequency. The period TOFF during which is turned off is kept constant, and the period TON during which it is turned on is variably controlled. That is,
In this power supply circuit, as a constant voltage control operation, an operation is performed to variably control the switching frequency, thereby performing resonance impedance control on the switching output, and at the same time, controlling the conduction angle of the switching element Q1 in the switching cycle (PWM control). ) Can also be seen as doing. Then, this composite control operation is
This is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Further, the power supply circuit of the present embodiment shown in FIG. 1 is provided with a high voltage generating circuit 4 having a step-up transformer HVT and a high voltage rectifying circuit. First, a primary winding N4 is wound on the primary side of the step-up transformer HVT. This primary winding N4 is connected to the secondary winding N2 (formed by connecting the first secondary winding N2A and the second secondary winding N3 in series) of the insulating converter transformer PIT described above. Connected in parallel. As described above, the resonance voltage V3 is generated in the secondary winding N2 by the action of the series resonance capacitor Cs, and the resonance voltage V3 is transmitted to the primary winding N4. Then, on the secondary side of the step-up transformer HVT, the alternating high voltage excited by the resonance voltage V3 obtained in the secondary winding N4 is used to finally generate the high DC voltage EHV as described later. Generate and output.

FIG. 6 is a sectional view of the step-up transformer HVT, and the structure of the step-up transformer HVT will be described with reference to FIG. The step-up transformer HVT shown in FIG.
Similarly to the flyback transformer FBT shown in FIG. 11, the square core 30 is formed by combining the magnetic legs of the two U-shaped cores CR10 and CR20 so as to face each other. And these two U-shaped cores CR1
A gap G is provided at a portion where the 0 end and the end of the U-shaped core CR20 face each other. Then, as shown in the figure, by attaching a low-voltage winding bobbin LB and a high-voltage winding bobbin HB to one of the magnetic legs of the rectangular core 30, these low-voltage winding bobbin LB and high-voltage winding bobbin HB
For the primary winding N4 and the secondary high-voltage winding NH, respectively.
V is divided and wound. The primary winding N4 is wound around the low-voltage winding bobbin LB, and a plurality of secondary high-voltage windings NHV are wound around the high-voltage winding bobbin HB by inserting and winding up the interlayer film F. Will be.

However, this case is different from the flyback transformer FBT shown in FIG. 11 in that a litz wire is used for the primary winding N4. Step-up transformer HVT
The alternating voltage generated in the primary winding N4 is the secondary output of the insulating converter transformer PIT, that is, its frequency corresponds to the switching frequency of the switching element Q1. Since this switching frequency uses a predetermined frequency range, for example, within a range of about several tens kHz to 200 kHz, FIG.
The frequency becomes higher than the frequency (horizontal synchronization frequency fH) of the primary current of the flyback transformer FBT shown in FIG.
In this case, if a litz wire is used for the primary winding N4 as in the present embodiment, generation of an eddy current in the primary winding N4 can be prevented.

In this embodiment, the case where the secondary high-voltage winding NVH of the step-up transformer HVT is wound by interlayer winding is shown. The step-up transformer HVT of the present invention is not limited to the interlayer winding, and may be, for example, a high-voltage bobbin HB.
Can be divided into a plurality of regions, and the secondary side high-voltage windings NHV can be wound around each divided region, that is, can be wound by so-called divided winding. That is, the plurality of secondary high-voltage windings NHV wound around the high-pressure bobbin HB may be wound in a state in which they are insulated from each other.

On the secondary side of the step-up transformer HVT shown in FIG. 1, for example, five sets of secondary high-voltage windings NVH1, NVH2, NVH
3, NHV4 and NHV5 are wound in a divided state.
The primary winding N4 and the secondary high-voltage winding NHV are wound so that the polarity (winding direction) is the same. For example, the winding start end of the secondary-side high-voltage winding NHV1 is connected to one end of the high-voltage capacitor CHV1, and the winding end is connected to the anode of the high-voltage rectifier diode DHV1. The cathode of the high voltage rectifier diode DHV1 is connected to the other end of the high voltage capacitor CHV1.

In such a connection form, as a result, a double voltage half-wave rectifier circuit composed of [secondary high voltage winding NHV1, high voltage rectifier diode DHV1, high voltage capacitor CHV1] is formed. The voltage doubler rectification operation of such a voltage doubler half-wave rectifier circuit is as follows. First, during the period when the high-voltage rectifier diode DHV1 is turned off, an operation of charging the high-voltage capacitor CHV1 with the rectified current rectified by the high-voltage rectifier diode DHV1 is obtained. Then, during the period in which the high-voltage rectifier diode DHV1 is turned on, an operation is obtained in which the voltage across the high-voltage capacitor CHV1 is added to the induced voltage induced in the secondary high-voltage winding NHV1. Thereby, [the secondary high voltage winding NVH
1, high voltage rectifier diode DHV1, high voltage capacitor CHV1]
In the voltage doubler rectifier circuit, the voltage corresponding to almost twice the induced voltage induced in the secondary high voltage winding NHV1 is obtained.

In the power supply circuit according to the present embodiment, the above-mentioned double voltage half-wave is applied to the secondary high voltage windings NHV2, NHV3, NHV4 and NHV5 wound on the secondary side of the step-up transformer HVT. A double voltage half-wave rectifier circuit similar to the rectifier circuit is formed. That is, on the secondary side of the step-up transformer HVT, [the secondary side high voltage winding NHV1, the high voltage rectifier diode DH
V1, high voltage capacitor CHV1], [secondary high voltage winding NVV2,
High voltage rectifier diode DHV2, high voltage capacitor CHV2],
[Secondary high voltage winding NHV3, high voltage rectifier diode DHV3, high voltage capacitor CHV3], [Secondary high voltage winding NHV4, high voltage rectifier diode DHV4, high voltage capacitor CHV4], [Secondary high voltage winding NHV5, high voltage rectifier diode] DHV5 and the high voltage capacitor CHV5].

In the circuit shown in FIG.
A multiple voltage rectifier circuit in which a set of voltage doubler half wave rectifier circuits are connected in series is formed. When the multiplying voltage rectifier circuit composed of these five sets of voltage doubler rectifier circuits performs a rectifying operation, the smoothing capacitor CHVO1 receives the induced voltage induced on the secondary high voltage winding NHV via the high voltage rectifier diode DO1. Almost one
The charging operation is performed by the voltage corresponding to 0 times. As a result, a high-level DC voltage corresponding to ten times the voltage level obtained in each of the secondary high-voltage windings NHV1 to NHV5 is obtained at both ends of the smoothing capacitor CHVO1. That is, a 10-fold voltage half-wave rectifier circuit is formed as the multiple voltage rectifier circuit in this case. And the DC high voltage obtained across the smoothing capacitor CHVO1 is
DC high voltage EHV used as anode voltage of CRT
It is used as

In the circuit shown in FIG. 1, each secondary high voltage winding N provided on the secondary side of the step-up transformer HVT
No current resonance operation is caused by HV1 to NHV5 and each of the high voltage capacitors CHV1 to CHV5. This is because the parallel resonance frequency caused by the leakage inductance component of the secondary winding N2 of the insulated converter transformer PIT and the capacitance component of the secondary side series resonance capacitor Cs is increased by the step-up transformer HV.
This is because the frequency is selected to be sufficiently lower than the series resonance frequency of the leakage inductance components of the secondary high voltage windings NHV1 to NHV5 of T and the capacitance components of the high voltage capacitors CHV1 to CHV5.

Further, on the secondary side of the step-up transformer HVT,
For example, in order to obtain a focus voltage EFV of a CRT, a series connection circuit of [resistor R1, variable resistor R2, resistor R3] is connected in parallel with a series circuit including five sets of voltage doubler half-wave rectifier circuits.

Here, for example, the insulation converter transformer P
The voltage level of the secondary DC output voltage EO1 output from the secondary side of the IT is 135 V, and the voltage level of the DC high voltage EHV output from the secondary side of the step-up transformer HVT is 3
When actually configuring the circuit shown in FIG. 1 so that 2 kV can be obtained, the secondary side series resonance capacitor Cs = 0.1
5 μF, primary winding N4 of the step-up transformer HVT = 20T,
Secondary high voltage windings NHV1 to NHV5 = 500T, high voltage capacitors CHV1 to CHV5 = 100PF / 5kV, and smoothing capacitor CHVO1 = 1000PF / 40kV are selected. In addition,
At this time, each secondary-side high-voltage winding NVH1 of the step-up transformer HVT
It is desirable that the voltage level of the induced voltage induced in .about.NHV5 is 5 kV or less.

FIGS. 2 and 3 show operation waveforms of the power supply circuit of the present embodiment shown in FIG. 2 shows operation waveforms when, for example, the AC input voltage VAC is 100 V and the load current IHV flowing on the secondary side of the step-up transformer HVT is 2 mA, and FIG. 3 shows the operation waveform when the AC input voltage VAC is 100 V and the step-up transformer HVT.
FIG. 7 is a diagram showing operation waveforms when the load current IHV flowing through the secondary side of FIG.

When the load current IHV flowing on the secondary side of the step-up transformer HVT is 2 mA, the switching operation is performed by the switching element Q1 performing the switching operation by the series resonance circuit (NB, CB) as the self-excited oscillation drive circuit. A primary-side parallel resonance voltage Vcr as shown in FIG. 2A is obtained at both ends of the parallel connection circuit of the element Q1 // parallel resonance capacitor Cr by the action of the parallel resonance circuit. As shown in the figure, the primary side parallel resonance voltage Vcr has a waveform that becomes a sine wave pulse during the period TOFF when the switching element Q1 is on and is zero during the period TOFF when the switching element Q1 is off, as shown in FIG. It corresponds to. Also, the collector of the switching element Q1 is shown in FIG.
The collector current IC having the waveform shown in FIG. 2 flows, and the resonance current I1 having the waveform shown in FIG. 2B flows through the primary winding N1 of the insulating converter transformer PIT.

The switching output is transmitted to the secondary side of the insulated converter transformer PIT by the on / off operation of the switching element Q1. As a result, currents I2 and I3 having waveforms as shown in FIGS. 2D and 2E flow on the secondary side of the insulating converter transformer PIT, and a current flows through the primary winding N4 of the step-up transformer HVT. A resonance voltage V3 as shown in FIG. 2 (f) is applied.

When the load current IHV flowing through the secondary side of the step-up transformer HVT is 0 mA, the switching element Q1
// A parallel resonance voltage Vc on the primary side as shown in FIG.
r is obtained, and a collector current IC having a waveform as shown in FIG. 3C flows through the collector of the switching element Q1. Accordingly, a resonance current I1 having a waveform as shown in FIG. 3B flows through the primary winding N1 of the insulating converter transformer PIT.

Then, by the on / off operation of the switching element Q1, currents I2 and I3 having waveforms as shown in FIGS. 3D and 3E flow on the secondary side of the insulating converter transformer PIT. Thus, the resonance voltage V3 as shown in FIG. 3 (f) is applied to the primary winding N4 of the step-up transformer HVT.
Is applied.

Comparing the operation waveforms shown in FIGS. 2 and 3, when the high-voltage load connected to the secondary side of the step-up transformer HVT is 64 W (IHV = 2 mA), the switching frequency of the switching element Q 1 is 125 kHz. And
As shown in FIG. 2, the ON period TON of the switching element Q1 is controlled to 5 μs, and the OFF period TOFF is controlled to 3 μs. On the other hand, when the high-voltage load connected to the secondary side of the step-up transformer HVT becomes 0 W (IHV = 0 mA), the switching frequency of the switching element Q1 becomes 13
0 kHz, and as shown in FIG.
The period TOFF during which 1 is turned off is controlled to 3.5 μs, and the period TON during which it is turned on is controlled to 4.2 μs.

A comparison between the circuit of the present embodiment having such a configuration shown in FIG. 1 and the conventional circuit shown in FIG. 11 shows that the circuit shown in FIG. The horizontal output circuit 40 is controlled by the secondary DC voltage EO1.
Is raised to obtain a DC high voltage EHV from the high voltage generation circuit 50. On the other hand, in the power supply circuit of the present embodiment shown in FIG. 1, the resonance voltage V3 output from the secondary side of the insulating converter transformer PIT is input to the high voltage generation circuit 4 and the resonance voltage V3 is boosted. Thus, a DC high voltage EHV is obtained. That is, in the circuit of the present embodiment shown in FIG. 1, the secondary-side output of the insulated converter transformer PIT functioning as a switching power supply circuit is directly input to the high-voltage generating circuit 4. The high voltage generating circuit 4 obtains the high DC voltage EHV without interposing the horizontal output circuit 40 for converting the DC output voltage EO1 of the switching power supply circuit 10 into the flyback pulse voltage as in the circuit shown in FIG.

As a result, the conventional circuit shown in FIG. 11 has a power conversion efficiency of about 70% when obtaining the DC high voltage EHV from the input voltage, whereas the conventional circuit shown in FIG. In the power supply circuit of the embodiment, the power conversion efficiency is 87.
The power conversion efficiency can be improved to 5%, and the power conversion efficiency can be improved by about 17.5%. actually,
For example, when a high-voltage load power of 64 W is output from the circuit shown in FIG. 11 and the circuit of the present embodiment shown in FIG.
While the circuit shown in FIG. 11 requires 91.4 W of input power, the circuit of the present embodiment shown in FIG. 1 requires only 73.1 W of input power and about 18.3 W of input power. The power was reduced.

In the circuit of this embodiment shown in FIG. 1, the resonance voltage V3 output from the secondary side of the insulating converter transformer PIT is input to the primary winding N4 of the step-up transformer HVT. The resonance voltage waveform input to the primary side of the step-up transformer HVT has a rectangular shape whose positive and negative levels are almost symmetric as shown in FIG. As a result, the induced voltage output from the secondary side of the step-up transformer HVT also has a rectangular waveform. When such an induced voltage is rectified by the high-voltage rectifier diodes DHV1 to DHV5, the conduction angle of the high-voltage rectifier diode DHV is wider than that in the case shown in FIG.

Accordingly, when the circuit of the present embodiment is applied to a television receiver or the like, as shown in FIG. 5, when the high-voltage load power is 0 W (IHV = 0 mA), the voltage level of the DC high voltage EHV is 32 kV. It has become. On the other hand, even if the high-voltage load power increases to 64 W (IHV = 2 mA),
The voltage level of the DC high voltage EHV is about 31 kV.
In other words, in the circuit of the present embodiment, the range of the high-voltage load power (0
W to 64 W) in the voltage level width Δ of the DC high voltage EHV.
EHV becomes about 1.0 kV.

That is, in the circuit shown in FIG. 11, the high-voltage load power is from no load (IHV = 0 mA) to 64 W (IHV =
The voltage level width ΔEHV of the DC high voltage EHV at the time of fluctuating to 2 mA) is 1.5 kV, but in the circuit of the present embodiment shown in FIG.
Can be suppressed to 1.0 kV. Therefore, by applying the circuit of the present embodiment to, for example, a television receiver and supplying a DC high voltage EHV to the anode electrode of the CRT, the electron beam output from the CRT by the DC high voltage EHV can be obtained. Since the amplitude fluctuation in the horizontal direction can be suppressed, it is not necessary to provide a zooming correction circuit or the like for the horizontal output circuit of the television receiver.

The switching frequency of the step-up transformer HVT corresponds to the switching frequency of the switching element Q1, and is not synchronized with, for example, the cycle of the horizontal synchronizing signal fH of the video signal. Thereby, the secondary high voltage winding NVH1 of the step-up transformer HVT is generated by the leakage magnetic flux and the leakage inductance from the step-up transformer HVT.
Ringing does not occur in the induced voltage of .about.NHV5. Therefore, even when the circuit of this embodiment is applied to a television receiver, for example, raster ringing does not occur on the screen of a CRT, and even if ringing occurs, the circuit of this embodiment Since the ringing current component is not superimposed on the horizontal deflection current IDY since it is formed independently of the horizontal deflection circuit of the high voltage generation circuit 4, raster ringing does not occur on the screen of the CRT.

The primary winding N4 of the step-up transformer HVT
DC component is not superimposed on the resonance current flowing through
The shape of the core of the step-up transformer HVT can be reduced in size, and it is not necessary to increase the diameter of the primary winding N4, so that the shape can be reduced in size. Further, since the peak current value of the current flowing through the switching element Q1 is also reduced, the heat generation of the switching element Q1 is suppressed, and it is not necessary to take measures such as attaching a heat sink to the switching element Q1.

The circuit configuration of the power supply circuit of the present invention is not limited to the circuit configuration shown in FIG. FIG. 9 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG. In this figure, the voltage resonance type converter provided on the primary side has a separately excited configuration, for example, one MOS-F
A switching element Q21 based on ET is provided. The drain of the switching element Q21 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1, and the source is connected to the primary side ground. Here, the parallel resonance capacitor Cr is connected in parallel between the drain and the source. Further, a clamp diode DD is connected in parallel between the drain and the source.

The switching element Q21 is driven by the oscillation / drive circuit 2 so that the switching operation described above with reference to FIG. 1 is obtained. That is, the control circuit 1 supplies a current or voltage of a level fluctuated according to the fluctuation of the DC output voltage EO1 to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, a switching drive signal (voltage) whose cycle is varied according to the output level from the control circuit 1 is applied to the gate of the switching element Q21 so as to stabilize the DC output voltage EO1. Output to As a result, the switching frequency of the switching element Q21 is varied. In this case, as described in FIG. 1, the period during which the switching element Q21 is off is constant, and the period during which the switching element Q21 is on is varied. The generated switching drive signal is output.

In this case, the starting circuit 3 is supplied with the rectified and smoothed voltage Ei obtained from the smoothing capacitor Ci as an operating power source.
The start-up circuit 3 executes an operation for starting the oscillation / drive circuit 2 by the start-up voltage obtained in the winding N5 additionally wound around the IT.

On the secondary side of the insulating converter transformer PIT, a tap is provided for the secondary winding N2. Then, a bridge rectifier circuit DBR and a smoothing circuit are connected to a first secondary winding N2A formed by a winding end end and a tap of the secondary winding N2 via a series connection of a secondary side series resonance capacitor Cs. A full-wave rectifier circuit composed of a capacitor CO1 is connected. That is, also in this case, as the secondary winding N2 of the insulating converter transformer PIT, the second secondary winding N3 is provided at one end of the first secondary winding N2A for obtaining the secondary DC output voltage EO1. It is formed by winding up. Then, the primary winding N4 of the step-up transformer HVT of the high-voltage generating circuit 5 is connected in parallel to the secondary winding N2 composed of the first secondary winding N2A and the second secondary winding N3. I have.

The secondary side of the step-up transformer HVT of the high-voltage generating circuit 5 shown in FIG.
HV1 to HV3 are divided and wound independently. The winding direction of the secondary high-voltage windings NHV1 to NHV3 has the same polarity (winding direction) as that of the primary winding N4, as in the circuit shown in FIG.

The double voltage half-wave rectifier circuit as described above is formed for each of the three sets of secondary high voltage windings NHV1 to NHV3 wound on the secondary side of the step-up transformer HVT. ing. That is, on the secondary side of the step-up transformer HVT, [secondary high voltage winding NHV1, high voltage rectifier diode DHV1, high voltage capacitor CHV1], [secondary high voltage winding NHV1]
HV2, high voltage rectifier diode DHV2, high voltage capacitor CHV
2], [Secondary high voltage winding NVH3, high voltage rectifier diode DHV
3, a high voltage capacitor CHV3], and three sets of voltage doubler half-wave rectifier circuits are formed. These three sets of voltage doubler rectifier circuits are connected in series to form a multiple voltage doubler rectifier circuit. Therefore, as the multiple voltage rectifier circuit in this case, a six-fold voltage half-wave rectifier circuit for obtaining a DC voltage EHV corresponding to a level approximately six times the induced voltage induced in each of the secondary high voltage windings NHV1 to NHV3. Is formed.

In the circuit having such a configuration, the number of turns (number of turns) of the secondary winding N2 of the insulating converter transformer PIT is larger than the number of turns of the secondary winding N2 of the circuit shown in FIG. It is configured to rotate. Therefore, the voltage level input to the primary winding N4 of the step-up transformer HVT is higher than, for example, the circuit shown in FIG. For this reason, in the circuit shown in FIG. 9, for example, a six-fold voltage rectifier circuit having a voltage multiple smaller than that of the ten-fold voltage rectifier circuit which is the multiple voltage rectifier circuit shown in FIG. 1 is provided on the secondary side of the step-up transformer HVT. Even if provided, it is possible to obtain a DC high voltage EHV having substantially the same voltage level as the DC high voltage EHV output from the circuit shown in FIG. In the case of such a configuration, the same effect as the circuit of the present embodiment shown in FIG. 1 can be obtained, and the circuit shown in FIG.
Since the number of secondary high-voltage windings NHV independently wound on the secondary side of the step-up transformer HVT can be reduced, the size of the step-up transformer HVT can be further reduced.

FIG. 10 is a diagram showing a circuit configuration of a power supply circuit according to a third embodiment of the present invention. In addition,
In FIG. 10, the configuration on the primary side is the same as the configuration in FIG. 1, and thus the same portions are denoted by the same reference numerals and description thereof will be omitted.

The isolated converter transformer PI shown in FIG.
On the secondary side of T, a center tap is provided for the secondary winding N2. Then, a full-wave rectifier circuit including a bridge rectifier circuit DBR and a smoothing capacitor CO1 is connected to a first secondary winding N2A formed by a winding end and a center tap of the secondary winding N2. A DC output voltage EO1 is obtained. Then, the primary winding N4 of the step-up transformer HVT is connected in parallel to the secondary winding N2. In this case, the end of the second secondary winding N3 and the primary winding NVT of the step-up transformer HVT are connected. A secondary side series resonance capacitor Cs is inserted in series with the line N4.

On the secondary side of the step-up transformer HVT, for example, two sets of secondary-side high-voltage windings NVH1, NHV2 are wound independently. In this case, the secondary high-voltage winding NHV1
-NHV2 have the same polarity (winding direction) as the primary winding N4. In this case, the step-up transformer HV
The winding start end of the secondary-side high-voltage winding NHV1 of T is connected to a high-voltage rectifier diode DH
Connected to the connection point between the anode of V1 and the cathode of the high-voltage rectifier diode DHV2, and connected to the high-voltage capacitor CHV
It is connected to the connection point of the anode of the high-voltage rectifier diode DHV3 and the cathode of the high-voltage rectifier diode DHV4 through the series connection of the two. On the other hand, the end of the secondary high-voltage winding NHV1 is connected to the connection point between the negative electrode of the smoothing capacitor CHVO1 and the positive electrode of the smoothing capacitor CHVO2. The negative electrode of the smoothing capacitor CHVO1 and the smoothing capacitor CHVO2
High voltage rectifier diode DHV2
And the cathode of the high voltage rectifier diode DHV3 are connected. Smoothing capacitor CHVO1 and smoothing capacitor CHV
O2 is the negative electrode of the smoothing capacitor CHVO1 and the smoothing capacitor C
After being connected in series with the positive electrode of HVO2, the positive electrode of the smoothing capacitor CHVO1 is connected to the cathode of the high-voltage rectifier diode DHV1, and the negative electrode of the smoothing capacitor and CO2 is connected to the secondary side ground.

In such a connection form, a voltage doubler rectifier circuit composed of a set of [high voltage capacitor CHV1, high voltage rectifier diodes DHV1, DHV2, smoothing capacitor CHVO1] for secondary high voltage winding NVH1, and [high voltage capacitor CHV2] , A high voltage rectifier diode DHV3, DHV4, and a smoothing capacitor CHVO2], and the outputs (smoothing capacitors CHVO1, CHVO2) of these voltage doubler rectifier circuits are connected in series. . Therefore, as a whole rectifier circuit combining these voltage doubler rectifier circuits, a smoothing capacitor CHVO1-a smoothing capacitor CHV connected in series
At both ends of O2, an output voltage corresponding to four times the alternating voltage obtained in the secondary high-voltage winding NVV1 is obtained. That is, a quadruple voltage full-wave rectifier circuit capable of obtaining an output corresponding to a voltage level approximately four times the induced voltage is formed in one secondary high-voltage winding NHV1.

The operation of the above-described quadruple voltage full-wave rectifier circuit is as follows. The quadruple voltage full-wave rectifier circuit performs the rectification operation by inputting the obtained alternating voltage to the secondary high voltage winding NHV1. At this time, the [high voltage capacitor CHV1, the high voltage rectifier diodes DHV1, DHV2, the smoothing capacitor CHVO1] ] Will be described below. First, during the period when the high-voltage rectifier diode DHV1 is turned off and the high-voltage rectifier diode DHV2 is turned on, the rectified current rectified by the high-voltage rectifier diode DHV2 by the leakage inductance of the secondary high-voltage winding NHV1 and the high-voltage capacitor CHV1 is converted to the high-voltage capacitor. The operation of charging CHV1 is obtained. During the period in which the high-voltage rectifier diode DHV2 is turned off and the high-voltage rectifier diode DHV1 is turned on to perform the rectification operation, the voltage induced in the secondary-side high-voltage winding NHV1 is applied with the potential of the high-voltage capacitor CHV1. Then, the operation is performed to charge the smoothing capacitor CHVO1.

By performing the rectification operation as described above, a DC voltage (rectified smoothed voltage) corresponding to almost twice the induced voltage of the secondary high voltage winding NHV1 is obtained in the smoothing capacitor CHVO1. In addition, [High Voltage Capacitor CHV
2. A voltage doubler rectifier circuit composed of a set of high-voltage rectifier diodes DHV3, DHV4 and a smoothing capacitor CHVO2] has a similar operation, so that both ends of the smoothing capacitor CHVO2 have approximately two times the induced voltage of the secondary high-voltage winding NHV1. A DC voltage corresponding to the double is obtained. Thereby, in the quadruple voltage full-wave rectifier circuit, the smoothing capacitor CHV connected in series is connected.
At both ends of O1-CHVO2, a DC voltage corresponding to four times the induced voltage induced in the secondary high voltage winding NHV1 is obtained.

In the circuit shown in FIG. 10, the above-mentioned quadruple voltage full-wave rectifier circuit is formed also for the secondary high voltage winding NHV2 wound on the secondary side of the step-up transformer HVT. Thus, a multiple voltage rectifier circuit is formed by connecting these two sets of quadruple voltage rectifier circuits in series. That is, in this case, as a multiple voltage rectifier circuit, the secondary side high voltage windings NHV1,
An eight-fold voltage rectifier circuit for obtaining a DC high voltage EHV (rectified smoothed voltage) corresponding to a level approximately eight times the induced voltage induced in the NHV2 is formed.

In a circuit having such a configuration,
The configuration is such that the number of turns (number of turns) of the secondary winding N2 of the insulating converter transformer PIT is greater than the number of turns of the secondary winding N2 of the circuit shown in FIG. Therefore, the voltage level input to the primary winding N4 of the step-up transformer HVT is higher than, for example, the circuit shown in FIG. Therefore, in the circuit shown in FIG. 10, for example, an eight-fold voltage rectification circuit having a voltage multiple smaller than that of the ten-fold voltage rectification circuit which is the multiple voltage rectification circuit shown in FIG. 1 is provided on the secondary side of the step-up transformer HVT. Even if it is provided, it becomes possible to obtain a DC high voltage EHV having substantially the same voltage level as the DC high voltage EHV output from the circuit shown in FIG. In such a configuration, the same effect as that of the circuit of the present embodiment shown in FIG. 1 can be obtained, and moreover, the circuit of the present embodiment shown in FIG. It is possible to reduce the size.

In the present embodiment, a 10-fold voltage rectifier circuit, 6
Although the case where the voltage doubler rectifier circuit and the eight-fold voltage rectifier circuit are provided is described as an example, the present invention is not limited to the rectifier circuit having such a configuration. Various rectifier circuits configured to obtain a DC high voltage EHV in which the induced voltage level induced in the voltage winding NHV corresponds to a predetermined voltage level can be applied.

In the present embodiment, a case where a rectifying circuit of a full-wave rectifying system is provided as a rectifying circuit for obtaining a secondary output voltage EO1 on the secondary side of the insulating converter transformer PIT will be described as an example. However, the present invention is not limited to the rectifier circuit having such a configuration, and various configurations of the secondary-side rectifier circuit of the insulated converter transformer PIT according to the present invention can be considered.

[0087]

As described above, in the switching power supply circuit of the present invention, the resonance voltage obtained on the secondary side of the insulated converter transformer constituting the switching power supply circuit as a complex resonance type is converted into the primary voltage of the boosting transformer. Direct input to the side. Then, after boosting the resonance voltage in the boosting transformer, the DC high voltage generating means obtains a DC high voltage that is set to a predetermined high voltage level. Therefore, if the switching power supply circuit of the present invention is applied to a television receiver, for example, when obtaining a high DC voltage to be supplied to the anode of a cathode ray tube, the flyback pulse voltage obtained from the secondary side DC output voltage , There is no need to convert to an alternating voltage again.
As a result, the power conversion efficiency when a DC high voltage is obtained from the input voltage can be improved.

Further, according to the present invention, even when the high-voltage load fluctuates, the DC high voltage output by the DC high voltage generating means can have a smaller range of voltage fluctuation than in the prior art. Therefore, if the present invention is applied to, for example, a high voltage supply unit of a television receiver, it becomes possible to suppress, for example, horizontal amplitude fluctuations of an electron beam output from a cathode ray tube.

Further, since the alternating voltage of the insulating converter transformer is input to the primary side of the step-up transformer, no DC component is superimposed on the primary side current flowing through the primary side of the step-up transformer. , And the diameter of the primary winding can be reduced. As a result, the size and weight of the step-up transformer can be reduced. Also, in this case, the peak value of the current flowing through the switching element is reduced, and the amount of heat generated by the switching element is also reduced.

Further, a secondary winding of the insulating converter transformer is provided with a second secondary winding with respect to a first secondary winding for obtaining an alternating voltage inputted to the DC output voltage generating means. The winding is wound up, and an alternating voltage is input from the second secondary winding to the primary winding of the step-up transformer. In this case, the voltage level of the voltage input to the primary winding of the step-up transformer can be increased, and the number of secondary-side high-voltage windings of the step-up transformer can be reduced.
It is possible to reduce the size of the step-up transformer.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 3 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 4 is a diagram showing operation waveforms of the step-up transformer of the present embodiment.

FIG. 5 is a diagram showing a relationship between a DC high voltage output from the high voltage generation circuit of the present embodiment and a high voltage load current.

FIG. 6 is a cross-sectional view illustrating a configuration of a step-up transformer according to the present embodiment.

FIG. 7 is a cross-sectional view illustrating a configuration of an insulating converter transformer.

FIG. 8 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 9 is a circuit diagram illustrating a configuration example of a power supply circuit according to a second embodiment of the present invention.

FIG. 10 is a circuit diagram illustrating a configuration example of a power supply circuit according to a third embodiment of the present invention.

FIG. 11 is a circuit diagram showing a configuration of a conventional high voltage generation circuit and its peripheral circuits.

FIG. 12 is a waveform chart showing an operation of a main part of the circuit shown in FIG. 11;

FIG. 13 is a diagram showing operation waveforms of a flyback transformer.

14 is a diagram showing a relationship between a high DC voltage and a high load current output from the circuit shown in FIG. 11;

FIG. 15 is a cross-sectional view showing a configuration of a flyback transformer provided in the circuit shown in FIG.

[Explanation of symbols]

Reference Signs List 1 control circuit, 2 oscillation / drive circuit, 3 start-up circuit, 4 56 high-voltage generation circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, PRT orthogonal control (drive) transformer,
HVT step-up transformer, Cr primary side parallel resonance capacitor, Cs secondary side series resonance capacitor, N1 N4 primary winding, N2 secondary winding, NHV secondary high voltage winding, NC control winding, NB drive winding, ND resonance Current detection winding, CB resonance capacitor, DBR bridge rectifier circuit, DO1 DO2
Rectifier diode, DHV1 ~ DHV4 High voltage rectifier diode,
CHV1 ~ CHV5 High voltage capacitor, CO1 CHVO1 CHVO2
Smoothing capacitor

Claims (6)

[Claims] A switching means for intermittently outputting an input DC input voltage; an insulating converter transformer for transmitting an output of the switching means to a secondary side; A primary-side voltage resonance circuit inserted so as to be a resonance type; and a secondary-side series resonance formed by connecting a secondary-side series resonance capacitor in series to the secondary winding of the insulating converter transformer. A DC output voltage generating means configured to obtain a secondary DC output voltage by inputting an alternating voltage obtained to a secondary winding of the insulating converter transformer and performing a rectification operation; Constant voltage control means for performing constant voltage control by varying the switching frequency of the switching element according to the level of the secondary DC output voltage A step-up transformer that transmits the alternating voltage input to the primary side to the secondary side to obtain a step-up voltage obtained by stepping up the alternating voltage on the secondary side; And a DC high voltage generation means configured to obtain a DC high voltage that is a predetermined high voltage level by inputting the obtained boosted voltage and performing a rectification operation. 2. A secondary winding of the insulating converter transformer, a first secondary winding wound to obtain an alternating voltage input to the DC output voltage generating means, and a step-up transformer. And a second secondary winding wound around the first secondary winding so as to obtain an alternating voltage input to the primary winding. The switching power supply circuit according to claim 1. 3. The high-voltage direct-current generating means includes a plurality of secondary high-voltage windings independently wound on a secondary side of the step-up transformer, and a plurality of secondary high-voltage windings, respectively. And a plurality of voltage doubler rectifier circuits provided so as to obtain an output voltage having a level corresponding to approximately twice the input boosted voltage. The switching power supply circuit according to claim 1, further comprising: a multiplying voltage rectifier circuit. 4. The high-voltage direct-current generating means includes a plurality of secondary high-voltage windings independently wound on a secondary side of the step-up transformer, and a plurality of secondary high-voltage windings, respectively. And a plurality of quadruple voltage rectifier circuits provided so as to obtain an output voltage having a level corresponding to almost four times the inputted boosted voltage. The switching power supply circuit according to claim 1, further comprising: a multiplying voltage rectifier circuit formed. 5. The switching power supply circuit according to claim 1, wherein in the step-up transformer, the secondary high-voltage winding is wound by interlayer winding. 6. The switching power supply circuit according to claim 1, wherein in the step-up transformer, the secondary high-voltage winding is wound by split winding.