JP2002064981A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To save energy at the time of outputting a constant DC voltage from a switching power supply circuit to be adapted to a television receiver. SOLUTION: When it is required to obtain the DC output voltages E04 to E06 from an AC voltage generated in a secondary coil N5B provided at the secondary of an insulated converter transformer PIT, a control current into the control coils NC2 to NC4 of each universal control transformer PRT-2 to PRT-4 is controlled with control circuits 2 to 4 depending on the level change of the DC output voltages E04 to E06, and the inductance of coils NR2 to NR4 to be controlled is variably controlled. As a result, the DC output voltages E04 to E06 can made constant, and power loss generated by such conversion of the DC output voltages E04 to E06 to the constant voltages can be lowered.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit suitable for various video equipment such as a color television receiver and a projector.

[0002]

2. Description of the Related Art In a video apparatus such as a television receiver or a projector, for example, an analog IC (Integrated C
Some are provided with an ircuit-based circuit block and a digital IC-based circuit block. A video device provided with such an analog IC or digital IC circuit block is provided with a constant voltage power supply for supplying a constant operating voltage to these circuit blocks.

FIG. 8 is a diagram showing a configuration of a switching power supply circuit provided in, for example, a large color television receiver as an example of a conventional power supply circuit provided in the above-described video equipment. is there. In the power supply circuit shown in FIG. 8, a rectified smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC is generated from a commercial AC power supply (AC input voltage VAC) by a bridge rectifier circuit Di and a smoothing capacitor Ci. As the switching converter which receives and switches the rectified smoothed voltage Ei (DC input voltage), there is provided a self-excited type voltage resonance type converter having a single switching element Q1 and performing a so-called single-end switching operation. ing.

The switching element Q1 is driven by a self-excited oscillation drive circuit comprising a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB, and its switching frequency is determined by the drive winding NB and the resonance capacitor CB. Is determined by the resonance frequency of the resonance circuit composed of The startup resistor RS is provided for supplying a startup current obtained on the rectifying / smoothing line to the switching element Q1 when the commercial AC power is turned on.

As shown, a clamp diode DD1 and a primary-side parallel resonance capacitor Cr are connected to the switching element Q1, and the capacitance of the primary-side parallel resonance capacitor Cr itself and the primary winding of the insulating converter transformer PIT are connected. The primary side parallel resonance circuit of the voltage resonance type converter is formed by the leakage inductance L1 on the N1 side.

[0006] The orthogonal control transformer PRT-1 is a saturable reactor in which a resonance current detection winding ND, a drive winding NB, and a control winding NC1 are wound. The orthogonal control transformer PRT-1 is provided for driving the switching element Q1 and for controlling the constant voltage.

[0007] Insulation converter transformer (Power Isolation
Transformer) PIT transmits the switching output of switching element Q1 to the secondary side. Although a detailed description is omitted here, a loose coupling state can be obtained by forming a gap with respect to the core in the insulating converter transformer PIT.

As shown, a secondary winding is formed on the secondary side of the insulating converter transformer PIT so as to wind up the secondary windings N2, N3, N4 and N5. In this case, as shown in the figure, the connection between the secondary windings N4 and N5 is connected to the secondary ground, and the secondary winding is connected between the secondary ground and the winding end of the secondary winding N2. The side parallel resonance capacitor C2 is connected in parallel.

That is, in the power supply circuit shown in FIG. 8, the primary side of the insulated converter transformer PIT is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side obtains a voltage resonance operation. Voltage resonance circuit is provided. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

The secondary side parallel resonance capacitor C
A half-wave rectifying / smoothing circuit comprising a rectifier diode DO1 and a smoothing capacitor CO1 is provided for a secondary winding (N2 + N3 + N4) connected in parallel to the secondary winding 2 and the horizontal winding of the television receiver. A DC output voltage EO1 (135 V) for deflection is obtained.

A half-wave rectifying / smoothing circuit comprising a rectifier diode DO2 and a smoothing capacitor CO2 is provided for the secondary winding (N3 + N4) comprising the secondary windings N3 and N4. From the DC output voltage EO2 for vertical deflection
(15 V), and a rectifier diode DO3 and a smoothing capacitor CO3 are connected to the secondary winding N5 as shown in the figure, and a half-wave rectifying and smoothing circuit composed of the rectifier diode DO3 and the smoothing capacitor CO3 Similarly, a DC output voltage EO3 (-15 V) for vertical deflection is obtained.

That is, on the secondary side of the insulating converter transformer PIT, the DC output voltages EO2, E03 (± 15 V) for vertical deflection are derived from the induced voltage induced in the secondary winding (N3 + N4) and the secondary winding N5. ). Therefore, the number of secondary windings (N2 + N3) and the number of secondary windings N5
Have the same number of turns.

In this case, the secondary side DC output voltage EO1 is also branched and input to the control circuit 1. The control circuit 1 uses the DC output voltage EO2 as the operating voltage and varies the level of the control current (DC current) flowing through the control winding NC1 in accordance with the level change of the DC output voltage EO1, so that the orthogonal control transformer PRT- The control unit variably controls the inductance LB of the driving winding NB wound around the drive coil NB. Thereby, the resonance condition of the resonance circuit of the self-excited oscillation drive circuit formed including the inductance LB of the drive winding NB changes. This is an operation of varying the switching frequency of the switching element Q1, and this operation is intended to make the DC output voltage output from the secondary side constant. In addition, even in such a configuration of the constant voltage control having the orthogonal control transformer PRT-1, since the primary side switching converter is of the voltage resonance type, the switching frequency can be controlled simultaneously with the switching frequency. It can be seen that the control of the conduction angle (PWM control) of the switching element Q1 is performed. Then, this composite control operation is
This is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Further, in the power supply circuit shown in FIG. 8, a DC output voltage EO4 (9V) to be supplied to an analog IC circuit block is obtained from the output of the secondary winding (N3 + N4), and the secondary winding is obtained. DC output voltage EO5 (5V) supplied to the digital IC circuit block from the output of N4
I'm trying to get

In this case, the output of the secondary winding (N3 + N4) is connected to the inductor L21 (4.7) in order to reduce power loss.
rectifier diode DO4 and smoothing capacitor C
The half-wave rectifying / smoothing circuit composed of O4 receives the DC output voltage EO7 (11V) once.
To be converted to And, this DC output voltage E
A DC output voltage EO4 (9 V) to be output to the analog IC circuit block is obtained from O7. The output of the secondary winding N4 is input to a half-wave rectifying / smoothing circuit including a rectifier diode D05 and a smoothing capacitor C05.
In this half-wave rectifying and smoothing circuit, the DC output voltage EO8 (6.
After conversion to 5 V), DC output voltages EO5 (5V) and EO6 (3.3V) to be output to the circuit blocks of the digital IC system are obtained from the DC output voltage EO8.

By the way, the analog IC system and the digital I
DC output voltage E supplied to each circuit block of C system
O4 to EO6 need to be set to a constant voltage so that the voltage fluctuation is within ± 2%. However, even in the combined control type switching power supply circuit as shown in FIG. 8, the DC output voltage level output from the secondary side slightly varies with the variation of the secondary side load power Po. Become. For example, as shown in FIG. 10, when the secondary-side load power Po becomes a light load, the DC output voltages EO2 (15V) and EO8
The voltage level of (6.5 V) slightly decreases.

For this reason, in the power supply circuit shown in FIG. 8, the DC output voltage EO4 (9V) whose voltage fluctuation is constant within ± 2% from the DC output voltage EO7 (11V).
And a constant voltage circuit for obtaining a voltage variation of ± 2 from the DC output voltage EO8 (6.5 V).
% DC output voltage EO5 (5V), E
Each of the voltage regulating circuits for obtaining O6 (3.3 V) is provided.

The constant voltage circuit has, for example, an output current of 2
When the output current is set to A or less, a three-terminal series regulator IC is used, and when the output current is set to 2 A or more, a step-down converter using a chopper regulator IC is used.

In the case of the power supply circuit shown in FIG. 8, the maximum rating of the DC output voltage EO4 is 9 V / 1.5 A, and the output current is 2 A or less. Circuit is a 3-terminal series regulator I
± 2% composed of C-1 and smoothing capacitor CO41
DC output voltage EO4 (9V ± 0.18
V).

The maximum rating of the DC output voltage EO5 is 5 V
/1.5 A, and the output current is 2 A or less.
The DC output voltage EO5 (5 V
± 0.1 V).

On the other hand, the maximum rating of the DC output voltage EO6 is 3.3 V / 3 A, and the output current is 2 A or more. Therefore, in this case, the DC output voltage EO8 is determined by the ferrite bead inductor FB. Via PWM (Pulse Wi
dth Modulation) The DC-DC converter 11 constituted by a step-down chopper circuit of a control method
A DC output voltage EO6 (3.3 V ± 0.07 V) is obtained in which the voltage fluctuation is constant within ± 2%.

The DC-DC converter 11 includes a chopper regulator IC-3, a flywheel diode D1
1 and an inductor L22 (20 μH), and the output voltage output via the inductor L22 is fed back to the chopper regulator IC,
The switching operation is controlled to make the output voltage level constant.

However, such a DC-DC converter 1
In No. 1, since the switching operation has a rectangular waveform, the noise level generated by the switching operation increases. For this reason, a ferrite bead inductor FB is provided in the preceding stage of the chopper regulator IC-3, and a ceramic capacitor Cn is provided in the subsequent stage to suppress the generated switching noise. Also, D
Since the DC output voltage of the C-DC converter 11 includes a harmonic ripple voltage component, the output voltage line includes an electric field capacitor CO61, CO62 and an inductor L23.
(3.3 μH) is provided to remove high frequency ripple voltage components.

FIG. 9 shows operation waveforms of the power supply circuit shown in FIG. FIGS. 9A to 9F show that the total load power of the DC output voltages EO1 to EO6 is 200 W after making the voltage fluctuations of the DC output voltages EO4 to EO6 constant within ± 2%. FIG. 9 shows operation waveforms under the conditions shown in FIG.
(G) to (l) show operation waveforms under the condition that the total load power of the DC output voltages EO1 to EO6 is 100 W.

When the total load power is 200 W, the switching frequency of the switching element Q1 is, for example, 7
It is controlled to be 1.4 kHz, and the ON / OFF period TON / TOFF of the switching element Q1 is 10 μs / 4 μs. The resonance voltage V1 generated at both ends of the primary side parallel resonance capacitor Cr by the on / off operation of the switching element Q1 is shown in FIG. 9A, and is a sine during the period TOFF when the switching element Q1 is off. A wavy pulse waveform is obtained. At this time, a collector current ICP as shown in FIG. 9B flows through the switching element Q1.

When the switching element Q1 is turned on, the clamp diode DD1 and the switching element Q1 are turned on.
A damper current (negative direction) flows through the base-collector, and a damper current period (0.5 .mu.s) in which the damper current flows becomes a ZVS (Zero Volt Switching) region, and the switching element Q1 in the ZVS region.
Will be turned on.

By such a switching operation, the voltage V2 generated at both ends of the secondary parallel resonance capacitor C2 provided on the secondary side of the insulating converter transformer PIT.
Has a resonance waveform as shown in FIG. The voltage V3 generated at both ends of the secondary winding (N3 + N4) is shown in FIG.
The resonance waveform shown in (d) is obtained, and the secondary winding (N3 +
N4), an output current I3 flows as shown in FIG. The voltage V5 generated at both ends of the secondary winding N5 is
A resonance waveform as shown in FIG.

On the other hand, when the total load power is 100 W, the switching frequency of the switching element Q1 is controlled to, for example, 100 kHz, and the ON / OFF period TON / TOFF of the switching element Q1 is 6 μs / 4 μs.
Becomes In this case, a resonance voltage V1 as shown in FIG. 9G is generated at both ends of the primary side parallel resonance capacitor Cr, and a collector current ICP as shown in FIG. 9H flows through the switching element Q1. .

Also in this case, the voltage V2 generated across the secondary side parallel resonance capacitor C2 by the switching operation of the switching element Q1 has a resonance waveform as shown in FIG. N3 + N4) has a resonance waveform as shown in FIG. 9 (j), and a current I3 as shown in FIG. 9 (e) flows from the winding end of the secondary winding N3. The voltage V5 generated at both ends of the secondary winding N5 has a resonance waveform as shown in FIG.

[0030]

By the way, in the power supply circuit shown in FIG. 8, a voltage regulation circuit for obtaining a DC output voltage EO4 to EO6 whose voltage variation is regulated within ± 2% is used. Terminal series regulator IC-1, IC-
2, and a DC-DC converter including a chopper regulator IC-3, so that these regulators IC-1, IC-3, and DC
-Power loss occurs in the DC converter 11.

For example, to obtain the DC output voltage EO4,
In the terminal series regulator IC-1, about 3 W of power loss occurs, and in the three-terminal series regulator IC-2 for obtaining the DC output voltage EO5, about 2.3 W
Power loss occurs. Further, the DC-DC converter 11 for obtaining the DC output voltage EO6 has a DC-DC power conversion efficiency of about 90%, so that a power loss of about 1.2 W occurs. Therefore, when obtaining the DC output voltages EO4 to EO6 in the power supply circuit shown in FIG. 8, a power loss of about 6.5 W occurs in total.

Also, a three-terminal series regulator IC-
1, it is necessary to attach a heat sink to the IC-2, and the DC-DC converter 11 includes a ferrite bead inductor FB, as a countermeasure component for suppressing the switching noise generated by the switching operation,
Since it is necessary to provide the ceramic capacitor Cn, there is also a disadvantage that the cost of parts increases as the number of parts increases.

[0033]

Therefore, a switching power supply circuit according to the present invention is configured as follows in consideration of the above-mentioned problems. That is, a switching unit formed with a switching element for intermittently outputting the input DC input voltage, and a primary-side parallel resonance circuit that makes the operation of the switching unit a voltage resonance type are formed. A primary side parallel resonance capacitor is provided, and is provided for transmitting an output of the primary side to the secondary side. A primary side winding is wound on the primary side, and at least a first secondary side is wound on the secondary side. A secondary winding having a winding portion and a second secondary winding portion formed so as to wind up with respect to the first secondary winding is wound, and the primary winding is wound. An insulated converter transformer that provides the required degree of coupling, which is loosely coupled between the wire and the secondary winding, and a secondary parallel resonance capacitor connected in parallel to the secondary winding Secondary resonance circuit formed by the A first DC output voltage generating means formed including a side parallel resonance circuit and configured to obtain a DC output voltage by performing a half-wave rectification operation on an alternating voltage obtained from a secondary winding; In accordance with the DC output voltage level, the switching frequency of the switching element is variably controlled, and the OFF period in the switching cycle is fixed, and the ON time is varied to drive the switching element. A first constant voltage control unit configured to perform voltage control. Then, a half-wave rectifier circuit that performs a half-wave rectification operation on each of the alternating voltages obtained from the second secondary winding is provided, and is configured to obtain second, third, and fourth DC output voltages. It is inserted between the second DC output voltage generating means, the second secondary winding, and the half-wave rectifier circuits provided for obtaining the second, third, and fourth DC output voltages, respectively. A control winding comprising controlled windings and control windings for controlling the inductances of the controlled windings, respectively, according to second, third and fourth DC output voltage levels;
There is provided a second constant voltage control means for performing the constant voltage control of the second, third, and fourth DC output voltages by variably controlling the inductance of the controlled winding.

That is, according to the present invention, when obtaining the second to fourth DC output voltages from the alternating voltage generated in the second secondary winding of the insulated converter transformer, the second transformer constituted by the control transformer is used. The constant voltage control means makes the second to fourth DC output voltages constant, thereby reducing power loss in the switching power supply circuit.

[0035]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure has a configuration as a complex resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. A DC input voltage Ei obtained by smoothing an input voltage input from a commercial AC power supply (not shown) through a bridge rectifier circuit by a smoothing capacitor Ci is input to the power supply circuit illustrated in FIG.

The switching converter which inputs and outputs the DC input voltage Ei is a single switching element Q
1 and a voltage resonance type converter that performs a switching operation in a so-called single-ended manner by a self-excited system. In this case, the switching element Q1
A high breakdown voltage bipolar transistor (BJT; junction transistor) is used.

The base of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via the base current limiting resistor RB and the starting resistor RS, and the emitter is grounded to the primary side ground. Between the base of the switching element Q1 and the primary side ground, a driving winding NB, a resonance capacitor CB,
A series resonant circuit for driving self-excited oscillation, which is composed of a series connection circuit of base current limiting resistors RB, is connected. Further, the base of the switching element Q1 and the negative electrode (1
The clamp diode DD1 inserted between the secondary side ground) forms a path for a clamp current flowing when the switching element Q1 is turned off. The collector of the switching element Q1 is connected to one end of a primary winding N1 formed on the primary side of the insulating converter transformer PIT, and the emitter is grounded.

A primary parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. The primary side parallel resonance capacitor Cr forms a primary side parallel resonance circuit of the voltage resonance type converter by its own capacitance and the leakage inductance L1 of the primary side winding N1. Although the detailed description is omitted here, when the switching element Q1 is turned off, the operation of the primary side parallel resonance circuit causes
Voltage V generated at both ends of primary side resonance capacitor Cr
1 is actually a sinusoidal pulse waveform to obtain a voltage resonance type operation.

The orthogonal control transformer PRT-1 is a saturable reactor in which the resonance current detection winding ND, the drive winding NB, and the control winding NC1 are wound, and the switching element Q1
And for constant voltage control. Although not shown, the orthogonal control transformer PRT-1 has a three-dimensional structure by joining the ends of two magnetic legs of two double U-shaped cores having four magnetic legs. Form the core. Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction.
It is configured to be wound in a direction orthogonal to the resonance current detection winding ND and the drive winding NB.

In this case, the orthogonal control transformer PRT-1
In the above, the resonance current detecting winding ND is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1, so that the switching output of the switching element Q1 passes through the primary winding N1. And transmitted to the resonance current detection winding ND. Then, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is generated in the drive winding NB. This drive voltage is output from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. This allows
The switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

Insulation converter transformer (Power Isolation
Transformer) PIT transmits the switching output of switching element Q1 to the secondary side. As shown in FIG. 6, the structure of the insulating converter transformer PIT includes an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other.
The primary winding N1 and the secondary winding N2 are wound around the center magnetic leg of the EE type core using the split bobbin B, respectively. A gap G is formed for the center magnetic leg as shown in the figure. Thereby, loose coupling with a required coupling coefficient is obtained. Gap G is the center magnetic leg of E-shaped cores CR1 and CR2.
It can be formed by making it shorter than the outer magnetic leg of the book. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

Incidentally, the insulation converter transformer PIT
Of the secondary winding includes a primary winding N1 and a secondary winding N2.
Of the primary winding N1 and the secondary winding N1 by the polarity (winding direction) of the rectifier diode D0 and the polarity of the alternating voltage excited in the secondary winding.
2 with respect to the mutual inductance M with the inductance L2, the operation mode of + M (polarization mode; forward operation) and the operation mode of -M (depolarization mode;
Flyback operation). For example, FIG.
Mutual inductance becomes + M when equivalent to the circuit shown in FIG. 7A, and becomes −M when equivalent to the circuit shown in FIG. 7B.

In the power supply circuit shown in FIG. 1, the rectifier diode DO1 is connected during the period in which the polarity of the primary winding N1 and the secondary windings N2, N5A, N5B of the insulating converter transformer PIT are in the + M operation mode. The charging operation of the smoothing capacitors CO1 to CO3 is performed via .about.DO3.

As shown, a secondary winding N2 serving as a first secondary winding and a secondary winding serving as a second secondary winding are provided on the secondary side of the insulating converter transformer PIT. The secondary winding is formed by winding up N5A and N5B.
In this case, as shown, the secondary winding N5A and the secondary winding N
5B is connected to the secondary side ground, and a secondary side parallel resonance capacitor C2 is connected between the secondary side ground and the winding end of the secondary winding N2. Have been. That is, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding (N2 + N5A).

In this case, the insulation converter transformer PIT
On the secondary side, a secondary parallel resonance circuit is formed by the leakage inductance (L2 + L5A) of the secondary winding (N2 + N5A) and the capacitance of the secondary parallel resonance capacitor C2. Thereby, a voltage resonance operation is obtained on the secondary side of the insulating converter transformer PIT, and the alternating voltage induced on the secondary side of the insulating converter transformer PIT has a resonance voltage waveform.

The secondary winding (N2 + N5A) is provided with a half-wave rectifying / smoothing circuit composed of a rectifying diode DO1 and a smoothing capacitor CO1, and this half-wave rectifying / smoothing circuit outputs a DC output signal for horizontal deflection of a television receiver. Voltage EO1 (135V)
I'm trying to get The secondary winding N5A is provided with a half-wave rectifying / smoothing circuit including a rectifying diode DO2 and a smoothing capacitor CO2, and obtains a DC output voltage EO2 (15 V) for vertical deflection from the half-wave rectifying / smoothing circuit. I have to.

Further, the secondary winding N5B is provided with a half-wave rectifying / smoothing circuit including a rectifier diode DO3 and a smoothing capacitor CO3. In this case, the secondary winding N5B is connected to the cathode of the rectifier diode DO3 at the winding start end and to the negative electrode of the smoothing capacitor CO3 at the anode. Output voltage EO3 (−1)
5V). That is, on the secondary side of the insulating converter transformer PIT, the secondary windings N5A, N5A
DC output voltages EO2 and E03 (± 15 V) for vertical deflection are obtained from the induced voltage induced in 5B. In this case, the number of turns of the secondary windings N5A and N5B is the same.

That is, in the power supply circuit shown in FIG. 1, a primary side parallel resonance circuit for providing a voltage resonance type switching operation is provided on the primary side, and a secondary side resonance circuit for obtaining the voltage resonance operation is provided on the secondary side. A composite resonance type switching converter provided with a side parallel resonance circuit is configured.

As described above with reference to FIG. 6, the structure of such a composite resonance type switching converter has a gap G with respect to the insulated converter transformer PIT.
Is formed and loose coupling is performed by a required coupling coefficient, thereby achieving a state in which the state is hardly saturated. For example, the insulation converter transformer P
If no gap G is provided for IT, the isolated converter transformer PIT is used during flyback operation.
Is likely to become saturated and the operation becomes abnormal, and it is difficult to expect that the rectification operation on the secondary side is performed properly.

The above-described DC output voltage EO1 is supplied to the control circuit 1
Is also branched and input. The control circuit 1 is composed of, for example, an error amplifier and the like, and has a DC output voltage EO2 (1
5V) as the operating voltage and the insulation converter transformer PI
By changing the level of the control current (DC current) flowing through the control winding NC1 of the orthogonal control transformer PRT-1 according to the level change of the DC output voltage EO1 output from the secondary side of T, the orthogonal control is performed. The inductance LB of the drive winding NB wound around the transformer PRT-1 is variably controlled. As a result, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes, and the switching element Q1
This is the operation of changing the switching frequency of 1. By this operation, for example, the DC output voltages EO1 to EO3 output from the secondary side of the insulating converter transformer PIT are stabilized.

When the orthogonal control transformer PRT-1 for variably controlling the inductance LB of the drive winding NB is provided as in the power supply circuit of the present embodiment shown in FIG. The period TOFF during which the element Q1 is off is kept constant, and the period TON during which the element Q1 is on is variably controlled. In other words, in the power supply circuit shown in FIG. 1, as the constant voltage control operation, the switching frequency is variably controlled to perform the resonance impedance control on the switching output, and at the same time, the conduction angle control (PWM) of the switching element Q1 in the switching cycle. That is, the operation as the composite control method for performing the control is performed.

Further, the power supply circuit shown in FIG. 1 is provided with three sets of orthogonal control transformers PRT-2, PRT-3 and PRT-4 each constituted by a ferrite transformer. In this case, each of the orthogonal control transformers PRT-2〜
The PRT-4 includes controlled windings NR2, NR3, NR4, respectively.
And a saturable reactor wound with control windings NC2, NC3, and NC4, and DC output voltages EO4, EO5, and EO6, which will be described later.
Is performed.

These orthogonal control transformers PRT-2 to PRT-P
As a structure of RT-4, as shown in FIG.
Two double U-shaped cores 50a, 5 having two magnetic legs
The three-dimensional core (ferrite core) 50 is formed by joining the ends of the magnetic legs 0b to each other. Then, a controlled winding NR is wound around predetermined two magnetic legs of the three-dimensional core 50, and a control winding NC is wound in a direction orthogonal to the controlled winding NR. It is constituted by doing.
In this case, a gap G is provided at a portion where the U-shaped cores 50a and 50b face each other.

FIG. 3B is a diagram showing an example of the inductance superposition characteristics of the orthogonal control transformers PRT-2 to PRT-4 shown in FIG. 3A. Note that FIG.
As an example of the inductance superposition characteristic shown in (b),
The U-shaped cores 50a and 50b shown in FIG. 3A have a magnetic core cross-sectional area of 4 × 4 mm, a controlled winding NR = 6T (turn),
It is assumed that the control winding NC is 1000 T and the gap G is 75 μm.

In this case, each orthogonal control transformer PRT2
As shown, PRT-4 flows through the controlled winding NR by changing the inductance LR of the controlled winding NR in accordance with the level of the control current (DC current) flowing through the control winding NC. Variable control of the current I4 is performed.

Each of the orthogonal control transformers PRT2 to PRT-
One end of each of the controlled windings NC2 to NC4 provided in the winding 4 is connected to the winding start end of the secondary winding N5B of the insulating converter transformer PIT. And the orthogonal control transformer PRT-2
The other end of the controlled winding NC2 is connected to the anode of a rectifier diode DO4 composed of, for example, a Schottky diode, and a half-wave rectifying and smoothing circuit composed of the rectifier diode DO4 and the smoothing capacitor CO4 outputs a DC output voltage EO4 (second DC output voltage). Similarly, by connecting the other end of the controlled winding NC3 of the orthogonal control transformer PRT-3 to the anode of a rectifier diode DO5 such as a Schottky diode, a half-wave rectifier composed of the rectifier diode DO5 and the smoothing capacitor CO5 is provided. A DC output voltage EO5 (third DC output voltage) is obtained from the smoothing circuit. The other end of the controlled winding NC4 of the orthogonal control transformer PRT-4 is connected to a rectifying diode DO6 such as a Schottky diode.
Connected to the anode of this rectifier diode DO6
A DC output voltage EO6 (fourth DC output voltage) is obtained from a half-wave rectifying / smoothing circuit composed of a rectifying / smoothing capacitor C06.

The DC output voltages EO4 to EO6
Are also branched and input to the control circuits 2, 3, and 4, respectively. The control circuits 2 to 4 are also constituted by an error amplifier such as a shunt regulator for which temperature compensation or the like is performed, and the DC output voltage EO2 is input to each of the control circuits 2 to 4 as an operating voltage. .

In the control circuit 2, the DC output voltage EO4
, The orthogonal control transformer PRT-2
Variably controls the level of the control current (DC current) flowing through the control winding NC2 of the control winding NC2 to control the inductance L of the controlled winding NR2.
R2 is made variable. As a result, the current I4 flowing through the rectifier diode DO4 via the controlled winding NR2 of the orthogonal control transformer PRT-2 is controlled to make the voltage level of the DC output voltage EO4 constant within 9V ± 0.18V. Like that.

Similarly, in the control circuits 3 and 4, the control windings N of the orthogonal control transformers PRT-3 and PRT-4 are changed according to the level changes of the DC output voltages EO5 and EO6, respectively.
The inductances LR3 and LR4 of the controlled windings NR3 and NR4 are varied by variably controlling the level of the control current (DC current) flowing through C3 and NC4. As a result, the current flowing through the controlled windings NR3 and NR4 of the respective orthogonal control transformers PRT-2 and PRT-3 is controlled, and the DC output voltage E
O5 is set to a constant voltage within 5V ± 0.1V, and the DC output voltage EO6 is set to a constant voltage within 3.3V ± 0.07V.

Each of the orthogonal control transformers PRT2 to PRT2
RC snubber circuits 5a, 5b and 5c are provided between the anodes of the rectifier diodes DO4 to DO6 to which the controlled windings NR2 to NR4 of RT-4 are connected and the secondary side ground, respectively.

Each of the RC snubber circuits 5a to 5c has a capacitor CS2, CS3, C3,
S4 and resistors RS2, RS3, RS4 are connected in series to suppress the high-frequency ringing current contained in the current flowing through each of the rectifier diodes DO4 to DO6. This is because, for example, the rectifier diodes DO4 to DO6 are in a conductive state, and the current flows through the controlled windings NR2 to NR4 of the orthogonal control transformers PRT-2 to PRT-4. When DO4 to DO6 become non-conductive,
The rectifier diode D4 is formed by the capacitance of the depletion layer between the anode electrode and the cathode electrode of each of the rectifier diodes DO4 to DO6 and the inductances LR2 to LR4 of the controlled windings NR2 to NR4.
A high frequency ringing current is generated in O4 to DO6. Such a high-frequency ringing current exceeds the breakdown voltage of the rectifier diodes DO4 to DO6. Therefore, in the power supply circuit shown in FIG. 1, high frequency ringing current is suppressed by providing RC snubber circuits 5a to 5c to the anodes of the rectifier diodes DO4 to DO6.

According to experiments, when the power supply circuit shown in FIG. 1 is actually constructed, the secondary side parallel resonance capacitor C2 =
0.01 μF, secondary winding N2 of insulating converter transformer PIT = 40T, secondary windings N5A and N5B = 5T, controlled winding NR2 of orthogonal type control transformers PRT-2 to PRT-4 =
8T, NR3 = 10T, NR4 = 12T, capacitor CS2 ~
CS4 = 4700PF and resistors RS2 to RS4 = 680Ω are selected.

In the power supply circuit shown in FIG. 1, a DC output voltage EO4 of 9 V is used as an operating voltage for the analog IC.
However, it is also possible to obtain a DC output voltage EO4 of 12 V for an analog IC by providing an active clamp circuit instead of the DC output voltage EO4 of 9 V. Similarly, it is also possible to obtain an operating voltage of 2.5 V for a digital IC.

Here, as an example of the operation waveforms of the switching power supply circuit shown in FIG. 1, the operation waveforms in the case where the above components are used are shown in FIG. This figure 2
In (a) to (e), the total load power of the DC output voltages EO1 and EO2 is set to 200 W after making the voltage fluctuations of the DC output voltages EO4 to EO6 within ± 2%. Operation waveforms under the conditions are shown, and FIGS. 2F to 2J show operation waveforms under the condition that the total load power of the DC output voltages EO1 and EO2 is 100 W.

The total load power of the power supply circuit shown in FIG.
When the power is set to 0 W, the switching frequency of the switching element Q1 is controlled to be, for example, 71.4 kHz, and the ON / OFF period TON / TOFF of the switching element Q1 is controlled.
Is 10 μs / 4 μs. Then, the on / off operation of the switching element Q1 causes the parallel resonance capacitor C
The resonance voltage V1 generated at both ends of r is shown in FIG. 2 (a), and a period TOFF during which the switching element Q1 is turned off.
Thus, a sinusoidal pulse waveform having a peak voltage of 600 VP is obtained. At this time, a collector current ICP as shown in FIG. 2B flows through the switching element Q1.

When the switching element Q1 is turned on, the clamp diode DD1 and the switching element Q1 are turned on.
, A damper current (negative direction) flows through the base-collector. The damper current period (2 μs) during which the damper current flows becomes the ZVS region, and the switching element Q1 is turned on in the ZVS region.

By such a switching operation, the secondary winding N5B of the isolated converter transformer PIT is connected to FIG.
2C, a voltage V4 as shown in FIG. 2 (d) is generated at the anode of the rectifier diode DO4, and a current I4 flowing through the rectifier diode DO4 is shown in FIG. The resonance waveform becomes as shown in e).

On the other hand, when the total load power is 100 W, the switching frequency of the switching element Q1 is controlled to be, for example, 100 kHz, and the ON / OFF period TON / TOFF of the switching element Q1 is 6 μs / 4 μs.
Becomes In this case, a resonance voltage V1 as shown in FIG. 2F is generated at both ends of the parallel resonance capacitor Cr, and a collector current ICP as shown in FIG. 2G flows through the switching element Q1.

Also in this case, when the switching element Q1 is turned on, a damper current (negative direction) flows through the clamp diode DD1 and the base-collector of the switching element Q1, and the damper current period (1.5 μs) is Z.
In the VS region, the switching element Q1 is turned on.

By such switching operation, the secondary winding N5B of the isolated converter transformer PIT is connected to the secondary winding N5B of FIG.
2H, a resonance voltage V5 as shown in FIG. 2 (i) is generated at the anode of the rectifier diode DO4, and a current I4 flowing through the rectifier diode DO4 is changed as shown in FIG. A resonance waveform as shown in FIG.

FIGS. 2A to 2E and FIG. 2F
As can be seen from the operation waveforms shown in (j) to (j), in the power supply circuit shown in FIG.
When the switching frequency fluctuates to 0 W, the switching frequency of the switching element Q1 varies from 71.4 kHz to 100 kHz. However, in varying the switching frequency, the on-period of the switching element Q1 is varied while the off-period is kept constant. It is understood from the control that the switching operation of the switching element Q1 is controlled by the complex control method. Further, it can be seen that the operation waveform of each part is a resonance waveform.

When the operation waveform of the conventional power supply circuit shown in FIG. 9B is compared with the operation waveform of the power supply circuit of the present embodiment shown in FIG. 2B, the maximum load power ( 20
0W), the ZVS region in the damper current period in which the damper current flows through the switching element Q1 is 0.5 μS
It can be seen from FIG.

As described above, the switching power supply circuit according to the present embodiment shown in FIG. 1 constitutes a switching converter as a complex resonance type, and the voltage fluctuation from the secondary side of the isolated converter transformer PIT. Is ± 2%
In order to obtain DC output voltages EO4 to EO6 within the range, orthogonal control transformers PRT-2 to PRT-4 are provided. Then, based on the level changes of the DC output voltages EO4 to EO6, the quadrature control transformers PRT-2 to PRT-2 to PR
By variably controlling the inductances LR2 to LR4 of the controlled windings NR2 to NR4 provided at T-4, the DC output voltages EO4 to EO6 are made constant.

For this reason, the power loss generated when obtaining the DC output voltages EO4 to EO6 at constant voltage from the power supply circuit shown in FIG. 1 is mainly caused by the orthogonal control transformers PRT-2 to PRT-2.
The control power loss is T-4. For example, at the time of maximum load power (at 200 W), each orthogonal control transformer P
Since the DC control currents IC2 to IC4 flowing through the control windings NC2 to NC4 of the RT-2 to PRT-4 are about 40 mA,
The control power loss in each of the orthogonal control transformers PRT-2 to PRT-4 is about 0.6 W, and the power supply circuit shown in FIG. The control power loss is about 1.8W.

The orthogonal type control transformers PRT-2 to PRT-P
Since the number of windings of the controlled windings NR2 to NR4 of the RT-4 is small, the power loss in each of the controlled windings NR2 to NR4 can be neglected. Since the iron loss of the ferrite cores provided in the PRT-2 to PRT-4 is slightly added, and power loss occurs in the resistors RS2 to RS4 forming the RC snubber circuits 5a to 5c, the power supply shown in FIG. The total power loss that occurs when obtaining the constant-voltage DC output voltages EO4 to EO6 from the circuit is about 2.4 W.

On the other hand, the power loss that occurs when obtaining the DC output voltages EO4 to EO6 at constant voltage from the power supply circuit shown in FIG. 8 is about 6.5 W, as described above.
Therefore, comparing the power loss in the conventional power supply circuit shown in FIG. 8 with the power loss in the power supply circuit shown in FIG. 1, the power supply circuit shown in FIG. The loss can be reduced. Therefore, in the power supply circuit shown in FIG. 1, the AC input power can be reduced by about 4.5 W as compared with the conventional power supply circuit shown in FIG. 8, thereby achieving energy saving.

Further, in the power supply circuit shown in FIG. 1, like the conventional power supply circuit shown in FIG.
Since a three-terminal series regulator for obtaining O5 is not required, a heat sink attached to the three-terminal series regulator is not required.

In the conventional power supply circuit shown in FIG. 8, the DC-DC converter 11 for obtaining the DC output voltage EO6 is used.
Since the operating waveform is a rectangular waveform, switching noise is generated along with the switching operation.Therefore, a countermeasure component for suppressing the switching noise and a π-type filter circuit for removing a high-frequency ripple voltage are provided. It had to be provided. On the other hand, in the power supply circuit shown in FIG. 1, the operation waveform of each unit becomes a resonance waveform, and the operation waveform of each unit becomes smooth, so that the switching noise accompanying the switching operation is suppressed and the switching noise is suppressed. And a π-type filter circuit for removing high-frequency ripple voltage are not required. For example, three sets of smoothing electrolytic capacitors are required in the conventional power supply circuit shown in FIG. Will be done. Therefore, in the power supply circuit according to the present embodiment shown in FIG.
As a result, the number of parts can be reduced, so that the cost of parts can be reduced.

Further, in the power supply circuit according to the present embodiment, at the time of the maximum load power (200 W), Z in the damper current period in which the damper current flows through switching element Q1.
The VS area has been expanded from the conventional 0.5 μS to 2 μS. This means that the stable operation region of the switching element Q1 is expanded with respect to fluctuations in the AC input voltage and the load power. Therefore, when the power supply circuit of the present embodiment shown in FIG. 1 is compared with the conventional power supply circuit shown in FIG. 8, the power supply circuit of the present embodiment is more effective against fluctuations in AC input voltage and load power. The control range can be expanded.

Further, the power supply circuit of the present invention is not limited to the circuit configuration shown in FIG. FIG. 4 is a diagram illustrating a secondary-side configuration of a switching power supply circuit according to a second embodiment of the present invention. The same parts as those of the power supply circuit shown in FIG. The configuration of the primary circuit of the power supply circuit shown in FIG. 4 is the same as that of the self-excited type voltage resonance converter shown in FIG.

In the power supply circuit shown in FIG. 4, one end of each of the controlled windings NR2 to NR4 forming each of the orthogonal control transformers PRT-2 to PRT-4 has a winding end of the secondary winding N5A of the insulating converter transformer PIT. This is different from the switching power supply circuit shown in FIG. That is, in the power supply circuit shown in FIG. 4, the resonance voltage V3 generated in the secondary winding N5A is changed by each orthogonal control transformer PRT-
2 to PRT-4 to output to the rectifier diodes DO4 to DO6, so that both ends of the smoothing capacitors CO4 to CO6
The constant voltage DC output voltages EO4 to EO6 are obtained.

According to an experiment, when the power supply circuit shown in FIG. 4 is actually constructed, the secondary side parallel resonance capacitor C
2 = 0.01 μF, the secondary winding N2 of the insulating converter transformer PIT = 40T, the secondary windings N5A and N5B = 5T, and the controlled winding N of the orthogonal control transformers PRT-2 to PRT-4.
R2 = 6T, NR3 = 8T, NR4 = 10T, capacitor CS2
.About.CS4 = 4700 PF and resistors RS2 to RS4 = 470.OMEGA.

Here, as an example of the operation waveform of the switching power supply circuit shown in FIG. 4, an operation waveform in the case where the above-mentioned components are used is shown in FIG. This figure 5
(A) to (d) show the operation waveforms under the condition that the total load power of the DC output voltage EO1 and the DC output voltage EO2 is 200 W after the DC output voltages EO4, EO5, and EO6 are made constant. 5 (e) to 5 (h) show the DC output voltage E
The operation waveform under the condition that the total load power of O1 and the DC output voltage EO2 is 100 W is shown.

The total load power of the power supply circuit shown in FIG.
When it is set to 0 W, a resonance voltage V2 as shown in FIG. 5A is generated at both ends of a secondary parallel resonance capacitor C2 provided on the secondary side of the insulating converter transformer PIT. A resonance voltage V3 as shown in FIG. 5B is obtained from the next winding N5A. A voltage V4 as shown in FIG. 5C is generated at the anode of the rectifier diode DO4, which is the other end of the controlled winding NR2 of the orthogonal control transformer PRT-2, and flows through the rectifier diode DO4. The current I4 has a resonance waveform as shown in FIG.

On the other hand, when the total load power is 100 W, both ends of the secondary side parallel resonance capacitor C 2
5E, a resonance voltage V2 as shown in FIG. 5F is generated from the secondary winding N5A.
3 is obtained. And the orthogonal control transformer PRT-2
Rectifier diode DO4 at the other end of the controlled winding NR2
A voltage V4 as shown in FIG. 5 (g) is generated at the anode and a current I4 flowing through the rectifier diode DO4 is
A resonance waveform as shown in FIG.

Also in this case, FIGS. 5 (a) to 5 (d)
5 (e) to 5 (h), the periods of the resonance voltage V2 obtained from the secondary side of the isolated converter transformer PIT are 6 μs / 8 μs and 6 μs, respectively.
Since μs / 5 μs, it can be seen that the switching operation of the switching element Q1 is also controlled by the complex control method in this case. Further, it can be seen that the operation waveform of each part is a resonance waveform.

Therefore, in the power supply circuit shown in FIG. 4, similarly to the power supply circuit shown in FIG. 1, a switching converter of a complex resonance type is formed, and then the secondary side of the insulating converter transformer PIT is provided. By providing the orthogonal control transformers PRT-2 to PRT-4, the DC output voltages EO4 to EO6 can be made constant. Therefore, also in this case, the total power loss due to the constant voltage of each of the DC output voltages EO4 to EO6 from the power supply circuit shown in FIG.
4 W, which is approximately 4.1 W lower than the power loss of the conventional power supply circuit shown in FIG. Therefore, AC input power can be reduced by about 4.5 W, and energy can be saved accordingly.

Also, in this case, a three-terminal series regulator is not required, so that a heat sink attached to the three-terminal series regulator and a countermeasure component for suppressing switching noise and high-frequency ripple voltage are not required. Accordingly, the number of parts can be reduced, and the cost of parts can be reduced.

Further, although not shown in the waveform diagram of FIG. 5, also in this case, the ZVS of switching element Q1
Since the area is enlarged, it becomes possible to expand the control range with respect to the fluctuation of the AC input voltage and the load power as compared with the conventional power supply circuit shown in FIG.

In the above-described embodiment, the configuration of the primary circuit of the power supply circuit is described as a self-excited voltage resonance converter. However, this is merely an example, and the present invention is not limited thereto. Can be constituted by, for example, a separately excited voltage resonance converter.

Further, in this embodiment, the orthogonal control transformer PRT is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the orthogonal control transformer PRT, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, as in the case of the orthogonal control transformer, for example, 4
A three-dimensional core is formed by combining two sets of double U-shaped cores having two magnetic legs. Then, the control winding NC1 and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, one of the control winding NC1 and the drive winding NB is wound around two magnetic legs adjacent to each other among the four magnetic legs, The other winding is wound around two magnetic legs which are considered to be in a diagonal positional relationship. When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. As a result, the current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, The power loss of the switching element can be further reduced.

[0092]

As described above, the switching power supply circuit according to the present invention, when obtaining the second to fourth DC output voltages from the alternating voltage generated in the second secondary winding of the insulating converter transformer, The second to fourth DC output voltages are made constant by a second constant voltage control means constituted by a control transformer. In the case of such a configuration, since the power loss due to the constant voltage of the second to fourth DC output voltages can be reduced as compared with the conventional power supply circuit, the AC input power can also be reduced. It is possible to achieve energy saving.

Further, unlike the conventional power supply circuit, it is not necessary to provide a three-terminal series regulator or a DC-DC converter to obtain the constant second to fourth DC output voltages. A heat sink attached to the DC-DC converter becomes unnecessary, and a countermeasure component for suppressing switching noise and high-frequency ripple voltage generated in the DC-DC converter becomes unnecessary. Therefore, the number of components can be reduced as compared with the conventional power supply circuit, and the component cost can be reduced accordingly.

Furthermore, in the present invention, the ZVS region, which is a stable operation region of the switching element, can be expanded as compared with the conventional power supply circuit. It is possible to extend the control range for the fluctuation of

[Brief description of the drawings]

FIG. 1 is a diagram showing a configuration of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 3 is a diagram showing a configuration of an orthogonal control transformer provided in the power supply circuit shown in FIG. 1 and an inductance superposition characteristic thereof.

FIG. 4 is a diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention.

FIG. 5 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 6 is a cross-sectional view showing a structure of the insulating converter transformer.

FIG. 7 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 8 is a diagram showing a configuration of a conventional power supply circuit.

9 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 10 is a diagram showing a change in DC output voltage with respect to load power of the power supply circuit shown in FIG. 8;

[Explanation of symbols]

1-4 control circuit, 5a-5c RC snubber circuit, Ci
Smoothing capacitor, Cr primary side parallel resonance capacitor,
C2 secondary parallel resonance capacitor, CB resonance capacitor, CS2-CS4 capacitor, DD1 clamp diode, DO1-DO6 rectifier diode, N1 primary winding, N2
N5A N5B secondary winding, NB drive winding, NC1 to NC4
Control winding, NR2 to NR4 Controlled winding, PIT isolation converter transformer, PRT-1 to PRT-4 orthogonal control transformer, Q1 switching element, R1 R2 RS2 to
RS4 RB resistance

Claims (3)

[Claims] 1. A switching means formed with a switching element for intermittently outputting an input DC input voltage, and a primary-side parallel resonance circuit having a voltage resonance type operation of the switching means. A primary side parallel resonance capacitor provided in such a manner as to be provided for transmitting the output of the primary side to the secondary side, a primary side winding is wound on the primary side, and at least the A secondary winding having a secondary winding portion and a second secondary winding portion formed so as to be wound up with respect to the first secondary winding is wound. An insulation converter transformer configured to obtain a required degree of coupling that is loosely coupled with respect to the primary winding and the secondary winding; and a secondary parallel to the secondary winding. Formed by connecting resonant capacitors in parallel A secondary side parallel resonance circuit is formed including the secondary side parallel resonance circuit, and is configured to obtain a DC output voltage by performing a half-wave rectification operation on an alternating voltage obtained from the secondary side winding. The first DC output voltage generating means, the switching frequency of the switching element is variably controlled in accordance with the DC output voltage level, and the ON period is variable while the OFF period in the switching cycle is constant. The first constant voltage control means for performing the constant voltage control by switching driving of the switching element as described above, and the alternating voltage obtained from the second secondary winding, each of which is a half-wave. A second DC output voltage generating means provided with a half-wave rectifier circuit for performing a rectifying operation, and configured to obtain second, third, and fourth DC output voltages; and the second secondary winding And a controlled winding inserted between a half-wave rectifier circuit provided to obtain the second, third, and fourth DC output voltages, and controls the inductance of the controlled winding. And a control transformer comprising a control winding that controls the inductance of the controlled winding according to the second, third, and fourth DC output voltage levels. And a second constant voltage control means for performing constant voltage control of the third and fourth DC output voltages. 2. In order to obtain the second, third, and fourth DC output voltages, an anode of a rectifier diode forming a half-wave rectifier circuit and a secondary-side ground are provided between the anode and the rectifier diode. The switching power supply circuit according to claim 1, wherein a snubber circuit is connected to each of the switching power supply circuits. 3. The switching power supply circuit according to claim 1, wherein said second secondary winding is a winding for obtaining a vertical deflection voltage of a television receiver.