EP3183891B1 - Calcule de coefficients d'un filtre rif pour un filtre d'un dispositif de beamforming - Google Patents

Calcule de coefficients d'un filtre rif pour un filtre d'un dispositif de beamforming Download PDF

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EP3183891B1
EP3183891B1 EP15753373.8A EP15753373A EP3183891B1 EP 3183891 B1 EP3183891 B1 EP 3183891B1 EP 15753373 A EP15753373 A EP 15753373A EP 3183891 B1 EP3183891 B1 EP 3183891B1
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frequency
filters
forming
filter
calculation means
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EP3183891A1 (fr
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Andreas Franck
Christoph SLADECZEK
Albert ZHYKHAR
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Fraunhofer Gesellschaft zur Forderung der Angewandten Forschung eV
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Fraunhofer Gesellschaft zur Forderung der Angewandten Forschung eV
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R1/00—Details of transducers, loudspeakers or microphones
    • H04R1/20—Arrangements for obtaining desired frequency or directional characteristics
    • H04R1/32—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only
    • H04R1/40—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers
    • H04R1/403—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers loud-speakers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R1/00—Details of transducers, loudspeakers or microphones
    • H04R1/20—Arrangements for obtaining desired frequency or directional characteristics
    • H04R1/32—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only
    • H04R1/40—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers
    • H04R1/406—Arrangements for obtaining desired frequency or directional characteristics for obtaining desired directional characteristic only by combining a number of identical transducers microphones
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R3/00—Circuits for transducers, loudspeakers or microphones
    • H04R3/005—Circuits for transducers, loudspeakers or microphones for combining the signals of two or more microphones
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R2201/00—Details of transducers, loudspeakers or microphones covered by H04R1/00 but not provided for in any of its subgroups
    • H04R2201/40—Details of arrangements for obtaining desired directional characteristic by combining a number of identical transducers covered by H04R1/40 but not provided for in any of its subgroups
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R2430/00—Signal processing covered by H04R, not provided for in its groups
    • H04R2430/20—Processing of the output signals of the acoustic transducers of an array for obtaining a desired directivity characteristic
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R3/00—Circuits for transducers, loudspeakers or microphones
    • H04R3/12—Circuits for transducers, loudspeakers or microphones for distributing signals to two or more loudspeakers

Definitions

  • the present invention is concerned with the calculation of FIR filter coefficients for beamforming filters of a transducer array, such as e.g. an array of microphones or speakers.
  • Beamforming technologies such as those used in the audio field, specify - in the case of a microphone array for the evaluation of the individual signals of the microphones or in the case of a loudspeaker array for the reproduction of the signals of the individual speakers - as the signals of an individual Filtering with a respective discrete-time filter are subjected.
  • the resulting FIR filters accurately map the given frequency response in the frequency raster given by the DFT, but the frequency response can assume arbitrary values between the raster points. This often leads to useless designs with strong oscillations of the resulting frequency response.
  • the length of the FIR filter automatically results from the resolution of the frequency response specification (and vice versa).
  • Filters created using Frequency Sampling Design are susceptible to temporal aliasing, ie periodic convolution of impulse responses (eg [Smi11]). Additional techniques such as zero-padding the DFTs or windowing the generated FIR filters may need to be used.
  • An alternative approach is to determine the FIR coefficients in a one-step process directly in the time domain [MDK11].
  • the radiation behavior of the array for a given pattern of frequencies is represented directly as a function of the FIR coefficients of all transducers (ie loudspeakers / microphones) and formulated as a single optimization problem, by the solution of the optimal filter coefficients for all beamforming filters are determined simultaneously.
  • the problem here is the size of the optimization problem, both in terms of the number of variables to be optimized (filter length times the number of beamforming filters) and with respect to the dimension of the determination equations and, if necessary, secondary conditions.
  • the latter dimension is usually proportional to both the number of frequency raster points and the spatial resolution used to determine the desired beamformer response.
  • this method is limited to low element count arrays and very small filter orders. For example, [MSK11] microphone arrays of six elements and a filter length of 8 are used.
  • EP 1 919 251 A1 shows a beamforming filter, designed as an FIR filter and applied to data coming from an array of sensors.
  • the characteristics of the filters can be determined in the frequency domain.
  • the FIR filter coefficients are then derived from the determined characteristics.
  • the object of the present invention is to provide a concept for calculating FIR filter coefficients for the beamforming filters of a transducer array, which is more effective, such as in relation to the ratio between achieved beamforming quality and the amount of computation required. This object is solved by the subject matter of the appended independent claim.
  • One idea on which the present application is based is to have realized that the effectiveness of calculating FIR filter coefficients for beamforming filters for transducer arrays, such as arrays of microphones or loudspeakers, can be increased if the calculation is performed in two stages is, on the one hand by the calculation of frequency domain filter weights of the beamforming filters in a predetermined frequency raster, ie coefficients describing the transmission function of the beamforming filters in the frequency domain or for a respective frequency or for a sinusoidal input signal with a respective frequency to target frequency responses for the To obtain beamforming filters such that application of the beamforming filters to the array approximates a desired direction selectivity and followed by a calculation of the FIR filter coefficients for the beamforming filters, ie coefficients describing the impulse response of the beamforming filters in the time domain such that frequency responses of the beamforming filters approximate the intermediate frequency responses.
  • the two-step nature allows for independent choice of the frequency resolution resulting from the discrete Fourier transform of the Implus responses described by the FIR filter coefficients. Furthermore, specific secondary conditions can be specified both in the calculation of the beamforming drive weights in the frequency domain and in the calculation of the time domain FIR filter coefficients, in order to influence the respective calculation in a targeted manner.
  • Fig. 1 shows first an example of an array 10 of loudspeakers 12, which is to be enabled by the application of beamforming filters (BFF) 14 to have a desired directional selectivity, ie, for example, to radiate in a particular direction 16.
  • BFF beamforming filters
  • the number N of speakers 12 may be two or more.
  • the loudspeaker 12 n is connected to a common audio input 18 via its corresponding beam-forming filter 14 n .
  • the audio signal s () at the input 18 is a discrete-time audio signal consisting of a sequence of audio samples and the beamforming filters 14 n are configured as FIR filters and thus fold the audio signal with the impulse response of the respective beamforming filter 14 n as defined by the FIR filter coefficients of the respective beamforming filter 14 n .
  • h is BFF n ( i ) the filter coefficients of the FIR filter 14 n with the FIR order I BFF n or the filter length I BFF n +1 is.
  • the loudspeaker array 10 emits the audio signal at the input 18 with a desired directional selectivity, such as in the desired direction 16.
  • Fig. 1 by way of example only, the loudspeakers 12 n are arranged equidistantly in a line and the array 10 is a linear array of loudspeakers.
  • a two-dimensional arrangement of loudspeakers would also be conceivable, as well as a non-uniform distribution of the loudspeakers 12 in the array 10 and also an arrangement deviating from an arrangement along a straight line or a plane.
  • the emission direction 16 can be measured, for example, by an angular deviation of the direction 16 from a mid-perpendicular of the straight line or the surface along which the loudspeakers 12 are arranged.
  • the radiation should preferably be audible at a particular location in front of the array 10.
  • the filter coefficients h of the beamforming filters 14 n can also be selected more precisely such that the directional characteristic or directional selectivity of the array 10 not only experiences a maximum in a certain direction 16 when irradiated, but also fulfills other desired criteria, such as an angular one Beamwidth, a particular frequency response in a direction 16 maximum radiation or even a specific frequency response as far as an area that includes the direction 16 and directions around it affected.
  • Fig. 2 shows such a microphone array.
  • the microphone array of Fig. 2 is exemplarily also provided with the reference numeral 10, but in each case consists of microphones 20 1 ... 20 N together.
  • each microphone generates a received audio signal S BFF i and is connected via a respective beamforming filter 14 n to a common output node 22 for outputting the received output signal s ', so that the filtered audio signals s ' BFF s the beamforming filter 14 n additively contribute to the audio signal s'.
  • an adder 24 is connected between the outputs of the beamforming filters 14 n and the common output node 22.
  • h is BFF n again the FIR filter coefficients of the beamforming filters 14 n .
  • the microphone array 10 Fig. 2 has a desired directional or directional characteristic to primarily or exclusively absorb the sound scene from a particular direction 16, so that it is reflected in the output signal s', the direction 16 again as in the case of Fig. 1 can be defined by the angulare deviation ⁇ or in the two-dimensional case ⁇ and ⁇ from a perpendicular of the array 10 and the desired direction selectivity may possibly also be more accurate than just specifying a direction of maximum sensitivity, namely more precisely in terms of spatial dimension or frequency dimension.
  • Fig. 3 now shows an embodiment of an apparatus for calculating FIR filter coefficients for the beamforming filter of a transducer array, such as an array of microphones, as for example in Fig. 2 was shown, or speakers, such as in Fig. 1 was shown.
  • a transducer array such as an array of microphones, as for example in Fig. 2 was shown, or speakers, such as in Fig. 1 was shown.
  • the device is indicated generally at 30 and may be implemented, for example, in software executed by a computer, in which case all of the devices and modules described below may be, for example, different sections of a computer program.
  • An implementation in the form of a dedicated hardware e.g. in the form of an ASIC or in the form of a programmable logic circuit, e.g. an FPGA, however, is also possible.
  • the device 30 calculates the FIR filter coefficients 32, such as the h BFF just mentioned above n for the beamforming filters 14 n , specific to the array 10, to which the device 30 has interfaces to obtain information about the array 10 or information about the desired direction selectivity.
  • the device 30 receives converter data 34 from outside, which will be described in more detail below and for example the positions and orientations of the transducer elements, ie for example the loudspeakers or microphones, and their individual direction-selective sensitivity or emission characteristic and / or specify the frequency response.
  • Other information relates, for example, to the desired directional selectivity. For example, shows Fig.
  • the device 30 obtains data 36 which indicate the desired directional behavior of the array 10, such as a direction of maximum radiation and possibly more precise information, such as the radiation behavior or the sensitivity around the just mentioned maximum radiation / sensitivity.
  • the data 36 are supplemented, for example, by further data 38 which can be specified from the outside of the device 30 and, for example, the desired transfer characteristic or the frequency response of the array 10 in the direction of the radiation or sensitivity of the array 10, ie the frequency-dependent desired description the sensitivity or emission intensity of the array adjusted with the final FIR filter coefficients in a certain direction or directions.
  • FIR filter coefficients 32 Other information may also be provided to the device 30 for calculating the FIR filter coefficients 32, such as specifications regarding robustness to be maintained of the calculated FIR filter coefficients against deviations of the transducer data 34 from actual physical conditions of an actual array 10, these specifications in Fig. 3 40, as well as data on a frequency restriction, 42, whose exemplary meaning for the calculation will be described below and possibly related to the transducer data 34.
  • all of the information 34 to 42 used for the device 30 of FIG Fig. 3 can be specified from outside, are optional.
  • the device 30 could also be specific to a particular array setup, and it would also be possible for the device to be specific to particular settings of the other data. In the case of an input facility, this may be implemented via an input interface, such as via user input interfaces of a computer or read interfaces of a computer, such that, for example, the data is read from one or more particular files.
  • the device of Fig. 3 includes first calculating means 44 and second calculating means 46.
  • the first calculating means 44 calculates frequency-domain filter weights of the beamforming filters, ie complex-valued samples of the transfer function of the beamforming filters. They are used to set a target frequency response for the beamforming filters.
  • the first calculation device 44 calculates the frequency range drive weights in a frequency raster defined by specific, not necessarily equidistant frequencies ⁇ 1 ... ⁇ k , such that it has a transfer function H BFF n describe the beamforming filter which approximates the desired direction selectivity when applying such beamforming filters to the array 10.
  • the first calculation device uses, for example, a suitable optimization algorithm, for example a method for solving linear, quadratic or convex optimization problems.
  • the frequency raster may, for example, in accordance with requirements of the beamforming application, such as different requirements for the accuracy of the radiation specification in certain frequency ranges, or according to other requirements, such as the subsequent and below-mentioned FIR time domain design method, such as necessary sampling rate for the definition of the desired frequency response, to be selected.
  • the second calculation means 46 is for determining the FIR filter coefficients of the beamforming filters which describe the impulse response of the beamforming filters.
  • the second calculation device 46 carries out the calculation in such a way that the frequency responses of the beamforming filters, as they correspond to the FIR filter coefficients via the relationship between transfer function and impulse response, approximate the target frequency responses predetermined by the first calculation device 44.
  • the second calculation device 46 also uses optimization according to the following embodiment description, which can again be implemented as a method for solving linear, quadratic or convex optimization problems.
  • the first calculation means 44 performs the calculation by solving a first optimization problem, after which a deviation between a directional selectivity of the array, as reflected by the frequency range drive weights H BFF n ( ⁇ k ), and the desired direction selectivity that may be predetermined by the data 34 and / or 38 is minimized. As it is in Fig.
  • the first calculation means 44 may use the robustness specifications 40 as a constraint of the optimization problem, and the conversion data 34 is used to determine the relationship between the optimization variables, namely the frequency domain drive weights H BFF n ( ⁇ k ) on the one hand and the resulting directional selectivity on the other hand to set or define.
  • the second calculator 46 may also solve an optimization problem to make the calculation. After the second optimization problem on which the second calculation device 46 is based, a deviation from the target frequency responses H BFF n ( ⁇ k ) minimized in the frequency domain.
  • the FIR filter coefficients to be calculated by the second calculator 46 correspond to the impulse response, and the device 46 tries to calculate them by optimization according to the following embodiments such that the transfer functions corresponding to these impulse responses correspond to the transfer functions H BFF n ( ⁇ k ) as close as they have been calculated by the first calculating means 44. From the description below In this case, it will become clear that, advantageously, in the optimization of the calculation device 46, ancillary conditions provided specifically for this optimization can be taken into account, as specified by the data 42.
  • a target frequency response modification device 48 optionally provided between the two calculation devices 44 and 46, which optionally modifies the target frequency responses of the beamforming filters as determined by the first calculation device 44 before being used as the approximation target by the second calculation device 46 are used.
  • Various modification options are described. They are designed to not cause the effectiveness of the calculation of the FIR filter coefficients 32 by the computing device 46 or even the calculation of lower-quality FIR filter coefficients.
  • the device 30 may optionally include an optional modification device 50 for modifying the calculated FIR filter coefficients, as calculated by the second calculation device 46, so that the respective modification is taken into account.
  • Fig. 3 will also exemplarily describe a possible modular structure for the first calculation device 44 and the target frequency response modification device 48, but the respective modular design is merely exemplary.
  • the filter design process as implemented by the device 30, provides for a variety of individual interrelated measures and precautions in accordance with the embodiments described below. All in all, they enable the generation of particularly stable, robust control filters or beamforming filters. The operation of the device 30 will now be described in detail. However, some of the measures can also be left out depending on the application.
  • the transducer characteristics ie the properties of, for example, microphones or loudspeakers
  • the transducer data 34 describes the transducer characteristics that are typically derived from measurements or modeling, such as simulation.
  • the transducer data 34 may represent, for example, the directional and frequency dependent transfer function of the transducers of (in the case of loudspeakers) and to (in the case of sensors or microphones, respectively) different points in space.
  • a module 52 of the computing device 44 may perform directional characteristic interpolation, ie, interpolation of the transducer data 34 to determine the transfer function of the transducers from / to points or Allow directions that are not included under the original data 34, ie, not in the original records.
  • the converter data of the module 52 thus obtained are used in two function blocks or modules 54 and 56 of the first calculation device 44, namely a delay and sum beamformer module and an optimization module 56.
  • the delay and sum (delay and summation Beamformer module 54 calculates a delay and an amplitude weight, ie, for each transducer n , based on a desired specification for the directional behavior of the transducer array, which specifies, for example, a desired magnitude in the emission direction or emission directions, using the individual transducers of the array in the respective direction frequency-independent quantities, such as a time delay and a gain factor per converter 12 and 14.
  • the optimization 56 operates in the frequency domain.
  • the frequency range shaping coefficients already mentioned above or the frequency range control weights H BFF n ie frequency dependent quantities can be improved by the inclusion of specific converter transfer functions, especially if they deviate greatly from an ideally assumed behavior, such as a monopole characteristic.
  • transducer data 34 obtained by measurements often have pure delays, such as acoustic propagation, and a delay extraction module 58 connected between the directional characteristic interpolation module 52 and the optimization module 56 could be provided to remove common delay times of all transducer data.
  • the desired complex sound radiation which is described by the objective function, is not necessarily limited to directions.
  • Other arguments are also possible, for example, such as the desired emission along a line or across a surface / volume.
  • robustness refers to the property of showing only a relatively small deterioration of the radiation behavior in the case of deviations of the transducer array 10 or of the transmission system, eg due to deviations of the control filters from ideal behavior, positioning errors of the transducers in the array or deviations from the modeled transmission behavior
  • a measure of robustness frequently used for microphone arrays is the so-called "White Noise Gain” [BW01, MSK09], ([WNG]), which results as the quotient of the signal magnitude in the direction of incidence and the L 2 standard of the drive weights for the array .
  • This measure can also be usefully used for loudspeaker arrays [MK07], whereby here the signal magnitude in the desired emission direction assumes the role of the magnitude in the direction of incidence.
  • the calculation device 44 has another module, namely a module 58, which determines the final specification of the frequency response of the transducer array in the emission direction from the acquired reference magnitude response of the module 54 in combination with the specifications 38 for the desired frequency response or desired frequency response. That is, the starting point for the determination of the module 58 is the frequency response of the transducer array, as determined by the DSB values of the module 54, ie the frequency response that results for the transducer array with the DSB values in the corresponding direction. From this Magnitudengang starting, are made by the module 58 modifications. For example, modifications are made to the reference magnitude response, for example to equalize the frequency response.
  • the directivity of the array can be increased within certain limits (either globally or for certain frequencies) by reducing the magnitude response in the direction of emission from the reference magnitude response.
  • the use of the DSB reference design and its WNG value allows a good estimation of the robustness properties of the final design specification.
  • psychoacoustic findings flow into the frequency response determination 58.
  • the knowledge can be exploited that certain frequency ranges of a signal are more important for the perception of a sound event, and that therefore a less advantageous, because less directed, radiation in other frequency ranges can be compensated or less perceptible by a targeted increase in these frequency ranges .
  • this equalization is, for one thing, independent of the signal and is also limited to only one emission characteristic, that is to say is not based on psychoacoustic masking between different emission characteristics or audio signals.
  • optimization is then performed in module 56.
  • the beamforming filter is designed here in the frequency domain for a series of discrete frequencies ⁇ k .
  • optimization methods based on convex optimization are preferably used [M07, MSK09].
  • the optimization-based approach allows for numerous constraints that affect both the radiation achieved and the driving power can relate to. For example, a limitation on the minimum white noise gain can be set. In the same way it is possible to set maximum amounts for the control weights in order to limit the control of the individual transducers.
  • Fig. 4 The previous description of the possible implementation of the operation of the first calculation means 44 summarized once again in an illustrative manner, becomes Fig. 4 directed.
  • the starting point of the target frequency response calculation by the calculation device 44 forms the desired direction selectivity, which in Fig. 4 described with ⁇ is and is provided with the reference numeral 70.
  • the desired direction selectivity ⁇ is illustrated here by way of example as a function ⁇ dependent on the emission angle ⁇ . However, as stated above, the directionality may be defined other than angular.
  • Fig. 4 by a dashed ⁇ , the desired direction selectivity 70 can be defined in space sense, not just in one plane. Top right is in Fig. 4 indicated how the angles ⁇ and ⁇ could be defined.
  • ⁇ could depend on ⁇ .
  • this frequency response ⁇ is to be distinguished from the frequency response H n ( ⁇ k ) for the individual beamforming filters, as they are to be calculated by the calculation device 44. Both act like a filter with a transfer function as determined by the dependence on ⁇ , but the frequency response ⁇ is affected by the finally calculated frequency responses H n of the individual beamforming filters.
  • the desired direction selectivity 70 is now to be achieved with the particular transducer array.
  • Fig. 4 In the upper right corner loudspeakers were used as the elements of the array, but as already mentioned, an array of other transducers, such as microphones, is also possible.
  • the array is thus determined from certain transducer positions, transducer orientations, a transducer frequency response, wherein the frequency response may in turn be direction-dependent, and / or a directional dependence of the radiation or the sensitivity, which in turn may be frequency dependent.
  • module 54 a pair of values ⁇ n and a n , namely frequency-independent delay ⁇ and a gain value a, which is also frequency-independent, are now determined for each transducer n, so that assuming only these frequency-independent values in the BFFs the transducer n are applied, the directional selectivity ⁇ '72 results, which is direction-dependent, that is, for example, depending on ⁇ and optionally ⁇ and frequency-dependent, ie depending on ⁇ .
  • the determination in module 54 is performed so that the desired direction selectivity 70 is achieved or approximated where possible. This is only to a limited extent, of course, as per converter yes only be determined frequency independent delay and gain n.
  • ⁇ '72 now serves as a starting point for the actual desired direction selectivity 74, which will later be the basis of optimization 56. From the directional selectivity 72 one uses the foreknowledge that it is due to its DSB property is robust. The module 58 now modifies the directionality ⁇ '72 to be closer to the desire for a particular frequency dependence of directional selectivity.
  • the optimization target 74 of the optimization 56 is thus a frequency-dependent and direction-dependent directional selectivity ⁇ Ziei and the optimization 56 is carried out in such a way that it finds target frequency responses or transfer functions H n ( ⁇ k ) for the beamforming filters n .
  • Optimization goal 74 is achieved or approximated as well as possible by its use in the beamforming filters of transducer array 10, ie, a deviation according to a certain criterion is minimized
  • Optimization 56 could thus be regarded as a fine adjustment of beamforming filter transfer functions 76 which when used for the beamforming filters, to d frequency independent delays and gains are equivalent.
  • the DSB design is actually used only for the optimization target formulation and the frequency domain optimization 56 may start independently of the DSB design.
  • the DSB design is not adapted or used as a basis, thus serving merely as a template for the desired frequency response in the emission direction, ie for the optimization target definition, and the optimization algorithm 56 starts "from Scratch ", ie without knowledge of the DSB weighting.
  • the optimization 56 is carried out, may be suitably set by the application.
  • the variables to be optimized are 2 ⁇ N ⁇ K, where N is the number of converters and K is the number of frequency samples for which optimization 56 is performed.
  • the optimized target frequency responses 78, which result from the optimization 56, can be achieved by optionally additionally subjecting the optimization to additional constraints, such as, for example, ancillary conditions regarding compliance with certain robustness criteria, as specified by the data 40.
  • the optimization 56 can therefore in particular be a quadratic program with a secondary condition that does not fall below a certain robustness measure.
  • the frequency responses of the individual drive filters n result from the drive weights H n ( ⁇ k ) obtained in the optimization 56, since the weights of the filter are taken in each case.
  • These filters often contain a significant delay or delay, which manifests itself for example by the phase or group delay. This delay is a hindrance for the further processing stages, such as in particular the subsequent optimization in the second calculation means 46.
  • the optional smoothing step described below is also made more difficult or requires a significantly higher resolution of the frequency raster in the optimization 56 in the first calculation means, since the smoothing is a Determination of the continuous phase involved by a phase unwrapping. The greater the increase in the phase function contained in the frequency response, the more difficult is the correct detection and subsequent compensation of the phase jumps. This has a negative effect on the correctness of the "phase-unwrapping" algorithms.
  • the local optimization target ie target frequency response 78
  • the local optimization target ie target frequency response 78
  • the local optimization target ie target frequency response 78
  • the phase terms caused by delays are eliminated as far as possible become.
  • Further requirements of the optimization step in the calculation device 46 will be described in more detail below.
  • the causality of the resulting filters is not relevant at this stage of the design process. It is possible to work with non-causal, near zero-phase transfer functions desired frequency responses for the drive filter. The causality can be made causal again after the FIR design (by reinserting the extracted delays, possibly supplemented by additional delays).
  • Fig. 5 illustrates once again the operation of the delay adjustment module 80 of the modifier 48.
  • the starting point is the set of target frequency responses 78 to be modified, ie H n ( ⁇ k ).
  • Fig. 5 Illustratively shows the phase curve 82 of H n ( ⁇ k ). It has exemplary phase jumps 84.
  • the 2 ⁇ phase-shift-adjusted phase profile is shown at 86 and this can be approximated by a linear curve 88, such as a least square fit, where the linear portion 88 has a slope corresponding to a frequency-independent delay ⁇ ' ⁇ equivalent.
  • Fig. 5 at 90 shows the phase characteristic of the thus modified target frequency responses H ' n ( ⁇ k ).
  • the delays ⁇ ' ⁇ are reserved or stored.
  • the frequency-domain smoothing by the module 92 has the following meaning.
  • the frequency responses 78 and H ' n ( ⁇ k ) of the drive filters n generated by the optimization-based filter design typically have large fluctuations of the magnitude and also of the phase.
  • Such design specifications are difficult to implement in an FIR filter design or require a very high FIR filter order or FIR length of the beamforming filters. In the latter case, although a good agreement with the given interfaces can be achieved, however, strong overshoot phenomena frequently occur between the interpolation points ⁇ k , which deteriorate the frequency response of the resulting beamformer. Even from psychoacoustic considerations, it often does not make sense to map such narrowband fluctuations.
  • the desired frequency responses 78 of the drive filters are subjected to a smoothing algorithm. This is done, for example, for psychoacoustic considerations with a frequency-dependent window width of, for example, 1/3 octave or 1/6 octave [HN00].
  • the smoothing is carried out separately for magnitude and phase, ie a separate smoothing of the magnitude transfer function (to be more precise of the "zero-phase" frequency response (cf., for example, [Sar93, SI07]) and the continuous ( It is possible that magnitude and phase are generated and independently by a phase unwrapping algorithm in the module 92 from the complex frequency response H ⁇ ( ⁇ k ) or H ' n ( ⁇ k ) is smoothed by convolution with a frequency-dependent smoothing filter, also referred to as "window.”
  • the phase unwrapping in the module 92 may, in the case of the presence of the module 80, occur if necessary because phase unwrapping was already performed in module 80.
  • optimization methods for linear, quadratic or more generally convex compression problems can be used, and this optimization problem may be accompanied by constraints, which for example concern the course of the transfer function of the beamforming filters, ie a secondary condition that the The transfer function or the frequency range of the beamforming filters is concerned if the optimization in the calculation device 46 otherwise relates to the FIR filter coefficients of the beamforming filters which correspond to the impulse response of the beamforming filters as optimization variables.
  • the modification device 50 It is responsible for "integrating" the modification by the module 80, ie the leveling of the phase response of the target frequency responses of the beamforming filters, back into the FIR filter coefficients obtained by the optimization in the calculation means 46, if necessary, by adding a kind of delay Recombination performed, such as the zeros insertion described in more detail below, after which the FIR filter coefficients are prefixed zeros. This will be described below. Fig.
  • FIG. 6 shows by way of example by a double arrow that the FIR filter coefficients h BFF obtained by the optimization in the calculation device 46 n Beamforming filter mode, the impulse response of the respective beam-forming filter n describe and pass over an FFT or Fourier transforms in the transfer function H n ( ⁇ ) of the respective beamforming filter or correspond to her.
  • Fig. 6 shows by way of example at 96 the impulse response and at 98 exemplarily the 2 ⁇ phase-jump-corrected phase characteristic of the transfer function.
  • f 0 Hz
  • a dedicated specification of the radiation behavior is not useful, especially when modulating real converter.
  • very high frequencies such as relative to the spatial aliasing frequency of the array, usually no meaningful guidelines possible: 1) the formation of pronounced side lobes can not be prevented by a corresponding desired characteristic. 2)
  • the width of the "beam" of the desired direction of radiation decreases with increasing frequency.
  • the optimization process in the calculator 46 i. the optimization-based design of FIR filters, the introduction of frequency ranges or frequency sections for which no specifications are made, i. for which there is no desired frequency response or target frequency response, i. for which no optimization goal is specified. Such areas may be referred to as transition bands or "do not care bands".
  • transition bands or do not care bands For the considered beamforming applications, however, it can be seen that even very narrow frequency ranges without design specification or without optimization target in the optimization of the second calculation means 46 lead to uncontrolled behavior of the designed FIR filters, such as e.g. to an extremely high magnitude and to fluctuations of the beamforming filter frequency response in these frequency sections.
  • Fig. 3 the optional possibility, according to which restrictions 42 are made for the optimization target with regard to the frequency response of these frequency sections.
  • the frequency sections may be selected depending on characteristics of the converter.
  • the frequency constraint 42 is included in the optimization problem to be solved by the calculation means 46 by specifying a maximum magnitude as a constraint for the convex optimization problem: ... under the constraint that
  • X represents a discretized representation of the transition or "do not care" bands, ie those frequency segments for which there is no optimization target in the optimization in the second calculation means 46, and
  • the aim of the optimization in the second calculation device 46 is the frequencies obtained by the frequency domain design for the beamformers, which are denoted above as H n ( ⁇ k ) and H ' n ( ⁇ k ) and H " n ( ⁇ k ), respectively hereinafter referred to as the desired frequency response with the variable name ⁇ ( ⁇ ) to generate FIR filters, with which the filtering of the source signals, ie the speaker signals in the case of a loudspeaker array, as shown in FIG Fig. 1 and the microphone output signals in the case of a microphone array as shown in FIG Fig. 2 shown can be done.
  • a mathematical optimization method is used, which can be, for example, a method of convex optimization.
  • the frequency response H (w) of the designed FIR filter h (i) is determined so that ⁇ ( ⁇ ) is approximated as best as possible, ie that the error with respect to a selectable standard p becomes minimal.
  • the optimization problem can generally be represented in the following form: minimize H i ,, 0 ⁇ i ⁇ I ⁇ H ⁇ - H ⁇ ⁇ ⁇ p under the constraint that ⁇ constraint s >
  • the suffix ⁇ constraint (s)> is optional. Secondary conditions do not have to be present, but may be present, as described above with regard to the high-frequency restrictions, as an example. A single constraint is also possible. In general, these constraints represent a variety of possible constraints, including, but not limited to, the frequency response or coefficients of the FIR filter.
  • the target frequency responses for the time domain optimization in the second calculation device 46 which arise in the context of the frequency domain optimization 56 (or with modification 80 and / or 92) are generally complex-valued and have a non-trivial, in particular neither linear nor minimal-phase, frequency response.
  • the optimization problem of the above equation (2) corresponds to a filter design problem for FIR filters with arbitrary phase characteristics.
  • a large number of methods are described in the literature, for example in [PR95, KM95; KM99].
  • the delays contained in the filters ⁇ ( ⁇ ) and ⁇ ( ⁇ ) ie the linear term of the phase response which has been adjusted by 2 ⁇ phase jumps, are of particular importance.
  • the desired function ⁇ ( ⁇ ) should be adjusted so that the linear component of the phase is as close to 0 as possible. This is effectively realized by the modifications 80 and 50.
  • the modification device 50 optionally integrates the previously compensated delay components back into the drive filters.
  • the integration of the delays ⁇ ' ⁇ into the filters n is bypassed by applying the pure delays ⁇ ' n at runtime of the beamforming application be applied to the input or output signals of the control filters by means of suitable signal processing means, such as digital delay lines (English Delay Lines ). In this case it is only necessary to ensure that the impulse responses of the obtained FIR filters are causal, ie that the indices of the impulse responses begin at zero.
  • Such a modification does not require active arithmetic operations at run-time, but merely corresponds to the introduction of a constant implementation-related delay for all drive filters n. It should be ensured that this delay is constant for all drive filters of a beamformer.
  • frequency domain optimization 56 could also use a hybrid design approach.
  • an optimization-based approach for obtaining the frequency domain drive functions H m ( ⁇ k ) as described so far is combined with a design corresponding to the DSB design as calculated in the module 54, wherein the DSB Design approach is used for the high frequencies.
  • the aim is to reduce the required filter order while improving robustness. It is exploited that for high frequencies, the emission characteristics of the transducer array due to the spatial aliasing can not be completely controlled.
  • a DSB design approach is used for frequencies above a fixed fundamental frequency, such as a frequency that is relatively close to the spatial aliasing frequency of the transducer array.
  • the frequency domain specification of the complete filter is composed of two parts: the frequency responses obtained by optimization up to the cutoff frequency and the frequency responses corresponding to those of the DSB for the frequencies above it.
  • the combination of both methods is followed by the subsequent smoothing already described above and the optimization-based FIR design.
  • a critical step here is the alignment of the signal delays of both design approaches. For example, it is possible to use a least-square fitting to determine a delay offset for the DSB in such a way that the delay jumps of the individual control filters are minimized in the root mean square.
  • the hybrid design approach allows for more robust high frequency radiation, resulting in less erratic fluctuations in performance without significant loss of performance, with partially improved low frequency directivity, and consistent filter ordering.
  • the reason for this can be assumed that the degrees of freedom provided by a certain filter order are better used with the hybrid design approach for the frequency ranges in which the characteristic can be influenced, while for high frequencies in which due to spatial aliasing hard restrictions for the Suppression of unwanted radiation, less resources are spent.
  • Fig. 7 illustrates again the hybrid design approach:
  • the transfer function to be used for time domain optimization in the second calculating means 46 is composed of the obtained according to the frequency range optimization 56 transfer function, as far as a portion of lower audio frequencies is concerned 100, where the the frequency-independent pairs of values ⁇ ⁇ a ⁇ corresponding transfer functions H n be used in the higher audio frequency section 102.
  • Section 100 and section 102 can be concatenated to a cutoff frequency ⁇ cross connect to each other, for example, corresponds to the spatial aliasing frequency border of the transducer array or from the latter deviates less than 10%. It is also possible that, as indicated by dashed dots, low-frequency section and high-frequency sections 100 and 102 overlap each other.
  • the time-domain optimization transfer function finally to be used could be obtained, for example, by averaging between both transfer functions (DSB design and optimization result of FIG. 56).
  • the above embodiments thus described a possibility for the creation of a design of robust FIR filters for beamforming applications. From complex-valued frequency responses of the individual beamforming filters, FIR filters with arbitrary phase responses can be generated. Particular value of the above embodiments is that robustness properties of the beamformer can be obtained.
  • the present invention can be used in a variety of beamforming applications, e.g. in loudspeaker arrays for room-selective sonication, for generating "quiet zones" or for reproducing surround material via loudspeaker lines (soundbars).
  • the above embodiments may also be used by microphone arrays to record sound directionally selective.
  • aspects have been described in the context of a device, it will be understood that these aspects also constitute a description of the corresponding method, so that a block or a component of a device is also to be understood as a corresponding method step or as a feature of a method step. Similarly, aspects described in connection with or as a method step also provide a description of a corresponding one
  • Some or all of the method steps may be performed by a hardware device (or using a hardware device), such as a microprocessor, a programmable computer, or an electronic circuit. In some embodiments, some or more of the most important method steps may be performed by such an apparatus.
  • the inventive set of FIR filter coefficients 32 for the beamforming filters may be stored on a digital storage medium, or may be transmitted on a transmission medium such as a wireless transmission medium or a wired transmission medium such as the Internet.
  • embodiments of the invention may be implemented in hardware or in software.
  • the implementation may be performed using a digital storage medium, such as a floppy disk, a DVD, a Blu-ray Disc, a CD, a ROM, a PROM, an EPROM, an EEPROM or FLASH memory, a hard disk, or other magnetic disk or optical memory are stored on the electronically readable control signals that can cooperate with a programmable computer system or cooperate such that the respective method is performed. Therefore, the digital storage medium can be computer readable.
  • some embodiments according to the invention include a data carrier having electronically readable control signals capable of interacting with a programmable computer system such that one of the methods described herein is performed.
  • embodiments of the present invention may be implemented as a computer program product having a program code, wherein the program code is operable to perform one of the methods when the computer program product runs on a computer.
  • the program code can also be stored, for example, on a machine-readable carrier.
  • inventions include the computer program for performing any of the methods described herein, wherein the computer program is stored on a machine-readable medium.
  • an embodiment of the method according to the invention is thus a computer program which has a program code for performing one of the methods described herein when the computer program runs on a computer.
  • a further embodiment of the inventive method is thus a data carrier (or a digital storage medium or a computer-readable medium) on which the computer program is recorded for carrying out one of the methods described herein.
  • a further embodiment of the method according to the invention is thus a data stream or a sequence of signals, which represent the computer program for performing one of the methods described herein.
  • the data stream or the sequence of signals may be configured, for example, to be transferred via a data communication connection, for example via the Internet.
  • Another embodiment includes a processing device, such as a computer or a programmable logic device, that is configured or adapted to perform one of the methods described herein.
  • a processing device such as a computer or a programmable logic device, that is configured or adapted to perform one of the methods described herein.
  • Another embodiment includes a computer on which the computer program is installed to perform one of the methods described herein.
  • Another embodiment according to the invention comprises a device or system adapted to transmit a computer program for performing at least one of the methods described herein to a receiver.
  • the transmission can be done for example electronically or optically.
  • the receiver may be, for example, a computer, a mobile device, a storage device or a similar device.
  • the device or system may include a file server for transmitting the computer program to the recipient.
  • a programmable logic device eg, a field programmable gate array, an FPGA
  • a field programmable gate array may cooperate with a microprocessor to perform one of the methods described herein.
  • the methods are performed by any hardware device. This may be a universal hardware such as a computer processor (CPU) or hardware specific to the process, such as an ASIC.

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  • Health & Medical Sciences (AREA)
  • Otolaryngology (AREA)
  • Physics & Mathematics (AREA)
  • Engineering & Computer Science (AREA)
  • Acoustics & Sound (AREA)
  • Signal Processing (AREA)
  • General Health & Medical Sciences (AREA)
  • Circuit For Audible Band Transducer (AREA)
  • Obtaining Desirable Characteristics In Audible-Bandwidth Transducers (AREA)

Claims (12)

  1. Dispositif pour calculer les coefficients de filtre FIR pour les filtres de formation de faisceau d'un réseau de transducteurs (10), avec:
    un premier moyen de calcul (44) destiné à calculer des pondérations de filtre dans le domaine de la fréquence des filtres de formation de faisceau (141, ..., 14N) pour une plage de fréquences prédéterminée pour obtenir des réponses de fréquence cibles (78) pour les filtres de formation de faisceau, de sorte qu'une application des filtres de formation de faisceau au réseau de transducteurs (10) se rapproche d'une sélectivité de direction souhaitée (36; 38; 70; 74); et
    un deuxième moyen de calcul (46) destiné à calculer les coefficients de filtre FIR (32) pour les filtres de formation de faisceau, de sorte que les réponses de fréquence des filtres de formation de faisceau se rapprochent des réponses de fréquence cibles,
    qui présente par ailleurs un moyen de modification de réponse de fréquence cible (48) qui est couplé entre le premier moyen de calcul (44) et le deuxième moyen de calcul (46) pour modifier les réponses de fréquence cibles des filtres de formation de faisceau telles qu'elles sont obtenues par le premier moyen de calcul (44), de sorte que le deuxième moyen de calcul (46) calcule les coefficients de filtre FIR pour les filtres de formation de faisceau de sorte que les réponses de fréquence des filtres de formation de faisceau se rapprochent des réponses de fréquence cibles de forme modifiée par le moyen de modification de réponse de fréquence cible (48), la modification comportant
    un lissage de la plage de fréquences (92), et/ou
    pour chaque filtre de formation de faisceau, un nivellement (80) d'une réponse de phase (86) nettoyée de 2π sauts de phase de la réponse de fréquence cible du filtre de formation de faisceau respectif par élimination d'une part de fonction de phase linéaire (88) et mémorisation d'un retard pour le filtre de formation de faisceau respectif qui correspond à une augmentation de la part de fonction de phase linéaire.
  2. Dispositif selon la revendication 1, dans lequel le premier moyen de calcul (44) est conçu pour effectuer le calcul en résolvant un premier problème d'optimisation, après quoi est minimisé un écart entre une sélectivité de direction du réseau telle qu'elle résulte des pondérations de filtre dans le domaine de la fréquence et la sélectivité de direction souhaitée (74).
  3. Dispositif selon la revendication 2, dans lequel le premier moyen de calcul (44) est conçu de sorte que le premier problème d'optimisation soit un problème d'optimisation convexe.
  4. Dispositif selon la revendication 2 ou 3, dans lequel le premier moyen de calcul (44) est conçu pour effectuer le calcul dans une première plage de basses fréquences audio (100) en résolvant le premier problème d'optimisation pour obtenir des réponses de fréquence cibles de gamme de basses fréquences pour les filtres de formation de faisceau, et dans une deuxième plage de hautes fréquences audio (102) en calculant les retards globaux de fréquence et de pondérations d'amplitude pour le réseau en fonction de la sélectivité de direction souhaitée, et en combinant ensuite les réponses de fréquence cibles de plage de basses fréquences avec les réponses de fréquence cibles de plage de hautes fréquences qui correspondent aux retards globaux de fréquence et aux pondérations d'amplitude.
  5. Dispositif selon l'une des revendications précédentes, dans lequel le dispositif présente par ailleurs un moyen de modification de coefficient de filtre FIR (50) qui est conçu pour soumettre les coefficients de filtre FIR (32) tels qu'ils sont calculés par le deuxième moyen de calcul à un décalage dans le domaine temporel qui correspond au retard mémorisé pour le filtre de formation de faisceau respectif.
  6. Dispositif selon l'une des revendications précédentes, dans lequel le deuxième moyen de calcul (46) est conçu pour effectuer le calcul en résolvant un deuxième problème d'optimisation, après quoi est minimisé un écart entre les réponses de fréquence des filtres de formation de faisceau qui correspondent aux coefficients de filtre FIR et les réponses de fréquence cibles.
  7. Dispositif selon la revendication 6, dans lequel le deuxième moyen de calcul (46) est conçu de sorte que le deuxième problème d'optimisation soit un problème d'optimisation convexe.
  8. Dispositif selon la revendication 6 ou 7, dans lequel le deuxième moyen de calcul (46) est conçu de sorte que le deuxième problème d'optimisation pondère l'écart de manière sélective en fréquence ou prédétermine des seuils de tolérance fonction de la fréquence pour l'écart.
  9. Dispositif selon l'une des revendications 6 à 8, dans lequel le deuxième moyen de calcul (46) est conçu de sorte que le deuxième problème d'optimisation présente, comme contrainte secondaire, une limitation de la quantité des réponses de fréquence des filtres de formation de faisceau qui correspondent aux coefficients de filtre FIR, dans au moins un segment de fréquences dans lequel l'écart n'est pas minimisé.
  10. Dispositif selon l'une des revendications 1 à 9, dans lequel une résolution de fréquence des filtres de formation de faisceau telle qu'elle est définie par les coefficients de filtre FIR est différente d'une résolution de fréquence de la plage de fréquences pour laquelle sont calculées les pondérations de filtre dans le domaine de la fréquence des filtres de formation de faisceau.
  11. Procédé pour calculer des coefficients de filtre FIR pour les filtres de formation de faisceau d'un réseau de transducteurs (10), avec le fait de:
    calculer les pondérations de filtre dans le domaine de la fréquence des filtres de formation de faisceau (141,..., 14N) pour une plage de fréquences prédéterminée pour obtenir des réponses de fréquence cibles (78) pour les filtres de formation de faisceau, de sorte qu'une application des filtres de formation de faisceau au réseau de transducteurs (10) se rapproche d'une sélectivité de direction souhaitée (36; 38; 70; 74); et
    modifier les réponses de fréquence cibles des filtres de formation de faisceau, où la modification comporte
    un lissage de la plage de fréquences (92), et/ou
    pour chaque filtre de formation de faisceau, un nivellement (80) d'une réponse de phase (86) nettoyée de 2n sauts de phase de la réponse de fréquence cible du filtre de formation de faisceau respectif par élimination d'une part de fonction de phase linéaire (88) et mémorisation d'un retard pour le filtre de formation de faisceau respectif qui correspond à une augmentation de la part de fonction de phase linéaire; et
    calculer les coefficients de filtre FIR (32) pour les filtres de formation de faisceau de sorte que les réponses de fréquence des filtres de formation de faisceau se rapprochent des réponses de fréquence cibles de forme modifiée par le moyen de modification de réponse de fréquence cible (48).
  12. Programme d'ordinateur avec un code de programme pour mettre en oeuvre le procédé selon la revendication 11 lorsque le programme est exécuté sur un ordinateur.
EP15753373.8A 2014-08-22 2015-08-21 Calcule de coefficients d'un filtre rif pour un filtre d'un dispositif de beamforming Active EP3183891B1 (fr)

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PCT/EP2015/069291 WO2016026970A1 (fr) 2014-08-22 2015-08-21 Calcul de coefficient de filtre fir pour un filtre de formation de faisceau

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