EP2870659A1 - Antenna system for broadband satellite communication in the ghz frequency range, comprising dielectrically filled horn antennas - Google Patents

Abstract

The invention relates to an antenna system comprising at least four horn antennas which are at least partially filled with a dielectric. The effective wavelength in the horn antennas increases according to the dielectric properties of the filling, and the horn antennas can support much larger bandwidths than if they had no filling. Although dielectric fillings result in parasitic losses due to the dielectric, said losses remain comparatively small especially in small horn antennas for use in the Ka band. The permittivity of the dielectric is selected in such a way that the dielectrically filled horn antennas can be optimally operated even in the lowest service frequency band of the antenna system.

Description

Antenna system for broadband satellite communication in the GHz frequency range with dielectrically filled horns

The invention relates to an antenna system for broadband communication between earth stations and satellites,

especially for mobile and aeronautical applications.

 The demand for wireless broadband data transmission channels with very high data rates, especially in the field of mobile satellite communications, is steadily increasing. However, especially in the aeronautical field, there is a lack of suitable antennas, which in particular can fulfill the conditions required for mobile use, such as small dimensions and low weight. For the directional, wireless

Data communication with satellites (e.g., in the Ku or Ka band) also places extreme demands on the

Transmission characteristic of the antenna systems, since a disturbance of adjacent satellites must be reliably excluded.

 In aeronautical applications, the weight and size of the antenna system are very important as they reduce the payload of the aircraft and add extra weight

Operating costs cause.

The problem therefore is to provide antenna systems that are as small and lightweight as possible, which nevertheless satisfy the regulatory requirements for transmitting and receiving operation when operating on mobile carriers.

 The regulatory requirements for the transmission operation arise e.g. from the standards 47 CFR 25.209, 47 CFR 25.222, 47 CFR 25.138, ITU-R .1643, ITU-R p.524-7, ETSI EN 302 186 or ETSI EN 301 459. All of these regulatory requirements are designed to ensure that Transmission mode of a mobile satellite antenna no interference may occur adjacent satellites. These are typically envelopes

 (Envelopes or masks) of maximum spectral power density as a function of the distance angle to the target satellite

Are defined. The for a given distance angle

given values may not be exceeded in the transmission mode of the antenna system. This leads to strict

Requirements for the angle-dependent antenna characteristic. With increasing distance angle of the target satellites the

CONFIRMATION COPY Antenna gain drops sharply. This can be physically only by very homogeneous amplitude and phase assignments of

Antenna can be achieved. Typically, therefore

Parabolic antennas used, which have these properties.

For most mobile applications, especially on

Airplanes, parabolic mirrors are very poorly suited because of their size and because of their circular aperture. In commercial aircraft, for example, the

Antennas mounted on the fuselage and may therefore have only the lowest possible height because of the additional air resistance.

Antennas which as sections of paraboloid

 ("Banana-shaped mirror") are executed, although possible, but have geometriebedingt only a very small

Efficiency.

 Antenna fields, which are composed of individual radiators and have suitable feed networks, however, can in any geometry and any length to

Aspect Ratio be executed without the

Antenna efficiency suffers. In particular, you can

Antenna fields are realized very low altitude.

However, in antenna fields, especially when the reception frequency band and the transmission frequency band go far

apart (such as in the Ka band with reception frequencies at about 18 GHz - 21 GHz and transmission frequencies at about 28 GHz - 31 GHz) the problem lies in the fact that the individual radiators of the fields must support a very large bandwidth.

It is known that horns are the distance

most efficient single radiators in fields are. In addition, horns can be designed broadband.

In the case of antenna fields which are constructed from horn radiators and are fed with pure waveguide networks, however, the known problem of significant parasitic sidelobes (so-called "grating lobes" or "grating lobes") occurs in the antenna pattern. These grating lobes result from the fact that the beam centers (phase centers) of the antenna elements, which form the antenna field, because of the dimension of the

Hohleiternetzwerke by design too large

Have distance to each other. This can, especially at

Frequencies above about 20 GHz, below certain

Beam angles for the positive interference of the antenna radiators and thus for the unwanted emission of

electromagnetic power in unwanted

Lead solid angle ranges. In addition, the reception and transmission frequencies are far apart in terms of frequency and the distance between the beam centers must be

are designed for regulatory reasons according to the minimum useful wavelength of the transmission band, then the horns are regularly so small that the receiving band can no longer be supported by them.

In the Ka band, for example, the minimum useful wavelength is only about learning. So that the beam elements of the antenna array are tight, so no parasitic side lobes

(Grating lobes) may occur, the aperture area of a

quadratic horn only amount to approx. However, conventional horns of this size have only a very low performance in the reception band at approx. 18 GHz - 21 GHz, since they have to be operated close to the cut-off frequency because of the finite aperture angle. The Ka-band can no longer support such horns or their efficiency decreases very much in this band.

In addition, the horns generally have to support two orthogonal polarizations, which further restricts the geometrical margin, since an orthomode signal converter, so-called transducers, becomes necessary at the horn output. An embodiment of the orthomode signal converter in waveguide technology fails regularly because at higher GHz frequencies not enough space is available.

If the horns packed tightly in fields, then another problem is that in the available space behind the horn field no more efficient feed networks can be accommodated.

It is well known the feed networks for fields of

Horn radiators, which are implemented in high-power technology, only very small dissipative losses. In the optimal case, the individual horns of the fields of

Waveguide components fed and the entire food network is also made of waveguide components. In the case that the reception and the transmission band are far apart in frequency, however, the problem arises that conventional

Waveguides can then no longer support the required frequency bandwidth.

 For example, in Ka band, the required bandwidth is more than 13 GHz (18 GHz - 31 GHz). conventional

Rectangular waveguides can not support such a large bandwidth efficiently.

This results in the following problems for mobile, in particular aeronautical satellite antennas of small size, which must be solved simultaneously: 1. regulatory compliant antenna diagram without parasitic side lobes (grating lobes) in the transmission frequency band, the

Operation of the antenna with maximum spectral power density allowed,

 2. High antenna efficiency both in the reception band and in the transmission band, even with small individual radiator dimensions,

3. Efficient feed networks, which take up the smallest possible installation space and generate the lowest possible dissipative losses,

4. The most compact and space-saving design of the antenna with the highest possible antenna efficiency.

If these problems are solved by a suitable arrangement, then, even if only a limited space for a small antenna is available, a powerful

System for large bandwidths will be provided.

It is known that with antennas which as fields of

Single emitters are designed to grating-lobe free

Antenna diagrams can then be achieved when the

Phase centers of the individual emitters are less than a wavelength of maximum useful frequency apart. In addition, it is known that the side lobes of the antenna diagram are due to parabolic amplitude assignments of such antenna fields

can be suppressed (e.g., J.D. Kraus and R.J. Marhefka, "Antennas: for all applications," 3rd ed., McGraw-Hill series in electrical engineering, 2002)

Amplitude assignments can be optimally adapted to the regulatory mask for a given antenna size

Antenna diagram (e.g., DE 10 2010 019 081 AI, Seifried, Wenzel et al.).

The object of the invention is to provide a broadband antenna system in the GHz frequency range, in particular for aeronautical applications, which provides a regulatory compliant transmission mode over a large transmission range

Bandwidth with maximum spectral power density allowed while receiving a high

Antenna efficiency and a low inherent noise has.

This object is achieved by the antenna system according to claim 1.

According to the invention, the antenna system consists of at least four horns, which are at least partially filled with a dielectric. According to the dielectric properties of the filling then increases the effective wavelength in the Horns and these are able to support much larger bandwidths than would be the case without filling. Although dielectric fillings lead to parasitic

Losses through the dielectric, but especially for very small horns, these losses remain comparatively small. For applications in the Ka band, a dielectric filling with a dielectric constant of about 2 is sufficient. With horns only a few centimeters deep, this leads to losses of <0.2 dB when using suitable materials.

In an advantageous embodiment of the antenna system, the dielectric constant (permittivity number) of the dielectric with which the horn radiators are filled is selected such that the frequency-filled horn radiators are frequency-wise

lowest useful band of the antenna system is still optimal

can be operated. The prerequisite for this is that the wavelength belonging to the lowest usable frequency in the dielectric still sufficiently far above the lowest

Operating frequency, which is given by the geometric dimensions of the horn, is removed.

 If the horns are designed such that no parasitic side lobes (grating lobes) occur in the antenna pattern or at least in a section through the antenna pattern of the antenna system, then the dimension of the aperture of the horns is smaller in at least one direction

at most equal to the wavelength λ _{3 of} a reference frequency, which lies in the transmission band of the antenna.

Now is λ _Ε the free space wavelength of the lowest

Useful frequency, then applies to the corresponding wavelength in the dielectric

if ε is the permittivity (permittivity) of the

Dielectric at the corresponding frequency called

Now, a horn only works satisfactorily if at least one aperture dimension is close to the wavelength of the lowest useful frequency, because only then does a regular antenna pattern of the single radiator result.

 Thus, must

λε, medium * A _s (2)

apply what together with equation (1) to the condition

leads, However, condition (3) applies to individual insulated horns and is too strict for antenna systems which are composed of multiple horns, since the mutual

electromagnetic couplings of the horns not

are considered.

 If the mutual couplings are taken into account, then it has been found that it is optimal for an antenna system, which consists of several dielectrically filled horns, if the condition ε> ^ - (4) is satisfied. If the antenna system is designed according to condition (4), then the antenna system, even if e.g. the receiving and transmitting bands of the antenna are far apart in frequency, with very good performance parameters

feature .

According to an advantageous further development of the invention, the individual radiators support a first and a second polarization and the two polarizations are mutually orthogonal.

After a further advantageous development of

Invention, the first and second polarization are linear

Polarizations.

The signals of the two orthogonal polarizations are carried in separate feed networks, which has the advantage that with the aid of corresponding components, such as e.g. Polarizers or 90 ° hybrid couplers, both linearly polarized signals and circularly polarized signals sent, or

can be received.

In order that the antennas may have the smallest possible size and nevertheless a regulatory compliant transmission mode with maximum spectral power density is possible, is also in accordance with an advantageous development of the

Invention provided that at least part of the

Single radiator is dimensioned so that for direct

adjacent single radiator the distance of the phase centers of the single radiator less than or equal to the wavelength of the

highest transmission frequency is when no parasitic

Side lobes (grating lobes) may occur (reference frequency in the transmission band).

If at least four adjacent individual radiators are also located in different directly adjacent modules, then at least one direction is defined by the antenna field, so that for this direction the distance of the phase centers of the Single radiator is less than or equal to the wavelength of the highest transmission frequency, at which no parasitic

Side lobes (grating lobes) may occur.

In this direction, preferably along a straight line through the antenna field, are directly adjacent

Single radiator then tight, so no parasitic

Otherwise, these grating lobes would lead to a strong reduction of the "grating lobes" in the corresponding section through the antenna pattern

regulatory allow spectral power density.

As a single radiator come in principle all known

Radiation elements, which two orthogonal polarizations

support, in question. These are e.g. rectangular or

Horn radiator.

It is also advantageous if the modules have an at least approximately rectangular geometry, that is to say that Νχ = ^η x ri _k contain individual radiators, where Ν, n, i, 1, k are straight

 Numbers are, ^ N, = Ngilt and N is the total number of

 Single radiator is. Such rectangular modules can be assembled in a space-saving manner to antenna fields. In addition, the rectangular modules can be fed in a relatively simple manner with binary microstrip networks.

In order to realize antennas with the lowest possible dissipation losses, it is advantageous for the individual radiators

Form horn horns, which are among the lowest loss antennas. Both horns with a rectangular and a round aperture opening can be used. If grating-lobes are not to occur in any section through the antenna pattern, horns are square

Aperturöffnung advantageous, the size of Aperturöffnun is then chosen so that the distance between the phase centers of directly adjacent horns less than or equal to the

Wavelength of the highest transmission frequency is the reference frequency at which no grating lobes may occur.

If the horns of the antenna system are tight and stepped, then in a further advantageous embodiment the apertures of the horns and the steps are designed such that the horns function optimally both in the receiver and in the transmission band of the antenna.

This can be achieved in that the horns have a rectangular aperture whose two edge lengths are smaller or at most equal to the free space wavelength of a

Reference frequency are, which lies in the transmission band of the antenna. The area available for the antenna system is optimally utilized and a maximum antenna gain is achieved.

In order that the stepped horns are optimally adapted in the receiving band, the horns are designed so that they have at least one rectangular cross section between the aperture (horn opening) and the horn end, for the longer edge k applies

k> - ^ - where λ _Ε the free space wavelength of the lowest usable frequency, and ε the dielectric constant (permittivity) of the

denoted dielectric filling. This ensures that the lowest usable frequency (typically the

lowest reception frequency) is above the lower limit frequency ("cut-off" frequency) of the horn.

In order to achieve optimum adaptation in the transmission band, the horns also have at a deeper point over another rectangular cross section for the longer edge k _T applies

 k> - ^ -

where λ _τ denotes the free space wavelength of the highest useful frequency. This is the highest at Hornende

Usable frequency of the antenna (typically the maximum

Transmission frequency) above the "cut-off" frequency.

In order to achieve the widest possible bandwidths, it is also advantageous if the individual radiators are designed as horn radiators in such a way that in the two polarization planes with symmetrical geometrical constrictions, i.

Constrictions are equipped and fed separately at its output for each of the two orthogonal polarizations on the belonging to the respective polarization direction geometric constriction. Such geometric

Constrictions can greatly increase the range of horns

enlarge.

If the transmitting and receiving bands are far apart in terms of frequency then, according to a further advantageous

By adjusting the width and length of the steps, as well as the number of stages, then the antenna can be optimally adapted to the respective usable frequency bands.

In order to achieve a high cross-polarization decoupling, it is also advantageous if the horns are designed so that they support two orthogonal linear polarizations. With such horns, insulations far exceeding 40 dB can be achieved. Especially with signal codings with high spectral efficiency are such isolation values

required.

A further improvement in the reception power, especially in the case of very small horn radiators, can be achieved by providing the individual horn radiators with a dielectric

Cross-septum or a dielectric lens can be equipped. The insertion loss (Sn) in the reception band can be significantly reduced by such structures, even if the aperture areas of the individual radiators are already so small that a free-space wave without these additional dielectric structures would already be almost completely reflected.

Since in parallel-fed individual radiators, the dissipative losses, such as by a dielectric filling occur only once, are for a further advantageous

Further development of the invention, the horns of the

Antenna field fed in parallel. This is most effective when the microstrip lines and the waveguides are constructed as binary trees, since the number of required power dividers in the general case of arbitrary values of

Total number of individual radiators N and any values of the number of individual radiators in a module Ni is so minimal.

The binary trees are in the general case neither

completely still completely symmetrical.

Applies, however, according to an advantageous development of the invention for all modules of the antenna system or at least for most of the modules N _i = 2 "', with rii a whole

Number, then the number of required power dividers can be further reduced, because then at least part of the binary trees is complete.

It is particularly favorable if, in addition, N = 2 ", where n is an integer, then the feed networks of the

Antenna system can be designed as complete and fully symmetrical binary trees and all individual radiators can equal length feeder lines, i. also similar

Dampings, have.

It is also advantageous if the microstrip lines are located on a thin substrate and are guided in closed metallic cavities, wherein the cavities are typically filled with air. A substrate becomes typically referred to as thin when its thickness is smaller than the width of the microstrip lines.

This coaxial line-like structure with typically air as a filling leads to comparatively low-loss

High-frequency lines. So it has been shown that the

dissipative losses of such lines e.g. at Ka-band

Frequencies are only about a factor of 5 to 10 higher than the losses of waveguides. Since these lines are used only for relatively short distances, the absolute losses remain relatively small. Also the

Noise contribution of such lines to the inherent noise of the system remains relatively small.

 The production of densely packed antenna systems can be greatly facilitated by the fact that they consist of several layers

are constructed and are the microstrip networks of the two orthogonal polarizations between different layers. The modules of the antenna system can then be assembled from a few layers. Advantageously, the layers of aluminum or similar electrically conductive materials, which can be structured with the known structuring method (milling, etching, lasers, wire erosion, water cutting, etc.). The microstrip line networks are deposited on a substrate by known etching techniques

structured.

 Advantageously, the cavities through which the microstrip lines are routed directly to the

textured metallic layers. If the cavities are designed as notches or depressions in the metallic layers respectively above and below the microstrip line, then the microstrip line lies together with its

Substrate in a cavity, which consists of two half-shells. The walls of the cavity can be electrically closed by the substrate with electrical

 Vias is provided. "Fences" of Vias in such arrangements can cause loss

almost completely prevent electromagnetic power.

Are the receiving and transmitting bands of the antenna

Frequently far apart in terms of frequency, then it may be the case that standard waveguide (rectangular waveguide) the

can no longer support the required bandwidth. In this case, it is advantageous, the waveguide along the

Propagation direction of the electromagnetic wave with

to provide geometric constrictions (constrictions). By such constrictions, the useful bandwidth can be greatly increased. The number and arrangement of constrictions depend on the design of the antenna system. For very large useful bandwidths so-called double-ridged waveguides are advantageous, which is a significantly larger

Can have bandwidth as a standard waveguide. These waveguides have a geometric constriction parallel to the supported polarization direction, which prevents the formation of parasitic higher modes.

 At very high useful frequencies or very dense

Single radiators, an advantageous development of the invention is that dielectrically filled waveguides are used for the waveguide supply networks. Such

Waveguides require much less space than

air-filled waveguide. Depending on the requirements of the installation space can also be a part or a whole

Waveguide network consist of dielectric filled waveguides. Also a partial filling is possible.

For further processing of the signals, e.g. by coupling a low-noise amplifier (LNA) to the receiving feed network and / or a

Power Amplifier ("High Power Amplifier", HPA) to the transmit feed network, it may be advantageous the

Equip food networks with frequency diplexers. Such frequency diplexers separate the reception from the transmission band. In particular waveguide diplexers are advantageous because they can achieve a very high isolation and are also very low attenuation.

At which point the frequency diplexers are inserted into the feed networks depends on the particular application. For example, e.g. conceivable that each module of the antenna array is equipped with a diplexer directly at its output or input. At the input and output of these diplexers are then all signal combinations in pure form: polarization 1 in the receiving band, polarization 2 in the receiving band, polarization 1 in the transmission band and polarization 2 in the transmission band. The modules can then be interconnected by four corresponding waveguide feed networks. These

Embodiment has the advantage that the waveguide feed networks frequency must not be very broadband, because they must be suitable only for signals of Empfangsbzw, the transmission band.

However, it is also conceivable that the frequency diplexer

only be mounted respectively at the input and output of the waveguide networks. Such an embodiment saves space, but typically requires a broadband design of the waveguide networks.

For applications that need to transmit or receive in different polarizations, or in applications where it is the polarization of the transmit or the receive signal changes dynamically ("polarization diversity"), it is

advantageous if both the intra-modular

Microstrip line networks as well as the inter-modular waveguide networks are designed so that they can simultaneously support the transmitting and the receiving band.

 If the antenna is provided with frequency diplexers connected to a suitable radio frequency switching matrix, then dynamic switching between the orthogonal polarizations is possible

("Polarization switching").

Such embodiments are particularly advantageous if the antenna is to be used in satellite services which work with the so-called "spot beam" technology. "Spot beam" technology is produced on the earth's surface

Coverage areas (cells) of relatively small area (typical diameter in Ka band approx. 200km -300km). To be able to use the same frequency bands in neighboring cells

 ("Frequency re-use"), adjacent cells are distinguished only by the polarization of the signals.

When using the antenna on fast-moving carriers, in particular on aircraft, then typically very many and very fast cell changes take place and the antenna must be capable of the polarization of the receiving or

Switching transmission signals quickly.

 On the other hand, if the antenna is used in satellite services in which the polarization of the received or transmitted signal is fixed and changes neither temporally nor geographically, then it is advantageous if the first intra-modular

Microstrip line network and the associated inter-modular waveguide network on the receiving band of the Antennne, and the second intra-modular microstrip line network and the associated inter-modular waveguide network are designed for the transmission band of the antenna system.

 This embodiment has the advantage that the respective feed networks can be optimized for the respective usable frequency band, and thus a very low-loss antenna system of very high performance is created.

Are the radiating elements of the antenna system on two

designed orthogonal linear polarizations, then according to an advantageous embodiment of the invention, the

Food networks equipped with so-called 90 ° hybrid couplers. 90 ° hybrid couplers are four-ports which convert two orthogonal linearly polarized signals into two orthogonal circularly polarized signals and vice versa. With such Arrangements it is then possible to send or receive circularly polarized signals.

Alternatively, the antenna array for receiving and transmitting circularly polarized signals with a

so-called polarizer. Typically, these are suitably structured metallic layers ("layers") lying in a plane approximately perpendicular to the propagation direction of the electromagnetic wave, the metallic structure acting in such a way that it is capacitive in one direction and inductive in the orthogonal direction two orthogonally polarized signals, this means that a phase difference is imposed on the two signals: if the phase difference is now set to be just 90 ° passing through the polarizer, then two orthogonal linearly polarized signals are converted into two orthogonal circularly polarized signals ,

vice versa .

In order to obtain large useful bandwidths, the polarizer advantageously consists of several layers, which are mounted at a certain distance (typically in the region of a quarter wavelength) from each other.

A particularly suitable embodiment of the polarizer is a multi-layer meander polarizer. Here are the usual patterning metallic

Meander structures of suitable dimensions on one

typically textured thin substrate. The way

structured substrates are then applied to foam boards

glued or laminated to sandwiches. As foams are e.g. low-loss closed-cell foams such as Rohacell or XPS in question.

Advantageous here is a sequence of foam boards,

Superimposed on adhesive films and structured substrates and pressed with a press. In a relatively simple manner then creates a suitable polarizer light weight.

According to a further advantageous embodiment of the invention, very high useful bandwidths and high cross polarization isolation are achieved if the polarizer is not accurate

is mounted perpendicular to the propagation direction of the electromagnetic wave in front of the antenna field, but slightly tilted. In these arrangements, the typical distance of the polarizer to the aperture surface of the antenna array is in the range of a wavelength of the useful frequency and the tilt angle

opposite the aperture plane in the range of 2 ° to 10 °.

Because the antenna pattern of the

Antenna system in the transmission band under a regulatory given small mask size, and small antennas can only be transmitted with high spectral power densities if the diagram is as close as possible to the mask, it may be advantageous to use the antenna system with amplitude amplitude tapering

Mistake. Parabole amplitude assignments of the aperture are particularly suitable in the case of flat aperture openings for this purpose. Parabole amplitude assignments are thereby

in that the power contributions of the individual radiators increase from the edge of the antenna field toward the middle and extend z. B. results in a parabolic-like course.

Such amplitude assignments of the antenna field lead to a suppression of the side lobes in the antenna pattern and thus to a higher regulatory allowed spectral

Power density.

 Since the side lobes in applications in geostationary

Satellite services only along a tangent to the

geostationary orbit must be suppressed at the location of the target satellite, the amplitude occupancy of the antenna system is preferably designed so that they at least along the direction through the antenna system in which the

Radiation elements are tight, acts. Here are the

Radiation elements in the direction dense, in which the distance of the phase centers of the individual radiators is less than or equal to the wavelength of the highest transmission frequency at which no significant parasitic side lobes (grating lobes) may occur.

In addition, other benefits and features of the

present invention from the description more preferred

Embodiments can be seen. The features described therein may be implemented alone or in combination with one or more of the features mentioned above. The following description of the preferred embodiments will be made with reference to the accompanying drawings

Drawings .

BRIEF DESCRIPTION OF THE FIGURES

Fig. La-b show schematically an inventive

Antenna module, which consists of a field of 8 x 8

Individual emitters exists;

Fig. 2a-b show exemplary

 Microstrip feed networks for an 8 x 8

Antenna module; Fig. 3a-d illustrate schematically the exemplary structure of an antenna according to the invention from antenna modules and the

Networking of the modules by waveguide networks;

 4a-d show the detailed structure of a single quadruple toothed horn radiator;

Fig. 5 schematically illustrate the detailed structure of a 2 x 2

Antenna module of four-toothed ("quad-ridged")

Horn radiator is;

Figures 6a-b show an exemplary 8x8 antenna module consisting of dielectrically filled horns;

Figures 7a-d illustrate the exemplary detailed construction of a single dielectrically filled horn radiator;

Fig. 8 shows schematically the detailed structure of a 2 x 2 module of dielectrically filled horns;

Fig. 9 shows a module according to the invention which is provided with a dielectric grid for improving the impedance matching;

10a-b show a module according to the invention in ply technique;

Figures lla-d show the detailed structure of a module according to the invention in layering technology;

Fig. 12 shows schematically the vacuum model of a

module according to the invention;

FIG. 13 shows the exemplary construction of a waveguide power splitter composed of double-ridged waveguides; FIG.

Fig. 14 schematically shows a position of a polarizer;

Fig. 15a-b show an example of a schematic

Amplitude assignment of an antenna system according to the invention and the resulting maximum regulatory spectral EIRP density;

Fig. 16 shows a possible construction of a fixed polarization antenna system according to the invention of the transmitting and receiving signals in the form of a block diagram;

Fig. 17 shows a possible construction of a variable polarization antenna system according to the invention of the transmitted and received signals using 90 ° hybrid couplers in the form of a block diagram; Fig. 18 schematically shows the construction of a variable polarization antenna system of the transmission and reception signals of the present invention using a polarizer in the form of a block diagram.

The exemplary illustrated in the drawings

Embodiments of the antenna and its components are explained in more detail below.

Fig. 1 illustrates an exemplary embodiment of a

Antenna module of an antenna according to the invention. The

Single emitters 1 are here designed as rectangular horns, which can support two orthogonal polarizations.

The intra-modular microstrip line networks 2, 3 for the two orthogonal polarizations are located between different layers.

The antenna module consists of a total of 64 primary

Single emitters 1 which in an 8 x 8 antenna field

are arranged (N = 64). The dimensions of the individual radiators and the size of their aperture surfaces are chosen so that the distance of the phase centers of the individual beam elements along both major axes is less than X _m i _n , where A _min denotes the wavelength of the highest useful frequency. This distance ensures that parasitic sidelobes, so-called "grating lobes", can not occur in any direction in the antenna diagram up to the highest usable frequency (reference frequency).

In the exemplary case of the antenna module shown in Figure 1, both microstrip line networks set a 64: 1

Power dividers because they merge the signals from 64 individual emitters. An exemplary internal organization of the two microstrip line networks is shown in FIG. 2

shown.

However, embodiments are also conceivable for which the modules comprise a smaller or larger number of horns. For K / Ka band antennas e.g. 4 x 4 modules are optimal. The microstrip networks then represent a 16: 1 power divider that receives the signals from 16

Single radiators merges. The microstrip lines in this case are relatively short and their noise contribution therefore remains small.

Depending on the application, an antenna with optimum performance parameters can be constructed by appropriate design of the module sizes. Advantageously, the modules are only made as big as necessary to feed them with waveguides. The parasitic noise contribution of

Microstrip lines are thereby minimized.

The two microstrip line networks 2, 3 couple the merged signals, each separated by polarization, into microstrip-to-waveguide couplings 4, 5, as shown in FIG. 1b. Through this waveguide couplings 4, 5 can be any number of modules with the help of

Waveguide networks are coupled efficiently and low attenuation to an antenna system according to the invention.

FIG. 2 shows two exemplary microstrip line networks 2, 3 for feeding the individual radiators 1 of the 8 × 8

Antenna module of Fig. 1. Both networks are designed as binary 64: 1 power dividers.

The two mutually orthogonal microstrip-to-waveguide couplings 6, 7, the orthogonal polarized signals in the individual horns of the 8 x 8 module or coupled. The sum signal is sent to the

Waveguide couplings 4a and 5a in waveguide one or

decoupled. Since the two microstrip line networks 2, 3 are typically superimposed in two planes, waveguide feedthroughs 4b and 5b are also located on the corresponding board in order to create an opening and the connection to the waveguide couplings 4a and 5a, respectively.

The microstrip line networks 2, 3 can be made by any known method. Whereby low-loss substrates for antennas are particularly suitable.

FIG. 3 shows by way of example how different antenna modules 8 can be coupled to antenna systems according to the invention.

Antenna systems according to the invention consist of a number M of modules, where M must be at least two. In Fig. 3 are examples of modules with Ni = 8 x 8 = 64 (i = 1, ..., 16)

Single radiators 1 shown. M is equal to 16 and the modules are arranged in an 8 x 2 array (see Fig. 3a), which is a rectangular antenna with N = ^ N, = 64 x 16 = 1024

Single radiators results.

However, other arrangements of the modules and other module sizes are also conceivable. Thus, the modules may e.g. also be arranged in a circle. Also, not all modules must have the same size (number of individual emitters).

The modules 8 are now connected to each other by means of the waveguide networks 9, 10. For this purpose, the corresponding Waveguide coupling points 11, 12 of the waveguide networks 9, 10 with the corresponding waveguide couplings 4, 5 (see Fig. Lb) of the individual modules 8 connected.

The waveguide networks 9, 10 themselves each represent an M: 1 power divider, so that the two orthogonally polarized signals are fed to the sum ports 13, 14 in the

Antenna system fed or from the antenna system

can be disconnected.

Depending on the application and the required frequency bandwidth, waveguides 9, 10 can be provided with a wide variety of waveguides, such as, for example, waveguide networks. Conventional rectangular or round waveguides or broad-band toothed ("ridged") waveguides are used, and dielectrically filled waveguides are also conceivable.

So it can be e.g. be beneficial to the part of the

 Waveguide network, which connects directly to the waveguide coupling 4, 5, to fill with a dielectric. The dimensions of the dielectrically filled waveguides are then significantly reduced, so that their space requirement is minimal.

The antenna shown in Fig. 3 is thus constructed according to claim 1:

The antenna consists of an antenna field of W

Single emitters 1 each individual emitter 1 two

can support independent orthogonal polarizations and N the total number of individual radiators 1 of the antenna field

designated .

In addition, the antenna array is composed of modules 8, each module containing N ± single radiators and ^ N ,. = N holds.

 i

In the embodiment of Fig. 3 this is in addition that each module Ni = nj _k xn contains individual radiators, N _if n, i, 1, k are integers and N ^, = Ngilt.

The individual radiators 1 are dimensioned (see Fig. 1), that for at least one direction through the antenna field of the

Distance of the phase centers of the horns is less than or equal to the wavelength of the highest transmission frequency at which no grating lobes may occur.

The individual radiators 1 are supplied separately by microstrip lines for each of the two orthogonal polarizations (see Fig. 2, microstrip-to-waveguide couplings 6, 7). The microstrip lines of one orthogonal polarization are the first intra-modular

 Microstrip line network 2 connected and the

 Microstrip lines of the other orthogonal polarization are to the second intra-modular

 Microstrip line network 3 connected.

The first micro-strip intra-modular network 2 is coupled to the first inter-modular waveguide network 9 and the second micro-strip intra-modular network 3 is coupled to the second inter-modular waveguide network 10 such that the first inter-modular waveguide network 9 receives all of the one orthogonal signals Polarization at the first sum port 13 merges and the second inter-modular waveguide network 10 all signals of the other orthogonal polarization at the second summing port 14 merges.

In addition, the microstrip line networks 2, 3 and the waveguide networks 9, 10 are here as complete and

fully symmetrical binary trees are constructed so that all individual radiators 1 are fed in parallel.

FIGS. 3c and 3d show a physical realization of a corresponding antenna system. The modules 8 consist of single radiators 1 and have two different sizes, i. the number of individual radiators 1 per module 8 is not the same for all modules 8. The middle four modules 8 each have 8 individual emitters 1 more than the other four modules 8. As a result, the height of the antenna system at the left and right edges is less than in the central area. Such embodiments are particularly advantageous when the antenna system must be optimally adapted to an aerodynamic radome.

The modules 8 are fed separately with two waveguide networks 9 and 10 for each polarization. The waveguide networks 9, 10 are located in two separate layers behind the modules and the modules are connected to the waveguide networks 9, 10 through the coupling points 11, 12, which are coupled to the waveguide couplings of the modules 4, 5 ,. Both waveguide networks 9, 10 are realized here as cutouts.

If the transmission and reception bands of the antenna system are widely separated in terms of frequency, then the case may arise that the dimensions of the individual radiators 1 of the field must be so small that the lower one of the two

Frequency bands close to the cutoff frequency of the

Single radiator 1 is coming, or even below it.

Conventional horns, for example, can do this Frequency band then no longer support or their efficiency decreases sharply.

For example, at a K / Ka band operation the

Receive frequency band at about 19GHz - 20GHz and the

Transmitting frequency band at approx. 29GHz - 30GHz. To the condition that the antenna diagram in the transmission band free of

parasitic sidelobes ("grating lobes") is to fulfill, the aperture of the single emitters 1 may be at most learning x learning large (A _m in is learning).

 For example, conventional dual polarized horns work with an aperture of only 1cm x 1cm

However, at 19GHz - 20GHz (A _max = 1.58cm) almost no more, because an acceptable impedance matching to the free space is no longer possible. In addition, the horn would have to be operated very close to the lower cutoff frequency ("cutoff" frequency), which would lead to very high dissipative losses and to a very low antenna efficiency.

It may therefore be advantageous that the primary

Single radiator 1 as a ridged horn

are executed. Such horns can greatly extended compared to conventional horns

Own frequency bandwidth.

 The impedance matching of such toothed horns to the free space then takes place according to the method of antenna physics. The toothed horns can be designed so that they have two

support orthogonal polarizations. This is e.g. This is achieved by virtue of the fact that the horns are quadripartite ("quad-ridged"), the signals of the orthogonal ones

Polarizations are separated

 Microstrip line networks 2, 3 supplied and removed.

 FIG. 4 a schematically shows the detailed construction of a horn radiator equipped with symmetrical geometric constrictions using the example of a four-toothed tooth

Horn 1. The horn 1 consists of three segments (layers), with the two between the segments

Microstrip networks 2,3 are located.

The horns 1 are symmetrical with geometric

Constrictions 15, 16 corresponding to the orthogonal

Polarization directions are provided, which extend along the propagation direction of the electromagnetic wave.

Such horns are referred to as "toothed" horns.

Shown in Fig. 4a is an exemplary quadruple

serrated single horn, the broadband two orthogonal

Can support polarizations. As shown in the sections in Figs. 4b and 4c, the geometric constrictions are executed in a stepped manner and the

Distance of the constrictions 15, 16 to each other decreases in the direction of the coupling and decoupling. As a result, a very large frequency bandwidth can be achieved. In particular, horns 1 can be realized, which also

Frequency-wise far-reaching transmit and receive tapes can support without substantial losses in the efficiency. An example of this are K / Ka band satellite antennas. Here the reception band lies at 18 GHz - 21 GHz and the transmission band at 28 GHz - 31 GHz.

The depth, width and length of the steps depend on the desired frequency bands and can be numerical

Simulation methods are determined.

The coupling or decoupling of the signals on the

 Microstrip line networks 2, 3 typically occur at the narrowest point of the constrictions 15, 16 for the respective polarization direction, which is a very broadband

Impedance matching allowed.

Fig. 4d shows schematically a part of the longitudinal section through a toothed horn at the location of two opposing constrictions 16. The constrictions 16 are stepped

and the distance di opposing stages decreases from the aperture of the horn (top) to the horn end (down) down.

In addition, the horn itself is stepped (see Fig. 4a-c), so that at each stage, the edge length j of the horn opening in the corresponding cross-section also decreases from the aperture of the horn to the horn end.

The distances di and the associated edge lengths ai, or at least a part thereof, are now designed so that the associated lower cutoff frequency of the respective toothed waveguide section is below the lowest

Use frequency of the horn is located. Only if this condition is met can the electromagnetic wave of the

corresponding wavelength in the horns up to

Waveguide to microstrip line coupling penetrate, and there are coupled or disconnected.

Since the dissipative damping when approaching the lower

Limit frequency increases sharply, the distances d and the associated edge lengths are advantageously chosen so that a sufficient distance to the cutoff frequency remains and the attenuation is not too high. In addition, it must be taken into account that in antenna systems which consist of several horns, mutual

Couplings of the radiators are effective.

 In Fig. 5, the inventive structure of a 2 x 2 antenna module is shown schematically, which consists of four quadruple

toothed horns 1, four couplings 17 on the

Microstrip line networks 2, 3, two separate for each of the two orthogonal polarizations

 Microstrip line networks 2, 3, and outcoupling of the microstrip line networks 2, 3 on the waveguide coupling 4, 5 has. The constrictions as symmetrical teeth 15, 16 of the horns 1 are also shown.

The two orthogonally polarized signals pole 1 and pole 2, their reception or radiation from the horns. 1

supported by the training or

Einkopplungsstellen 17 in the corresponding

 Microstrip line network 2, 3 is fed or extracted from this.

The microstrip line networks 2, 3 in turn are designed as a binary: 1 power divider and couple the

Sum signals in the waveguide 4, 5.

The distance of the phase centers of two adjacent horns 1 in the vertical direction is smaller than Ä _min , so that at least in this direction in the antenna diagram no

unwanted parasitic side lobes ("grating lobes") can occur and the horns are close in this direction.

In the example shown in FIG. 5, the phase centers of the horns 1 coincide with the beam centers of the horns 1. In general, however, this is not

necessarily the case. However, the position of the phase center of a horn 1 of any geometry can with

numerical simulation methods are determined.

 For the coupling and decoupling of the signals supported by the toothed horns 1 microstrip lines are due to their known broadband in a special way. In addition, microstrip lines require very little space, so that high-efficiency, broadband horn antenna systems whose antenna patterns have no parasitic side lobes ("grating lobes"), even for very high frequencies (for example, 30 GHz - 40 GHz) can be realized.

In Fig. 6, the antenna modules of dielectric filled horns 18 are constructed. The with a dielectric 19 filled horns 18 are here exemplified in an 8 x 8 antenna array and are on the

 Microstrip line networks 2 and 3 coupled together.

The microstrip line networks 2, 3 couple the

Sum signals in the waveguide couplings 4, 5.

FIGS. 7a-c show the internal structure of a single horn radiator 18 completely filled with a dielectric. Like the horn 18 itself, the dielectric packing (dielectric) 19 also consists of three segments, each defined by the microstrip line networks 2, 3.

 So that the individual radiators 1 can support two widely spaced frequency bands, they are designed stepped in their interior, as shown in the sections Fig. 7b-c is exemplified. The extraction or coupling of the highest frequency band is typically carried out at the narrowest or lowest point by the

Microstrip line network 3, which is furthest away from the aperture opening of the single radiator 1. The lower frequency band is switched on or coupled in at a further point to the aperture opening, by a microstrip line network 2.

 The depth, width and length of the steps depends on the desired frequency bands and can also be used here

numerical simulation methods are determined.

Are the two input or decoupling of the

However, the horn 1 can also be designed so that both inputs and outputs can support both the transmit and the receive frequency band.

The dielectric filling body 19 is also designed to match exactly stepped. The shape of the filling body 19 on the aperture surface depends on the

electromagnetic requirements to the antenna pattern of the single radiator 1. The filler 19 can be performed flat as shown at the aperture opening. However, others, e.g. curved inwards or outwards, versions possible.

As dielectrics come a variety of known materials such as Teflon, polypropylene, polyethylene, polycarbonate, or polymethylpentene in question. For simultaneous coverage of the K and the Ka band, for example, a dielectric with a sufficient Dielectric constant of about 2 (eg Teflon,

Polymethylpentene).

 In the exemplary embodiment illustrated in FIG. 7, the horn antenna 18 is completely filled with a dielectric 19. However, there are also embodiments with only

partial filling possible.

The advantage of using dielectrically filled horns is that the grains themselves have a much less complex internal structure than in the case of toothed horns.

However, in order to display high-efficiency antennas even at very high GHz frequencies, it is also conceivable, e.g.

fourfold toothed horn with a dielectric to fill. Other horn geometries with dielectric filling or partial filling are also possible.

In Fig. 7d, an advantageous embodiment of a stepped executed dielectrically filled horn radiator, which has a rectangular aperture is shown schematically.

Fig. 7d shows the view of the horn from above (top view) with the aperture edges ki and .2, and the longitudinal sections through the horn along the lines A-A 'and B-B'.

The horn is now designed so that a first

rectangular section through the horn exists, the opening of which has a long edge k _E , and a second

Cross section through the horn exists whose opening has a long edge k _s .

Now lies the receiving band of the antenna system in frequency lower than the transmission band and the edge k _{E is} now selected so that the associated lower limit frequency of a dielectric filled waveguide with a long edge k _E below the lowest useful frequency of the receiving band

Antenna system is located, then the horn can the

Support reception band.

In addition, if the edge k _s chosen so that the associated lower limit frequency of a dielectric filled waveguide with a long edge k _{s is} below the lowest useful frequency of the transmission band of the antenna system, then the

Horns also support the transmission band, and this is true even if the reception band and transmission band are far apart.

Since, in FIG. 7d, the edge k _{s is} orthogonal to the edge k _E , two orthogonal are simultaneously produced by such a horn supports linear polarizations, since the corresponding waveguide modes are linearly polarized and orthogonal to each other.

Such horns running in such a stepped manner can also be operated correspondingly without or only with partial dielectric filling, and that the embodiment shown in FIG. 7 d can be adapted to any number of rectangular horn cross sections and thus to any number of

Nutzbändern can be extended.

If the horns of the antenna system are now close, ie should no parasitic side lobes (grating lobes) occur in the antenna pattern of the antenna system, then in a further advantageous embodiment, the edge lengths ki and k _{2 of} the rectangular aperture of the horns so

chosen that both ki and k _{2 are} smaller or highest equal to the wavelength of the reference frequency, which is in the transmission band of the antenna.

 In this case, the available space is then optimally utilized and a maximum antenna gain is achieved.

FIG. 8 shows an exemplary 2 × 2 antenna module that consists of four dielectrically filled horns 18. As illustrated in FIG. 7b-c, the inputs and / or outputs in the microstrip line networks 2, 3 are completely embedded in the dielectric 19 here. Otherwise, the module does not differ from the corresponding toothed module

Horn radiators, as shown in Fig. 5, the

Microstrip line networks 2, 3 are connected to the waveguide couplings 4, 5, respectively.

 In Fig. 9, a further advantageous embodiment is shown. Here, the module is equipped with a dielectric grid 20 extending over the entire aperture opening. Such dielectric gratings 20 can greatly improve the impedance matching, in particular at the lower frequency band of the individual radiators 1, by virtue of their close proximity to the aperture openings of the individual radiators 1

Reduce wavelength.

In the example shown in Fig. 9, this is achieved

ensures that dielectric crosses are located above the centers of the aperture openings of the individual radiators. However, other embodiments such as cylinders, spherical body, cuboid etc. are possible. Also, the dielectric grid 20 need not be regular or periodic. For example, e.g.

conceivable that the grid for the horn 1 at the edge of the Antenna has a different geometry than for the horn 1 in the center. Thus, for example, edge effects could be modeled.

FIG. 10a-b illustrates an exemplary module that is incorporated in FIG

Layer technology is constructed. By this technique, modules according to the invention can be produced particularly cost-effectively. In addition, even at very high frequencies (high

Tolerance requirements) ensures the reproducibility of the modules.

The first layer consists of an optional polarizer 21, which is used in circularly polarized signals. The polarizer 21 converts linearly polarized signals into circularly polarized and vice versa, depending on the polarization of the incident signal. So be on the antenna system

incident circularly polarized signals in linear

converted polarized signals, so they from the

Horn radiators of the module can be received lossless. On the other hand, the linearly polarized signals radiated from the horns are converted into circularly polarized signals and then radiated into the clearance.

The next two layers form the front part of the

Horn radiator field, which includes the primary horn structures 22 without input or output unit.

 The following layers 23a, 2 and 23b form the input or

Decoupling the first linear polarization from the

Horns of the field. The microstrip line network 2 of the first polarization and its substrate are embedded in metallic carriers (layers) 23a, 23b. The carriers 23a, 23b have recesses (notches) at the locations where a microstrip line runs (see also FIG.

Reference numeral 25).

Likewise, the microstrip line network 3 of the second, orthogonal polarization with its substrate is embedded in the carriers 23b, 23c.

In the last position, the waveguide terminations 24 of the horns and the waveguide outcouplings 4 and 5.

The primary horn structures 22, the carriers 23a-c and

Waveguide terminations 24 are electrically conductive and can be inexpensively with the known methods of

Metalworking e.g. made of aluminum (e.g., milling, laser cutting, water jet cutting, electroerodizing).

However, it is also conceivable to produce the layers of plastic materials, which are then completely or partially coated with an electrically conductive layer (eg galvanic or chemical). For the production of the plastic layers, for example, the known injection molding methods can also be used. Such embodiments have the advantage over layers of aluminum or other metals that a considerable weight reduction can result, which is particularly the case with applications of the antenna system

Aircraft is beneficial.

 With this layer technique, even at very high GHz frequencies, it becomes a highly efficient and cost-effective solution

Antenna module provided.

The postage technique described can be used both for

Use antenna modules made of toothed horns as well as modules made of dielectrically filled horns in the same way.

Figs. 11a-d show the detailed structure of the microstrip line networks 2, 3 embedded in the metallic carriers. The recesses (notches) 25 are made such that the

Microstrip lines 26 of the microstrip network 2, 3 run in closed metallic cavities. The microwave losses are thereby minimized.

 Since with finite thickness of the substrates (board) the

Microstrip lines 26 between the metallic layers remains a gap through which microwave power could escape, is also provided to provide the substrates with metallic vias 27 at the edges of the notches, so that the metallic supports are galvanically connected, and so the cavities be completely closed electrically. Are the vias 27 along the

Microwave lines 26 sufficiently tight, then no

Microwave power escape more.

 Preferably, the plated-through holes 27 are flush with the metallic walls of the cavity 25. In addition, if a thin, low-loss substrate (board material) is used, the electromagnetic properties of such a structure are similar to those of an air-filled coaxial line.

In particular, a very broadband microwave line is possible and parasitic higher modes are not

capable of propagation. In addition, even at very high GHz frequencies, the tolerance requirements are low.

With very thin substrates (for example, <20μπΐ) and correspondingly low use frequencies may be due to the

Through holes are also dispensed with, since even without vias then virtually no microwave power can escape through the then very narrow slots. The Hornstrahlereinkopplungen or -auskopplungen 6, 7 are integrated directly into the metallic carrier.

Fig. 12 shows the vacuum model of an exemplary 8 x 8 antenna module. The horns 1 are tightly packed and still leaves more than enough space for the

Microstrip line networks 2, 3, and for the

Waveguide terminations 28 of the single radiator 1 and the

Waveguide couplings 4, 5. A dielectric grating 20 is mounted in front of the aperture plane.

In a further advantageous embodiment, the

Waveguide networks, which couple the modules together made up of toothed waveguides. This has the advantage that toothed waveguide a much larger

Frequency bandwidth can have as conventional

Waveguide or can be designed specifically for different utility bands.

An exemplary network of doubly toothed waveguides is shown schematically in FIG. The rectangular ones

Waveguides are provided with symmetrical geometric constrictions 29, which are supplemented at the location of the power divider by vertical constrictions 30.

 The design of the toothed waveguide and the corresponding power divider can be done with the methods of numerical

Simulation of such components take place, depending on the

Requirements for the network.

It does not necessarily have to use double-toothed waveguides. Also, e.g. simply toothed or fourfold toothed waveguides are conceivable.

In one embodiment, not shown, the

Waveguides of the inter-modular waveguide networks completely or partially filled with a dielectric. Such fillings can with the same frequency use the space requirement in

Decrease significantly compared to unfilled waveguides. This results in very compact, space-optimized antennas, which are particularly suitable for applications on aircraft. It can be both standard waveguide and waveguide with geometric constrictions with a

Dielectric be filled.

In a further advantageous embodiment, the

Antenna equipped with a multilayer meander polarizer. Fig. 14 shows by way of example a position of such

Polarizers. In order to achieve axis ratios of the circularly polarized signals in the vicinity of 1 (0 dB), multilayer meander polarizers are used.

In an embodiment, not shown, several of the layers shown in Fig. 14 are arranged in parallel planes one above the other. Between the layers is a low-loss layer of foam material (e.g., Rohacell, XPS) having a thickness in the region of one quarter of a wavelength. With lower axle ratio requirements, however, fewer layers may be used. Likewise, more layers can be used if the axis ratio requirements are high.

 An advantageous arrangement is a 4-layer meander polarizer with the axial ratios of less than 1 dB can be achieved, which is usually sufficient in practice.

 The design of the meander polarizers depends on the useful frequency bands of the antenna system and can be done with methods of numerical simulation of such structures.

 The meandering lines 31 are in the embodiment of FIG. 14 at an angle of about 45 ° to the main axes of the antenna. This leads to incident, linearly polarized along a major axis signals are converted into circularly polarized signals. Depending on which main axis the

Signals are polarized linearly produces a left circularly polarized or a right circularly polarized signal.

Since the meander polarizer is a linear device, the process is reciprocal, i. in the same way, left- and right-circularly polarized signals become linear

converted polarized signals.

It is also conceivable, for the polarizers others

to use geometric structures as meanders. There are a variety of passive geometric conductor structures known, with which linear polarized can be converted into circularly polarized signals. It depends on the application, which structures are most suitable for the antenna.

As shown in FIG. 10, the polarizer 21 may be placed in front of

Aperture opening can be installed. This makes it possible in a relatively simple manner, the antenna for both linear

polarized signals as well as for circularly polarized

Signals without the need to change the internal structure.

In a further advantageous embodiment, the

Antenna equipped with a parabolic amplitude assignment, which is realized by a corresponding design of the power dividers of feed networks. Since the antenna pattern must be below a mask prescribed by regulations, such amplitude assignments can achieve much higher maximum permitted spectral EIRP densities in the transmit mode than without such assignments. This is especially important for antennas with a small aperture area

Advantage, since the maximum regulatory compliant spectral EIRP density is directly proportional to the achievable data rate and thus to the cost of a corresponding service.

In Fig. 15a such an amplitude assignment is shown schematically. The power contributions of the individual horns fall from the center of the aperture to the edge. In Fig. 15a, this is exemplified by different degrees of blackening (dark: high power contribution, bright: low power contribution). Here, the contributions to performance fall in both main axis directions (azimuth and elevation). This results in an approximately optimally matched to the regulatory mask antenna pattern for all angles of rotation ("skew").

 Depending on the requirements of the antenna diagram, however, it may also be sufficient to occupy the aperture in one direction only.

It is also conceivable that the amplitude occupancy only runs parabolically in the area around the antenna center, but increases again when approaching the edge, so that a closed curve exists around the antenna center and the power contributions of the individual radiators from the center of the antenna to each point of this curve fall off. Such amplitude assignments can

especially for non-rectangular antennas of advantage.

In Fig. 15b is an example of one in both

Main axis directions parabolic amplitude occupancy of a rectangular 64 x 20 Ka-band antenna following maximum

Regulatory compliant spectral EIRP density (EIRP SD) as a function of the angle of rotation around the

 Without parabolic coverage, the EIRP SD would be about 8 dB lower in the range of 0 ° skew to about 55 ° skew, and in the range of about 55 ° skew to about 90 ° skew by about 4 dB lower.

Figs. 16-18 show the basic structure of a series of antenna systems according to the invention with different

Functional scope in the form of block diagrams.

The antenna system, the basic structure of which is shown in FIG. 16, is particularly suitable for applications in the K / Ka band (reception band about 19.2 GHz -20.2 GHz, transmission band about 29 GHz -30 GHz), in which the polarizations of the transmitting and the Received signal are fixed and orthogonal to each other (ie, the polarization direction of the signals does not change).

 Since the K / Ka band typically employs circularly polarized signals, a polarizer 21 is initially provided. This is followed by an antenna field 32, which is constructed either of four-toothed ("quad-ridged") horn radiators or of dielectrically filled horn radiators

Aperture openings of the individual horns typically have dimensions less than 1 cm in this frequency range.

The antenna array 32 is according to the invention in modules

organized, each individual emitter over two

Polarizations has separate microstrip line couplings 33, which in turn after

Polarizations separated to two

 Microstrip network 36 are connected.

Since the polarization of the transmit and receive signals is fixed and is typically orthogonal to one another, it is provided here that the microstrip line network 36 has one polarization on the transmit band and the

 Microstrip line network 36 of the other polarization interpreted on the receiving band.

 This has the advantage that the microstrip network

36 of the receiving band can be designed for minimum losses, and thus the G / T of the antenna becomes optimal.

In the exemplary construction of FIG. 16, the polarizer 21 is oriented such that the signals in the transmission band 34 are right-handed circular and the signals in the reception band 35 are circularly polarized left-handed.

The signals separated by polarization and frequency band of the two microstrip line networks 36 of the individual modules are now using microstrip line-to-waveguide couplings

37 coupled in two waveguide networks 38.

Again, it is envisaged that the two waveguide networks

38 to the appropriate band that they should support.

For example, different waveguide cross sections can be used for the receive band waveguide network and the transmit band waveguide network. In particular, enlarged waveguide cross sections can be used, which greatly reduce the dissipative losses in the waveguide networks and thus can significantly increase the efficiency of the antennas. Furthermore, a receive band frequency filter 39 is provided to protect the low noise receive amplifier, which is typically mounted directly on the receive band output of the antenna, from being overdriven by the strong transmit signals.

In order to achieve the regulatory sideband suppression in the transmission band is also an optional

Transmission band filter 40 is provided. This is e.g. then

required if a transmit band power amplifier (HPA), not shown, does not have a sufficient filter at its output.

The structure shown in Fig. 16 of an antenna system according to the invention has a further, in particular for

Satellite antennas, very important advantage. Since the transmit band feed network and the receive band feed network are completely separated both at the microstrip line level and at the waveguide level, it becomes possible to use different amplitude assignments for the two networks.

 Thus, e.g. the receive band feed network is homozygously occupied, i. the power contributions of all horns of the antenna are equal in the receiving band and all power dividers both at the level of the receiving band

Microstrip line network as well as at the level of the receive waveguide network are symmetrical 3dB power dividers when the feed network is constructed as a complete and fully symmetric binary tree.

Because homogeneous amplitude assignments to the maximum possible

Antenna gain is achieved so that the antenna in the receiving band is maximally efficient and the ratio of antenna gain and noise G / T of the antenna is maximum.

On the other hand, the transmit band feed network can thus be parabolic regardless of the receive band feed network

Amplitude assignment are provided that the regulatory compliant spectral EIRP density is maximum.

Although reduced by such parabolic

 Amplitude assignments of the antenna gain, but this is not critical, because this is due to the design limited only to the transmission band and does not affect the receiving band.

The essential features of satellite antennas, especially satellite antennas of small size, are the G / T and the maximum regulatory compliant spectral EIRP density. The G / T is directly proportional to the data rate that can be received via the antenna. The maximum regulatory EIRP spectral density is directly proportional to the data rate that can be transmitted with the antenna.

With antenna systems which are constructed according to FIG. 16, both features can be optimized independently of each other.

For very small satellite antennas, this results in another advantage. Namely, there is a problem that the width of the main beam in the reception band can become so large that not only signals of the target satellite but also signals of adjacent satellites are received. The signals from neighboring satellites then effectively act like one

additional noise contribution, which leads to a considerable

Degradation of the effective G / T can lead.

In antenna systems according to the invention, which are constructed in accordance with FIG. 16, this problem can at least be solved

be partially solved. Namely, if e.g. the Empfangsband- feed network not homogeneous amplitude occupied, but

hyperbolic amplitude, then decreases the width of the

Main beam of the antenna. Hyperbolic amplitude assignments are characterized by the fact that the power contributions of the individual radiators of the antenna field increase from the middle to the edge.

By means of a hyperbolic amplitude occupancy, at least in a partial area of the antenna system, it is possible to achieve that the intensity of the interference signals received by neighboring satellites by the antenna decreases and the effective G / T increases in such an interference scenario.

 In Fig. 17, the structure of an inventive

Antenna system shown in the form of a block diagram showing the simultaneous operation with all four possible

Polarization combinations of the signals allowed.

The antenna system initially consists of an antenna array 41 of broadband, dual polarized horns, e.g. fourfold toothed horns, which are organized according to the invention in modules.

In contrast to the embodiment shown in FIG. 16, however, no polarizer is used here, but each horn emits two orthogonal linear polarized signals, which however, when operating with circularly polarized signals, will also be the complete one

Information included. The essential difference from the embodiment in FIG. 16 consists in the fact that at the level of the feed networks, it is not separated into a receive band and a transmit band feed network, but the signals are separated only according to their different polarization.

All signals 42 of the same polarization are after

Decoupling 33 from the antenna field in the first

Microstrip line network merged all signals of orthogonal polarization 43 in the second

Microstrip line network.

 The two microstrip line networks 36 are designed such that they both the transmission band and the

Support reception band. An optimization of

Food networks on one of the tapes is here only

limited possible. However, all four polarization combinations are simultaneously available for this.

While the microstrip line networks 36 of the present invention are typically already broadband by design (coaxial line like construction)

Receive and transmit band can support simultaneously, after the transition 37 microstrip-to-waveguide waveguide networks 44, if very large bandwidths

are required to be specially designed for this purpose. This can e.g. done by the toothed waveguide described in Fig. 13. However, it is also possible, e.g. to use dielectric filled waveguide.

For the separation of receive band and transmit band signals, two frequency diplexers 45, 46 are provided, one for each polarization. The frequency diplexers 45, 46 are e.g.

low attenuation waveguide diplexer.

When operating with linearly polarized signals, all linear polarization combinations are then simultaneously available at the output of the two diplexers: two orthogonally polarized linear signals both in the reception band 49 and in the transmission band 50.

When operating with circularly polarized signals

additionally two 90 ° hybrid couplers 47, 48, one for the

Reception 49 and one for the transmission band 50, provided with the aid of which at the output of the frequency diplexer 45, 46 present linear polarized signals, circular

polarized signals can be combined. The 90 °

Hybrid couplers 47, 48 are e.g. low-attenuation

Waveguide coupler. At the output of the two 90 ° hybrid couplers 47, 48 are then all four possible circularly polarized signals (in

Receive 49 and transmit band 50 each right-handed and left-handed circular) simultaneously.

If appropriate RF switches and / or RF couplers are installed between diplexer 45, 46 and 90 ° hybrid couplers 47, 48 and thus the linearly polarized signals are coupled out, then the antenna system can also be used for simultaneous operation with four different linearly and four different circularly polarized signals be used. Also many other combination possibilities and the appropriate ones

Antenna configurations are possible.

 In Fig. 18, the structure of an inventive

Antenna system in the form of a block diagram, which has the same functional scope as that in FIG. 16

has shown antenna, but is organized differently.

In the structure of Fig. 18 is for operation with circular

Polarized signals used a polarizer 21 instead of the 90 ° hybrid coupler 47, 48 of the structure of FIG. 17.

 The feed networks 36, 44 process again two orthogonal polarizations separated from each other (in this case left-circular and rechtszikular) and are each designed correspondingly broadband for the receiving band and the transmission band.

At the output of the frequency diplexers 45, 46 are then directly the four polarization combinations of circularly polarized signals simultaneously. At the frequency-diplexer 45 for the first circular polarization the signal in the receive and transmit band, at

Frequency diplexer 46 for the second (to the first orthogonal) circular polarization the signal in the receive and transmit band.

By using two 90 ° hybrid couplers (not shown), which are connected to the diplexers 45, 46 similarly to the construction of FIG. 17, the structure according to FIG. 18 can also be designed for the operation of linearly polarized signals, or it is possible with the corresponding circuit matrix a simultaneous operation with circular and linearly polarized signals.

The advantage of the construction of FIG. 18 is that when operating with circularly polarized signals no 90 °

Hybrid couplers are needed. This may vary depending on the application e.g. Save space or weight. Also can be under

Circumstances arise cost advantages.

The advantage of the construction according to FIG. 17, however, lies in the fact that when operating with circularly polarized signals Axial ratio of the circularly polarized signals on the respective power contributions at the input of the 90 ° hybrid coupler 47, 48 is in principle freely adjustable.

This can e.g. then be advantageous if the antenna is operated under a radome. It is known that, especially for high GHz frequencies, radomes through the radome material and the radome curvature can have polarization anisotropies that cause the axis ratio of circularly polarized signals to be greatly altered as it passes through the radome.

This effect causes the

 Cross polarization isolation can fall sharply, which can severely degrade the achievable channel separation and ultimately leads to a degradation of the achievable data rate.

A structure of the antenna of Fig. 17 now allows this

Axis ratio of the circularly polarized signals, e.g. in the transmission mode, adjust so that a subsequent, caused by the Radomdurchgang polarization distortion

is compensated. A degradation of

 Cross-polarization isolation thus does not take place effectively.

Claims

claims

1. antenna system,

with at least four individual radiators (18), the

Single radiators (18) are horns, which are at least partially filled with a dielectric (19).

 2. Antenna system according to claim 1, characterized in that the horn radiator (18) is completely filled with dielectric (19).

3. Antenna system according to one of the preceding claims, characterized in that the dielectric constant of the dielectric (19) is greater than or at least equal to the ratio of

Free space wavelength of the lowest usable frequency for

Free space wavelength of a reference frequency, wherein the reference frequency is in the transmission band of the antenna system.

4. Antenna system according to one of the preceding claims, characterized in that the dielectric constant of the dielectric (19) is between 1.8 and 3, preferably between 1.9 and 2.1.

5. Antenna system according to one of the preceding claims, characterized in that the dielectric (19) per filled

Horn radiator (19) consists of at least three parts between which the feed (2, 3) of the horn (18) is arranged.

6. Antenna system according to claim 2, characterized in that the horns (18) are aligned in a plane and an at least approximately rectangular aperture

possess, whose longer edge is smaller than 1.5 cm, preferably smaller than 1 cm.

7. Antenna system according to one of the preceding claims, characterized in that the horn radiator (18) as stepped

Horn are executed.

8. Antenna system according to claim 7, characterized in that in the horns (18) in at least two

at least one step is mounted opposite walls.

9. Antenna system according to one of the preceding claims, characterized in that the horn radiators (18) have an approximately rectangular aperture, both of which

Edge lengths are smaller or at most equal to the wavelength of a reference frequency, wherein the reference frequency is in the transmission band of the antenna.

10. Antenna system according to claim 9, characterized in that the horns (18) are designed stepped, with at least a first rectangular cross section, the

Opening has a longer edge, which is greater than or equal to half the ratio of the free space frequency of the lowest usable frequency to the root of the dielectric constant of the dielectric (19), and the horns (18) have at least a second rectangular cross section, the opening of a longer Edge, which is greater than or equal to at most half the ratio of the free space wavelength of the highest useful frequency to the root of the dielectric constant of the dielectric (19).

11. Antenna system according to one of the preceding claims, characterized in that the horns (18) support a first and a second polarization and the two

Polarizations are orthogonal to each other.

12. Antenna system according to claim 11, characterized in that the first and second polarization are linear polarizations.

 13. Antenna system according to claim 11, characterized in that the horn radiator (18) for the first polarization with a first microstrip line network (2) and for the second polarization with a second

 Microstrip line network (3) and the microstrip line network (2) for the first polarization and the microstrip line network (3) for the second

Polarization respectively separate microstrip line networks (2, 3) form.

14. Antenna system according to one of the preceding claims, characterized in that at least part of the

Horn radiator (18) is dimensioned so that the distance of the phase centers of two directly adjacent horns (18) smaller or at most equal to the wavelength of a

Reference frequency is, which lies in the transmission band of the antenna system.

 15. Antenna system according to one of the preceding claims, characterized in that a part or all horns (18) are equipped with a dielectric cross-septum and / or a dielectric lens.

16. Antenna system according to one of the preceding claims, characterized in that the Mikrostreifenleitungs- networks (2, 3) are constructed as binary trees, so that the horns (18) are fed in parallel.

17. Antenna system according to one of the preceding claims, characterized in that the Mikrostreifenleitungs- networks (2, 3) are located on a thin substrate and in cavities (25) are guided, whose walls at least

partially electrically conductive.

 18. Antenna system according to one of the preceding claims, characterized in that the antenna system

various layers (22, 23a, 23b, 23c, 24) and the microstrip line network (2) of one polarization and the microstrip line network (3) of the other

Polarization are separated from each other between the layers (23a-c) are located.

 19. Antenna system according to claim 18, characterized in that the different layers (22, 23a, 23, b, 23c, 24) are made of metal, the microstrip lines (26) of the

Microstrip line networks (2, 3) are guided in cavities, which are designed as notches (25) in the layers (23a, 23, b, 23c), wherein in each case a notch (25) above and below the microstrip line (26).

 20. Antenna system according to claim 17, characterized in that a substrate of the microstrip line networks (2, 3) with metallic plated-through holes (27) is provided such that an electrical contact between the walls of the cavities (25) is produced.

 21. Antenna system according to one of the preceding claims, characterized in that the waveguide networks (9, 10) with frequency diplexers (45, 46) are equipped which the signals of the transmission band (34) from the signals of the

Disconnect receive band (35) so that they are separated

can be further processed.

 22. Antenna system according to one of the preceding claims, characterized in that the dimensions of

Microstrip lines of the microstrip line networks (2, 3) are chosen so that both the transmission band and on the receiving band of the antenna system are supported.

23. Antenna system according to one of claims 1 to 21, characterized in that the dimensions of

 Microstrip lines of the microstrip line networks (2, 3) are chosen so that the first microstrip line network (2) is designed for the receiving band of the antenna, and the second microstrip line network (3) on the

Transmission band of the antenna is designed.

24. Antenna system according to claim 23, with a homogeneous amplitude occupancy in the receiving band, so that the power contributions of all Horn antenna (18) are approximately equal, and with an amplitude occupancy in the transmission band, so that the power contributions of at least a portion of the horns (18) are different from each other.

25. Antenna system according to one of the preceding claims, with 90 ° hybrid coupler (47, 48) in the feed networks, so that from linearly polarized signals circularly polarized

Signals can be generated.

 26. Antenna system according to one of the preceding claims, characterized in that the horn radiator (18) with a polarizer (21) for receiving and transmitting circular

polarized signals are equipped.

 27. Antenna system according to claim 26, characterized in that the polarizer (21) is designed as a multi-layer meander polarizer and is mounted in front of the apertures of the horn (18).

28. Antenna array with a plurality of antenna systems according to one of the preceding claims, which are coupled together with waveguide networks (9, 10).

29. An antenna array according to claim 28, characterized in that a first waveguide network (9) combines all the signals of the first polarization and a second waveguide network (10) merges all the signals of the second polarization.

30. Antenna field according to one of claims 28 to 29, characterized in that at least a part of the

 Waveguide networks (9, 10) via at least one geometric constriction along the propagation direction of

electromagnetic wave features.

31. Antenna array according to claim 30, characterized in that at least a part of the Hohleiternetzwerke (9, 10) is designed as a single-ridged or double-ridged waveguide.

32. Antenna array according to one of claims 28 to 31, characterized in that at least part of the waveguide networks (9, 10) is completely or partially filled with a dielectric.

33. Antenna field according to one of claims 28 to 32, characterized in that the dimensions of the waveguides of

Waveguide networks (9, 10) are selected so that both the transmission band and on the receiving band of the antenna system are supported.

34. Antenna field according to one of claims 28 to 32, characterized in that the dimensions of the waveguide of

Waveguide networks (9, 10) are selected so that the first waveguide network (9) is designed for the receiving band of the antenna system, and the second waveguide network (10) are designed for the transmission band of the antenna system.

 35. Antenna array according to claim 34, with homogeneous

Amplitude assignment in the receiving band, so that the

Power contributions of all horns (18) are approximately equal, and with an amplitude occupancy in the transmission band, so that the power contributions of at least a portion of

Increase the horn (18) from the edge to the center of the antenna field.

36. Antenna field according to one of claims 28 to 35, characterized in that the amplitude assignment at least in the direction in which the horns (18) are tight, at least in the transmission frequency band follows an approximately parabolic curve, so that the power contributions of the horns (18) , which are at the edge of the antenna field are smaller than the

Power contributions of the horns (18), which lie in the middle of the antenna field.

 37. Antenna array according to one of claims 28 to 36, characterized in that the waveguide networks (9, 10) are constructed as binary trees, so that the individual

Antenna systems are powered in parallel.