EP2870658B1 - Antenna system for broadband satellite communication in the ghz frequency range, comprising horn antennas with geometrical constrictions - Google Patents

Description

The invention relates to an antenna system for broadband communication between earth stations and satellites, in particular for mobile and aeronautical applications.

The demand for wireless broadband data transmission channels with very high data rates, especially in the field of mobile satellite communications, is steadily increasing. However, especially in the aeronautical field, there is a lack of suitable antennas, which in particular can fulfill the conditions required for mobile use, such as small dimensions and low weight. For directional, wireless data communication with satellites (for example in the Ku or Ka band), there are also extreme demands on the transmission characteristics of the antenna systems, since interference with neighboring satellites must be reliably excluded.

In aeronautical applications, the weight and size of the antenna system are very important because they reduce the payload of the aircraft and cause additional operating costs.

The problem therefore is to provide antenna systems that are as small and lightweight as possible, which nevertheless satisfy the regulatory requirements for transmitting and receiving operation when operating on mobile carriers.

For example, the regulatory requirements for broadcasting arise from the standards 47 CFR 25.209, 47 CFR 25.222, 47 CFR 25.138, ITU-R M.1643, ITU-R S.524-7, ETSI EN 302 186 or ETSI EN 301 459. All these regulatory provisions are intended to ensure that no interference of adjacent satellites can occur in the directional transmission mode of a mobile satellite antenna. For this purpose, envelopes (envelopes or masks) of maximum spectral power density are typically defined as a function of the distance angle to the target satellite. The for a given distance angle given values may not be exceeded in the transmission mode of the antenna system. This leads to stringent requirements for the angle-dependent antenna characteristic. As the distance from the target satellite increases, the antenna gain must drop sharply. This can be achieved physically only by very homogeneous amplitude and phase assignments of the antenna. Typically, therefore, parabolic antennas are used which have these properties.

However, parabolic mirrors are poorly suited for most mobile applications, especially on airplanes, because of their size and because of their circular aperture. In commercial aircraft, for example, the antennas are mounted on the fuselage and therefore may only have the lowest possible height because of the additional air resistance.

Antennas which are designed as sections of paraboloid ("banana-shaped mirror"), although possible, but have geometriebedingt only a very low efficiency.

Antenna fields, which are constructed of individual radiators and have suitable feed networks, however, can be performed in any geometry and any length to aspect ratio without the antenna efficiency suffers. In particular, antenna fields of very low height can be realized.

In the case of antenna fields, however, especially when the reception frequency band and the transmission frequency band are far apart (such as in the Ka band with reception frequencies at about 18 GHz - 21 GHz and transmission frequencies at about 28 GHz - 31 GHz), the problem arises that the Single radiators of the fields have to support a very large bandwidth.

It is well known that horns are by far the most efficient single emitters in fields. In addition, horns can be designed broadband.

In the case of antenna fields which are constructed from horn radiators and are fed with pure waveguide networks, however, the known problem of significant parasitic sidelobes (so-called "grating lobes" or "grating lobes") occurs in the antenna pattern. These grating lobes are caused by the fact that the beam centers (phase centers) of the antenna elements which form the antenna field, due to the dimension of the Hohleiternetzwerke by design too far away from each other. This can, especially at frequencies above about 20 GHz, at certain beam angles to the positive interference of the antenna radiator and thus lead to the unwanted emission of electromagnetic power in unwanted solid angle ranges.

If reception and transmission frequencies are also far apart in terms of frequency, and if the distance between the beam centers has to be designed according to the minimum useful wavelength of the transmission band, then the horns regularly become so small that the reception band can no longer be supported by them.

In Ka band, for example, the minimum useful wavelength is only about 1cm. So that the radiation elements of the antenna field are dense, so no parasitic side lobes (grating lobes) occur, the aperture area of a square horn may only be about 1cm x 1cm. However, conventional horns of this size have only a very low performance in the reception band at approx. 18 GHz - 21 GHz, since they have to be operated close to the cut-off frequency because of the finite aperture angle. The Ka-receiving band can no longer support such horns or their efficiency decreases very much in this band.

In addition, the horns are generally intended to support two orthogonal polarizations, which further restricts the geometrical margin, since an orthomode transducer, so-called transducers, becomes necessary at the horn output. An embodiment of the orthomode signal converter in waveguide technology fails regularly because at higher GHz frequencies not enough space is available.

If the horns packed tightly in fields, then another problem is that in the available space behind the horn field no more efficient feed networks can be accommodated.

It is known that feed networks for fields of horns, which are implemented in high-power technology, produce only very small dissipative losses. In the optimal case, the individual horns of the fields are fed by waveguide components and the entire feed network also consists of waveguide components. In the case that the receiving and the transmitting band are far apart in terms of frequency, however, the problem arises that conventional waveguides can no longer support the then required frequency bandwidth.

For example, in the Ka band the required bandwidth is more than 13 GHz (18 GHz - 31 GHz). Conventional rectangular waveguides can not efficiently support such a large bandwidth.

1. 1. regulatorisch konformes Antennendiagramm ohne parasitären Nebenkeulen (Gitterkeulen) im Sendefrequenzband, das den Betrieb der Antenne mit maximaler spektraler Leistungsdichte erlaubt,
2. 2. Hohe Antenneneffizienz sowohl im Empfangsband als auch im Sendeband auch bei kleinen Einzelstrahlerdimensionen,
3. 3. Effiziente Speisenetzwerke, welche einen möglichst geringen Bauraum in Anspruch nehmen und möglichst geringe dissipative Verluste erzeugen,
4. 4. Möglichst kompakter und raumsparender Aufbau der Antenne bei gleichzeitig möglichst hoher Antenneneffizienz.

This results in the following problems for mobile, in particular aeronautical satellite antennas of small size, which must be solved simultaneously:

1. 1. regulatory compliant antenna diagram without parasitic side lobes (grating lobes) in the transmit frequency band, which allows the operation of the antenna with maximum spectral power density,
2. 2. High antenna efficiency both in the reception band and in the transmission band, even with small individual radiator dimensions,
3. 3. Efficient feed networks, which take up the smallest possible installation space and generate the lowest possible dissipative losses,
4. 4. The most compact and space-saving design of the antenna with the highest possible antenna efficiency.

If these problems are solved by a suitable arrangement, then even if only a limited space for a small antenna is available, a broadband powerful system can be made available.

It is known that with antennas which are designed as fields of individual emitters, grating-free free antenna diagrams can be achieved if the phase centers of the individual emitters are less than one wavelength of the maximum useful frequency. In addition, it is known that the side lobes of the antenna diagram can be suppressed by parabolic amplitude assignments of such antenna fields (eg JD Kraus and RJ Marhefka, "Antennas: for all applications", 3rd ed., McGraw-Hill series in electrical engineering, 2002 ). By special amplitude assignments, an antenna pattern which is optimally adapted to the regulatory mask for a given antenna size can be achieved (eg DE 10 2010 019 081 A1 ; Seifried, Wenzel et. al.).

According to the US Pat. No. 6,271,799 B1 US 6,360,860 discloses an antenna device with a dual polarized four-comb antenna horn having an electrically conductive line with first and second opposite ends along a horn axis. On an inside of the electrically conductive Line four electrically conductive webs are arranged. A printed circuit board containing a dielectric substrate is connected across the first end of the dual polarized four-comb antenna horn and across the horn axis. Further, an electrically conductive pattern is formed on the dielectric substrate defining feed elements for the dual polarized four-comb antenna horn.

The DE 10 2010 019 081 A1 discloses an antenna for broadband satellite communication consisting of a field of primary horns interconnected by a waveguide feed network.

The Korean patent application KR20100072693 describes an antenna for improving the transmission / reception module having a horn antenna unit and a plurality of mode change units, wherein the horn antenna unit has a rectangular opening side and a grating is formed on the inner surface of the horn antenna unit. The mode changing unit is formed inside the horn antenna unit as a staircase shape.

The object of the invention is to provide a broadband antenna system in the GHz frequency range, in particular for aeronautical applications, which allows a regulatory compliant transmission mode with maximum spectral power density at minimum dimensions and at the same time has a high antenna efficiency and low intrinsic noise in the receive mode.

This object is achieved by the antenna system according to claim 1 and the antenna array according to claim 13.

According to the invention, the antenna system consists of at least four horns, the horns supporting two mutually orthogonal linear polarizations and are equipped in both polarization planes with constrictions. By narrowing the horns in the two polarization planes with symmetrical geometrical constrictions along the propagation direction of the electromagnetic wave (provided with "teeth"), the bandwidth of the horns can be greatly increased. This makes it possible, even broad transmit and receive bands or in large frequency spacing, as with the Ka-band, to use existing transmit and receive tapes.

So that the individual serrated horns can still be optimally operated with widely spaced useful frequency bands, both the horn radiators and the constrictions will be designed stepwise. By suitable choice of the height and width of the stages of the horn and the stages of the constrictions Horn horns can then be optimally adapted to the Nutzfrequenzbänder impedance.

The distance between the opposite, stepped constrictions and the opening of the associated horn cross-section is then selected in a preferred embodiment such that this distance decreases from stage to stage from the aperture opening to the horn end and at each stage to the respective distance and to the respective horn opening lower Cut-off frequency is less than the lowest usable frequency.

To achieve high cross-polarization decoupling, it is also advantageous if the horns are designed to support two orthogonal linear polarizations. With such horns, insulations far exceeding 40 dB can be achieved. Especially with signal codings with high spectral efficiency such isolation values are required.

The lower limit frequency belonging to the respective distance and to the respective horn opening can be determined by numerical simulation methods.

Thus, in addition no parasitic side lobes (grating lobes) occur in the antenna diagram of the antenna system, the distance of the phase centers of directly adjacent horn is smaller or at most equal to the wavelength λ _{s of} the highest transmission frequency below which, for regulatory reasons, no grating lobes may occur.

It is also advantageous to choose the aperture of the horns rectangular, and preferably so that both edge lengths are less than or equal to λ _S. The available aperture area is thus optimally utilized and a maximum antenna gain is achieved.

gilt, wobei λ_E die Wellenlänge der niedrigsten Nutzfrequenz bezeichnet und p₁ zwischen 0,3 und 0,4 und p₂ zwischen 0,25 und 0,35 liegen.For antenna systems which consist of several horns has proved to be advantageous if the gradation of the horns and the stages of the constrictions are chosen so that at least for a part of the stages, for the distance d _{i of} the i-th stages of two opposite Constrictions and the associated edge length a _i of Hornstrahlerquerschnitts at the i-th stage (see. Fig. 4d ) $$d_i\le {\mathrm{<figure-callout\; id="1"\; label="p"\; filenames="imgf0001.png,imgf0003.png"\; state="\{\{state\}\}">p</figure-callout>}}_1\frac{2\pi }{{\lambda }_e}{a_i}^2-p_2{a}_i$$

where λ _E denotes the wavelength of the lowest usable frequency and p ₁ between 0.3 and 0.4 and p ₂ between 0.25 and 0.35.

In this case, not only a good impedance matching of the horn to the usable frequency bands can be achieved, but also a good impedance matching of the antenna system as a whole. This is true even if the usable frequency bands are far apart.

As has also been shown, especially for K / Ka band frequencies (reception band: approx. 18GHz - 21GHz, transmission band approx. 28GHz - 31GHz) a very good impedance matching can be achieved, if p ₁ = 0.35, p ₂ = 0 , 29 and 0.5cm <a ₀ <1cm , where a ₀ denotes the longer edge of the rectangular aperture of the horn.

und gleichzeitig $${\lambda }_S\ge a_0\ge \frac{{\lambda }_S}{2}$$

gilt, wobei hier p₁ =0,35 und p₂ =0,29 sind und λ_S die Wellenlänge der höchsten Nutzfrequenz bezeichnet.In a further advantageous embodiment, the apertures of the horns are approximately square with edge length a ₀ . Then the horns are dense along two orthogonal directions and the antenna system is very well impedance matched to the useful frequency bands if for at least a portion of the stages, for the distance d _{i of} the ith stages of two opposing constrictions and the associated edge length a _{i of} the horn cross section at the i-th stage (cf. Fig. 4d ) $$d_i\le {\mathrm{<figure-callout\; id="1"\; label="p"\; filenames="imgf0001.png,imgf0003.png"\; state="\{\{state\}\}">p</figure-callout>}}_1\frac{2\pi }{{\lambda }_e}{a_i}^2-p_2{a}_i$$

and at the same time $${\lambda }_S\ge a_0\ge \frac{{\mathrm{<figure-callout\; id="2"\; label="\lambda \; S"\; filenames="imgf0001.png,imgf0005.png"\; state="\{\{state\}\}">\lambda \; </figure-callout>}}_S}{2}$$

where p ₁ = 0.35 and p ₂ = 0.29 and λ _S denotes the wavelength of the highest usable frequency.

If conditions (1) and (3) are fulfilled for the antenna system's horns, then an antenna system results which does not have any parasitic sidelobes (grating lobes) in any section through the antenna diagram and can also have a maximum antenna gain in all usable frequency bands. Such antenna systems are particularly advantageous for aeronautical applications because they allow global deployment.

According to an advantageous further development of the invention, the individual radiators support a first and a second polarization and the two polarizations are mutually orthogonal. According to a further advantageous development of the invention, the first and second polarization are linear polarizations.

The signals of the two orthogonal polarizations are carried in separate feed networks, which has the advantage that with the aid of corresponding components, such as e.g. Polarizers or 90 ° hybrid couplers, both linearly polarized signals and circularly polarized signals can be sent or received.

So that the antennas can have the smallest possible size and nevertheless a regulatory compliant transmission mode with maximum spectral power density is possible, it is also provided according to an advantageous development of the invention that at least a portion of the individual radiator is dimensioned so that the distance of the directly adjacent individual radiator Phase centers of the individual radiator is less than or equal to the wavelength of the highest transmission frequency at which no parasitic sidelobes (grating lobes) may occur (reference frequency in the transmission band).

If at least four adjacent individual radiators are also located in different directly adjacent modules, then at least one direction is defined by the antenna field, so that for this direction the distance of the phase centers of the individual radiators is less than or equal to the wavelength of the highest transmission frequency, in which no parasitic side lobes ( grating-lobes) may occur.

In this direction, preferably along a straight line through the antenna field, directly adjacent individual radiators are then close, so that no parasitic side lobes ("grating lobes") can occur in the corresponding section through the antenna pattern. Otherwise, would These grating awards lead to a strong reduction of the regulatory allowed spectral power density.

In principle, all known radiation elements which support two orthogonal polarizations come into consideration as individual radiators. These are e.g. rectangular or round horns.

gilt und N die Gesamtzahl der Einzelstrahler ist. Solche rechteckigen Module lassen sich in platzsparender Weise zu Antennenfeldern zusammensetzen. Zudem können die rechteckigen Module in relativ einfacher Weise mit binär aufgebauten Mikrostreifenleitungsnetzwerken gespeist werden.It is also advantageous if the modules have an at least approximately rectangular geometry, ie N _i = n ₁ xn _k single radiators, where N _i , n, i, l, k are even numbers, ${\displaystyle \underset{i}{\Sigma }{N}_i=N}$

and N is the total number of individual emitters. Such rectangular modules can be assembled in a space-saving manner to antenna fields. In addition, the rectangular modules can be fed in a relatively simple manner with binary microstrip networks.

In order to realize antennas with the lowest possible dissipative losses, it is advantageous to design the individual radiators as horn radiators, which belong to the lowest-loss antennas. Both horns with rectangular and with a round aperture can be used. If grating lobes are not to occur in any section through the antenna pattern, square aperture horns are advantageous, the size of the aperture opening then being chosen such that the spacing of the phase centers of directly adjacent horns is less than or equal to the wavelength of the highest transmission frequency as the reference frequency. in which no grating praise may occur.

The horns (horns) can be advantageously carried out as a dielectrically filled horns. According to the dielectric properties of the filling, the effective wavelength in the horns increases and these are able to support much larger bandwidths than would be the case without filling. Although dielectric fillings lead to parasitic losses through the dielectric, these losses remain comparatively small, especially in the case of very small horns. For example, e.g. For applications in the Ka band, a dielectric filling with a dielectric constant of about 2 is sufficient. With horns only a few centimeters deep, this leads to losses of <0.2 dB when using suitable materials.

If the transmitting and receiving bands are far apart in terms of frequency then, according to a further advantageous embodiment of the invention, the horns are designed as stepped horns ("stepped horns"). By adjusting the width and length of the steps, as well as the number of stages, can then the antenna can be optimally adapted to the respective usable frequency bands.

A further improvement in the reception power, in particular in the case of very small horn radiators, can be achieved by equipping the individual horn radiators with a dielectric cross-septum or a dielectric lens. The insertion loss (S ₁₁ ) in the receiving band can be significantly reduced by such structures, even if the aperture areas of the individual radiators are already so small that a free-space wave without these additional dielectric structures would already be almost completely reflected.

Since in parallel-fed individual radiators, the dissipative losses, such as by a dielectric filling occur only once, the horns of the antenna array are fed in parallel according to a further advantageous development of the invention. This is most effective when the microstrip lines and the waveguides are constructed as binary trees, since the number of power dividers needed in the general case of arbitrary values of the total number of individual radiators N and arbitrary values of the number of individual radiators in a module N _i becomes so minimal.

The binary trees are in the general case neither completely nor completely symmetrical.

Applies, however, according to an advantageous development of the invention for all modules of the antenna system or at least for most of the modules N _i = 2 ^{n i} , with n _{i of} an integer, then the number of required power dividers can be further reduced, because then at least part of the binary trees is complete.

It is particularly favorable if, in addition, N = 2 ⁿ , where n is an integer. Then the feed networks of the antenna system can be designed as complete and fully symmetrical binary trees and all individual emitters can have equal length feeder lines, ie also similar attenuations.

It is also advantageous if the microstrip lines are located on a thin substrate and are guided in closed metallic cavities, wherein the cavities are typically filled with air. A substrate is typically referred to as being thin if its thickness is smaller than the width of the microstrip lines.

This coaxial line-like structure with typically air as a filling leads to comparatively low-loss high-frequency lines. Thus, it has been found that the dissipative losses of such lines e.g. at Ka-band frequencies are only about a factor of 5 to 10 higher than the losses of waveguides. Since these lines are used only for relatively short distances, the absolute losses remain relatively small. The noise contribution of such lines to the inherent noise of the system thus remains relatively small.

The production of densely packed antenna systems can be greatly facilitated by being constructed of multiple layers and by having the microstrip feed networks of the two orthogonal polarizations between different layers. The modules of the antenna system can then be assembled from a few layers. Advantageously, the layers of aluminum or similar electrically conductive materials, which can be structured with the known structuring method (milling, etching, lasers, wire erosion, water cutting, etc.). The microstrip line networks are patterned on a substrate by known etching techniques.

Advantageously, the cavities through which the microstrip lines are routed are structured directly with the metallic layers. If the cavities are designed as notches or depressions in the metal layers lying above and below the microstrip line, then the microstrip line lies together with its substrate in a cavity which consists of two half shells. The walls of the cavity can be electrically closed by providing the substrate with electrical vias. In such arrangements, "fences" by Vias can almost completely prevent the loss of electromagnetic power.

If the reception and transmission bands of the antenna are very far apart in terms of frequency, then it may be the case that standard hollow conductors (rectangular waveguides) can no longer support the required bandwidth. In this case, it is advantageous to provide the waveguides along the propagation direction of the electromagnetic wave with geometric constrictions (constrictions). By such constrictions, the useful bandwidth can be greatly increased. The number and arrangement of constrictions depend on the design of the antenna system.

For very large useful bandwidths so-called double-ridged waveguides are advantageous, which can have a significantly larger bandwidth than standard waveguide. These Waveguides have a geometric constriction parallel to the supported polarization direction, preventing the formation of parasitic higher modes.

At very high useful frequencies or very close individual emitters, an advantageous development of the invention is that dielectrically filled waveguides are used for the waveguide supply networks. Such waveguides require much less space than air-filled waveguide. Depending on the requirements of the installation space, a part or a whole waveguide network may additionally consist of dielectric filled waveguides. Also a partial filling is possible.

For further processing of the signals, e.g. By coupling a low noise amplifier (LNA) to the receive feed network and / or a high power amplifier (HPA) to the transmit feed network, it may be advantageous to feed the frequency diplexer feed networks equip. Such frequency diplexers separate the reception from the transmission band. In particular waveguide diplexers are advantageous because they can achieve a very high isolation and are also very low attenuation.

At which point the frequency diplexers are inserted into the feed networks depends on the particular application. For example, e.g. conceivable that each module of the antenna array is equipped with a diplexer directly at its output or input. At the input and output of these diplexers are then all signal combinations in pure form: polarization 1 in the receiving band, polarization 2 in the receiving band, polarization 1 in the transmission band and polarization 2 in the transmission band. The modules can then be interconnected by four corresponding waveguide networks. This embodiment has the advantage that the waveguide feed networks do not have to be very broadband in terms of frequency because they each only have to be suitable for signals of the receiving or transmitting band.

However, it is also conceivable that the frequency diplexers are mounted only at the input or output of the waveguide networks. Such an embodiment saves space, but typically requires a broadband design of the waveguide networks.

For applications in which to transmit or receive in different polarizations, or in applications in which the polarization of the transmit or the receive signal changes dynamically ("polarization diversity"), it is advantageous if both the intra-modular Microstrip lines networks as well as the inter-modular waveguide networks are designed so that they can simultaneously support the transmitting and the receiving band.

If the antenna is provided with frequency diplexers which are connected to a suitable radio frequency switching matrix, then dynamic switching between the orthogonal polarizations is possible (polarization switching).

Such embodiments are particularly advantageous when the antenna is to be used in satellite services, which work with the so-called "spot beam" technology. In "spot beam" technology, coverage areas (cells) of relatively small area are formed on the earth's surface (typical diameter in the Ka-band approx. 200km -300km). In order to be able to use the same frequency bands in neighboring cells ("frequency re-use"), adjacent cells are only distinguished by the polarization of the signals.

When using the antenna on fast-moving carriers, in particular on aircraft, then typically very many and very fast cell changes take place and the antenna must be able to quickly switch the polarization of the receive or transmit signals.

On the other hand, if the antenna is used in satellite services where the polarization of the transmit signal is fixed and does not change temporally or geographically, it is advantageous if the first intra-modular microstrip line network and the associated inter-modular waveguide network point to the Receiving band of the Antennne, and the second intra-modular microstrip network and the associated inter-modular waveguide network are designed for the transmission band of the antenna system.

This embodiment has the advantage that the respective feed networks can be optimized for the respective usable frequency band, and thus a very low-loss antenna system of very high performance is created.

If the radiation elements of the antenna system are designed for two orthogonal linear polarizations, then according to an advantageous embodiment of the invention, the feed networks are equipped with so-called 90 ° hybrid couplers. 90 ° hybrid couplers are four-ports which convert two orthogonal linearly polarized signals into two orthogonal circularly polarized signals and vice versa. With such arrangements, it is then possible to send or receive also circularly polarized signals.

Alternatively, the antenna array for receiving and transmitting circularly polarized signals can also be equipped with a so-called polarizer. Typically, these are suitably structured metallic layers ("layers") which lie in a plane approximately perpendicular to the propagation direction of the electromagnetic wave. The metallic structure acts in such a way that it acts capacitively in one direction and inductively in the orthogonal direction. For two orthogonally polarized signals, this means that a phase difference is imposed on the two signals. If the phase difference is now set to be just 90 ° when passing through the polarizer, then two orthogonal linearly polarized signals are converted to two orthogonal circularly polarized signals, and vice versa.

In order to obtain large useful bandwidths, the polarizer advantageously consists of several layers, which are mounted at a certain distance (typically in the region of a quarter wavelength) from each other.

A particularly suitable embodiment of the polarizer is a multi-layer meander polarizer. In this case, metallic meander structures of suitable dimensions are patterned on a typically thin substrate using the usual structuring methods. The substrates structured in this way are then glued onto foam boards or laminated to form sandwiches. As foams are e.g. low-loss closed-cell foams such as Rohacell or XPS in question.

It is advantageous here to superimpose a sequence of foam sheets, adhesive sheets and structured substrates on one another and to press them with a press. In a relatively simple manner then creates a suitable polarizer light weight.

According to a further advantageous embodiment of the invention, very high useful bandwidths and high cross-polarization isolations are achieved when the polarizer is not mounted exactly perpendicular to the direction of propagation of the electromagnetic wave in front of the antenna field, but slightly tilted. In these arrangements, the typical distance of the polarizer to the aperture surface of the antenna array is in the range of a wavelength of the useful frequency and the tilt angle to the aperture plane in the range of 2 ° to 10 °.

Since the antenna pattern of the antenna system in the transmission band must be below a regulatory mask, and for small antennas can only be sent with high spectral power densities, if the diagram as close as possible to the mask It may be advantageous to provide the antenna system with an amplitude amplitude tapering. Parabole amplitude assignments of the aperture are particularly suitable in the case of flat aperture openings for this purpose. Parabole amplitude assignments are characterized in that the power contributions of the individual radiators from the edge of the antenna field towards the center increase and z. B. results in a parabolic-like course.

Such amplitude assignments of the antenna field lead to a suppression of the side lobes in the antenna pattern and thus to a higher regulatory allowable spectral power density.

Since the side lobes in applications in geostationary satellite services only need to be suppressed along a tangent to the geostationary orbit at the location of the target satellite, the amplitude occupancy of the antenna field system is preferably designed to be at least along the direction through the antenna system in which the radiating elements are dense. acts. In this case, the beam elements are dense in the direction in which the distance of the phase centers of the individual radiators is less than or equal to the wavelength of the highest transmission frequency at which no significant parasitic side lobes (grating lobes) may occur.

In addition, other advantages and features of the present invention will be apparent from the description of preferred embodiments. The features described therein may be implemented alone or in combination with one or more of the features mentioned above. The following description of the preferred embodiments will be made with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE FIGURES

• Fig. 1a-b schematically show an inventive antenna module, which consists of a field of 8 x 8 individual radiators;
• Fig. 2a-b show exemplary microstrip feed networks for an 8x8 antenna module;
• Fig. 3a-d schematically illustrate the exemplary structure of an antenna according to the invention of antenna modules and the networking of the modules by waveguide networks;
• Fig. 4a-d show the detailed structure of a single quadruple-toothed horn radiator;
• Fig. 5 schematically illustrates the detailed structure of a 2 x 2 antenna module of four-toothed ("quad-ridged") horn radiator;
• Fig. 6a-b show an exemplary 8 x 8 antenna module, which consists of dielectrically filled horns;
• Fig. 7a-d illustrate the exemplary detailed construction of a single dielectrically filled horn radiator;
• Fig. 8 schematically shows the detailed structure of a 2 x 2 module of dielectrically filled horns;
• Fig. 9 shows a module according to the invention, which is provided for improving the impedance matching with a dielectric grid;
• Fig. 10a-b show a module according to the invention in layering technique;
• Fig. 11a-d show the detailed structure of a module according to the invention in layering technology;
• Fig. 12 schematically shows the vacuum model of a module according to the invention;
• Fig. 13 shows the exemplary construction of a waveguide power divider composed of double-ridged waveguides;
• Fig. 14 schematically shows a position of a polarizer;
• Fig. 15a-b show, by way of example, a schematic amplitude assignment of an antenna system according to the invention and the resulting maximum regulatory-compliant spectral EIRP density;
• Fig. 16 shows a possible construction of an antenna system according to the invention with fixed polarization of the transmitted and the received signal in the form of a block diagram;
• Fig. 17 shows a possible construction of a variable polarization antenna system according to the invention of the transmitted and received signals using 90 ° hybrid couplers in the form of a block diagram;
• Fig. 18 schematically shows the structure of an antenna system according to the invention with variable polarization of the transmitting and Receive signal using a polarizer in the form of a block diagram.

The exemplary embodiments of the antenna and its components shown in the drawings are explained in more detail below.

Fig. 1 represents an exemplary embodiment of an antenna module of an antenna according to the invention. The individual emitters 1 are designed here as rectangular horns, which can support two orthogonal polarizations.

The intra-modular microstrip line networks 2, 3 for the two orthogonal polarizations are located between different layers.

The antenna module consists of a total of 64 primary individual emitters 1 which are arranged in an 8 x 8 antenna field ( N _i = 64). The dimensions of the individual radiators and the size of their aperture surfaces are chosen so that the distance of the phase centers of the individual beam elements along both major axes is smaller than λ _min , where λ _min denotes the wavelength of the highest useful frequency. This distance ensures that parasitic sidelobes, so-called "grating lobes", can not occur in any direction in the antenna diagram up to the highest usable frequency (reference frequency).

In the exemplary case of in Fig. 1 As shown in the antenna module shown, both microstrip line networks provide a 64: 1 power splitter as they combine the signals from 64 individual emitters. An exemplary internal organization of the two microstrip transmission networks is in Fig. 2 shown.

However, embodiments are also conceivable for which the modules comprise a smaller or larger number of horns. For K / Ka band antennas e.g. 4 x 4 modules are optimal. The microstrip line networks then provide a 16: 1 power splitter that merges the signals from 16 individual emitters. The microstrip lines in this case are relatively short and their noise contribution therefore remains small.

Depending on the application, an antenna with optimum performance parameters can be constructed by appropriate design of the module sizes. Advantageously, the modules are only made as large as necessary in order to feed them with waveguides can. The parasitic noise contribution of the microstrip lines is thereby minimized.

The two microstrip line networks 2, 3 couple the merged signals into polarized-to-waveguide couplings 4, 5, respectively, according to polarization, as shown in FIG Fig. 1b is shown. By means of these waveguide couplings 4, 5, an arbitrarily large number of modules can be coupled efficiently and with low attenuation to form an antenna system according to the invention with the aid of waveguide networks.

Fig. 2 shows two exemplary microstrip line networks 2, 3 for feeding the individual radiator 1 of the 8 x 8 antenna module of Fig. 1 , Both networks are designed as binary 64: 1 power dividers.

By the two mutually orthogonal microstrip-to-Hohleiterkopplungen 6, 7, the orthogonally polarized signals in the individual horns of the 8 x 8 module or coupled. The sum signal is input or output to the waveguide couplings 4a and 5a in waveguides. Since the two microstrip line networks 2, 3 are typically superimposed in two planes, waveguide feedthroughs 4b and 5b are also located on the corresponding board in order to create an opening and the connection to the waveguide couplings 4a and 5a, respectively.

The microstrip line networks 2, 3 can be made by any known method. Whereby low-loss substrates for antennas are particularly suitable.

Fig. 3 shows by way of example how different antenna modules 8 can be coupled to antenna systems according to the invention.

Einzelstrahlern ergibt.Antenna systems according to the invention consist of a number M of modules, where M must be at least two. In Fig. 3 are exemplary modules with N _i = 8 x 8 = 64 (i = 1, ..., 16) single radiators 1 shown. M is equal to 16 and the modules are arranged in an 8 x 2 field (cf. 3a ), which is a rectangular antenna with $\mathrm{N}={\displaystyle \underset{i}{\Sigma }{N}_i=64\times 16=1024}$

Single radiators results.

However, other arrangements of the modules and other module sizes are also conceivable. Thus, the modules may e.g. also be arranged in a circle. Also, not all modules must have the same size (number of individual emitters).

The modules 8 are now connected to each other by means of the waveguide networks 9, 10. For this purpose, the corresponding waveguide coupling points 11, 12 of the waveguide networks 9, 10 with the corresponding waveguide couplings 4, 5 (see. Fig. 1b ) of the individual modules 8 connected.

The waveguide networks 9, 10 themselves each represent an M: 1 power divider, so that the two orthogonally polarized signals can be fed into the antenna system via the sum ports 13, 14 or be coupled out of the antenna system.

Depending on the application and the required frequency bandwidth, waveguides 9, 10 can be provided with a wide variety of waveguides, such as, for example, waveguide networks. Conventional rectangular or round waveguides or broad-banded ridged waveguides are used. Dielectric filled waveguide are conceivable.

So it can be e.g. be advantageous to fill the part of the waveguide network, which connects directly to the waveguide coupling 4, 5, with a dielectric. The dimensions of the dielectrically filled waveguides are then significantly reduced, so that their space requirement is minimal.

In the Fig. 3 illustrated antenna is thus constructed according to claim 1:
The antenna consists of an antenna array of N individual radiators 1, wherein each individual radiator 1 can support two independent orthogonal polarizations and N denotes the total number of individual radiators 1 of the antenna array.

gilt.In addition, the antenna array is constructed of modules 8, each module contains N _i single radiator and ${\displaystyle \underset{i}{\Sigma }{N}_i=N}$

applies.

gilt.In the embodiment of Fig. 3 In addition, it holds that each module contains N _i = n ₁ × n _k single radiators, N _i , n , i , l , k are integers and ${\displaystyle \underset{i}{\Sigma }{N}_i=N}$

applies.

The individual radiators 1 are dimensioned (s. Fig. 1 ) that for at least one direction through the antenna array, the distance of the phase centers of the horns is less than or equal to the wavelength of the highest transmission frequency at which no grating lobes may occur.

The individual radiators 1 are supplied separately by microstrip lines for each of the two orthogonal polarizations (see FIG. Fig. 2 Microstrip-to-waveguide couplings 6, 7).

The microstrip lines of one orthogonal polarization are connected to the first intra-modular microstrip line network 2 and the microstrip lines of the other orthogonal polarization are connected to the second intra-modular microstrip network 3.

The first micro-strip intra-modular network 2 is coupled to the first inter-modular waveguide network 9 and the second micro-strip intra-modular network 3 is coupled to the second inter-modular waveguide network 10 such that the first inter-modular waveguide network 9 receives all of the one orthogonal signals Polarization at the first sum port 13 merges and the second inter-modular waveguide network 10 all signals of the other orthogonal polarization at the second summing port 14 merges.

In addition, the microstrip line networks 2, 3 and the waveguide networks 9, 10 are constructed here as complete and fully symmetrical binary trees, so that all individual radiators 1 are fed in parallel.

The Figures 3c and 3d show a physical realization of a corresponding antenna system. The modules 8 consist of individual radiators 1 and have two different sizes, ie the number of individual radiators 1 per module 8 is not the same for all modules 8. The middle four modules 8 each have 8 individual emitters 1 more than the other four modules 8. As a result, the height of the antenna system at the left and right edges is less than in the central area. Such embodiments are particularly advantageous when the antenna system must be optimally adapted to an aerodynamic radome.

The modules 8 are fed separately with two waveguide networks 9 and 10 for each polarization. The waveguide networks 9, 10 are located in two separate layers behind the modules and the modules are connected to the waveguide networks 9, 10 through the coupling points 11, 12, which are coupled to the waveguide couplings of the modules 4, 5 ,. Both waveguide networks 9, 10 are realized here as cutouts.

If the transmission and reception bands of the antenna system are far apart in terms of frequency, then the case may arise that the dimensions of the individual radiators 1 of the field must become so small that the lower of the two frequency bands comes close to the cutoff frequency of the individual radiators 1, or even lower. Conventional horns, for example, can no longer support this frequency band or their efficiency drops sharply.

For example, in a K / Ka band operation, the receive frequency band is about 19GHz - 20GHz and the Transmitting frequency band at approx. 29GHz - 30GHz. In order to fulfill the condition that the antenna pattern in the transmission band is free of parasitic sidelobes ("grating lobes"), the aperture of the individual radiators 1 must not be more than 1 cm x 1 cm in size λ _min is 1 cm).

Conventional dual polarized horns, for example, with an aperture of only 1cm x 1cm, however, hardly work at 19GHz - 20GHz (λ _max = 1.58cm), because an acceptable impedance match to the free space is no longer possible. In addition, the horn would have to be operated very close to the lower cutoff frequency ("cutoff" frequency), which would lead to very high dissipative losses and to a very low antenna efficiency.

The primary individual radiators 1 are designed as ridged horns. Such horns have a much wider than conventional horns frequency bandwidth.

The impedance matching of such toothed horns to the free space then takes place according to the method of antenna physics. The serrated horns are designed to support two orthogonal polarizations. This is e.g. achieved in that the horns are serrated quadruple symmetrical ("quad-ridged"). The signals of the orthogonal polarizations are supplied and removed by separate microstrip line networks 2, 3.

Fig. 4a schematically shows the detailed structure of a equipped with symmetrical geometrical constrictions horn with the example of a four-tooth horn horn 1. The horn 1 consists of three segments (layers), which are located between the segments, the two microstrip lines networks 2.3.

The horns 1 are provided with symmetrical geometric constrictions 15, 16 corresponding to the orthogonal directions of polarization which extend along the propagation direction of the electromagnetic wave.

Such horns are referred to as "toothed" horns. Is shown in Fig. 4a an exemplary quadruple toothed single horn that can support broadband two orthogonal polarizations.

As in the cuts in Fig. 4b and 4c are shown, the geometric constrictions are executed stepped and the distance of the constrictions 15, 16 to each other decreases in the direction of the coupling and decoupling. As a result, a very large frequency bandwidth can be achieved. Especially horn horns 1 can be realized, which can also support frequency far distant transmitting and receiving tapes without significant losses in efficiency. An example of this are K / Ka band satellite antennas. Here the reception band lies at 18 GHz - 21 GHz and the transmission band at 28 GHz - 31 GHz.

The depth, width and length of the steps depend on the desired frequency bands and can be determined with numerical simulation methods.

The coupling or decoupling of the signals onto the microstrip network 2, 3 typically takes place at the narrowest point of the constrictions 15, 16 for the respective polarization direction, which allows a very broadband impedance matching.

Fig. 4d schematically shows a part of the longitudinal section through a toothed horn at the location of two opposing constrictions 16. The constrictions 16 are executed stepped and the distance d _{i of} opposite stages decreases from the aperture of the horn (above) to the horn end (down) down.

In addition, the horn itself is stepped (cf. Fig. 4a-c ), so that at each stage, the edge length a _{i of} the horn opening in the corresponding cross section from the aperture of the horn to the horn end also decreases.

The distances d _i and the associated edge lengths a _i , or at least a part thereof, are now designed so that the associated lower limit frequency of the respective toothed waveguide section is below the lowest useful frequency of the horn. Only when this condition is met can the electromagnetic wave of the appropriate wavelength penetrate into the horn to waveguide-to-microstrip line coupling, where it can be coupled in and out.

Since the dissipative damping greatly increases when approaching the lower limit frequency, the distances d _i and the associated edge lengths a _{i are} advantageously chosen so that a sufficient distance to the cutoff frequency remains and the attenuation does not become too high.

In addition, it must be taken into account that in antenna systems which consist of several horn radiators, mutual couplings of the radiators are effective.

As has been shown, a favorable embodiment can nevertheless be described by an analytical condition which includes the edge length a _{i of} the aperture in the corresponding cross section through the horn and the distance d _i .

In Fig. 5 3 schematically shows the structure according to the invention of a 2 × 2 antenna module consisting of four quadruple toothed horns 1, four outcouplings 17 on the microstrip line networks 2, 3, two microstrip line networks 2, 3 separated for each of the two orthogonal polarizations, and outcouplings of the microstrip line networks 2, 3 on the waveguide coupling 4, 5 has. The constrictions as symmetrical teeth 15, 16 of the horns 1 are also shown.

The two orthogonally polarized signals pol 1 and pol 2 whose reception or radiation is supported by the horns 1 are fed through the extraction or injection points 17 in the corresponding microstrip line network 2, 3 and extracted from this.

The microstrip line networks 2, 3 in turn are designed as binary 4: 1 power dividers and couple the sum signals into the waveguides 4, 5.

The distance of the phase centers of two adjacent horns 1 in the vertical direction is smaller than λ _min , so that at least in this direction in the antenna diagram no unwanted parasitic side lobes ("grating lobes") can occur and the horns are dense in this direction.

The phase centers of the horns 1 fall in the in Fig. 5 illustrated example with the beam centers of the horns 1 together. In general, however, this is not necessarily the case. However, the position of the phase center of a horn 1 of any geometry can be determined by numerical simulation methods.

For the coupling and decoupling of the signals supported by the toothed horns 1 microstrip lines are due to their known broadband in a special way. In addition, microstrip lines require very little space, so that highly efficient, broadband horn antenna systems whose antenna diagrams have no parasitic side lobes ("grating lobes"), even for very high frequencies (eg 30 GHz - 40 GHz) can be realized.

In Fig. 6 a further embodiment of the invention is shown. Here, the antenna modules of dielectric filled horns 18 are constructed. The horns 18 filled with a dielectric 19 are arranged here by way of example in an 8 × 8 antenna field and are coupled to one another via the microstrip line networks 2 and 3.

The microstrip line networks 2, 3 couple the sum signals into the waveguide couplings 4, 5.

In the Fig. 7a-c the internal structure of a completely filled with a dielectric individual horn radiator 18 is shown. Like the horn 18 itself, the dielectric packing (dielectric) 19 also consists of three segments, each defined by the microstrip line networks 2, 3.

So that the individual radiators 1 can support two widely spaced frequency bands, they are executed stepped in their interior, as in the sections Fig. 7b-c is shown by way of example. The extraction or coupling of the highest frequency band is typically at the narrowest or lowest point by the microstrip network 3, which is farthest from the aperture of the single radiator 1. The lower frequency band is switched on or coupled in at a further point to the aperture opening, by a microstrip line network 2.

The depth, width and length of the steps depend on the desired frequency bands and can also be determined with numerical simulation methods.

If the two input or output coupling points of the microstrip line networks 2, 3 spatially sufficiently close to each other, then the horn 1 can also be designed so that both inputs and outputs can support both the transmit and the receive frequency band.

The dielectric filling body 19 is also designed to match exactly stepped. The shape of the filling body 19 on the aperture surface depends on the electromagnetic requirements of the antenna pattern of the single radiator 1. The filler 19 can be performed flat as shown at the aperture opening. However, there are also other, for example, inward or outward curved versions possible.

As dielectrics come a variety of known materials such as Teflon, polypropylene, polyethylene, polycarbonate, or polymethylpentene in question. For example, for simultaneous coverage of the K and Ka tape, a dielectric having a dielectric constant of about 2 is sufficient (e.g., Teflon, polymethylpentene).

In the in Fig. 7 illustrated exemplary embodiment, the horn antenna 18 is completely filled with a dielectric 19. However, embodiments with only partial filling are also possible.

The advantage of using dielectrically filled horns is that the horns themselves have a much less complex internal structure than in the case of toothed horns.

However, in order to display high-efficiency antennas even at very high GHz frequencies, it is also conceivable, e.g. fourfold toothed horn with a dielectric to fill. Other horn geometries with dielectric filling or partial filling are also possible.

In Fig. 7d is an advantageous embodiment of a stepped executed dielectrically filled horn radiator, which has a rectangular aperture has shown schematically.

Fig. 7d shows the view of the horn from above (top view) with the aperture edges k ₁ and k ₂ , and the longitudinal sections through the horn along the lines AA 'and B-B'.

The horn is now designed so that there is a first rectangular cross-section through the horn, the opening of which has a long edge k _E , and a second cross-section exists through the horn, the opening of which has a long edge k _s .

If now the reception band of the antenna system is lower in frequency than the transmission band and the edge k _{E is} now chosen so that the associated lower limit frequency of a dielectric filled waveguide with a long edge k _{E is} below the lowest useful frequency of the receiving band of the antenna system, then the horn support the reception band.

In addition, if the edge k _s chosen so that the associated lower limit frequency of a dielectric filled waveguide with a long edge k _{s is} below the lowest usable frequency of the transmission band of the antenna system, then the horn can also support the transmission band, and this is true even if the reception band and transmission band are far apart.

Because in Fig. 7d the edge k _{s is} orthogonal to the edge k _E , two orthogonal linear polarizations are simultaneously supported by such a horn, since the respective waveguide modes are linearly polarized and orthogonal to one another.

Such stepped horn radiators can also be operated without or only with partial dielectric filling and that the in Fig. 7d illustrated embodiment can be extended to any number of rectangular horn sections and thus to any number of Nutzbändern.

If the horns of the antenna system are now close, ie should no parasitic side lobes (grating lobes) occur in the antenna pattern of the antenna system, then in a further advantageous embodiment, the edge lengths k ₁ and k _{2 of} the rectangular aperture of the horn are chosen so that both k ₁ as well as k _{2 are} smaller or highest equal to the wavelength of the reference frequency, which is in the transmission band of the antenna.

In this case, the available space is then optimally utilized and a maximum antenna gain is achieved.

Fig. 8 shows an exemplary 2 x 2 antenna module, which consists of four dielectrically filled horns 18. As in Figure 7b-c shown here are the inputs or outputs in the microstrip network 2, 3 completely embedded in the dielectric 19. Otherwise, the module does not differ from the corresponding module of toothed horns, as in Fig. 5 is shown, the microstrip line networks 2, 3 are connected to the waveguide couplings 4, 5 respectively.

In Fig. 9 is shown a further advantageous embodiment. Here, the module is equipped with a dielectric grid 20 extending over the entire aperture opening. Such dielectric gratings 20 can greatly improve the impedance matching, particularly at the lower frequency band of the single radiators 1, by reducing the effective wavelength in the vicinity of the aperture openings of the single radiators 1.

Im in Fig. 9 This is achieved by the fact that dielectric crosses are located above the centers of the aperture openings of the individual radiators. It is, however Other embodiments such as cylinders, spherical body, cuboid etc. possible. Also, the dielectric grid 20 need not be regular or periodic. For example, it is conceivable that the grating for the horns 1 at the edge of the antenna has a different geometry than for the horns 1 in the center. Thus, for example, edge effects could be modeled.

Fig. 10a-b represents an exemplary module, which is built in layer technology. By this technique, modules according to the invention can be produced particularly cost-effectively. In addition, even at very high frequencies (high tolerance requirements), the reproducibility of the modules is guaranteed.

The first layer consists of an optional polarizer 21, which is used in circularly polarized signals. The polarizer 21 converts linearly polarized signals into circularly polarized and vice versa, depending on the polarization of the incident signal. Thus, incident on the antenna system circularly polarized signals are converted into linearly polarized signals, so that they can be received lossless from the horns of the module. On the other hand, the linearly polarized signals radiated from the horns are converted into circularly polarized signals and then radiated into the clearance.

The next two layers form the front part of the horn radiation field, which comprises the primary horn structures 22 without coupling-in or coupling-out unit.

The following layers 23a, 2 and 23b form the coupling in and out of the first linear polarization from the horns of the field. The microstrip line network 2 of the first polarization and its substrate are embedded in metallic carriers (layers) 23a, 23b. The carriers 23a, 23b have recesses (notches) at the locations where a microstrip line runs (cf. Fig. 11d , Reference numeral 25).

Likewise, the microstrip line network 3 of the second, orthogonal polarization with its substrate is embedded in the carriers 23b, 23c.

In the last position, the waveguide terminations 24 of the horns and the waveguide outcouplings 4 and 5.

The primary horn structures 22, the carriers 23a-c and waveguide terminations 24 are electrically conductive and can be inexpensively produced using the known methods of metal working, for example, made of aluminum (eg milling, laser cutting, water jet cutting, electroeroding).

However, it is also conceivable to produce the layers from plastic materials, which are subsequently completely or partially coated with an electrically conductive layer (for example galvanically or chemically). For the production of the plastic layers, e.g. also the known injection molding process can be used. Such embodiments have the advantage over layers of aluminum or other metals that a significant weight reduction can result, which is particularly advantageous in applications of the antenna system on aircraft.

With this layer technology, a highly efficient and cost-effective antenna module is made available even at very high GHz frequencies.

The layer technique described can be used equally well for antenna modules made of toothed horns as well as modules made of dielectrically filled horns.

Fig. 11a-d 10 show the detail structure of the microstrip line networks 2, 3 embedded in the metallic carriers. The recesses (notches) 25 are designed so that the microstrip lines 26 of the microstrip line networks 2, 3 run in closed metallic cavities. The microwave losses are thereby minimized.

In addition, since with the finite thickness of the substrates (circuit board) of the microstrip lines 26 a gap remains between the metallic layers, could escape through the microwave power, it is provided to the substrates with metallic vias 27 at the edges of the notches, so that the metallic supports are galvanically connected, and so the cavities are completely electrically closed. If the plated-through holes 27 are sufficiently dense along the microwave lines 26, then microwave power can no longer escape.

Preferably, the plated-through holes 27 are flush with the metallic walls of the cavity 25. In addition, if a thin, low-loss substrate (board material) is used, the electromagnetic properties of such a structure are similar to those of an air-filled coaxial line. In particular, a very broadband microwave line is possible and parasitic higher modes are not capable of propagation. In addition, even at very high GHz frequencies, the tolerance requirements are low.

In the case of very thin substrates (eg <20 μm) and correspondingly low useful frequencies, the plated-through holes may also be dispensed with, as well as without them Vias then virtually no microwave power can escape through the then very narrow slots.

The Hornstrahlereinkopplungen or -auskopplungen 6, 7 are integrated directly into the metallic carrier.

Fig. 12 shows the vacuum model of an exemplary 8 x 8 antenna module. The horns 1 are densely packed and still leaves more than enough space for the microstrip line networks 2, 3, and for the waveguide terminations 28 of the individual radiator 1 and the waveguide couplings 4, 5. A dielectric grating 20 is mounted in front of the aperture plane.

In a further advantageous embodiment, the waveguide networks, which couple the modules together from toothed waveguides. This has the advantage that toothed waveguides can have a much greater frequency bandwidth than conventional waveguides or can be designed specifically for different useful bands.

An exemplary network of dual-toothed waveguides is shown in FIG Fig. 13 shown schematically. The rectangular waveguides are provided with symmetrical geometrical constrictions 29, which are supplemented by vertical constrictions 30 at the location of the power dividers.

The design of the toothed waveguides and corresponding power dividers can be done by the methods of numerical simulation of such components, depending on the requirements of the network.

It does not necessarily have to use double-toothed waveguides. Also, e.g. simply toothed or fourfold toothed waveguides are conceivable.

In one embodiment, not shown, the waveguides of the inter-modular waveguide networks are completely or partially filled with a dielectric. Such fillings can significantly reduce the space requirement compared to unfilled waveguides with the same useful frequency. This results in very compact, space-optimized antennas, which are particularly suitable for applications on aircraft. Both standard waveguides and waveguides with geometric constrictions can be filled with a dielectric.

In a further advantageous embodiment, the antenna is equipped with a multilayer meander polarizer.

Fig. 14 shows an example of a position of such a polarizer.

In order to achieve axis ratios of the circularly polarized signals in the vicinity of 1 (0 dB), multilayer meander polarizers are used.

In one embodiment, not shown, several of the in Fig. 14 layers shown in parallel planes arranged one above the other. Between the layers there is a low-loss layer of foam material (eg Rohacell, XPS) with a thickness in the region of a quarter of a wavelength. With lower axle ratio requirements, however, fewer layers may be used. Likewise, more layers can be used if the axis ratio requirements are high.

An advantageous arrangement is a 4-layer meander polarizer with the axial ratios of less than 1 dB can be achieved, which is usually sufficient in practice.

The design of the meander polarizers depends on the useful frequency bands of the antenna system and can be done with methods of numerical simulation of such structures.

The meandering lines 31 are in the embodiment of Fig. 14 at an angle of about 45 ° to the main axes of the antenna. This results in incident, linearly polarized along a major axis signals are converted into circularly polarized signals. Depending on which main axis the signals are linearly polarized, a left-circularly polarized or a right-circularly polarized signal is produced.

Since the meander polarizer is a linear device, the process is reciprocal, i. In the same way, left- and right-circularly polarized signals are converted into linearly polarized signals.

It is also conceivable to use other geometric structures as meanders for the polarizers. There are a variety of passive geometric conductor structures known, with which linear polarized can be converted into circularly polarized signals. It depends on the application, which structures are most suitable for the antenna.

As in Fig. 10 As shown, the polarizer 21 may be mounted in front of the aperture opening. This makes it possible in a relatively simple manner to use the antenna for both linearly polarized signals and for circularly polarized signals, without the need to change the internal structure for it.

In a further advantageous embodiment, the antenna is equipped with a parabolic amplitude assignment, which is realized by a corresponding design of the power divider of the feed networks. Since the antenna pattern must be below a mask prescribed by regulations, such amplitude assignments can achieve much higher maximum permitted spectral EIRP densities in the transmit mode than without such assignments. This is of great advantage, in particular for antennas with a small aperture area, since the maximum regulatory-compliant spectral EIRP density is directly proportional to the achievable data rate and thus to the cost of a corresponding service.

In Fig. 15a such an amplitude assignment is shown schematically. The power contributions of the individual horns fall from the center of the aperture to the edge. In Fig. 15a this is exemplified by different degrees of blackening (dark: high performance contribution, bright: low contribution to performance). Here, the contributions to performance fall in both main axis directions (azimuth and elevation). This results in an approximately optimally matched to the regulatory mask antenna pattern for all angles of rotation ("skew").

Depending on the requirements of the antenna diagram, however, it may also be sufficient to occupy the aperture in one direction only.

It is also conceivable that the amplitude occupancy only runs parabolically in the area around the antenna center, but increases again when approaching the edge, so that there is a closed curve around the antenna center and the power contributions of the individual radiators from the center of the antenna to each point of this curve fall off. Such amplitude assignments may be of particular advantage for non-rectangular antennas.

In Fig. 15b By way of example, the maximum regulatory compliant spectral EIRP density (EIRP SD), as a function of the angle of rotation about the main beam axis ("skew"), is shown by a parabolic amplitude distribution of a rectangular 64 × 20 Ka band antenna in both main axis directions. Without parabolic coverage, the EIRP SD would be about 8 dB lower in the range of 0 ° skew to about 55 ° skew, and about 4 dB lower in the range of about 55 ° skew to about 90 ° skew.

The Fig. 16-18 show the basic structure of a number of antenna systems according to the invention with different functional scope in the form of block diagrams.

The antenna system, whose basic structure in Fig. 16 is particularly suitable for applications in the K / Ka band (reception band approx. 19.2GHz -20.2GHz, transmission band approx. 29GHz -30GHz) where the polarizations of the transmission and reception signals are fixed and orthogonal to each other (ie Polarization direction of the signals does not change).

Since the K / Ka band typically uses circularly polarized signals, initially a polarizer 21 is provided. This is followed by an antenna field 32, which is constructed either of four-toothed ("quad-ridged") horn radiators or of dielectrically filled horn radiators. The aperture openings of the individual horns typically have dimensions smaller than 1cm x 1cm in this frequency range.

The antenna array 32 according to the invention is organized in modules, each individual radiator having two microstrip line couplings or outcouplings 33 separated by polarization, which in turn are connected to two microstrip line networks 36 separated by polarization.

Since the polarization of the transmit and receive signals is fixed and typically orthogonal to one another, it is envisaged here that the microstrip line network 36 of one polarization be placed on the transmit band and the microstrip line network 36 of the other polarization on the receive band.

This has the advantage that the microstrip network 36 of the receiving band can be designed for minimum losses, and thus the G / T of the antenna becomes optimal.

In the exemplary construction of Fig. 16 the polarizer 21 is oriented such that the signals in the transmission band 34 are right-handed circular and the signals in the reception band 35 are circularly polarized left-handed.

The signals separated by polarization and frequency band of the two microstrip line networks 36 of the individual modules are now coupled with microstrip line-to-waveguide couplings 37 in two waveguide networks 38.

Again, it is envisaged that the two waveguide networks 38 will be optimized for the corresponding band they are to support.

For example, different waveguide cross sections can be used for the receive band waveguide network and the transmit band waveguide network. In particular, you can enlarged waveguide cross-sections can be used, which greatly reduce the dissipative losses in the waveguide networks and thus can significantly increase the efficiency of the antennas.

Furthermore, a receive band frequency filter 39 is provided to protect the low noise receive amplifier, which is typically mounted directly on the receive band output of the antenna, from being overdriven by the strong transmit signals.

In order to achieve the regulatory sideband suppression in the transmission band, an optional transmission band filter 40 is also provided. This is e.g. required when a transmit band power amplifier (HPA), not shown, does not have a sufficient filter at its output.

The in Fig. 16 shown construction of an antenna system according to the invention has another, especially for satellite antennas, very important advantage. Since the transmit band feed network and the receive band feed network are completely separated from each other both at the microstrip line level and at the waveguide level, it becomes possible to use different amplitude assignments for the two networks.

Thus, e.g. the receive band feed network is homozygously occupied, i. the power contributions of all of the antenna's horns are the same in the receive band and all power dividers at both the receive band microstrip line level and the receive band waveguide network level are balanced 3dB power dividers when the feed network is constructed as a complete and fully symmetric binary tree.

Since homogeneous amplitude assignments lead to the maximum possible antenna gain, this ensures that the antenna is maximally efficient in the receiving band and the ratio of antenna gain and inherent noise G / T of the antenna becomes maximum.

On the other hand, the transmit band feed network can be provided with a parabolic amplitude assignment independently of the receive band feed network in such a way that the regulatory compliant spectral EIRP density becomes maximum.

Although reduced by such parabolic amplitude assignments of the antenna gain, but this is not critical, because this is due to the design limited only to the transmission band and does not affect the receiving band.

The essential features of satellite antennas, especially satellite antennas of small size, are the G / T and the maximum regulatory compliant spectral EIRP density.

The G / T is directly proportional to the data rate that can be received via the antenna. The maximum regulatory EIRP spectral density is directly proportional to the data rate that can be transmitted with the antenna.

With antenna systems, which accordingly Fig. 16 are constructed, both features can be optimized independently.

For very small satellite antennas, this results in another advantage. Namely, there is a problem that the width of the main beam in the reception band can become so large that not only signals of the target satellite but also signals of adjacent satellites are received. The signals from adjacent satellites then effectively act as an additional noise contribution, which can lead to a significant degradation of the effective G / T.

In antenna systems according to the invention, which accordingly Fig. 16 are constructed, this problem can be solved at least partially. If, for example, the reception band feed network is not homogeneously occupied by amplitude, but is occupied by hyperbol amplitude, then the width of the main beam of the antenna decreases. Hyperbolic amplitude assignments are characterized by the fact that the power contributions of the individual radiators of the antenna field increase from the middle to the edge.

By means of a hyperbolic amplitude occupancy, at least in a partial area of the antenna system, it is possible to achieve that the intensity of the interference signals received by neighboring satellites by the antenna decreases and the effective G / T increases in such an interference scenario.

In Fig. 17 the structure of an antenna system according to the invention is shown in the form of a block diagram, which allows simultaneous operation with all four possible polarization combinations of the signals.

The antenna system initially consists of an antenna array 41 of broadband, dual polarized horns, e.g. fourfold toothed horns, which are organized according to the invention in modules.

In contrast to the embodiment, which in Fig. 16 is shown, but here no polarizer is used, but Each horn radiator receives two orthogonal linear polarized signals which, however, also contain the full information when operating with circularly polarized signals.

The main difference to the embodiment in Fig. 16 consists in the fact that is not separated at the level of the feed networks in a receive band and a transmit band feed network, but the signals are separated only according to their different polarization.

All signals 42 of the same polarization are combined after the extraction 33 from the antenna field in the first microstrip network, all signals of the orthogonal polarization 43 in the second microstrip network.

The two microstrip line networks 36 are designed such that they support both the transmission band and the receiving band. An optimization of the feed networks on one of the tapes is possible here only to a limited extent. However, all four polarization combinations are simultaneously available for this.

While microstrip networks 36 of the present invention are typically already broadband by design (coaxial line like construction) to simultaneously support the receive and transmit bands, after transition 37 microstrip to waveguides, waveguide networks 44 must be used if very large bandwidths are required specially designed. This can be done by the in Fig. 13 described toothed waveguide done. However, it is also possible to use, for example, dielectrically filled waveguides.

For the separation of receive band and transmit band signals, two frequency diplexers 45, 46 are provided, one for each polarization. The frequency diplexers 45, 46 are e.g. low attenuation waveguide diplexer.

When operating with linearly polarized signals, all linear polarization combinations are then simultaneously available at the output of the two diplexers: two orthogonally polarized linear signals both in the reception band 49 and in the transmission band 50.

When operating with circularly polarized signals, two 90 ° hybrid couplers 47, 48, one for the receive 49 and one for the transmit band 50, are additionally provided, with the aid of which at the output of the frequency diplexers 45, 46 present linear polarized signals, circular polarized signals can be combined. The 90 ° hybrid couplers 47, 48 are, for example, low-attenuation waveguide couplers.

At the output of the two 90 ° hybrid couplers 47, 48 are then all four possible circularly polarized signals (in the receiving 49 and transmit band 50 each right-handed and left-handed circular) before simultaneously.

If appropriate RF switches and / or RF couplers are installed between diplexer 45, 46 and 90 ° hybrid couplers 47, 48 and thus the linearly polarized signals are coupled out, then the antenna system can also be used for simultaneous operation with four different linearly and four different circularly polarized signals be used. Many other combinations and the corresponding antenna configurations are possible.

In Fig. 18 the structure of an antenna system according to the invention in the form of a block diagram is shown, which has the same scope of functions as in Fig. 16 has shown antenna, but is organized differently.

Under construction Fig. 18 For operation with circularly polarized signals, a polarizer 21 is used instead of the 90 ° hybrid couplers 47, 48 of the design Fig. 17 ,

The feed networks 36, 44 process again two orthogonal polarizations separated from each other (in this case left-circular and rechtszikular) and are each designed correspondingly broadband for the receiving band and the transmission band.

At the output of the frequency diplexers 45, 46 are then directly the four polarization combinations of circularly polarized signals simultaneously. At the frequency-diplexer 45 for the first circular polarization, the signal in the receive and transmit band, at the frequency diplexer 46 for the second (to the first orthogonal) circular polarization, the signal in the receive and transmit band.

Through the use of two 90 ° hybrid couplers (not shown), which are similar in construction to the Fig. 17 can be connected to the diplexers 45, 46, also the structure after Fig. 18 be designed for the operation of linearly polarized signals, or it is possible with the corresponding circuit matrix, a simultaneous operation with circular and linearly polarized signals.

The advantage of the construction after Fig. 18 This is because when operating with circularly polarized signals no 90 ° hybrid couplers are needed. This may vary depending on the application eg space or weight saving. Also, there may be cost benefits.

The advantage of the construction after Fig. 17 On the other hand, in operation with circularly polarized signals, the axis ratio of the circularly polarized signals via the respective power contributions at the input of the 90 ° hybrid coupler 47, 48 is in principle freely adjustable.

This can e.g. then be advantageous if the antenna is operated under a radome. It is known that, in particular for high GHz frequencies, radomes by the radome material and the radome curvature may have polarization anisotropies which cause the axis ratio of circularly polarized signals to be greatly altered as it passes through the radome.

This effect has the consequence that the cross-polarization isolation can fall sharply, which can severely degrade the achievable channel separation and ultimately leads to a degradation of the achievable data rate.

A structure of the antenna after Fig. 17 now allows the axis ratio of the circularly polarized signals, for example, in the transmit mode to be adjusted so that a subsequent, caused by the Radomdurchgang polarization distortion is compensated. A degradation of the cross-polarization isolation thus does not take place effectively.

Claims (16)

1. Antenna system,
   having at least four single radiating elements (1), wherein the single radiating elements are in the form of horn antennas, and the horn antennas (1) support two mutually orthogonal linear polarizations and are equipped with constrictions (15, 16) in both polarization planes, characterized in that the walls of the horn antennas (1) and the geometric constrictions (15, 16) are at least to some extent of stepped design and the aperture of the horn antennas (1) is at any rate approximately rectangular, the interval between two opposite constrictions (15, 16) and the cross section of the horn antenna opening decreases from the aperture to the horn end of the horn antenna (1) from step to step, at least for some of these steps the steps are designed such that, for the interval d_i between the integer i-th steps, wherein i ≥ 0 in two opposite constrictions (15, 16) and the associated edge length a_i of the opening of the horn antenna cross section of the integer i-th step, $${\mathrm{d}}_{\mathrm{l}}\le {\mathrm{p}}_1\frac{2\pi }{{\lambda }_F}{{\mathrm{a}}_{\mathrm{l}}}^2-{\mathrm{p}}_2{\mathrm{a}}_{\mathrm{l}},$$

   

   holds when λ_E denotes the free-space wavelength of the lowest useful frequency of the antenna system, p₁ is between 0.3 and 0.4 and p₂ is between 0.25 and 0.35, wherein preferably p₁ = 0.35 and p₂ = 0.29.
2. Antenna system according to Claim 1, characterized in that the constrictions (15, 16) are in symmetrical form with respect to the horn antenna central axis.
3. Antenna system according to either of the preceding claims, characterized in that the steps in the walls of the horn antennas (1) and the steps in the symmetrical geometric constrictions (15, 16) are provided with optimum impedance matching to useful frequencies of the antenna system with respect to one another.
4. Antenna system according to Claim 3, characterized in that the interval between two opposite constrictions (15, 16) decreases from the aperture to the horn end of the horn antenna (1) from step to step, and on each step the lower cut-off frequency, associated with the respective interval, of the horn section associated with the respective step is lower than the lowest useful frequency of the antenna system.
5. Antenna system according to one of the preceding claims, characterized in that the aperture of the horn antennas (1) is at any rate approximately square,
   and $${\lambda }_S\ge a_0\ge \frac{{\lambda }_S}{2}$$

   

   holds, wherein a₀ denotes the edge length of the aperture and •_S denotes the free-space wavelength of the highest useful frequency of the antenna system and p₁ = 0.35 and p₂ = 0.29.
6. Antenna system according to one of the preceding claims, characterized in that some of the horn antennas (1) or all the horn antennas (1) are equipped with a dielectric cross septum and/or a dielectric lens.
7. Antenna system according to one of the preceding claims, characterized in that the horn antennas (1) are to some extent or completely filled with a dielectric (19).
8. Antenna system according to one of the preceding claims, characterized in that at least some of the horn antennas (1) are dimensioned such that the interval between the phase centers of two directly adjacent horn antennas (1) is less than or at most equal to the wavelength of a reference frequency that lies in the transmission band of the antenna system.
9. Antenna system according to one of the preceding claims, characterized in that the horn antennas (1) are fed by a first microstrip line (2) for the first of the orthogonal linear polarizations and by a second microstrip line (3) for the second of the orthogonal linear polarizations, and the microstrip lines (2) for the first polarization and the microstrip lines (3) for the second polarization each form separate microstrip line networks (2, 3).
10. Antenna system according to one of the preceding claims, characterized in that microstrip lines of the microstrip line networks (2, 3) are situated on a thin substrate and are routed in cavities, the walls of which are at least to some extent electrically conductive.
11. Antenna system according to one of the preceding claims, characterized in that the antenna system is constructed from various layers (22, 23a, 23b, 23c, 24) and a microstrip line network (2) for one polarization and a microstrip line network (3) for the other polarization are in this case situated separately from one another between the layers.
12. Antenna system according to Claim 11, characterized in that the different layers (22, 23a, 23b, 23c, 24) form an antenna module (8) and are made from metal, and the microstrip lines (26) of the microstrip line networks (2, 3) are routed in cavities that are designed as notches (25) in the layers (23a, 23b, 23c), one notch (25) being situated above and one being situated below the microstrip line (26).
13. Antenna array having a plurality of antenna systems according to one of the preceding claims that comprises waveguide networks (9, 10) and are coupled to one another thereby.
14. Antenna array according to Claim 13, characterized in that a first waveguide network (9) brings together all the signals of the first polarization and a second waveguide network (10) brings together all the signals of the second polarization.
15. Antenna array according to either of claims 13 and 14, characterized in that at least some of the waveguide networks (9, 10) have at least one geometric constriction (15, 16) along the direction of propagation of the electromagnetic wave.
16. Antenna array according to Claim 15, characterized in that at least some of the waveguide networks (9, 10) are designed as single-ridged or double-ridged waveguides.

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