EP2425490B1 - Broadband antenna system for satellite communication - Google Patents

Description

The invention relates to a broadband antenna system for communication between mobile carriers and satellites, in particular for aeronautical applications.

The demand for wireless broadband data transmission channels with very high data rates, especially in the field of mobile satellite communications, is steadily increasing. However, especially in the aeronautical field, there is a lack of suitable antennas, which in particular can fulfill the conditions required for mobile use, such as small dimensions and low weight. For directional, wireless data communication with satellites (eg in the Ku or Ka band), there are also extreme requirements on the transmission characteristics of the antenna systems, since interference with neighboring satellites must be reliably excluded.

In aeronautical applications, the weight and size of the antenna system is very important because it reduces the payload of the aircraft and causes additional operating costs.

The problem therefore is to provide antenna systems that are as small and lightweight as possible, which nevertheless satisfy the regulatory requirements for transmitting and receiving operation when operating on mobile carriers.

The regulatory requirements for the transmission operation arise z. From standards CFR 25.209, CFR 25.222, ITU-R M. 1643 or ETSI EN 302 186. All these regulatory requirements are intended to ensure that interference with adjacent satellites can not occur in the directional transmission of a mobile satellite antenna. Typically, envelopes of maximum spectral power density are defined as a function of the distance to the target satellite. The values specified for a certain distance angle must not be exceeded in the transmission mode of the antenna system. This leads to stringent requirements for the angle-dependent antenna characteristic. As an example, in Fig. 5a the requirement from CFR 22.209 for the angle-dependent antenna gain in the Ku band in the direction of the azimuth (tangential to the Clarke orbit) is shown (bold curve). As the distance from the target satellite increases, the antenna gain must drop sharply. This can be achieved physically only by very homogeneous amplitude and phase assignments of the antenna. Typically, therefore, parabolic antennas are used which have these properties. For mobile use, especially on aircraft, such antennas are not suitable. Here, to reduce the air resistance, rectangular or rectangular antenna apertures are used which have an aspect ratio height to width of at most 1: 4. Since parabolic mirrors have only very low efficiencies in such aspect ratios, for applications such. As on airplanes or cars, preferably antenna fields in question.

In antenna fields, however, the known problem of the so-called "grating lobes" occurs. Grating lobes are significant parasitic side lobes, which arise from the fact that the beam centers of the antenna elements that make up the antenna field, due to the design have to have a certain distance from each other. This leads at certain beam angles to the positive interference of the antenna radiators and thus to the unwanted emission of electromagnetic power in unwanted solid angle ranges. From the theory two-dimensional Antenna fields (eg JD Kraus and RJ Marhefka, "Antennas: for all applications", 3rd ed., McGraw-Hill series in electrical engineering, 2002 ) shows that significant parasitic grating lobes do not occur only if the beam centers of the antenna array are less than a wavelength of the minimum useful wavelength of each other.

Since antenna fields have to have a feed network, there is the practical problem of finding network and antenna field topologies which, on the one hand, meet the above requirement for the maximum distance between the beam centers and, on the other hand, occupy as little space as possible. In addition, the feed networks must be minimally dissipative in order to realize high antenna efficiencies and thus minimum antenna sizes.

Directed satellite communication also typically uses two independent signal polarizations to increase the data rate. The antenna system must therefore be able to process two independent polarizations simultaneously. Both in the transmitting and in the receiving mode, a high polarization separation is required so that there is no mixing and thus a loss of efficiency. In the transmission mode, there are also strict regulatory requirements for the polarization separation so that it can not interfere with neighboring transponders with orthogonal polarization (see, for example, CFR 25.209 or 25.222). In the case of antenna fields, it must therefore be ensured, on the one hand, that the primary radiator elements have sufficiently good polarization separation or preservation, and, on the other hand, that there is no undesired mixing of the orthogonal polarizations in the feed networks.

Especially in aeronautical applications, the required polarization decoupling with linearly polarized signals places very high demands on the antenna system. Because such systems are typically on the Fuselage are mounted and have a two-axis positioner, the antenna aperture is always with its azimuth axis in the plane of the aircraft. The aircraft level is typically a tangential plane to the earth's surface. If the aircraft position and satellite position are not of the same geographical length, then the antenna aperture, when directed at the satellite, will always be twisted by a certain angle, which depends on the geographic length, with respect to the plane of the Clarke orbit. This so-called geographic skew can not be compensated in mobile applications by a rotation of the antenna about an axis perpendicular to the aperture plane, as is possible with stationary terrestrial antennas. Thus, an aeronautical antenna system must be able to meet the regulatory requirements despite the unfavorable length to aspect ratio even in the presence of a geographic skew up to a certain angle of rotation of typically approximately ± 35 °.

1. 1. minimal mögliche Dimension zur Erfüllung der regulatorischen Anforderungen,
2. 2.höchste Antenneneffizienz bei minimalem Gewicht,
3. 3. große Bandbreite um das Empfangs- und das Sendeband abzudecken (z. B. Ku-Band Betrieb: 10, 7-12, 75 GHz und 13, 75-14, 5 GHz),
4. 4. sehr gute Richtcharakteristik,
5. 5. hohe Polarisationstrennung,
6. 6. Kompensation des geographischen skews durch Nachführung der Polarisationsebenen bei linear polarisierten Signalen.

This results in the following problems for mobile, in particular aeronautical satellite antennas, which must be solved simultaneously:

1. 1. minimum possible dimension to fulfill the regulatory requirements,
2. 2.highest antenna efficiency with minimum weight,
3. 3. wide bandwidth to cover the receive and transmit bands (eg Ku-band operation: 10, 7-12, 75 GHz and 13, 75-14, 5 GHz),
4. 4. very good directional characteristics,
5. 5. high polarization separation,
6. 6. Compensation of the geographic skew by tracking the polarization planes in linearly polarized signals.

State of the art:

It is known that antennas which are designed as fields of horns, over a very high efficiency feature. If fields are fed by horns with a network of waveguides, then the attenuation of electromagnetic waves through such networks can be very small. Such a field is z. B. in the patent US 5243357 proposed. However, this is a pure receiving antenna (column 1, line 10 ff.). The very high polarization decoupling necessary for operation as a transmitting antenna can not be achieved with the proposed network of square waveguides. In addition, the design of the spacing between the radiator elements is comparatively large, since the square waveguides must have dimensions in the range of half the wavelength of the useful frequency for efficient waveguiding and the centers of the radiating elements are therefore far more than one wavelength apart. It is known that this leads to significant side lobes (so-called "grating lobes") in the antenna characteristic. In pure reception mode these side lobes are harmless. A regulatory allowed transmission mode is not possible because z. For example, CFR 25.209 and CFR 25.222 set very stringent side lobe suppression requirements. An improvement of the polarization separation can be achieved by separate feed networks. So z. B. in the US 2005/0146477 proposed to use their own food network for the left circular and the right circular polarization. The radiator elements (here aperture crosses) must, however, be fed serially. This severely limits the usable bandwidth. A typical Ku-band operation, z. B. with a reception band of 10.7 GHz to 12.75 GHz and a transmission band of 14.0 GHz to 14.5 GHz, is not possible with such an arrangement. In z. B. US 5568160 It is also proposed to feed the distribution network with aperture crosses. However, primary antenna elements here are square horns. The feed network is divided into a network for horizontal polarization and a network for vertical polarization. A high polarization decoupling is thus possible. However, due to the design, the radiator centers are also comparatively far apart, so that parasitic side lobes occur. The same problem occurs the z. In US 6225960 . WO 2006/061865 and GB 2247990 proposed arrangements. In the US 6,201,508 It is proposed to apply a grid ("crossed septum", column 3, line 26) over each individual horn to homogenize the aperture. However, the design of the beam centers is far more than one wavelength away from each other due to their design, and parasitic sidelobes due to phase correlation continue to occur. Also, by design, the device has a significant height (extension perpendicular to the aperture plane), making it hardly usable for mobile and, in particular, aeronautical applications (Ku band "0.37 m", column 5, line 15).

Drawings:

Fig. 1a-c

illustrate the structure according to the invention of a horn field aperture and the schematic structure of the feed networks;

Fig. 2

shows the detailed structure of the aperture surface;

Fig. 3a-d

show the back side of an antenna according to the invention and the detailed construction of the horn field with the feed networks for two orthogonal linear polarizations;

Fig. 4a-b

exemplify an E-field divider and an H-field divider of the feed networks;

Fig. 5a-b

show a typical antenna diagram of an antenna according to the invention,

Fig. 6

shows the back of an antenna according to the invention with frequency diplexers and amplifiers;

Fig. 7

represents a waveguide module according to the invention for polarization tracking;

Fig. 8

shows an aeronautical antenna system with a two-axis positioner;

Fig. 9

represents a combined E-field and H-field divider, with the aid of which the antenna can be tracked with high precision.

The object of the invention is to provide a broadband antenna system, in particular for aeronautical applications, which, with minimal dimensions, permits a regulatory compliant transmission and reception operation and the precise alignment of the antenna with the target satellites.

folgt, wobei k und m ganze Zahlen sind und 2k+m=N₁ gilt, und die Leistungen p_i,j, i=1..N₁, j=1..N₂, die Leistungsbeiträge der einzelnen primären Hornstrahler bezeichnen, die Aperturbelegung durch symmetrische und asymmetrische binäre E- und H-Leistungsteiler (7, 8) in jedem der beiden Speisenetzwerke für jede der beiden orthogonalen Polarisationen realisiert ist, und die gesamte Aperturfläche von einem Phasenegalisierungsgitter (9) abgedeckt ist, wobei die Maschen (10) des Phasenegalisierungsgitters eine quadratische Dimension mit Kantenlänge b aufweisen und jedenfalls näherungsweise b = 1, h = 2 b und b < λ gilt, sodass in der Richtung N₁ die Stege des Gitters über der Stoßkante zweier benachbarter Hornstrahler liegen und in Richtung N₂ die Stege des Gitters sich jedenfalls näherungsweise genau in der Mitte der Aperturfläche der einzelnen Hornstrahler befinden.This object is achieved with the invention according to claim 1. Fig. 1a-c The antenna for broadband satellite communication, in particular for mobile applications, consists of a field of primary horns (1), which are interconnected by a waveguide feed network (2), wherein the antenna consists of a Number N = N ₁ x N ₂ primary horns with N ₁ > 4 N ₂ , N ₁ and N _{2 are} even integers, for the total aperture area A of the antenna A = L x H with L ≥ 4 H and L <N ₁ λ, where λ denotes the minimum free-space wavelength of the electromagnetic wave to be transmitted or received, the primary horns allow the reception and transmission of two orthogonal linearly polarized electromagnetic waves by crossing a rectangular aperture area a = 1 xh with 1 < h and 1 <λ and at least approximately square output (3), where L = N ₁ 1, H = N ₂ h, and A = N ₁ x N ₂ x 1 xh = L x H gi and the primary horns (1) are fed directly at their output (3) via rectangular waveguides (4, 5) such that one of the orthogonal linear polarizations is fed in and out parallel to the aperture surface and the other of the orthogonal linear polarizations via a waveguide septum (6) in a plane perpendicular to the aperture surface and is discharged, the horns of the primary horns compressed and are perpendicular to the aperture surface of a length 1 _H <1.5 λ, the waveguide feed network (2) from a feed network for one of the two orthogonal linear polarizations (4) and a separate feed network for the other of the two orthogonal linear polarizations (5), each of the two feed networks is constructed as a binary tree with binary E and H power dividers (7, 8), so that the last power divider at the lowest level of the binary tree is the power of two half-apertures, each with N / 2 primary horns for each of the two orthogonal polarizations separately symmetrically merges, the aperture of the antenna at least approximately the relation $${\mathrm{p}}_{\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="3"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{<}\mathrm{...}\mathrm{<}{\mathrm{p}}_{\mathrm{k}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{=}\mathrm{...}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{>}\mathrm{...}\mathrm{>}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}$$

where k and m are integers and 2k + m = N ₁ , and the powers p _{i, j} , i = 1..N ₁ , j = 1..N ₂ denote power contributions of the individual primary horns, the aperture occupancy is realized by symmetrical and asymmetrical binary E and H power dividers (7, 8) in each of the two feed networks for each of the two orthogonal polarizations, and the entire aperture area is covered by a phase equalization grating (9), the meshes (10 ) of the Phasenegalisierungsgitters have a quadratic dimension with edge length b and at least approximately b = 1, h = 2 b and b <λ is valid, so that in the direction of N _1, the webs of the grid over the abutting edge of two adjacent horn and are in the direction N ₂ the In any case, webs of the grid are located approximately exactly in the middle of the aperture area of the individual horns.

By dimensioning the horn field with a number N = N ₁ x N ₂ primary horns, where N ₁ > 4 N ₂ , and N ₁ and N _{2 are} even integers, a rectangular antenna aperture is achieved that meets the requirements of the lowest possible Height in mobile, especially aeronautical, use is sufficient. This dimensioning rule also ensures that upon rotation of the antenna about the main axis of the beam necessarily associated with the rotation expansion of the main lobe remains low within the +/- 35 ° angle range that is important for the application. With a length to side ratio of 4: 1, the expansion in the Ku transmission band (14 GHz-14.5 GHz) is only a few tenths of a degree.

The angular range for the geographic skew of +/- 35 ° is therefore of particular importance, because then z. B. in Ku-band, the entire North American continent with only one satellite can be covered. This leads to a significant reduction in the cost of providing a corresponding service.

If N ₁ and N _{2 are} even numbers, then the horn field can be fed efficiently with a bi-directional binary feed network.

The dimensioning rule for the length L of the horn field, L <N ₁ λ, ensures that no parasitic sidelobes occur in the azimuth direction, which are generated by too large a distance of the beam centers of the primary horns. The wavelength λ must be the smallest of the wavelength occurring in the transmission mode. In Ku-band broadcasting this z. For example, the wavelength is 14.5 GHz, so λ≈2.07 cm. Only by suppressing parasitic side lobes is a regulatory permissible transmission mode possible.

The primary horns have, as in Fig. 1b and Fig. 2 shown, a rectangular aperture area a, with a = 1 xh and 1 <h. The horn radiation field is then in accordance with the rules L = N ₁ l, H = N ₂ h, and
A = N ₁ × N ₂ × 1 × h = L × H, where A denotes the total aperture area of the field. Thus, the aperture surfaces a of the primary horns in azimuth and elevation are close together and are aligned with their short edge in the azimuth and with their long edge in the elevation direction. With 1 <λ it is then achieved that with dense horn occupancy no parasitic side lobes in the azimuth direction can occur. If z. B. for the Ku band Transmission mode selected in the frequency band 14 GHz-14.5 GHz 1 <λ _max and l≈λ _max ≈2.07 cm, then one obtains in the inventive choice of h = 2 1 and N ₁ > 4 N ₂ a horn antenna field minimum dimension that can meet the regulatory requirements. Is regulatory z. For example, for the 3dB width Δ _{3dB of} the main lobe in azimuth 2 °, then the known approximation formula Δ _3dB = 51 ° / L _λ (eg. JD Kraus and RJ Marhefka, "Antennas: for all applications", 3rd ed., McGraw-Hill series in electrical engineering, 2002, pa. 374 ) with L _λ = L / λ _max = N _{1, min} a minimum number N _{1, min} = 26. For the minimum number of N ₂ , N _{2, min} , then N _{2, min} ≤ 4, the rule that N ₁ and N _{2 are} even integers, respectively.

If, in addition, the rule of claim 1 is used, that the feed network is designed as a binary tree, then a horn radiation field results with N ₁ = 32 and N ₂ = 4, ie L≈64cm and H≈16cm. If the Aperturbelegung by symmetrical and asymmetrical binary E and H power divider now selected according to the invention, then the antenna pattern can comply with the regulatory requirements.

The sizing of the primary horns also ensures that they can have a quadratic output that supports two orthogonal linear polarizations. The square output (3) is fed by two rectangular waveguides lying in orthogonal planes. This geometry ensures effective polarization separation. In addition, the feeding tube lying in a plane perpendicular to the aperture plane is provided with a waveguide septum (6), which prevents the parasitic migration of the orthogonal polarization in this waveguide branch. The transition from the square output (3) of the primary horn to the input of the rectangular waveguide of the one linear polarization lying in the aperture plane is typically designed stepwise. This can also improve polarization separation and broadbandness. A typical embodiment of the signal extraction from the primary horns is shown in FIG Fig. 2 shown.

In order to keep the dimensions of the horn field as small as possible, the horns of the primary horns are compressed in the beam direction. Their length perpendicular to the aperture surface is only 1 _H <1.5 λ. This length is much smaller than the length which would result according to the known sizing of horn apertures and leads without Phasenegalisierungsgitter to a significant impedance mismatch to the free space wave and thus to considerable reflection losses. However, if the aperture is provided with a phase-adjusting grating according to the invention, then the horns can be dimensioned according to the invention without significant losses occurring. This leads to a considerable reduction in the size of the overall antenna. The phase gating in antennas according to the invention therefore not only has the task to homogenize the phase assignment of the aperture, but also serves for the impedance matching of the primary horn to the free-space wave impedance.

In order to achieve the greatest possible polarization separation and a maximum instantaneous bandwidth, a separate feed network is provided for each of the two orthogonal polarizations. The separate feed directly from the horn output also has the advantage that the two linear orthogonal polarizations can be processed completely separately and a high-precision phase adjustment can take place. This is necessary in order to be able to achieve the accuracy required for the polarization tracking of typically <1 ° over the entire instantaneous bandwidth of typically more than 3 GHz. Also, the separation of the transmitting and receiving band is facilitated by appropriate frequency diplexer.

The construction of food networks as binary trees, as shown schematically in Fig. 1c shown, allows the use of high-precision binary symmetric and asymmetric E-field and H-field power dividers (7, 8), as exemplified in Fig. 4a and Fig. 4b are shown. This high precision is necessary to get one for both Polarizations to achieve almost identical frequency response over the entire instantaneous bandwidth, which is necessary in order to achieve the necessary precision in polarization tracking can. By design, high-efficiency phasing can then be achieved by a suitable combination of waveguide pieces with coaxial cable pieces over the entire instantaneous bandwidth. In addition, this has the advantage that the amplitude and phase assignment of the aperture can be set very accurately. This is necessary in order to be able to reliably comply with the regulatory envelope over the entire required transmission bandwidth of typically more than 500 MHz. It has been shown that, in contrast to multiple power dividers, production-related tolerances in binary structures typically result in larger feeding structures. The waveguides (2) of the feed networks are dimensioned for both polarizations such that on the one hand as lossless waveguide over the entire instantaneous bandwidth is achieved, and on the other hand is minimized by a high integration density of the required space. In Ku band z. B. therefore waveguides are used whose aspect ratio is substantially smaller than the standard ratio 1: 2. In the in Fig. 1a illustrated embodiment, the waveguide (2) have only an aspect ratio of 6.5: 16. This has been shown to be sufficient to cover the entire instantaneous bandwidth of 10.7 GHz-12.75 GHz and 13.75 GHz-14.5 GHz. Compared with waveguides with standard dimensions, this results in a significant volume reduction of about 20% in the feed networks and a corresponding weight reduction. So has the in Fig. 3a-d illustrated embodiment for the Ku-band total only one depth (extension perpendicular to the aperture plane) of about 15 cm, which is particularly for aeronautical applications of great advantage.

It is envisaged to implement the feed networks such that the line divider at the lowest level signals the two half-apertures with N / 2 primary horns respectively merges. This has the advantage that this power divider can also be designed as a combined E-field and H-field divider. Thus, not only the sum signal of the two half-apertures but also the difference signal can be tapped directly at the aperture output. If the difference signal is processed accordingly this enables the high-precision alignment of the antenna on the target satellites. For Ku-band broadcasting in the US z. For example, the CFR 25.222 standard requires a targeting accuracy of <0.2 °. This is possible with conventional methods of "open loop" tracking with the aid of position data (eg via GPS and / or inertial detectors) only over short periods of time. Then the transmission mode must be interrupted and the antenna with the help of the received signal to be realigned.

If, on the other hand, the aperture is constructed so that it can provide the difference signal, accuracies can be achieved with the help of a "closed loop" tracking, which are permanently << 0.2 ° in time.

In Fig. 1c the schematic structure of the two feed networks for the two orthogonal linear polarizations is shown. Directly at the output (3) of the primary horns (1) the two polarisations are separated and fed in two separate feed networks (4) (solid lines) and (5) (dotted lines). Both feed networks are designed as binary trees with E-field dividers (7) and H-field dividers (8). At the lowest level, the signals from N / 2 primary horns are symmetrically combined. To measure the difference signal of the two aperture halves for both polarizations, the lowest-level divider may be implemented as a combined E-field and H-field divider (30).

gehorcht, wobei k und m ganze Zahlen sind und 2k+m=N₁ gilt, und die Leistungen p_i,j, i=1..N₁, j=1..N₂, die Leistungsbeiträge der einzelnen primären Hornstrahler bezeichnen. Es hat sich gezeigt, dass Amplitudenbelegungen, die dieser Relation gehorchen - sofern alle anderen erfindungsgemäßen Merkmale vorhanden sind - Antennendiagramme erzeugen, welche die typischen regulatorischen Envelopen (z. B. definiert in CFR 25.209 und ETSI EN 302 186) einhalten können. Diese Klasse von Amplitudenbelegungen hat zudem, zusammen mit den Dimensionierungsvorschriften für das Hornstrahlerfeld, die einzelnen primären Hornstrahler und das Phasenegalisierungsgitter des Anspruch 1 die Eigenschaft, dass bei zunehmendem geographischem skew Winkel keine parasitären "grating lobes" auftreten, sondern das Niveau der Nebenkeulen in Azimutrichtung über die gesamte instantane Bandbreite abnimmt. Dies ist ein wesentlicher Vorteil erfindungsgemäßer Anordnungen gegenüber bisher bekannten Anordnungen. Der Effekt ist in Fig. 5a und Fig. 5b für eine typische Ausführungsform und für eine Frequenz im Ku-Sendeband (14.25 GHz) dargestellt. Der Winkel theta bezeichnet dabei den Winkel entlang der Tangente an den Clark-Orbit an der Stelle, an der sich der geostationäre Satellit befindet, und der skew-Winkel den Rotationswinkel der Apertur senkrecht zur Strahlrichtung, wenn die Antenne auf diesen Satelliten ausgerichtet ist. Die fett eingezeichnete Kurve ("FCC") markiert die regulatorische Envelope nach CFR 25.209, die vom Antennengewinn "gain" nicht überschritten werden darf. Fig. 5a zeigt den Winkelbereich -180° bis +180°, Fig. 5b den Bereich um die Hauptkeule.It is also intended to provide the aperture with a hyperbolic amplitude assignment, which at least approximately the relation $${\mathrm{p}}_{}$$$${\mathrm{}}_{\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="3"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{<}\mathrm{...}\mathrm{<}{\mathrm{p}}_{\mathrm{k}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{=}\mathrm{...}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{>}\mathrm{...}\mathrm{>}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}$$

where k and m are integers and 2k + m = N ₁ , and the powers p _{i, j} , i = 1..N ₁ , j = 1..N ₂ denote power contributions of the individual primary horns. It has been found that amplitude assignments that obey this relation - if all other features according to the invention are present - produce antenna diagrams which can comply with the typical regulatory envelopes (eg defined in CFR 25.209 and ETSI EN 302 186). This class of amplitude assignments, in addition to the sizing specifications for the horn field, the individual primary horns and the phase gating of claim 1, has the property that, as the geographic skew angle increases, no parasitic grating lobes occur, but the level of sidelobes in the azimuth direction the entire instantaneous bandwidth decreases. This is a significant advantage of arrangements according to the invention over previously known arrangements. The effect is in Fig. 5a and Fig. 5b for a typical embodiment and for a frequency in the Ku broadcast band (14.25 GHz). The angle theta denotes the angle along the tangent to the Clark orbit at the location where the geostationary satellite is located, and the skew angle the angle of rotation of the aperture perpendicular to the beam direction when the antenna is aligned with that satellite. The bold curve ("FCC") marks the regulatory envelope according to CFR 25.209, which must not be exceeded by the antenna gain "gain". Fig. 5a shows the angle range -180 ° to + 180 °, Fig. 5b the area around the main lobe.

Aperture occupancy is realized by symmetric and asymmetrical binary E and H power splitters (7, 8) in each of the two feed networks for each of the two orthogonal polarizations, and thus is effective over the entire instantaneous bandwidth. This has the advantage that also in the receiving band a very high directivity is achieved and the parasitic irradiation of signals from neighboring satellites is greatly reduced. A typical embodiment of the feed networks is in Fig. 1c shown. Typical Embodiments of E-Field Dividers (7) and H-Field Dividers (8) are in the FIGS. 4a and 4b shown.

As in Fig. 1a, 1b and 2 In addition, it is provided that the entire aperture area is covered by a Phasenegalisierungsgitter (9), wherein the meshes (10) of the Phasenegalisierungsgitters have a quadratic dimension with edge length b and in any case approximately b = 1, h = 2 b and b <λ applies , so that in the direction of N _1, the webs of the grid over the abutting edge of two adjacent horns are located and in the direction of N _2, the webs of the grid are anyway approximately exactly in the middle of the aperture surface of the individual horns (1). The dimensioning b = 1 and thus b <λ ensures that the phase gating in the azimuth follows the periodicity of the horn field and thus no additional parasitic side lobes occur. In the elevation direction, the webs of the phase gating grating divide the aperture surfaces of the primary horns into two equal parts, as in FIG Fig. 1a shown. This arrangement has the advantage that the phase occupation of the field is homogenized in both directions and that no parasitic side lobes caused by phase correlation occur even when the aperture is rotated about the main radiation direction. Because the grid has square cells, even in the presence of a geographic skew, no distortion of the E-field and H-field vectors occurs, even if, as in arrangements according to the invention, the aperture areas of the primary horn have an aspect ratio of 1: 2. Thus, the number of required primary horns in the elevation direction can be halved, since then they need not have an extension in this direction which is smaller λ. The topological requirements for the feed networks are thereby simplified considerably and an additional volume or weight reduction is achieved.

The extension of the phase gating grating (9) in the direction perpendicular to the aperture surface is typically between λ / 4 and λ / 2. This expansion is determined by the extension l _H of the horns horn horns, which according to the invention <1.5 λ. By varying both lengths, the instantaneous bandwidth and the impedance matching to the free-space wave can be adjusted according to the respective requirements. Arrangements according to the invention have the advantage over fields of unmodified horns that an additional degree of freedom exists for the aperture design and the antenna performance of the strongly shortened horns can thus be optimized with respect to the available installation space.

Further advantageous embodiments of the antenna are described below.

folgt, wobei k und m ganze Zahlen sind und m ≥ 2 k, 2k+m=N₁ und jedenfalls näherungsweise p_i,j = p_2k+m+1-i,j für i=1..N₁/2 gilt, und die Leistungen p_i,j, i=1..N₁, j=1..N₂, die Leistungsbeiträge der einzelnen primären Hornstrahler bezeichnen. Mit dieser Klasse von trapezförmigen Amplitudenbelegungen wird erreicht, dass die Zahl der asymmetrischen Leistungsteiler der Speisenetzwerke minimiert werden und dennoch den regulatorischen Anforderungen genügt werden kann. Die Netzwerke werden dadurch erheblich fehlertoleranter und einfacher zu fertigen. Für das oben genannte Beispiel einer Apertur für das Ku-Band mit N₁=32 und N₂=4, ergibt sich z. B. m=16 und k=8, sodass im Prinzip nur 8 unterschiedliche asymmetrische Leistungsteiler notwendig sind. Dies stellt eine erhebliche Vereinfachung dar. Ein Beispiel eines gemessenen Antennendiagramms einer erfindungsgemäßen Antenne mit trapezförmiger Aperturbelegung ist in Fig. 5a und 5b dargestellt.With regard to the regulatory conformity and because of the simpler manufacturing, it is advantageous if the aperture of the antenna at least approximately the relation $${\mathrm{p}}_{}$$$${\mathrm{}}_{\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="3"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{<}\mathrm{...}\mathrm{<}{\mathrm{p}}_{\mathrm{k}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{=}\mathrm{...}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{>}\mathrm{...}\mathrm{>}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}$$

follows, where k and m are integers and m ≥ 2 k, 2k + m = N ₁ and at any rate approximately p _{i, j} = p _{2k + m + 1-i, j} for i = 1..N _1/2 applies , and the powers p _{i, j} , i = 1..N ₁ , j = 1..N ₂ , denote the power contributions of the individual primary horns. With this class of trapezoidal amplitude assignments it is achieved that the number of asymmetrical power dividers of the feed networks can be minimized and still meet the regulatory requirements. The networks become considerably more fault-tolerant and easier to manufacture. For the above-mentioned example of an aperture for the Ku band with N ₁ = 32 and N ₂ = 4, z. B. m = 16 and k = 8, so that in principle only 8 different asymmetrical power dividers are necessary. This represents a considerable simplification. An example of a measured antenna diagram of an antenna according to the invention with a trapezoidal aperture is shown in FIG Fig. 5a and 5b shown.

folgt, wobei k und m ganze Zahlen sind und m ≥ 2 k, 2k+m=N₁ und jedenfalls näherungsweise p_i,j = p_2k+m+1-i,j für i=1..N₁/2 gilt, und die Leistungen p_i,j, i=1..N₁, j=1..N₂, die Leistungsbeiträge der einzelnen primären Hornstrahler bezeichnen und die Leistungen p_i,j bis p_k,j sowie die Leistungen p_k+m,j bis p_2k+m,j jeweils linear voneinander abhängig sind, sodass die p_1,j bis p_k,j und die p_k+m,j bis p_2k+m,j jeweils zumindest näherungsweise auf einer Geraden liegen, und die Steigungen der beiden Geraden sich jedenfalls näherungsweise nur durch das Vorzeichen unterscheiden.A further simplification of the production can be achieved in that the aperture of the antenna at least approximately the relation $${\mathrm{p}}_{}$$$${\mathrm{}}_{\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{<}{\mathrm{<figure-callout\; id="3"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{<}\mathrm{...}\mathrm{<}{\mathrm{p}}_{\mathrm{k}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{=}\mathrm{...}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{1}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{2}\mathrm{.}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{3}\mathrm{.}\mathrm{j}}\mathrm{>}\mathrm{...}\mathrm{>}{\mathrm{<figure-callout\; id="2"\; label="p"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">p</figure-callout>}}_{\mathrm{2}\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{.}\mathrm{j}}$$

follows, where k and m are integers and m ≥ 2 k, 2k + m = N ₁ and at any rate approximately p _{i, j} = p _{2k + m + 1-i, j} for i = 1..N _1/2 applies , and the powers p _{i, j} , i = 1..N ₁ , j = 1..N ₂ , denote the power contributions of the individual primary horns and the powers p _{i, j} to p _{k, j} and the powers p _{k + m, j} to p _{2k + m, j are} each linearly dependent on each other such that the p _{1, j} to p _{k, j} and the p _{k + m, j} to p _{2k + m, j are} each at least approximately on a straight line, and the gradients of the two straight lines differ in any case approximately only by the sign.

A further advantageous embodiment is in Fig. 6 shown. If the antenna is used simultaneously for transmission and reception, then it is advantageous if the output of the feed network of each of the two orthogonal polarizations is connected by a waveguide (11) to a waveguide frequency diplexer (12) comprising the transmission frequency band from Receiving frequency band separates and the receiving frequency band output (13) of the two waveguide frequency diplexer (12) is in each case connected to a low-noise amplifier (14). There are provided waveguide components, since they can have the lowest attenuation and the highest isolation between the transmitting and receiving band. The receive frequency band output is each connected directly to a low noise amplifier, or preferably a waveguide, such that the parasitic noise performance through dissipative connections remains minimal.

Because of the low inherent noise of antennas according to the invention, cooled low-noise amplifiers can advantageously also be used here. In particular, with thermoelectrically cooled low-noise amplifiers or active or passive cryogenically cooled low-noise amplifiers, the receiving power of the antenna can be further increased.

In Fig. 7 A typical embodiment of a waveguide module for polarization tracking is shown. to Compensation of the geographic skew or other polarization rotations that are caused by corresponding movements of the antenna carrier, it is advantageous if the two orthogonal linearly polarized signals at the two outputs of the feed networks and / or at the outputs of the waveguide-frequency diplexer and / abut the outputs of the low-noise amplifier, are fed orthogonally into one or more waveguide modules, which consist of two along their axis connected hollow conductor pieces (15, 16) which against each other about the waveguide axis (17) motor-driven (18) by means of a transmission (19) can be rotated so that on the feed points (20) opposite side (21) of the waveguide modules in their polarization relative to the fed orthogonal linearly polarized signals rotated linearly polarized signals can be coupled out and so reconstruct the polarization of the incident waves t or the polarization of the waves to be transmitted can be controlled.

If the antenna is used for receiving and transmitting signals in different frequency bands, which may be far apart, then it is advantageous if the antenna has a waveguide module for polarization tracking for the transmit band and a separate waveguide module for polarization tracking for the receive band Is provided. The two waveguide modules can then be matched exactly to the corresponding band. As a result, a high-precision polarization tracking is achieved and caused by the frequency dispersion of the waveguide deviations can be minimized.

If the antenna is not only used for receiving and transmitting linearly polarized signals but also for receiving and / or transmitting circularly polarized signals, it is advantageous if the two orthogonally linearly polarized signals at the two outputs of the feed networks and / or at the outputs of the waveguide frequency diplexer and / or at the outputs of the low-noise amplifiers abutting with one or more 90 ° hybrid couplers are converted into orthogonal circularly polarized signals, so that also circularly polarized signals can be transmitted and / or received with the antenna. Also, with appropriate division of the transmit and receive signals, simultaneous operation with all four possible orthogonal polarizations (2 × linear + 2 × circular) is possible both in transmit mode and in simultaneous receive mode. An arrangement according to claim 1 thus has the highest possible variability.

In particular, for mobile applications, it is advantageous if the antenna is mounted on the elevation axis of a two-axis positioner and the waveguide modules for compensation of polarization rotations and / or the 90 ° hybrid coupler for the reconstruction of circularly polarized signals mounted on the azimuth platform of the positioner and the antenna and waveguide modules and / or the 90 ° hybrid couplers are interconnected with flexible high frequency cables. This arrangement of aperture and RF modules reduces the space required and facilitates integration, especially in aeronautical applications. A typical arrangement with a two-axis positioner is in FIG Fig. 8 shown. The Hornfeld aperture with feed network (22) is mounted on the elevation axis (23) and can be aligned by means of the elevation motor (24) and the elevation gear (25) in the elevation direction. With the aid of the azimuth motor (26), the antenna can be rotated about the azimuth axis (27). In the azimuth axis (27) a high-frequency rotary feedthrough with typically two channels is integrated. The electronics boxes (28) and (29) typically contain the control electronics for the positioner and additional high-frequency modules, such. B. modules according to claim 4 for polarization tracking. Also, boxes (28) and (29) may include processing electronics for high-precision tracking of the antenna, such as the electronics for processing the difference and sum signals of a combined E-field and H-field divisor. Because of the extreme environmental conditions to which particular hull-mounted aeronautical antennas are exposed, it may be advantageous if all or part of the antenna components are completely or partially silvered or coppered, all or part of the components are soldered and / or welded together and / or or glued, the antenna is provided with the exception of the aperture surface from the outside wholly or partially with a protective layer against the ingress of moisture, and in the plane between the primary horns (1) and the Phasenegalisierungsgitter (9) or in the plane of the horn outputs (3 ) a high frequency permeable waterproof film is introduced, which prevents the penetration of moisture into the primary horns and the waveguide feed network. Particularly in mobile applications, antennas according to the present invention typically consist of light metals such as aluminum or metallized plastic materials for reasons of weight reduction. To increase the antenna efficiency, it is advantageous to silver or copper these materials, since silver and copper have a very high RF conductivity. In order to ensure the required RF-tightness even at extreme rapid temperature changes, it is advantageous to solder at least critical parts of the aperture, to weld, or to glue, with the bonding typically electrically conductive adhesives are used. In addition, it may be necessary to protect the aperture against penetrating moisture, in particular condensation. Since it has been shown that the Phasenegalisierungsgitter must not be galvanically connected to the primary horns, it is advantageous to attach a necessary protective film between the plane of the primary horns and the Phasenegalisierungsgitter or in the plane of the horn outputs (3). This also has the advantage of very high mechanical stability even with strong changes in ambient air pressure.

However, to protect against moisture ingress can be applied to the Phasenegalisierungsgitter also from the outside a suitable RF-transmissive material. suitable Materials are in particular thin sheets of closed-cell foams (eg polystyrene, Airex, etc.). These plates can be glued and / or screwed to the surface of the phase gating grid with suitable flexible or viscoplastic adhesives, thus reliably preventing the ingress of moisture or other undesirable substances into the antenna. It is also advantageous hydrophobic and / or fungicidal equipment of the surface of the protective material as this prevents the unwanted colonization of biological organisms ("biological slime", fungi), which can adversely affect the high-frequency properties. The direct foaming of the openings of the Phasenegalisierungsgitters is possible.

Furthermore, it may be advantageous, especially for aeronautical applications, to provide the feed network with ventilation openings. Such vents may prevent condensate from accumulating inside the antenna, which may degrade the high frequency characteristics of the antenna. The ventilation openings are preferably attached to the long edge of the waveguide of the feed network, since only small high frequency currents flow. The dimension of the vents is typically much smaller than the wavelength for which the antenna is designed. However, the ventilation openings can also be mounted in the protective film of the Phasenegalisierungsgitters or in the Phasenegalisierungsgitter covering material, in which case larger openings can be realized. To prevent the ingress of dirt or other undesirable substances such. For example, to prevent oil, it may be advantageous to provide the vents with only water vapor permeable membranes (eg, oleophobic Gore membranes).

Fig. 9 represents a typical embodiment of a combined E-field and H-field divider, with the aid of which the antenna can be tracked with high precision. An advantageous Embodiment of the antenna is characterized in that the last waveguide power divider each of the two feed networks (4,5), which combines the signals of the two aperture halves with each N / 2 primary horns designed as a combined E and H divider (30) is such that both the sum signal (31) of the two symmetrical aperture halves and the difference signal (32) of the two symmetrical aperture halves is applied to this waveguide four-port and both the sum signal and the difference signal can be derived separately for each of the two orthogonal polarizations. Combined E-field and H-field divisors, so-called "magic teas", are four-element elements which, due to their geometric properties, provide both the sum signal of two supplied signals and the difference signal. Due to the binary structure of the feed networks it is possible in Hornfeld apertures according to the invention, instead of the last binary power divider to install a "magic tea". The difference signal can then be used either alone or together with the sum signal for high-precision alignment of the antenna on the target satellites. Since the difference signal disappears with exact alignment and the sum signal with exact alignment becomes maximum, z. B. the quotient of the signal powers P _difference / P _sum an extremely pronounced minimum (a so-called "zero") with exact alignment. In case of deviations from the exact orientation of the value of the quotient increases sharply and can be used for precise and fast tracking of the antenna. In addition, the phase of the RF signal at the differential port (32) has a zero crossing with exact alignment, so that the sign of the phase position indicates the direction in which the antenna must be tracked. Since the high-precision tracking in satellite antennas must in principle only along the Clarke orbit - the azimuth direction - must be made, it is sufficient to divide the aperture in half in the azimuth direction. In the elevation direction, "open loop" tracking is typically sufficient with the aid of position data and / or inertial detector data.

If the last power divider of the feed networks designed as a combined E-field and H-field divider (30), it is advantageous if the difference gate (32) of the combined E- and H-divider is equipped with a transmission band-cut filter, the prevents the penetration of transmission signals in the differential branch and the difference gate (32) is connected via the transmission band rejection filter with a low-noise amplifier. Since only the receiving signal must be used for high-precision tracking of the antenna by means of the signal of the differential gate, the low-noise amplifier which amplifies this signal can be effectively protected by a transmission band-cut filter from overdriving by the typically very strong transmission signal. Typically, this is a waveguide barrier filter is used because this class of components has only a very low attenuation. It is also advantageous to connect the low-noise amplifier directly to the transmit band blocking filter, preferably also through waveguides, as this can minimize signal loss. If the received signal strong enough then but also embodiments are conceivable in which the low-noise amplifier with a high-frequency cable, z. B. a coaxial line is connected to the transmission band blocking filter.

In particular, for mobile applications of the antenna, it is advantageous if the differential signals and / or a part of the sum signals of the two symmetrical aperture halves are forwarded to a processing electronics, which evaluates the strength and / or the phase position of the differential signals and / or the sum signals and these to the Passing control electronics of the antenna positioner, so that the control electronics can track the antenna so that the difference signal is minimal and so the antenna remains aligned with the target satellites when the antenna carrier moves relative to the target satellite. Due to the design, the antenna is optimally aligned with the target satellites when the received signal at the difference gate of the combined E-field and H-field divider becomes minimal. This optimality criterion can thereby in a simple way for high-precision tracking of the antenna at moving antenna carriers are used to be processed by a suitable electronic unit and forwarded to the controller of the antenna positioning system. Since the difference signal is permanently available in time, very high sampling rates and thus very fast tracking are possible even with very fast moving antenna carrier. Since the phase of the difference signal has a rapid zero crossing with optimum alignment with the target satellites, it is advantageous to also evaluate the phase position of the difference signal and to use for tracking. Typically, this allows an even higher precision in the tracking can be achieved than when only the strength of the difference signal is used. Since the antenna diagram of the Differenztors has two main lobes, which can show in the worst case on neighboring satellites, it is also advantageous to compare the difference signal in its strength and / or its phase position with the sum signal to exclude the parasitic interference of neighboring satellites in the tracking , In principle, by appropriate processing of the sum signal, since the antenna diagram of the Summentors has only a single, well-defined main lobe, parasitic interference terms are eliminated in the difference signal. This can be z. B. take place in that the difference signal is phase-aligned projected to the sum signal.

In order to track the antenna with high precision, in principle both beacon signals of the satellite and normal transponder signals can be used. In this case, a satellite beacon typically consists of a narrowband (<1 kHz) CW-like signal, while a normal transponder typically emits a broadband signal (in Ku-band, for example, 30 MHz), which is coded by phase coding (eg. QPSK) an information content is imprinted. In both cases, it may be advantageous to increase the signal to noise ratio of the difference gate signal and / or the summent signal by limiting the noise bandwidth. Also, the processing of high-frequency signals is facilitated by the fact that the processing electronics contains one or more fixed frequency mixer and / or one or more controllable frequency-variable mixer and one or more frequency filters for the difference signals and / or the sum signals, with which the difference signal or a part of the difference signal and / or the sum signal or a part of the sum signal in a defined baseband can be converted and processed there. By using controllable frequency-variable mixers ("frequency synthesizer"), the frequency range or transponder used for tracking can be specifically controlled.

For satellite signals of suitable strength, the difference signal and the sum signal in the baseband can be directly evaluated. For this purpose, it is advantageous if the strength of the difference signal and / or the sum signal in the baseband is measured with a suitable electronic circuit and transferred to the control electronics of the antenna positioner. In this case, standard electronic components, such as suitable amplifiers or power detectors, can be used, which are available at low cost for typical base bands in the MHz range.

In weak satellite signals or unfavorable satellite configurations, it may be advantageous if the difference signal and / or the sum signal in the baseband is digitized with an analog-to-digital converter and forwarded to a processor which has suitable evaluation methods to measure the strength and / or the Phase position of the difference signal and / or the sum signal to determine, and passes this information to the control electronics of the antenna positioner. By digitizing the signals, the software-controlled evaluation and thus the flexible adaptation to the respective conditions is possible. The processor can be z. B. consist of a specially programmed FPGA or a simple freely programmable arithmetic unit. To improve the signal quality z. B. software-implemented controllable filter can be used with the help of which the noise bandwidth can be optimized.

If the antenna signals are converted into a baseband for the purpose of high-precision tracking, digitized and forwarded to a processor, then it is advantageous in particular for aeronautical applications in which the antenna carrier (eg the aircraft) can move at very high speed. if the processor has an evaluation method with which the Doppler frequency shift of the difference signal and / or the sum signal occurring during rapid movements of the antenna carrier can be compensated. In contrast to the electronic implementation of Doppler tracking electronics, the software-implemented tracking is relatively inexpensive to implement in a suitable processor if the signals are already in digitized form. Since the maximum Doppler shift can be calculated over the maximum velocity of the antenna carrier, it is possible to configure a software-implemented filter accordingly. Then z. B. using an FFT ("Fast Fourier Transform") determines the current frequency of the signal, the noise bandwidth adjusted accordingly and the strength of the signal are measured.

Since the antenna aperture in mobile and in particular aeronautical applications typically can not be rotated about the beam axis, it may be advantageous if a polarization rotation of the difference signal and / or the sum signal of the two aperture halves due to the spatial position of the antenna carrier is reflected by one or more waveguide modules Claim 4 or in that the processor of the processing electronics has a suitable evaluation method can be compensated. As a result, a mixing of the signals of different polarization and thus a signal interference, which may affect the precise tracking prevented. In principle, this is depending on the application, two methods, the use of waveguide modules according to claim 4 and the software processing, available. Since the position of the antenna carrier, z. B. via GPS, typically known is, the polarization rotation can be calculated in a simple manner and can then be transferred to the control of the waveguide module or to the processor.

If the signals of the difference gate and the sum gate are present in digitized form, then it has been shown that it is advantageous if the evaluation method of the processor is to multiply two or more temporally successive values of the amplitude of the baseband difference signal and these products over a certain time .DELTA.t sum up to a sum S ₁ , in each case two or more temporally successive values of the amplitude of the baseband sum signal multiply and accumulate these products over a certain time .DELTA.t to a sum S ₂ , after the expiration of the period .DELTA.t the quotient S ₁ / S ₂ and / or another suitable function f (S ₁ , S ₂ ), the value obtained thereby by the method of the smallest distance or another suitable method with the standard curve f _N (δ , S ₁ , S ₂ ), thereby determining the value of the deviation angle δ and these n to pass to the control electronics of the antenna positioner. With the aid of this method even difference signals can be processed for which the noise power is higher than the signal power. If the time interval Δt is selected appropriately, all the noise components disappear in the multiplication correlator and the strength of the typically generalized periodic signal becomes visible. If the sum signal is processed accordingly, then z. B. the quotient S ₁ / S ₂ regardless of the respective signal amplitudes, which is a great advantage with changing signal strengths. The signal strength independent standard curve f _N (δ, S ₁ , S ₂ ) can be calculated by simple mathematical methods. However, for precise tracking, the standard curve may also be measured using the method and a suitable satellite transponder or beacon and then stored. Because of its simplicity, the method can also be used in z. B. analog electronics can be implemented.

Since, in particular, aeronautical antennas are typically mounted under an aerodynamically optimized radome, it may be necessary due to space constraints to modify the rectangular shape of apertures according to the invention. In particular, in order to maintain the necessary distance to the underside of the radome, a rounding of the corners of the aperture (horns with powers p ₁₁ , p _{1N 1} , p _{1N 2 ,} p _{N 2 N 1} in Fig. 1b ) become necessary. It has been found that a change in the horn edges or a reduction in the horn opening and even the complete removal of horns horns at the corners of the aperture hardly affects the performance of the antenna and its positive characteristics with respect to the antenna characteristics.

In a not illustrated embodiment, the antenna is constructed according to the invention, up to a total of N physical not realized _half primary horn radiator, which are located at the edge of the aperture, or changed in their outline or reduced realized, the associated cells of the Phasenegalisierungsgitters are correspondingly so modified that the edges of the cells continue to lie on the edges of the primary horns, the aperture allocation according to the invention is realized only for complete lines of the field of primary horns containing N ₁ primary horns (see. Fig. 1b ), and the binary tree structure of the two feed networks (cf. Fig. 1c ) is trimmed accordingly in the absence of primary horns.

Claims (18)

1. Aerial for broadband satellite communication, in particular for mobile applications, comprising an array of primary horn radiators (1) which are connected together by a waveguide feed network (2), wherein the aerial consists of a number N=N₁ x N₂ of primary horn radiators, where N₁ and N₂ are even integers, characterised in that N₁ > 4 N₂, for the total aperture area A of the aerial A=L x H, where L ≥ 4 H and L < N₁ λ, where λ denotes the minimum free-space wavelength of the electromagnetic wave to be transmitted or to be received, the primary horn radiators enable the reception and the transmission of two orthogonal linearly polarised electromagnetic waves in that they have a rectangular aperture area a = 1 x h where 1 < h and 1 < λ, and have an approximately square output (3), where L = N₁ 1, H = N₂ h and A = N₁ x N₂ x 1 x h = L x H, and the primary horn radiators (1) are fed directly at their output via rectangular waveguides (4, 5) such that one of the orthogonal linear polarisations is supplied and carried away parallel to the aperture area, and the other of the orthogonal linear polarizations is supplied and carried away via a waveguide septum (6) in a plane at right angles to the aperture area, the horns of the primary horn radiators are compressed and have a length 1_H < 1.5 λ at right angles to the aperture area, and the waveguide feed network (2) comprises a feed network for one of the two orthogonal linear polarizations and a separate feed network for the other of the two orthogonal linear polarizations, each of the two feed networks is in the form of a binary tree with binary E and H power dividers (7, 8), so that the respective last power divider on the lowest level of the binary tree combines the powers of two half-apertures, in each case with N/2 primary horn radiators for each of the two orthogonal polarizations, separately and symmetrically, the aperture configuration of the aerial in each case approximately follows the relationship: $${\mathrm{p}}_{\mathrm{1}\mathrm{,}\mathrm{j}}\mathrm{<}{\mathrm{p}}_{\mathrm{2}\mathrm{,}\mathrm{j}}\mathrm{<}{\mathrm{p}}_{\mathrm{3}\mathrm{,}\mathrm{j}}\mathrm{<}\mathrm{\dots }\mathrm{<}{\mathrm{p}}_{\mathrm{k}\mathrm{,}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{1}\mathrm{,}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{2}\mathrm{,}\mathrm{j}}\mathrm{=}\mathrm{\dots }\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{,}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{1}\mathrm{,}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{2}\mathrm{,}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{3}\mathrm{,}\mathrm{j}}\mathrm{>}\mathrm{\dots }\mathrm{>}{\mathrm{p}}_{\mathrm{2}\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{,}\mathrm{j}}$$

   

   
   where k and m are integers and 2k+m=N₁, and the powers pij, i=1 ... N₁, j=1 ... N₂, denote the power contributions of the individual primary horn radiators, the aperture configuration is formed by symmetrical and asymmetrical binary E and H power dividers (7, 8) in each of the two feed networks for each of the two orthogonal polarizations, and the entire aperture area is covered by a phase equalization grid (9), where the meshes (10) of the phase equalization grid (9) have a square dimension with an edge length b, and in each case approximately b=1, h=2 b and b < λ, such that in the direction N₁ the webs of the grid lie above the abutting edge of two adjacent horn radiators (1) and in the direction N₂, the webs of the grid are each located approximately precisely at the centre of the aperture area of the individual horn radiators (1).
2. Device according to claim 1, characterised in that the aperture configuration of the aerial in each case approximately follows the relationship: $${\mathrm{p}}_{\mathrm{1}\mathrm{,}\mathrm{j}}\mathrm{<}{\mathrm{p}}_{\mathrm{2}\mathrm{,}\mathrm{j}}\mathrm{<}{\mathrm{p}}_{\mathrm{3}\mathrm{,}\mathrm{j}}\mathrm{<}\mathrm{\dots }\mathrm{<}{\mathrm{p}}_{\mathrm{k}\mathrm{,}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{1}\mathrm{,}\mathrm{j}}\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{2}\mathrm{,}\mathrm{j}}\mathrm{=}\mathrm{\dots }\mathrm{=}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{,}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{1}\mathrm{,}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{2}\mathrm{,}\mathrm{j}}\mathrm{>}{\mathrm{p}}_{\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{+}\mathrm{3}\mathrm{,}\mathrm{j}}\mathrm{>}\mathrm{\dots }\mathrm{>}{\mathrm{p}}_{\mathrm{2}\mathrm{k}\mathrm{+}\mathrm{m}\mathrm{,}\mathrm{j}}$$

   

   
   where k and m are integers and m ≥ 2k, 2k+m=N₁ and, in each case approximately, pi,j=P_2k+m+1-i,j for i=1 ... N₁/2, and the powers p_i,j, i=1 ... N₁, j=1 ... N₂ denote the power contributions of the individual primary horn radiators.
3. Device according to any one of the preceding claims, characterised in that the output of the feed network of each of the two orthogonal polarizations is in each case connected by means of a waveguide (11) to a waveguide frequency diplexer (12), which separates the transmission frequency band from the reception frequency band, and the reception frequency band output (13) of the two waveguide frequency diplexers (12) is in each case connected to a low-noise amplifier (14).
4. Device according to any one of the preceding claims, characterised in that the two orthogonally linear polarized signals which are present at the two outputs of the feed networks and/or at the outputs of the waveguide frequency diplexers (12) and/or at the outputs of the low-noise amplifiers (14) are fed orthogonally into one or more waveguide modules, which consist of two waveguide pieces (15, 16) which are connected to one another along their axis and can be rotated, driven by motors, with respect to one another about the waveguide axis (17), so that on the opposite side (21) of the waveguide modules to the feed points (20) linearly polarized signals whose polarization has been rotated with respect to the orthogonally linear-polarized signals fed in can be output, and the polarization of the incident waves can thus be reconstructed, or the polarization of the waves to be transmitted can be controlled.
5. Device according to claim 4, characterised in that the aerial is equipped with a waveguide module for polarization tracking for the transmission band and with a waveguide module separate from the former for polarization tracking for the reception band.
6. Device according to any one of the preceding claims, characterised in that the two orthogonally linear-polarized signals, which are present at the two outputs of the feed networks (2) and/or at the outputs of the waveguide frequency diplexers (12) and/or at the outputs of the low-noise amplifiers (14), are converted by one or more 90° hybrid couplers to orthogonal circular-polarized signals, so that the aerial can also be used to transmit and/or receive circular polarized signals.
7. Device according to any one of the preceding claims, characterised in that the aerial is fitted onto the elevation axis (23) of a two-axis positioner, and the waveguide modules (15, 16) according to claim 4 and/or the 90° hybrid couplers according to claim 5 are fitted on the azimuth platform of the positioner, and the aerial and the waveguide modules (15, 16) and/or the 90° hybrid couplers are connected to one another by means of flexible high frequency cables.
8. Device according to any one of the preceding claims, in particular for aeronautical applications, characterised in that all or some of the components of the aerial are entirely or partially silver-plated or copper-plated, all or some of the components are soldered and/or welded and/or adhesively bonded to one another, the aerial with the exception of the aperture area is provided entirely or partially from the outside with a protective layer against the penetration of moisture, and a watertight film is introduced in the plane between the primary horns (1) and the phase equalization grid (9) or in the plane of the horn outputs (3) which film prevents the penetration of moisture into the primary horns and the waveguide feed network.
9. Device according to one of the preceding claims, characterised in that the last waveguide power divider of each of the two feed networks (4, 5) which combines the signals from the two aperture halves with in each case N/2 primary horn radiators (1) is designed as a combined E and H divider (30) such that both the sum signal (31) of the two symmetrical aperture halves and the difference signal (32) of the two symmetrical aperture halves are applied to this waveguide four-port network, and both the sum signal and the difference signal can be passed out separately for each of the two orthogonal polarizations.
10. Device according to claim 9, characterised in that the difference port (32) of the combined E and H divider is equipped with a transmission band stop filter, which prevents the transmission signals from entering the difference branch, and the difference port (32) is connected via the transmission band stop filter to a low-noise amplifier.
11. Device according to claim 9, in particular for mobile applications, characterised in that the difference signals and/or a portion of the sum signals of the two symmetrical aperture halves are passed to processing electronics, which evaluate the strength and/or the phase angle of the difference signals and/or of the sum signals and transfers/transfer them/this to the control electronics of the aerial positioner, such that the control electronics can readjust the aerial such that the difference signal is minimal, and the aerial thus remains aligned with the target satellites when the aerial carrier is moving relative to the target satellite.
12. Device according to claim 11, characterised in that the processing electronics for the difference signals and/or the sum signals contains one or more fixed frequency mixers and/or one or more controllable variable-frequency mixers and one or more frequency filters, by means of which the difference signal or a portion of the difference signal and/or the sum signal or a portion of the sum signal can be converted to a defined baseband and can be processed there.
13. Device according to claim 12, characterised in that the strength of the difference signal and/or of the sum signal in the baseband is measured by a suitable electronic circuit, and is transferred to the control electronics of the aerial positioner.
14. Device according to claim 12, characterised in that the difference signal and/or the sum signal is digitized in the baseband by an analogue/digital converter and is passed to a processor which has suitable evaluation methods for determining the strength and/or the phase angle of the difference signal and/or of the sum signal and for transferring this information to the control electronics of the aerial positioner.
15. Device according to claim 14, in particular for aeronautical applications, characterised in that the processor has an evaluation method by means of which it is possible to compensate for the Doppler frequency shift which occurs in the difference signal and/or in the sum signal when the aerial carrier is moving fast.
16. Device according to claim 9, characterised in that a polarization rotation of the difference signal caused by the spatial position of the aerial carrier and/or the sum signal of the two apertures halves can be compensated for by one or more waveguide modules according to claim 4, or by the processor in the processing electronics having a suitable evaluation method.
17. Device according to claim 14, characterised in that the evaluation method of the processor consists of multiplying two or more successive values of the amplitude of the baseband difference signal and adding these products over a specific time Δt to form a sum S₁, multiplying two or more successive values of the amplitude of the baseband sum signal in each case and adding these products over a specific time Δt to form a sum S₂, forming the quotient S₁/S₂ and/or some other suitable function f(S₁, S₂) after the time interval Δt has elapsed, comparing the value obtained in this way with the standard curve f_N (δ, S₁, S₂), which is known from a calibration measurement or calculation using the shortest-interval method or some other suitable method, determining in this way the value of the error angle δ and transferring the latter to the control electronics of the aerial positioner.
18. Device according to any one of the preceding claims, characterised in that up to a total of N₁/2 primary horn radiators, which are located at the edge of the aperture, are not physically implemented, or their border is changed or is reduced in size, the associated cells of the phase equalization grid are correspondingly modified such that the edges of the cells still lie on the edges of the primary horn radiators (1), the aperture configuration according to claim 1 or claim 2 is implemented only for complete rows in the array of primary horn radiators (1) which contain N₁ primary horn radiators (1), and the binary tree structure of the two feed networks (4, 5) is appropriately tailored when primary horn radiators are absent.

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