EP1678787A1 - Device and method for emitting and/or receiving electromagnetic radiation - Google Patents

Abstract

The invention relates to a device (100) and a method for emitting and/or receiving electromagnetic radiation. The aim of the invention is to improve corresponding generic devices and methods so that the angle ( THETA ) of the ray beams of the device (100) in elevation (E) can be adjusted in a simple and economical manner. For this purpose, the phase shift ( DELTA <) between the electromagnetic radiation emitted and/or received by various antenna elements (32, 34, 36, 38) or the angle ( THETA ) of emission and/or receipt of the electromagnetic radiation in elevation (E) is adjusted by variation of the effective permittivity ( epsilon eff), especially the spreading coefficient, of the line (20) and/or by arranging at least one element (50, 52, 54) that consists at least partially of a conductive material, especially at least partially of a metal, at a variable distance to the line (20) and/or to the antenna elements (32, 34, 36, 38).

Description

 Device and method for emitting and / or receiving electromagnetic radiation

Technical field

The present invention relates to a device for emitting and / or receiving electromagnetic radiation, in particular electromagnetic high-frequency radar radiation, comprising at least one single-layer or multi-layer, in particular also at least one metallic layer, substrate on which at least one planar line, in particular in the form of a ribbon line or in the form of a symmetrical or asymmetrical coplanar line or in the form of a microstrip line or in the form of a slot line or in the form of a coplanar two-band line, with at least two antenna elements, in particular beam elements, of which at least in particular is applied partially serial feed and / or in particular at least partially in-phase feed and / or in particular at least partially phase and / or amplitude symmetrical feed, for example

by means of direct or capacitive coupling of at least one feed network on the upper side of the substrate facing the antenna elements or

- By means of at least one slot electromagnetic coupling of at least one feed network from the underside of the substrate facing away from the antenna elements or

- Via at least one electrical feedthrough from the underside of the substrate facing away from the antenna elements, wherein at least one metallization layer can be arranged on the underside of the substrate facing away from the antenna elements.

The present invention further relates to a method for

Radiate and / or receive electromagnetic radiation, in particular electromagnetic H [och] F [requenz] radar radiation, by means of at least two antenna elements, in particular beam elements, via at least one planar line, in particular in the form of a ribbon line or in the form of a symmetrical line or asymmetrical coplanar line or in the form of a microstrip line or in the form of a slot line or in the form of a coplanar two-band line, in particular at least partially in series and / or in particular at least partially in phase and / or in particular at least partially in phase and / or amplitude symmetry, for example

by means of direct or capacitive coupling of at least one feed network on the upper side of at least one single-layer or multi-layer, in particular also at least one metallic layer, substrate having the antenna elements

by means of electromagnetic coupling of at least one feed network, which takes place through at least one slot in each case, from the underside facing away from the antenna elements of at least one single-layer or multi-layer, in particular also at least one metallic layer, substrate or

are fed via at least one electrical feedthrough from the underside facing away from the antenna elements of at least one single-layer or multilayer substrate, in particular also having at least one metallic layer, at least one on the underside of the substrate facing away from the antenna elements

Metallization layer can be provided.

State of the art

Sensing the surroundings of a means of transportation, in particular a motor vehicle, can in principle be done by means of LI [ght] D [etecting] A [nd] R [anging], by means of RA [dio] D [etecting] A [nd] R [anging] Video or ultrasound.

Radar sensors are becoming increasingly widespread on means of transportation, in particular on motor vehicles. Today's systems are used for automatic distance and / or

Cruise control; Future systems that are currently in the development stage should enable further functionalities, such as comfort systems, for example for stop-and-go operation, to safety systems that sharpen airbags and belt tensioners, optimize airbag deployment times or the

Collision warning and avoidance serve.

For such applications, there must be a large area around the

Means of transportation around or the entire environment of the means of transportation are scanned. For this, several

Sensors grouped around the means of transportation. The antennas of the commercially available automotive radar sensors at a frequency of 77 gigahertz are usually constructed as lens antennas; For future radar sensors at a frequency of 24 gigahertz and at a frequency of 77 gigahertz, planar antennas are being examined.

In this context, it is known from the prior art to use planar phased arrays in military radar systems:

In order to determine the angular positions of the target objects in the horizontal (azimuth A; cf. FIG. 1A, FIG. 1B and FIG. 1C), several beam lobes are formed during beam shaping on an analog level (cf. FIG. 1A and FIG. 1B). A phase-controlled group antenna G (“phased array”) with phase shifters P (cf. FIG. 1A) and with power divider L (cf. FIG. 1A) or with beam-shaping array is used for this purpose

Element or network S (cf. FIG. 1B) for generating the phase assignment, such as a Rotman / Archer / Gent lens, Butler matrix or Pale matrix.

The circuit-side outputs of the beam shaping network S (cf.

1B) can be mixed in parallel or serially into the baseband via a changeover switch and processed further by means of a processing unit V.

For beam shaping on a digital level (see FIG. 1C), the

Signals of all antenna columns for digital evaluation by means of successively connected low-noise amplifiers R (so-called L [ow] N [oise] A [mplifier]) and by means of low-pass filters T are mixed down into the baseband and digitized by means of analog / digital converters W.

The aforementioned concepts and principles are shown in Figure 1A, in Figure 1B and shown in Figure 1C each for the reception path.

In the vertical (elevation E; cf. FIG. 1A, FIG. 1B and FIG. 1C), a plurality of antenna elements are usually arranged one above the other, which are controlled within a column with a fixed phase and amplitude relationship to one another. A beam bundling in elevation E is thus achieved, which serves to increase the range and to suppress undesired targets which are at a very low or high altitude.

The group antenna G is usually built up planar on H [och] F [requenz] substrates, such as glass, ceramic or softboard. Patches are generally used as antenna elements of the group antenna G; alternatives are, for example, dipole or slot radiators. Current research is concerned with the transfer of these concepts into cost-effective systems for use in motor vehicles.

The installation of the radar sensors places high demands on the size and shape of the sensor, especially for the side area.

By using planar antennas, the sensor becomes flat. Since radar sensors cannot be installed behind the metallic outer walls of a vehicle, the (plastic) bumpers, plastic trims, scratches and scratches remain as installation space in the side area

Impact protection elements and spoilers.

In this context it should be borne in mind that the outer walls of

Motor vehicles are usually not exactly vertical. The radar sensor may therefore have to be installed at an angle because it is behind

Bumpers, trims and the like are available Installation space is not sufficient for vertical installation. The installation angles for the radar sensor generally differ for the different installation locations on a motor vehicle and / or between different motor vehicles.

For the S [hort] R [ange] R [adar] sensors currently being developed with, for example, four or six elements in elevation, the beam lobe in elevation is so wide that an inclined installation with a deviation of the order of about + five degrees to about + ten degrees from the vertical can be tolerated.

However, if planar short- to medium-range sensors or planar L [ong] R [ange] R [adar] - / A [daptive] C [ruise] C [ontrol] sensors are considered, the width of the beam is only increased in elevation be a few degrees to achieve the necessary antenna gain; then a beam lobe oriented as precisely as possible along the horizontal is absolutely necessary.

A beam deflection of three degrees upwards at a distance of thirty meters already leads to the maximum of the beam lobe

1.60 meters above the installation location of the sensor (cf. FIG. 2, on the basis of which the deflection of the beam lobe is visually illustrated in the case of an installation inclined by three degrees).

With regard to planar H [och] F [requenz] lines and planar antennas, planar H [och] F [requenz] lines, such as coplanar, are nowadays used for the construction of inexpensive H [och] F [requenz] circuits -, Microstrip, slot lines or the like used.

These three planar line types with the respective basic course of the electrical field of the basic mode are exemplary in FIG. 3A as a (symmetrical or asymmetrical) coplanar line (= so-called "coplanar waveguide"),

- In Figure 3B as a microstrip line (= so-called "microstrip line") and

- Outlined in Figure 3C as a slot line (= so-called "slot line").

In addition to the planar line types shown in FIG. 3A, in FIG. 3B and in FIG. 3C, there are a large number of other planar line types, such as ribbon lines or coplanar two-band lines (cf. for example R. K. Hoffmaπn, "Integrated Microwave Circuits",

Springer-Verlag, Berlin, 1983).

The following modifications can also occur:

- Metallization of the underside of the substrate; - multilayer substrates, whereby metallic layers can also occur;

- dielectric layers that cover the metallic conductor tracks.

Special microwave substrates, such as glass, ceramic or plastic, which can be mixed with fillers or reinforced with glass fibers, or the like, serve as the substrate. Planar antennas, for example with dipole, patch or slot radiators, are built up on this microwave substrate; Details on this are, for example, the presentation in P. Bhartia, K.V. S. Rao, R. S. Tomar, "Millimeter-Wave Microstrip and Printed Circuit Antennas", Artech House, Boston, London,

1991, removable.

FIG. 4A, FIG. 4B and FIG. 4C show possible configurations for the feeding of the planar antennas: in the case of the series feeding (so-called "series feed") according to FIG. 4A, an electrical path length occurs between the antenna elements, via which a fixed beam deflection in elevation can be set;

in the case of in-phase feeding (so-called “corporate feed”) according to FIG. 4B, all antenna elements are fed with the same phase, the amplitude usually decreasing symmetrically outwards in order to reduce the side lobes;

- A combination of the series feed (see FIG. 4A) and the in-phase feed (see FIG. 4B) is the phase and amplitude symmetrical feed according to FIG. 4C. Here the antenna elements are not necessarily fed in phase, but the phase deviations and the amplitude assignment are symmetrical, and the feed network is also smaller than in the case of in-phase feed (cf. FIG. 4B).

As can be seen from the two exemplary arrangements of a direct or capacitive series feed according to FIG. 5A and according to FIG. 5B, the antenna elements can be coupled directly to the feed network.

Alternatively, the antenna elements of the

Substrate underside

- by electromagnetic coupling (so-called slot coupling; see FIG. 6A) or

- serially fed via electrical high-frequency feedthroughs (so-called "vias"; see FIG. 6B) (see P. Bhartia, KVS Rao, RS Tomar, "millimeter-wave microstrip and printed circuit antennas") , Artech House, Boston, London, 1991).

The power distribution network is accordingly either on the same metal level as the antenna elements or on the Antenna elements opposite substrate side. In the latter case, the substrate can have an internal, locally interrupted metallization or can be constructed from a plurality of metallic and dielectric layers. Furthermore, the power distribution and feeding can be done on an inside

Substrate layer take place.

As far as the beam swiveling in elevation is concerned, the beam lobes can be swiveled in elevation by adjusting the phase relationship between the antenna elements in elevation, so that the

Beam lobes are aligned at a desired angle in the vertical (generally parallel to the horizontal plane) when the radar sensor is installed at an angle.

This beam deflection by phase shift between the

Radiator elements are illustrated in FIG. 7, with general principles and the functional relationship between the phase shift Δφ and the deflection angle Θ in SK Koul, B. Bhat, "Microwave and Millimeter Wave Phase Shifters", Volume 1 and Volume 2, Artech House, Boston , London, 1991.

In this context, the phase relationship between the radiator elements can be set by various measures (i) and / or (ii):

(i) A special design of the antenna or the feed network for each elevation angle can be implemented most simply by different line lengths in the feed network, via which the antenna elements are controlled.

This would require different H [och] F [requenz] boards for everyone Elevation angle desired by the user or for a certain number of sensibly graded elevation angles are produced and used in the corresponding sensors, which requires a considerable logistical and organizational effort in production and storage.

Mixing up the type plate or the H [och] F [requenz] board could also lead to the error that a sensor does not have the intended elevation angle; then the radar system will not work at all, with a reduced range or only under certain circumstances.

Such an error would be very difficult to find, because the wrong elevation angle cannot be recognized externally on the sensor, but only by opening the sensor and by inspecting the H [och] F [requenz] board precisely or by measuring the beam characteristics, which is practically impracticable in a motor vehicle workshop.

(ii) Electronically or otherwise adjustable phase shifters (cf. SK Koul, B. Bhat, "Microwave and Millimeter Wave Phase Shifters", Vol. 1 and Vol. 2, Artech House, Boston, London, 1991) are forbidden because of the number of phase shifters required, the associated costs and also the possibly increasing sensor size.

In the case of mechanically "trimmed" phase shifters, the aforementioned error can also occur here that the set elevation angle or the nameplate are mixed up.

The elevation angle of a radar sensor with electronic Controlled phase shifters could indeed be set to the correct value by exchanging information with the motor vehicle electronics, without errors occurring, but electronically controllable phase shifters are prohibited for cost reasons, as mentioned.

Presentation of the invention: task, solution, advantages

Based on the disadvantages and

The present invention is based on the inadequacies and, in recognition of the outlined prior art, the object of further developing a device of the type mentioned at the outset and a method of the type mentioned at the outset such that the angle of the beam lobes of the radar sensor can be adjusted in elevation in a simple and inexpensive manner is, the electronics and H [och] F [requenz] modules should remain unchanged for all realizable elevation angles.

Furthermore, the present invention is intended to rule out errors which arise as a result of confusing the phase shifter assembly and / or the nameplate or as a result of incorrect "trimming".

This object is achieved by a device with the features specified in claim 1 and by a method with the features specified in claim 18. Advantageous refinements and useful developments of the present invention are characterized in the respective subclaims.

The teaching according to the present invention is therefore based on Provision of one or more radar antennas that can be used in the manner essential to the invention for transmitting and / or receiving high-frequency electromagnetic radiation for non-perpendicular installation on or in means of transportation, in particular on or in motor vehicles.

The core of the present invention is the adjustment of the beam angle in elevation of the beam lobe of a radar antenna for means of transportation, in particular for motor vehicles, for which purpose the deliberate and targeted detuning of the at least one planar H [och] F [requenz] -

signal line

by changing the effective dielectric constant, in particular the coefficient of propagation, of the signal line (so-called "dielectric loading"), for example by means of at least one cap made of dielectric material, or

by attaching at least one element made of conductive material, for example at least one Ra [dar] dom [e] s made of metal, at a certain distance from the signal line or

- is used by combining these two technical measures.

The principle of so-called "dielectric loading" in mechanically controllable phase shifters is already known per se from the prior art (a simple way of realizing a mechanically controllable phase shifter is, for example, in S. K. Koul, B. Bhat,

"Microwave and Millimeter Wave Phase Shifters", Volume 1 and Volume 2, Artech House, Boston, London, 1991):

The principle of "dielectric loading" in mechanically controllable phase shifters consists in the effective dielectric constant of the

Change line. For this purpose, with planar lines, such as microstrip lines or striplines (see page 73 in SK Koul, B. Bhat, "Microwave and Millimeter Wave Phase Shifters", Vol. 1 and Vol. 2, Artech House, Boston, London, 1991), which surrounds the planar line Material changes, for example by sliding a plate of dielectric material over the line.

This principle can also be applied to other planar lines, such as coplanar lines, slot lines and a large number of symmetrical and asymmetrical strip lines; Analogously to this, the effective dielectric constant of a waveguide can also be changed by moving a piece of dielectric material within the waveguide (see page 75 in SK Koul, B. Bhat, "Microwave and Millimeter Wave Phase Shifters", Volume 1 and Volume 2, Artech House, Boston, London, 1991).

An alternative way of influencing the effective dielectric constant of a dielectric waveguide is to vary the distance of a conductive element from the waveguide. This principle is used in the document WO 00/54368 A1 from the prior art in order to pivot the beam by mechanical opening and closing

Moving a conductive plate over a dielectric waveguide to realize.

In contrast to the disclosure according to the publication WO 00/54368 A1, however, no dielectric is used in the present invention

Waveguide, but rather used a planar H [och] F [requenz] line, which in various embodiments, for example as a coplanar line (= so-called "coplanar waveguide"), as a microstrip line (= so-called "microstrip line"), as a slot line (= so-called "slot line") or as other symmetrical and / or asymmetrical strip lines, can be formed (cf. for the design of planar H [och] F [requenz] line also RK Hoffmann, "Integrated Microwave Circuits", Springer-Verlag, Berlin, 1983).

Compared to the prior art according to the publication WO 00/54368 A1, the new and inventive embodiment according to the present invention is advantageous in that the complicated processing of the dielectric waveguide on the substrate is eliminated.

There are also no transitions between the dielectric waveguide and the H [och] F [requenz] circuit which generates the transmit signal or processes the received signal. The H [och] F [requenzj circuit is expediently constructed with planar H [och] F [requenz] lines. The H [och] F [requenz] circuit and the planar H [och] F [requenz] lines, the phase (relation) and the antenna pattern of which are influenced by the dielectric cap, are advantageously located on the same substrate.

Furthermore, in the present invention - in contrast to the prior art according to the publication WO 00/54368 A1 - not only at least one conductive, in particular metallic, element, but alternatively or additionally, at least one dielectric element for influencing the phase (difference ) used between the individual beam elements of the radar sensor.

This is for a dielectric waveguide as described in the document

WO 00/54368 A1 is disclosed, in principle only possible to a very limited extent, because the waveguiding in the dielectric waveguide is based on the difference in the dielectric constant between the waveguide and the surrounding air. If a dielectric element were brought into the immediate vicinity of the dielectric waveguide, part of the

Power decoupled into the dielectric element and would go consequently undesirably lost.

A further delimitation criterion of the present invention for the disclosure according to the publication WO 00/54368 A1 is that the object known from the prior art relates to a "scanning" antenna, the beam lobe of which repetitively scans a certain angular range, whereas the present invention treated in a preferred manner the fixed setting of the beam by means of the cap of the (radar) sensor.

According to an essential development of the invention, both the present device and the present method can additionally

- the elevation angle, - the type designation of the sensor and / or

- The vehicle type and the installation location for which the sensor with its special elevation angle is intended, directly

- In at least one label of the preferably cap-shaped dielectric material and / or

- Be noted in at least one label of the preferably cap-shaped conductive element. Confusion of sensors is therefore excluded.

According to a preferred embodiment of the present invention, the exact setting of the different elevation angles

- over the distance of the dielectric cap and / or - over the distance of the conductive cap from the feed network (= so-called "feed network"). Alternatively or in addition to this, the exact adjustment of the different elevation angles can also take place via the material, in particular via the dielectric constant of the material, of the cap.

Again, alternatively or in addition to this, the exact setting of the different elevation angles can also be achieved by suitable structuring of the cap depending on the elevation angle, for example in the form of holes, in the form of grooves, in the form of columns, in the form of steps, in the form of Honeycomb and / or in the form of the like.

Structuring the dielectric or metal-coated cap with at least one periodic structure, for example with a P [hotonic] B [and] G [ap] structure, is particularly advantageous, so that a so-called

"Slow Wave" structure is created. With such a periodic structure, which has a pass band and stop band in frequency and is known per se, for example, from waveguides, particularly large phase shifts and thus particularly large elevation angles can be achieved.

In this context, the "slow wave" structure makes it possible to introduce the required phase shift in a direct connection between two patch elements [= antenna or beam (er) elements], without the need for bypass lines which run between the feeds of the antenna or beam (er) elements available space are difficult to accommodate and cause additional losses. A "slow wave" structure is particularly well suited for applications in the S [hort] R [ange] R [adar] because the "slow wave" structure is particularly broadband. Since the distance between the dielectric and / or conductive element and the high-frequency board comprising the substrate is to be set relatively precisely and is to be kept constant over the life of the sensor device according to the present invention, the tolerance range of this distance should be estimated in the

Range of a few tens of micrometers.

For this reason, the material of the dielectric body and / or the conductive body according to an expedient development of the present invention has a similar, optimally even the same thermal expansion coefficient as the material of the H [och] F [requenz] board, and in this case in particular how the material of the substrate.

If all dielectric and / or conductive elements or

Bodies for the different elevation angles are made of the same material or at least of a material that is similar in terms of thermal expansion behavior, the elevation angle can be adjusted by means of the structuring of the dielectric and / or conductive element discussed above.

According to a preferred embodiment of the present invention, the dielectric material and / or the conductive element can be mechanically, for example by clamping or screwing over spacers, or in direct contact with the H [och] F [requenz] circuit board, which can also be realized by selective contact surfaces be connected. An alternative or supplementary possibility is the selective or full-surface gluing of the dielectric and / or conductive body and H [och] F [requenz] board.

In an essential development of the present invention According to the invention and the present method, the dielectric material and / or the conductive element can also be constructed in several parts. For this purpose, for example, the element influencing the phases and thus the antenna pattern can be mounted above the feed or feed network or below the feed or feed network; at least one further, preferably cap-shaped element then protects the radar arrangement against environmental influences.

Alternatively or in addition to this, this can be the phases and thus the

Element influencing the antenna pattern can also be inserted into at least one recess in the cap, in order then to be mounted together with this cap over the feed network or under the feed network.

According to a particularly advantageous embodiment of the present invention, the transitions between areas that are out of phase and areas that are out of phase can be realized by gradual transitions between these areas. This means that the distance between the dielectric and / or metallic body

Planar conduction in the transition area preferably runs continuously, for example linearly trapezoidal, or varies in several small steps.

In this context, an exemplary

Design as Ra [dar] dom [e] or "radar dome") the metallization of the dielectric and / or metallic body in the area of the undisturbed piano lines can be omitted. The transition area to the specifically disturbed piano lines can, however, be metallized. In a preferred embodiment of the present invention, the feed network or feed network can be embodied in at least one other line type in order to have a greater influence on the phase by the dielectric material or by the conductive element. For example, the H [och] F [requenz] circuit can be constructed from microstrip lines (= so-called "microstrip lines"), whereas the feed network is coplanar in the area in which the phase and thus the antenna diagram are to be influenced.

This different embodiment is based on the fact that a larger proportion of the electromagnetic field in the air is conducted above the line in a coplanar or slot line than in a microstrip line; the influence of the dielectric cap or the conductive element is therefore greater.

In order to keep the radar beam in elevation at the same angle with different loads on a means of transportation without level control, in particular a motor vehicle without level control, the phase-influencing dielectric and / or conductive can

Element can be designed to be adjustable. Such adjustment can take place, for example, via at least one electric motor.

According to a preferred development of the present invention, the (radar) sensor has at least one coding element which is expediently accessible from the outside, such as at least one jumper or at least one switch.

Such a coding element is used for the purpose of the sensor

Angle evaluation communicated the installation position. Then the sensor can "upside down" and "upside down" depending on whether upward or downward beam deflection is desired.

In this way, the (radar) sensor only has to be designed for one type of cap element - dielectric or metal - and the beam deflection that can be achieved with such a type of cap element and only goes in one direction can be optimized or maximized.

The present invention further relates to at least one mechanically controllable phase shifter which is based on the variation of the spacing of at least one conductive element from at least one planar H [och] F [requenz] line, such as, for example, from at least one ribbon line,

- from at least one (symmetrical or asymmetrical) coplanar line (= so-called "coplanar waveguide"),

- from at least one microstrip line (= so-called "microstrip line"), - from at least one slot line (= so-called "slot line") or

- Based on at least one coplanar two-band line (for the definition of the line types: see page 93 in R. K. Hoffmann, "Integrated Microwave Circuits", Springer-Verlag, Berlin, 1983).

The present invention further relates to at least one dielectric waveguide in which the phase shift or the angle, in particular the elevation angle, of the radiation and / or reception of the electromagnetic radiation in elevation by variably spacing arrangement of at least one at least partially made of conductive material, in particular at least partially made of metal, formed element is adjustable. In this context, in the case of a dielectric waveguide, the arrangement of at least one conductive element is preferred over the arrangement of at least one dielectric element, since "dielectric loading" only functions to a very limited extent on a dielectric waveguide in that the waveguide of the dielectric waveguide is based on total reflection at the interface Air is based and the wave is no longer guided when the dielectric loading is increased due to one or more dielectric elements.

Finally, the present invention relates to the use of at least one device according to the type set out above and / or a method according to the type set out above in the automotive sector, in particular in the field of vehicle surroundings sensors, for example for measuring and determining the angular position of at least one object, as is also relevant in the context of a pre-crash sensor for triggering an airbag in a motor vehicle.

In this case, a sensor system, in particular a radar sensor system, is used to determine whether there will be a possible collision with the detected object, for example with another motor vehicle. If there is a collision, the speed and impact point of the collision are also determined.

Knowing this data, life-saving milliseconds can be obtained for the driver of the motor vehicle, in which preparatory measures can be carried out, for example, when the airbag is activated or when the belt system is tightened.

Other possible areas of application of the device and of the method According to the present invention, parking assistance systems, a blind spot detection or blind spot monitoring or a stop & go system as an extension to an existing device for adaptive automatic control of the driving speed, such as an A [daptive-] C [ruise-] C [ontrol] system (= system for adaptive

Cruise control).

Accordingly, the planar antenna system proposed according to the present invention can be used both in the L [ong] R [ange] R [adar] range and in A [daptive] C [ruise] C [ontrol] systems, for example the third

Generation, as well as in the S [hort] R [ange] R [adar] range.

In this context, L [ong] R [ange] R [adar] is generally understood to be a long-range radar for long-range functions, which is typically used at a frequency of 77 gigahertz for A [daptive] C [ruise] C [ontrol] functions becomes.

In principle, the S [hort] R [ange] R [adar] system with the antenna or antenna system proposed according to the present invention

Beam (er) elements and equipped with the dielectric or metallized, in particular cap-shaped bodies proposed according to the present invention, provided that the targeted adjustment of the elevation angle proves to be necessary.

This applies more to subsequent generations of the S [hort] R [ange] R [adar], if

- In particular, a stronger beam bundling in elevation in connection with an increase in range should take place at the receiving end or

- In particular, larger and therefore more focused on the transmission side Antenna arrays are used to further reduce the side lobes.

In this context, S [hort] R [ange] R [adar] is generally understood to be a short-range radar for short-range functions, which is typically used at a frequency of 24 gigahertz for parking assistance functions or for pre-crash functions to trigger an airbag.

Last but not least, the structure according to the present invention can be used in a S [hort] R [ange] R [adar] sensor in which the direction of the beam lobe is set in elevation by at least one vehicle-specific dielectric and / or conductive cap.

Brief description of the drawings

As already discussed above, there are various possibilities for advantageously designing and developing the teaching of the present invention. For this purpose, reference is made on the one hand to the claims subordinate to claims 1 and 18, and on the other hand further configurations, features and advantages of the present invention are explained in more detail below with reference to the exemplary embodiments illustrated by FIGS. 8A to 36.

It shows:

1A shows, in a partially schematic representation, a first arrangement for analog beam shaping via phase shifters according to the prior art; 1B is a partial schematic representation of a second arrangement for analog beam shaping via a beam shaping network according to the prior art;

1C shows, in a partially schematic representation, an arrangement for digital beam shaping according to the prior art;

2 shows a lateral representation of the deflection of the beam lobe when a radar sensor is installed obliquely according to the prior art;

3A shows a first device according to the prior art in cross-sectional representation (upper part of the picture) and in a top view (lower part of the picture), the planar line arrangement of which is designed as a coplanar line;

3B in cross-sectional representation (upper part of the picture) and in a top view (lower part of the picture) a second device according to the prior art, the planar line arrangement of which is designed as a microstrip line;

3C shows a third device according to the prior art, the planar line arrangement of which is designed as a slot line, in cross-sectional representation (upper part of the figure) and in a top view (lower part of the figure);

4A shows a schematic representation of a first possibility for feeding antenna elements in the form of a series feed according to the prior art; 4B shows a schematic representation of a second possibility for feeding antenna elements in the form of in-phase feeding according to the prior art;

4C shows a schematic representation of a third possibility for feeding antenna elements in the form of a phase and amplitude symmetrical feeding according to the prior art;

5A shows a top view of a first possibility for a direct or capacitive series feed of antenna elements according to the prior art;

5B is a top view of a second option for direct or capacitive series feeding of antenna elements according to the prior art;

6A in cross-sectional representation (upper right part of the picture), in side view (left part of the picture) and in a top view (lower right part of the picture) a first possibility for a series supply of antenna elements from the underside of the substrate by electromagnetic slot coupling according to the prior art;

6B in cross-sectional representation (upper right part of the picture), in side view (left part of the picture) and in a top view (lower right part of the picture) a second possibility for a series supply of antenna elements from the underside of the substrate via electrical H [och] F [requenz] bushings according to the state of the art; 7 shows a schematic illustration of an arrangement for beam deflection by phase shift between radiator elements according to the prior art;

8A shows a cross-sectional illustration of a first exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a coplanar line;

8B shows a cross-sectional illustration of the first exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a microstrip line;

8C shows a cross-sectional illustration of the first exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a slot line;

9A shows a cross-sectional illustration of a second exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a coplanar line;

9B shows a cross-sectional illustration of the second exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a microstrip line;

9C shows a cross-sectional illustration of the second exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a slot line;

10A shows a cross-sectional illustration of a third exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a coplanar line; 10B is a cross-sectional illustration of the third exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a microstrip line;

10C shows a cross-sectional illustration of the third exemplary embodiment of the device according to the present invention, the planar line arrangement of which is designed as a slot line;

11 shows a fourth exemplary embodiment of the device according to the present invention in cross-sectional representation (upper right part of the picture), in side view (left part of the picture) and in a top view (lower right part of the picture);

12 shows a fifth exemplary embodiment of the device according to the present invention in cross-sectional representation (upper right part of the image), in side representation (left part of the image) and in a top view (lower right part of the image);

13 shows a sixth exemplary embodiment of the device according to the present invention in cross-sectional representation (upper right part of the picture), in side view (left part of the picture) and in a top view (lower right part of the picture);

14 shows a seventh exemplary embodiment of the device according to the present invention in cross-sectional representation (upper right part of the image), in side representation (left part of the image) and in a top view (lower right part of the image);

15 shows a schematic representation of an eighth exemplary embodiment of the device according to the present invention; 16 shows a schematic representation of a ninth exemplary embodiment of the device according to the present invention;

17 shows a schematic representation of a device in which binary-shifted phase shift elements are installed;

18 shows a schematic illustration of a tenth exemplary embodiment of the device according to the present invention;

19 shows a schematic representation of an eleventh embodiment of the device according to the present invention;

20 shows a schematic illustration of a twelfth exemplary embodiment of the device according to the present invention;

21 shows a schematic representation of a thirteenth exemplary embodiment of the device according to the present invention;

22 shows a schematic illustration of a fourteenth exemplary embodiment of the device according to the present invention;

23 shows a schematic representation of a fifteenth exemplary embodiment of the device according to the present invention;

24 shows a schematic illustration of an exemplary embodiment of a simple feed network designed for simulation calculations according to the present invention; 25 shows a perspective illustration of an exemplary embodiment of a first simulation model of the arrangement with a simple feed network from FIG. 24 in the case of the provision of dielectric cap-shaped bodies according to the present invention;

26 shows a perspective illustration of an alternative embodiment to FIG. 25 of a simulation model of the arrangement with a simple feed network from FIG. 24 in the case of the provision of metallic cap-shaped bodies according to the present invention;

27 shows in three-dimensional plot the directivity measured in decibels in elevation of the arrangement with a simple feed network from FIG. 24 without a dielectric and / or metallic cap-shaped body according to the present invention;

FIG. 28 in two-dimensional graphic representation (so-called antenna diagram in elevation) shows the directivity in elevation of the arrangement with simple feed network from FIG. 24 plotted against the measured beam deflection angle, measured in decibels, without dielectric and / or metallic cap-shaped body according to the present invention for different frequencies;

29 shows in a two-dimensional graphical representation (so-called antenna diagram in elevation) the directivity in elevation of the arrangement, plotted against the beam deflection angle measured in degrees and measured in decibels 24 simple feed network of FIG. 24 without dielectric and / or metallic cap-shaped body according to the present invention, with dielectric cap-shaped bodies according to the present invention and with metallic cap-shaped body according to the present invention;

30 shows a perspective illustration of an exemplary embodiment of a second simulation model of an arrangement with a meandering feed network according to the present invention;

31 shows in two-dimensional graphic representation (so-called antenna diagram in elevation) the directivity in elevation of the arrangement with meandering feed network from FIG. 30 plotted against the beam deflection angle measured in degrees, measured in decibels, from FIG. 30 without dielectric and / or metallic cap-shaped body according to the present invention for different frequencies;

32 in a two-dimensional graphic representation (so-called antenna diagram in elevation) shows the directivity in elevation of the arrangement with a meandering feed network from FIG. 30 plotted against the beam deflection angle measured in degrees, measured in decibels, without dielectric and / or metallic cap-shaped body according to the present invention, with dielectric cap-shaped bodies according to the present invention and with metallic cap-shaped bodies according to the present invention; FIG. 33 in a two-dimensional graphic representation (so-called antenna diagram in elevation) shows the directivity in elevation of the arrangement with a meandering feed network from FIG. 30 plotted against the beam deflection angle measured in degrees and measured in decibels from FIG. 30 without dielectric and / or metallic cap-shaped body according to the present invention for different frequencies, with dielectric cap-shaped body according to the present invention for different frequencies and with metallic cap-shaped body according to the present invention for different frequencies;

34 shows a perspective illustration of an exemplary embodiment of a third simulation model of an arrangement with an in-phase feed network according to the present invention;

35 is a two-dimensional graphic representation (so-called antenna diagram in elevation) of the directivity plotted against the beam angle measured in degrees, measured in decibels in elevation of the arrangement with in-phase feed network from FIG. 34 without dielectric and / or metallic cap-shaped body according to the present invention, with dielectric cap-shaped body according to the present invention (beam deflection: "forwards") and with dielectric cap-shaped body according to the present invention (beam deflection: "backwards"); and

Fig. 36 in a two-dimensional graphic representation (so-called antenna diagram in elevation) against the in degrees measured beam deflection angle, directivity measured in decibels in elevation of the arrangement with in-phase feed network from FIG. 34 without dielectric and / or metallic cap-shaped body according to the present invention for different frequencies, with dielectric cap-shaped body according to the present invention (beam deflection: "forward" ) for different frequencies and with a dielectric cap-shaped body according to the present invention (beam deflection: "backwards") for different frequencies.

The same or similar configurations, elements or features are provided with identical reference symbols in FIGS. 1A to 36.

Best way to carry out the invention

In the following, the (radar) device 100 according to the present invention, which is especially designed for the short range, and a method related thereto for detecting, detecting and / or evaluating one or more objects will be explained by way of example.

In this context, the device 100, which functions as an antenna, can be used in an essential manner according to the invention for transmitting and / or receiving electromagnetic H [och] F [requenz] radar radiation.

For this purpose, the device 100 has a substrate layer 10 with a

Dielectric constant ε _r , ι on; A metallization layer 12 is applied to the underside 10u of the substrate 10 (cf. FIG. 3B: design according to the prior art; cf. FIG. 8B: first exemplary embodiment the present device 100; see. FIG. 9B: second exemplary embodiment of the present device 100; see. Figure 10B: third embodiment of the present device 100).

A planar configuration runs on the top 10o of the substrate 10

Feed network or feed network in the form of one or more lines 20; Examples are in FIGS. 3A, 3B, 3C (= designs according to the prior art) and in FIGS. 8A, 8B, 8C (= first exemplary embodiment of the present device 100), in FIGS. 9A, 9B, 9C (= second exemplary embodiment) of the present device 100) and in FIGS. 10A, 10B, 10C (= third exemplary embodiment of the present device 100) each show three different planar line types with the respective basic course of the electric field of the basic mode, namely - in FIGS. 3A, 8A, 9A , 10A a symmetrical coplanar line (= so-called "coplanar waveguide"),

- In Figures 3B, 8B, 9B, 10B a microstrip line (= so-called "microstrip line") and

- In Figures 3C, 8C, 9C, 10C, a slot line (= so-called "slot line").

The planar linework 20 leads to a plurality of antenna or beam elements 32, 34, 36, 38, likewise applied to the substrate-shaped H [och] F [requenz] board 10 (cf. FIGS. 4A, 4B, 4C, 5A, 5B , 6A, 6B: designs according to the prior art; see FIG. 11: fourth embodiment of the present device 100; see FIG. 12: fifth embodiment of the present device 100; see FIG. 13: sixth embodiment of the present device 100; see FIG. 14: seventh embodiment of the present device 100; see FIG. 15: eighth embodiment of the present device 100; see FIG. 16: ninth embodiment the present device 100; see. Figure 17: Device with three binary graded phase shift elements 60, 62, 64; see. Figure 18: tenth embodiment of the present device 100; see. Figure 19: eleventh embodiment of the present device 100; see. Figure 20: twelfth embodiment of the present device 100; see. Figure 21: thirteenth embodiment of the present device 100; see. Figure 22: fourteenth embodiment of the present device 100; see. Figure 23: fifteenth embodiment of the present device 100).

These emitter elements 32, 34, 36, 38 can be supplied in various ways, for example as a serial supply 22s (so-called "series feed": see FIGS. 4A, 5A, 5B, 6A, 6B: designs according to the prior art; see FIG. 11: fourth embodiment of the present device 100; see FIG. 12: fifth embodiment of the present device 100; see FIG. 13: sixth embodiment of the present device 100; see FIG. 14: seventh embodiment of the present device 100; see FIG. Figure 15: eighth embodiment of the present device 100; see Figure 22: fourteenth embodiment of the present

Device 100; see. Figure 23: fifteenth embodiment of the present device 100).

In such a series feed 22s, a direct or capacitive coupling of the feed or feed network takes place on the

Top side 10o of the substrate 10 (see FIGS. 5A, 5B: designs according to the prior art; see FIG. 11: fourth embodiment of the present device 100; see FIG. 12: fifth embodiment of the present device 100).

As an alternative to such a direct or capacitive coupling of the Feed network on the top 10o of the substrate 10, serial feed 22s can also take place from the bottom 10u of the substrate 10 by means of electromagnetic coupling of the feed network through a slot 32s, 34s, 36s, 38s (see FIG. 6A: Execution according to the prior art Technology; see FIG. 13: sixth exemplary embodiment of the present device 100; see FIG. 22: fourteenth embodiment of the present device 100; see FIG. 23: fifteenth embodiment of the present device 100).

As an alternative to such an electromagnetic coupling of the feed network from the bottom 10u of the substrate 10, a serial feed 22s can also be provided from the bottom 10u of the substrate 10 via an electrical feedthrough 32d, 34d, 36d, 38d (see FIG. 6B: version according to the prior art; see FIG. 14: seventh exemplary embodiment of the present device 100).

An alternative or supplementary method to feeding the antenna elements 32, 34, 36, 38 to the method of the series feed 22s is the in-phase feed 22g (= so-called "corporate feed": see FIG

4B: execution according to the prior art; see. Figure 17: Device with three binary graded phase shift elements 60, 62, 64; see. Figure 18: tenth embodiment of the present device 100; see. Figure 19: eleventh embodiment of the present device 100; see. Figure 20: twelfth embodiment of the present device 100; see. Figure 21: thirteenth embodiment of the present device 100).

A further alternative or supplementary method for feeding the antenna elements 32, 34, 36, 38 to the method of series feeding 22s and / or to the method of in-phase feeding 22g is the phase and Amplitude-symmetrical feed 22p (cf. FIG. 40: design according to the prior art; cf. FIG. 16: ninth exemplary embodiment of the present device 100).

The essence of the present invention can now be seen in the fact that the beam angle in elevation E of the radar antenna or radar device 100 provided for a motor vehicle 200 according to the present invention can be adjusted by the planar H [high] F [requency] signal line 20 and is specifically detuned.

This deliberate and deliberate detuning of the planar H [och] F [requenz] signal line 20 and thus the deliberate and deliberate influencing of the phase difference Δφ between the antenna elements 32, 34, 36, 38 and the resulting antenna diagram takes place in accordance with the first exemplary embodiment of the present invention FIGS. 8A, 8B, 8C by changing the effective dielectric number ε _e f _f , that is to say the propagation coefficient of the signal line 20 (so-called "dielectric loading"), by a cap made of dielectric material 40 with a dielectric constant ε _r , ₂ > 1 in one a certain distance above the planar signal line 20 is arranged.

Here, by increasing the dielectric constant ε _r , 2 of the dielectric material 40 above the line 20, the coefficient of propagation on the line 20 and thus the

Phase difference Δφ between two radiator elements 32, 34 and 34, 36 and 36, 38 can be increased.

The deliberate and targeted detuning of the planar H [och] F [requenz] signal line 20 and thus the deliberate and targeted influencing of the

Phase difference Δφ between the antenna elements 32, 34, 36, 38 and the resulting antenna pattern in the second exemplary embodiment of the present invention according to FIGS. 9A, 9B, 9C is carried out by attaching a plate-shaped or layered element 50 made of conductive material at a certain distance from the signal line 20.

In this case, by attaching the conductive element 50 above the line 20 with air in the intermediate space, the coefficient of expansion on the line 20 and thus the phase difference Δφ between two radiator elements 32, 34 or 34, 36 or 36, 38 can be reduced.

If - as in the case of the second exemplary embodiment according to FIGS. 9A, 9B, 9C - the phase difference .DELTA..phi. And thus the elevation angle .phi. Is set by a metallic element 50, then this metallic element 50 can be favorably replaced by a partial or complete one

Produce metallization of a plastic cap.

The deliberate and targeted detuning of the planar H [och] F [requenz] signal line 20 and thus the deliberate and targeted influencing of the phase difference Δφ between the antenna elements 32, 34, 36, 38 and the resulting antenna diagram is carried out in accordance with the third exemplary embodiment of the present invention 10A, 10B, 10C by combining these two technical measures (= dielectric element + conductive element) in the form of a cap made of dielectric material 40, the side of which facing away from the line 20 is coated with a conductive layer 50s. As an alternative to this, a variant is also conceivable in which the conductive element 50 is coated with one or more dielectric layers 40s.

The "dielectric loading" by means of the dielectric cap 40 (cf. figures

8A, 8B, 8C) or the attachment of the conductive element 50 (cf. figures 9A, 9B, 90) or the combination of these two technical measures (cf. FIGS. 10A, 10B, 10C) takes place by means of a corresponding one which is dependent on the desired elevation angle Θ

- Design of the dielectric cap 40 (see FIGS. 8A, 8B, 80) or - Design of the conductive cap 50 (see FIGS. 9A, 9B, 90) or

- Design of the dielectric cap 40 with conductive layer 50s (cf. FIGS. 10A, 10B, 10C) of the sensor 100 (for comparison, the respective undisturbed line 20 known from the prior art is shown in FIGS. 3A, 3B, 30).

With the aid of these three principles described above, individual phase shifters are not controlled according to the invention, but practically the entire feed network or feed network is detuned or larger parts of the feed network or feed network are detuned; for this reason the feed network is at least partially constructed as a series feed 22s (so-called "series feed") (see page 161 in P. Bhartia, KVS Rao, RS Tomar, "Millimeter-Wave Microstrip and Printed Circuit Antennas", Artech House, Boston, London, 1991).

To realize different elevation angles Θ only a different cap has to be fitted; the electronic and high-frequency modules of sensor 100 are the same for all elevation angles,, which is the case for directly coupled antenna elements 32, 34, 36 with series feed in FIG. 11 (= fourth exemplary embodiment of the present device 100) and in FIG 12 (= fifth exemplary embodiment of the present device 100) is illustrated.

In this case, the just designed dielectric cap 40, which is arranged at a relatively large distance from the circuit board 10, influences the line 20 and which extends between the beam elements 32, 34, 36 thus the phase Δφ of line 20 little.

In contrast, the graduated dielectric and / or partially metallized cap 40 according to FIG. 12 influences the line 20 running between the beam elements 32, 34, 36 and thus the

Phase Δφ of line 20 is stronger, the transition 40t between the area 40b (located on the left in FIG. 12), which influences the phase Δφ on line 20 (= phase "detuned" area), and that (located on the right in FIG. 12) located) area 40n, which does not influence the phase Δφ on the line 20 (= phase-wise "untuned" area), is carried out gradually. This means that the distance between the dielectric cap 40 and the line 20 in the transition region 40t varies continuously, namely linearly trapezoidal (see FIG. 12).

As the further the respective representation of the fourth

11 and the fifth embodiment according to FIG. 12, it is possible on the one hand to arrange the feed network on the same metallization level as the beam elements 32, 34, 36, 38, which means a direct or capacitive series supply of the directly coupled beam elements 32,

34, 36, 38 means (see pages 133 f. In P. Bhartia, K.V. S. Rao, R. S. Tomar, "Millimeter-Wave Microstrip and Printed Circuit Antennas", Artech House, Boston, London, 1991).

The dielectric cap 40 or the conductive cap 50 then simultaneously forms a ra [dar] dom [e] or a “radar dome”, that is to say a dome-shaped weather protection for the patch elements, which is permeable to electromagnetic radiation, for example in the form of a plastic cladding for the Radar 100 antenna system.

On the other hand, the feed network can also, like the respective one 13 shows the sixth exemplary embodiment according to FIG. 13 and the seventh exemplary embodiment according to FIG. 14, on which the side of the substrate 10 opposite the radiator elements 32, 34, 36, 38 are built.

The emitters 32 or 34 or 36 or 38 are in this case

by means of electromagnetic coupling through slots 32s or 34s or 36s or 38s (cf. sixth exemplary embodiment according to FIG. 13) or

by means of electromagnetic coupling through high frequency feedthroughs 32d or 34d or 36d or 38d (so-called "vias") or excited by the like, the dielectric cap 40 determining the elevation angle Θ on the back, that is located on the side of the sensor 100 remote from the beam.

This means that the phase Δφ and the resulting antenna pattern are influenced by the dielectric and / or metallized cap 40 in the series feed according to FIG. 13 (= sixth embodiment) and according to FIG. 14 (= seventh embodiment) through the substrate 10.

In this case, too, is the transition 40t between the area 40b (located on the left in FIG. 13 or in FIG. 14) which influences the phase Δφ on the line 20 (= phase "detuned" area) and the (located in FIG. 13 or the area on the left in FIG. 14)

40n, which does not influence the phase .DELTA..phi. This means that the distance between the dielectric cap 40 and the line 20 in the transition region 40t varies continuously, namely linearly trapezoidal (see FIGS. 13 and 14). While the eighth exemplary embodiment of the device 100 according to FIG. 15 illustrates the beam deflection caused by a plate-shaped body 40 made of dielectric material of the dielectric constant ε _r , 2 in the case of serial feed 22s (so-called "series feed"), FIG. 16 shows the ninth

Embodiment of the device 100, the beam deflection with phase-symmetrical feed 22p (see also the illustration in FIG. 4C from the prior art).

Due to its symmetry, the phase (and amplitude-symmetrical feed 22p) has advantageous properties in that a simpler design of the feed can be achieved for a power distribution that drops outwards from the center, particularly with regard to a reduction in the side lobes of the phase and amplitude symmetrical feed 22p

Symmetry advantageously has little or no "squint" in elevation E.

As shown in the ninth embodiment in FIG. 16, the respective phase difference Δφ between the antenna elements

32, 34, 36, 38 - on one side (= upper area in FIG. 16) of the central feed of such a feed or feed network by "dielectric loading" by means of the dielectric cap 40 and - on the other side (= lower area in Figure 16) of the central feed of such a feed or feed network can be reduced by providing a conductive body 50. The elevation angle Θ can thus also be set for this feed network.

In Figure 17, in Figure 18, in Figure 19, in Figure 20 and in Figure 21 are five Different variants of an in-phase feed 22g are shown, which do not require a phase difference of 360 degrees between the antenna elements 32, 34, 36, 38 and thus in particular for broadband radar systems (so-called U [ltra] W [ide] B [and] radar systems) and for broadband communication systems

(so-called U [ltra] W [ide] B [and] communication systems) are suitable.

In this context, U [ltra] W [ide] B [and] systems are generally understood to mean radar and communication systems which work with pulsed signals, the pulse length of which is very short and the latter

Bandwidth is therefore very large.

This is done in the dining network

a first, binary-shifted phase shift element 60 causing a phase shift of 2Δφ,

a second, phase-shifting phase shifting element 62, which effects a phase shift of Δφ, and

- A third, phase-shifting phase shifting element 64 causing a phase shift of Δφ is installed (see FIG. 17) by a certain beam deflection nΔφ (with n =

0 for the first beam element 32 or n = 1 for the second beam element 34 or n = 2 for the third beam element 36 or n = 3 for the fourth beam element 38).

By

a first dielectric element 40, which causes a phase shift of 2Δφ and is suitably structured,

a second, suitably structured dielectric element 42 causing a phase shift of Δφ and a third, suitably structured dielectric element 44 causing a phase shift of Δφ can be graded using the three binary

Phase shift elements 60, 62, 64 built-in phase shift nΔφ

are either compensated so that the beam deflection is reduced or even disappears (cf. tenth exemplary embodiment according to FIG. 18),

- or be amplified, so that the beam deflection is exemplarily increased to 2nΔφ (with n = 0, 1, 2, 3) (cf. eleventh embodiment according to FIG. 19).

Here, the three dielectric elements 40, 42, 44 are designed as suitably structured dielectric caps, the first dielectric cap 40 [<--> phase shift 2 Δφ] being twice as long as the second dielectric cap 42 [<--> phase shift Δφ] and how the third dielectric cap 44 [<-> phase shift Δφ] is formed.

Instead of using dielectric elements 40, 42, 44, it is also possible to use conductive elements 50, 52, 54 to compensate or to amplify the beam deflection, namely in such a way that

a first conductive element 50, which causes a phase shift of 2 (-Δφ) and is suitably structured,

a second, suitably structured conductive element 52, which causes a phase shift of -Δφ and

- a third, suitably structured conductive element 54, which causes a phase shift of -Δφ and which can be compensated for by means of the three binary graded phase shift elements 60, 62, 64, phase shift n Δφ - so that the beam deflection is reduced or even disappears (cf. zwölfes Embodiment according to Figure 20),

- or can be amplified so that the beam deflection is exemplarily increased to 2n Δφ (with n = 0, 1, 2, 3) (cf. thirteenth exemplary embodiment according to FIG. 21).

Here, the three conductive elements 50, 52, 54 are designed as suitably structured metallic caps, the first metallic cap 50 [<--> phase shift 2 (-Δφ)] being twice as long as the second metallic cap 52 [<--> Phase shift -Δφ] and how the third metallic cap 54 [<--> phase shift -Δφ] is formed.

The comparison of Figure 18 (= tenth embodiment) with Figure 20 (= twelfth embodiment) and a comparison of Figure 19 (= eleventh embodiment) with Figure 21 (= thirteenth)

Embodiment) apparent, respectively opposite arrangement of the elements 40, 42, 44 or 50, 52, 54 influencing the phase shift nΔφ on the individual branches of the feed network or feed network when the beam deflection disappears in FIGS. 18 and 20 or in relation to FIG. 17 The doubled beam deflection in FIGS. 19 and 21 is explained by the fact that the effective dielectric constant e ^ on the feed network and thus the phase shift nΔφ between the antenna elements 32, 34, 36, 38 - is increased by the dielectric materials 40, 42, 44 (cf. 18 and 19), which corresponds to an electrical extension of the planar line system 20, and

is reduced by the conductive materials 50, 52, 54 (cf. FIG. 20 and FIG. 21), which corresponds to an electrical shortening of the planar line system 20. Two variants of the present invention in the form of a meandering feed network, that is to say in the form of a meandering guide of the feed line 20 to influence the phases to a greater extent, and the resulting antenna diagram are shown in FIG. 22 (= fourteenth embodiment of the device 100) and in FIG. 23 (= fifteenth embodiment) of the device 100).

Thus, the electrical path length between the beam (er) elements 32, 34, 36, 38 can be a multiple of half the wavelength by the

Fields of the beam (s) elements 32, 34, 36, 38 are aligned antiparallel to each other (see FIG. 22) or parallel to each other (see FIG. 23), each using an electromagnetic slot coupling as an example from the rear of the H [och] F [ requenz] board 10 takes place.

The following is a detailed theoretical explanation of the structure and the functional principle of the present invention, the beam deflection Θ in elevation E being discussed first:

According to FIG. 7, the beam angle Θ is related to the phase shift Δφ between two antenna or beam (s) elements 32, 34, 36, 38 as follows (cf. SK Koul, B. Bhat, "Microwave and Millimeter Wave Phase Shifters", Volume 1 and Volume 2, Artech House, Boston, London, 1991): Δφ = Δφ ₂ - Δφ _! = (ω / c) a sinΘ

For the propagation coefficient of the lossless line the following applies approximately (for lines for T [ransversal] E [lectro] M [agnetic] waves, that is for lines for electromagnetic waves without

Field proportions in the direction of expansion even apply exactly): \ 1/2 1/2 ß = ω ^* (L-cy = ω (μo εoεβff)

It can thus be found for the ratio Δφ ₂ / Δφi of the phase shift Δφ between two elements as a function of the effective dielectric constant ε ^ for two configurations of the dielectric cap 40 (see FIGS. 11 and 12):

Δφ ₂ / Δ (pι = ß ₂ l / ßι l = / ε _e ff.i) 1/2

This gives the beam deflection Θ to:

Θ = arcsin {Δφ ₁ / [2πa [(ε _{eff, 2} / ε _eff, ι) 1 ^1/2 -1]},

where the distance a of the dielectric element 40 is normalized to the wavelength λ: a '= a / λ.

For a vanishing beam deflection (Θ equal to zero degrees) without influence of the dielectric constant ε _r , _{2 of} the dielectric material 40, the result is between two antenna elements 32, 34 and 34, 36 and 36,

38 a phase difference Δφi = 2ττ.

If a non-vanishing beam deflection (Θ not equal to zero degrees) is realized both upwards and downwards and only "dielectric loading" (<-> provision of at least one dielectric body 40) is to be used, then a phase difference Δφ! <2ττ selected, because "dielectric loading" can only extend a line 20 electrically.

It should also be borne in mind that in addition to the

Propagation coefficient ß also changes the line impedance Z: 1/2 -1/2 Z = (L7C) ~ Eeff-

A certain mismatch is usually tolerable. This mismatch determines the maximum achievable beam deflection Θ unless configurations are found in which the capacitance C and the inductance L 'change in a similar manner.

Such a configuration can be given in a manner essential to the invention by partial or complete metallization of at least one plastic cap which then functions as a metallic element 50 for adjusting the elevation angle ((cf. second exemplary embodiment according to FIGS. 9A, 9B, 9C).

Otherwise, there remains the possibility essential to the invention, the

To increase the length of the line 20 or the phase shift Δφ between two antenna or beam (er) elements 32, 34, 36, 38 to Δφ = n 2π (cf. tenth exemplary embodiment according to FIG. 18 and eleventh exemplary embodiment according to FIG. 19).

As far as achievable changes in the effective dielectric constant εeff of the planar line 20 are concerned, it can generally be stated that the effective dielectric number of a microstrip line (= so-called "microstrip line") generally deviates less than the dielectric constant ε _r , ι of the substrate 10 than this at the dielectric constant one

(symmetrical or asymmetrical) coplanar line (= so-called "coplanar waveguide") or the case of the dielectric constant of a slot line (= so-called "slot line").

Estimates for the effective dielectric constant of such

Piano guides can be found on pages 151 and 176 in RE Collin, "Foundations for Microwave Engineering", 2nd edition, McGraw-Hill International Editions, New York etc., 1992.

For a coplanar line and for a slot line with infinitely thin metallization and with air above the substrate 10, the effective dielectric constant is

where ε _r , ι is the dielectric constant of the substrate 10.

For a microstrip line, the effective dielectric constant ε ^ depends on the thickness h of the substrate 10 and on the width w of the microstrip. In the case of infinitely thin metallization and with air above the substrate 10, the following applies: ε _βff = 0.5 ^ + 1) + 0.5 (ε _r, ι - 1) (1 + 12h / w) ^"1/2 + 0, 02 (ε _rι1 - 1) (1-w / h) ² for w <h; ε _θff = 0.5 ^* (ε _r .ι + 1) + 0.5 (ε _r , ι - 1) (1+ 12h / wV ^{1 2} for w <h.

This means that the effective dielectric constant ε _e ff of the microstrip line is always greater than the effective dielectric constant ε _e ιτ of the coplanar line or the slot line.

From the above equations it follows that "dielectric loading" with a material 40 whose dielectric constant ε _r , _{2 is} equal to the dielectric constant ε _r, ι of the substrate 10 results in an effective maximum

Dielectric constant ε _eff can be achieved, which is equal to the dielectric constant ε _{rι1 of} the substrate 10.

For coplanar lines or slot lines with a dielectric cap 40, whose dielectric constant ε _r , _{2 is} greater than that

Dielectric constant ε _r .ι of the substrate 10 is an effective maximum Dielectric constant £ _< # = 0.5 (ε _r, ι + ε _{r, 2} ) achievable, a second conductive level has to be added for microstrip lines (see FIGS. 10A, 10B, 10C), so that a symmetrical strip line results.

For the microstrip line with "dielectric loading", the effective dielectric constant εe _ff always remains smaller than for the same "dielectric loading" for the coplanar line or slot line.

If, on the other hand, a conductive element 50 is placed over the line 20, the effective dielectric constant εeff can theoretically be reduced to the value one. Exact results can be obtained with simulation programs.

The table below gives some examples.

Line 20 configuration without cap 40 Line 20 with cap 40

Line 20 a 'Z ₀ Δ (pι ε _r , ι _θ f, ι εr.2 εeff, 2 Z Θ

Microstrip 0.64 50 2π 3 2.44 3 3 45 9.8 ° (SRR)

Coplanar 0.64 50 2ττ 3 2 3 3 41 20.6 °

Microstrip 0.5 50 2π 3 2.44 3 3 45 12.6 °

Microstrip 0.5 50 2ττ 3 2.44 4 3.24 38 41.0 °

Coplanar 0.5 50 2π 3 2 3 3 41 26.7 °

Microstrip + 0.5 50 2ττ 3 2.44 11 2.2 ge53 -5.8 ° conductive element 50 estimates

Microstrip + 0.5 50 2ττ 3 2.44 1 2.0 ge55 -10.9 ° conductive element 50 estimates

Microstrip 0.4 50 2π 3 2.44 3 3 45 15.8 °

The microstrip line (S [hort] R [ange] R [adar]; eight millimeters at one

24 gigahertz frequency) refers to a fifty ohm line at a frequency of 24 gigahertz to 10 mil Ro3003. With a jump from fifty ohm line impedance to 41 ohm line impedance, an adjustment of Sn = -20 decibels can be achieved. If a large setting range for the beam deflection Θ is required with a slight mismatch, it is advisable to reduce the distance a of the antenna or beam (s) elements 32, 34, 36, 38 in elevation E.

In order to explain and verify the functional principle of the present invention, various results based on simulations are presented below, the serial feed being considered first:

For the beam (er) element of the S [hort] R [ange] R [adar] sensor (slot-coupled patch), a provisional, non-optimized design for a series feed is calculated; Accordingly, a (simple) feed network or feed network for is in FIG

Simulation calculations are shown, whereby the same power is taken as a basis for all four patches, that is to say the power output on all antenna or beam (er) elements is nominally the same; the distance between the antenna or beam (er) elements is λ _s = 8 millimeters or Aq ^ = 2ττ.

In the design for the series supply, it is important that all connections between the branches to the antenna or beam elements (running vertically in the circuit diagram) are made as long as possible with a similar line impedance (between forty ohms and fifty ohms) be so that the influence of the dielectric and / or conductive cap is as uniform as possible. For this reason, the line to the last element is transformed to the impedance level of 45 ohms (the respective impedance levels are indicated on the lines). This design for series feeding is implemented in a H [igh] F [requency] S [structure] S [imulator] model as part of a finite element simulation program for electromagnetic waves in three-dimensional structures, using slot-coupled patch elements.

This HFSS simulation model for four slot-coupled, series-fed patches is shown in FIG. 25, which also contains the Ra [dar] dom [e] and adhesive for the Ra [dar] dom [e]. A separate simulation calculation is carried out for the position of the reference planes at the branches of the branch lines to the patches; all branch lines are extended accordingly by 350 micrometers.

The simulation calculations of the influence of the dielectric and / or metallized cap are carried out in two configurations:

- The entire space below the feed or dining network is filled with a dielectric; There is air in the level of the metallization of twenty micrometers thick, ie in the space next to the conductor tracks (cf. FIG. 25);

- A cap, which is gradually brought up to the conductor track in the area of the distribution network, is attached below the feed network (see FIG. 26, in which the HFSS simulation model, namely only lines, coupling slots and cap, for simulation calculations on the influence of a metallic cap is shown).

FIG. 27 shows a three-dimensional plot of the directivity measured in decibels in elevation of the arrangement with a simple feed or feed network without a dielectric and / or conductive cap in one

Frequency of 24 gigahertz. FIG. 28 shows the directivity plotted against the beam deflection angle measured in degrees (from the z-axis) and measured in decibels in elevation of the arrangement with a simple feed or feed network without a dielectric and / or conductive cap. Due to the serial feed, the beam angle is frequency-dependent, the different frequencies 22 gigahertz, 24 gigahertz, 26 gigahertz and 28 gigahertz being considered.

In Figure 29 are against the beam deflection angle measured in degrees

(from z-axis) applied directivities measured in decibels in elevation of the arrangement with simple feed or feed network at a frequency of 24 gigahertz compiled for the following different configurations: - array without dielectric and / or metallized cap;

- completely covering dielectric cap with dielectric constant ε = 3, lying directly on the conductor tracks (cf. FIG. 25);

- completely covering dielectric cap with dielectric constant ε = 4.2, lying directly on the conductor tracks (cf. FIG. 25); and - metallic cap at a distance of one hundred micrometers from the conductor tracks (cf. FIG. 26), which is gradually brought up to the distribution network in the edge region.

In this context, there is a swivel range of approximately + ten degrees, as already explained above.

After the functional principle of the present invention in the case of a general series feed has been explained and verified above on the basis of simulation results, various results, partly based on simulations, for the

Considering the case of a meandering serial feed: Figure 30 shows a meandering feed or feed network analogous to Figure 22 (= fourteenth embodiment of the device 100) and to Figure 23 (= fifteenth embodiment of the device 100) that the antenna or beam (er) elements or patches with an electrical

Path length of Δφi = 4ττ (corresponding to 2λ _s , i.e. twice the wavelength of the substrate) connects in order to achieve the greatest possible deflection of the beam lobe.

Furthermore, the distance of the antenna or beam (er) elements or

Patches reduced to six millimeters (corresponding to 0.5 λ at 25 gigahertz), which further increases the deflection of the beam lobe. The feed or feed network according to FIG. 30 generates an output distribution of 0.25 / 1/1 / 0.25 with an amplitude assignment of 0.5 / 1/1 / 0.5. This reduces the side lobes to about -20 decibels below the main lobe maximum; the main lobe also widens.

FIG. 31 shows the directivity plotted against the beam deflection angle measured in degrees (from the z axis) and measured in decibels in elevation of the arrangement with a meandering feed or feed network without a dielectric and / or conductive cap, the different frequencies being 22 gigahertz, 24 gigahertz, 26 gigahertz and 28 gigahertz can be considered.

Due to the longer line length between the patches compared to FIG. 28, the frequency dependence of the beam angle becomes stronger. The spacing of the antenna or beam (er) elements or patches of six millimeters corresponds to half a free space wavelength of 26 gigahertz. Higher frequencies are not included in Figure 31 because

"grating lobes" occur. In FIG. 32, the directivities, measured in decibels, and measured in decibels in elevation of the arrangement with a meandering feed or feed network at a frequency of 24 gigahertz at a frequency of 24 gigahertz are compiled for the following different configurations against the beam deflection angle (from z-axis).

- array without dielectric and / or metallized cap;

- dielectric cap with dielectric constant ε = 2;

- dielectric cap with dielectric constant ε = 3; - metallic cap at a distance of two hundred micrometers from the conductor tracks; and

- metallic cap at a distance of four hundred micrometers from the conductor tracks.

In this context, a metallic cap at a short distance deteriorates the beam shape, so that a metallic cap at a distance of two hundred micrometers can achieve a beam deflection of -7 degrees.

For larger distractions, the meandering dining network would have to be specially designed for the use of a metallic cap. In contrast, a dielectric cap in this arrangement causes a very strong beam deflection, which is already greater than fifteen degrees for a dielectric constant ε = 2 and reaches an angle of thirty degrees for a dielectric constant ε = 3.

In FIG. 33, directivities, measured in decibels, are plotted against the beam deflection angle measured in degrees (from the z axis) in elevation of the arrangement with a meandering feed or feed network in a frequency range from 24 gigahertz to 26

Gigahertz for the following different configurations compiled:

- Array without dielectric and / or metallized cap at a frequency of 24 gigahertz;

- array without dielectric and / or metallized cap at a frequency of 25 gigahertz;

- Array without dielectric and / or metallized cap at a frequency of 26 gigahertz;

- dielectric cap with dielectric constant ε = 2 at a frequency of 24 gigahertz; - dielectric cap with dielectric constant ε = 2 at a frequency of 25 gigahertz;

- dielectric cap with dielectric constant ε = 2 at a frequency of 26 gigahertz;

- dielectric cap with dielectric constant ε = 3 at a frequency of 24 gigahertz;

- dielectric cap with dielectric constant ε = 3 at a frequency of 25 gigahertz; and

- dielectric cap with dielectric constant ε = 3 at a frequency of 26 gigahertz.

In this context, the frequency-dependent angle difference of the beam maxima decreases for large beam deflections, but remains very large there as well.

While the two arrangements outlined above

(simple feed or feed network according to FIGS. 24 to 29; meandering feed or feed network according to FIGS. 30 to 33) for this reason primarily for narrow band applications, such as for a long-range radar (so-called L [ong] R [ange ] R [adar], typically for one at a frequency of 77

Gigahertz working, regulating the distance, that is suitable for an A [daptive] C [ruise] C [ontrol] system) with a planar antenna, in FIGS. 24, 25 and 26 an in-phase feed network with a binary graded phase difference analogous to FIG. 17 and FIG. 18 (= tenth embodiment of the device 100), to Figure 19 (= eleventh embodiment of the

Device 100), to Figure 20 (= twelfth embodiment of the device 100) and to Figure 21 (= thirteenth embodiment of the device 100) examined.

Figure 34 shows an in-phase feed network, all

In principle, antenna elements are in phase, that is, they are supplied with a vanishing phase shift (Δφ! = 0).

In order to obtain a large beam deflection, the line lengths from the patches to the first branch are approximately eight millimeters, that is to say the line lengths from the patches to the first branch correspond to approximately λ _s . The line length between the first branch and the second branch is about ten millimeters to about twelve millimeters.

A phase shift of 35 degrees between the antenna or beam elements is inserted into the in-phase feed network, which can just be compensated for on the above-mentioned line lengths by a dielectric cap with a dielectric constant ε = 3 (analogously to FIG. 18).

This results in a beam deflection of ten degrees without a dielectric and / or conductive cap. The amplitude assignment is again 0.5 / 1/1 / 0.5 (see FIG. 30), the distance between the antenna or beam (er) elements is 5.4 millimeters to one another. The areas below the conductor tracks are the areas of the dielectric cap which are used for the beam deflection "forwards" or "forwards" and "backwards" or "backwards".

In Figure 35 are against the beam deflection angle measured in degrees

(from z-axis) applied directivities measured in decibels in elevation of the arrangement with in-phase feed or feed network at a frequency of 24 gigahertz compiled for the following different configurations: - array without dielectric and / or metallized cap (beam deflection in the design);

- dielectric cap with dielectric constant ε = 3 (beam deflection "forward"); and

- dielectric cap with dielectric constant ε = 3 (beam deflection "backwards").

In this context, it is possible to exactly compensate the beam deflection of approximately ten degrees, which is predetermined by the line lengths, by means of a first dielectric cap; on the other hand, a second dielectric cap shaped differently from the first dielectric cap can more than double the beam deflection.

In FIG. 36, the directivities, measured in decibels, and measured in decibels in elevation of the arrangement with in-phase feed or feed network in a frequency range from twenty gigahertz to 28 gigahertz for the following different configurations are compiled against the beam deflection angle (from z-axis).

- Array without dielectric and / or metallized cap at a frequency of twenty gigahertz; - Array without dielectric and / or metallized cap at a frequency of 22 gigahertz; - Array without dielectric and / or metallized cap at a frequency of 24 gigahertz;

- Array without dielectric and / or metallized cap at a frequency of 26 gigahertz; - Array without dielectric and / or metallized cap at a frequency of 28 gigahertz;

- dielectric cap in the "swivel forward" area with dielectric constant ε = 3 at a frequency of twenty gigahertz;

- dielectric cap in the "swivel forward" area with dielectric constant ε = 3 at a frequency of 22 gigahertz;

- dielectric cap in the "swivel forward" area with dielectric constant ε = 3 at a frequency of 24 gigahertz;

- dielectric cap in the "swivel forward" area with dielectric constant ε = 3 at a frequency of 26 gigahertz; - dielectric cap in the area "swivel forward" with dielectric constant ε = 3 at a frequency of 28 gigahertz;

- dielectric cap in the "swivel back" area with dielectric constant ε = 3 at a frequency of twenty gigahertz;

- dielectric cap in the "swivel back" area with dielectric constant ε = 3 at a frequency of 22 gigahertz;

- dielectric cap in the "swivel back" area with dielectric constant ε = 3 at a frequency of 24 gigahertz;

- dielectric cap in the "swivel back" area with dielectric constant ε = 3 at a frequency of 26 gigahertz; and - dielectric cap in the "swiveling back" area with dielectric constant ε = 3 at a frequency of 28 gigahertz.

In this context, the arrangement without a dielectric and / or metallized cap and for the dielectric cap which compensates for the beam deflection (area "swiveling back") results in a relatively small variation in the beam lobe maximum with the Frequency.

The variation of the maximum with the frequency when swiveling "forward" is relatively small, but - like the side lobes - can still be optimized; such optimization includes in particular

Shape and placement of the dielectric and / or conductive caps,

- Phase errors on the antenna or beam (er) elements and

- The distance of the antenna or beam (er) elements to each other.

In summary, the above illustrate

Exemplary embodiments of three different feed networks (simple feed or feed network according to FIGS. 24 to 29; meandering feed or feed network according to FIGS. 30 to 33; in-phase feed network with a binary graded phase difference according to FIGS. 34 to 36) the potential of the latter

Invention proposed setting of the elevation angle of a planar radar antenna.

A column of four slot-coupled patches at a frequency of 24 gigahertz is used for the simulation calculations, these patches being available as optimized antenna or beam (s) elements for the simulation. The limitation to four antenna or beam (er) elements keeps the effort for the simulation within limits.

However, the beam lobe of this column is so wide that there is only a difference in the directivities of a few decibels in the swivel range, so that the effort - not least because of the additional losses due to the swivel - would not be worthwhile for this configuration; however, these simulations do

Effect of beam swiveling clearly. Furthermore, a better one Suppression of the side lobes and the "grating lobes" can be realized by an optimized design of the dining network.

If planar antennas are used in the medium range and for L [ong] R [ange] R [adar] applications, columns with about twenty antenna or beam (er) elements must be used in order to be able to achieve the necessary antenna gains at all. The beam is then only a few degrees wide, and an installation of about five degrees to about ten degrees from the plumb line can no longer be tolerated under any circumstances.

The simple serial feed has the greatest relevance for a narrowband L [ong] R [ange] R [adar]. Here, however, the angular range that can be achieved by "dielectric loading" is limited. Remedy can be found through the use

- of materials with a higher dielectric constant,

- of alternative line types, for example coplanar lines, or

- Be created by so-called "slow wave" structures and / or by so-called P [hotonic] B [and] G [ap] structures in the dielectric or conductive, in particular cap-shaped body.

In the exemplary embodiment of the in-phase feed network (see FIGS. 34 to 36), the potential for broadband systems and for a large swivel range is also shown, although the

Dining networks become quite complex and large.

With regard to the detectability of the present invention on

This product is verified by opening and comparing two radar sensors for different installation angles, for example from two different motor vehicles. If the boards, on to which the feed network and the antennas are located, are identical and if the dielectric or conductive, in particular cap-shaped bodies differ, then the proof is completed.

In the event that the conductor tracks and / or the antenna or

Beam (er) element θ are provided with an opaque coating (in this case it is not visible whether the boards are identical or not), the coating has to be removed, for example by means of a solvent, or X-ray images of H [och] F to produce [requenz] boards.

If the dielectric or metallized, in particular cap-shaped bodies, for example as a result of painting, look identical and also have identical dimensions, then the dielectric constant of the dielectric or metallized, in particular cap-shaped bodies must be determined; there are suitable measuring techniques for this.

Claims

Expectations

1. Device (100) for emitting and / or receiving electromagnetic radiation, in particular electromagnetic H [och] F [requenz] radar radiation, comprising at least one single-layer or multilayer, in particular also at least one metallic layer, substrate (10) , on which at least one planar line (20), in particular in the form of a ribbon line or in the form of a symmetrical or asymmetrical coplanar line (20k) or in the form of a microstrip line (20m) or in the form of a slot line (20s) or in the form of a coplanar two-band line , with at least two antenna elements (32, 34, 36, 38), in particular beam elements, whose in particular at least partially serial feed (22s) and / or in particular at least partially in phase feed (22g) and / or in particular at least partially phase and / or amplitude-symmetrical feed (22p), for example by means of direct or capacitive coupling one of at least one feed network on the upper side (10o) of the substrate (10) facing the antenna elements (32, 34, 36, 38) or - by means of at least one slot (32s, 34s, 36s, 38s), which is at least one electromagnetic network, at least one feed network from the underside (10u) of the substrate (10) facing away from the antenna elements (32, 34, 36, 38) or via at least one electrical feedthrough (32d, 34d, 36d, 38d) from the one from the antenna elements (32, 34 , 36, 38) facing away from the bottom (10u) of the substrate (10) takes place, whereby at least one metallization layer (12) can be arranged on the underside (10u) of the substrate (10) facing away from the antenna elements (32, 34, 36, 38), so that the phase shift (Δφ) between that of various antenna elements (32, 34, 36, 38) emitted and / or received electromagnetic radiation or the angle (Θ), in particular the elevation angle, the radiation and / or reception of the electromagnetic radiation in elevation (E) by varying the effective dielectric constant ( ε _eff ), in particular the propagation coefficient, the line (20) and / or by arranging at least one at least partially made of conductive material, in particular at least partially, at a distance from the line (20) and / or the antenna elements (32, 34, 36, 38) partially made of metal, formed elements (50, 52, 54) is adjustable.

2. Device according to claim 1, characterized in that the effective dielectric constant (ε _eff ) of the line (20) and thus the phase shift (Δφ) between the antenna elements (32, 34, 36, 38) through to the line (20) and / or to the antenna elements (32, 34, 36, 38) variably spaced arrangement of dielectric, in particular cap-shaped material (40, 42, 44) on the top (10o) of the substrate facing the antenna elements (32, 34, 36, 38) (10), in particular above the line (20) with air between the dielectric material (40, 42, 44) and the line (20), and / or on the underside (10u) of the substrate (10) facing away from the antenna elements (32, 34, 36, 38), in particular below the line (20) with air between the dielectric material (40, 42, 44) and the line (20), can be varied, in particular enlarged.

3. Device according to claim 1 or 2, characterized in that the effective dielectric constant (ε _eff ) of the line (20) and thus the phase shift (Δφ) between the antenna elements (32, 34, 36, 38) through to the line (20) and / or to the antenna elements (32, 34, 36, 38) at a variable spacing from the arrangement of the particularly cap-shaped conductive element (50, 52, 54), for example in the form of a partially or completely metallized plastic cap, on the antenna elements (32, 34 , 36, 38) facing top (10o) of the substrate (10), in particular above the line (20) with air between the conductive element (50, 52, 54) and the line (20), and / or - on the of The underside (10u) of the substrate (10) facing away from the antenna elements (32, 34, 36, 38), in particular below the line (20) with air between the conductive element (50, 52, 54) and the line (20), can be varied , in particular can be reduced.

4. The device according to at least one of claims 1 to 3, characterized in that the dielectric material (40) has at least one conductive layer (50s) and / or - that the conductive element (50) has at least one dielectric layer (40s).

5. The device according to at least one of claims 1 to 4, characterized in that the type designation of the device (100), - the type designation of the means of transport (200), in particular the motor vehicle for which the device (100) is provided, the elevation angle ( Θ) of the device (100) and / or the installation location in the means of transportation (200) for which the device (100) is provided with the selected elevation angle (Θ) on the dielectric material (40, 42, 44) and / or on the conductive element (50, 52, 54) is noted, in particular stamped.

6. The device according to at least one of claims 1 to 5, characterized in that the phase shift (Δφ) between the antenna elements (32, 34, 36, 38) and / or the angle (Θ), in particular the elevation angle, the radiation and / or receiving the electromagnetic radiation in elevation (E) by varying the distance of the metal-coated (50s) or dielectric material (40, 42, 44) from the feed network, by means of the dielectric constant (ε _r , ₂ ) of the metal-coated (50s) or dielectric material ( 40, 42, 44) and / or by means of a, in particular periodic, structuring of the metal-coated (50s) or dielectric material (40, 42, 44) depending on the elevation angle (period), for example through holes, through grooves, through columns, through steps , by honeycombs, by a P [hotonic] B [and] G [ap] structure to form a "slow wave" structure or by the like, is exactly adjustable, in particular can be enlarged exactly.

7. The device according to at least one of claims 1 to 6, characterized in that the dielectric material (40, 42, 44) and / or the conductive element (50, 52, 54) have the same or similar thermal expansion coefficient as the material of the particular substrate (10) designed as a high [F] requence circuit board.

8. The device according to at least one of claims 1 to 7, characterized in that the dielectric material (40, 42, 44) and / or the conductive element (50, 52, 54) in direct contact, in particular via point contact areas, or - is connected via at least one spacer with the optional use of clamps and / or screws or by means of selective or full-area gluing in connection with the substrate (10), which is designed in particular as a H [och] F [requenz] board.

9. The device according to at least one of claims 1 to 8, characterized in that the dielectric material (40, 42, 44) and / or the conductive element (50, 52, 54) is constructed in one part or in several parts, for example by - at least a dielectric and / or conductive element influencing the phase shift (Δφ) and / or the elevation angle (Θ) is arranged above the feed network and / or under the feed network and at least one further influencing the phase shift (Δφ) and / or the elevation angle (Θ) dielectric and / or conductive element against the device (100) in particular Protects environmental influences.

10. The device according to claim 9, characterized in that the (further) dielectric and / or conductive element in at least one recess of the dielectric material (40, 42, 44) and / or the conductive element (50, 52, 54) can be used and can be mounted together with the dielectric material (40, 42, 44) and / or with the conductive element (50, 52, 54) above the feed network and / or below the feed network.

11. The device according to at least one of claims 1 to 10, characterized in that the distance of the dielectric material (40, 42, 44) and / or the conductive element (50, 52, 54) to the line (20) in the metallized transition region in particular (40, 42, 44t and 50t) from the areas (40b and 50b) influencing the phase shift (Δφ) and / or the elevation angle (Θ) to those not influencing the phase shift (Δφ) and / or the elevation angle (Θ) , in particular non-metallized areas (40n or 50n) increases gradually, for example step-like, and / or continuously, for example linearly trapezoidal.

12. The device according to at least one of claims 1 to 11, characterized in that in the case of an at least partially phase and / or amplitude symmetrical feed (22p) on one side of the central feed of the feed network, the phase shift (Δφ) and / or the elevation angle ( Θ) can be enlarged by means of the dielectric material (40, 42, 44) and on the other side of the central feed of the feed network the phase shift (Δφ) and / or the elevation angle (Θ) can be reduced by means of the conductive element (50, 52, 54).

13. The device according to at least one of claims 1 to 12, characterized in that for influencing the phase shift (Δφ) and / or the elevation angle (Θ) to a greater extent, the feed network in another type, for example in the form of a coplanar line (20k) or in In the form of a slot line (20s) or in the form of a coplanar two-band line, it is designed as the line (20) itself, which is, for example, in the form of a microstrip line (20m).

14. The device according to at least one of claims 1 to 13, characterized in that in particular in the case of particularly broadband radar systems or U [ltra] W [ide] B [and] radar systems for setting a certain beam deflection (Θ) in the feed network at least one Binary graded phase shift element (60, 62, 64) is installed, which can be compensated by means of the dielectric material (40, 42, 44) and / or by means of the conductive element (50, 52, 54) so that the deflection of the beam lobe can be reduced , or can be amplified so that the deflection of the beam can be increased.

15. The device according to at least one of claims 1 to 14, characterized in that to influence the phase shift (Δφ) and / or the elevation angle (Θ) to a greater extent, the line (20), in particular the feed line, is meandering, so that the electromagnetic fields of the antenna elements (32, 34, 36, 38) are aligned anti-parallel to one another or parallel to one another and / or the electrical path length between the antenna elements (32, 34, 36, 38) is a multiple of half the wavelength of the emitted and / or received electromagnetic radiation.

16. The device according to at least one of claims 1 to 15, characterized in that the dielectric material (40, 42, 44) and / or the conductive element (50, 52, 54) is in particular adjustable via at least one electric motor to the to keep emitted and / or received electromagnetic radiation in elevation (E) at the same angle (Θ), in particular at the same elevation angle, even with different loads on the means of transportation (200).

17. The device according to at least one of claims 1 to 16, characterized by at least one coding element, in particular accessible from outside the device (100), such as at least one jumper and / or at least one switch, for communicating and storing the installation position of the device (100). ,

18. Method for emitting and / or receiving electromagnetic radiation, in particular electromagnetic high-frequency radar radiation, by means of at least two antenna elements (32, 34, 36, 38), in particular beam elements, which have at least one planar trained line (20), in particular in the form of a ribbon line or in the form of a symmetrical or asymmetrical Coplanar line (20k) or in the form of a microstrip line (20m) or in the form of a slot line (20s) or in the form of a coplanar two-band line, in particular at least partially in series (22s) and / or in particular at least partially in phase (22g) and / or in particular at least partially phase and / or amplitude symmetrical (22p), for example by means of direct or capacitive coupling to at least one feed network on the upper side (10o) facing the antenna elements (32, 34, 36, 38) of at least one single-layer or multi-layer, in particular also at least one metallic layer, comprising substrate (10) or by means of at least one feed network from at least one single-layer underside (1 Ou) facing away from the antenna elements (32, 34, 36, 38) by at least one slot (32s, 34s, 36s, 38s) or multilayer, in particular also having at least one metallic layer n substrate (10) or via at least one electrical feedthrough (32d, 34d, 36d, 38d) from the underside (10u) facing away from the antenna elements (32, 34, 36, 38) of at least one single-layer or multi-layer, in particular also at least a substrate (10) having a metallic layer is fed, at least one on the underside (10u) of the substrate (10) facing away from the antenna elements (32, 34, 36, 38)

Metallization layer (12) can be provided

characterized ,

that the phase shift (Δφ) between that radiated by different antenna elements (32, 34, 36, 38) and / or received electromagnetic radiation or the angle (Θ), in particular the elevation angle, the radiation and / or reception of the electromagnetic radiation in elevation (E) is set, - by the effective dielectric constant (ε _eff ), in particular the coefficient of propagation, which Line (20) is varied and / or by variably spacing at least one element (50, 52, 54) formed at least partially from conductive material, in particular at least partially from metal, from line (20) and / or antenna elements (32, 34 , 36, 38) is arranged.

19. The method according to claim 18, characterized in that the effective dielectric constant (ε _e ff) of the line (20) and thus the phase shift (Δφ) between the antenna elements (32, 34, 36, 38) varies, in particular is increased, by dielectric, in particular cap-shaped material (40, 42, 44) on the top (10o) of the substrate (10) facing the antenna elements (32, 34, 36, 38), in particular above the line (20) with air between the dielectric Material (40, 42, 44) and the line (20), and / or on the underside (10u) of the substrate (10) facing away from the antenna elements (32, 34, 36, 38), in particular below the line (20) with air between the dielectric material (40, 42, 44) and the line (20), variably spaced from the line (20) and / or to the antenna elements (32, 34, 36, 38).

20. The method according to claim 18 or 19, characterized in that the effective dielectric constant (ε _eff ) of the line (20) and thus the phase shift (Δφ) between the antenna elements (32, 34, 36, 38) is varied, in particular reduced, by the conductive element (50, 52, 54), which is in particular cap-shaped, for example in the form of a partially or completely metallized plastic cap, on which the antenna elements (32, 34, 36, 38) facing upper side (10o) of the substrate (10), in particular above the line (20) with air between the conductive element (50, 52, 54) and the line (20), and / or on that of the antenna elements (32, 34, 36, 38) facing away from the bottom (10u) of the substrate (10), in particular below the line (20) with air between the conductive element (50, 52, 54) and the line (20), variably spaced from Line (20) and / or to the antenna elements (32, 34, 36, 38) is arranged.

21. The method according to at least one of claims 18 to 20, characterized in that the phase shift (Δφ) between the antenna elements (32, 34, 36, 38) and / or the angle (Θ), in particular the elevation angle, the radiation and / or reception of the electromagnetic radiation in elevation (E) is precisely adjusted, in particular precisely increased, by varying the distance of the metal-coated (50s) or dielectric material (40, 42, 44) from the feed network, by changing the dielectric constant (ε _r , 2 ) the metal-coated (50s) or dielectric material (40, 42, 44) is selected and / or by the metal-coated (50s) or dielectric material (40, 42, 44), for example through holes, through grooves, through columns, through steps, through honeycombs, through a P [hotonic] B [and] G [ap] structure with a "slow wave" structure or the like, depending on the elevation angle (Θ) in particular periodically structured.

22. The method according to at least one of claims 18 to 21, characterized in that in the case of an at least partially phase and / or amplitude symmetrical feed (22p) on one side of the central feed of the feed network, the phase shift (Δφ) and / or the elevation angle ( Θ) is increased by means of the dielectric material (40, 42, 44) and the phase shift (Δφ) and / or the elevation angle (Θ) is reduced by means of the conductive element (50, 52, 54) on the other side of the central feed of the feed network.

23. The method according to at least one of claims 18 to 22, characterized in that, in particular in the case of particularly broadband radar systems or U [ltra] W [ide] B [and] radar systems, a specific beam deflection (Θ) is set by entering the feed network at least one binary graded phase shift element (60, 62, 64) is installed, which is compensated for by means of the dielectric material (40, 42, 44) and / or by means of the conductive element (50, 52, 54) such that the deflection of the beam lobe is reduced, or is increased so that the deflection of the beam is increased.

24. The method according to at least one of claims 18 to 23, characterized in that the dielectric material (40, 42, 44) and / or the conductive element (50, 52, 54) in particular over at least one electric motor is adjusted in order to keep the emitted and / or received electromagnetic radiation in elevation (E) at the same angle (Θ), in particular at the same elevation angle, even with different loads on the means of transportation (200).

25. Use of at least one device (100) according to at least one of claims 1 to 17 and / or a method according to at least one of claims 18 to 24 for sensing, in particular for radar sensing, the surrounding area of a means of transportation (200), in particular a motor vehicle, for example for the purpose of object-specific measurement of the distance and / or the speed of at least one object in the vicinity of the means of transportation (200), the automatic adjustment of the distance and / or the speed of the means of transportation (200), the stop-and-go operation of the means of transportation (200), increasing the safety when operating the means of transportation (200) with regard to sharpening the airbag and / or belt tensioner, optimizing the triggering time of the airbag and / or belt tensioner or warning of and avoiding a collision, for example with a other means of transportation (202).