Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
  • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
  • H04L27/38—Demodulator circuits; Receiver circuits
  • H04L27/3845—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier
  • H04L27/3854—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier using a non - coherent carrier, including systems with baseband correction for phase or frequency offset
  • H04L27/3863—Compensation for quadrature error in the received signal
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/06—Receivers
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L25/00—Baseband systems
  • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
  • H04L25/0202—Channel estimation
  • H04L25/022—Channel estimation of frequency response
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2647—Arrangements specific to the receiver only

Definitions

• This invention relates generally to communication channel estimation and, more particularly, to systems and methods for improving the use of quadrature modulation unbiased training sequences in the training of receiver channel estimates, by removing quadrature imbalance errors.
• FIG. 1 is a schematic block diagram of a conventional receiver front end (prior art).
• a conventional wireless communications receiver includes an antenna that converts a radiated signal into a conducted signal. After some initial filtering, the conducted signal is amplified. Given a sufficient power level, the carrier frequency of the signal may be converted by mixing the signal (down-converting) with a local oscillator signal. Since the received signal is quadrature modulated, the signal is demodulated through separate I and Q paths before being combined. After frequency conversion, the analog signal may be converted to a digital signal, using an analog-to- digital converter (ADC), for baseband processing. The processing may include a fast Fourier transform (FFT).
• FFT fast Fourier transform
• a unique direction in the constellation (e.g., the I path) is stimulated, while the other direction (e.g., the Q path) is not.
• the same type of unidirectional training may also be used with pilot tones. Note: scrambling a single modulation channel (e.g., the I channel) with ⁇ 1 symbol values does not rotate the constellation point, and provides no stimulation for the quadrature channel.
• FIG. 2 is a schematic diagram illustrating quadrature imbalance at the receiver side (prior art). Although not shown, transmitter side imbalance is analogous.
• the Q path is the reference.
• the impinging waveform is cos(wt + ⁇ ), where ⁇ is the phase of the channel.
• the Q path is down-converted with — sin(wt).
• the I path is down-converted with (l+2 ⁇ )cos(wt+ 2 ⁇ ).
• 2 ⁇ and 2 ⁇ are hardware imbalances, respectively a phase error and an amplitude error.
• the low pass filters Hi and HQ are different for each path.
• the filters introduce additional amplitude and phase distortion. However, these additional distortions are lumped inside 2 ⁇ and 2 ⁇ . Note: these two filters are real and affect both +w and —w in an identical manner. [0011] Assuming the errors are small:
• the first component on the right hand side, cos(wt), is the ideal I path slightly scaled.
• the second component, - 2 ⁇ .sin(wt) is a small leakage from the Q path.
• the errors result in the misinterpretation of symbol positions in the quadrature modulation constellation, which in turn, results in incorrectly demodulated data.
• Wireless communication receivers are prone to errors caused by a lack of tolerance in the hardware components associated with mixers, amplifiers, and filters. In quadrature demodulators, these errors can also lead to imbalance between the I and Q paths, resulting in improperly processed data.
• a training signal can be used to calibrate receiver channels.
• a training signal that does not stimulate both the I and Q paths does not address the issue of imbalance between the two paths.
• An unbiased training signal can be used to stimulate both the I and Q paths, which results in a better channel estimate.
• channel estimates are derived from predetermined information associated with the positive or reference (+f) subcarriers. Better channel estimates can be obtained if the negative or mirror (-f) subcarriers are also used.
• a frequency smoothing can be applied to a training signal that acts as a means of removing channel bias errors.
• a method for supplying a frequency- smoothed communications training signal.
• the method generates a frequency-smoothed unbiased training signal in a quadrature modulation transmitter.
• the frequency-smoothed unbiased training signal includes a plurality of pilot signal products, where each pilot signal product includes complex plane information represented by a reference frequency subcarrier, multiplying complex plane information represented by mirror frequency subcarrier. The sum of the plurality of pilot signal products is equal to zero.
• the method supplies the frequency- smoothed unbiased training signal so that it may be transmitted within a single symbol period.
• the frequency-smoothed unbiased training signal includes a plurality of adjacent reference frequency subcarriers and a plurality of adjacent mirror frequency subcarriers.
• the frequency- smoothed unbiased training signal may include a group of adjacent reference frequency subcarriers, without intervening subcarriers, and a plurality of adjacent mirror frequency subcarriers, without intervening subcarriers.
• the frequency- smoothed unbiased training signal may be represented as follows:
• a method for calculating a channel estimate using a frequency-smoothed unbiased training signal accepts a frequency-smoothed unbiased training sequence in a quadrature demodulation receiver.
• the frequency- smoothed unbiased training sequence includes a plurality of a plurality of pilot signal products, wherein each pilot signal product includes predetermined complex plane information (p) represented by a reference frequency subcarrier (f), multiplying predetermined complex plane information (p m ) represented by mirror frequency subcarrier (-f).
• the sum of the plurality of pilot signal products is equal to zero.
• the method processes the frequency- smoothed unbiased training signal, generating a plurality of processed symbols (y) representing complex plane information. Each processed symbol (y) is multiplied by a conjugate of a corresponding reference signal (p*), and a frequency- smoothed channel estimate (h) is obtained.
• FIG. 1 is a schematic block diagram of a conventional receiver front end (prior art).
• FIG. 2 is a schematic diagram illustrating quadrature imbalance at the receiver side (prior art).
• FIG. 3 is a schematic block diagram depicting an exemplary data transmission system.
• FIG. 4 is a schematic block diagram of a system or device for supplying a frequency- smoothed unbiased training signal.
• FIG. 5 is a diagram depicting a simple example of a frequency- smoothed unbiased training signal.
• FIG. 6 is a diagram depicting a second example of frequency- smoothed unbiased training signal.
• FIG. 7 is a diagram depicting an unbiased training signal enabled as a group of pilot symbols accompanying communication symbols.
• FIG. 8 is a diagram depicting a frequency-smoothed unbiased training signal enabled as a preamble preceding non-predetermined communication data.
• FIG. 9 is a schematic block diagram of a system or device for calculating a channel estimate using a frequency-smoothed unbiased training signal.
• FIG. 10 depicts the performance achieved by applying unbiased training signal algorithms to the WiMedia UWB standard.
• FIG. 11 is a flowchart illustrating a method for supplying a frequency- smoothed communications training signal.
• FIG. 12 is a flowchart illustrating a method for calculating a channel estimate using a frequency- smoothed unbiased training signal.
• a component may be, but is not limited to being, a process running on a processor, generation, a processor, an object, an executable, a thread of execution, a program, and/or a computer.
• an application running on a computing device and the computing device can be a component.
• One or more components can reside within a process and/or thread of execution and a component may be localized on one computer and/or distributed between two or more computers.
• these components can execute from various computer readable media having various data structures stored thereon.
• the components may communicate by way of local and/or remote processes such as in accordance with a signal having one or more data packets (e.g., data from one component interacting with another component in a local system, distributed system, and/or across a network such as the Internet with other systems by way of the signal).
• a signal having one or more data packets (e.g., data from one component interacting with another component in a local system, distributed system, and/or across a network such as the Internet with other systems by way of the signal).
• data packets e.g., data from one component interacting with another component in a local system, distributed system, and/or across a network such as the Internet with other systems by way of the signal.
• DSP digital signal processor
• ASIC application specific integrated circuit
• FPGA field programmable gate array
• a general-purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine.
• a processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.
• a software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art.
• a storage medium may be coupled to the processor such that the processor can read information from, and write information to, the storage medium.
• the storage medium may be integral to the processor.
• the processor and the storage medium may reside in an ASIC.
• the ASIC may reside in the node, or elsewhere.
• the processor and the storage medium may reside as discrete components in the node, or elsewhere in an access network.
• FIG. 3 is a schematic block diagram depicting an exemplary data transmission system 300.
• a baseband processor 302 has an input on line 304 to accept digital information form the Media Access Control (MAC) level.
• the baseband processor 302 includes an encoder 306 having an input on line 304 to accept digital (MAC) information and an output on line 308 to supply encoded digital information in the frequency domain.
• An interleaver 310 may be used to interleave the encoded digital information, supplying interleaved information in the frequency domain on line 312.
• the interleaver 310 is a device that converts the single high speed input signal into a plurality of parallel lower rate streams, where each lower rate stream is associated with a particular subcarrier.
• An inverse fast Fourier transform (IFFT) 314 accepts information in the frequency domain, performs an IFFT operation on the input information, and supplies a digital time domain signal on line 316.
• a digital-to-analog converter 318 converts the digital signal on line 316 to an analog baseband signal on line 320.
• a transmitter 322 modulates the baseband signal, and supplies a modulated carrier signal as an output on line 324.
• alternate circuitry configurations capable of performing the same functions as described above would be known by those with skill in the art.
• a receiver system would be composed of a similar set of components for reverse processing information accepted from a transmitter. [0038] FIG.
• the system 400 comprises a transmitter or signal generator means 402 having an input on line 404 to accept training information, typically in digital form. For example, the information may be supplied from the MAC level.
• the transmitter 402 has an output on line 406 to supply a quadrature modulated frequency- smoothed unbiased training signal.
• the transmitter 402 may include a transmitter subsystem 407, such as a radio frequency (RF) transmitter subsystem that uses an antenna 408 to communicate via an air or vacuum media.
• RF radio frequency
• the transmitter subsystem 407 includes an in-phase (I) modulation path 410, or a means for generating I modulation training information.
• the transmitter subsystem 407 also includes a quadrature (Q) modulation path 412, or a means for generating Q modulation training information.
• I path information on line 404a is upconverted at mixer 414 with carrier fc, while Q path information on line 404b is upconverted at mixer 416 with a phase shifted version of the carrier (fc + 90°).
• the I path 410 and Q path 412 are summed at combiner 418 and supplied on line 420.
• the signal is amplified at amplifier 422 and supplied to antenna 408 on line 406, where the frequency-smoothed unbiased training signals are radiated.
• the I and Q paths may alternately be referred to as I and Q channels.
• a frequency-smoothed unbiased training signal may also be referred to as a frequency-balanced training sequence, and is part of a larger class of balanced or unbiased training signals described in parent applications, and in detail below.
• the frequency-smoothed unbiased training signal includes a plurality of pilot signal products, where each pilot signal product includes complex plane information represented by a reference frequency subcarrier, multiplying complex plane information represented by mirror frequency subcarrier. The sum of the plurality of pilot signal products is equal to zero.
• the transmitter 402 supplies the frequency- smoothed unbiased training signal within a single symbol period.
• the components of the frequency- smoothed training signal are serially supplied or supplied in batches, and collected, in a memory (not shown) for example. Once the entire FSTS is collected, it can be supplied for use within a single symbol period.
• the memory/collection and combinations means may be considered to be part of the transmitter 402, even if they are enabled in separate modules or devices (not shown). It should also be understood that in some aspects the transmitter 402 acts as a signal generation, while the actual sending of the FSTS over a communication medium is performed by other modules or devices.
• the transmitter 402 also sends quadrature modulated (non-predetermined) communication data.
• the frequency- smoothed unbiased training signal is used by a receiver (not shown) to create channel estimates, which permit the non-predetermined communication data to be recovered more accurately.
• the quadrature modulated communication data is sent subsequent to sending the unbiased training sequence.
• the unbiased training sequence is sent concurrently with the communication data in the form of pilot signals.
• the system is not limited to any particular temporal relationship between the training signal and the quadrature modulated communication data.
• a message is a grouping of symbols in a predetermined format.
• a message may have a duration of several symbols periods. One or more symbols may be transmitted every symbol period.
• Some messages include a preamble preceding the main body of the message. For example, a message may be formed as a long packet containing many OFDM, CDMA, or TDMA symbols.
• the FSTS may be comprised of 2, or more than 2 pilot signal products.
• the transmitter 402 generates a frequency- smoothed unbiased training signal including a plurality of adjacent reference frequency subcarriers and a plurality of adjacent mirror frequency subcarriers.
• the reference subcarriers and corresponding mirror subcarriers are within a relatively close (spectrum-wise) proximity.
• the frequency- smoothed unbiased training signal may include a group of adjacent reference frequency subcarriers, without intervening subcarriers, and a plurality of adjacent mirror frequency subcarriers, without intervening subcarriers.
• An intervening subcarrier may, for example, be a subcarrier carrying communication (non-predetermined) data or other information not associated with the training signal.
• the group includes all the reference and mirror subcarriers in the FSTS.
• the frequency- smoothed unbiased training signal may be represented as follows:
• the frequency- smoothed unbiased training signal may be represented using weighted pilot signal products as follows:
• FIG. 5 is a diagram depicting a simple example of a frequency- smoothed unbiased training signal.
• a first pilot signal product has a reference subcarrier 500 at frequency + f representing information as a first complex plane value, and a mirror subcarrier 502 at frequency — f representing the first complex plane value.
• the subcarrier "arrows" can be thought of as phasors having an amplitude of 1 and an angle of 90 degrees.
• a second pilot signal product has a reference subcarrier 504 at frequency (f+1), adjacent frequency +f, representing the first complex plane value, and a mirror subcarrier 506 at frequency — (f + 1), adjacent the frequency -f, representing the first complex plane value +180 degrees.
• the arrow representing mirror subcarrier 506 would have an amplitude of 1 and an angle of 270 degrees.
• all the subcarriers have the same value, normalized to 1, in this example, it should be understood that more complex variations of this example may use non-uniform amplitudes.
• the FSTS is not limited to the use of only 90 degree and 270 degree angles.
• the same methodology would apply to a FSTS with more than 2 pilot signal products.
• the depicted FSTS could be modified to add a third pilot signal product (not shown) with subcarrier at frequency (f-1), adjacent frequency +f, representing the first complex plane value, and a mirror subcarrier at frequency — (f - 1), adjacent the frequency -f, representing the first complex plane value +180 degrees.
• FIG. 6 is a diagram depicting a second example of frequency- smoothed unbiased training signal.
• a first pilot signal product has a reference subcarrier 600 at frequency + f representing information as a first complex plane value, and a mirror subcarrier 602 at frequency — f representing the first complex plane value.
• the subcarrier "arrows" can be thought of as phasors having an amplitude of 1 and an angle of 90 degrees.
• a second pilot signal product has a reference subcarrier 604 at frequency (f+1), adjacent frequency +f, representing the first complex plane value + 90 degrees, and a mirror subcarrier 606 at frequency — (f + 1), adjacent frequency -f, representing the first complex plane value -90 degrees.
• the arrow representing reference subcarrier 604 would have an amplitude of 1 and an angle of 180 degrees
• the mirror subcarrier 606 would have an amplitude of 1 and an angle of 0 degrees.
• the subcarriers have the same value in this example, normalized to 1, and it should be understood that more complex variations of this example may use non-uniform amplitudes.
• the FSTS is not limited to the use of only 0 degree, 90 degree, and 180 degree angles.
• the same methodology would apply to a FSTS with more than 2 pilot signal products.
• the depicted FSTS could be modified to add a third pilot signal product (not shown) with subcarrier at frequency (f-1), adjacent frequency +f, representing the first complex plane value + 90 degrees, and a mirror subcarrier at frequency — (f - 1), adjacent the frequency -f, representing the first complex plane value -90 degrees.
• FIG. 7 is a diagram depicting an unbiased training signal enabled as a group of pilot symbols accompanying communication symbols.
• the transmitter accepts (non-predetermined) communication data.
• an unbiased frequency-smoothed training signal is generated with P pilot signal products, along with (N — P) communication data symbols (subcarrier s).
• N subcarriers are supplied in one symbol period, including the frequency-smoothed unbiased training signal and quadrature modulated communication data.
• Many communications systems such as those compliant with IEEE 802.11 and UWB use pilot tones for channel training purposes.
• the components of the frequency-smoothed training signal (FSTS), or the communication data symbols, or both may be serially supplied or supplied in batches, and collected, in a memory (not shown). Once all the symbols in the symbol period are collected, they can be supplied for use within a single symbol period.
• the memory/collection and combinations means may be considered to be part of the transmitter, even if they are enabled in separate modules or devices.
• FIG. 8 is a diagram depicting a frequency-smoothed unbiased training signal enabled as a preamble preceding non-predetermined communication data.
• the transmitter supplies the frequency- smoothed unbiased training signal in a first symbol period using a group of reference frequency subcarriers and corresponding mirror frequency subcarriers.
• the transmitter accepts communication data, generates quadrature modulated communication data on the group of reference frequency subcarriers and corresponding mirror frequency subcarriers, and supplies the quadrature modulated communication data in a second symbol period, subsequent to the first symbol period.
• the preamble may be comprised of a plurality of symbol periods, with an FSTS used in some or all of the preamble symbol periods.
• communication data may be supplied in a plurality of symbol periods (as shown) following the preamble.
• UWB Ultra Wideband
• an Ultra Wideband (UWB) system uses 6 symbol periods transmitted prior to the transmission of communication data or a beacon signal. Therefore, one or more of the 6 symbol periods may be used for the transmission of a FSTS.
• the transmitter of Fig. 4, or elements of the transmitter may be enabled as a processing device for generating a frequency-smoothed unbiased training signal.
• the processing device would comprise a signal generator module having an input to accept training information and an output to supply a quadrature modulated frequency- smoothed unbiased training signal.
• the frequency- smoothed unbiased training signal would include a plurality of pilot signal products, where each pilot signal product includes complex plane information represented by a reference frequency subcarrier, multiplying complex plane information represented by mirror frequency subcarrier. Also as above, the sum of the plurality of pilot signal products is equal to zero.
• the signal generator module would supply the frequency- smoothed unbiased training signal within a single symbol period.
• FIG. 9 is a schematic block diagram of a system or device for calculating a channel estimate using a frequency-smoothed unbiased training signal.
• the system or device 900 comprises a quadrature demodulation receiver or receiving means 902 having an input on line 904 to accept a frequency-smoothed unbiased training signal.
• the receiver 902 may be an RF device connected to an antenna 905 to receive radiated information.
• the receiver may alternately receive the unbiased training sequence via a wired or optical medium (not shown).
• the receiver 902 has an in-phase (I) demodulation path 906 for accepting I demodulation training information.
• a quadrature (Q) demodulation path 908 accepts Q demodulation training information.
• the receiver 902 includes analog-to digital converters (ADC) 909, a fast Fourier transformer (FFT) 910, a deinterleaver 912, and a decoder 914.
• ADC analog-to digital converters
• FFT fast Fourier transformer
• deinterleaver 912 a deinterleaver 912
• decoder 914 The receiver supplies training information in response to receiving the FSTS.
• the frequency- smoothed unbiased training sequence includes a plurality of a plurality of pilot signal products.
• a pilot signal product includes predetermined complex plane information (p) represented by a reference frequency subcarrier (f), multiplied by predetermined complex plane information (p m ) represented by mirror frequency subcarrier (-f). The sum of the plurality of pilot signal products is equal to zero.
• a processor or processing means 916 has an input on line 918 to accept the training information, the processor generates a plurality of processed symbols (y) representing complex plane information. The processor 916 multiplies each processed symbol (y) by a conjugate of a corresponding reference signal (p*), and supplies a frequency- smoothed channel estimate (h) at an output on line 920.
• the receiver 902 supplies the training information as the output of the ADCs 909.
• the FFT, deinterleaver, and decoder processes, or their equivalent, are performed by processor 916.
• the FSTS is comprised of 2 or more pilot signal products.
• the receiver accepts a frequency-smoothed unbiased training signal including a plurality of adjacent reference frequency subcarriers and a plurality of adjacent mirror frequency subcarriers. The meaning of "adjacent" is dependent upon subcarrier spacing, frequency, and other modulation characteristics.
• the frequency- smoothed unbiased training signal includes a group of adjacent reference frequency subcarriers, without intervening subcarriers, and a plurality of adjacent mirror frequency subcarriers, without intervening subcarriers. This group may include all, or only a subset of all the subcarriers in the FSTS.
• the received frequency- smoothed unbiased training signal may be expressed as the transmitted FSTS, as follows:
• the received FSTS may include weighted pilot signal products as follows:
• the receiver 902 may accept an unbiased frequency- smoothed training signal with P pilot signal products and (N — P) communication data symbols in the same symbol period, and supply both training information and communication data, also see FIG. 7.
• the receiver 902 accepts a frequency- smoothed unbiased training signal in a first symbol period, with a group of reference frequency subcarriers and corresponding mirror frequency subcarriers.
• the receiver also accepts quadrature modulated communication data on the group of reference frequency subcarriers and corresponding mirror frequency subcarriers in a second symbol period, subsequent to the first symbol period, and supplies communication data, see FIG. 8.
• the receiver of FIG. 9 may also be enabled as a processing device for calculating a channel estimate using a frequency-smoothed unbiased training signal.
• the processing device comprises a receiver module having an input to accept a frequency-smoothed unbiased training sequence and an output to supply training information.
• the frequency-smoothed unbiased training sequence includes a plurality of pilot signal products, where each pilot signal product includes predetermined complex plane information (p) represented by a reference frequency subcarrier (f), multiplying predetermined complex plane information (p m ) represented by mirror frequency subcarrier (-f). Also as above, the sum of the plurality of pilot signal products is equal to zero.
• a calculation module has an input to accept the training information.
• the calculation module generates a plurality of processed symbols (y) representing complex plane information, multiplies each processed symbol (y) by a conjugate of a corresponding reference signal (p*), and supplies a frequency- smoothed channel estimate (h) at an output.
• Training signals whether enabled in a preamble or as pilot signals are similar in that the information content of transmitted data is typically predetermined or "known" data that permits the receiver to calibrate and make channel measurements. When receiving communication (non- predetermined) data, there are 3 unknowns: the data itself, the channel, and noise. The receiver is unable to calibrate for noise, since noise changes randomly. Channel is a measurement commonly associated with delay and multipath.
• the errors resulting from multipath can be measured if predetermined data is used, such as training or pilot signals. Once the channel is known, this measurement can be used to remove errors in received communication (non-predetermined) data. Therefore, some systems supply a training signal to measure a channel before data decoding begins.
• the channel can change, for example, as either the transmitter or receiver moves in space, or the clocks drift.
• many systems continue to send more "known” data along with the "unknown” data in order to track the slow changes in the channel.
• the transmitter of FIG. 4 and the receiver of FIG. 9 may be combined to form a transceiver.
• the transmitter and receiver of such a transceiver may share elements such as an antenna, baseband processor, and MAC level circuitry.
• the explanations made above are intended to describe a transceiver that both transmits unbiased training sequences and calculates unbiased channel estimates based upon the receipt of unbiased training sequences from other transceivers in a network of devices.
• Modern high data rate communication systems transmit signals on two distinct channels, the in-phase and quadrature-phase channels (I and Q).
• the two channels form a 2D constellation in a complex plane.
• QPSK and QAM are examples of constellations.
• the I and Q channels may be carried by RF hardware that cannot be perfectly balanced due to variations in RF components, which results in IQ imbalance.
• the imbalance issued are even greater.
• IQ imbalance distorts the constellation and results in crosstalk between the I and Q channels: the signal interferes with itself. Increasing transmission power does not help, since self-generated interference increases with the signal power.
• the signal-to-noise ratio reaches an upper bound that puts a limit on the highest data rate attainable with a given RF hardware.
• SINR signal-to-noise ratio
• a costly solution is to use fancier, more expensive hardware.
• a possibly less costly solution is to digitally estimate IQ imbalance and compensate for it.
• the concepts of digital estimation and compensation algorithms have been previously advanced in the art. However, the solutions tend to be expensive because they do not rely on a special type of training sequence. These solutions often only consider imbalance at one side, usually at the receiver.
• OFDM Orthogonal Frequency Division Multiplexing
• time domain systems which study end-to-end imbalance, from transmitter to receiver.
• OFDM Orthogonal Frequency Division Multiplexing
• the imbalance is modeled as a function of frequency, taking into account variations in the frequency response of the filters.
• Two kinds of enhancements are presented: one with zero cost that eliminates the interference from the channel estimate by using an unbiased training sequence. Substantial gains are achieved because the error of the channel estimate is often more detrimental to performance than the error in the data itself.
• a second, relatively low cost, enhancement compensates for data distortion, if more gain is needed.
• a model of the IQ imbalance is provided below. Analysis is provided to show how conventional channel estimation using unbiased training sequences can mitigate part of the IQ imbalance. Then, a straightforward extension is provided to calculate the IQ imbalance parameters, proving that the algorithms are effective. Using the estimated parameters, a simple compensation algorithm is presented to mitigate data distortion. Simulation results for WiMedia's UWB are also given, as well as suggestions to amend the standard.
• IQ imbalance arises when the power (amplitude) balance or the orthogonality (phase) between the in-phase (I) and quadrature-phase (Q) channels is not maintained. IQ imbalance is therefore characterized by an amplitude imbalance 2 ⁇ and a phase imbalance 2 ⁇ .
• a complex symbol x is transmitted and received via the I and Q channels.
• the symbol x is received intact. But in the presence of IQ imbalance, a noisy or distorted version is likely received.
• B is the imbalance matrix.
• the second row is obsolete since it is a duplicate version of the first row. But it gives a same size and type input and output so imbalance blocks at transmitter and receiver can be concatenated, as described below.
• the imbalance matrix at the transmitter is defined by Bt, and at the receiver it is defined by B 1 -.
• a one-tap channel is considered, suitable for OFDM.
• a one-tap channel h in appropriate matrix form is
• IQ imbalance and channel combine to create a global channel h', plus an undesired distortion or interference characterized by a global imbalance parameter ⁇ '.
• the global imbalance parameter ⁇ ' changes when the channel changes, and may need to be estimated regularly.
• the condition is considered where the symbol x, rather than spanning the entire complex plane, is restricted to a given (ID) axis.
• the axis may be associated with BPSK modulation, the real axis, the imaginary axis, or any axis in between.
• x * kx may be written, where k is a complex constant (a rotation), and
• the interference created by IQ imbalance does not show up at the same frequency f, but rather at the mirror frequency — f, and vice versa. What is transmitted at -f creates interference on frequency +f. If signal x m is the signal transmitted at frequency -f, where index m denotes a quantity at mirror frequency — f, then at frequency — f the following is obtained
• the IQ imbalance parameters ⁇ and ⁇ are here a function of frequency. This models an imbalance due to different low-pass (base-band) or band-pass (IF) filters in the system.
• the I and Q paths cannot have the exact same filters and, hence, the imbalance varies with frequency.
• This kind of imbalance exists but it is very expensive to compensate.
• An equalizer and an extension of the model to deal with different convolutions on different channels are required. So in the time domain, bulk or average imbalance is used. Frequency domain systems are able to take advantage of the plain equalizer structure and model the imbalance on a per frequency basis.
• the second row is no longer obsolete.
• the model deals, in one shot, with a pair of mirror frequencies.
• a one-tap channel h at frequency f, and h m at frequency -f is modeled by the matrix
• h', h m ' are the global channel taps, and ⁇ ', ⁇ m ' are the global imbalance parameters.
• the imbalance parameters change when the channels change and may need to be estimated regularly.
• the model (12) is stimulated with pilot tones. At frequency +f, the pilot p is transmitted, and at frequency — f, the pilot p m . Assuming, without loss of generality, that the pilots have a unit norm (the channel carries the effective power), the conventional channel estimate at frequency f is obtained by de-rotating by p *
• the training sequence consists of an even number of symbols, and it is enough to ensure each pair adds up to zero
• Examples of simple sequences that satisfy the condition are given in Table 1. These types of training sequences are denoted as unbiased training sequences because, on one hand, unbiased channel estimates are produced, and on the other, the training signals equally spans the I and Q dimensions of the complex plane in time domain. For example, an unbiased training sequence is not concentrated along just the real axis.
• the channel estimate may be smoothed over adjacent subcarriers within one symbol.
• the cyclic prefix is designed to be short, and the channel is supposed to vary slowly from tone to tone.
• the filters in the RF chain should have short temporal response and their frequency response also varies slowly, i.e., the IQ imbalance varies slowly across subcarriers.
• the same channel smoothing techniques can be used to smooth and improve the imbalance parameter estimate.
• unbiased training sequences there is no interaction between the channel estimate and the imbalance estimate.
• Each estimated can be independently smoothed.
• a nearly unbiased training sequence can be obtained by applying the summation from equation (14) over groups of 2 or more adjacent subcarriers. Then smoothing automatically cancels all or part of the interference from mirror frequencies.
• One solution is to rotate the pilot by 90 degrees on the adjacent subcarrier (moving in mirror directions on the positive and negative frequencies).
• the channel estimate (h v ) is equal to the original channel h plus some undesired terms:
• index i now denotes adjacent subcarriers rather than different symbol periods.
• Index j is the center pilot, and w ⁇ is the weighting function. Then, the formula w ⁇ Bi m pi * pi m * can be written as a convolution:
• the weight function w is known and is dependent upon how the channel is averaged.
• the pilots p and p m are unknowns.
• B m is unknown in this equation and depends on the physical hardware (IQ imbalance). But since B m is unknown at the time the p and p m are being optimized, and if it is assumed the averaging over adjacent frequencies means that Bm does not vary noticeably, then is can be assumed that B m is a constant independent of index i. Hence, the term B m can be dropped from the equation to obtain
• the power can remain constant as the index i changes.
• A contains dependent terms. That is, A(i) and A(im) are equal terms. So only half of "A” is unknown, which makes the problem a bit harder to solve. But if w is symmetric around the origin, then the matrix and vector can be wrapped around in a mirror fashion. What remains is a half-size matrix, and vector W and A'. Then, the optimal convolution WA is:
• L transmissions Xi, L noise terms Ni and L observations Y i may be respectively concatenated into the 2 by L matrices
• the unknown is H'.
• the LS estimator is
• Model (17) can be viewed as unknown information H' sent via 2 consecutive transmissions over 2 vectors (rows of ⁇ ) in an L dimension space. We denote by ⁇ ⁇ , M ⁇ and 3 j respectively row j of .Y, A and J , where j ⁇ ⁇ 1,2 ⁇ . Models (12) and (17) can be written
• the LS estimator consists of projecting onto each vector, in a parallel way to the other vector in order to cancel interference.
• a very good result is obtained when the 2 vectors are orthogonal, i.e., when dot product (14) is zero.
• Unbiased training sequences are by definition, training sequences that verify this condition. Other sequences use non-orthogonal vectors and suffer a loss of performance function of the angle between the vectors Vi and Y 2 .
• Many OFDM systems currently use a very poor kind of training sequences where ⁇ lv ⁇ 2 are collinear, and it is impossible to properly estimate the 4 entries in H'. These training sequences tend to estimate noisier versions of the channels h' and h' m .
• H'-H' This is a 2 by 2 matrix, i.e., 4 error values. Each value can be isolated by multiplying left and right with combinations of the vectors M ⁇ ⁇ and K ⁇ ⁇ ⁇ .
• EA ⁇ A *H is an identity matrix, or more generally a diagonal matrix with elements ⁇ 2 and ⁇ m 2 , it can be shown that the MSE of h' and * m ' are, respectively, the first and second diagonal elements of ⁇ 2 (.1.V 11 ) 1 .
• the MSE are, respectively, the first and second diagonal element of Om 2 ClW 11 )- 1 .
• the total MSE is 2( ⁇ 2 + ⁇ m 2 )tr(.l ,V 1 *)- 1 .
• tr(,Y.V H ) 2L.
• ⁇ IAj subject to ⁇ ⁇ j is constant.
• the total MSE has been minimized, and the resulting MSE per element is either ⁇ 2 /L or ⁇ m 2 /L. But this MSE per element is likely to be the best that can be obtained, even if a unique vector transmission is used. The MSE is unlikely to be improved for a 2 vector transmissions, and therefore the MSE per element has been minimized.
• the unbiased training sequences plus conventional channel estimator are the MMSE of all LS estimators.
• the IQ imbalance parameters may be estimated (as described previously) and applied to compensate for data distortion.
• the model is the same as any 2-tap channel with cross-correlations. Any channel equalization algorithm can be fitted.
• a simple equalization algorithm is presented suitable for the ubiquitous bit-interleaved coded QAM and fading channels.
• the ZF technique consists of computing
• Table 3 summarizes the ZF algorithm with noise enhancement avoidance.
• FIG. 10 depicts the performance achieved by applying unbiased training signal algorithms to the WiMedia UWB standard.
• the highest data rate, 480 Mbps, is simulated in IEEE 802.15.3's channel model CM2 (indoor pico-environment of about 4 meters). Shadowing and band hopping are turned off.
• the figure shows the Packet Error Rate (PER) as a function of Eb/No. The performance degrades quickly without any form of compensation.
• Table 4 lists the loss of various algorithms with respect to ideal case.
• WiMedia UWB loss from IQ imbalance at PER of 10 2 Current StandardUnbiased TrainingCompensation
• End-to-end IQ imbalance and channel combine to form a global 2 by 2 channel matrix.
• the use of unbiased training sequences achieves considerable gains at no cost.
• the unbiased training sequences automatically cancel end-to-end self-generated interference from the channel estimate.
• such training sequences are ideal for estimating IQ imbalance parameters, and a simple algorithm is given to compensate for data distortion: Zero- Forcing with noise enhancement avoidance.
• WiMedia UWB benefits from the following enhancement: the conventional biased training sequence that consists of 6 symbols exclusively transmitted on the I channel can be divided in 2 halves to create an unbiased sequence. The first 3 symbols are sent on the I channel, and the last 3 symbols are sent on the Q channel. By uniformly spanning the complex plane, an unbiased training sequence is created with large gains for high data rates. For backward compatibility, this scheme may be reserved for high data rate modes and signaled via the beacons, or the training sequence type may be blindly detected.
• the subcarriers f and -f can be assigned to different users. Considerable interference can arise if power control drives one user to high power level. It is therefore a good idea to locate the pilots of different users on mirror subcarriers. The pilots should satisfy the unbiased training sequence criterion. Each user automatically benefits without any extra effort. The pilots may hop to different locations while maintaining mirror positions.
• the time domain formulas can be extended to Code Division Multiple Access (CDMA) with a Rake equalizer combining several one-tap channels. Unbiased training sequences automatically improve the channel estimate per tap.
• CDMA Code Division Multiple Access
• a simple unbiased training sequence for CDMA consists of constantly rotating the complex symbols by 90 degrees.
• the theory can be extended to other time domain systems (e.g., TDMA) besides CDMA.
• the channel estimate is obtained by the convolution of the received signal with a matched filter, which is mirror version of the complex conjugate of the FSTS. In other words, ignoring the AWGN, it is a convolution of the channel, the transmitted FSTS, and the matched filter. It can be shown that the channel estimate contains a self-interference (bias) term generated by the IQ imbalance. By considering the equation in frequency domain, the self-interference term can be made to nearly vanish if the FSTS is carefully chosen. Indeed, by using the above- described FSTSs, the self-interference tends to cancel after summing up the values from adjacent tones (assuming slow channel variations). Hence, an unbiased training signal for time domain systems can be designed with frequency domain constraints on adjacent tones.
• FIG. 11 is a flowchart illustrating a method for supplying a frequency- smoothed communications training signal. Although the method is depicted as a sequence of numbered steps for clarity, the numbering does not necessarily dictate the order of the steps.
• Step 1100 the terms “generating”, “deriving”, and “multiplying” refer to processes that may be enabled through the use of machine-readable software instructions, hardware, or a combination of software and hardware.
• the method starts at Step 1100.
• Step 1102 generates a frequency- smoothed unbiased training signal in a quadrature modulation transmitter.
• the frequency-smoothed unbiased training signal includes a plurality of pilot signal products, where each pilot signal product includes complex plane information represented by a reference frequency subcarrier, multiplying complex plane information represented by mirror frequency subcarrier. The sum of the plurality of pilot signal products is equal to zero.
• Step 1104 supplies the frequency- smoothed unbiased training signal within a single symbol period.
• the components of the FSTS may be supplies serially or in batches, and stored until a complete FSTS is collected. In this aspect, a subsequent step (not shown) would transmit the collected FSTS is a single symbol period.
• Step 1102 generating a frequency-smoothed unbiased training signal in Step 1102 including a generating plurality of adjacent reference frequency subcarriers and a plurality of adjacent mirror frequency subcarriers.
• the FSTS may be comprised of 2 or more pilot signal products.
• Step 1102 generates the frequency-smoothed unbiased training signal that includes with a group of adjacent reference frequency subcarriers, without intervening subcarriers, and a plurality of adjacent mirror frequency subcarriers, without intervening subcarriers.
• the frequency- smoothed unbiased training signal may be represented as follows, as depicted in Step 1102b:
• the frequency-smoothed unbiased training signal may be comprised of a first pilot signal product with a reference subcarrier at frequency + f representing information as a first complex plane value, and a mirror subcarrier at frequency — f representing the first complex plane value.
• the FSTS further includes a second pilot signal product with a reference subcarrier at frequency (f+1), adjacent frequency +f, representing the first complex plane value, and a mirror subcarrier at frequency — (f + 1), adjacent the frequency -f, representing the first complex plane value +180 degrees.
• the frequency- smoothed unbiased training signal may be generated as a first pilot signal product with a reference subcarrier at frequency + f representing information as a first complex plane value, and a mirror subcarrier at frequency — f representing the first complex plane value.
• the FSTS also includes a second pilot signal product with a reference subcarrier at frequency (f+1), adjacent frequency +f, representing the first complex plane value + 90 degrees, and a mirror subcarrier at frequency — (f + 1), adjacent frequency -f, representing the first complex plane value -90 degrees.
• generating the unbiased frequency- smoothed training signal in Step 1102 includes generating P pilot signal products.
• Step 1103 generates (N - P) communication data symbols.
• Step 1104 supplies the FSTS and communication data symbols is a single symbol period.
• Step 1106 transmits N subcarriers in one symbol period, including the frequency-smoothed unbiased training signal and quadrature modulated communication data.
• Step 1102 generates a frequency-smoothed unbiased training signal using a group of reference frequency subcarriers and corresponding mirror frequency subcarriers
• Step 1104 supplies the frequency-smoothed training signal in a first symbol period
• Step 1108 generates quadrature modulated communication data on the group of reference frequency subcarriers and corresponding mirror frequency subcarriers
• Step 1110 supplies the quadrature modulated communication data in a second symbol period, subsequent to the first symbol period.
• the above- described flowchart may also be interpreted as an expression of a machine-readable medium having stored thereon instructions for a frequency- smoothed communications training signal. The instructions would correspond to Steps 1100 through 1110, as explained above.
• FIG. 12 is a flowchart illustrating a method for calculating a channel estimate using a frequency- smoothed unbiased training signal.
• the method starts at Step 1200.
• Step 1202 accepts a frequency- smoothed unbiased training sequence in a quadrature demodulation receiver.
• the frequency- smoothed unbiased training sequence includes a plurality of pilot signal products, where each pilot signal product includes predetermined complex plane information (p) represented by a reference frequency subcarrier (f), multiplying predetermined complex plane information (p m ) represented by mirror frequency subcarrier (-f).
• the sum of the plurality of pilot signal products is equal to zero.
• Step 1204 processes the frequency- smoothed unbiased training signal, generating a plurality of processed symbols (y) representing complex plane information.
• Step 1206 multiplies each processed symbol (y) by a conjugate of a corresponding reference signal (p*).
• Step 1208 obtains a frequency-smoothed channel estimate (h).
• the processed symbols (y) are associated with the reference subcarrier.
• Step 1204 may processes the frequency- smoothed unbiased training signal, generating a plurality of processed symbols (y m ) representing complex plane information associated with the mirror subcarrier.
• Step 1206 multiplies each processed symbol (y m ) by a conjugate of a corresponding reference signal (p m *), and Step 1208 obtains a frequency- smoothed channel estimate (h m ) associated with the mirror subcarrier.
• the above-mentioned steps find both the (h) and (h m ) channel estimates.
• Step 1202 accepts plurality of adjacent reference frequency subcarriers and a plurality of adjacent mirror frequency subcarriers.
• the FSTS may be comprised of 2 or more pilot signal products.
• Step 1202 accepts the frequency- smoothed unbiased training signal including a group of adjacent reference frequency subcarriers, without intervening subcarriers, and a plurality of adjacent mirror frequency subcarriers, without intervening subcarriers.
• the frequency- smoothed unbiased training signal may be represented as follows:
• the frequency- smoothed unbiased training signal may be represented as follows:
• the frequency-smoothed unbiased training signal may be comprised of a first pilot signal product with a reference subcarrier at frequency + f representing information as a first complex plane value, and a mirror subcarrier at frequency — f representing the first complex plane value.
• the FSTS further includes a second pilot signal product with a reference subcarrier at frequency (f+1), adjacent frequency +f, representing the first complex plane value, and a mirror subcarrier at frequency — (f + 1), adjacent the frequency -f, representing the first complex plane value +180 degrees.
• the frequency- smoothed unbiased training signal may be generated as a first pilot signal product with a reference subcarrier at frequency + f representing information as a first complex plane value, and a mirror subcarrier at frequency — f representing the first complex plane value.
• the FSTS also includes a second pilot signal product with a reference subcarrier at frequency (f+1), adjacent frequency +f, representing the first complex plane value + 90 degrees, and a mirror subcarrier at frequency — (f + 1), adjacent frequency -f, representing the first complex plane value -90 degrees.
• Step 1202 accepts the unbiased frequency- smoothed training signal as P pilot signal products in a symbol period. Then, Step 1203 accepts (N — P) communication data symbols in the (same) symbol period. [00134] Alternately, Step 1202 accepts a frequency- smoothed unbiased training signal with a group of reference frequency subcarriers and corresponding mirror frequency subcarriers. Then, Step 1210 accepts quadrature modulated communication data on the group of reference frequency subcarriers and corresponding mirror frequency subcarriers, subsequent to the receipt of the frequency-smoothed unbiased training signal.

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