EP1406349B1 - Active wide-band reception antenna with regulation of the receiving level - Google Patents

Abstract

The device has a passive part with output connections connected to amplifier stage input connections and reduced internal impedance if a defined reception level is exceeded. The amplifier input stage amplifying element's high impedance control port has a HF connection to the passive part. An electronic element reduces the reception level in a transfer network whose linearising input admittance is lower when the HF input signal is attenuated. The device consists of a passive part (1) whose output connections are connected to the input connections of an amplifier stage (21) and its internal impedance is reduced when a defined reception level is exceeded. The amplifier-input stage contains a 3-pole amplifying element (2) whose high impedance control port (15) has a high frequency connection to the first connection of the passive part. The amplifier has an electronic element (32) for reducing the reception level in a transfer network (31), whose linearising input admittance (7') is lower when the high frequency input signal is attenuated.

Description

The invention relates to an active broadband receiving antenna, comprising a passive antenna part 1 with a frequency-dependent effective length l _e , whose output terminals are connected to the input terminals of an amplifier circuit 21 high frequency. Electrically long antennas or antennas that are in direct coupling with electrically large bodies have a frequency-dependent open-circuit voltage when excited with an electrical field strength held constant above the frequency, which is expressed by the effective length l _e (f). Especially in the frequency range above 30 MHz, the antenna noise temperature T _A in terrestrial environment - coming from low frequencies - has fallen so far that for bipolar transistors from the side of the passive antenna part, a source impedance in the vicinity of the optimal impedance for the transistor impedance Z _opt is required for noise adaptation, so as not to suffer any significant sensitivity loss due to the transistor noise. The basic form of an active antenna of this kind is in Fig. 2b shown and is known for example from DT-AS 23 10 616, the DT-AS 15 91 300 and AS 1919749. In active broadband antennas, which are not channel selective, but are tuned to a frequency band, such as the FM radio frequency band broadband is it is necessary to transform the antenna impedance Z _S (f) of a short radiator in Z _A (f) into the vicinity of Z _opt (see VHF range in DT-AS 23 10 616) or to design the radiator itself in such a way that the antenna impedance Z _S (f) itself lies in the vicinity of Z _opt (see FM range in AS 1919749). This leads both in electrically large and in electrically small antennas to a frequency-dependent open circuit voltage at the transistor input, which expresses as a strong frequency-dependent effective length l _e (f) of the passive antenna part, resulting in connection with the frequency dependence of the voltage division factor between Z _opt and the deviating input resistance of the transistor results in the need to smooth the resulting frequency response of the received signal at the load resistor Z _L by means of a matching circuit at the output of the active circuit. This is also necessary to protect the subsequent receiving system against non-linear effects due to level overload.

In the case of broadband receiving antennas, the high electrical field strengths in the vicinity of the transmitter, for example also by on-board transmitters, can lead to strong interference due to intermodulation and limiting effects in the electronic amplifier of the active receiving antenna, since this is highly sensitive in terms of high sensitivity and in terms of broadband compliance with the electrical Characteristics is dimensioned. The technique used is usually very complex, with the effort with increasing demand for intermodulation strength increases greatly. In active receiving antennas, which use a rectifier circuit with control circuit to determine the signal level, but cheaper amplifier can be used, since they are able to lower the gain of the active receiving antenna when a predetermined receive level is exceeded in order to receive interference by intermodulation and To avoid limiting effects in the amplifier and in the secondary circuit.
Narrow bandwidth receivers typically do not need to be protected from nonlinear effects by level overload. In the U.S. Patent 4,875,019 is z. Example, a receiver preamplifier with a matched to the fixed frequency for the narrow-band reception of Loran signals input resonant circuit specified. Measures to reduce the reception level are therefore not provided.

In the DE 43 23 014 describes an active broadband antenna in which the antenna impedance to be measured is transformed by means of a low-loss transformation network into the optimal source impedance of the subsequent electronic amplifier to achieve an optimal signal-to-noise ratio. To protect the subsequent receiving system against non-linear effects due to level overload is often necessary to lower the gain of the active antenna. In the DE 43 23 014 the exceeding of a predetermined reception level is detected by means of a rectifier circuit and lowered by means of a control amplifier, the gain of the active antenna. This is done by providing a passive, signal dampening network that bridges the active antenna portion and reducing the gain of the active receive antenna by providing the signal path through the electronic amplifier at its input, or at its output or at its output Input and output is separated by the electronic switch. In this case, the load occurring at the amplifier input by the bridging signal-damping network together with the switching measures to be applied there is distracting.

The basic form of active antennas with a transformation network at the amplifier input, as they are used as broadband antennas for the FM area, is in Fig. 2b shown and is known, for example from DT-AS 23 10 616, the DT-AS 15 91 300. Active antennas according to this prior art, for example, mounted on a large scale above the high frequency range with antenna arrangements in a motor vehicle window, together with a heater for the window heating , such as in EP 0 396 033 . EP 0 346 591 and in EP 0 269 723 described. The structures of the heating fields used as passive antenna part 1 are not originally intended for use as an antenna vehicle parts, which are only slightly changed due to their function for heating. If such an antenna element, an active antenna according to the prior art as in Fig. 2b realized, the existing impedance on the heating field by means of a primary matching circuit in the vicinity of the impedance Z _opt for noise adaptation to transform and to smooth the frequency response of the active antenna by means of an output-side matching network. This approach requires the relatively complicated dimensioning of two filter circuits, which can not be done separately for each filter for an advantageous overall behavior of the active antenna due to the mutual dependence of each other. In addition, the amplifier circuit is not intended to provide sufficient linearity characteristics as a simple amplifying element as in Fig. 2b can be designed, whereby the creative freedom of the two matching networks is considerably narrowed. In addition, the design of two filters associated with increased effort. Another significant disadvantage of an active antenna of this kind is the load of the matching circuit connected to the heating field with downstream amplifier, if from the same heating field several active antennas are designed to form an antenna diversity system or a group antenna with special Richteigenschaften or other purposes. This disadvantageous situation exists in all antenna arrangements whose passive antenna parts are in appreciable electromagnetic radiation coupling with one another. For example, according to the state of the art in a multi-antenna scanning diversity system formed from the heating field at the connection points formed for the antenna amplifier switching diodes attached, which in each case only those matching circuit with amplifier turns on, whose signal is turned on to the receiver and which the other connection points unlock. This leads in such systems to a considerable effort and to the additional requirement of the synchronous switching of the diodes, which is exactly synchronized with the antenna selection.

An active broadband receiving antenna which does not possess the latter disadvantages is disclosed in the non-prepublished European patent application EP 02004597 , published as EP 1 246 294 , in Fig. 1 described. However, it has the remaining disadvantage that there is no effective means for lowering the gain of the active antenna when exceeding a predetermined reception level for protection against non-linear effects.

The object of the invention is therefore to design an active broadband receiving antenna according to claim 1 so that an effective means for reducing the gain of the active antenna is given when exceeding a predetermined reception level for protection against non-linear effects.

This object is achieved by the features of claim 1.

The advantages attainable with the invention consist in particular in the reduction of the economic outlay and in the simplicity to achieve an optimum in terms of signal-to-noise ratio and the risk of non-linear effects optimal received signal. The achievable by the features of the main claim high linearity of the three-pole amplifying element 2 allow to make the reduction of the gain of the active antenna at the output of this element in conjunction with a simultaneously achieved increase in the linearizing negative feedback. Due to the omission of a primary matching network in conjunction with the input-side high impedance of the amplifier circuit results in a very advantageous freedom in the design of complex multi-antenna systems whose passive antenna parts are in radio coupling to each other. This results in the advantageous property that for multiple antenna arrangements, the multiple coupling out of received signals from a passive antenna arrangement with several connection points, which are in electromagnetic radiation coupling to each other, given by the formation of active antennas no significant mutual influence of the received signals. In connection with the diversity arrangement, the above-mentioned switching diodes for enabling

Embodiments of inventive active broadband receiving antennas and antenna systems are shown in the drawings and will be described in more detail below. In detail shows:

Fig. 1:

Active broadband receiving antenna according to the invention with an amplifier circuit 21 connected directly to the passive antenna part 1 with a three-pole amplifying element 2, with input admittance 7 of the transmission network 31 with adjustable transmission element 34 located in the source line, eg in the form of a series resistor implemented as an adjustable electronic element 32, a downstream low-loss filter circuit 3 and an effective resistor 5 and control amplifier 33 on the output side.

Fig. 2:

• a) Electrical equivalent circuit diagram of an active broadband receiving antenna according to the invention with series noise source u _r and negligible in effect Parallelrauschstromquelle i _r a field effect transistor as a three-pole amplifying element 2 with an outside of the transmission range on the input side high-impedance low-loss filter circuit 3rd
• b) Electrical equivalent circuit diagram of a prior art active broadband receiving antenna with noise matching network and frequency dependent effective length of the passive antenna part 1 at the connection point of the transistor and output matching network for smoothing the frequency response.

3:

Active broadband receiving antenna according to Fig. 1 However, with an adjustable transmission member 34 having a plurality of series-connected resistors 35 each having a resistance 35 in parallel and designed as a switching diode 36 adjustable electronic element 34 for lowering the reception level in stages.

4:

Active broadband receiving antenna as in the FIGS. 1 and 3 However, with an adjustable transmission member 34 of a transformer 38 with available in stages gear ratio (t), switching diodes 36 as adjustable electronic elements 32 for setting a large transmission ratio (t) and thus a large ratio of the input voltage U _E to the output voltage U _A at large Reception levels.

Fig. 5:

Active broadband receiving antenna as in the FIGS. 1 . 3 and 4 but with an adjustable longitudinal adjustable element 30 as a frequency-dependent dipole 47 with a similar to the input admittance 7 of the low-loss filter circuit 3, but substantially with a two-pole mean 46 smaller by a frequency-independent factor (t-1) than the Eingangsadmittanz 7 of the low-loss filter circuit 3 with a frequency-dependent two-terminal 47 connected in parallel switching diode 36th

Fig. 6:

Active broadband receiving antenna as in Fig. 4 with amplifier unit 11 with the noise figure F _v as a further circuit; Design of the real part G of the effective at small receive levels admittance 7 as sufficiently large, so that the noise contribution of the amplifier unit 11 is smaller than the noise contribution of the field effect transistor. 2

Fig. 7:

Active broadband receiving antenna as in Fig. 2a with a plurality of low-loss filter circuits, which are controlled via switching diodes 36 alternatively between the input and the output of the transmission network 31 for alternative reduction of the internal gain of the active antenna.

Fig. 8:

Active broadband receiving antenna as in Fig. 6 but with a filter circuit 3 with fixed blind elements 20 and with dummy elements 20a, which are switched on and off by means of adjustable electronic elements 32 for lowering the internal gain.

Fig. 9:

1. a) aus einem Eingangs-Feldeffekttransistoi 13 und einem Bipolartransistor 14 in Emitterfolgerschaltung
2. b) aus einem Eingangs-Bipolartransistor 49 und einem weiteren Bipolartransistor 50 in Emitterfolgerschaltung
3. c) aus einem Eingangstransistor und einem weiteren Transistor zur hochfrequenten Nachführung der Drain- bzw. der Kollektorelektrode des Eingangstransistors.
4. d) aus einer kombinierten Transistorschaltung zur elektronischen Steuerung der Ruhespannungsquelle U_D0 45 und des Ruhestroms I_S0 50 des Eingangstransistors im Zusammenhang mit der Absenkung der inneren Verstärkung der aktiven Antenne aufgrund zu hoher Empfangspegel.

Design of the three-pole reinforcing element 2 as an extended three-pole reinforcing element

1. a) from an input field effect transistor 13 and a bipolar transistor 14 in emitter follower circuit
2. b) of an input bipolar transistor 49 and another bipolar transistor 50 in emitter follower circuit
3. c) from an input transistor and a further transistor for high-frequency tracking of the drain and the collector electrode of the input transistor.
4. d) from a combined transistor circuit for the electronic control of the rest voltage source U _D0 45 and the quiescent current I _S0 50 of the input transistor in connection with the lowering of the internal gain of the active antenna due to high reception levels.

Fig. 10:

Passive antenna part 1 with a connection point 18, the two terminals are up against the ground terminal, with a field effect transistor 2a and another field effect transistor 2b and a transformer designed as an isolation transformer 38 with switching diodes 36 for setting the transmission ratio

Fig. 11:

Design of several transmission frequency bands over several separate transmission paths for the respective frequency bands. Each of the transmission paths is assigned an adjustable transmission element 34, 34 'and a control amplifier 33, 33' frequency-selectively.

Fig. 12:

Arrangement as in Fig. 11 but with control amplifiers 33, 33 'selectively driven in the receiver 44 for controlling the adjustable transmission elements 34, 34' in the active antenna.

Fig. 13:

Group antenna for the design of directivity with a passive antenna array 27 with electrical radiation coupling between the connection points 18, which are each connected to an amplifier circuit 21 and a high-frequency line 10 and whose signals are combined in the antenna combiner 22. There is a common control amplifier 33 for monitoring the high-frequency received signal 8 at the antenna output.

Fig. 14:

Scanning diversity antenna system with an arrangement as in Fig. 13 but with electronic switches 25 in place of the antenna combiner 22 and each one spare load resistor 26 for loading the non-switched antenna branches. There is a common control amplifier 33 for monitoring the selected high-frequency received signal.

Fig. 15:

Scanning diversity antenna system formed from printed on the window heating fields with diversity moderately positioned junction 18 to achieve diversity Independent received signals 8. There is a common control amplifier 33 in to monitor the selected high-frequency received signal in the electronic switch 25 is present.

Fig. 16:

Scanning diversity antenna system as in Fig. 15 , but with separately determined susceptible values 23 for improving the diversity-related independence of the received signals of the passive antenna parts 1. Each active antenna is assigned a separate control amplifier 33.

Fig. 17:

Active antenna according to the invention, but with a transformer 24 with sufficiently high-impedance primary inductance and sufficiently large gear ratio for broadband increase of the effective length 1 _e .

Fig. 18:

• a) and b): Exemplary antenna configurations of possible passive antenna parts 1
• c) impedance profiles of the antenna structures A1, A2 and A3 in the impedance plane in the frequency range of 76 to 108 MHz and hatched areas for R _A <R _amine and R _A > R _Amax
• d) Real parts of the antenna impedances according to c) with permissible value range R _Amin <R _A <R _Amax

Fig. 19:

1. a) course of the series reactances X ₁ and X ₃ and the parallel susceptance B _{2 of} the T-filter arrangement according to the invention in Fig. 6b over the frequency with the example of the broadband coverage of the radio areas VHF radio and VHF and UHF-Femsehrundfunk.
2. b) Electrical equivalent circuit diagram of an antenna according to the invention for the frequency ranges mentioned under a).

In Fig. 1 an antenna according to the basic form of the invention is shown. By way of example, the heating field of a motor vehicle printed on a window pane shows that the passive antenna part 1 can not be shaped in such a way that it is suitable for use As antenna in the meter and Dezimeterwellenbereich has particular desired properties and thus has a random frequency dependence of both the effective length I _e and their impedance according to their geometric structure and the metallic border of the window. The essence of the present invention is to realize an active antenna, which allows to absorb this randomness of the frequency dependence of the given passive antenna part 1 with the help of a low-cost, easy to detect and easy to implement active antenna and with respect to noise, linearity and Frequency response to make free and between the incident wave with the electric field strength E and the high-frequency received signal 8 to achieve a predetermined frequency response. According to the present invention, the receive voltage present at a connection point 18 is fed to the amplifier circuit 21, which contains at the input a three-pole amplifying element 2, preferably an element with the character of a field effect transistor 2, which is negative-feedback in its source line to the input admittance 7 of a low-loss filter circuit 3, which is completed at its output with an effective resistance 5. In an antenna of this type, the input admittance 7 according to the invention, for example, to be designed such that the strong frequency dependence, which is the receive idle voltage, expressed by the effective length l _{e of} the thus designed passive antenna part 1 in the high-frequency received signal 8 is largely balanced. In order to reduce the reception level in the range of excessive reception field strengths, an adjustable longitudinal element 30 is provided in the adjustable transmission element 34, which acts as a through-connection in the range of small reception levels. If the longitudinal element 30 is set to high impedance in the region of too high a reception level, it causes on the one hand the lowering of the high-frequency received signal 8 as well as an increase in the impedance acting countercurrently in the source line of the transistor or a reduction of the admittance 7 'present there. Thus, the field effect transistor 2 is linearized by the measure and protects the continuative circuit from excessive reception levels.

zu erfüllen. Moderne Gallium-Arsenid-Transistoren besitzen im Vergleich zur übrigen Beschaltung vernachlässigbare Kapazitäten C₁ und C₂ und eine im Hinblick auf die vorgesehene Anwendung vernachlässigbare Wirkung von i_r als Ursache für die bei Rauschanpassung solcher Transstoren extrem kleinen Rauschtemperatur T_N0. Der äquivalente Rauschwiderstand ist vom Ruhestrom abhängig und kann oberhalb 30 MHz breitbandig mit 30 Ohm und weniger angesetzt werden. Für das Beispiel einer Antenne für den UKW-Frequenzbereich und einer dort vorherrschende Antennentemperatur von ca. 1000 K ist somit im Hinblick auf die Rauschempfindlichkeit für den Realteil der komplexen Antennenimpedanz, welcher bei verlustarmem Feldeffekttransistor 2 den Strahlungswiderstand darstellt, innerhalb des Übertragungsfrequenzbereichs ausschließlich R_A(f) > ca. 10 Ohm als hinreichende Bedingung zu fordern.The operation and the design principle of an antenna according to the invention are based on the electrical equivalent circuit diagrams of FIGS. 2a and 5 explains:
The suitability of a predefined passive antenna part 1 for the design of a sufficiently noise-sensitive active antenna can be estimated on the basis of the antenna temperature prevailing in the transmission frequency range. Field effect transistors usually have an extremely small parallel noise _{current source} i _r so that their contribution i _r * Z _A at negligible small gate-source and gate-drain capacitances C ₂ and C ₁ and the occurring in practice antenna impedances Z _A compared to the series noise _{voltage source} u _{r of} the field effect transistor, expressed by its equivalent noise _resistance R _f , is always negligibly small. The sensitivity requirement is thus reduced to the fact that the noise voltage source u _r ² = 4kT _o BR _AF in relation to the received noise voltage source u _rA ² = 4kT _A BR _A , which is given by the antenna temperature T _A and the real part R _{A of} the antenna impedance Z _A is smaller or at most the same size. With equally large noise contributions, the sufficient sensitivity criterion for negligible capacitances C ₁ , C ₂ is therefore merely the requirement that is easy to test $${\mathrm{R}}_{\mathrm{A}}\mathrm{>}{\mathrm{R}}_{\ddot{\mathrm{a}}\mathrm{F}}\mathrm{*}{\mathrm{<figure-callout\; id="0"\; label="T"\; filenames="imgf0007.png,imgf0008.png"\; state="\{\{state\}\}">T</figure-callout>}}_{\mathrm{0}}\mathrm{/}{\mathrm{T}}_{\mathrm{A}}$$

to fulfill. Modern gallium arsenide transistors have negligible capacitances C ₁ and C ₂ in comparison with the other circuits and a negligible effect of i _r as a cause for the noise temperature T _N0, which is extremely small in the case of noise adaptation of such transistors. The equivalent noise resistance depends on the quiescent current and can be set above 30 MHz broadband with 30 ohms and less. For the example of an antenna for the FM frequency range and a prevailing antenna temperature of about 1000 K is thus in terms of noise sensitivity for the real part of the complex antenna impedance, which represents the radiation resistance at low-loss field effect transistor 2, within the transmission frequency range exclusively R _A ( f)> about 10 ohms as sufficient condition to demand.

In Fig. 5 the noise contribution of an amplifier unit 11 is considered at the end of the high-frequency line 10 connected to the low-loss filter circuit 3 on the output side. With sufficient gain in the amplifier circuit 21, this contribution is kept correspondingly small. To protect the downstream amplifier unit 11 against non-linear effects, it is often necessary to make this gain largely frequency-independent within the transmission frequency range. This is achieved by corresponding preferably lossless transformation of the effective effective resistance 5 at the output of the low-loss filter circuit 3 into a suitably frequency-dependent input admittance 7. Is the frequency dependence required due to the frequency dependency of the effective length l _e (f) for the input admittance 7 As is known, a circuit of reactances for the low-loss filter circuit 3 can be found, which largely corresponds to this requirement.

The inventive criterion for the exemplary design of a necessary and frequency-independent reception power within the transmission frequency range for the terrestrial broadcasting reception of an active vehicle antenna with respect to the reception power in the downstream receiving arrangement on the basis of Figure 5 explained. The largely frequency-independent reception behavior is to be demanded, on the one hand not to significantly reduce the sensitivity of the overall system by the noise contribution of the active antenna downstream receiving system and on the other hand to avoid non-linear effects due to gain peaks as a result of frequency-dependent reception behavior within a transmission frequency range. The active antenna downstream receiving system is Figure 5 represented by the amplifier unit 11 with the noise figure F _v . Its noise contribution to the total noise is in Figure 5 shown as the equivalent noise _resistance R _AV at the input of the amplifier circuit 21, where: $$R_{\ddot{a}V}=\frac{\left(F_V-1\right)}{4\cdot G\left(f\right)}$$

Herein, G (f) denotes the frequency-dependent real part of the input admittance 7 of the low-loss filter circuit 3. This noise contribution is then insignificant with respect to the inevitable received sound of the rushing with T _A R _A, if: $$G\left(f\right)\succ \frac{(F_V-1)\cdot {\mathrm{<figure-callout\; id="0"\; label="T"\; filenames="imgf0007.png,imgf0008.png"\; state="\{\{state\}\}">T</figure-callout>}}_0}{4\cdot T_A}\cdot \frac{1}{R_A\left(f\right)}$$

In order to satisfy the sensitivity condition, in an advantageous embodiment of an active antenna according to the invention, the frequency dependence of the real part G (f) of the input admittance 7 of the low-loss filter circuit 3 is reciprocal to the frequency response of the real part R _A (f) of the complex antenna impedance. For the example of an FM radio receiver with F _V ~ 4, G (f) <1 / (3 * R _A (f)) would have to be chosen approximately. On the other hand, in order to protect the receiver from excessive reception levels, it is desirable not to select the power amplification of the active antenna substantially greater than the optimum overall system sensitivity, and thus G (f) approximately as indicated in the right part of equation (3).

The great advantage associated with the invention is that the frequency response given for G (f) from R _A (f) can therefore be easily fulfilled because neither the input impedance of the low-impedance filter circuit 3, which is given by 1 / g _{m of} the field-effect transistor 2 is still the effective effective resistance 5 at the output of the low-loss filter circuit 3 unavoidable essential reactive components have. This results in the advantageously free configurability of the frequency response of the active antenna according to the present invention. In contrast, in an active antenna according to the prior art in Fig. 2b the frequency-dependent radiator impedance Z _S (f) exists forcibly and inseparably as source impedance of the primary-side transformation network. Their frequency behavior limits the achievable bandwidth of the impedance transformed into the vicinity of Z _opt and thus the bandwidth of the signal-to-noise ratio at the output of the active circuit.

bzw. für die Referenzantenne bei der Bezugsfrequenz: $$D_{\mathit{pm}}=\frac{1}{N}{\displaystyle \sum _{n=1}^N}{D}_p\left({\mathrm{\Phi }}_n,,\Theta \right)$$

In the following, the exemplary configuration of the frequency characteristic of G (f) of an active vehicle antenna according to the invention is described, if it is required that the received power P _a at the input of the receiving system connected downstream of the active antenna is greater by a factor V than with a passive reference antenna , For example, a passive rod antenna on the vehicle at their resonance length. Due to the forcibly different directional diagrams, this factor is based on the azimuthal average values at a defined constant elevation angle θ of the wave incidence. Comparative azimuthal directivity measurements using an antenna measuring path with vehicle rotation on the passive antenna part 1 and on the comparison antenna result in N angular steps for a full revolution and with the directivity D _a (φ _n , θ) of the given passive antenna part 1 and according to the directivity factor D _p (φ _n , 0) of the passive reference antenna in each case for the n-th angular step, the following azimuthal mean values for the directivity factors: $$D_{\mathit{at\; the}}\left(f\right)=\frac{1}{N}{\displaystyle \underset{n=1}{\overset{N}{\Sigma }}}{D}_a\left({\mathrm{\Phi }}_n,,\mathrm{\Theta },,f\right)$$

or for the reference antenna at the reference frequency: $$D_{\mathit{pm}}=\frac{1}{N}{\displaystyle \underset{n=1}{\overset{N}{\Sigma }}}{D}_p\left({\mathrm{\Phi }}_n,,\Theta \right)$$

wobei l_em ²(f) den bei jeder Frequenz auftretenden azimutalen Mittelwert der quadratischen fektiven Länge des passiven Antennenteils 1 unter Berücksichtigung der sich mit D_am(f) gem. Gleichung (2) ergebenden effektiven Fläche des passiven Antennenteils 1 wie folgt darstellt: $${l_{\mathit{em}}}^2\left(f\right)=\frac{1}{N}{\displaystyle \sum _{n=1}^N}{l_{\mathit{en}}}^2\left(f\right)=\frac{{\mathit{\lambda }}^2}{\mathit{\pi }}\cdot \frac{R_A\left(f\right)}{Z_0}\cdot D_{\mathit{am}}\left(f\right)$$

The active antenna downstream receiving system, which in Fig. 5 is represented by the amplifier unit 11, is usually based on the line impedance Z _{L of} the high-frequency line system. The average azimuthal reception power in the load resistor 9 results in a sufficiently large slope g _{m of} the input characteristic of the field effect transistor 2 to: $$P_{\mathit{at\; the}}=\frac{1}{2}\cdot {\mathrm{<figure-callout\; id="2"\; label="e"\; filenames="imgf0003.png,imgf0005.png"\; state="\{\{state\}\}">e</figure-callout>}}^2\cdot {l_{\mathit{em}}}^2\left(f\right)\cdot G\left(f\right)$$

where l _em ² (f) is the azimuthal average value of the quadratic fective length of the passive antenna part 1 occurring at each frequency, taking into account with D _am (f) according to FIG. Equation (2) resulting in the passive antenna part 1 as follows: $${l_{\mathit{em}}}^2\left(f\right)=\frac{1}{N}{\displaystyle \underset{n=1}{\overset{N}{\Sigma }}}{l_{\mathit{s}}}^2\left(f\right)=\frac{{\mathit{<figure-callout\; id="2"\; label="\lambda "\; filenames="imgf0003.png,imgf0005.png"\; state="\{\{state\}\}">\lambda </figure-callout>}}^2}{\mathit{\pi }}\cdot \frac{R_{\mathrm{<figure-callout\; id="0"\; label="A\; f\; Z"\; filenames="imgf0007.png,imgf0008.png"\; state="\{\{state\}\}">A\; </figure-callout>}}\left(f\right)}{Z_{}}\cdot D_{\mathit{at\; the}}\left(f\right)$$

The mean azimuthal reception power of the passive reference antenna with D _{pm is} from equation (5): $$P_{\mathit{pm}}=\frac{{\mathit{<figure-callout\; id="2"\; label="\lambda "\; filenames="imgf0003.png,imgf0005.png"\; state="\{\{state\}\}">\lambda </figure-callout>}}^2}{\mathrm{8th}\cdot \mathit{\pi }}\cdot \frac{e^2}{{\mathrm{<figure-callout\; id="0"\; label="Z"\; filenames="imgf0007.png,imgf0008.png"\; state="\{\{state\}\}">Z</figure-callout>}}_0}\cdot D_{\mathit{pm}}$$

Taking into account the amplification requirement P _am / P _pm = V, the frequency profile for G (f) to be demanded according to the invention results in: $$G\left(f\right)=\frac{1}{R_A\left(f\right)}\cdot \frac{{\mathit{D}}_{\mathit{pm}}}{D_{\mathit{at\; the}}\left(f\right)}\cdot V$$

Η is the case of a lossy passive antenna part 1 to the Efficiency in equation (8) of the directivity factor D η to replace _at (f) by _the D (f) *. The remaining sizing rules are not changed.

gilt, dann ist der Rauschbeitrag des der aktiven Antenne nachgeschalteten Empfangssystems zum Gesamtrauschen vernachlässigbar klein. Ist zusätzlich die in Gleichung (1) angegebene Bedingung erfüllt, dann ist die Empfindlichkeit ausschließlich durch die Richtwirkung des passiven Antennenteils 1 und von der herrschenden Störeinstrahlung abhängig. Die minimal notwendige mittlere azimutale Strahlungsdichte S_am für ein Signal-Störverhältnis = 1 lautet dann: $$S_{\mathit{am}}\left(f\right)=\frac{k\cdot T_A\cdot B}{D_{\mathit{am}}\left(f\right)}\cdot \frac{4\cdot \mathit{\pi }}{{\mathit{\lambda }}^2}$$

und steigt mit 1/η an, wenn D_am(f) durch D_am(f)* η zu ersetzen ist.For the case of approximately the same azimuthal mean values D _pm and D _am (f), the frequency dependence of G (f) is to be made proportional to 1 / R _a (f). Is V chosen so large that $$\frac{{\mathit{D}}_{\mathit{pm}}}{D_{\mathit{at\; the}}\left(f\right)}\cdot V»\frac{(F_V-1)\cdot {\mathrm{<figure-callout\; id="0"\; label="T"\; filenames="imgf0007.png,imgf0008.png"\; state="\{\{state\}\}">T</figure-callout>}}_0}{4\cdot T_A}$$

is true, then the noise contribution of the active antenna downstream receiving system to the total noise is negligible. If, in addition, the condition given in equation (1) is satisfied, then the sensitivity depends exclusively on the directivity of the passive antenna part 1 and on the prevailing interference radiation. The minimum necessary average azimuthal radiation density S _am for a signal-to-noise ratio = 1 is then: $$S_{\mathit{at\; the}}\left(f\right)=\frac{k\cdot T_A\cdot B}{D_{\mathit{at\; the}}\left(f\right)}\cdot \frac{4\cdot \mathit{<figure-callout\; id="2"\; label="\pi \; \lambda "\; filenames="imgf0003.png,imgf0005.png"\; state="\{\{state\}\}">\pi \; </figure-callout>}}{{\mathit{\lambda }}^{}}$$

and increases with 1 / η, if D _am (f) is replaced by D _am (f) * η.

Taking into account the interfering radiation emanating from the vehicle itself, the selection of a passive antenna part 1 suitable for an antenna according to the invention as a vehicle-mounted structure in conjunction with the condition for R _A (f) given in equation (1) and discussed in more detail below can therefore be made accurately take place that the ratio T _A / D _am (f) for the transmission frequency range is determined to be sufficiently large.

In Figs. 18a and 18b For example, antenna configurations of possible passive antenna parts 1 of active antennas according to the invention are given. At the connection points 18 are in the complex impedance level in Fig.18c illustrated impedance curves Z _A (f) as a function of the frequency before. The area marked by hatching in the left margin of the diagram is one-sided by the value R _Amin = const. bounded. Impedance curves that run outside of the area marked in this way thus fulfill the condition of the negligible noise of the field effect transistor 2 specified in equation (1) in the presence of a specific interference radiation according to T _A. The diagram convincingly shows the advantage of an active antenna according to the invention over an active antenna according to FIG Fig. 2b in the prior art, which is that without input side matching means all antenna structures meet this condition without input-side transformation means. In the Fig. 18c are the real parts of in FIGS. 18a and b represented passive antenna parts 1 on the frequency of 76 to 108 MHz applied. The frequency response of the real part of the invention to be designed input admittance 7 at the input of the low-loss filter circuit 3 is therefore inverted to those in Fig. 18d shown curves according to aspects as they were discussed in connection with the equations (3) and (8) to make.

In the case of the amplifier circuit 21 according to the invention, due to possible nonlinear effects, such as intermodulation, there is naturally also an upper limit to the magnitude of the tolerable voltage which is effective at the input and which results in the reception field over the effective length l _e . The maximum tolerable voltage can be increased by selecting a suitable field effect transistor 2 and by selecting a suitable operating point and by other circuit measures known per se. According to the invention, equation (6) can be assigned to a maximum tolerable azimuthal mean value l _em with a known azimuthal directivity factor D _am (f) a maximum tolerable active component R _Amax . The value range with R _A > R _{Amax that} is not allowed for sizing is in the Figures 18c and 18d also marked hatched. The radiation resistances R _{A of} the impedance values of particularly favorable structures for use as a passive antenna part 1 are therefore outside the hatched value range with R _Amin <R _A <R _Amax .

In a further advantageous embodiment of the invention, a predetermined antenna structure by using a low-loss transformer with the transmission ratio ü, as in Fig. 17 specified, added, which forms the passive antenna part 1 together with the antenna structure - eg a heating field on the window pane. The broadband transmission ratio is advantageously selected such that the impedance measurable at the output of the transformer is placed with its real part in the value range with R _Amin <R _A <R _Amax . It is advantageous here to make the primary inductance sufficiently high-impedance.

The linearity requirement is met by a sufficiently large negative feedback through the input admittance 7 located in the source line. This requires a comparatively low negative feedback in the transmission range, which is dimensioned in accordance with the amplification requirement, for example, according to equation (8), but which is as large as possible outside the transmission range. In an advantageous embodiment of the invention, preferably T-half filters or T-filters or chain circuits of such filters are used to implement such low-loss filter circuits 3. Such filters are shown in their basic structure in the figures. To correspond to a more complicated frequency response of the G (f), the individual elements be supplemented by other reactive elements. In the interest of the input-side high impedance and the blocking effect in the stopband, it is expedient to form series or parallel branch each from a combination of reactances such that both the absolute value of a reactance in series branch 28 and the absolute value of a susceptance in parallel branch 29 each within a Transmission frequency range is sufficiently small and outside such a sufficiently large ( Fig. 19b ).

In a further advantageous application of the invention, it is proposed to store corresponding basic structures for low-loss filter circuits 3 with initially unknown values for the reactive elements in a modern digital computer for different characteristic courses of G (f) and to measure both the impedance Z _{A of} the passive antenna part 1 and to determine the azimuthal average D _on the directivity by measurement or calculation and also be saved in the digital computer. The frequency profile of G (f) thus determined using equation (8) enables the subsequent concrete determination of the reactive elements of the low-loss filter circuit 3 for a suitably selected basic filter structure using known variational calculation strategies for the predetermined amplification V of the active antenna.

Particularly in those antenna systems in which a plurality of antennas are formed, such as in antenna diversity systems, group antenna systems or multi-range antenna systems, it is helpful in a vorteihaften development of the invention, as in Fig. 6 indicated to make the amplifier unit 11 as the active output stage of the amplifier circuit 21. This can be provided with an output resistance equal to the characteristic impedance Z _{L of} conventional coaxial cables. The effective effective resistance 5 is formed by the input impedance of the amplifier unit 11. G (f) is to be designed analogously to the above-mentioned embodiments with the aid of a low-loss filter circuit 3 completed with this impedance.

Due to the inefficiency of the adjustable transmission element 34 in the case of low reception levels, this sensitivity analysis is not impaired. The voltage reduction after the first amplifying element of the active antenna is advantageous in particular because it allows an optimum effect with regard to the frequency dependence of the expected intermodulation interference. The influence on the sensitivity of the entire receiving system is thus determined only by the influence of the increased voltage to the voltage reduction noise figure of the subsequent circuit.

In the following, different forms of lowering the internal amplification of the active antenna are compared. In the FIGS. 1 . 2a and 3 the voltage reduction takes place via a longitudinal element 30, which is frequency independent. As a result, receive signals at frequencies in which low-impedance real parts of the antenna impedances are present and accordingly large values of the input admittance G (f) are designed according to the invention are attenuated more strongly than receive signals at frequencies with a high-impedance real part of the antenna impedances. When using a frequency-independent longitudinal element 30, a mean resistance value must therefore be selected for the reduction at high reception levels, which is too small for intermodulation received signals at frequencies with a large real part of the antenna impedances and too large at frequencies with a small real part of the antenna impedances. This involves the risk that intermodulation received signals at frequencies with a large real part of the antenna impedances due to the smaller negative feedback effect cause too much intermodulation and on the other hand, the remaining gain at frequencies with a small real part of the antenna impedance is too small and the arrangement is too insensitive at these frequencies.

In an advantageous embodiment of the invention, therefore, such forms of adjustable transmission elements 34 are proposed, which reduce the admittances set at small reception levels 7 independent of frequency by a suitable factor. In the amplifier components available today, for example, for the VHF range and an application in motor vehicles, a level reduction between 20 * log (t) = 10 dB and 20 * log (t) = 20dB is practicable. As a result, the internal gain of the active antenna is frequency-independently reduced by a desired factor and the above-mentioned frequency-dependent intermodulation effect does not occur. According to the invention, this is achieved for example by a transformer arrangement as in Fig. 4 and in Fig. 6 reached. For this purpose, the frequency-independent transmission ratio of the transformer with the help of divided windings and the illustrated switching diodes 36 designed as adjustable electronic elements 32 in stages adjustable. If the gear ratios are selected correctly, the appropriate values for the conductance G (f) in the admittance 7 or 7 'can be selected for the range of low or high reception levels. For increasing the linearity and the current driving range of the three-pole amplifying element 2, see FIG Fig. 6 provided to raise the quiescent current in this element together with the lowering of the internal gain of the active antenna.

Another method for achieving a frequency-independent negative feedback is by the arrangement in Fig. 5 given. Here, the frequency-independent lowering of the high-frequency received signals 8, the adjustable longitudinal element 30 is designed as a frequency-dependent dipole 47. This is similar to a Eingangsadmittanz 7 of the low-loss filter circuit 3, but essentially with a frequency-independent factor t-1 smaller Zweipoladmittanz 46 than the Eingangsadmittanz 7 of the transmission network 31 at low receive levels. By parallel connection of a switching diode 36 to the frequency-dependent two-terminal 47, the two-pole admittance 46 is effective by the setting in the off state and the two-pole admittance 46 is bridged in their setting in Durchlaßzustand takes place at locked switching diode 36 is a lowering of the high-frequency received signals 8 to a substantially frequency-independent factor t = U _E / U _A.

In a further advantageous embodiment of the invention, the transmission network 31 with filter character in Fig. 8 designed as a low-loss filter circuit 3 with fixed blind elements 20. In this case, switchable dummy elements 20a are used which are switched on and off with the aid of adjustable electronic elements 32 such that falls below a predetermined receive level, the desired frequency dependence of the greater conductance G (f) of effective at the source terminal 24 Eingangsadmittanz 7 for higher internal gain of the active Antenna is given on the one hand. On the other hand, when exceeding a predetermined reception level, the desired frequency dependence of the correspondingly reduced frequency conductance G '(f) is set with the same frequency dependence of the input admittance 7' for the lowered internal amplification of the active antenna.

In the transmission network 31 in the advantageous arrangement in Fig. 7 There are a plurality of low-loss filter circuits 3, 3a present, which are connected via switching diodes 36 alternatively between the input and the output of the transmission network 31. Their Eingangsadmittanzen 7, 7b for small reception levels or 7 ', 7b' for large reception levels are each formed with fixed blind elements 20 such that with the help of the switching diodes 36 falls below a predetermined reception level, the desired frequency dependence of the conductance G (f) of the effective higher internal gain input admittance 7 effective at the source terminal 24, and the desired frequency dependence of the conductance G '(f) of the active antenna lower input gain 7' active at the source terminal 24 are applied to a given receive level is.

At the in Fig. 10 illustrated embodiment of an active antenna according to the invention, the passive antenna part 1 is designed with a connection point 18, whose two terminals are up against the mass 0. Each of the two terminals is connected to a respective control terminal 15a or 15b of a three-pole amplifying element 2. The source terminals 24a and 24b are connected to the primary side of a transformer 38 designed as an isolating transformer, the secondary side of which has different outputs for forming different transformation ratios t. The adjustable transmission member 34 is thus formed from the transformer and the switching diodes 36. The drain terminals 53a and 53b of the three-pole amplifying elements 2a and 2b are connected to the ground O, respectively.

In a further advantageous embodiment of the invention, the three-pole reinforcing element 2, as in Fig. 9a , designed as an extended three-pole reinforcing element. To increase the effective transconductance of the transmission line, the extended element of an input field effect transistor 13, whose source of the bipolar transistor 14 is driven in the emitter follower circuit and the emitter terminal 12, the source electrode of the extended three-pole amplifying element 2 is formed combined.

In a further advantageous embodiment, the three-pole reinforcing element 2 is in Fig. 9b combined as an extended three-pole amplifying element of an input bipolar transistor 49 and another bipolar transistor 50 in emitter follower circuit. The emitter terminal 12 of the bipolar transistor 50 forms the source terminal 24 of the three-pole amplifying element 2. When set sufficiently low quiescent current in the input bipolar transistor 49, the required high impedance at low input capacitance and a sufficiently low parallel noise current is achieved. A much larger adjusted quiescent current in the further bipolar transistor 50 causes a sufficiently large slope of the transfer characteristic for the entire element.

In Fig. 9c is the three-pole amplifying element 2 designed as an extended three-pole amplifying element of an input bipolar transistor 49 and input field effect transistor 13, whose collector terminal or drain terminal to the source or. Emitter terminal of an additional transistor 51 is connected and whose base or gate terminal is connected to the emitter or source terminal of the input bipolar transistor 49 and input field effect transistor 13, respectively. Through this connection, the source terminal 24 of the three-pole amplifying element 2 is formed. An extended three-pole amplifying element of this form prevents by voltage tracking at the drain or collector terminal of the input transistor, the disturbing influence of a voltage-dependent capacitance between the control electrode and the drain and collector electrode.

In Fig. 9d the three-pole amplifying element 2 is designed as an extended three-pole amplifying element, in which an electronically controllable quiescent current source I _S0 or / and an electronically controllable rest voltage source U _{D0 is} present. As a result, the quiescent current I _S0 or / and the quiescent voltage U _D0 in the input bipolar transistor 49 or input field-effect transistor 13 are set increased when large reception levels occur in connection with the inventive lowering of the internal amplification of the active antenna.

For the design of several transmission frequency bands are in Fig. 11 a plurality of bipolar transistors 14, 14 'for expanding the three-pole amplifying element 2 and for the combined formation of a plurality of three-pole amplifying elements 2, 2'. The base electrodes are connected to the source of a common input transistor 13 and to the source of an extended three-pole amplifying element, respectively FIGS. 9a to 9d connected. The bipolar transistors 14,14 'are each connected in emitter follower circuit to the input of a low-loss filter circuit 3,3' to form separate transmission paths for the respective frequency bands. In each of the transmission paths are each an adjustable transmission element 34,34 'and a control amplifier 33,33', which is supplied via filter measures only the respective transmission path associated frequency band from the high-frequency received signal 8 respectively. The control signal 42, 42 'is in each case the associated adjustable transmission element 34,34' supplied. In contrast to this are in Fig. 12 the control signals 42, 42 'by means of selection and control amplifier 33, 33' in Receiver 44 derived from the output signal of the active antenna and the active antenna via control lines 41 supplied.

In a particularly advantageous embodiment of the invention, the present active antenna is used repeatedly in an antenna system whose passive antenna parts 1 with frequency-dependent and with respect to incident waves by amount and or only in phase different directional diagrams of the effective lengths l _e possess, however, in electromagnetic Radiation coupling to each other and together form a passive antenna array 27 with multiple connection points 18a, b, c. According to the invention, each is connected in each case to an amplifier circuit 21 according to the invention and supplemented to form an active antenna according to the invention. Due to the high impedance of the amplifier inputs is given by the coupling of the high-frequency received signals 8 to the passive antenna parts 1 no significant mutual influence of the receiving voltages. Such an antenna arrangement is generally in Fig. 13 shown. The present at the output of the amplifier circuit 21 receive signals 8 are superimposed weighted for the design of a group antenna arrangement with predetermined receiving properties with respect to directivity and antenna gain without retroactivity to the voltage applied to the passive antenna parts 1 high-frequency reception signals in a Antennencombiner 22 present for this purpose. There may advantageously a common control amplifier 33, the control signals 42a, b, c the transmission networks 31 a, b, c is fed into the active antennas for lowering the summed high-frequency received signal 8, perform the level monitoring. In a further advantageous embodiment of such a group antenna arrangement, the level monitoring and attenuation take place separately in each active antenna with the aid of a control amplifier 33 respectively accommodated there.

When using an antenna according to the invention as an active window pane antenna, it is advantageously possible to accommodate the amplifier circuit 21 invisible in the very narrow edge area of the vehicle window. Therefore, it is desirable to miniaturize the part to be attached to the connection point 18 and to attach only the parts of the amplifier circuit 21 which are functionally necessary there. The other parts of the low-loss filter circuit 3 are placed remotely and turned on via the high-frequency line 10.

In a further advantageous embodiment of the invention, the active antenna is designed as a multi-region antenna for several frequency ranges. For this purpose are in Fig. 19a for the frequency ranges FM radio broadcasting and VHF and UHF television broadcasting the fundamental frequency characteristics of reactances X ₁ , X ₃ and the susceptibility B _{2 of} a T-filter arrangement of the in Fig. 19b specified low-loss filter circuit 3 exemplified. The T-filter configuration in this case ensures the input-side high-impedance of the low-loss filter circuit 3 to achieve a sufficiently large negative feedback of the field effect transistor 2 in the stopband areas. The low-loss filter circuit 3 is designed as a T-half filter or T-filter or as a chain circuit such filter whose or series branch or parallel branch is formed in each case from a combination of reactances such that both the absolute value of a reactance in the series branch 28 than Also, the absolute value of a susceptance in the parallel branch 29 each within a transmission frequency range sufficiently small and outside such is sufficiently large and the high-frequency received signal 8 is supplied to the control amplifier 33 at the output and from the control signal 42, the adjustable transmission element 34 is controlled.

In order to compensate for effects of even-numbered non-linearity and the resulting interband frequency conversions in the amplifier circuit 21, another field effect transistor 2 having the same electrical properties is used in a further advantageous embodiment of the invention in addition to the field effect transistor 2. Here, the input terminals of the amplifier circuit 21 are formed by the two control terminals of the field effect transistors 15a and 15b, and the input of the low-loss filter circuit 3 is connected to the source terminals 19a and 19b. A Umsymmetrierglied in the low-loss filter circuit 3 is used for the Umsymmetrierung the high-frequency received signals 8. Such. Circuit can advantageously also be connected to a connection point 18 with two connections leading to ground voltage.

The efficiency of antenna diversity systems is determined by the number of available, mutually independent antenna signals. This independence is reflected in the correlation factor between the received voltages occurring in a Rayleigh wave field during travel. In a particularly advantageous embodiment of the invention, a plurality of active receiving antennas according to the invention in an antenna diversity system for Vehicles used, wherein the passive antenna parts 1 are selected such that their present in a Rayleigh reception field idle at the connection points 18 received signals E * l _{e are} as independent of each other in terms of diversity. Such systems, in which the connection points 18 are selected from this point of view and taking into account technical aspects of the vehicle, are exemplary in the FIGS. 15 and 16 shown. Due to the electromagnetic radiation couplings existing between the connection points 18, this independence then applies only to the connection points 18 operated at idling. By wiring the connection points 18 with the amplifier circuits 21 according to the invention, the high-frequency received signals 8 are tapped at the antenna outputs due to their negligible capacitive input conductance. The diversity of the independence of the received signals at the connection points 18 is thus not affected by this measure in an advantageous manner and this independence is therefore in the same way for the received signals 8 at the antenna outputs. Thus, mutually independent receive signals 8 are available at the antenna outputs for selection in a scanning diversity system or for further processing in one of the other known diversity methods.

In contrast, the wiring of pad 18 would be implemented with a prior art transform circuit Fig. 2b cause a dependence of the antenna signals on the antenna output via the currents flowing at the connection point 18. This relationship is explained in more detail below for a passive antenna part 1 with two connection points 18:

Are U01 and U02 the no-load voltage amplitudes at the connection points 18 of a passive antenna arrangement 27 in FIG Fig. 14 In the reception field and Z11, Z22, the antenna impedances measured there are also Z12 and the interaction impedance due to coupling of the connection point 18 and Y1 and Y2 are the input admittances of the amplifiers, with which the connection point 18 are loaded, results for the voltage amplitudes occurring under this load at the connection points 18 the following relationship: $$\begin{array}{l}\left(\begin{array}{c}U1\\ U2\end{array})=\frac{1}{N}\cdot \left(\begin{array}{cc}1-Z22\cdot Y2& Z12\cdot Y2\\ Z12\cdot Y1& 1-Z11\cdot Y1\end{array}\right)\cdot (\begin{array}{c}U10\\ U20\end{array}\right)\\ \mathit{With}\\ N=1-Z11\cdot Y1-Z22\cdot Y2+Z11\cdot Z22\cdot Y1\cdot Y2-Z{12}^2\cdot Y1\cdot Y2\end{array}$$

The correlation factor between the voltage amplitudes U1 and U2, and thus also between the antenna output voltages, results with the aid of the time average values of the voltages U1 and U2: $$\rho =\frac{\widetilde{U1\cdot U2}}{\sqrt{\widetilde{U{1}^2\cdot U{2}^2}}}$$

For the case assumed here, independent idling-receiving voltage amplitudes U10 and U20 result when traveling in the Rayleigh receiving field. This is expressed by a vanishing correlation factor, ie: $$\rho =\frac{\widetilde{U10\cdot U20}}{\sqrt{\widetilde{U{10}^2\cdot U{20}^2}}}=0$$

If the input admittances of the amplifiers with which the connection points 18 are loaded are negligibly small according to the invention, ie Y1 = 0 and Y2 = 0, the voltages U1 and U2 from equation (11) result as follows: $$\left(\begin{array}{c}U1\\ U2\end{array}\right)=\frac{1}{N}\cdot \left(\begin{array}{cc}1& 0\\ 0& 1\end{array}\right)\cdot \left(\begin{array}{c}U10\\ U20\end{array}\right)$$

The interactions in the unit matrix occupied by the number 0 in equation (13) show that the vanishing decorrelation in the voltages U1 and U2 described in equation (13) is retained in an amplifier circuit 21 according to the invention. The evaluation of equation (11), on the other hand, results in a combination of the two no-load voltages via the interaction parameters Z12 * Y2 or Z12 * Y1 with the respective voltages under load, since then: $$\begin{array}{l}U1=\left(1-Z22\cdot Y2\right)\cdot U10+Z12\cdot Y2\cdot U20\\ \mathit{respectively}\mathrm{,}\\ U2=\left(1-Z11\cdot Y1\right)\cdot U20+Z12\cdot Y1\cdot U10\end{array}$$

It will be appreciated that if coupling 18 is not permanently coupled, i. not vanishing Z12, the correlation factor disappears only if Y1 = Y2 = 0.

On the other hand, the foregoing considerations indicate that with the mutual interdependence of the open circuit voltages U10 and U20, specific values for Y1 and Y2 can be found which reduce or eliminate the interdependence in the amplifier input voltages U1 and U2 through the transformation described in Equation 15. In an advantageous embodiment of the invention, it is therefore, as in Fig. 16 indicated, the passive antenna array 27 at their terminals 18 by suitable Leitwerte - preferably susceptors 23 for reasons of noise sensitivity - to connect such that the correlation between the voltages at the connection points 18 in the interest of greater diversity efficiency is smaller. Active antennas according to the invention have the decisive advantage that the definition of such suitable reactive elements can be made largely independent of sensitivity considerations. Because for each of the different connection points 18 resulting radiation resistances R _A (f) in each case no exact balance is required, but it is only to require that they are the in Fig. 18 belong to the permitted range of values described. To lower too high reception level can be as in Fig. 15 the level of the selected signal is supplied to a common control amplifier 33 in the electronic switch 25, in which a control signal 42 is formed and fed to the transmission networks 31 in the amplifier circuits 21 of the active receiving antennas for lowering the selected high-frequency received signal 8. In a further embodiment, as in Fig. 16 In each case, a separate control amplifier 33 for monitoring the high-frequency received signal 8 at the relevant antenna output is assigned to the amplifier circuits 21 of the active antennas.

List of terms Active broadband receiving antenna with receive level control

• Mass 0
• Passive antenna part 1
• three-pole reinforcing element 2
• Low-loss filter circuit 3
• Exit 4
• Effective impedance 5 of the secondary circuit
• Entrance 6
• Input admittance 7
• High-frequency received signal 8
• Load resistor 9
• High frequency line 10
• Amplifier unit 11
• Emitter connection 12
• Input field effect transistor 13
• Bipolar transistor 14
• Control connection 15
• Junction 18
• Fixed blind element 20
• switchable blind element 20a
• Amplifier circuit 21
• Antenna combiner 22
• Susceptance 23
• Source connection 24
• Electronic changeover switch 25
• passive antenna arrangement 27
• Serial branch 28
• Parallel branch 29
• adjustable longitudinal element 30
• Transmission network 31
• adjustable electronic element 32
• Control amplifier 33
• Adjustable transmission member 34
• Resistance 35
• Switching diode 36
• Adjustable resistance 37
• Transmitter 38
• Controllable DC source 40
• Control line 41
• Control signal 42
• Receiver 44
• Controllable DC voltage source 45
• Two pole admittance 46
• frequency-dependent dipole 47
• Two-pole filter 48
• Input bipolar transistor 49
• another bipolar transistor (50
• additional transistor 51
• Drain connection 53

• Input impedance Z
• Noise figure F _v
• Conductivity G
• effective length le
• Wavelength λ
• Boltzmann constant k
• Characteristic impedance of free space Z ₀
• Measuring bandwidth B
• Gear ratio t
• Input voltage U _E
• Output voltage U _A
• Gate-source capacitance C2
• Gate-drain capacitance C1
• Noise temperature TN0
• Ambient temperature T0

Claims (39)

1. Active wide-band receiver antennae consisting of a passive antenna component (1), its output connectors, consisting of a primary connector (18) and a secondary connector (1') are connected with the input connectors of a amplifier circuit (21), whereby the input circuit of the amplifier circuit (21) contains a three-pole amplifier element (2), whose high-impedance control connector (15) is connected to the primary connector (18) of the passive antenna (1), whereby, further, in the connection between the source connector (24) of the three-pole amplifier element (2) and the second connector (1') of the passive antenna element (1) the initial admittance (7) transmission networks (31) with filter character have a degenerating and linearizing effect and the transmission network (31) is loaded with the subsequent circuit at one output (4)
   with the following further features:

   - in the transmission network (31) there is at least one adjustable electronic element (32) for the variable reduction of the reception level,

   - in the case of low-frequency reception signals (8), the transmission network (31) features a low level of loss

   - the linearizing initial admittance (7') of the transmission network (31) is lower if the adjustable electronic element (32) is set for a reduction of the high-frequency receiving signal (8), causing the reduction in the amplification of the active antenna at the output of the three-pole amplifying element 2 in connection with a simultaneously achieved increase in linearizing degeneration and

   - to reduce excessive receiving levels there is a regulating amplifier (33) to which the highfrequency reception signal (8) is fed and whose regulating signal (42) controls an adjustable transfer element (34) (Fig. 1).
2. Active wide-band receiver antenna according to claim 1 characterized by the fact that the transmission network (31), which has the character of a filter when the electronic element (32) or electronic elements (32) is/are set for small high-frequency reception levels, is formed as a low-loss filter circuit that is loaded with the effective impedance (5) of the subsequent circuit at its output (4) and the dummy element of the low-loss transmission network (31) is selected so that the frequency dependency of the conductance G(f) of the effective initial admittance (7) at the input of the transmission network (31) is set in such a way that when amplification is specified for the active antenna, the frequency response resulting from the frequency-dependent effective length l_e of the passive antenna part (1) in the high-frequency receiving signal (8) within a broad frequency band is organized according to freely chosen aspects.
3. Active wide-band receiver antenna according to one of claims 1 to 2, characterized by the fact that the transmission network (31), which has the character of a filter from the chain circuit, consists of an adjustable transmission element (34) and a low-loss filter circuit (3) with a fixed dummy element that is loaded with the effective impedance (5) of the subsequent circuit at its output (4) and the adjustable transmission element (34) when a specified receiving level for non-frequency-dependent and low-loss signal transmission is formed and the dummy elements of the low-loss transmission network (31) are selected so that the frequency dependency of the conductance G(f) of the effective initial admittance (7) at the input of the source connection (24) is set in such a way that when amplification is specified for the active antenna, the frequency response resulting from the frequency-dependent effective length I_e of the passive antenna part (1) in the high-frequency receiving signal (8) within a broad frequency band is organized according to freely chosen aspects. (Fig. 1)
4. Active wide-band receiver antenna according to one of claims 1 to 3 characterized by the fact that the transmission network (31), which has the character of a filter takes the form of a fixed dummy element (20) and that there is at least one connectible dummy element (20a) that can be switched on and off using an adjustable electronic element (32), so that if a specified receiving level is not reached, the required frequency dependency of the conductance G(f) of the effective initial admittance (7) at the input of the source connection (24) is set for higher amplification for the active antenna, and, if a given receiving level is reached, the required frequency dependency of the conductance G(f) of the effective initial admittance (7) at the input of the source connection (24) applies to the reduced amplification of the active antenna. (Fig. 8).
5. Active wide-band receiver antenna according to one of claims 1 to 4 characterized by the fact that the transmission network (31), which has the character of a filter, has a sufficiently small dummy portion b(f) in the initial conductance (7) when a given reception level is not reached and when particular transmission patterns are required to avoid non-linear effects. (Fig. 1).
6. Active wide-band receiver antenna according to one of claims 1 to 5 characterized by the fact that with all settings of the at least one adjustable electronic element (32) the level of the effective degenerating input admittance (7, 7') outside the utilizable frequency band in the stop frequency range of the transmission network (31) connected to the source connector (24) with filter character to avoid non-linear effects with all settings of the settable electronic element (32) or settable electronic elements (32) is sufficiently small.
7. Active wide-band receiver antenna according to one of claims 1 to 6 characterized by the fact that the transmission network (31) is formed from the chain circuit of an adjustable transmission element (34) formed as a transmission block and a low-loss filter circuit (3) and the ration (t:1) of the input voltage (U_E) to the output voltage (U_A) of the adjustable transmission element (34) is set to a sufficient level with the help of a longitudinal element (30) contained there is a specified reception level is exceeded. (Figs. 1, 2a, 3, 4, 5, 6, 10, 11, 12).
8. Active wide-band receiver antenna according to claim 7 characterized by the fact that the adjustable longitudinal element (30) is implemented by an adjustable resistor (37), for example a diode with the character of a PIN diode (Fig. 1).
9. Active wide-band receiver antenna according to claim 5 characterized by the fact that the adjustable longitudinal element (30) is formed by a resistor (35) or several resistors switched in sequence (35) with an adjustable electronic element (32) switched in parallel with the resistor (35) and executed as a switching diode (36), whose setting in the block state renders the associated resistor fully effective and whose setting in the open state bridges the switching diode (36) of the resistor, so that when the switching diode (36) or switch diodes (36) are under control, the reception level drops in increments (Fig. 2a, 3).
10. Active wide-band receiver antenna according to claim 7 characterized by the fact that for the purpose of the non-frequency-dependent reduction of reception signals (8), the adjustable longitudinal element (30) takes the form of a frequency-dependent two-pole network (47) with a two-pole admittance (46) similar to the input admittance of the low-loss filter circuit (3), but mainly smaller than the input admittance of the low-loss filter circuit (3) by a non-frequency-dependent factor (t-1) with a switching diode (36) switched in parallel with the frequency-dependent two pole network (47), whose blocked state setting renders the two pole admittance (46) effective and whose the open state setting bridges the two-pole admittance (46), so that when the switching diode (36) is blocked, the high-frequency reception signals (8) are reduced by a factor (t) that is largely independent of the frequency.
11. Active wide-band receiver antenna according to claim 10 characterized by the fact that the frequency-dependent two-pole network (47) is formed by the input admittance of a two-terminal filter circuit (48) that is arranged according to the structure of the low-loss frequency circuit (3), at least in terms of the major dummy elements, and whose dummy elements are chosen to have a higher impedance than the corresponding dummy elements of the low-loss filter circuit (3) by a non-frequency-dependent factor (t-1) and the two-pole filter circuit (48) is terminated with an impedance of the same factor as the effective impedance (5) of the subsequent circuit (Fig. 8).
12. Active wide-band receiver antenna according to one of claims 1 to 6 characterized by the fact that the adjustable transmission element (34) contains a repeater (38) with incremental transmission ratio (t) and switching diodes (36) exist as adjustable electronic elements (32) which are controlled in such a way that at high receiving levels the transmission ratio (t), and therefore the ratio of the input voltage UE to the output voltage UA of the adjustable transmission element (34) is set to a suitably high level (Fig. 4, 6)
13. Active wide-band receiver antenna according to one of claims 1, 2, 4 to 6 characterized by the fact that there are several low-loss filter circuits (3) in the transmission network (31) with filter character, which are switched by means of switching diodes (36) or, alternatively, between the input and output of the transmission network (31) and whose input admittance (7, 7') are formed with fixed setting dummy elements (20) in such a way that, when a prescribed reception level is not reached, the required frequency-dependency of the conductance G(f) of the effective input admittance (7) at the source connector (24) for higher amplification of the active antenna, and, when a prescribed reception level is exceeded, the required frequency-dependency of the conductance G(f) of the effective input admittance (7') for reduced amplification of the active antenna at the source connection (24) are achieved with the help of the switching diodes (36) (Fig. 7).
14. Active wide-band receiver antenna according to one of claims 1 to 13 characterized by the fact that the 3-pole amplifying element (2) takes the form of a field effect transistor whose high impedance control port (15) is formed by the gate whose source port (24) is formed by the source and whose sink port (53) is formed by the drain port.
15. Active wide-band receiver antenna for use above 30 MHz according to claim 14 characterized by the fact that the field-effect transistor (2) exhibits a negligible source of parallel noise current i_r, a very small gate drain capacity C₁ and a very small gate source capacity C₂ and a negligible 1/f noise and whose minimum noise temperature T_N0 during noise adjustment is substantially smaller than the ambient temperature T₀ (Fig. 2).
16. Active wide-band receiver antenna according to one of claims 1 to 13 characterized by the fact that the 3-pole amplifying element (2) takes the form of an extended three-pole amplifying element, consisting of an input field effect transistor (13) whose source controls the bipolar transistor (14) in emitter successor circuit and whose emitter port (12) forms the source electrode of the extended field effect transistor (2) (Fig. 9a).
17. Active wide-band receiver antenna according to one of claims 1 to 13 characterized by the fact that the three-pole amplifier element (2) takes the form of an extended three-pole amplifier element, consisting of an input bipolar transistor (49) whose emitter controls another bipolar transistor (50) in emitter successor circuit and whose emitter port (12) forms the source port (24) of the three-pole amplifier element (2) and whose idle current is set at a lower level in the input bipolar transistor (49) than in the remaining bipolar transistor (50) (Fig. 9b).
18. Active wide-band receiver antenna according to one of claims 1 to 13 characterized by the fact that the three-pole amplifier element (2) takes the form of an extended three-pole amplifier element, consisting of an input bipolar transistor (49) or bipolar input effect transistor (13) whose collector port or drain port is connected to the source or emitter port of an additional transistor (51) and whose base or gate port is connected to the emitter or source port of the input bipolar transistor (49) or input field effect transistor (14), so that this connection forms the source connection (24) of the three-pole amplifier element (2) (Fig. 9c).
19. Active wide-band receiver antenna according to one of claims 1 to 13 characterized by the fact that the three-pole amplifier element (2) takes the form of an extended three-pole amplifier element containing an electronically controllable idle current source (I_S0) and/or an electronically controllable idle voltage source (U_D0) by means of which, in the context of the reduction in the amplification of the active antenna because of the excessively high reception level (in line with the invention), the idle current (J_S0) and/or the idle voltage (U_D0) in the input bipolar transistor (49) or input field effect transistor (13) is set at a higher level (Fig. 9d, 6).
20. Active wide-band receiver antenna according to one of claims 1 to 13 characterized by the fact that the passive part of the antenna (1) has a connection point (18) whose two ports are high in comparison with ground (0), each of which is connected to a control port (15, 16) of a three-pole amplifier element (2) whose source ports (24 a, b) are connected to the primary side of a repeater (38) in the form of an isolating transformer, whose secondary side has different outputs for arranging different transmission ratios (t) and from which - together with switching diodes (36) - form the adjustable transmission element (34) and connect the sink ports (53) to ground (0) (Fig. 10)
21. Active wide-band receiver antenna according to one of claims 1 to 13 characterized by the fact that there are several bipolar transistors (14, 14') for expanding the three-pole amplifier elements (2) and for the combined formation of several three-pole amplifier elements (2, 3') used in producing several transmission frequency bands whose basic electrodes are connected to the source electrode of a common input transistor (13, 49) or to the source port of an expanded three-pole amplifier element according to claims 16, 17, 18 and 19 and which are connected in a subsequent emitter circuit with the input of a low-loss filter circuit (3, 3') for form isolated transmission routes for the relevant frequency bands and each of the transmission routes contains an adjustable transmission element (34, 34') and a control amplifier (33. 33') to which only the frequency band from the high-frequency reception signal (8) assigned to the relevant transmission route by means of filter measures is supplied and the control signal (42, 42') is supplied to the assigned adjustable transmission element (34, 34') (Fig. 11).
22. Active wide-band receiver antenna according to claims 21 characterized by the fact that the control signals (42, 42') are derived from the output signal of the active antenna through a selection medium and control amplifier (33, 33') and are supplied to the active antenna by means of control lines (41) (Fig. 12).
23. Active wide-band receiver antenna according to one of claims 1 to 22 characterized by the fact that there are several passive antenna parts (1) with frequency-dependent and, in respect of incident waves according to extent and phase different radiation patterns for the effective lengths I_e which coupled by electromagnetic radiation and together form a passive antenna setup (27) with several connector points (18 a, b, c), each of which is switched with an amplifier circuit (21 a, b, c) and is supplemented to form an active antenna in line with the invention, so that the connection of the amplifier circuits (21 a, b, c) at the passive antenna parts (1) does not cause noticeable mutual influence on the reception voltages and the high-frequency reception signals (8 a, b, c) are brought together in an antenna combiner (22) and the active receiver antennae each contain a control amplifier (33) for monitoring the high frequency reception signal (8) at the antenna output.
24. Active wide-band receiver antenna according to one of claims 1 to 22 characterized by the fact that there are several passive antenna parts (1) with frequency-dependent and, in respect of incident waves according to extent and phase different radiation patterns for the effective lengths 1_e which coupled by electromagnetic radiation and together form a passive antenna setup (27) with several connector points (18 a, b, c), each of which is switched with an amplifier circuit (21 a, b, c) and is supplemented to form an active antenna in line with the invention, so that the connection of the amplifier circuits (21 a, b, c) at the passive antenna parts (1) does not cause noticeable mutual influence on the reception voltages and the high-frequency reception signals (8 a, b, c) are brought together in an antenna combiner (22) and there is a common control amplifier (33) whose control signal (42 a, b, c) is supplied to the transmission networks (31 a, b, c) in the active antennae to reduce the accumulated high-frequency reception signal (8) (Fig. 13).
25. Active wide-band receiver antenna according to claim 24 characterized by the fact that the active receiving antennae are used in an antenna diversity system for cars and the passive antenna parts (1) have been selected so that their reception signals in a Rayleigh reception field are as independent from one another as possible and the high-frequency reception signals (8) are made available for selection in a scanning diversity system or for further processing in one of the other known diversity processes in a non-reactive way, i.e. without influencing the diversity-based independence of the reception signals (Fig. 14).
26. Active wide-band receiver antenna according to claim 25 characterized by the fact that the active receiving antennae are used in an antenna diversity system for cars and the passive antenna parts (1) have been selected so that their reception signals in a Rayleigh reception field are as independent from one another as possible and the high-frequency reception signals (8) are made available for selection in a scanning diversity system or for further processing in one of the other known diversity processes in a non-reactive way, i.e. without influencing the diversity-based independence of the reception signals and the level of the selected signal is fed to a common control amplifier (33) in which a control signal ((42) is formed and fed to the transmission networks (31) in the active reception antennae to reduce the selected high-frequency reception signal (8) (Fig. 14, 15).
27. Active wide-band receiver antenna according to claim 25 characterized by the fact that the active reception antennae (21) each contain a control amplifier (33) for monitoring the high-frequency reception signal (8) at the antenna output (Fig. 16).
28. Active wide-band receiver antenna according to claims 25 and 26 characterized by the fact that in order to improve the diversity-based independence of the reception signal of the passive antenna parts (1) their connection points (18) have specially determined dummy conductance values (23) in parallel with the input of the amplifier circuit (21) (Fig. 16).
29. Active wide-band receiver antenna according to claims 1 to 28 characterized by the fact that when the transmission network (31) is set for small high frequency signals (8), the effective conductance at the output (4) of the low-loss filter circuit (3) is determined by the input resistor of a high-frequency line (10) with the load resistance (9) applied at the end at the load resistance (9) is formed by the input impedance of a further amplifier unit (11) with noise coefficient Fu and the teal part G of the effective admittance (7) is chosen to be sufficiently great so that the noise value of the amplifier unit (11) is less than the noise value of the field effect transistor (2) Fig. 5).
30. Active wide-band receiver antenna according to claims 1 to 29 characterized by the fact that for the wide-band creation of favourable transmission conditions in the filter circuit (3) there us a repeater (24) with suitable transmission ratio U.
31. Active wide-band receiver antenna according to claims 1 to 19 characterized by the fact that frequency-selective transmission routes are formed in the low-loss filter circuit (3) on the basis of signal branching for the purpose of frequency-selective exclusion of high-frequency reception signals (8) for different transmission frequency bands at several outputs.
32. Active wide-band receiver antenna according to claims 1 to 31 characterized by the fact that the passive antenna layout (27) takes the form of conductor structures on a plastic carriers set into a groove in a conducting car body or mounted on a car window, e.g. in the form of one or more heated areas and/or conductor structures separate from the heating and these conductor structures have several connector points (18) for forming passive antenna parts (1) for connecting amplifier circuits (21) (Fig. 15, 16).
33. Active wide-band receiver antenna according to claims 1 to 31 characterized by the fact that the passive antenna layout (27) takes the form of a largely cohesive conducting surface for suppressing the transmission of radiation in the infrared range with a sufficiently low surface resistance on the window of a car and in order to exclude reception signals at suitably positioned connecting points (18) with amplifier circuits (21) on the edge of the conducting surface not connected to the conducting bodywork whose high frequency reception signals (8) are supplied by means of high frequency lines (10) to produce a beam antenna for an antenna combiner (22) or to form a scanning diversity system for an electronic switch (25) or to form a diversity setup that operates according to some other process.
34. Active wide-band receiver antenna according to claims 1 to 31 characterized by the fact that the passive antenna part is derived from a part of the car that was not originally intended for use an antenna and whose design cannot be changed very much and on which there is a connection point (18) for forming a passive antenna part (1) and for the applicable polarization and elevation of an incident wave in the usable frequency range a particular azimuth mean value D_m of the directivity is determined and the real part R_A of impedance Z_A of the passive antenna part (I) in the transmission frequency range is in the range between R_Amin and a maximum value R_Amax (Fig. 18 a, b, c).
35. Active wide-band receiver antenna according to claims 1 to 34 characterized by the fact that there is a modem digital computer and both the impedance Z_A of the passive antenna part (I) is recorded by measurement or calculation, as well as the azimuth mean value D_m of the directivity determined by measurement or calculation are stored in the digital computer and in which for various characteristic possible frequency patterns of antenna impedances suitable basic structures for low-loss filter circuits (3) are stored in the digital computer (3) and with the help of known strategies for calculating variations the dummy elements of the low-loss filter circuit (3) are determined for a specified mean gain in the active antenna.
36. Active wide-band receiver antenna according to claims 1 to 35 characterized by the fact that the low-loss filter circuit (3) takes the form of a T semi-filter or T filter and/or of a chain of such filters, whose series and/or parallel branch are formed from a combination of dummy resistors in such a way that both the absolute value of a resistance in the series branch (28) and the absolute value of a susceptance in the parallel branch (29) in each case is sufficiently small within a transmission frequency range and sufficiently large outside of such a range and that high frequency reception signal (8) at the output is fed to the control amplifier (33) and whose regulating signal (42) controls the adjustable transmission element (34) (Fig. 19 a, b).
37. Active wide-band receiver antenna according to claims 1 to 36 characterized by the fact that for the purposes of the separation of a miniaturized front-end of the active antenna in the low-loss filter circuit (3) a high-frequency line (10) as a element that transforms the effective admittance (7) on a frequency-dependent basis (Fig. 5).
38. Active wide-band receiver antenna for ultra-short wave reception in cars according to one of claims 1 to 32 characterized by the fact that the passive antenna part (I) takes the form of a conductor structure printed on a dielectric carrier, such as a window or plastic backing and the low-loss filter circuit (3) as a band-pass with admission in the ultra-short wave range and high input impedance outside the ultra-short wave frequency range (fig. 1),
39. Active wide-band receiver antenna according to claims 1 to 38 characterized by the fact that there is a repeater (24) with sufficiently high-impedance primary inductivity and a suitably chosen transmission ratio for the wide-band increase of the effective length I_e of the passive antenna part (1) between its connecting point (18) and the input of the amplifier circuit (21) (Fig. 17).

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