EP1487052B1 - Antenna system in the aperture of an electrical conducting car body - Google Patents

Abstract

The arrangement has an aperture length (L) selected to be so small that the resonant frequency of the aperture (1) exceeds the center frequency of the working frequency range, a capacitive tuning element for tuning the aperture resonance to a resonant frequency close to the center frequency and a coupling element for coupling the antenna connection points to the electromagnetic fields in the aperture with minimal residual inductive effects of magnetic reactive power.

Description

The invention relates to an antenna arrangement in the substantially rectangular or. trapezoidal aperture 1 of an electrically conductive vehicle body in the meter wave range z. B. for FM reception.

The invention is based on an antenna system, such as in the DE 195 35 250 A1 in Figure 4a using the example of a roof segment for a small vehicle is described. The antennas specified there (5.6) for frequencies up to the meter range are preferably formed as conductor structures of thin wire. Due to the limited space available in vehicle construction spaces for the segments described there are primarily roof segments or segments in the trunk lid in question wherein the aperture length L through the vehicle width and their aperture width B by other vehicle technical predetermined conditions, such as the sunroof, the rollover security, etc. is restricted. This leads, in particular in the range of meter waves, to the fact that the aperture length L is often smaller than half the operating wavelength and the aperture width B must be chosen smaller than 1/10 of the operating wavelength. In this case, with the in the DE 195 35 250 A1 In Figure 4a proposed antennas (5,6) the task of a low-loss adaptation with the greatest possible bandwidth can not be realized. Even with larger passenger vehicles is for the aperture length L more than 90 cm hardly available. This means in the VHF range at a center frequency of f _m = 97 MHz relative to the wavelength of this frequency relative aperture length L of L / λ, = 0.3 at a relative bandwidth of the VHF range of (f _max -f _min ) / f _m = 0.211. For the FM band in Japan with its center frequency of f _m = 83 MHz, this means a relative aperture length L of L / λ = 0.25 relative to the wavelength of this frequency for a relative bandwidth of the FM band of (f _max -f _min ) / f _m = 0.17. The proposed antennas have the disadvantage of narrowband when adapted to the usual in the antenna technology impedances or the bandwidth of the adaptation can only be achieved through losses. For example, the operating frequency ranges in the form of the above-mentioned frequency bands at the predetermined small relative aperture length L of L / λ = 0.3 and L / λ = 2.5 can not be covered sufficiently low loss; ie the product of efficiency and bandwidth is too small

An antenna arrangement for tuning a conductively bounded aperture is known from US Pat US 3,210,766 , Resonance tuning can only be achieved there if a variable capacitance is introduced between opposing points exclusively in the middle of the longitudinal edges of the aperture. The arrangement specified therein aims exclusively at generating a resonance of the structure - without consideration of the resulting bandwidth of the resonance. The disadvantages associated therewith result from the restriction to introduce the tuning element in the middle of the longitudinal direction of the aperture and to allow no design diversity of the conductive connection between the central, opposite points. Another disadvantage is that the solution specified there is aimed exclusively at the generation of resonances with the aid of a variable capacitive element for discrete variable resonance frequencies and thus a broadband resonance for covering a whole frequency range, such as the VHF frequency range is not possible.

The object of the invention is therefore to avoid the disadvantage of given at low loss adaptation of the antenna narrow band in such aperture antennas.

This object is achieved by means of the features of the main claim.

Fig. 1a)

Aussparung mit der Aperturlänge L und der Aperturbreite B im leitenden Dach eines Kfz zur Bildung einer Antenne nach der Erfindung

Fig. 1b)

azimutales Strahlungsdiagramm bei Horizontalpolarisation bei Frequenzen unterhalb der Apertur-Eigenresonanz

Fig. 2a)

Frequenzverlauf der Leerlauf-Empfangsspannung am Ankoppelelement 3 zum Nachweis der Eigenresonanzfrequenz f_s der Apertur

Fig. 2b)

Anordnung zur Feststellung der Eigenresonanzfrequenz f_s

Fig. 2c)

Frequenzverlauf der Leerlauf Empfangsspannung einer Antenne nach der Erfindung am Ankoppelelement 3 zum Nachweis der durch Verstimmung reduzierten Resonanzfrequenz f_o

Fig. 2d)

Antenne nach der Erfindung mit einer auf die niedrigere Resonanzfrequenz f_o abgestimmten Apertur mit dem kapazitiven Abstimmelement 5

Fig. 3

1. a) Ersatzschaltbild zur Erläuterung der die Bandbreite reduzierenden Wirkung einer induktiven Komponente im kapazitiven Abstimmelement 5.
2. b) verlustlose Impedanztransformation auf das gewünschte Impedanzniveau bei Frequenzen unterhalb der Eigenresonanz der Apertur.

Fig. 4

Reduzierung der Bandbreite in Abhängigkeit von der Verstimmung fo/fs bei verschiedenen unerwünschten induktiven Effekten im kapazitiven Abstimmelement 5 als Parameter

1. a) Verhältnis von b_ro mit induktivem Effekt zu b_ropt ohne induktiven Effekt jeweils bei f_o
2. b) Verhältnis von b_ro bei f_o mit induktivem Effekt zu b_rs bei der Apertur-Eigenresonanz f_s

Fig. 5

1. a) Realisierung eines kapazitiven Abstimmelements 5 mit induktivitätsarmem Leiter 9 und Ankoppelelement 3 mit kapazitiver Ankopplung 23 und Parallelresonanzkreis 21 zur Gestaltung eines Zweikreis- Resonanzbandfilter -Verhaltens
2. b) Antennenimpedanz an der Antennenanschlußstelle 4 in a) für den FM-Bereich in Japan

Fig. 6

Nachweis der Breitbandigkeit auch bei größerer Bedeckung der Aperturlänge L mit einem induktivitätsarmen Leiter

1. a) Anordnung des induktivitätsarmen Leiters 9 mit kapazitiven Bauelementen 12 und von ihm getrenntem kapazitiven Koppelelement 3 mit Antennenanschlussstelle 4
2. b) Impedanzverlauf für die Anordnung in a) an der Antennenanschlussstelle 4
3. c) wannenartig ausgebildeter induktivitätsarmer Leiter 9 mit Dielektrikum ε_r zur Ausbildung der zur Abstimmung benötigten verteilten Kapazität zwischen Wannenrand 19 und Aperturrand 13. Die Mikrowellenantenne 24 nutzt die Wanne als Grundfläche

Fig. 7

1. a) Anordnung wie in Fig. 6a, jedoch mit kapazitivem Ankoppelelement 3 mit einer einfachen Transformationsschaltung
2. b) Impedanzverlauf für die Anordnung in a) an der Antennenanschlussstelle 4 für das UKW-Band als Betriebsfrequenzbereich

Fig. 8

1. a) Anordnung mit galvanisch mit der Fahrzeugkarosserie verbundenen flächigem Leiter 22 als mögliche leitende Grundfläche 25 für eine Mikrowellenantenne bei einem kombinierten Antennensystem
2. b) Impedanzverlauf für die Anordnung in a) an der Antennenanschlussstelle 4 für das FM- Band in Japan als Betriebsfrequenzbereich

Fig. 9

Grundformen für die Ausbildung von Ankoppelelementen 3

1. a) als magnetischer Dipol 20
2. b) als elektrischer Dipol 26

Fig. 10

Nachweis der Breitbandigkeit auch bei nahezu über die gesamte Aperturlänge L eingebrachte leitende Fläche 17 als induktivitätsarmer Leiter 9 bei kombinierter Verwendung als Ankoppelelement 3 mit Antennenanschlussstelle 4

1. a) Anordnung
2. b) Impedanzverlauf für die Anordnung in a) zur anschließend breitbandigen Transformation für den UKW-Bereich

The invention is further explained below with reference to some embodiments in the figures. Show it:

Fig. 1a)

Recess with the aperture length L and the aperture width B in the conductive roof of a vehicle to form an antenna according to the invention

Fig. 1b)

Azimuthal radiation pattern with horizontal polarization at frequencies below the aperture self-resonance

Fig. 2a)

Frequency curve of the idle receive voltage at the coupling element 3 for detecting the natural resonance frequency f _{s of} the aperture

Fig. 2b)

Arrangement for determining the natural resonant frequency f _s

Fig. 2c)

Frequency response of the idle receiving voltage of an antenna according to the invention on the coupling element 3 for detecting the reduced by detuning resonance frequency f _o

Fig. 2d)

Antenna according to the invention with an adapted to the lower resonance frequency f _o aperture with the capacitive tuning element. 5

Fig. 3

1. a) equivalent circuit diagram for explaining the bandwidth-reducing effect of an inductive component in the capacitive tuning element. 5
2. b) lossless impedance transformation to the desired impedance level at frequencies below the natural resonance of the aperture.

Fig. 4

Reduction of the bandwidth as a function of the detuning fo / fs with various undesired inductive effects in the capacitive tuning element 5 as a parameter

1. a) Ratio of b _ro with inductive effect to b _ropt without inductive effect at f _o
2. b) Ratio of b _ro at f _o with inductive effect to b _rs at the aperture self-resonance f _s

Fig. 5

1. a) Realization of a capacitive tuning element 5 with low inductance conductor 9 and coupling element 3 with capacitive coupling 23 and parallel resonant circuit 21 for designing a two-circuit resonant band filter behavior
2. b) Antenna impedance at the antenna connection point 4 in a) for the FM area in Japan

Fig. 6

Demonstration of broadband even with greater coverage of the aperture length L with a low-inductance conductor

1. a) arrangement of the low-inductance conductor 9 with capacitive components 12 and separated from it capacitive coupling element 3 with antenna connection point. 4
2. b) impedance curve for the arrangement in a) at the antenna connection point 4
3. c) trough-shaped inductance-poor conductor 9 with dielectric ε _r to form the required for tuning distributed capacitance between Bath rim 19 and aperture edge 13. The microwave antenna 24 uses the tub as a base

Fig. 7

1. a) arrangement as in Fig. 6a but with capacitive coupling element 3 with a simple transformation circuit
2. b) impedance curve for the arrangement in a) at the antenna connection point 4 for the VHF band as operating frequency range

Fig. 8

1. a) arrangement with galvanically connected to the vehicle body planar conductor 22 as a possible conductive base 25 for a microwave antenna in a combined antenna system
2. b) impedance characteristic for the arrangement in a) at the antenna connection point 4 for the FM band in Japan as the operating frequency range

Fig. 9

Basic forms for the formation of coupling elements 3

1. a) as a magnetic dipole 20th
2. b) as an electric dipole 26

Fig. 10

Detection of the broadband even with almost over the entire aperture length L introduced conductive surface 17 as a low-inductance conductor 9 in combined use as a coupling element 3 with antenna connection point. 4

1. a) arrangement
2. b) impedance curve for the arrangement in a) for subsequent broadband transformation for the VHF range

The radiation associated with an antenna in an aperture of the predetermined type is determined at aperture lengths significantly below the half-wave resonance mainly by the currents at the aperture edge. With an antenna of this kind, for example in the roof of a motor vehicle ( Fig. 1a ), therefore, for frequencies below the aperture resonance results in a horizontal radiation pattern, as shown in Figure 1b). This directional diagram, which applies to the horizontal polarization, is independent in form of any excitation of the aperture, provided the aperture does not exceed the aperture resonance. Antenna structures, which are introduced into the aperture, thus subject at these frequencies in terms of their own contribution to radiation of the given by the boundary of the aperture dominance of the edge currents. For this reason, it is necessary to design the antenna structures introduced into the aperture in such a way that excitation of the edge currents of the aperture which is as low-loss as possible and the possible bandwidth is reduced as little as possible.

An aperture of the type described has a high pass-like character with respect to its radiation properties, with different beam patterns and relatively large bandwidths with good efficiency can be achieved with relatively slim antenna conductors at frequencies above the aperture self-resonance especially with a larger width of the aperture with different antenna structures and positions. This has been demonstrated in the past by numerous forms of window pane antennas in automobiles.

To illustrate the teaching of the invention, the following description assumes the example of an aperture of length L = 0.9 m and B = 0.2 m. In Fig. 2b this aperture is considered with the coupling line 3 with junction 4. Due to the distributed effect of all influences, the following mathematical relationships are not accurate. However, they describe the phenomena occurring with sufficient accuracy and make it possible to translate the given teaching into practice on the basis of the readable tendencies.

festgestellt. Die Resonanzfrequenz ergibt sich bei Gleichheit der elektrischen, das ist die durch die elektrischen Felder in der Apertur verursachte Blindleistung mit der magnetischen, das ist die durch die magnetischen Felder in der Apertur hervorgerufenen Blindleistung. Bei Frequenzen unterhalb der Resonanzfrequenz, also bei den hier zutreffenden kurzen Aperturlängen, ist die elektrische Blindleistung in der Apertur zu klein, um die gewünschten resonanzartigen Randströme hervorzurufen. Erfindungsgemäß wird dieses Defizit an elektrischer Blindleistung durch ein kapazitives Abstimmelement 5 aufgehoben, so dass die resonanzartigen Ströme nunmehr bei einer niedrigeren Frequenz f_o erzeugt sind, welche durch die resonanzartige Überhöhung der effektiven Höhe in Fig. 2c nachgewiesen ist. Aufgrund der bei der niedrigeren Frequenz f_o kleineren auf die Blindleistung bezogenen Strahlungsdämpfung der Apertur ist die relative Aperturbandbreite $$b_{\mathit{ro}}=\frac{f_1-f_2}{\sqrt{f_1{f}_2}}=\frac{f_1-f_2}{f_o}$$

kleiner als bei der Eigenresonanz f_s der Apertur. Bezeichnet man mit P_ma die magnetische Blindleistung bei der neuen Resonanzfrequenz f_o, so ist die für die Verstimmung notwendige elektrische Blindleistung AP_e gegeben durch: $$\frac{\mathrm{\Delta }{P}_e}{P_{\mathit{ma}}}=1-{\left(\frac{f_o}{f_s}\right)}^2$$

welche mit größer werdender Verstimmung anwächst. Die optimale relative Bandbreite, welche bei dieser Maßnahme für die Resonanzüberhöhung der Aperturströme bei f_o erreicht werden kann, ist gegeben durch das Verhältnis aus der gesamten magnetischen Blindleistung P_ma zur abgestrahlten Leistung P im Sendefall. $$b_{\mathit{ropt}}=\frac{P_{\mathit{ma}}}{P}$$

Erfindungsgemäß wirkt das kapazitive Abstimmelement 5 mit seiner wirksamen Kapazität AC in Fig. 3a zwischen den Berandungspunkten A und A', wobei der an dieser Stelle gestrichelt angegebene Leitwert G_A die wirksame Strahlungsdämpfung der Anordnung repräsentiert. Der Zusammenhang zwischen den die Strahlungsdämpfung repräsentierenden Leitwerten ergibt sich aus dem Spannungsverhältnis Uc zu U_A wie folgt: $$G_A\approx G_c{\left(\frac{U_c}{U_A}\right)}^2$$

und der Zusammenhang zwischen den wirksamen Kapazitäten ist gegeben aus: $$\mathrm{\Delta }C=\mathrm{\Delta }{C}_c\frac{G_A}{G_c}$$

First, the frequency dependence of the received voltage when irradiated in the main receiving direction as effective height h _eff in Fig. 2a considered. In this case, the maximum current allocation occurs at the natural resonant frequency f _{s of} the aperture, which is expressed in a maximum value of the open circuit voltage measured at the coupling point, measured as the effective height. Here, a relative bandwidth b _rs determined by the radiation attenuation and the reactive power ratios becomes as follows $$b_{\mathit{rs}}=\frac{f_1-f_2}{\sqrt{{\mathrm{<figure-callout\; id="1"\; label="f"\; filenames="imgf0002.png,imgf0003.png"\; state="\{\{state\}\}">f</figure-callout>}}_1{\mathrm{<figure-callout\; id="2"\; label="f"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">f</figure-callout>}}_2}}=\frac{f_1-f_2}{f_s}$$

detected. The resonant frequency is given by the electrical equality, that is the reactive power caused by the electric fields in the aperture, which is the reactive power produced by the magnetic fields in the aperture. At frequencies Below the resonant frequency, ie at the short aperture lengths applicable here, the reactive electric power in the aperture is too small to produce the desired resonance-like edge currents. According to the invention, this deficit of electrical reactive power is canceled by a capacitive tuning element 5, so that the resonance-like currents are now generated at a lower frequency f _o , which is due to the resonance-like elevation of the effective height in Fig. 2c is proven. Due to the lower reactive-power radiation attenuation of the aperture at the lower frequency f _o , the relative aperture bandwidth is $$b_{\mathit{ro}}=\frac{f_1-f_2}{\sqrt{{\mathrm{<figure-callout\; id="1"\; label="f"\; filenames="imgf0002.png,imgf0003.png"\; state="\{\{state\}\}">f</figure-callout>}}_1{\mathrm{<figure-callout\; id="2"\; label="f"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">f</figure-callout>}}_2}}=\frac{f_1-f_2}{f_O}$$

smaller than the natural resonance f _{s of} the aperture. If the magnetic reactive power at the new resonance frequency f _o is denoted by P _ma , then the reactive electric power AP _e required for detuning is given by: $$\frac{\mathrm{\Delta }{P}_e}{P_{\mathit{ma}}}=1-{\left(\frac{f_O}{f_s}\right)}^2$$

which increases with increasing nuisance. The optimum relative bandwidth, which can be achieved in this measure for the resonance peaking of the aperture currents at f _o , is given by the ratio of the total magnetic reactive power P _ma to the radiated power P in the transmission case. $$b_{\mathit{R\; opt}}=\frac{P_{\mathit{ma}}}{P}$$

According to the invention, the capacitive tuning element 5 acts with its effective capacitance AC in Fig. 3a between the boundary points A and A ', wherein the guide value G _{A shown} dashed at this point represents the effective radiation damping of the arrangement. The relationship between the conductance values representing the radiation attenuation results from the stress ratio Uc to U _A as follows: $$G_A\approx G_c{\left(\frac{U_{\mathrm{<figure-callout\; id="2"\; label="c\; U\; A"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">c\; </figure-callout>}}}{U_A}\right)}^2$$

and the relationship between the effective capacities is given by: $$\mathrm{\Delta }C=\mathrm{\Delta }{C}_c\frac{G_A}{G_c}$$

As the distance d _A increases, the voltage U _A sharply decreases in relation to the voltage U _C towards the end of the aperture 1, so that both the effective capacitance ΔC and the conductance corresponding to the radiation at this point are determined according to equations (4) and (5) increases strongly. In the arrangements in Fig. 3 the effective capacitances are each represented by the series connection of an inductance L _p or L _pc and a capacitance C _p or C _pc .

An essential element of the present invention is to make the effective capacitance at the selected location in the aperture extremely low induction, that is, with the smallest possible inductive influence. If the influence of the series inductance is negligible, the bandwidth of the resonance peak of the electric and magnetic fields in the aperture is largely independent of the position d _A for the attachment of the capacitive tuning element. In this case, at the frequency f _o, the maximum relative bandwidth b _{ropt results} . If the inductive reactive power P _mp in the element L _p can not be neglected compared to the magnetic reactive power P _ma generated by the edge currents of the aperture, the relative bandwidth at the frequency f _{o is} reduced to the value b _ro approximately according to the following relationship: $$b_{\mathit{ro}}=\frac{P}{\mathrm{\Sigma }{P}_m}=\frac{P}{P_{\mathit{ma}}+P_{\mathit{mp}}}=\frac{{/}_{{\mathrm{<figure-callout\; id="1"\; label="P\; ma\; P"\; filenames="imgf0002.png,imgf0003.png"\; state="\{\{state\}\}">P\; </figure-callout>}}_{\mathit{ma}}}_P^{}}{\left(1+{}^{P_{\mathit{mp}}}{/}_{P_{\mathit{ma}}}\right)}=\frac{b_{\mathit{R\; opt}}}{1+{}^{P_{\mathit{mp}}}{/}_{P_{\mathit{ma}}}}$$

ergibt sich zusammen mit Gleichung (2) eingesetzt in Gleichung (6) für die relative Bandbreite $$\begin{array}{ll}{b}_{\mathit{ro}}& =\frac{b_{\mathit{ropt}}}{1+\left[1-{\left({}^{f_o}{/}_{f_s}\right)}^2\right]\frac{P_{\mathit{mp}}}{\mathrm{\Delta }{P}_e}}\\ & =\frac{b_{\mathit{ropt}}}{1+\left[1-{\left({}^{f_o}{/}_{f_s}\right)}^2\right]{\omega }_o^2\mathrm{\Delta }{\mathit{CL}}_p}\end{array}$$

With $$\frac{P_{\mathit{mp}}}{P_{\mathit{ma}}}=\frac{\mathrm{\Delta }{P}_{\mathit{e}}}{P_{\mathit{ma}}}\frac{P_{\mathit{mp}}}{\mathrm{\Delta }{P}_{\mathit{e}}}$$

is given together with equation (2) in equation (6) for the relative bandwidth $$\begin{array}{ll}{b}_{\mathit{ro}}& =\frac{b_{\mathit{R\; opt}}}{1+\left[1-{\left({}^{{\mathrm{<figure-callout\; id="2"\; label="f\; O"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">f\; </figure-callout>}}_O}{/}_{f_s}\right)}^2\right]\frac{P_{\mathit{mp}}}{\mathrm{\Delta }{P}_e}}\\ & =\frac{b_{\mathit{R\; opt}}}{1+\left[1-{\left({}^{{\mathrm{<figure-callout\; id="2"\; label="f\; O"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">f\; </figure-callout>}}_O}{/}_{f_s}\right)}^2\right]{\omega }_{\mathrm{<figure-callout\; id="2"\; label="O"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">O</figure-callout>}}^2\mathrm{\Delta }{\mathit{CL}}_p}\end{array}$$

des aus 4 und C_p bestehenden Resonanzkreises der Frequenz f_o kommt, umso stärker wird die Bandbreite bei f_o eingeengt. Damit gilt ferner: $$\begin{array}{c}{b}_{\mathit{ro}}\end{array}=\frac{b_{\mathit{ropt}}}{1+\frac{\left[1-{\left({}^{f_o}{/}_{f_s}\right)}^2\right]}{\left[{\left(\frac{f_p}{f_o}\right)}^2-1\right]}}$$

Thus, the bandwidth is considerably reduced by the influence of L _p , which increases with increasing detuning. The closer the resonance frequency f _p $$f_p=\frac{1}{\sqrt{C_p<figure-callout\; id="4"\; label="\; L\; p"\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">\; </figure-callout>L_p{}_{}}}$$

4 and C _p resonance circuit of the frequency f _o comes, the more the bandwidth is narrowed at f _o . Thus, the following also applies: $$\begin{array}{c}{b}_{\mathit{ro}}\end{array}=\frac{b_{\mathit{R\; opt}}}{1+\frac{\left[1-{\left({}^{f_O}{/}_{f_s}\right)}^2\right]}{\left[{\left(\frac{f_p}{f_O}\right)}^2-1\right]}}$$

In Fig. 4a is the bandwidth reduction as a function of the influence of the occurring in L _p undesired reactive magnetic power as a function of the frequency ratio f _o / f _s for different values of C _p / ΔC and P _mp / P _ma shown. Additionally is in Fig. 4b the influence of the unwanted reactive magnetic power on the ratio of the relative bandwidth b _ro at the frequency f _o to the relative aperture bandwidth b _rs at natural resonant frequency f _s , taking into account that at low frequencies the optimally achievable bandwidth for the current resonance with the cube of the Frequency gets smaller. It is therefore all the more important not to reduce the bandwidth of the antenna arrangement by further disadvantageous coupling to the aperture. As the distance d _A from the center increases, compliance with the condition Pmp / OP _e << 1 becomes more and more difficult. This is apparent from the following equation (11) in conjunction with equation (4). Because for equal influence of the inductance L _p applies: $$L_p=L_{\mathit{pc}}\cdot \frac{G_c}{G}$$

For this reason, the capacitive tuning element, in particular when tuned outside the aperture center, must be designed to be particularly non-inductive according to the invention. From the It is clear from the above that a thin antenna conductor inserted into the aperture is not suitable for supplying the reactive power AP _e necessary for the tuning to the aperture 1, since this is impossible due to its self-inductance without the bandwidth-reducing reactive magnetic power P _mp .

The invention will be explained further using the example of an aperture 1 in a vehicle body 2 with an aperture length L of 90 cm and an aperture width B of 20 cm. The aim in this example is to create an antenna for an operating frequency range according to the VHF range in Europe or according to the FM frequency range in Japan. If the capacitive tuning element 5 as in Fig. 2d placed in the center of the aperture length L in the aperture 1, so is sufficient at this high-impedance point, a capacitance C _pc of 5 pF to reduce the natural resonance frequency f _s = 116 MHz of the aperture 1 to f _o = 90 MHz. This goes out Fig. 2c out. The relative bandwidth of the aperture resonance is reduced from b _rs = 0.2 to bro = 0.08. The effective conductance G _c ( Fig. 3b ) is without capacitive detuning in the case of the Apertureigenresonanz f _s about 1 mS and is reduced with the considered detuning to the resonant frequency f ₀ to about 0.54 mS. Together with the reactive power ratios changed at the lower frequency, the indicated detuning results in the relatively large reduction of the relative bandwidth b _{ro of} the aperture resonance. For the positioning of the coupling element 3 with antenna connection point 4, the conductance of 0.54 mS corresponding to a resistance of 1.86 kΩ is too high a value to realize a simple lossless matching circuit. For this reason, it is technically much cheaper to position the coupling element 3 such that the impedance level available there is of the order of the desired antenna impedance, with increasing distance d _D from the center line of the aperture 1, the conductance G in the FIGS. 3a and 3b strongly increases. This impedance level is determined by the conductance G in Fig. 3c determines which represents at the points D and D 'the total radiation attenuation of the aperture, which is analogous to equation (3) that the impedance level strongly decreases according to the relationship to the aperture end and by selecting a suitable distance d _D to the desired value can be adjusted. For the conductance G approximates: $$G\approx G_c{\left(\frac{U_{\mathrm{<figure-callout\; id="2"\; label="c\; U\; D"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">c\; </figure-callout>}}}{U_D}\right)}^2$$

This transformation, which can be regarded as a practically lossless measure, makes it possible, for example, to design an equivalent resonant band filter with two resonant circuits, as described in US Pat Fig. 5a is shown. Here, the aperture 1 acts as a tuned to the frequency f _o resonant circuit. With the help of in Fig. 5a shown coupling capacitance 23 in the coupling element 3 together with the low-loss dummy elements 21, which are connected in parallel as the second resonant circuit of the antenna connection point 4, the in Fig. 5b produce lossy broadband impedance curve shown.

This covers with a broadband loop in the vicinity of the optimum impedance for noise matching to a transistor the low in comparison to the natural resonant frequency of the aperture 1 FM band in Japan (76 to 90 MHz, operating frequency range). In the following it will be shown that the aperture resonance can be produced equivalently in different ways, without the coupling element 3 having to be changed except for fine-tuning measures. The low-inductance conductor 9 can be designed as a flat conductor with a sufficiently large conductor width 11. In this case, concentrated capacitive components 12 can be used to bridge the point of interruption, it being advantageous to avoid undesired inductive effect to use a plurality of such capacitive components 12 distributed over the conductor width 11.

Another possibility of designing the capacitive tuning element 5 with the desired effective capacitance ΔC is the design of the interruption point 6 as a slot capacitance, which can be set by selecting a suitable conductor slot width 14.

Another advantageous possibility of designing the capacitive tuning element 5 is shown in FIG Fig. 5a shown. Here, the capacitive tuning element 5 is introduced into an appreciable distance d _A in the aperture. 1 Because of the considerably larger capacitance C _p than the capacitance C _pc required for center mounting, the influence of the inductance L _{p is} considerably greater there than that of an inductance L _{pc of the} same size when mounted centrally (see equation 11). Therefore, a flat configuration of the low-inductance conductor 9 is advantageous. With a suitable choice of the capacitive component 7 with introduction of concentrated capacitive components 12 at a given edge distance 10 or with a suitable choice of a conductor slot width 14 in the sufficiently large selected conductor width 11 can be in Fig. 5b Achieve shown impedance curve. All possibilities shown in the figures for tuning the aperture resonance are practically equivalent.

In a further advantageous embodiment of the invention, the capacitive tuning element 5 as a larger conductive surface 17 with a longitudinal dimension up to half an aperture length L as low-inductance conductor 9 in the aperture 1, as in Fig. 6a , brought in. The desired overall capacitive effect is formed by the edge distance 10 between the boundary of this conductive surface 17 and the aperture edges 13 in conjunction with suitably distributed concentrated capacitive devices 12. In particular for the design of combined antenna systems in the aperture 1, it is advantageous to form the conductive surface 17 of the capacitive tuning element 5 in a trough-like manner for receiving further antennas for other frequency ranges. This trough can advantageously be designed as a conductive base 25 of microwave antennas 24 ( Fig. 6c ). To lead out the connection lines from the aperture 1, these are made high impedance for the meter wave frequency range by throttling.

It should be noted that due to the remaining small edge distance 10, the contribution of the area of the aperture bridged with the trough to the formation of the self-inductance contributes less and the capacitance has to be correspondingly increased; however, the basic properties of the tuned aperture are preserved. Of course, similarly to the conductive surface 17, which is in the form of a conductive well, it is not necessary to attach the coupling element 3 in the plane of the vehicle body surrounding the aperture 1. Rather, it may also be placed in a recessed manner on a dielectric carrier material in the aperture 1.

The coupling element 3 with its antenna connection point 4 for coupling to the resonance-like excessive magnetic field or to the resonance-like excessive electric field in the aperture 1, with a coupling element 3 with the character of a magnetic dipole 20 or with a coupling element 3 with the character of electric dipole 26 ( Fig. 9a, Fig. 9b ). Magnetically acting coupling elements 3 for decoupling the strong magnetic fields at the end of the aperture 1 are additionally in the FIGS. 2b, 2d and 3a, 3b shown. The decoupling with an electrical monopoly goes out Fig. 8a out. The associated impedance curve in Fig. 6a shows the broadbandness of this arrangement at the antenna junction 4, which advantageously the transformation into the desired impedance curve in Fig. 7b with the in Fig. 7a indicated simple low-loss reactive elements allows.

A particularly advantageous coupling to the aperture 1 is the above-mentioned capacitive coupling for the design of an equivalent resonant band filter with two circles, as described, for example, in US Pat FIG. 5a is shown. A particularly advantageous variant of the embodiment of the coupling element 3 with regard to the design of combination antennas is in Fig. 8a shown. There, the substantially elongated conductor 22 is galvanically connected at its one end to the aperture edge 13. In planar design of the elongated conductor 22, this can be advantageously used as a conductive base 25 of microwave antennas 24 in a combined antenna system. Due to the galvanic coupling, the lead-out of the connection lines of the microwave antennas 24 can take place without problems.

In a further advantageous embodiment of the invention, the capacitive tuning element 5 is combined with the coupling element 3 in that in the aperture 1 over a large part of the aperture length L, a conductive surface 17 is introduced as a low-inductance conductor 9. The vote is made by suitable design of the edge distance 10 in conjunction with the distributed introduction of concentrated capacitive elements 12. Due to the increased concentration of magnetic fields in the immediate vicinity of the edge is not too small edge distance 10 hardly a disadvantageous decrease in the self-inductance connected as a magnetic energy storage of the aperture , The desired antenna impedance can be adjusted with suitable positioning of the antenna connection point 4. This impedance is in Fig. 10b and shows a broadband loop in the frequency range of 80 to 110 MHz. By conventional circuit measures, such a broadband impedance can be transformed into a desired impedance curve, for example for the VHF range.

List of terms

• Aperture 1
• Vehicle body 2
• Coupling element 3
• Antenna connection point 4
• capacitive tuning element 5
• Interruption point 6
• capacitive component 7
• Distance 8
• low-inductance conductor 9
• Edge distance 10
• Conductor width 11
• capacitive components 12
• Aperturrand 13
• Conductor slot width 14
• LMK receiving antenna element 15
• LMK connection point 16
• conductive surface 17
• isolated gap 18
• Tub rim 19
• magnetic dipole 20
• low-loss dummy elements 21
• elongated conductor 22
• Coupling capacity 23
• Microwave antennas 24
• conductive base 25
• electric dipole 26
• Series inductance 27
• effective capacity ΔC
• Aperture length L
• Natural resonance frequency fs
• Reactive power Pmp
• generated reactive power Pma
• Resonant frequency fo
• Distance dA
• Distance dD

Claims (15)

1. Antenna arrangement in the generally rectangular/trapezoidal aperture (1), with aperture length L and aperture width W where W < L/3, of an electrically conductive vehicle body (2) for the very high frequency (VHF) range, whereby:

   - the aperture length L is selected to be so small that the self-resonant frequency (fs) of the aperture (1) is larger than the centre frequency of the service band;

   - a capacitive tuning element (5) to tune the aperture resonance to a resonance frequency fo near this centre frequency and a coupling element (3) to connect the antenna pick-up point (4) to the resonance-type excessive electromagnetic fields present in the aperture (1);

   - the capacitive tuning element (5) is positioned as a capacitively active connection between opposing points (A, A') on the longitudinal edges of the aperture (1) in an initial finite interval (dA) to the centre of the aperture length L and designed with low inductance such that the magnetic reactive power (Pmp) of this connection, caused by the residual inductive effect, is as small as possible in comparison to the magnetic reactive power (Pma) generated by the magnetic fields in the aperture (1):

   - the interval (8) between the two opposing points is bridged with a low-inductance conductor (9), which must be disconnected at one break point (6) at least, and

   - a capacitive component (7) is present on every one of the break points (6), of which there must be at least one, to bridge same; the capacitive value of this component is selected to be sufficiently large that the delivery of the electrical reactive power (Pe) required to tune the aperture (1) to the desired resonance frequency fo is ensured.
2. Antenna arrangement according to Claim 1, characterised in that, in particular where the initial interval (dA) has larger values, the low-inductance conductor (9) is realised as a flat conductor with a sufficiently large conductor width (11) and that for low-inductance, capacitive bridging of the minimum one break point (6) one or, where required, several concentrated capacitive components (12) distributed across the conductor width (11) are used.
3. Antenna arrangement according to one of Claims 1 to 2, characterised in that, only one break point (6) is present on one of the aperture edges (13), such that the entire area of the low-inductance conductor (9) is galvanically connected to the vehicle body (2).
4. Antenna arrangement according to one of Claims 1 to 3, characterised in that, the minimum one break point (6) of the flat low-inductance conductor (9) is a slit with an appropriate conductor slit width (14) with respect to the slit capacity effective between the slit edges, such that the required capacitive effect is achieved with the selected conductor width (11).
5. Antenna arrangement according to one of Claims 1 to 3, characterised in that, to construct the capacitive tuning element (5), the low-inductance conductor (9) is designed as a conducting surface (17) over a large part of aperture length L in the aperture (1), the tuning is provided by a suitably designed edge interval (10) in conjunction with the distributed concentrated capacitive components (12) and the low-inductance conductor (9) combined is used as a coupling element (3).
6. Antenna arrangement according to one of Claims 1 to 2, characterised in that, to construct the capacitive tuning element (5), the low-inductance conductor (9) is provided with small cross-section dimensions near the centre of the aperture length L and the capacitive effect is provided by activating a concentrated capacitive component (7) or, where there are several break points (6), several concentrated capacitive components (7).
7. Antenna arrangement according to one of Claims 1 to 3, characterised in that, to construct the capacitive tuning element (5), a conducting surface (17) with a length dimension of up to half the aperture length L is provided as a wide low-inductance conductor (9) in the aperture (1), that the minimum one break point (6) is provided by the interval between the edges of this conducting surface (17) and the aperture edges (13) and that the suitable capacitive overall effect is achieved through low-inductance bridging with several, distributed concentrated capacitive components (12).
8. Antenna arrangement according to one of Claims 1 to 5, characterised in that, to construct the capacitive tuning element (5), the conducting surface (17) is bowl-shaped, that the one minimum break point (6) takes the form of a continuous dielectric, insulated gap (18) between the bowl edge (19) and the aperture edge (13) and that the gap (18) is formed by shaping and by filling with a suitable dielectric material such that it is possible to tune the aperture resonance to the desired resonance frequency fo.
9. Antenna arrangement according to one of Claims 1 to 8, characterised in that, the coupling element (3) to connect to the resonance-type excessive magnetic field is positioned in the aperture as an antenna element with the character of a magnetic dipole (20).
10. Antenna arrangement according to one of Claims 1 to 8, characterised in that, the coupling element (3) to connect to the resonance-type excessive electrical field is positioned in the aperture as an antenna element with the character of an electric dipole (26).
11. Antenna arrangement according to one of Claims 1 to 8, characterised in that, the coupling element (3) is primarily executed as an elongated conductor and arranged with its antenna pick-up point (4) between two opposing points of the aperture edges (13) at an interval of dD from the centre of aperture length L, whereby this interval dD is selected to be correspondingly large to achieve a sufficiently low impedance level and that the coupling element (3) contains a serial coupling capacity to connect to the aperture (1) as the first resonant circuit of a capacitively coupled two-circuit band filter and that the second resonant circuit of the two-circuit band filter is formed by low-loss dummy elements (21) parallel to the antenna pick-up point (4).
12. Antenna arrangement according to Claim 11, characterised in that, the coupling element (3) additionally contains a series inductance (26), where the inductivity value of the latter in conjunction with the coupling capacity (23) and the low-loss dummy elements (21) creates a three-circuit band filter which increases the bandwidth.
13. Antenna arrangement according to Claim 11, characterised in that, the largely elongated conductor (22) in the coupling element (3) is galvanically connected at one end with an aperture edge (13) and designed flat so that it can be used as the conducting base (25) for microwave antennae (24) for frequencies which are orders of magnitude higher.
14. Antenna arrangement according to Claims 1 to 5 and 7 to 13, characterised in that, the conducting surface (17) is designed as a capacitive tuning element (5) and serves similarly as a conducting base (25) for microwave antennae (24) attached to it for frequencies that are orders of magnitude higher and that the connection lines of the microwave antennae (24) to be lead out of the aperture (1) are each designed using throttling to be highly resistive for the VHF range.
15. Antenna arrangement according to Claims 1 to 2, 4, 5 and 7 to 13, characterised in that, a capacitive long, medium and short-wave receiver antenna element (15) is present in the aperture (1) and that the shielding effect of the low-inductance conductor (9) with respect to reception of the low long, medium and short-wave frequencies is largely neutralised by an arrangement of several break points (6).

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