EP1376746B1 - Nicht-abstimmbares rechteckiges dielektrisches Wellenleiterfilter - Google Patents

Nicht-abstimmbares rechteckiges dielektrisches Wellenleiterfilter Download PDF

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Publication number
EP1376746B1
EP1376746B1 EP03007045A EP03007045A EP1376746B1 EP 1376746 B1 EP1376746 B1 EP 1376746B1 EP 03007045 A EP03007045 A EP 03007045A EP 03007045 A EP03007045 A EP 03007045A EP 1376746 B1 EP1376746 B1 EP 1376746B1
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EP
European Patent Office
Prior art keywords
waveguide
filter
substrate
dielectric
microstrip
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Expired - Lifetime
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EP03007045A
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English (en)
French (fr)
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EP1376746A1 (de
Inventor
Paolo Bonato
Giorgio Dr. Carcano
Lino De Maron
Danilo Gaiani
Fabio Morgia
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Siemens Holding SpA
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Siemens Mobile Communications SpA
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/207—Hollow waveguide filters
    • H01P1/208—Cascaded cavities; Cascaded resonators inside a hollow waveguide structure
    • H01P1/2088—Integrated in a substrate
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P11/00—Apparatus or processes specially adapted for manufacturing waveguides or resonators, lines, or other devices of the waveguide type
    • H01P11/007—Manufacturing frequency-selective devices
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P5/00—Coupling devices of the waveguide type
    • H01P5/08—Coupling devices of the waveguide type for linking dissimilar lines or devices
    • H01P5/10—Coupling devices of the waveguide type for linking dissimilar lines or devices for coupling balanced with unbalanced lines or devices
    • H01P5/107—Hollow-waveguide/strip-line transitions

Definitions

  • the present invention relates to the sector of the technique concerning the implementation of microwave filters, and specifically to a no-tuning filter in rectangular dielectric wave guide.
  • a typical band pass filter operating at the microwave frequencies includes a resonant hollow cavity consisting of metallic waveguide having rectangular cross section, delimited at its ends by metallic walls.
  • the cavity has a predetermined length, generally half wavelength ⁇ G at resonance or its multiples.
  • Input and output couplings are also obtained by appropriate means, similar to probes, to excite the right standing mode in the hollow cavity .
  • the signal to be filtered is inlet in the cavity through the first probe and the filtered signal is collected by the second probe .
  • resonant hollow cavities can be employed; these cavities are separated by metal walls with an opening along one of the transverse axis ("iris"), for instance the shorter axis, to obtain an inductive coupling.
  • iris transverse axis
  • An alternative implementation similar from the electrical point of view, foresees the use of one sole waveguide containing cylindrical conductors of appropriate diameter, arranged transversally to the waveguide, along the longitudinal axis and ⁇ G /2 apart. Said conductors are called "inductive post" , they act as impedance inverters and enable the synthesis of the selected desired bandpass response .
  • the mentioned filters generally have a large size and allow to obtain high values for the unloaded quality coefficient Q o and therefore low insertion losses in the desired bandpass frequency range, but require manufacturing techniques complex and expensive from a mechanical point of view. Said filters are also difficult to integrate with the circuits of microwave transceivers, manufactured nowadays in planar technique; therefore additional electrical and mechanical interconnection elements become necessary. Very often, the filters in metallic waveguide require also a fine tuning to be made manually by a skilled operator, through appropriate regulation elements.
  • a traditional way to reduce the overall dimensions of filters based on hollow waveguide is to fill the cavities with a material having high dielectric constant ⁇ r and low dielectric losses, that is with a material having small tan ⁇ values, where ⁇ is the loss angle appropriately defined.
  • the filling with dielectric material partially reduces the value of the quality factor Q o , therefore a compromise criterion shall be defined between the reduction of the overall dimensions of the cavity and the major insertion losses that can be tolerated for the filter.
  • a filter implemented as just mentioned still shows, the drawbacks of the previous air filled filter, mainly relating to the cost of the mechanical working and subsequent calibration.
  • ⁇ G the wavelength characteristic of the resonant mode
  • the two ⁇ G /4 resonators are inductively coupled through interposition of an appropriate segment of reduced cross section dielectric waveguide along the longitudinal axis, in which an H mode of evanescent type (that damps at a short distance) propagates.
  • Two rectangular shaped metal electrodes are required on two side faces without metal coating, to realize the input/output ports. The filter thus obtained, despite its compactness and reduced dimensions, has some drawbacks.
  • a first drawback is that very high dielectric constant material must be used to confine the electrical field mainly inside the filtering structure, because non metal coated walls would otherwise irradiate the power. This involves a low value of the quality factor Q 0 limiting the frequency range in which this solution is applicable.
  • a second disadvantage is due to the difficulty in realizing the connections between the I/O electrodes of the filter and the conductive lines of the remaining circuits employing it. In fact, said connections foresee welds on orthogonal plans requiring accurate manual operations that do not fit an automatic "surface mounting" manufacturing process.
  • the European patent application EP-A1-1024548 discloses a dielectric filter in which three or more resonators are integrally formed in a rectangular parallelepiped dielectric block completely metallized on its surfaces with exception of two unmetallized dielectric crowns around respective metal patches constituting the input/output electrodes at the two end resonators. Through holes are formed for adjusting the coupling between adjacent resonators.
  • the unmetallized dielectric crowns constitute two dielectric windows; a first one for injecting an input signal on the metal patch into the dielectric cavity, and a second one for extracting a filtered signal from the cavity and making available it on the metal patch.
  • this filter is not suitable to be integrated with a different layout on the same surface of a common dielectric surface, in particular a microstrip layout.
  • the connection to the I/O electrodes shall be performed by wire bonding or equivalent means.
  • the filter in fig. 1 includes a segment of dielectric waveguide made of four contiguous ⁇ G /2 resonators.
  • the waveguide is delimited by a metal coating MET deposited on the upper face of the sub-layer SUB, by a ground plan deposited on the opposite face, and at its longitudinal sides by the crown of peripheral metal coated holes.
  • MET metal coating
  • the transverse spacing among the holes is calculated to obtain the desired inductive coupling between adjacent sections.
  • two identical input/output sections CPW can be seen, each consisting, of a coplanar line ending in a transition TRA towards the rectangular dielectric waveguide.
  • Coplanar lines and relative transitions are obtained removing the metal coating MET from the substrate SUB, as shown in the figure, each transition corresponding to the two shorter segments of coplanar line, which terminate on the metal coating MET and are arranged at right angle versus the segment of longitudinal coplanar line.
  • This kind of filter has been specifically developed for connections to coplanar line circuits, generally used only for millimetre wave applications, a narrow range of microwaves.
  • This paper presents a novel MIC (microwave integrated circuit) design methodology, which consistently applies the H-plane discontinuity structures to integrate a multi-function PCB (printed-circuit board) fabrication process.
  • Two distinct types of waveguide namely: microstrip and metallic rectangular waveguide, are simultaneously integrated on the same substrate(s) through the interface mode converters.
  • the specific design is directed to embody an X-band bandpass filter prototype of the fifth-order (five waveguide sections).
  • the filter includes two tapered microstrips at the two sides of the rectangular metallic waveguide filled with the dielectric of the substrate either constituting a microstrip-to-waveguide transition or vice versa.
  • Six H-plane slits are milled at each side of the rectangular waveguide, symmetrically in respect of the longitudinal axis. Each couple of slits faced at the two sides of the rectangular waveguide behaves as an inductive element (impedance inverter) which controls the coupling between the delimited waveguide sections, for obtaining the desired frequency response.
  • scope of the present invention is to overcome the drawbacks of the known art and to propose a filter in dielectric waveguide that could be completely integrated in microstrip circuits, realized on the same substrate of the waveguide, eliminating the parasitic effects of additional connections.
  • Particular scope of the invention is that to provide an alternative solution to the planar filter disclosed in the lastly cited paper (Tzuang), which constitutes the nearest prior art.
  • scope of the present invention is a microwave filter in metallized dielectric rectangular waveguide, as described in claim 1.
  • the filter implemented according to the subject invention has:
  • Figure 2 shows the dielectric waveguide filter of the present invention.
  • a rectangular shaped central metal coating can be seen on the front side of the dielectric substrate, that extends for the whole width of the substrate up to reaching the two edges, where it continues connecting to a metal coating completely covering the rear face of the substrate (not shown in the figure) to form a resonant dielectric waveguide GDL-RIS.
  • Two metal coatings isosceles triangle shaped with the vertexes in a relevant short micro-strip for the input/output signals, extend from the shorter sides of the metal coating towards the edges of the substrate.
  • the filter has a symmetrical structure along the two axis of the front side of the dielectric substrate. The first striking thing is the compactness and elegance of the filter object of the invention and the fact that it has no tuning devices.
  • Figure 3 shows the front view of a dielectric substrate 1 duly metal coated in such a way as to include the filter of the previous figure not yet separated from the rest of the substrate including other copies of the same filter.
  • the front side metallization includes the two short microstrips 2 and 2' whose length continuously widens to form the triangular metal shapes 3 and 3' connected to the opposite sides of the central metal coating 4, having rectangular shape, corresponding to the upper wall of the dielectric guide GDL-RIS.
  • Two metal-coated grooves 5 and 5' delimit the dielectric waveguide guide GDL-RIS at the sides for all its length and over, if preferred for technological purposes .
  • Figure 4 shows the upper face of the filter of figure 2 , maintaining the same description of the previous figure 3 for the different elements.
  • the scope of this figure is to highlight the dimensions having a functional value.
  • the two smaller external holes F3 and F4 have 0,5 mm diameter and are placed close to the two longitudinal ends of the metal coating 4.
  • Figures 5 and 6 show the pattern of the transverse electrical fields along two cross sections of the substrate of fig.2 matching the dielectric waveguide GDL-RIS and the micro-strip 2 (or 2'), respectively.
  • the two figures highlight the ground plan 6, common to the micro-strip 2 (or 2') and to the dielectric waveguide GDL-RIS, which completely covers the rear side of the substrate 1 wich is continuously connected to the front side metallization visible in figure 4 .
  • the lines of the electrical field have trends coinciding with a "quasi-TEM" propagation mode in micro-strips 2 and 2' and TE 10 in the dielectric guide GDL-RIS. Of course, the two different modes must be well coupled between each other.
  • the triangular metal coatings 3 and 3' attain the double purpose of transforming the "quasi-TEM" mode of the microstrips 2 and 2' into the TE 10 mode of the waveguide GDL-RIS, simultaneously adjusting the impedance seen at the common ends of the two structures.
  • the lines of the transverse electrical field in the different structures represented in figures 5 and 6 are approximately oriented in the same direction and share a same profile, therefore the microstrip appears a suitable way to excite the dielectric waveguide.
  • the metal coatings 3 and 3' improve the above-mentioned suitability, making the two profiles of the electrical field more compatible between them in the filter operating frequency band.
  • the mentioned metal coatings have the additional characteristic to operate a mode transition, distinguishing from the simple "tapers” that perform the sole impedance adjustment. It is known that the propagation constant ⁇ of the TE 10 mode of the rectangular guide depends only on the width a ( fig.4 ) and not on the thickness b ( fig.5 ) of the guide, therefore the guide GDL-RIS thickness can be reduced without affecting the propagation constant, thus enabling to implement dielectric waveguide and microstrip circuits on the same substrate reducing the losses due to interconnections.
  • the filter of the example is a bandpass of the Chebyshev type, having 7,6 GHz central frequency and bandwidth at 20dB Return Loss of approximately 200 MHz.
  • the frequency response we wanted to realize is represented by the measurement of the scattering parameter S 21 and S 11 shown in figure 7 .
  • the design of the filter takes place in three steps: firstly A) the dimensions of the dielectric waveguide GDL-RIS and the first confidence level of the via-holes' diameters are calculated ; afterwards, B) the dimensions of transitions 3 and 3' are calculated; finally C) the filter as a whole is optimised.
  • the background for the design of the two steps A) and B) is largely supplied in the three volumes mentioned in the introduction.
  • the width a is such that the waveguide allows the propagation of the fundamental mode TE 10 for the frequencies included in the passband of the filter.
  • the length Lgdl-ris of the guide GDL-RIS depends on the shape and selectivity of the band pass filtering function we want to synthesize.
  • the problem of the synthesis of a lumped elements bandpass filter is to calculate the parameters of a prototype filter made of a cascade of concentrated constant resonant sections, each section consisting of a branch L s , C s, series, connected to a branch L p , C p , parallel; the cascade being supplied by the signal generator and ending on the matched load.
  • the "distributed" physical filter corresponding to the lumped elements prototype filter is realized selecting a waveguide length Lgdl-ris n-times ⁇ G /2 long for an "n" resonator prototype filter, and drilling n+1 "inductive post" acting as many inductive impedance inverters: these metallized via-holes are placed among adjacent ⁇ G /2 resonators .
  • the diameter of the metal coated holes is calculated based on the inductance value needed for a correct impedance inversion. This method leads to a first approximation project of the filter, which can be immediately verified through a generic linear simulation "tool" for a first design optimisation.
  • step B) the problem is to obtain the dimensions TL and T of the metal coatings 3 and 3' such that the impedance adjustment is optimised in the whole band of the filter. Since said metal coatings correspond to "taper" transitions, their dimensioning can avail of the teachings relevant to the same developed, for instance, in the corresponding sections of the third volume mentioned above (Collins) and of the relevant formula. From the theory we notice that the reflection factor ⁇ i at the "taper" input closed on a load (that in this case is the input impedance of the waveguide GDL-RIS) is expressed through a complex mathematical equation of the integral type evaluated on the "taper" profile.
  • ⁇ i is the function expressing the variation of the normalized impedance Z according to the size TL considered variable (see figure 4 ). Such a function will clearly depend on the profile selected for the "taper” and on the type of line used. Any profile of the transition 3 and 3', provided that it increases as the guide GDL-RIS approaches, can be considered as a progressive widening of the microstrips 2 and 2'. For the linear microstrip profile in of figure 4 the function Z(TL) is well known. An aspect having great importance in the design of a "taper” is to summarize the function Z(TL) that supplies the desired trend in frequency for the reflection factor ⁇ i .
  • Step C) is required by the complexity of the filtering structure and by the need to eliminate any manual tuning after the manufacturing of the filters themselves.
  • a linear simulation tool is inadequate, while it is profitable to have the optimisation made by an electromagnetic simulator for tri-dimensional structures (3-D) such as for instance, that corresponding to the version 5.6 of "Agilent HFSS” developed by Agilent Technologies Inc., located at Palo Alto, California.
  • Figure 7 shows two superimposed diagrams with the measured frequency response of the transmission ( S 21 ) and reflection ( S 11 ) scattering parameters S 21 of the filter shown in figure 2 .
  • These measures have been obtained employing a vectorial networks analyser, like HP8510C , equipped with Wiltron "Universal Test Fixture” calibrated with "Calibration kit - 36804" using a TRL technique, and 25 mils alumina reference standards.
  • the diagrams show that insertion losses are only 0,9dB at 7,6 GHz band centre frequency and the return losses are higher than 20dB in the 200 MHz band around the central frequency.
  • the filter of the example fits to the following generalizations:
  • the manufacturing method of the filter of figure 2 avails of the usual deposit techniques of thin metal layers on dielectric substrates .
  • the election technique is the one availing of the cathode deposit, or sputtering, of a metal multi-layer over an alumina substrate , on which multi-layer, a gold layer is then added according to galvanic or chemical method, after masking with fotoresist and subsequent removal.
  • the sputtering and the subsequent deposit of gold enables also to coat inside the holes F1, F2, F3, and F4 and the longitudinal grooves 5 and 5', the Applicant holds some patents in this respect.
  • a more economic technique avails of the silver serigraphic deposit on the top and bottom sides of the substrate ; the same operation enables the simultaneous deposit of silver in the mentioned holes and grooves.

Claims (5)

  1. Mikrowellenfilter, das ein dielektrisches Substrat (1) umfasst, das eine Metallisierung (2, 3, 4, 3', 2', 5, 5', 6) trägt, die geeignet ist, einen metallischen rechteckigen Wellenleiter (GDL-RIS), der mit dem Dielektrikum des Substrats gefüllt ist und in seiner Grundmode in Resonanz schwingt, und zwei sich verjüngende Mikrostreifen-Wellenleiter-Übergänge als Ein-/Ausgangsstrukturen (2, 3; 3', 2') an den zwei Enden des besagten rechteckigen Wellenleiters zu bilden, welches eine vorgegebene Anzahl von aneinander angrenzenden Wellenleiterabschnitten aufweist, die miteinander durch Kopplungsmittel (F1, F2, F3, F4) gekoppelt sind, die als induktive Element wirken, um die gewünschte Bandbreite von 3 dB zu erhalten, wobei die besagte Metallisierung die Seitenwände (5, 5') des besagten rechteckigen Wellenleiters (GDL-RIS) vollständig bedeckt,
    dadurch gekennzeichnet, dass:
    - die besagten Kopplungsmittel metallisierte Kontaktlöcher (F1, F2, F3, F4) sind, die in Abständen von λG/2 voneinander entlang der längs verlaufenden Symmetrieachse des dielektrischen Wellenleiters (GDL-RIS) angeordnet sind, wobei λG die Wellenlänge der besagten Grundmode ist.
  2. Mikrowellenfilter nach Anspruch 1, dadurch gekennzeichnet, dass sich die Breite der besagten sich verjüngenden Mikrostreifen-Wellenleiter-Übergänge (3, 3') linear vergrößert.
  3. Mikrowellenfilter nach Anspruch 1, dadurch gekennzeichnet, dass sich die Breite der besagten sich verjüngenden Mikrostreifen-Wellenleiter-Übergänge (3, 3') entsprechend einer parabolischen Funktion vergrößert.
  4. Mikrowellenfilter nach Anspruch 1, dadurch gekennzeichnet, dass sich die Breite der besagten sich verjüngenden Mikrostreifen-Wellenleiter-Übergänge (3, 3') entsprechend einer exponentiellen Funktion vergrößert.
  5. Mikrowellenfilter nach einem der vorhergehenden Ansprüche, dadurch gekennzeichnet, dass es einen Teil einer Mehrschichtstruktur bildet, die ein zweites dielektrisches Substrat aufweist, das seine eigenen Mikrostreifen-Schaltungen trägt und mit der Mikrostreifen-Struktur (2, 3; 3', 2') des besagten Filters verbunden ist, die auf dem zweite Substrat umgedreht montiert wird.
EP03007045A 2002-06-27 2003-03-27 Nicht-abstimmbares rechteckiges dielektrisches Wellenleiterfilter Expired - Lifetime EP1376746B1 (de)

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Application Number Priority Date Filing Date Title
ITMI20021415 2002-06-27
IT2002MI001415A ITMI20021415A1 (it) 2002-06-27 2002-06-27 Filtro non sintonizzabile in guida d'onda dielettrica rettangolare

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EP1376746A1 EP1376746A1 (de) 2004-01-02
EP1376746B1 true EP1376746B1 (de) 2006-08-23

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AT (1) ATE337622T1 (de)
DE (1) DE60307733T2 (de)
ES (1) ES2271406T3 (de)
IT (1) ITMI20021415A1 (de)

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WO2011069980A1 (fr) 2009-12-07 2011-06-16 Eads Defence And Security Systems Dispositif de transition hyperfréquence entre une ligne à micro-ruban et un guide d'onde rectangulaire
CN102280679A (zh) * 2010-06-13 2011-12-14 中兴通讯股份有限公司 金属化开槽基板集成波导
US8130063B2 (en) 2008-03-27 2012-03-06 Her Majesty the Queen in right of Canada, as represented by The Secretary of State for Industry, Through the Communications Research Centre Canada Waveguide filter
CN102763269A (zh) * 2009-11-27 2012-10-31 亚洲大学校产学协力团 使用衬底集成波导的移相器
WO2019221885A1 (en) * 2018-05-18 2019-11-21 Intel Corporation Reduced dispersion dielectric waveguides

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US8258892B2 (en) * 2008-02-19 2012-09-04 The Royal Institution For The Advancement Of Learning/Mcgill University High-speed bandpass serial data link
IT1398678B1 (it) * 2009-06-11 2013-03-08 Mbda italia spa Antenna a schiera di slot con alimentazione in guida d'onda e procedimento di realizzazione della stessa
CN101834339A (zh) * 2010-04-23 2010-09-15 电子科技大学 一种基片集成波导结构延迟线
JP5948844B2 (ja) * 2011-12-14 2016-07-06 ソニー株式会社 導波路およびこれを備えたインターポーザ基板ならびにモジュールおよび電子機器
KR101257845B1 (ko) * 2012-09-18 2013-04-29 (주)대원콘크리트 지금분쇄슬래그를 이용한 레진콘크리트 몰탈 조성물 및 레진콘크리트 원심력관
CN105098304B (zh) * 2014-05-20 2018-11-16 中国科学院微电子研究所 一种滤波器及其形成方法
SE541861C2 (en) 2017-10-27 2019-12-27 Metasum Ab Multi-layer waveguide, arrangement, and method for production thereof
CN108923104B (zh) * 2018-06-21 2024-04-19 云南大学 高选择性基片集成间隙波导带通滤波器
CN109687068B (zh) * 2018-07-17 2023-09-01 云南大学 宽带sigw带通滤波器
CN110071352B (zh) * 2019-04-29 2020-12-25 中国科学技术大学 全磁壁三角形滤波器
CN110085955B (zh) * 2019-05-09 2023-12-22 云南大学 超宽带isgw带通滤波器
SE544108C2 (en) * 2019-10-18 2021-12-28 Metasum Ab Multi-layer filter, arrangement, and method for production thereof
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Publication number Priority date Publication date Assignee Title
US8130063B2 (en) 2008-03-27 2012-03-06 Her Majesty the Queen in right of Canada, as represented by The Secretary of State for Industry, Through the Communications Research Centre Canada Waveguide filter
CN102763269A (zh) * 2009-11-27 2012-10-31 亚洲大学校产学协力团 使用衬底集成波导的移相器
CN102763269B (zh) * 2009-11-27 2015-09-16 亚洲大学校产学协力团 使用衬底集成波导的移相器
WO2011069980A1 (fr) 2009-12-07 2011-06-16 Eads Defence And Security Systems Dispositif de transition hyperfréquence entre une ligne à micro-ruban et un guide d'onde rectangulaire
CN102280679A (zh) * 2010-06-13 2011-12-14 中兴通讯股份有限公司 金属化开槽基板集成波导
WO2011157022A1 (zh) * 2010-06-13 2011-12-22 中兴通讯股份有限公司 金属化开槽基板集成波导
WO2019221885A1 (en) * 2018-05-18 2019-11-21 Intel Corporation Reduced dispersion dielectric waveguides

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ITMI20021415A0 (it) 2002-06-27
EP1376746A1 (de) 2004-01-02
ES2271406T3 (es) 2007-04-16
DE60307733T2 (de) 2007-10-11
DE60307733D1 (de) 2006-10-05
ATE337622T1 (de) 2006-09-15
ITMI20021415A1 (it) 2003-12-29

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