EP0205570B1 - Ligne de transmission a multi-conducteur dielectrique compose - Google Patents

Ligne de transmission a multi-conducteur dielectrique compose Download PDF

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Publication number
EP0205570B1
EP0205570B1 EP86900457A EP86900457A EP0205570B1 EP 0205570 B1 EP0205570 B1 EP 0205570B1 EP 86900457 A EP86900457 A EP 86900457A EP 86900457 A EP86900457 A EP 86900457A EP 0205570 B1 EP0205570 B1 EP 0205570B1
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EP
European Patent Office
Prior art keywords
layer
dielectric
transmission line
line structure
dielectric layer
Prior art date
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Expired - Lifetime
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EP86900457A
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German (de)
English (en)
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EP0205570A1 (fr
Inventor
Hermann Brian Sequeira
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Martin Marietta Corp
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Martin Marietta Corp
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Priority claimed from US06/683,535 external-priority patent/US4677404A/en
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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P3/00—Waveguides; Transmission lines of the waveguide type
    • H01P3/02—Waveguides; Transmission lines of the waveguide type with two longitudinal conductors
    • H01P3/08—Microstrips; Strip lines
    • H01P3/081—Microstriplines
    • H01P3/082—Multilayer dielectric
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/10—Auxiliary devices for switching or interrupting
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/32—Non-reciprocal transmission devices
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P3/00—Waveguides; Transmission lines of the waveguide type
    • H01P3/16—Dielectric waveguides, i.e. without a longitudinal conductor
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/20—Non-resonant leaky-waveguide or transmission-line antennas; Equivalent structures causing radiation along the transmission path of a guided wave

Definitions

  • This invention relates generally to high frequency transmission lines and more particularly to planar type waveguide structures for millimeter wave applications.
  • Guided wave transmission lines are widely used to channel the flow of high-frequency electrical energy. Common examples of these transmission lines are the coaxial line, the hollow metallic waveguide and the optical fiber. All these waveguiding structures are useful in long link applications. However, in situations where the distance between the transmitting and receiving points is below a few inches, as in an integrated circuit, planar transmission lines offer an attractive alternative to these types of transmission lines. A variety of planar transmission line configurations are possible.
  • metallic conductors play a primary role in the waveguiding process.
  • a dielectric strip plays a primary role in the waveguiding process.
  • This invention is directed to a multi-layer dielectric slab structure including a substrate layer and a guiding layer, with the substrate layer being bounded on the bottom side by a metal ground plane.
  • a dielectric loading strip metallized on the top face comprises the top layer and forms the remainder of the structure.
  • the layers of dielectric are primarily chosen with respect to their permittivities to keep the propagating energy away from the conductor surfaces, and thereby reduce conductor losses.
  • Such a structure is known from document US-A-4 028 643 up. cit ..
  • the structure according to Claim 1 is characterised by the features: the conductive coating on the top surface of the dielectric loading strip permits single mode propagation over a relatively wide band and radiation losses due to coupling of the desired mode to the substrate modes are reduced and the polarization of the dominant TM01 mode is such as to render the structure of the transmission line relatively insensitive to small deviations from parallelism among the different interfaces of the structure.
  • Figure 1A shows a perspective view of a known prior art microstrip line
  • a mode is a spatial distribution of energy across the cross-section of the guiding structure.
  • a waveguiding structure can propagate several modes. Each of the modes has a characteristic cut-off frequency below which the waveguiding structure will not support it. It is customary to choose the cross sectional dimensions of the waveguiding structure such that, over the frequency range of interest, only one mode will be supported. This mode is often the one with the lowest cut-off frequency and is called the dominant mode and the cut-off frequency for the next higher order mode represents the useful bandwidth for the waveguiding structure. Accordingly, it is customary to design structures for the widest possible bandwidth consistent with single-mode operation as described above.
  • the strip dielectric waveguide structure of Itoh disclosed in the above referenced U.S. Pat. 4,028,643, also has limitations. For one, its dominant mode is actually an admixture of two orthogonal polarization states. Consequently, some leakage via coupling to the parasitic substrate modes is unavoidable. Further, this coupling grows stronger as one progresses to lower frequencies since then, the difference between the guiding structure and the region external to it cannot be distinguished by the propagating energy. Additionally, the structure has no inherent mechanism for propagating direct current energy.
  • FIG. 1A a conducting strip 10 is mounted on a dielectric 11 which is coated on its bottom surface with a metallic ground plane 12.
  • a conducting strip 10 is mounted on a dielectric 11 which is coated on its bottom surface with a metallic ground plane 12.
  • two parallel conductors 13 are placed upon dielectric 14.
  • FIG 1C a coplanar waveguide configuration is shown in which three parallel conductors 15 are placed on dielectric 16, the two outer conductive strips acting as a ground plane.
  • coplanar conductive strips 17 are mounted on dielectric 18, but the edges of the strips are not coextensive with the edges of the dielectric slab, as they were in the slotline of Figure 1B.
  • microstrip structure shown in Figure 1A has proven the most versatile and successful among the prior art configurations using metallic strips.
  • the microstrip type of transmission line has been successfully used in applications up to 60GHz, but even at those frequencies, some of the problems associated with its use are evident.
  • the substrate modes are suppressed by choosing a dielectric substrate that is thin enough. At 60 GHz, a typical substrate thickness must not exceed 0.203 mm (0.008 inches). At higher frequencies, even thinner substrates must be used.
  • the impedance of a transmission line in microstrip is primarily determined by the ratio of the conductor strip width W, to the dielectric thickness h, i.e. W/h.
  • the value of W is bounded at the upper end by the requirement that it be small compared to the wavelength of the propagating energy at the frequency in question.
  • the lower bound on W is determined by the accuracy and reproducibility with which a narrow line can be fabricated.
  • a fourth problem concerns the thermal properties of the structure. Ironically, this consideration leads to the conclusion that the substrate is not thin enough. If the dielectric substrate is a semiconductor on which truly planar transmitting sources are integrated, then the heat generated within these sources would have to be removed if the device is to survive operation. Unfortunately, most electrical insulators are also thermal insulators, diamond and beryllium oxide being exceptions, and consequently, the heat generating device would be thermally isolated from a heat sink, unless the substrate was made very thin.
  • Planar dielectric waveguides offer more convenient substrate and guide dimensions, and also have low loss.
  • a dielectric strip guide is shown in which a dielectric strip 19 is mounted on a dielectric slab 20 which is coated on its bottom surface with metallic ground plane 21.
  • An inverted strip guide is shown in Figure 2B in which a dielectric strip 22 is sandwiched between dielectric slab 23 and metallic ground plane 24.
  • a key feature of planar dielectric structures is that they have very low loss at frequencies where the structures of Figure 1A- Figure 1D cannot be used at all.
  • planar dielectric waveguides have been used at optical frequencies spanning the infrared to visible range. This is in part due to the outright absence of conductors or the relative remoteness of the conductor surfaces from the propagating energy.
  • planar dielectric structures of Figures 2A and 2B suffer from the inherent limitation of being multi-modal in that they all support at least two modes. Any attempt to realize single mode operation usually results in a mode that is too weakly bound to the structure to be of any practical use.
  • the very close separation in the cut-off frequencies between the dominant TM0 mode and the next higher TE0 mode forces either an acceptance of a very narrow band waveguiding structure, or a dual-mode guide.
  • the coupling between these two modes can result in high radiation loss at discontinuities and bends, as well as increased coupling to the spurious substrate modes mentioned in connection with microstrip and dielectric strip waveguide.
  • planar dielectric structures are their extreme sensitivity to the condition of the interface.
  • any roughness in the surfaces of the guiding or cladding media or any bubbles trapped between them during the bonding process can have a profound influence on the losses due to random scattering from these centers at the boundaries.
  • a substrate 30 whose permittivity is ⁇ s and whose thickness is d s is clad on one side, i.e. the bottom face, by a metal ground layer 31.
  • the other side or top face of the substrate 30 is bonded to a dielectric guiding slab layer 32 whose permittivity is ⁇ g and whose thickness is h.
  • a relatively narrower dielectric loading strip 33 of width W, thickness d l and permittivity ⁇ l is bonded to the other side or top face of guiding slab 32.
  • the propagating direction for the electrical energy is along its longitudinal axis.
  • the upper face of the loading strip 33 is clad with a relatively thin metal layer or coating 34, which covers at least one third of the width W of the loading strip and extends uniformly, periodically or aperiodically along its length depending on the needs of the user.
  • the permittivity ⁇ g of the guiding layer 32 is made to be greater than both the permittivity ⁇ s of the substrate 30 and the permittivity ⁇ l of strip 33 due to the fact that energy being propagated seeks the medium with the highest permittivity.
  • the nature and thickness of metal ground layer 31 and metal cladding layer 34 are not critical.
  • a waveguide structure in accordance with the configuration shown in Figure 4 was constructed with a guiding slab layer 32 of RT "Duroid" 6010 and both the substrate dielectric 30 and strip 33 of alumina. "Duroid” is a trademark of Rogers Corp. for filled tetrafluoroeythylene material.
  • the line loss in this waveguide was measured at 94GHz and was found to be only 15,75 dB/m (0.4db/inch), compared with a loss of 98,4 dB/m (2.5db/inch) at that frequency for microstrip, an improvement of almost six to one.
  • the transmission line of Figure 4 combines the wideband feature of microstrip and the low loss characteristic of planar dielectric waveguides. Like the planar dielectric waveguide, it has no "sidewalls" so that scattering losses are reduced. Ohmic conductor losses are reduced substantially below those in an equivalent microstrip structure, permitting operation at higher frequencies. Further, the amount by which they are reduced increases as the frequency is increased. In the example presented, the conductor losses are 56% of their microstrip contributions at 75GHz. At 100GHz, they are 33% of their microstrip contributions. This is a significant result since conductor losses are known to increase with increasing frequency.
  • the dominant TM0 mode is not widely separated from the TE0 mode.
  • the addition of metal conductors 31 and 34 in the manner shown is particularly significant in the instant invention in that the separation between these modes is widened, thus permitting single-mode operation over a wider band.
  • the condition for mode separation is somewhat more complicated but nevertheless calculable when ⁇ s ⁇ ⁇ l .
  • the dominant mode is the TM0 mode whose polarization is such that the electric field is largely at right angles to the metal conductors as required. This entire phenomenon was not previously anticipated.
  • the top conducting strip 34 serves as a reflector to confine the propagating energy to region 1 of the dielectric layers even at low frequencies.
  • the substrate modes in the "wings" of the structure (region 2) have no such confinement means. Consequently, the effective dielectric constant of the wings will decrease with frequency whereas it will remain substantially constant in the region under the strip. This difference in the effective dielectric constants is what produces the confinement of the propagating energy which will therefore be well guided even at low frequencies and dc.
  • the thickness of the microstrip substrate 11 of the known prior structure shown in Figure 1 is limited because of the need to suppress the spurious substrate modes, as indicated previously.
  • the most troublesome among these modes is the TE0 mode for a grounded dielectric slab.
  • the structure shown in Figure 4 suppresses the propagation of this TE0 mode, thus permitting the use of thicker substrates 30 at a given operating frequency than would be possible with a comparable microstrip guide. As a result, losses at waveguide bends and waveguide discontinuities will be greatly reduced.
  • the dimensions of the waveguide ( Figure 4) at the frequency range of 75-100GHz will be larger than those of a microstrip structure ( Figure 1) designed for that range.
  • the choices of the guiding layer 32 thickness h, the substrate dielectric 30 thickness d s , and the loading strip 33 thickness d l are determined by the desired frequency of operation and by the dielectric permittivities ⁇ g , ⁇ s and ⁇ l .
  • a larger difference ⁇ g - ⁇ s or ⁇ g - ⁇ l will lead to smaller values of h, d s and d l , respectively.
  • any or all of the dielectric elements 30, 32 and 33 of the structure shown in Figure 4 may be semiconductors.
  • the conventional shunt and series excitation in microstrip line are well known.
  • the excitation source may be located at the interface 29 between guiding layer 32 and substrate dielectric 30 or at the interface 31 between guiding layer 32 and strip 33.
  • the excitation source would be oriented with its current transport direction parallel to the desired direction of propagation of the energy i.e. parallel to the longitudinal axis of the strip.
  • the transmission line of Figure 4 resembles microstrip with the closeness of the resemblance under the designer's control. At low frequencies, its behavior is identical to microstrip.
  • the structure of the present invention may be viewed as a means to extend the frequency of operation of microstrip circuits without having to change the substrate thickness.
  • a 1,778 mm (0,070 inches) thick conventional microstrip configuration is only usable from dc to 14GHz while the 1,778 mm (0,070 inches) compound dielectric slab in accordance with the invention and as presented in the design example above is usable from dc to 100GHz.
  • the characteristic impedance of the transmission line shown in Figure 4 is determined primarily by the ratio of the width W of the loading strip 33 to the effective guiding layer thickness, (which is always somewhat larger than the actual thickness h), when the width is small compared to the wavelength. Changes in width W can also be used to provide impedance matching and frequency filtering. For widths comparable to or larger than the wavelength, no complicated field analysis is required to define the impedance level, which is dependent on the operating frequency. However, this change need not be very large. In the design example presented, a 50 ⁇ line at 75GHz becomes a 64 ⁇ line at 100GHz. Smaller variations in impedance are possible with alternative designs at the cost of higher line loss.
  • FIG. 5 shows the special case of a symmetric linear taper so that loading strip 36 has the cross section of an isoceles trapezoid.
  • Strip 36 rests on guiding slab layer 37 which is in turn mounted on substrate dielectric layer 38.
  • Ground plane 39 is coated on the bottom of layer 38 while a metal cladding layer 40 is formed on the top surface of the tapered loading strip 36.
  • tapers may also be used for strip 36 such as concave and convex circular, concave and convex hyperbolic, exponential, etc.
  • the technique of tapering the strip 36 has some additional latent advantages: (i) It permits a wider range of conductor linewidths and can be realized without running into the mechanical difficulty of having to mount very thin strips edge-on on the guiding layer; (ii) The tapered sides "soften" the discontinuity at the strip's edge. This has the effect of focusing the energy towards the center of the strip. This focusing effect is increased if the taper is such that the slope at any point on it relative to the vertical is greater than the critical angle for that interface. The tapered sides also increase the separation between the TE0 and TM0 modes beyond the effect previously described. Thus, a wider operating bandwidth is permitted. Alternatively, for a given operating bandwidth, the conductor losses may be reduced even further.
  • One of the anticipated disadvantages is the sensitivity of the propagating energy to imperfections of the tapered sides. This is expected to be more critical for high impedance (narrow conductor width) lines. However, such sensitivity will have a smaller impact than any corresponding effect in a competitive planar dielectric waveguide.
  • the wide bandwidth afforded by the waveguide of this invention makes the medium ideally suited for digital transmission.
  • one or more of the dielectric layers 30, 32, 33 of Figure 4 may be replaced by a non-reciprocal medium, including, for example, a ferroelectric or ferrimagnetic material such as Barium titanate or a ferrite.
  • a ferroelectric or ferrimagnetic material such as Barium titanate or a ferrite.
  • the relatively small volumes in which the propagating waves are confined would enable one to use smaller amounts of control energy, and yet maintain the control energy density (energy/volume) at high enough levels to manipulate the guided energy. In practice, this means that one may use smaller magnetic field strengths to manipulate the high-frequency energy in devices such as ferrite phase shifters and modulators as well as in circulators and isolators.
  • the heat dissipation problem outlined earlier can be more effectively overcome by using materials, such as BeO, that are electrical insulators but thermal conductors for the substrate dielectric and/or dielectric strip. Since these materials can be brought in direct contact with the power-generating device, they can serve as a low thermal resistance path between the device and a heat sink.
  • materials such as BeO, that are electrical insulators but thermal conductors for the substrate dielectric and/or dielectric strip. Since these materials can be brought in direct contact with the power-generating device, they can serve as a low thermal resistance path between the device and a heat sink.
  • semiconductor materials can be used as one of the layers such as the loading layer 33, in the transmission line structure shown in Figure 4.
  • Semiconductor materials can be used as one or more of the layers. This even includes the guiding layer 32. Furthermore, active and passive sources can be integrated into the semiconductor. Active devices should be aligned so that their current path is colinear with the long axis of the transmission line in order that energy may be effectively coupled to and from the line.
  • the waveguide structure of the subject invention thus has the potential for realizing complete circuit and system functions on a single semiconductor wafer; in other words, it is compatible with monolithic integration.

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Abstract

Une ligne de transmission ayant une structure diélectrique en tranches à plusieurs couches comprend: une couche diélectrique de support (30) avec une épaisseur ds et une constante diélectrique epsilons; un plan conducteur de mise à la terre (31) sous la surface inférieure de la couche diélectrique de support (30); une couche diélectrique de support (30); une couche diélectrique de guidage (32) avec une épaisseur h et une constante diélectrique epsilong, epsilong étant supérieure à epsilons, attachée à la surface supérieure de la couche diélectrique de support (30); au moins une couche (33) allongée de chargement diélectrique étroite avec une largeur W, une épaisseur dl, et une constante diélectrique epsilonl, epsilong étant supérieure à epsilonl, attachée à la surface supérieure de la couche diélectrique de guidage (32); et un revêtement conducteur (34) sur la surface supérieure de la couche allongée (32) à mode unique dans une bande relativement large. Des pertes par rayonnement dues à l'accouplement entre le mode désiré et les modes du support, ainsi qu'aux conducteurs, sont réduites et la polarisation du mode dominant est telle qu'elle rend cette structure relativement insensible à de petites déviations du parallélisme entre les différentes interfaces.

Claims (12)

  1. Structure de ligne de transmission comprenant :
    (a) une première couche diélectrique (30) d'épaisseur prédéterminée ds et de permittivité prédéterminée εs, comportant des surfaces parallèles inférieure et supérieure ;
    (b) un plan de masse conducteur (31) sur la surface inférieure de ladite première couche diélectrique ;
    (c) une seconde couche diélectrique (32) d'épaisseur prédéterminée h et de permittivité εg, avec εg > εs, présentant des surfaces supérieure et inférieure parallèles et fixée à ladite première couche diélectrique de manière que la surface inférieure de ladite seconde couche diélectrique soit fixée à la surface supérieure de ladite première couche diélectrique ;
    (d) au moins une troisième couche diélectrique allongée relativement étroite (33) d'épaisseur dl et de permittivité εl, avec εg > εl, présentant des surfaces parallèles supérieure et inférieure, ladite surface inférieure étant fixée à la surface supérieure de ladite seconde couche diélectrique ; et caractérisée par
    (e) un revêtement conducteur (34) sur ladite surface supérieure de ladite troisième couche diélectrique (33), de manière qu'une propagation monomode soit permise sur une bande relativement large et que les pertes par rayonnement, dues au couplage du mode désiré aux modes du substrat et au conducteur, soient réduites et que la polarisation du mode dominant TM₀ soit choisie de manière à rendre la structure relativement insensible aux petits écarts de parallélisme entre les différentes interfaces.
  2. Structure de ligne de transmission selon la revendication 1, dans laquelle ladite première couche diélectrique (30) constitue une couche de substrat, ladite seconde couche diélectrique (32) constitue une couche formant plaque de guidage coextensive avec ladite couche de substrat, et ladite troisième couche diélectrique (33) constitue une couche de bande diélectrique de chargement ayant une largeur sensiblement inférieure à la largeur desdites couches de substrat et de guidage.
  3. Structure de ligne de transmission selon la revendication 2, dans laquelle ledit revêtement conducteur (34) recouvre au moins partiellement ladite surface supérieure de ladite couche de bande diélectrique de chargement.
  4. Structure de ligne de transmission selon la revendication 2, dans laquelle ledit revêtement conducteur (34) couvre au moins un tiers de la largeur de ladite surface supérieure de ladite couche (33) de bande de chargement.
  5. Structure de ligne de transmission selon la revendication 2, dans laquelle ledit revêtement conducteur (34) s'étend uniformément, périodiquement ou non périodiquement le long de sa longueur.
  6. Structure de ligne de transmission selon la revendication 2, dans laquelle une au moins desdites couches est constituée d'une couche de matériau semiconducteur.
  7. Structure de ligne de transmission selon la revendication 2, dans laquelle une au moins desdites couches constitue un milieu dissymétrique.
  8. Structure de ligne de transmission selon la revendication 2, dans laquelle une au moins desdites couches est en plus thermiquement conductrice.
  9. Structure de ligne de transmission selon la revendication 2, dans laquelle la largeur de ladite couche (33) de bande de chargement est modifiée de manière sélective pour assurer l'adaptation d'impédance et le filtrage de fréquence.
  10. Structure de ligne de transmission selon la revendication 2, dans laquelle ladite couche de bande de chargement (33) comporte deux surfaces latérales parallèles.
  11. Structure de ligne de transmission selon la revendication 2, dans laquelle ladite couche (33) de bande de chargement comporte deux surfaces latérales non parallèles.
  12. Structure de ligne de transmission selon la revendication 2, dans laquelle ladite couche (33) des bandes de chargement comprend deux surfaces latérales effilées.
EP86900457A 1984-12-19 1985-12-12 Ligne de transmission a multi-conducteur dielectrique compose Expired - Lifetime EP0205570B1 (fr)

Applications Claiming Priority (12)

Application Number Priority Date Filing Date Title
US06/683,535 US4677404A (en) 1984-12-19 1984-12-19 Compound dielectric multi-conductor transmission line
US06/801,533 US4835543A (en) 1984-12-19 1985-11-27 Dielectric slab antennas
US06/801,535 US4843353A (en) 1984-12-19 1985-11-27 Dielectric slab transistions and power couplers
US06/801,536 US4689584A (en) 1984-12-19 1985-11-27 Dielectric slab circulators
US06/801,537 US4835500A (en) 1984-12-19 1985-11-27 Dielectric slab optically controlled devices
US06/801,534 US4689585A (en) 1984-12-19 1985-11-27 Dielectric slab signal isolators
US801535 1991-12-02
US683535 1996-07-17
US801536 1997-02-18
US801534 1997-02-18
US801537 2001-03-08
US801533 2004-03-17

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EP0205570A1 EP0205570A1 (fr) 1986-12-30
EP0205570B1 true EP0205570B1 (fr) 1993-09-29

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EP (1) EP0205570B1 (fr)
DE (1) DE3587607T2 (fr)
WO (1) WO1986003891A2 (fr)

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DE102007041125B3 (de) * 2007-08-30 2009-02-26 Qimonda Ag Sensor, Verfahren zum Erfassen, Messvorrichtung, Verfahren zum Messen, Filterkomponente, Verfahren zum Anpassen eines Transferverhaltens einer Filterkomponente, Betätigungssystem und Verfahren zum Steuern eines Betätigungsglieds unter Verwendung eines Sensors
US7782066B2 (en) 2007-08-30 2010-08-24 Qimonda Ag Sensor, method for sensing, measuring device, method for measuring, filter component, method for adapting a transfer behavior of a filter component, actuator system and method for controlling an actuator using a sensor
CN102782933A (zh) * 2010-03-12 2012-11-14 欧姆龙株式会社 信号线路的构造、制造方法及使用该信号线路的开关

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WO2006059934A1 (fr) * 2004-12-01 2006-06-08 Telefonaktiebolaget Lm Ericsson (Publ) Ligne microruban et procede de production
FR2966982B1 (fr) 2010-10-27 2012-12-07 St Microelectronics Sa Ligne de transmission pour circuits electroniques
DE102014213849A1 (de) * 2014-07-16 2016-01-21 Siemens Aktiengesellschaft Verfahren zur Übertragung eines Signals, Signalübertragungseinrichtung und Messeinrichtung
CN111370830B (zh) * 2020-03-23 2021-07-16 武汉光谷信息光电子创新中心有限公司 差分共面波导传输线
CN116154437B (zh) * 2022-09-09 2024-05-14 电子科技大学 一种短毫米波高功率法拉第隔离器

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WO1986003891A3 (fr) 1988-01-14
DE3587607T2 (de) 1994-02-10
WO1986003891A2 (fr) 1986-07-03
DE3587607D1 (de) 1993-11-04
EP0205570A1 (fr) 1986-12-30

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