EP0153372A1 - Convertisseur continu-continu proportionnel auto-commande avec protection contre les surintensites de courant - Google Patents

Convertisseur continu-continu proportionnel auto-commande avec protection contre les surintensites de courant

Info

Publication number
EP0153372A1
EP0153372A1 EP19840903151 EP84903151A EP0153372A1 EP 0153372 A1 EP0153372 A1 EP 0153372A1 EP 19840903151 EP19840903151 EP 19840903151 EP 84903151 A EP84903151 A EP 84903151A EP 0153372 A1 EP0153372 A1 EP 0153372A1
Authority
EP
European Patent Office
Prior art keywords
voltage
transistor
winding
power
polarity
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP19840903151
Other languages
German (de)
English (en)
Inventor
Richard W. Lawrence
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Astec Components Ltd
Original Assignee
Astec Components Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Astec Components Ltd filed Critical Astec Components Ltd
Publication of EP0153372A1 publication Critical patent/EP0153372A1/fr
Withdrawn legal-status Critical Current

Links

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3376—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02H—EMERGENCY PROTECTIVE CIRCUIT ARRANGEMENTS
    • H02H7/00—Emergency protective circuit arrangements specially adapted for specific types of electric machines or apparatus or for sectionalised protection of cable or line systems, and effecting automatic switching in the event of an undesired change from normal working conditions
    • H02H7/10—Emergency protective circuit arrangements specially adapted for specific types of electric machines or apparatus or for sectionalised protection of cable or line systems, and effecting automatic switching in the event of an undesired change from normal working conditions for converters; for rectifiers
    • H02H7/12—Emergency protective circuit arrangements specially adapted for specific types of electric machines or apparatus or for sectionalised protection of cable or line systems, and effecting automatic switching in the event of an undesired change from normal working conditions for converters; for rectifiers for static converters or rectifiers
    • H02H7/1213—Emergency protective circuit arrangements specially adapted for specific types of electric machines or apparatus or for sectionalised protection of cable or line systems, and effecting automatic switching in the event of an undesired change from normal working conditions for converters; for rectifiers for static converters or rectifiers for DC-DC converters

Definitions

  • the present invention relates generally to DC-DC converters and more particularly to a method and apparatus for protecting switching circuit components such as power transistors within such converters from overcurrent conditions and thus damage due to overheating.
  • DC-DC converters can provide one or more output voltages derived from an unregulated DC voltage provided by a DC power source.
  • the DC power source conventionally develops the unregulated DC voltage by rectifying a voltage provided by an AC power source.
  • the DC-DC converter may find particular usefulness as the power supply for computer circuits.
  • computer circuits and their electronic hardware require well regulated DC voltages for proper operation. For example, should the output voltages of the computer's power supply become momentarily unregulated, i.e. producing a voltage whose value is outside of specified operating ranges, data bit errors may be generated within the computer.
  • a DC-DC converter may also find particular usefulness in other types of devices which have electromechanical components operable from a DC voltage power source.
  • the DC-DC converter may be able to develope one or more output voltages of high power capability, typically in the range of 30-50 volts, which need not be well regulated but should be current limited to protect the components of the DC-DC converter from potential damage due to
  • One such electromechanical device is a printer which has various electromechanical components whose current demands are not steady state but vary as a function of the particular printing function being performed, e.g., the typing of a character. Although the average power consumption by the printer may be relatively low, such printing functions may require very high current pulses for short periods of time. It is forseen that computers and printers will be provided as an integrated system preferably operable from a single power supply, or that a separate printer will be powered by the same power supply that powers a computer. In a DC-DC converter power supply designed to power such a system, a feedback regulating means of some sort would normally be provided to limit power transfer through the power transformer of the power supply during periods of excessive load.
  • the present invention is directed at overcoming the above problems by providing a DC-DC converter that provides a high power substantially unregulated DC voltage, wherein the converter compensates for overcurrent conditions as a predetermined function of the amount of excessive current being demanded at the power supply output and provides a simple means for protecting the converter from output short circuits.
  • This substantially unregulated DC is then used to power the printer or other electromechanical load that is relatively insensitive to voltage regulation.
  • a second DC-DC converter stage may also be fed by this unregulated DC voltage according to the present invention, which functions to generate the well regulated voltages required by the computer or other more sensitive electronic hardware.
  • output load overcurrent "protection is added to the primary side of the power transformer of a self-driven proportional DC-DC converter.
  • the DC-DC converter is current limited by periodically momentarily suspending operation of the switching circuit on the primary side of the DC-DC converter in response to excessive current loads experienced on the secondary side of the DC-DC converter.
  • the frequency of these momentary operational suspensions is a predetermined function of the level of the excessive current.
  • the output voltage of the DC-DC converter may then be used to power a printer and also to power a second stage, as mentioned above, such as a second DC-DC power supply, for developing one or more well regulated, low power output voltages usable by a computer, or the like.
  • a second stage such as a second DC-DC power supply
  • the output voltage regulation of this second stage remains relatively
  • FIG. 1A illustrates a self-driven proportional DC-DC converter power supply
  • FIG. IB is a schematic diagram of a circuit which provides overcurrent protection of the DC-DC converter of FIG. 1A according to the present invention
  • FIG. 2 illustrates two modes of operation of the DC-DC converter illustrated in FIG. 1;
  • FIG. 3 illustrates an overall functional block diagram of a two stage DC-DC power supply utilizing the overcurrent protection circuit of the present invention.
  • a self-driven proportional DC-DC converter includes a pair of complementary power transistors which alternately turn on to push or pull current through the primary winding of a power transformer.
  • the base of each power transistor when on, is driven by current proportional to the current through the primary winding.
  • the voltage at the node common to the primary winding of the power transformer and each of the power transistors is a square wave voltage having a peak to peak amplitude determined by the differential DC voltage applied to the converter.
  • the overcurrent protection circuit of the present invention clamps the voltage at the base of one of the power transistors when a high output current demand is detected. By clamping the base voltage for a predetermined time, the normal, alternate on-off operation of each power transistor is stopped for the duration of the predetermined time. By delaying the transition from having one power transistor on to having the other transistors on, the current through the primary winding is limited and hence so is the current delivered to the load. After the predetermined time has expired, the alternate on-off operation of the power transistors is resumed with the cycle continuing until the next overcurrent condition is detected.
  • the output voltage developed by the DC-DC converter is thus useful for supplying high current pulses and also provide an unregulated source DC
  • SUBSTITUTE S voltage to a secondary power supply stage for developing low level, well regulated voltages.
  • FIG. 1 there is shown a typical self-driven, proportional DC-DC converter 10.
  • a differential, unregulated DC voltage is applied to converter 10, the high voltage Vn being applied to voltage rail 12 and the low voltage V ⁇ being applied to voltage rail 14.
  • a pair of energy storage capacitors 16 and 18, are connected in series between rails 12 and 14, and operate to store the energy applied to each of voltage lines 12 and 14.
  • a pair of power transistors Ql and Q2 are alternately turned on to push or pull current through a primary winding 20 of a current transformer 22. For example, when power transistor Ql is on, the direction of current is into the dotted side of primary winding 20 and returns to the node 23 between the series connection of energy storage capacitors 16 and 18.
  • a secondary winding 24 of transformer 22 forward biases a diode 26 and the direction of current is out of the dotted side of secondary winding 24.
  • This current is applied to the inductance of an C filter 28 and charges the capacitor C to provide a "raw" output DC voltage V 0 .
  • the output current of secondary winding 24 returns to its center tap 25.
  • SUBSTITUTE SHEET also applied to the inductance of LC filter 28 and charges the capacitance C to provide the output voltage
  • the turns ratio of primary winding 20 to secondary winding 24 (measured end tap to end tap) is selected to be 25:12.
  • a pair of diodes 32 and 34 clamp the voltage at the dotted side of primary winding 20 to the high voltage VJJ on voltage rail 12 and the low voltage VL on voltage rail 14.
  • the voltage at the dotted side of primary winding 20 is a square wave whose peak to peak voltage goes substantially "rail to rail" between the high voltage V H and the low voltage L.
  • Each of power transistors Ql and Q2 are driven by a base drive current which is proportional to the current through primary winding 20 and thus their respective collectors.
  • a base drive transformer 36 is provided to apply a base drive current to each of power transistors Ql and Q2 in proportion to the current through primary winding 20.
  • Transformer 36 has a first winding 38 coupled in series with the dotted side of primary winding 20, a second winding 40 coupled to the base of power transistor Ql and a third winding 42 coupled to the base of power transistor Q2.
  • Second and third windings 40 and 42 are arranged with respect to first winding 38 so that current into the dotted side of primary winding 20 drives a current into the base of power transistor Ql whereas a current out of the dotted side of primary winding 20 drives a current into the base of power transistor Q2 through third winding 42.
  • the dotted side of first winding 38 is coupled to the dotted side of primary winding 20
  • the dotted side of second winding 40 is disposed away
  • the turns ratio of first winding 38 to each of second winding 40 and third winding 42 is 1:5. It should be obvious, however, that the turns ratio is determined by the large signal DC gain factor, or "beta", of power transistors Ql and Q2.
  • a pair of diodes 44 and 46 have their respective anodes coupled to the emitters of power transistors Ql and Q2. These diodes are provided to raise slightly the emitter voltage of each of power transistors Ql and Q2 to speed the turnoff response time of each of these transistors. The means by which power transistors Ql and Q2 are held off during an output load overcurrent condition is described in detail hereinbelow.
  • the dotted side of second winding 40, the cathode of diode 44 and the collector of power transistor Q2 are coupled at a common node 48.
  • the cathode of diode 46 and the undotted end of third winding 42 are coupled to voltage rail 14.
  • the collector of power transistor Ql is coupled to voltage rail 12.
  • Waveform 48a illustrates the timing of DC-DC converter 10 during a normal mode of operation and preferably has a frequency of at least 20 kHz. This choice of frequency eliminates annoying audio frequency noise that might be created by DC-DC converter 10 if it were operated at some lower frequency.
  • each of power transistors Ql and Q2 is off.
  • An initial start up current is generated by means of a resistor 50 which charges a capacitor 52.
  • capacitor 52 When capacitor 52 is charged to a voltage sufficient to trigger a diac 54, capacitor 52 discharges through diac 54 to apply a current pulse to the base of power transistor Q2 causing transistor Q2 to turn on in response thereto.
  • power transistor Q2 turns on, a current is drawn out of the dotted side of primary winding 20 and into the dotted side of first winding 38. This current develops a proportional current out of the dotted side of third winding 42 and into the base of transistor Q2, thereby maintaining transistor Q2 on once the current out of diac 54 goes away.
  • a resistor 56 and a diode 58 are coupled in series between the node between resistor 50 and capacitor 52 and node 48.
  • Diode 58 is arranged to become forward biased when the voltage of node 48 goes low so that capacitor 52 is discharged into node 48.
  • capacitor 52 prevents the voltage across capacitor 52 from triggering diac 54 during the free-running mode.
  • Resistor 50, capacitor 52 and diac 54 thus comprise just a starting circuit for DC-DC converter 10.
  • Resistor 56 and diode 58 provide means for inhibiting operation of this starting circuit during the free running operation of DC-DC converter 10.
  • a snubber circuit In parallel with primary winding 20 is a snubber circuit comprising a resistor 60 and a capacitor 61.
  • the snubber circuit filters voltage spikes appearing across primary winding 20 during transitions of the direction of current therethrough.
  • Power transformer 22 also includes a first feedback winding 64 and a feedback resistor 62.
  • Feedback resistor 62 is coupled between the base of power transistor Ql and the dotted side of feedback winding 64.
  • the other side of feedback winding 64 is coupled to the undotted side of first winding 38 of transformer 36.
  • the current through primary winding 20 is proportionally smaller.
  • the primary current through first winding 38 may be insufficient to provide an adequate base drive current to the appropriate one of power transistors Ql and Q2 through second winding 40 and third winding 42, respectively. Therefore, one of power transistors Ql or Q2 may not properly turn on.
  • Feedback winding 64 couples sufficient feedback current to the base of power transistor Ql to drive transistor Ql or Q2 into an on state during times of low current demand.
  • transistor Ql is assisted to cycle by means of winding 64
  • transistor Q2 is also assisted in its operation by winding 64 in the following manner.
  • the voltage across third winding 42 is increased, which assists in the complementary turn off and turn on of power transistor Q2.
  • feedback winding 64 and feedback resistor 62 provide an exemplary means for maintaining DC-DC converter 10 in a free running state at very light loads.
  • a resistor 68 is positioned in series between node 48 and first winding 38 of transformer 36.
  • Resistor 68 provides means for developing a voltage which is proportional to the current through primary winding 20.
  • resistor 68 is a small resistance, e.g. 0.20-0.25 ohms, and has a power rating sufficient to handle the expected maximum values of the primary current.
  • the voltage across resistor 68 is used in the operation of the overcurrent protection circuit shown in FIG. IB, as described hereinbelow. As seen in FIG. 1A, points A and B shown at resistor 68 correspond to similarly marked points in FIG. IB.
  • a second feedback winding 66 of power transformer 22 is also provided for connection to the overcurrent protection circuit.
  • the dotted side of feedback winding 66 is coupled to point B and its other side is coupled at a point C.
  • Point C also designates a point common to FIG. 1A and FIG. IB.
  • the overcurrent protection circuit comprises means for providing a signal which clamps the base voltage of power transistor Ql.
  • a point D indicates the point common to FIG. 1A and IB wherein said clamping signal appears.
  • FIG. IB there is shown an overcurrent protection circuit 70.
  • Circuit 70 provides a preferred means for practicing the method and apparatus of the present invention. As described above, circuit 70 is shown with external connections A-D for connection to the like referenced connections of DC-DC converter 10 in FIG. 1A.
  • Overcurrent protection circuit 70 is primarily responsive to the voltage across resistor 68.
  • An overcurrent condition such as a short circuit, will be indicated by the absolute value of the voltage across resistor 68 exceeding a predetermined voltage.
  • the polarity of the voltage across resistor 68 reverses on each half cycle, depending on whether transistor Ql or transistor Q2 is on.
  • Circuit 70 includes a timing circuit comprising a resistor 72 and a capacitor 74 coupled in series between points A and B, and therefore in parallel with resistor 68.
  • the voltage across resistor 68 develops a current through resistor 72 which charges capacitor 74.
  • the time constant of the timing circuit is determined by the values of resistor 72 and capacitor 74, and is selected so that when the voltage across resistor 68 is equal to or greater than the preselected voltage, the voltage on capacitor 74, depending on polarity, will turn on one of transistors Q3 or Q4 prior to a next transition in the on-off states of transistors Ql and Q2.
  • the emitters of transistors Q3 and Q4 are coupled to the potential at point B.
  • capacitor 74 develops the base-emitter voltage for each of transistors Q3 and Q4.
  • transistor Q3 When transistor Ql is on, and the voltage across resistor 68 exceeds the predetermined voltage, the voltage drop from points A to B has a positive polarity and after capacitor 74 has been charged, transistor Q3 turns on. When transistor Q3 turns on, its collector current lowers the voltage at the base of a transistor Q5 by means of a voltage divider formed by resistors 76 and 78. The base-emitter junction of transistor Q5 then becomes forward biased to turn on transistor Q5, which in turn elevates the voltage at its collector. The collector voltage of transistor Q5 forward biases a diode 80 and causes a further base current to be coupled through a resistor 82 to the base of transistor Q3.
  • transistors Q3 and Q5 form a latch wherein transistor Q3 turns on transistor Q5 and transistor Q5 maintains transistor Q3 on.
  • the emitter current for transistor Q5 is provided through a forward biased diode 84, which is coupled at point C to the undotted side of feedback winding 66 in FIG. 1A.
  • the polarity of the voltage across points B and G in FIG. IB is determined by the voltage of feedback winding 66.
  • diode 84 When diode 84 is forward biased, a capacitor 88 also becomes charged.
  • second winding saturates, it is clamped for a preselected delay time to zero volts to delay a transition between the turn off of transistor Ql and the turn on of transistor Q2.
  • the preselected delay time is determined by the RC time constant of capacitor
  • FIG. 2 at 48b illustrates a delayed transition between the turning off of transistor Ql and the turning on of transistor Q2.
  • the voltage at the collector of transistor Q5 is applied through resistor 86 to the base of a transistor Q6.
  • the voltage at point D (the base of power transistor Ql) is at a higher potential then the voltage at point A (node 48) . Therefore, transistor Q6 remains off.
  • transistor Q6 senses this and turns on.
  • the base voltage of power transistor Ql is thereby clamped to the collector voltage of transistor Q6, such that the voltage at point A and point D is substantially equal.
  • the voltage across second winding 40 cannot change polarity, but is forced by transistor Q6 to remain at zero volts, keeping the magnetic flux stored within transformer 36.
  • the voltage across third winding 42 also cannot change polarity.
  • the duration of this delay time is preselected and is . determined by capacitor 88 and resistor 86. Since the voltage across feedback winding 66 has reversed its polarity when transistor Ql turned off, diode 84 becomes reverse biased.
  • transistor Q5 since transistor Q5 is still on, capacitor 88 discharges a current through the emitter of transistor Q5 which is coupled through resistor 86 to the base of transistor Q6. Thus point D, the base of transistor Ql, remains clamped to the emitter voltage of transistor Q6 while capacitor 88 is discharging. After capacitor 88 has discharged, transistor Q5 turns off thereby allowing transistor Q6 to turn off. When transistor Q6 goes off, the stored flux in transformer 36 is enabled to change polarity, allowing the polarity across second and third windings 40 and 42 to reverse to thereby turn on transistor Q2.
  • the voltage across resistor 68 is of reverse polarity e.g., when transistor Q2 is on, and is developing a voltage indicative of excessive current in the opposite through primary winding 20, the voltage at the base of transistor Ql is also clamped so that second winding 40 does not change polarity. However, the voltage at the base of transistor Ql is clamped at the time when transistor Q2 turns off when third winding 42 saturates.
  • Transistor Q4 and a transistor Q7 form a latch as hereinabove described with respect to transistor Q3 and Q5 but are operable when the polarity of the voltage across resistor 68 is reversed.
  • a diode 85 becomes forward biased which couples a current from feedback winding 66 to charge a capacitor 90 and to bias the base of a transistor Q8 through a resistor 92.
  • transistor Q8 then turns on, if there has been an overcurrent detected by transistor Q4, and will clamp the voltage at the base of transistor Ql to store the present polarity of magnetic flux in transformer 36 as hereinabove described.
  • SUBSTITUTE SHEET power transistor is. turned off is illustrated.
  • the symmetrical nature of overcurrent protection circuit 70 is obvious from a study of FIG. IB. Those circuit elements associated with transistors Q4, Q7 and Q8 function as hereinabove described with reference to the symmetrical circuit elements associated with transistors Q2, Q5 and Q6.
  • Waveform 48d in FIG. 2 illustrates the operation of the overcurrent protection circuit 70 at maximum detected overcurrent.
  • the circuit 70 operates to generate an off delay or suspense time as hereinbaove described. Consequently, only a minimum amount of power is coupled to the output of converter 10 when in this operating mode.
  • the overcurrent protection means of the present invention is useful in a power supply system as illustrated in FIG. 4.
  • DC-DC converter 10 and overcurrent protection circuit 70 may provide an output voltage both to a printer 100 and a further DC-DC converter power supply 102. Since power supply 102 operates from a substantially unregulated input DC voltage, any sudden power surges required by printer 100 will not affect the regulated DC output voltages illustrated in FIG. 4 as being generated by power supply 102.
  • the unregulated DC voltage applied to DC-DC converter 10 may be developed in a conventional manner from a diode bridge 104 cascaded with an LC filter stage 106 at its input and a differential RLC filter stage 108 at its output. AC power may be applied to LC filter stage 106 through a fuse 110.
  • One input of diode bridge 104 may be switched to the center node of the input of differential RLC filter 108 to provide selection between an AC power source at 23Ov and an AC power source at 115v.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

Dans un convertisseur continu-continu auto-commandé (10), l'enroulement primaire (20) d'un transformateur de puissance (22) est commandé par une paire de transistors de puissance complémentaires (Q1 et Q2). La base de chaque transistor de puissance (Q1 ou Q2) est commandée alternativement par un enroulement de commande de base associé (40 ou 42) d'un transformateur de courant (38), chacun desquels développe un courant de commande de base proportionnel au courant primaire. L'un des enroulements de commande de base (40 ou 42) se sature normalement pour couper le transistor qui lui est associé (Q1 ou Q2), provoquant ainsi une inversion de la polarité de l'autre enroulement de commande de base (40 ou 42), qui active à son tour le transistor de puissance associé (Q1 ou Q2). Un circuit de protection contre les surintensités de courant (70) détecte tout courant excessif dans le primaire, résultant d'une surcharge ou d'un court-circuit dans le secondaire du transformateur de puissance (22) et immobilise la tension d'un enroulement de commande de base (40 ou 42) pendant un laps de temps présélectionné après la saturation de l'enroulement de commande de base associé (40 ou 42) pour empêcher une inversion de polarité de l'autre enroulement de commande de base. A l'expiration du laps de temps présélectionné, la tension peut de nouveau varier, permettant l'inversion de la polarité de l'autre enroulement de commande de base (40 ou 42), ainsi que la reprise du fonctionnement normal. Un deuxième convertisseur continu-continu (102) peut être placé sur le secondaire du transformateur de puissance (22) pour assurer une amélioration de la régulation de tension.
EP19840903151 1983-08-11 1984-08-13 Convertisseur continu-continu proportionnel auto-commande avec protection contre les surintensites de courant Withdrawn EP0153372A1 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US52263683A 1983-08-11 1983-08-11
US522636 1983-08-11

Publications (1)

Publication Number Publication Date
EP0153372A1 true EP0153372A1 (fr) 1985-09-04

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EP19840903151 Withdrawn EP0153372A1 (fr) 1983-08-11 1984-08-13 Convertisseur continu-continu proportionnel auto-commande avec protection contre les surintensites de courant

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Families Citing this family (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1130751B1 (fr) * 2000-03-02 2016-06-22 THOMSON multimedia S.A. Alimentation de puissance à découpage avec un circuit de protection contre une surtension
EP1130750A1 (fr) * 2000-03-02 2001-09-05 THOMSON multimedia S.A. Circuit de protection pour une alimentation de puissance à découpage
US11699948B2 (en) * 2020-08-07 2023-07-11 Delta Electronics, Inc. Power supply system with self-excited drive function
US11600986B2 (en) * 2020-08-07 2023-03-07 Delta Electronics, Inc. Safety shutdown apparatus with self-driven control, power supply system, and method of controlling the same

Family Cites Families (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3337787A (en) * 1964-09-24 1967-08-22 Walter F Joseph Direct current power supply having plural regulated outputs
US3859586A (en) * 1973-09-26 1975-01-07 Bell Telephone Labor Inc Overcurrent protection circuit utilizing peak detection circuit with variable dynamic response
JPS5512261Y2 (fr) * 1974-11-14 1980-03-17
US3919621A (en) * 1974-11-27 1975-11-11 Motorola Inc Control system for inverter circuits
JPS524035A (en) * 1975-06-28 1977-01-12 Tohoku Metal Ind Ltd Dc stabilizing power source
US4092708A (en) * 1976-12-29 1978-05-30 Honeywell Information Systems Inc. Power supply with overcurrent protection
US4104715A (en) * 1977-03-10 1978-08-01 Acme Electric Corp. Alternating current to alternating current converter apparatus
US4424556A (en) * 1981-08-13 1984-01-03 Matsushita Electric Industrial Co., Ltd. Self-oscillating DC to DC converter
US4453193A (en) * 1982-10-12 1984-06-05 General Electric Company Overcurrent protection for push-pull circuits

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
See references of WO8500933A1 *

Also Published As

Publication number Publication date
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18D Application deemed to be withdrawn

Effective date: 19861104

RIN1 Information on inventor provided before grant (corrected)

Inventor name: LAWRENCE, RICHARD, W.