CN114264877B - Sine wave phase difference accurate measurement circuit and measurement method thereof - Google Patents
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Abstract
The invention discloses a sine wave phase difference accurate measurement circuit and a measurement method thereof, wherein the device comprises a first branch and a second branch, the first branch comprises a first coupler, a first AGC amplitude automatic gain controller, a numerical control phase shifter, an AD8302 phase difference measurement module and a measurement result processing module which are sequentially connected, the second branch comprises a second coupler, a second AGC amplitude automatic gain controller, an AD8302 phase difference measurement module and a measurement result processing module which are sequentially connected, the first branch and the second branch share the AD8302 phase difference measurement module and the measurement result processing module, the output of the second AGC amplitude automatic gain controller is directly connected to the AD8302 phase difference measurement module, and the measurement result processing module CPU is also electrically connected with the numerical control phase shifter. The circuit is an embedded circuit, has low cost, small volume and strong practicability, and the method has high measuring speed and high accuracy.
Description
Technical Field
The invention relates to a communication electric wave accurate measurement technology, in particular to a sine wave phase difference accurate measurement circuit and a measurement method thereof.
Background
In the field of communication, almost all signals are transmitted by electric waves, with the transmission of sine waves as the most basic signal for the carrier wave of radio waves, clock oscillators of large electronic systems, and so forth. Because of the extremely wide bandwidth of optical communication transmission, more and more radio frequency signals are directly transmitted by optical fibers. For example, in the transmission and the reception of phased array radar, the reception of multipath short wave antenna signals, the reception of satellite signals and the like, the A/D, D/A conversion is not needed during long-distance transmission, and the radio frequency signals can be directly modulated into optical fibers for transmission. In a transmission system of multiple radio frequency signals, the phase is an important index, and some of the multiple radio frequency signals are required to have consistent phase, and some of the multiple radio frequency signals are required to have fixed phase difference, or when the phase between the signals drifts, the phase needs to be measured in time for calibration. At present, most of the phase of the measurement signal is measured by a special instrument, but the special instrument is generally large in size and is not suitable for real-time monitoring and mobile measurement. It is therefore necessary to perform accurate, embedded phase difference measurements at the terminals of the signal transmission.
Disclosure of Invention
The invention aims at overcoming the defects of the prior art and provides a sine wave phase difference accurate measurement circuit and a measurement method thereof. The circuit is an embedded circuit, has low cost, small volume and strong practicability, and the method has high measuring speed and high accuracy.
The technical scheme for realizing the aim of the invention is as follows:
the utility model provides a sine wave phase difference accurate measurement circuit, including first branch road and second branch road, first branch road includes the first coupler that connects gradually, first AGC range automatic gain controller, numerical control phase shifter, AD8302 phase difference measurement module and measurement result processing module, the second branch road includes the second coupler that connects gradually, second AGC range automatic gain controller, AD8302 phase difference measurement module and measurement result processing module, wherein, AD8302 phase difference measurement module and measurement result processing module are shared to first branch road and second branch road, the output of second AGC range automatic gain controller directly inserts AD8302 phase difference measurement module, measurement result processing module cpu still is connected with numerical control phase shifter electricity, adopt the mode of shifting the phase difference of two-way signals of first branch road and second branch road to measure, the shortcoming of measuring inherent binary nature of chip, nonlinearity and error imbalance is eliminated to the adoption numerical control phase shifter, can realize positive and negative phase shift, carry out feedback through the measurement result of AD8302, the direction and the size of numerical control phase shifter, avoid the nonlinear area of phase measurement chip, remove binary phase difference and error value and direct measurement result to be transmitted to the actual phase difference value in order to the digital phase difference control form after the direct conversion of the phase difference value of phase difference measurement module, the actual measurement value is calculated to the actual measurement result, the phase difference value is directly transmitted to the digital phase difference value of the actual measurement circuit, the actual measurement value is obtained in the form after the digital phase difference value is directly controlled to the phase difference value is calculated to the actual measurement value, and the actual measurement value is calculated, and the actual value is directly used for measuring the phase difference value is directly to be directly in the phase measurement value.
The first coupler and the second coupler are low-coupling-degree low-insertion-loss couplers, so that the measurement result is ensured not to influence the communication of the main channel signals.
The AGC amplitude automatic gain controller limits the amplitude of the measured signal to a range acceptable to the AD8302 and remains unchanged.
The sine wave phase difference accurate measurement method comprises the sine wave phase difference accurate measurement circuit, and the method comprises the following steps:
1) Setting the number of phase shifters to 0, recording the original phase difference voltage value of the first branch signal B and the second branch signal A measured by the AD8302 as M0, wherein the original phase difference of two branches corresponding to the M0 is theta degrees, and at the moment, the theta has positive and negative possible values;
2) Controlling a numerical control phase shifter to increase the phase of the first branch signal B by 21 degrees, and recording the voltage value measured by the AD8302 as M1;
3) Setting the phase shifter degree to 0, and restoring the phase of the first branch signal B to the initial phase;
4) Controlling a numerical control phase shifter to reduce the phase of the first branch signal B by 21 degrees, and recording the voltage value measured by AD8302 as M2;
5) The absolute value of the difference between M1 and M0 and the absolute value of the difference between M2 and M0 are obtained through calculation, the direction with the larger absolute value is taken as a judgment basis to judge the growth slope of the phase difference, so that whether theta is positive or negative is known, and the accurate value of theta is measured, but the accurate value is not necessarily obtained;
6) If θ is not within the range of ± (60 °,120 °), the measured voltage value M0 has a larger error, and the numerical control phase shifter is controlled to increase the degree of the first branch signal B by Δθ, so as to move the phase difference between the first branch signal B and the second branch signal a within the range of ± (60 °,120 °), and the phase difference value becomes "θ+Δθ";
7) Because of the difference of measurement errors after interval conversion, the phase difference value of the first branch signal B and the second branch signal a needs to be measured again through the AD8302 to obtain a final phase difference voltage M3, and the degree corresponding to M3 at this time is recorded as θ ', θ' =θ+Δθ, because θ 'is in the range of ± (60 °,120 °), the error is smaller, and the value is relatively accurate, and because Δθ is also an accurate value, the original value "θ=θ' - Δθ" is also an accurate value.
By the steps, the problems of nonlinearity and unbalanced error of the measurement result are solved, and the measurement result is more accurate.
Compared with the prior art, the technical scheme has the following advantages:
1) The circuit is an embedded circuit, has smaller volume and can be easily embedded into various use occasions compared with a heavy measuring instrument;
2) The transmission of the measured main signal is not affected, the phase change of the main signal can be monitored and reflected in real time, and a judgment basis is provided for a subsequent circuit;
3) The measurement accuracy is high, and the measurement accuracy can reach 0.228 degrees theoretically by using 12-bit A/D sampling;
4) Compared with the direct measurement by using the AD8302 chip, the method eliminates the measurement defects and has more accurate and precise measurement results.
The circuit is an embedded circuit, has low cost, small volume and strong practicability, and the method has high measuring speed and high accuracy.
Drawings
FIG. 1 is a graph showing the phase difference measurement response of AD8302 in the example;
FIG. 2 is a diagram showing the relationship between the phase difference measurement errors of the AD8302 in the embodiment;
FIG. 3 is a schematic circuit diagram of an embodiment;
FIG. 4 is a schematic diagram of an improved phase difference measurement scheme in an embodiment;
fig. 5 is a schematic diagram of a correction scheme of the phase difference measurement result in the embodiment.
Detailed Description
The present invention will now be further illustrated, but not limited, by the following figures and examples.
Examples:
referring to fig. 3, a sine wave phase difference accurate measurement circuit comprises a first branch and a second branch, wherein the first branch comprises a first coupler, a first AGC amplitude automatic gain controller, a numerical control phase shifter, an AD8302 phase difference measurement module and a measurement result processing module which are sequentially connected, the second branch comprises a second coupler, a second AGC amplitude automatic gain controller, an AD8302 phase difference measurement module and a measurement result processing module which are sequentially connected, the first branch and the second branch share the AD8302 phase difference measurement module and the measurement result processing module, the output of the second AGC amplitude automatic gain controller is directly connected with the AD8302 phase difference measurement module, the measurement result processing module cpu is further electrically connected with the numerical control phase shifter, the phase difference of signals of the first branch and the second branch is measured by adopting a phase shifting mode, the method has the advantages that the defects of inherent binarycity, nonlinearity and error imbalance of a measuring chip are eliminated, positive and negative phase shifting can be realized by adopting a numerical control phase shifter, feedback is carried out through the measuring result of the AD8302, the moving direction and the moving size of the numerical control phase shifter are controlled, the nonlinear area of a phase measuring chip is avoided, the binarycity is removed, the error imbalance is avoided, the AD8302 measures the phase difference of two paths of measured signals, the phase difference is transmitted to a measuring result processing module in the form of a voltage value, the digital measuring result is obtained through 12-bit A/D conversion, an actual measuring value is obtained after calculation, and the digital measuring result and the actual measuring value can be directly transmitted to an upper computer for processing or directly transmitted to a phase control circuit for controlling the phase in real time.
The first coupler and the second coupler are low-coupling-degree low-insertion-loss couplers, so that the measurement result is ensured not to influence the communication of the main channel signals.
The AGC amplitude automatic gain controller limits the amplitude of the measured signal to a range acceptable to the AD8302 and remains unchanged.
The sine wave phase difference accurate measurement method comprises the sine wave phase difference accurate measurement circuit, and the method comprises the following steps:
1) Setting the number of phase shifters to 0, recording the original phase difference voltage value of the first branch signal B and the second branch signal A measured by the AD8302 as M0, wherein the original phase difference of two branches corresponding to the M0 is theta degrees, and at the moment, the theta has positive and negative possible values;
2) Controlling a numerical control phase shifter to increase the phase of the first branch signal B by 21 degrees, and recording the voltage value measured by the AD8302 as M1;
3) Setting the phase shifter degree to 0, and restoring the phase of the first branch signal B to the initial phase;
4) Controlling a numerical control phase shifter to reduce the phase of the first branch signal B by 21 degrees, and recording the voltage value measured by AD8302 as M2;
5) The absolute value of the difference between M1 and M0 and the absolute value of the difference between M2 and M0 are obtained through calculation, the direction with the larger absolute value is taken as a judgment basis to judge the growth slope of the phase difference, so that whether theta is positive or negative is known, and the accurate value of theta is measured, but the accurate value is not necessarily obtained;
6) If θ is not within the range of ± (60 °,120 °), the measured voltage value M0 has a larger error, and the numerical control phase shifter is controlled to increase the degree of the first branch signal B by Δθ, so as to move the phase difference between the first branch signal B and the second branch signal a within the range of ± (60 °,120 °), and the phase difference value becomes "θ+Δθ";
7) Because of the difference of measurement errors after interval conversion, the phase difference value of the first branch signal B and the second branch signal a needs to be measured again through the AD8302 to obtain a final phase difference voltage M3, and the degree corresponding to M3 at this time is recorded as θ ', θ' =θ+Δθ, because θ 'is in the range of ± (60 °,120 °), the error is smaller, and the value is relatively accurate, and because Δθ is also an accurate value, the original value "θ=θ' - Δθ" is also an accurate value.
Simulation experiment:
the measuring chip adopted in the experiment is AD8302 of Adenode company, the measuring frequency range of the chip is LP-2.7 GHz, the amplitude range of input signals is minus 60 dBm-0 dBm, the phase difference detection range is 0-180 degrees, the corresponding output voltage change range is 0V-1.8V, the sensitivity of the output voltage is 10 mV/degree, the measuring error is smaller than 0.5 degrees, and when the phase difference delta phi=0 degree, the output voltage is 1.8V; when ΔΦ=180°, the output voltage was 30mV.
As shown in fig. 1, the amplitude of the input signal is-30 dBm, the input frequency is respectively 100MHz, 900MHz, 1900MHz, 2200MHz and 2700MHz, it can be seen that the same voltage value of the output corresponds to two different phases, for example, the output voltage of 0.8V corresponds to-100 ° and 100 ° respectively, which has two values, meaning that the result obtained by directly measuring the two paths of phase difference by using a chip is equivalent to taking absolute value, and can not directly judge which path of phase is larger and which path of phase is smaller, in addition, the corresponding curves of the frequencies are slightly different, and are concentrated between the phase difference ranges (-20 °,20 °) and between the frequency measurement result curves of minus and plus (160 °,180 °), and the curve linearity is measured only between the frequency measurement results of minus or plus (20 °,160 °), which is relatively consistent, which is a problem of measuring nonlinearity;
as shown in fig. 2, the phase difference measurement result corresponding to the signal with the input amplitude of-30 dBm and the frequency of 2200MHz is represented, wherein the abscissa represents the phase difference, the left ordinate is the same as the measurement result voltage of fig. 4, the right ordinate is the magnitude of the error corresponding to each phase difference, the measurement error between ± (60 °,120 °) of the measurement result is close to 0 and is the smallest error interval, and the measurement error gradually increases beyond this range, so that the AD8302 has a problem of unbalanced measurement error.
In the embodiment, the coupler with lower coupling degree couples the signal on the main channel to the measuring circuit with weak power, so that the transmission power of the main channel is hardly affected; at this time, because the power of the detected signal is very small, the power amplification needs to be carried out, the AGC amplitude gain automatic control circuit can limit the signal amplitude to a value acceptable by a measuring chip, and no matter whether the power of the previous signal changes or not, the amplitude of the signal output after the AGC circuit is unchanged; the measured signal of the first branch is input to the measuring channel 1 of the AD8302 through the numerical control phase shifter, the measured signal of the second branch is directly input to the measuring channel 2 of the AD8302, in a measuring result processing module, the CPU controls the numerical control phase shifter to move, so as to eliminate the measurement binaryzation, nonlinearity and error imbalance of the AD8302 measuring chip, and finally the measuring result is reported to an upper computer or directly sent out to a phase control circuit at the rear end to serve as a reference.
In the example, the power of the signals output by the two paths of main signals after photoelectric conversion is 5dBm, and the frequency is 200MHz; the coupler is a connector of an SMA-KFD19C type three-joint, the coupling degree is-20 dB, the coupling output power is-15 dBm, the insertion loss is 0.3dB, the main signal output power is 4.7dBm, the AGC module uses a circuit with a main chip of AD8367, the input signal power can be stabilized to be-10 dBm output, the power supply is +/-5V, the types of the connectors at two ends are SMA-KFD19C type, the numerical control phase shifter module uses a main chip of PE44820, the phase shift can be controlled through a serial port or a parallel port, the phase shift range is (0 DEG, 358.6 DEG), the phase shift resolution is 1.4 DEG, the input-output interface connector is SMA-KFD19C type, the signal control and power interface connector is J30J-9ZKP type, the power supply is +5V, the phase measurement module main chip is AD8302, two interface connectors are SMA-KFD19C, the power supply is + -5V, the measurement result processing module main CPU chip is GD32F407VGT6, the measurement result input interface is a feedthrough capacitor, the communication interface model is J30J-15ZKP, +5V power supply, the module can obtain a phase measurement value through the A/D sampling port and convert the phase measurement value into a digital mode, the phase movement of the phase shifter can be controlled through a program output serial port instruction, information can be output to an upper computer or a phase control functional unit at the rear end, the modules are connected through a radio frequency cable with shielding, and the connector model is SMA-JB2.
The two curves shown in FIG. 4 are idealized graphs of FIG. 1, assuming that the first measured phase shifter has a phase shift value of 0 and the measured original phase difference has an output voltage of M, then according to the measured binary characteristic of FIG. 1, two points H and H 'are corresponding to two phase differences θ and θ' on a line segment with bilateral symmetry, and the two angles are symmetric by a positive angle and a negative angle of 0 °, and by observation, the slopes of the left line segment and the right line segment are different, the slope of the left line segment is positive, and the output voltage value increases with the phase differenceThe positive slope is increased, the negative slope is increased to the right, the output voltage value is reduced along with the increase of the phase difference, so that the slope property can be judged to fall on the line segment through the judgment of the movement of the H point, the measured signal B is increased by delta theta by adopting the numerical control phase shifter in fig. 3 on the assumption that the H point falls on the positive slope line segment on the left as shown in the left diagram of fig. 4, the phase difference is changed from theta to theta 1, and the measurement result is changed from M to M 1 The H point becomes H 1 Dots, here in two cases: one is H 1 Will not cross the 0 deg. peak, if M 1 >M, the measurement result increases along with the increase of the phase difference, so that the H can be proved to fall on the left positive slope line segment; another case, as noted in the left-hand diagram of FIG. 4, is H 1 Crossing the 0 ° peak, at which time M 1 The relationship with M is not readily determined, assuming Δm1=m-M 1 Then Δm1 may be positive or negative or 0, and the position of the H point cannot be determined by increasing θ, which may be determined by decreasing θ, decreasing Δθ by the measured signal B by the phase shifter, changing the phase difference from θ to θ2, and changing the measurement result from M to M 2 The H point becomes H 2 Point, let Δm2=m-M 2 At this time, Δm2 is necessarily>0, that is, the measurement result decreases with decreasing phase difference, so that it can be determined that H falls on the left positive slope line according to the slope relationship, and the same method can also determine whether H falls on the right negative slope line as in the right graph of fig. 4.
In the judging process, because the H point has the 'mountain turning' passing through the 0-degree peak point or the 'valley crossing' passing through the + -180-degree low valley point, the method is to judge two times, namely, increasing the delta theta degree for the first time and reducing the delta theta degree for the second time, obtaining the difference value of the two measurement results, and obtaining the delta M 1 And the absolute value of delta M2 is large, so that the concrete position of H can be judged by judging the slope of the signaling result, in order to avoid the mountain-turning and valley-crossing conditions of the H point as much as possible, delta theta is smaller, and experiments prove that 21 degrees are proper, and the binaryzation of the measurement result can be eliminated through the operation, so that the accurate measurement result is obtained.
It can be seen from FIG. 2 that the measurement error between + -60 deg. and 120 deg. is close to 0, which is the smallest error interval if it is usedThe phase difference of the measured signals is exactly within the range of the interval, that is, the measured result is a relatively accurate true value, if the measured result does not fall within the range, the measured error gradually increases towards the two poles, and if the measured result does not fall within the range, the original measured results H of B and A are assumed to fall outside minus plus or minus (60 degrees, 120 degrees), as shown in fig. 5, the phase shifter in fig. 3 can be used for shifting the phase so as to fall within minus or plus (60 degrees, 120 degrees), a relatively accurate result is obtained, then the phase shifting value is subtracted, and the phase difference is the true phase difference, taking the left diagram of fig. 5 as an example, if the H point falls on the left positive slope line segment, but falls on the interval (-180 degrees, minus 120 degrees), in order to measure the most accurate as far as possible, the H point is shifted to the middle of the interval (-120 degrees, -60 degrees), namely the-90 degrees point, the phase shifter is shifted rightwards by delta theta minus 90 degrees, namely, the H is shifted to the right 1 Point, measure H 1 The phase corresponding to the point is θ1, note that because of the measurement error θ1 in θ is not equal to-90 °, the final true phase difference should be "θ1- Δθ" i.e., "θ1+90° +θ".
Through the operation, the original value is moved to the section of + - (60 DEG, 120 DEG) for measurement, the range of nonlinear variation under different frequencies in fig. 1 is avoided, so that the measurement of different frequencies is also linear and accurate, and the section requirement of the minimum phase measurement error in fig. 2 is met.
The AD8302 chip is adopted to measure the phase difference, so that the defects of binaryzation, nonlinearity, error imbalance and the like exist, the signal transmission of a main channel cannot be influenced during measurement is ensured, the binaryzation of a measurement result is overcome to ensure the accuracy of the measurement result, and the nonlinearity and the error imbalance of the measurement result are overcome to ensure the accuracy of the measurement result.
Claims (1)
1. A sine wave phase difference accurate measurement method adopts a sine wave phase difference accurate measurement circuit, the sine wave phase difference accurate measurement circuit comprises:
the device comprises a first branch and a second branch, wherein the first branch comprises a first coupler, a first AGC amplitude automatic gain controller, a numerical control phase shifter, an AD8302 phase difference measuring module and a measuring result processing module which are sequentially connected, the second branch comprises a second coupler, a second AGC amplitude automatic gain controller, an AD8302 phase difference measuring module and a measuring result processing module which are sequentially connected, the AD8302 phase difference measuring module and the measuring result processing module are shared by the first branch and the second branch, the output of the second AGC amplitude automatic gain controller is directly connected with the AD8302 phase difference measuring module, and a measuring result processing module CPU is further electrically connected with the numerical control phase shifter;
the first coupler and the second coupler are low-coupling degree and low-insertion loss couplers;
the AGC amplitude automatic gain controller limits the amplitude of the detected signal to be within an acceptable range of the AD8302 and keeps unchanged;
characterized in that the method comprises the following steps:
1) Setting the digital control phase shifter degree to 0, recording the original phase difference voltage value of the first branch signal B and the second branch signal A measured by the AD8302 as M0, and recording the original phase difference of two branches corresponding to M0 as theta degrees, wherein the theta has positive and negative possible values;
2) Controlling a numerical control phase shifter to increase the phase of the first branch signal B by 21 degrees, and recording the voltage value measured by the AD8302 as M1;
3) Setting the digital control phase shifter degree to 0, and restoring the phase of the first branch signal B to the initial phase;
4) Controlling a numerical control phase shifter to reduce the phase of the first branch signal B by 21 degrees, and recording the voltage value measured by AD8302 as M2;
5) The absolute value of the difference between M1 and M0 and the absolute value of the difference between M2 and M0 are obtained through calculation, the direction with the larger absolute value is taken as a judgment basis to judge the growth slope of the phase difference, so that whether theta is positive or negative is known, and the accurate value of theta is measured, but the accurate value is not necessarily obtained;
6) If θ is not within the range of ± (60 °,120 °), the measured voltage value M0 has a larger error, and the digitally controlled phase shifter is controlled to increase the degree of the first branch signal B by Δθ, and the phase difference value becomes "θ+Δθ";
7) Because of the difference of measurement errors after interval conversion, the phase difference value of the first branch signal B and the second branch signal a is measured again through the AD8302, so as to obtain a final phase difference voltage M3, the degree corresponding to M3 is recorded as θ ', θ' =θ+Δθ, θ 'is an accurate value within the range of ± (60 °,120 °), Δθ is also an accurate value, and the original value "θ=θ' - Δθ" is also an accurate value.
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