CN110086342B - Switch converter and control method thereof - Google Patents
Switch converter and control method thereof Download PDFInfo
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- CN110086342B CN110086342B CN201910435277.XA CN201910435277A CN110086342B CN 110086342 B CN110086342 B CN 110086342B CN 201910435277 A CN201910435277 A CN 201910435277A CN 110086342 B CN110086342 B CN 110086342B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0048—Circuits or arrangements for reducing losses
- H02M1/0054—Transistor switching losses
- H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Power Engineering (AREA)
- Dc-Dc Converters (AREA)
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Abstract
The invention discloses a switching converter and a control method thereof, wherein the switching converter comprises an input power supply, an output voltage, a power supply common ground, a switching tube Q1, a switching tube Q2, a switching tube Q3, a switching tube Q4, an inductor L1 and a capacitor C1; the drain of the switching tube Q1 and the drain of the switching tube Q3 are connected to the positive input power supply, the source of the switching tube Q1 and the drain of the switching tube Q2 are connected to one end of an inductor L1, the source of the switching tube Q3 and the drain of the switching tube Q4 are connected to the other end of the inductor L1, the source of the switching tube Q4 is connected to one end of a capacitor C1, and the source of the switching tube Q2 and the other end of the capacitor C1 are connected to the power supply common ground. The invention realizes the reverse polarity of input and output voltage, and the opening of all the switching tubes ZVS, the efficiency is high; when the absolute value of the ratio of the input voltage to the output voltage is large, the inductor L1 can be demagnetized quickly, the current waveform of the inductor L1 is changed from a triangle to a quadrangle, and the high-frequency and high-efficiency operation of the switching converter is realized.
Description
Technical Field
The present invention relates to switching power supplies, and more particularly, to a switching converter circuit and a control method thereof.
Background
Fig. 1 shows a conventional Buck _ Boost circuit, which has a large current effective value when the circuit works in an intermittent mode, and a MOS transistor Q1 is hard-switched, so that the conduction loss of D1 is large.
Fig. 2 shows a CUK circuit, which is a fourth-order or even higher-order circuit, the dynamic process is complex, the output is easy to overshoot, the effective value of the current of the circuit is large when the circuit works in the discontinuous mode, the MOS transistor Q1 is hard switched, and the conduction loss of D1 is large.
The conventional Buck _ Boost circuit and the CUK circuit belong to reversed polarity circuits, and input and output voltages are reversed. However, the problems that the effective value of current is large and the conduction loss is large when the circuit works in an intermittent mode exist; the MOS transistor Q1 is hard-switched and has a large switching loss, and therefore, is not suitable for high-voltage input and high-frequency application. When the absolute value of the ratio of the discontinuous mode to the input/output voltage is greater than 2, the demagnetization time of the inductor L1 is too long, and the demagnetization time is proportional to the magnitude of the output current, which causes a problem that it is difficult to compromise between the large current output and the high frequency output.
Disclosure of Invention
In view of the technical defects of the existing reverse polarity circuit, the invention provides the switching converter circuit and the control mode thereof, the circuit works in the discontinuous mode to reduce the effective value of current, and the problem of large conduction loss is solved; all switching tubes realize ZVS (zero voltage switching) turn-on, and when the absolute value of the ratio of the discontinuous mode to the input-output voltage is larger than 2, the problems that the demagnetization time of the inductor L1 is too long, large current output is difficult and high frequency is difficult are solved.
In order to achieve the purpose, the invention adopts the following technical scheme:
a switching converter comprises an input power supply, an output voltage, a power supply common ground, a switching tube Q1, a switching tube Q2, a switching tube Q3, a switching tube Q4, an inductor L1 and a capacitor C1; the drain of the switching tube Q1 and the drain of the switching tube Q3 are connected to the positive input power supply, the source of the switching tube Q1 and the drain of the switching tube Q2 are connected to one end of an inductor L1, the source of the switching tube Q3 and the drain of the switching tube Q4 are connected to the other end of the inductor L1, the source of the switching tube Q4 is connected to one end of a capacitor C1, and the source of the switching tube Q2 and the other end of the capacitor C1 are connected to the power supply common ground.
Preferably, the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4 are MOS tubes, triodes or IGBTs.
The first control method of the switching converter includes:
when the previous period is finished, the switching tube Q1 is turned off, because the current of the inductor L1 is negative, the current IL of the inductor L1 charges the output capacitor Coss1 of the switching tube Q1, the output capacitor Coss2 of the switching tube Q2 discharges, the voltage at one end of the inductor L1 is reduced to 0V from Vin, and the switching tube Q2 realizes ZVS on; because the switching tube Q3 is in a conducting state, the voltage across the inductor L1 is Vin, the Vin voltage excites the inductor L1, the current IL of the inductor L1 rises, and the switching tube Q3 is turned off again according to the requirement of closed-loop control; the current IL of the inductor L1 charges an output capacitor Coss3 of the switching tube Q3 and discharges the output capacitor Coss4 of the switching tube Q4, so that the voltage at the other end of the inductor L1 is reduced from Vin to Vo, and the switching tube Q4 realizes ZVS switching-on; the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, then the switching tube Q2 is turned off according to the requirement of closed-loop control, the current IL of the inductor L1 charges the output capacitor Coss2 of the switching tube Q2 and discharges the output capacitor Coss1 of the switching tube Q1, so that the voltage at one end of the inductor L1 is increased to Vin from 0V, and the switching tube Q1 realizes ZVS on; the voltage at the two ends of the inductor L1 is Vin-Vo, the Vin-Vo voltage demagnetizes the inductor L1, the current IL of the inductor L1 drops to negative current, then the switch tube Q4 is turned off, the current IL of the inductor L1 charges the output capacitor Coss4 of the switch tube Q4, the output capacitor Coss3 of the switch tube Q3 is discharged, the voltage at the other end of the inductor L1 is increased to Vin from Vo, and the switch tube Q3 realizes ZVS on; therefore, ZVS (zero voltage switch) switching-on is realized in one period by the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4. And then the switching tube Q1 is turned off according to the requirement of closed-loop control, and the switching tube Q1 is turned on in the next period.
The working process is described in connection with fig. 5 as follows:
stage t 0-t 1: at time t0, switching tube Q2 is turned on, the voltage across inductor L1 is Vin, inductor L1 is excited, current IL of inductor L1 rises, and switching tube Q3 is turned off at time t 1;
stage t 1-t 2: after the switching tube Q3 is turned off, the current IL in the inductor L1 charges the output capacitor Coss3 of the switching tube Q3, and discharges the output capacitor Coss4 of the switching tube Q4. At the time t2, the voltage at the other end of the inductor L1 is reduced from Vin to Vo, and the switching tube Q4 realizes ZVS on;
stage t 2-t 3: the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, and the switching tube Q2 is turned off at the time t 3;
stage t 3-t 4: the current IL of the inductor L1 charges an output capacitor Coss2 of the switching tube Q2, discharges the output capacitor Coss1 of the switching tube Q1, the voltage at one end of the inductor L1 rises from 0V to Vin at the time of t4, and the switching tube Q1 realizes ZVS switching-on;
stage t 4-t 5: the current IL of the inductor L1 has one phase change, and the switching tube Q4 is turned off at the time t5 from positive to negative;
stage t 5-t 6: the current IL of the inductor L1 charges an output capacitor Coss4 of the switching tube Q4, discharges the output capacitor Coss3 of the switching tube Q3, the voltage at the other end of the inductor L1 rises from Vo to Vin at the time of t6, and the switching tube Q3 realizes ZVS on;
stage t 6-t 7: the voltage across the inductor L1 is Vin, and the voltage difference is zero, so the current IL of the inductor L1 remains unchanged, and the switching tube Q1 is turned off at time t 7;
t 7-t 0+ Tx stage: the current IL of the inductor L1 charges an output capacitor Coss1 of the switching tube Q1, the output capacitor Coss2 of the switching tube Q2 discharges, the voltage at one end of the inductor L1 is reduced to 0V from Vin at the time of t0+ Tx, and the switching tube Q2 realizes ZVS switching-on;
the cycle is ended and the next duty cycle is started and the above stages are repeated.
As an improvement of the first control method, when the load becomes light, the time from the switching transistor Q2 being turned on to the switching transistor Q4 being turned off starts to decrease, and the time from the switching transistor Q3 being turned on to the switching transistor Q1 being turned off becomes longer.
The second control method of the switching converter includes:
when the previous period is finished, the switching tube Q4 is turned off, and since the current of the inductor L1 is negative, the current IL of the inductor L1 charges the output capacitor Coss4 of the switching tube Q4, the output capacitor Coss3 of the switching tube Q3 discharges, the voltage at the other end of the inductor L1 rises from Vo to Vin, and the switching tube Q3 realizes ZVS switching-on; because the switching tube Q2 is in a conducting state, the voltage across the inductor L1 is Vin, the Vin voltage excites the inductor L1, the current IL of the inductor L1 rises, and the switching tube Q3 is turned off again according to the requirement of closed-loop control; the current IL of the inductor L1 charges an output capacitor Coss3 of the switching tube Q3 and discharges the output capacitor Coss4 of the switching tube Q4, so that the voltage at the other end of the inductor L1 is reduced from Vin to Vo, and the switching tube Q4 realizes ZVS switching-on; the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, then the switching tube Q2 is turned off according to the requirement of closed-loop control, the current IL of the inductor L1 charges the output capacitor Coss2 of the switching tube Q2 and discharges the output capacitor Coss1 of the switching tube Q1, so that the voltage at one end of the inductor L1 is increased to Vin from 0V, and the switching tube Q1 realizes ZVS on; the voltage at two ends of the inductor L1 is Vin-Vo, the Vin-Vo voltage demagnetizes the inductor L1, the current IL of the inductor L1 is reduced to zero, the switching tube Q1 is turned off, the output capacitor Coss1 of the switching tube Q1 starts to charge, the output capacitor Coss2 of the switching tube Q2 discharges, the current IL of the inductor L1 is reduced to negative current from zero, the voltage at one end of the inductor L1 is reduced to 0V from Vin, and the switching tube Q2 realizes ZVS on; therefore, ZVS (zero voltage switch) switching-on is realized in one period by the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4. And then the switching tube Q4 is turned off according to the requirement of closed-loop control, and the switching tube Q4 is turned on in the next period.
The working process is described in connection with fig. 6 as follows:
stage t 0-t 1: at time t0, switching tube Q3 is turned on, the voltage across inductor L1 is Vin, inductor L1 is excited, current IL of inductor L1 rises, and switching tube Q3 is turned off at time t 1;
stage t 1-t 2: after the switching tube Q3 is turned off, the current IL in the inductor L1 charges the output capacitor Coss3 of the switching tube Q3, and discharges the output capacitor Coss4 of the switching tube Q4. At the time t2, the voltage at the other end of the inductor L1 is reduced from Vin to Vo, and the switching tube Q4 realizes ZVS on;
stage t 2-t 3: the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, and the switching tube Q2 is turned off at the time t 3;
stage t 3-t 4: the current IL of the inductor L1 charges an output capacitor Coss2 of the switching tube Q2, discharges the output capacitor Coss1 of the switching tube Q1, the voltage at one end of the inductor L1 rises from 0V to Vin at the time of t4, and the switching tube Q1 realizes ZVS switching-on;
stage t 4-t 5: the current IL of the inductor L1 drops to zero at time t5, at which time the switching tube Q1 is turned off;
stage t 5-t 6: an output capacitor Coss1 of a switch tube Q1 is charged, an output capacitor Coss2 of a switch tube Q2 is discharged, the current IL of an inductor L1 is reduced from zero to negative current, the voltage at one end of the inductor L1 is reduced from Vin to 0V at the time of t6, and ZVS (zero voltage switching on) is realized by the switch tube Q2;
stage t 6-t 7: the voltage at the two ends of the inductor L1 is Vo, the Vo reversely excites the inductor L1, and the switching tube Q4 is turned off at the time t 7;
t 7-t 0+ Tx stage: the current of the inductor L1 discharges the output capacitor Coss3 of the switching tube Q3, the output capacitor Coss4 of the switching tube Q4 is charged, the voltage at the other end of the inductor L1 rises from Vo to Vin at the time of t0+ Tx, and the switching tube Q3 realizes ZVS switching-on;
the cycle is ended and the next duty cycle is started and the above stages are repeated.
As an improvement of the second control method, characterized in that: when the load becomes light, the time from the switching tube Q3 being turned on to the switching tube Q1 being turned off begins to decrease, and the time from the switching tube Q2 being turned on to the switching tube Q4 being turned off becomes longer.
The invention also provides another switch converter with the same inventive concept, and the technical scheme is as follows:
a switching converter comprises an input power supply, an output voltage, a power supply common ground, a diode D1, a switching tube Q2, a switching tube Q3, a switching tube Q4, an inductor L1 and a capacitor C1; the cathode of the diode D1 and the drain of the switching tube Q3 are connected to the positive input power supply, the anode of the diode D1 and the drain of the switching tube Q2 are connected to one end of the inductor L1, the source of the switching tube Q3 and the drain of the switching tube Q4 are connected to the other end of the inductor L1, the source of the switching tube Q4 is connected to one end of the capacitor C1, and the source of the switching tube Q2 and the other end of the capacitor C1 are connected to the power supply common ground.
Preferably, the switching tube Q2, the switching tube Q3 and the switching tube Q4 are MOS tubes, triodes or IGBTs.
The control method of the switching converter comprises the following steps:
when the previous period is finished, the switching tube Q4 is turned off, and since the current of the inductor L1 is negative, the current IL of the inductor L1 charges the output capacitor Coss4 of the switching tube Q4, the output capacitor Coss3 of the switching tube Q3 discharges, the voltage at the other end of the inductor L1 rises from Vo to Vin, and the switching tube Q3 realizes ZVS switching-on; because the switching tube Q2 is in a conducting state, the voltage across the inductor L1 is Vin, the Vin voltage excites the inductor L1, the current IL of the inductor L1 rises, and the switching tube Q3 is turned off again according to the requirement of closed-loop control; the current IL of the inductor L1 charges an output capacitor Coss3 of the switching tube Q3 and discharges the output capacitor Coss4 of the switching tube Q4, so that the voltage at the other end of the inductor L1 is reduced from Vin to Vo, and the switching tube Q4 realizes ZVS switching-on; the voltage at two ends of an inductor L1 is Vo, an inductor L1 is demagnetized, the current IL is reduced, then a switch tube Q2 is turned off according to the requirement of closed-loop control, the current IL of the inductor L1 charges an output capacitor Coss2 of the switch tube Q2, the voltage at one end of an inductor L1 is increased from 0V to Vin, the voltage at two ends of the inductor L1 is clamped by Vin-Vo, when the current IL is reduced to zero, the current IL is reversed, the output capacitor Coss2 of the switch tube Q2 starts discharging, the voltage at one end of the inductor L1 is reduced from Vin to 0V, and the ZVS is turned on by a switch tube Q2; therefore, ZVS (zero voltage switch) switching-on is realized in one period by the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4. And then the switching tube Q4 is turned off according to the requirement of closed-loop control, and the switching tube Q4 is turned on in the next period.
The working process is described in connection with fig. 8 as follows:
stage t 0-t 1: at time t0, switching tube Q3 is turned on, the voltage across inductor L1 is Vin, inductor L1 is excited, current IL of inductor L1 rises, and switching tube Q3 is turned off at time t 1;
stage t 1-t 2: after the switching tube Q3 is turned off, the current IL in the inductor L1 charges the output capacitor Coss3 of the switching tube Q3, and discharges the output capacitor Coss4 of the switching tube Q4. At the time t2, the voltage at the other end of the inductor L1 is reduced from Vin to Vo, and the switching tube Q4 realizes ZVS on;
stage t 2-t 3: the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, and the switching tube Q2 is turned off at the time t 3;
stage t 3-t 4: the current IL of the inductor L1 charges an output capacitor Coss2 of the switch tube Q2, the voltage at one end of the inductor L1 rises from 0V to Vin, the voltage at two ends of the inductor L1 is clamped by Vin-Vo, when the current IL drops to zero, the current IL is reversed, the output capacitor Coss2 of the switch tube Q2 starts to discharge, the voltage at one end of the inductor L1 drops from Vin to 0V at the time of t4, and the switch tube Q2 realizes ZVS on;
stage t 4-t 5: the voltage at the two ends of the inductor L1 is Vo, the Vo reversely excites the inductor L1, and the switching tube Q4 is turned off at the time t 5;
t 5-t 0+ Tx stage: the current IL of the inductor L1 charges the output capacitor Coss4 of the switching tube Q4, discharges the output capacitor Coss3 of the switching tube Q3, the voltage at the other end of the inductor L1 rises from Vo to Vin at the time t0+ Tx, and the switching tube Q3 realizes ZVS switching-on at the time t0+ Tx;
the cycle is ended and the next duty cycle is started and the above stages are repeated.
Description of the meaning of the terms:
drain electrode of the switching tube: for the MOS tube, a drain electrode is referred, for the triode, a collector electrode is referred, for the IGBT, a drain electrode is referred, and other switching tubes can correspond to each other according to the knowledge of a person skilled in the art and are not listed one by one;
source electrode of the switching tube: for the MOS transistor, the source electrode, the emitter electrode, and the source electrode, the other switching transistors may correspond to each other according to the knowledge of those skilled in the art, and are not listed.
Compared with the prior art, the invention has the following beneficial effects:
1) the circuit works in an intermittent mode, and ZVS (zero voltage switching) of all MOS (metal oxide semiconductor) tubes is realized;
2) the waveform of the inductor current is changed from a triangle to a quadrangle, the effective value of the inductor current is reduced under the same output power, the conduction loss is reduced, the efficiency is improved, and the large current output is easy to realize;
3) when the absolute value of the ratio of the input voltage to the output voltage is larger, the demagnetization time of the inductor L1 is greatly shortened by the turn-off of the MOS transistor Q2, high frequency is realized, and the inductance value and the capacitance value of the capacitor are reduced by the high frequency, so that the size of a power supply is reduced, and the cost is reduced.
Drawings
FIG. 1 is a schematic diagram of a conventional Buck _ Boost circuit;
FIG. 2 is a schematic diagram of a CUK circuit;
FIG. 3 is a schematic circuit diagram of a first embodiment of the present invention;
FIG. 4 is a graph of input-to-output voltage ratio versus switching frequency;
FIG. 5 is a timing diagram illustrating a first operation of the first embodiment of the present invention;
FIG. 6 is a timing diagram illustrating a second operation of the first embodiment of the present invention;
FIG. 7 is a schematic circuit diagram of a second embodiment of the present invention;
FIG. 8 is a timing diagram illustrating operation of the second embodiment of the present invention.
Detailed Description
First embodiment
Fig. 3 is a schematic circuit diagram of a first embodiment of the present invention. The power supply comprises an input power supply positive Vin, an output voltage negative Vo, a power supply common ground GND, a MOS tube Q1, a MOS tube Q2, a MOS tube Q3, a MOS tube Q4, an inductor L1 and a capacitor C1; the drain of the MOS transistor Q1 and the drain of the MOS transistor Q3 are connected to the input power positive Vin, the source of the MOS transistor Q1 and the drain of the MOS transistor Q2 are connected to one end of the inductor L1, the source of the MOS transistor Q3 and the drain of the MOS transistor Q4 are connected to the other end of the inductor L1, the source of the MOS transistor Q4 is connected to one end of the capacitor C1, and the source of the MOS transistor Q2 and the other end of the capacitor C1 are connected to the power common ground GND.
Coss1, Coss2, Coss3 and Coss4 in fig. 3 are output capacitances of the MOS transistor Q1, the MOS transistor Q2, the MOS transistor Q3 and the MOS transistor Q4, respectively, and fig. 3 also shows body diodes of the MOS transistor Q1, the MOS transistor Q2, the MOS transistor Q3 and the MOS transistor Q4.
It should be noted that: it is common practice for those skilled in the art to replace MOS transistor Q1, MOS transistor Q2, MOS transistor Q3, and MOS transistor Q4 with other types of switching transistors such as transistors and IGBTs.
Fig. 4 shows the waveform of the current IL in the inductor L1 and the output current Io when the conventional Buck-Boost circuit operates in the discontinuous mode,
according to the formula
The rising slope of the current IL is
The falling slope of the current IL is
The rise and fall times of the current IL are the same, corresponding to a duty cycle T1.
When in use
When the rising slope of the current IL is
The falling slope of the current IL is
The inductance of inductor L1 is changed to make the rising slope of current IL sum
The rising slope of the current is the same, the falling time of the current IL is
The working period is T2 which is 2 times of the falling time of the time current IL and is larger than T1.
When in use
When the rising slope of the current IL is
The falling slope of the current IL is
The inductance of inductor L1 is changed to make the rising slope of current IL sum
The rising slope of the current is the same, the falling time of the current IL is
The working period is T3 and is larger than T2, which is 4 times of the falling time of the current IL.
Therefore, in the discontinuous mode, the larger the absolute value of the ratio of the input voltage to the output voltage is, the larger the corresponding switching period is, the smaller the frequency is, the more difficult the high frequency is to be realized, and when the absolute value of the ratio of the input voltage to the output voltage is greater than 2, the better beneficial effect of the invention can be ensured.
Fig. 5 shows a first operation sequence of the first embodiment for a switching converter with Vin voltage of 75V, Vo voltage of minus 12V, inductor L1 of 1uH, and output current of 20A, which is specifically as follows:
stage t 0-t 1: at time t0, MOS transistor Q2 is turned on, the voltage across inductor L1 is Vin, inductor L1 is excited, current IL of inductor L1 rises, and MOS transistor Q3 is turned off at time t 1;
stage t 1-t 2: after the MOS transistor Q3 is turned off, the current IL in the inductor L1 charges the output capacitor Coss3 of the MOS transistor Q3 and discharges the output capacitor Coss4 of the MOS transistor Q4. At time t2, the voltage of the circuit node SW2 (i.e., at the other end of the inductor L1) is reduced from Vin to Vo, and the MOS transistor Q4 realizes ZVS switching on;
stage t 2-t 3: the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, and the MOS transistor Q2 is turned off at the time t 3;
stage t 3-t 4: the current IL of the inductor L1 charges the output capacitor Coss2 of the MOS transistor Q2, discharges the output capacitor Coss1 of the MOS transistor Q1, and at time t4, the voltage of the circuit node SW1 (i.e., one end of the inductor L1) rises from 0V to Vin, so that the MOS transistor Q1 realizes ZVS switching on;
stage t 4-t 5: the current IL of the inductor L1 has one phase change, and the MOS tube Q4 is turned off at the time t5 from positive to negative;
stage t 5-t 6: the current IL of the inductor L1 charges the output capacitor Coss4 of the MOS transistor Q4, discharges the output capacitor Coss3 of the MOS transistor Q3, and at time t6, the voltage of the circuit node SW2 (i.e., the other end of the inductor L1) rises from Vo to Vin, so that the MOS transistor Q3 realizes ZVS switching on;
stage t 6-t 7: the voltage across the inductor L1 is Vin, and the voltage difference is zero, so the current IL of the inductor L1 remains unchanged, and the MOS transistor Q1 is turned off at time t 7;
t 7-t 0+ Tx stage: the current IL of the inductor L1 charges the output capacitor Coss1 of the MOS transistor Q1, the output capacitor Coss2 of the MOS transistor Q2 discharges, and at the time t0+ Tx, the voltage of the circuit node SW1 (i.e., one end of the inductor L1) is reduced from Vin to 0V, and the MOS transistor Q2 realizes ZVS switching-on;
the cycle is ended and the next duty cycle is started and the above stages are repeated.
As an improvement of the first control method, characterized in that: when the load becomes light, the stages t 0-t 1, t 2-t 3 and t 4-t 5 start to decrease, and the stages t 6-t 7 become longer. The decreasing value or increasing value of each stage is related to the input-output voltage value, the inductance of inductor L1, the setting of the optimum efficiency point, the switching frequency, etc., and a trend is illustrated here.
The cycle is ended and the next duty cycle is started and the above stages are repeated.
Since the circuit operates periodically, Tx in t0+ Tx mentioned above means a time length of X cycles.
It can be seen from figure 5 that the waveform of the current IL of inductor L1 is quadrilateral,under the same output power, compared with the triangular waveform in the prior art, the peak value of the inductive current is reduced, the effective value is reduced, so the conduction loss is reduced, the efficiency is improved, and the formula is shown in the specification
Obtaining L & ltdi & gt & ltN & gt dB & ltAe & gt, reducing di by the current peak value, and reducing the effective sectional area Ae of the magnetic core of the inductor under the condition that the inductance L of the inductor, the turn number N of the inductor and the dB of the magnetic core are not changed, so that the size of the magnetic core is reduced; if di is reduced under the same output ripple requirement, the capacitance value of the required filter capacitor is reduced, and the size of the capacitor is reduced; the switching-off of the MOS tube Q2 greatly shortens the demagnetization time of the inductor L1, realizes high frequency, and further reduces the inductance value and the capacitance value of the capacitor due to the high frequency; the size of the power supply is reduced, and the cost is reduced.
Fig. 6 shows a second operation sequence of the first embodiment, which is as follows:
stage t 0-t 1: at time t0, MOS transistor Q3 is turned on, the voltage across inductor L1 is Vin, inductor L1 is excited, current IL of inductor L1 rises, and MOS transistor Q3 is turned off at time t 1;
stage t 1-t 2: after the MOS transistor Q3 is turned off, the current IL in the inductor L1 charges the output capacitor Coss3 of the MOS transistor Q3 and discharges the output capacitor Coss4 of the MOS transistor Q4. At time t2, the voltage of the circuit node SW2 (i.e., at the other end of the inductor L1) is reduced from Vin to Vo, and the MOS transistor Q4 realizes ZVS switching on;
stage t 2-t 3: the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, and the MOS transistor Q2 is turned off at the time t 3;
stage t 3-t 4: the current IL of the inductor L1 charges the output capacitor Coss2 of the MOS transistor Q2, discharges the output capacitor Coss1 of the MOS transistor Q1, and at time t4, the voltage of the circuit node SW1 (i.e., one end of the inductor L1) rises from 0V to Vin, so that the MOS transistor Q1 realizes ZVS switching on;
stage t 4-t 5: the current IL of the inductor L1 drops to zero at time t5, at which time the MOS transistor Q1 is turned off;
stage t 5-t 6: an output capacitor Coss1 of the MOS transistor Q1 is charged, an output capacitor Coss2 of the MOS transistor Q2 is discharged, the current IL of the inductor L1 is reduced from zero to negative current, the voltage of a circuit node SW1 (namely one end of the inductor L1) is reduced from Vin to 0V at the time t6, and the ZVS on of the MOS transistor Q2 is realized;
stage t 6-t 7: the voltage at the two ends of the inductor L1 is Vo, the Vo reversely excites the inductor L1, and the MOS tube Q4 is turned off at the time t 7;
t 7-t 0+ Tx stage: the current of the inductor L1 discharges the output capacitor Coss3 of the MOS transistor Q3, charges the output capacitor Coss4 of the MOS transistor Q4, the voltage of the circuit node SW2 (namely the other end of the inductor L1) rises from Vo to Vin at the time t0+ Tx, and the MOS transistor Q3 realizes ZVS opening;
the cycle is ended and the next duty cycle is started and the above stages are repeated.
As can be seen from fig. 6, the waveform of the current IL of the inductor L1 is also quadrilateral, and the object of the invention is also achieved.
It should be noted that, in addition to the switching converter having Vin voltage of 75V, Vo voltage of minus 12V, inductor L1 of 1uH, and output current of 20A, the switching converter having other parameters has a similar operation timing chart, and the waveform of current IL of inductor L1 is also a quadrangle, which is different in amplitude at each time point.
In addition, both the above two operation sequences are for the application scenario when the load is fully loaded, and in the actual application scenario, the load often becomes lighter, and at this time, the efficiency of the circuit at light load can be improved by mode switching, and the improvement method is as follows:
1. when the load becomes light (namely the output current is reduced to a certain value), the drive of the Q1 is closed, the drive loss is reduced, and the efficiency is improved;
2. when the load becomes lighter (i.e., when the output current becomes smaller to a certain value), the phases t0 to t1, t2 to t3 and t4 to t5 are reduced more, and the phases t6 to t7 are lengthened more, so that the total switching period is maintained substantially unchanged. But the effective current IL time is reduced, so that the effective value of the current IL is larger, the conduction loss is larger, the Q2 is in a continuous conduction state at this time, the driving loss is further reduced, the circuit is changed into a traditional Buck _ Boost circuit, the MOS tube Q4 realizes a synchronous rectification function, the waveform of the current IL of the inductor L1 is changed into a common triangle from a quadrilateral plus a longer t 6-t 7 stage, the effective value of the current IL is reduced under the same output current, and the efficiency is improved.
Second embodiment
Fig. 7 is a circuit schematic of a second embodiment of the present invention. On the basis of the first embodiment, the MOS transistor Q1 is replaced by a diode D1, a cathode of the diode D1 is connected to a drain of the MOS transistor Q3 and the input power positive Vin, and an anode of the diode D1 is connected to a drain of the MOS transistor Q2 and one end of an inductor L1.
The time for the diode D1 to flow current is relatively short, compared with the MOS tube scheme, the conduction loss is not increased too much, one-way floating drive is omitted, the drive loss is reduced, the drive circuit is simplified, and the LED driving circuit is suitable for medium and small current output scenes.
The present embodiment can also obtain better implementation effect when the absolute value of the ratio of the input voltage to the output voltage is greater than 2, and for the switching converter with Vin voltage of 75V, Vo voltage of negative 12V, inductor L1 of 1uH, and output current of 20A, fig. 8 shows the operation timing sequence of the second embodiment, which is as follows:
stage t 0-t 1: at time t0, MOS transistor Q3 is turned on, the voltage across inductor L1 is Vin, inductor L1 is excited, current IL of inductor L1 rises, and MOS transistor Q3 is turned off at time t 1;
stage t 1-t 2: after the MOS transistor Q3 is turned off, the current IL in the inductor L1 charges the output capacitor Coss3 of the MOS transistor Q3 and discharges the output capacitor Coss4 of the MOS transistor Q4. At time t2, the voltage of the circuit node SW2 (i.e., at the other end of the inductor L1) is reduced from Vin to Vo, and the MOS transistor Q4 realizes ZVS switching on;
stage t 2-t 3: the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, and the MOS transistor Q2 is turned off at the time t 3;
stage t 3-t 4: the current IL of the inductor L1 charges an output capacitor Coss2 of the MOS transistor Q2, the voltage of a circuit node SW1 (namely one end of the inductor L1) rises to Vin from 0V, the voltage at two ends of the inductor L1 is clamped by Vin-Vo, when the current IL drops to zero, the current IL is reversed, the output capacitor Coss2 of the MOS transistor Q2 starts to discharge, the voltage of a circuit node SW1 (namely one end of the inductor L1) drops to 0V from Vin at the time of t4, and the ZVS on of the MOS transistor Q2 is realized;
stage t 4-t 5: the voltage at the two ends of the inductor L1 is Vo, the Vo reversely excites the inductor L1, and the MOS tube Q4 is turned off at the time t 5;
t 5-t 0+ Tx stage: the current IL of the inductor L1 charges the output capacitor Coss4 of the MOS transistor Q4, discharges the output capacitor Coss3 of the MOS transistor Q3, and increases the voltage of the circuit node SW2 (i.e., the other end of the inductor L1) from Vo to Vin at time t0+ Tx, so that the MOS transistor Q3 realizes ZVS switching on at time t0+ Tx;
the cycle is ended and the next duty cycle is started and the above stages are repeated.
As can be seen from fig. 8, the waveform of the current IL of the inductor L1 is also quadrilateral, and the object of the invention is also achieved.
In this embodiment, switching converters with other parameters may also be selected, and the efficiency of the circuit may also be improved by the mode switching, which is not described herein.
The above embodiments should not be construed as limiting the present invention, and the scope of the present invention should be determined by the scope of the appended claims. It will be apparent to those skilled in the art that many equivalent substitutions, modifications and alterations can be made without departing from the spirit and scope of the invention, such as fine tuning of the circuit by simple series-parallel connection of devices, etc., depending on the application, and such modifications and alterations should also be considered as the scope of the invention.
Claims (3)
1. A control method of a switching converter is used for controlling the switching converter, and the switching converter comprises an input power supply positive voltage, an output voltage negative voltage, a power supply common ground, a switching tube Q1, a switching tube Q2, a switching tube Q3, a switching tube Q4, an inductor L1 and a capacitor C1; the switch tube Q1 is connected in parallel with the output capacitor Coss1, the switch tube Q2 is connected in parallel with the output capacitor Coss2, the switch tube Q3 is connected in parallel with the output capacitor Coss3, the switch tube Q4 is connected in parallel with the output capacitor Coss4, the drain of the switch tube Q1 and the drain of the switch tube Q3 are connected to the positive input power supply, the source of the switch tube Q1 and the drain of the switch tube Q2 are connected to one end of the inductor L1, the source of the switch tube Q3 and the drain of the switch tube Q4 are connected to the other end of the inductor L1, the source of the switch tube Q4 is connected to one end of the capacitor C1, and the source of the switch tube Q2 and the other end of the capacitor C1;
the method is characterized in that: when the previous period is finished, the switching tube Q1 is turned off, because the current of the inductor L1 is negative, the current IL of the inductor L1 charges the output capacitor Coss1 of the switching tube Q1 and discharges the output capacitor Coss2 of the switching tube Q2, the voltage at one end of the inductor L1 is reduced to 0V from Vin, and the switching tube Q2 realizes ZVS on; because the switching tube Q3 is in a conducting state, the voltage across the inductor L1 is Vin, the Vin voltage excites the inductor L1, the current IL of the inductor L1 rises, and the switching tube Q3 is turned off again according to the requirement of closed-loop control; the current IL of the inductor L1 charges an output capacitor Coss3 of the switching tube Q3 and discharges the output capacitor Coss4 of the switching tube Q4, so that the voltage at the other end of the inductor L1 is reduced from Vin to Vo, and the switching tube Q4 realizes ZVS switching-on; the voltage at the two ends of the inductor L1 is Vo, the inductor L1 is demagnetized, the current IL is reduced, then the switching tube Q2 is turned off according to the requirement of closed-loop control, the current IL of the inductor L1 charges the output capacitor Coss2 of the switching tube Q2 and discharges the output capacitor Coss1 of the switching tube Q1, so that the voltage at one end of the inductor L1 is increased to Vin from 0V, and the switching tube Q1 realizes ZVS on; the voltage at the two ends of the inductor L1 is Vin-Vo, the Vin-Vo voltage demagnetizes the inductor L1, the current IL of the inductor L1 drops to negative current, then the switch tube Q4 is turned off, the current IL of the inductor L1 charges the output capacitor Coss4 of the switch tube Q4, the output capacitor Coss3 of the switch tube Q3 is discharged, the voltage at the other end of the inductor L1 is increased to Vin from Vo, and the switch tube Q3 realizes ZVS on; therefore, ZVS (zero voltage switching) switching-on is realized in one period by the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4; and then the switching tube Q1 is turned off according to the requirement of closed-loop control, and the switching tube Q1 is turned on in the next period.
2. The control method of the switching converter according to claim 1, characterized in that: the switching tube Q1, the switching tube Q2, the switching tube Q3 and the switching tube Q4 are MOS tubes, triodes or IGBTs.
3. The control method of the switching converter according to claim 1 or 2, characterized in that: when the load becomes light, the time from the switching tube Q2 being turned on to the switching tube Q4 being turned off begins to decrease, and the time from the switching tube Q3 being turned on to the switching tube Q1 being turned off becomes longer.
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| CN110661402B (en) * | 2019-09-29 | 2021-03-02 | 广州金升阳科技有限公司 | Self-adaptive ZVS circuit and control method thereof |
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| CN110011537A (en) * | 2019-05-09 | 2019-07-12 | 广州金升阳科技有限公司 | A kind of switch converters and its control method |
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| US9654003B1 (en) * | 2015-12-29 | 2017-05-16 | Texas Instruments Incorporated | Methods and apparatus for resonant energy minimization in zero voltage transition power converters |
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| US10177658B2 (en) * | 2016-04-14 | 2019-01-08 | Texas Instruments Incorporated | Methods and apparatus for adaptive timing for zero voltage transition power converters |
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| CN110011537A (en) * | 2019-05-09 | 2019-07-12 | 广州金升阳科技有限公司 | A kind of switch converters and its control method |
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