CN107438047A - The phase noise based on decision-feedback corrects compensation method certainly in a kind of single-carrier frequency domain equalization system - Google Patents
The phase noise based on decision-feedback corrects compensation method certainly in a kind of single-carrier frequency domain equalization system Download PDFInfo
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- H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
- H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
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- H04L27/3872—Compensation for phase rotation in the demodulated signal
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Abstract
The invention discloses the phase noise based on decision-feedback in a kind of single-carrier frequency domain equalization system from compensation method is corrected, it is related to wireless communication field；The present invention carries out phase noise rough estimate using pilot tone in receiving terminal to reception signal, complete the coarse compensation of phase noise, SC FDE demodulation is carried out after obtaining the signal after coarse compensation, planetary then is entered to decision-feedback signal and remaps processing, decision-feedback phase noise is estimated；And decision-feedback phase noise is corrected as threshold value using pilot point phase noise, the signal finally received using phase noise compensation after amendment, and obtained thermal compensation signal is demodulated and adjudicated.The present invention realizes simply have relatively low PAPR, be advantageous to the miniaturization and low power consumption of communication terminal in transmitting terminal；Extra processing unit need not be added in receiving terminal, suitable for a variety of phase noise models, effectively suppresses influence of the phase noise to error rate of system performance, improves the bit error rate performance of communication system.
Description
Technical field
The present invention relates to wireless communication field, based on judgement in specifically a kind of single-carrier frequency domain equalization system (SC-FDE)
The phase noise of feedback corrects compensation method certainly.
Background technology
In a wireless communication system, information reaches receiver by the electromagnetic wave modulated in space propagation.With wireless
The continuous development of communication, it is desirable to which communication terminal has the characteristics that miniaturization and low-power consumption.
SC-FDE is attracted wide attention with the signal processing mode of similar OFDM (OFDM) technology.SC-FDE
Using low-complexity frequency domain balancing technique, ability of anti-multipath is suitable with OFDM, but because SC-FDE is not carried out in transmitting terminal
IFFT processing, power peak and the average ratio (PAPR) of transmission signal are less than OFDM, therefore can use single carrier is ripe to penetrate
Frequency technology, reduce the cost of the nonlinear devices such as power amplifier.SC-FDE typically combines with quadrature amplitude modulation (QAM) to be made
With transmitting terminal only need to carry out constellation mapping to data, it is not necessary to other complicated modulation, there is relatively low system complexity, this
Comply fully with the demand of wireless terminal miniaturization.Therefore need to carry out more application of the SC-FDE technologies in radio communication scene
Add in-depth study.
In a wireless communication system, transmitting terminal and receiving terminal be required for producing corresponding carrier wave with complete corresponding radio frequency with
Frequency spectrum conversion between base band.But there is certain difference in the oscillator of generation carrier wave, cause carrier frequency with phase-locked loop
The random difference of short time with target frequency be present, in turn result in caused signal and random phase saltus step occurs, show as phase
Position noise.And as the rise of communications band, phase noise are obvious all the more to the performance impact of system.
Rare due to low-frequency range frequency spectrum resource, radio communication develops towards higher frequency range, this cause phase noise into
To influence the very important factor of error rate of system performance；Communication terminal then develops towards the direction of miniaturization, low-power consumption, SC-
FDE systems have relatively low PAPR and relatively low system complexity, can meet the needs of terminal low-power consumption, miniaturization simultaneously.Cause
This needs to study the phase noise compensation method for being suitable for SC-FDE systems.
The content of the invention
The present invention can influence the bit error rate performance of SC-FDE systems this problem for phase noise, in order to effectively be lifted
The bit error rate performance of SC-FDE systems, there is provided the phase noise based on decision-feedback corrects compensation certainly in a kind of SC-FDE systems
Method；
This method carries out phase noise rough estimate using pilot tone in receiving terminal to reception signal first, completes phase noise
Coarse compensation；SC-FDE demodulation is carried out after obtaining the signal after coarse compensation；Then planetary is entered to decision-feedback signal and remaps place
Reason, estimates decision-feedback phase noise；And by the use of pilot point phase noise as threshold value to decision-feedback phase noise
Corrected, the signal finally received using phase noise compensation after amendment, and obtained thermal compensation signal is demodulated and
Judgement.
Comprise the following steps that：
Step 1: for some Bitstream signal of wireless communication system transmitting terminal, obtained using 16QAM map modulations
NsymIndividual 16QAM symbols；
16QAM symbols represent as follows：pk, k=1,2 ..., Nsym
Step 2: 16QAM symbols are divided into data block by SC-FDE systems, and insertion is only respectively in each data block head and the tail
Special word, form sub-block；
The length of each data block forms by 496 16QAM symbols；The length of each unique word is 16；Each data
Block forms a sub-block together with end to end unique word therewith；Each sub-block has 528 numerical chracters；
Step 3: the numerical chracter of each sub-block is radiation-curable for antenna after the conversion of D/A modular converters respectively
Analog signal；
For m-th of numerical chracter s on i-th of sub-block after addition unique wordi,mIncluding three parts：
Part I is m-th of unique word u of the sub-blocki,m；Part II is m-th of transmission of the sub-block
Symbol zi,m；Part III is m-th of pilot point p of the sub-blocki,m；NuwRepresent the length of unique word；NdRepresent the data
The length of block；Pi represents the spacing value of two neighboring pilot point；Pn represents the quantity of pilot point；
Step 4: each analog signal reaches receiving terminal by white Gaussian noise, it is converted into respectively by A/D modular converters
The time-domain signal of receiving terminal；
Time-domain signal r corresponding to m-th of numerical chracter in i-th of sub-block structurei,mIt is as follows：
For the phase noise of m-th of numerical chracter on i-th of sub-block of transmitting terminal, ni,mFor i-th of sub-block
White Gaussian noise component on upper m-th of time-domain signal；
Step 5: being directed to i-th of sub-block, receiving terminal carries out phase noise rough estimate using pilot linear interpolation, obtains
The rough estimate phase noise of each numerical chracter into the sub-block.
Head and the tail numerical chracter phase noise first in i-th of sub-block of rough estimate, it is specially：
For 16 first unique words of the sub-block, the phase noise average value of 16 symbols is taken as i-th of subnumber
According to the phase noise of first symbol of first unique word of block；Similarly, phase corresponding to 16 symbols of the tail unique word of the sub-block
Position noise, the phase noise averaged as the tail unique word last symbol of the sub-block；
The phase noise of each symbol is calculated as follows in unique word：
δ2Represent the power of white Gaussian noise；
The phase noise of remaining 526 bit digital symbol of i-th of sub-block of rough estimate, it is specially：
First, the phase noise of each pilot point in the sub-block is calculated, then, using pilot point linear interpolation method, slightly
Estimate the phase noise of each numerical chracter between two neighboring pilot point.
The phase noise of each pilot point is calculated as follows：
Step 6: i-th of sub-block is directed to, using the rough estimate phase noise of each numerical chracter to respective time domain
Signal carries out phase noise compensation, obtains the data signal after each numerical chracter coarse compensation.
Data signal after m-th of numerical chracter coarse compensation of i-th of sub-blockIt is as follows：
Represent the conjugation of m-th of numerical chracter rough estimate phase noise of i-th of sub-block；
Step 7: carrying out SC-FDE demodulation to the data signal after each coarse compensation of i-th of sub-block, demodulated
Signal afterwards.
SC-FDE demodulation includes FFT, MMSE frequency domain equalizations and IFFT；
Step 8: being carried out to signal of all sub-blocks after SC-FDE is demodulated from correcting, the phase from after correcting is obtained
Position noise；
Comprise the following steps that：
Step 801, by after SC-FDE is demodulated signal carry out 16QAM demappings obtain binary bit stream；
Step 802, binary bit stream used into 16QAM map modulations, the 16QAM symbols after being compensated again；
Step 803, the 16QAM symbols after compensation are divided into data block, insertion is unique respectively in each data block head and the tail
Word, form the sub-block after compensation；
Step 804, for just compensation after i-th of sub-block, m-th of numerical chracterWith reference to the time domain before rough estimate
Signal ri,mPhase noise is tried to achieve from the feedback phase noise delta Φ in rectification module~ i,m；
Utilize the numerical chracter after first compensationWith time-domain signal ri,mI-th of sub-block, m-th of feedback phase is asked for make an uproar
Sound ΔΦ~ i,m。
* conjugation is represented.
Step 805, in each pilot point of i-th of sub-block, 4 are obtained from correction threshold values after just compensating；
4 are respectively from correction threshold value：The phase minimum Φ of pilot point phase noisemin,p,i,m, pilot point phase noise
Phase maximum Φmax,p,i,m, pilot point phase noise amplitude minimum value | Φ |min,p,i,mWith the width of pilot point phase noise
It is worth maximum
|Φ|max,p,i,m。
Step 806, the phase noise point using 4 mistaken verdicts from each feedback phase noise of correction threshold value searching.
Detailed process is as follows：
When the phase of certain feedback phase noise is less than the phase minimum Φ of pilot point phase noisemin,p,i,m, or be more than
The phase maximum Φ of pilot point phase noisemax,p,i,m, or the amplitude of the feedback phase noise is less than pilot point phase noise
Amplitude minimum value | Φ |min,p,i,m, or amplitude is more than the amplitude maximum of pilot point phase noise | Φ |max,p,i,m, then should
Feedback phase noise is the phase noise point of mistaken verdict.
Step 807, the phase noise point for mistaken verdict, replaced according to alternative rule from corresponding phase noise value
Generation.
Detailed process is as follows：
Substitute for the first time as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is wrong
During the phase noise point that erroneous judgement is determined, substituted with the phase noise of previous numerical chracter；
Second of replacement is as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is wrong
During the phase noise point that erroneous judgement is determined, it is averaged with the phase noise of previous numerical chracter and the phase noise of the latter numerical chracter
Value substitutes；
Third time substitutes as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is wrong
During the phase noise point that erroneous judgement is determined, it is averaged with the phase noise of the first two numerical chracter and the phase noise of final two digits symbol
Value substitutes；
By that analogy；
Step 808, using the time-domain signal before each phase noise after correction and each self-corresponding rough estimate, try to achieve from rectifying
Data signal after positive compensation simultaneously carries out SC-FDE demodulation, return to step 801；
With the increase of iterations, the phase noise of the symbol of mistaken verdict is corrected, and iterations is more, and mistake is sentenced
Point certainly is fewer, and error rate of system performance gets a promotion therewith.
The advantage of the invention is that：
1), the phase noise based on decision-feedback from backoff algorithm is corrected, is being sent out in a kind of single-carrier frequency domain equalization system
Sending end is realized simply, and has relatively low PAPR, is advantageous to the miniaturization and low power consumption of communication terminal；
2), the phase noise based on decision-feedback from backoff algorithm is corrected, is connecing in a kind of single-carrier frequency domain equalization system
Receiving end need not add extra processing unit, by making decisions feedback to demodulated signal from correction process, can be applied to
A variety of phase noise models, it can effectively suppress influence of the phase noise to error rate of system performance, greatly improve communication system
Bit error rate performance.
Brief description of the drawings
Fig. 1 is the schematic diagram that the phase noise based on decision-feedback corrects backoff algorithm certainly in a kind of SC-FDE of the present invention；
Fig. 2 is the flow chart that the phase noise based on decision-feedback corrects backoff algorithm certainly in a kind of SC-FDE of the present invention；
Fig. 3 is SC-FDE systems division data block of the present invention and inserts the schematic diagram that unique word forms sub-block；
Fig. 4 is that the present invention is carried out to all sub-blocks from the algorithm flow chart corrected；
Fig. 5 is performance map of the present invention using the SC-FDE bit error rates under 16QAM modulation systems contrast different situations.
Embodiment
The specific implementation method of the present invention is described in detail below in conjunction with the accompanying drawings.
As shown in figure 1, the present invention, for certain transmission bit signal, it is binary bit stream to be located at the form at A, is passed through
The 16QAM symbols at B are obtained after 16QAM constellation mappings；16QAM symbols are divided into data block by SC-FDE systems, and each
The unique word that data block head and the tail insertion identical length is 16, obtains the sub-block that length is 512；Sub-block passes through D/A
Module and awgn channel reach receiving terminal；Receiving terminal obtains analog signal in the time-domain digital letter at C after the conversion of A/D modules
Number；
Time-domain digital signal at C obtains the coarse compensation phase noise at D after pilot phase noise coarse compensation, thick to mend
Repay the time-domain digital signal at phase noise binding signal C and data signal at receiving terminal E after coarse compensation is calculated, pass through successively
The frequency domain representation of time-domain signal after coarse compensation at F is obtained after crossing FFT module, frequency after the equilibrium at G is obtained after frequency domain equalization
Domain signal, the SC-FDE demodulated signals at H are obtained after IFFT modules, two are obtained after hard decision is 16QAM demappings
System bit stream, again using 16QAM map modulations, obtain the 16QAM symbols after being compensated at J；Reinsert unique word and obtain K
Sub-block after place's compensation, revised phase noise at L is obtained from after correcting by phase noise.
The time-domain signal at phase noise compensation C corrected at L is compensated rear signal, returns at E；Pass through SC- successively
FDE is demodulated, loop iteration.
As shown in Fig. 2 comprise the following steps that：
Step 1: for some Bitstream signal of wireless communication system transmitting terminal, obtained using 16QAM map modulations
NsymIndividual 16QAM symbols；
16QAM symbols represent as follows：pk, k=1,2 ..., Nsym
Step 2: 16QAM symbols are divided into data block by SC-FDE systems, and insertion is only respectively in each data block head and the tail
Special word, form sub-block；
Ideally, the frequency spectrum of unique word sequence should have impartial or approximate impartial amplitude in all frequencies,
With produce wider bandwidth, stable frequency response can be uniformly subject to each frequency content ensured in channel
Detection.Block data structure after addition unique word is as shown in figure 3, each the length of data block is by 496 16QAM set of symbols
Into；The length of each unique word is 16；Each data block forms a sub-block together with end to end unique word therewith；Often
Individual sub-block has 528 numerical chracters.
Step 3: the numerical chracter of each sub-block is radiation-curable for antenna after the conversion of D/A modular converters respectively
Analog signal；
Baseband digital signal is transformed to the radiation-curable analog signal of antenna by D/A modular converters.
For m-th of numerical chracter s on i-th of sub-block after addition unique wordi,mIncluding three parts：
Part I is m-th of unique word u of the sub-blocki,m；Part II is m-th of transmission of the sub-block
Symbol zi,m；Part III is m-th of pilot point p of the sub-blocki,m；NuwRepresent the length of unique word；NdRepresent the data
The length of block；Pi represents the spacing value of two neighboring pilot point；Pn represents the quantity of pilot point.Pilot point is known to receiving terminal
Transmitting terminal time-domain digital symbolic point.
Step 4: each analog signal reaches receiving terminal by white Gaussian noise, it is converted into respectively by A/D modular converters
The time-domain signal of receiving terminal；
Signal communication environments are the environment of white Gaussian noise (AWGN) in the present invention, and only consider transmitting terminal phase noise
Influence.Signal at C for transmission signal by effect of phase noise and after Gaussian white noise channel, i-th of subdata agllutination
Time-domain signal r corresponding to m-th of numerical chracter on structurei,mIt is as follows：
For the phase noise of m-th of numerical chracter on i-th of sub-block of transmitting terminal, ni,mFor i-th of sub-block
White Gaussian noise component on upper m-th of time-domain signal.
Step 5: being directed to i-th of sub-block, receiving terminal carries out phase noise rough estimate using pilot linear interpolation, obtains
The rough estimate phase noise of each numerical chracter into the sub-block.
Signal at D is just compensation phase noise sequence, and the phase noise value of each symbolic point is estimated by linear interpolation.Before
The average phase noise of both ends unique word is worth as the first value of signal at D and end afterwards, as centre after pilot point linear interpolation
Noise spot.Head and the tail numerical chracter phase noise first in i-th of sub-block of rough estimate, it is specially：
For 16 first unique words of the sub-block, the phase noise average value of 16 symbols is taken as i-th of subnumber
According to the phase noise of first symbol of first unique word of block；Similarly, phase corresponding to 16 symbols of the tail unique word of the sub-block
Position noise, the phase noise averaged as the tail unique word last symbol of the sub-block；
The phase noise of each symbol is calculated as follows in unique word：
δ2Represent the power of white Gaussian noise；
The phase noise of remaining 526 bit digital symbol of i-th of sub-block of rough estimate, it is specially：
First, the phase noise of each pilot point in the sub-block is calculated, then, using pilot point linear interpolation method, slightly
Estimate the phase noise of each numerical chracter between two neighboring pilot point.
The phase noise of each pilot point is calculated as follows：
Step 6: i-th of sub-block is directed to, using the rough estimate phase noise of each numerical chracter to respective time domain
Signal carries out phase noise compensation, obtains the data signal after each numerical chracter coarse compensation.
Data signal after m-th of numerical chracter coarse compensation of i-th of sub-blockIt is as follows：
ΔΦi,mThe phase noise of m-th of time-domain signal for i-th of sub-block of coarse compensation receiving terminal is represented,Represent the conjugation of m-th of numerical chracter rough estimate phase noise of i-th of sub-block；
Step 7: carrying out SC-FDE demodulation to the data signal after each coarse compensation of i-th of sub-block, demodulated
Signal afterwards.
SC-FDE demodulation includes FFT, MMSE frequency domain equalizations and IFFT；
It is the frequency-region signal after FFT at F, is expressed from the next：
It is the signal after MMSE frequency domain equalizations at G, is the time-domain signal after IFFT at H,
Step 8: being carried out to signal of all sub-blocks after SC-FDE is demodulated from correcting, the phase from after correcting is obtained
Position noise；
As shown in figure 4, comprise the following steps that：
Step 801, by after SC-FDE is demodulated signal carry out 16QAM demappings obtain binary bit stream；
16QAM demappings i.e. hard decision, the binary stream after hard decision is obtained at I.
Step 802, binary bit stream used into 16QAM map modulations, the 16QAM symbols after being compensated again；
Step 803, the 16QAM symbols after compensation are divided into data block, insertion is unique respectively in each data block head and the tail
Word, form the sub-block after compensation；
The sub-block reinserted after unique word after obtaining the compensation at K has identical structure with signal at C.
Step 804, for just compensation after i-th of sub-block, m-th of numerical chracterWith reference to the time domain before rough estimate
Signal ri,mPhase noise is tried to achieve from the feedback phase noise delta Φ in rectification module~ i,m；
Utilize the numerical chracter after first compensationWith time-domain signal ri,mAsk for i-th of sub-block, m-th of feedback phase
Noise delta Φ~ i,m。
* conjugation is represented.
Step 805, in each pilot point of i-th of sub-block, 4 are obtained from correction threshold values after just compensating；
Because pilot point is known to receiving terminal, it is ensured that pilot point phase noise is correct, is made an uproar using the phase of pilot point
Sound carries out threshold value setting：4 are respectively from correction threshold value：The phase minimum Φ of pilot point phase noisemin,p,i,m：
The phase maximum Φ of pilot point phase noisemax,p,i,m：
The amplitude minimum value of pilot point phase noise | Φ |min,p,i,m：
The amplitude maximum of pilot point phase noise | Φ |max,p,i,m：
Step 806, the phase noise point using 4 mistaken verdicts from each feedback phase noise of correction threshold value searching.
Detailed process is as follows：
The present invention has carried out phase noise coarse compensation to signal first in receiving terminal, and then signal after coarse compensation is solved
Reconcile and adjudicate, if certain symbol has been mapped to the constellation point of mistake when re-starting 16QAM constellation mappings, then obtain
ΔΦ~ i,mTo have may greatly exceed from correction threshold value, so when the phase of certain feedback phase noise is made an uproar less than pilot point phase
The phase minimum Φ of soundmin,p,i,m, or the phase maximum Φ more than pilot point phase noisemax,p,i,m, or the feedback
The amplitude of phase noise is less than the amplitude minimum value of pilot point phase noise | Φ |min,p,i,m, or amplitude is more than pilot point phase
The amplitude maximum of noise | Φ |max,p,i,m, then the symbolic point is the phase noise point of mistaken verdict.
Step 807, the phase noise point for mistaken verdict, replaced according to alternative rule from corresponding phase noise value
Generation.
The present invention corrects phase noise until obtaining acceptable result using iterative manner；Detailed process is as follows：
Substitute for the first time as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is corresponding
Symbolic point when being the phase noise point of mistaken verdict, substitute a misjudgement point phase with the phase noise of previous numerical chracter and make an uproar
Sound；
ΔΦ~ i,k=ΔΦ~ i,k-1
K represents misjudgement point position；
Second of replacement is as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is corresponding
Symbolic point when being the phase noise point of mistaken verdict, with the phase noise of previous numerical chracter and the latter numerical chracter
Phase noise average value substitutes；
It is from correction condition transformation：
Third time substitutes as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is corresponding
Symbolic point when being the phase noise point of mistaken verdict, with the phase noise of the first two numerical chracter and final two digits symbol
Phase noise average value substitutes；
By that analogy, it is from correction condition transformation：
L is iterations, l>1.
Step 808, substituted using each correction after each phase noise to before each self-corresponding receiving terminal rough estimate when
Domain signal compensates, and tries to achieve from the data signal after correction compensation and carries out SC-FDE demodulation, return to step 801；
With the increase of iterations, the phase noise of the symbol of mistaken verdict is corrected, and iterations is more, and mistake is sentenced
Point certainly is fewer, and error rate of system performance gets a promotion therewith.
As shown in figure 5, being emulated using 16QAM modulation systems, each sub-block length is N=Nd+Nuw=512,
Data symbol Nd=496, unique block Nuw=16；" No-PHNC " represents not carry out any processing to phase noise in figure；
" PRC-Only " represents only to carry out coarse compensation to signal；" SC-PNC " represents that feedback is made decisions to signal to be compensated from correction；
" No-PHN " represents system not by bit error rate performance during effect of phase noise；When using decision-feedback one is carried out from correction algorithm
During secondary iteration, " SC-PNC-Iter1 " performance is substantially better than " PRC-Only "；And " SC-PNC-Iter4 " relative to " SC-PNC-
Iter2 " is 10 in the bit error rate-5When can obtain 1.5dB advantage.
Claims (4)
1. from compensation method is corrected, its feature exists the phase noise based on decision-feedback in a kind of single-carrier frequency domain equalization system
In comprising the following steps that：
Step 1: for some Bitstream signal of wireless communication system transmitting terminal, N is obtained using 16QAM map modulationssymIt is individual
16QAM symbols；
Step 2: 16QAM symbols are divided into data block by SC-FDE systems, and insertion is unique respectively in each data block head and the tail
Word, form sub-block；
Step 3: the numerical chracter of each sub-block is respectively the radiation-curable simulation of antenna after the conversion of D/A modular converters
Signal；
For m-th of numerical chracter s on i-th of sub-block after addition unique wordi,mIncluding three parts：
<mrow>
<msub>
<mi>s</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>=</mo>
<mfenced open = "{" close = "">
<mtable>
<mtr>
<mtd>
<msub>
<mi>u</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
</mtd>
<mtd>
<mrow>
<mo>(</mo>
<mn>1</mn>
<mo>&le;</mo>
<mi>m</mi>
<mo>&le;</mo>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>,</mo>
<msub>
<mi>N</mi>
<mi>d</mi>
</msub>
<mo>+</mo>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>+</mo>
<mn>1</mn>
<mo>&le;</mo>
<mi>m</mi>
<mo>&le;</mo>
<msub>
<mi>N</mi>
<mi>d</mi>
</msub>
<mo>+</mo>
<mn>2</mn>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>)</mo>
</mrow>
</mtd>
</mtr>
<mtr>
<mtd>
<msub>
<mi>z</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
</mtd>
<mtd>
<mrow>
<mo>(</mo>
<mrow>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>+</mo>
<mn>1</mn>
<mo>&le;</mo>
<mi>m</mi>
<mo>&le;</mo>
<msub>
<mi>N</mi>
<mi>d</mi>
</msub>
<mo>+</mo>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
</mrow>
<mo>)</mo>
</mrow>
</mtd>
</mtr>
<mtr>
<mtd>
<msub>
<mi>p</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
</mtd>
<mtd>
<mrow>
<mrow>
<mo>(</mo>
<mrow>
<mi>m</mi>
<mo>=</mo>
<mi>p</mi>
<mi>i</mi>
<mo>*</mo>
<mi>p</mi>
</mrow>
<mo>)</mo>
</mrow>
<mo>,</mo>
<mn>1</mn>
<mo>&le;</mo>
<mi>p</mi>
<mo>&le;</mo>
<mi>p</mi>
<mi>n</mi>
</mrow>
</mtd>
</mtr>
</mtable>
</mfenced>
</mrow>
Part I is m-th of unique word u of the sub-blocki,m；Part II is m-th of transmission symbol of the sub-block
zi,m；Part III is m-th of pilot point p of the sub-blocki,m；NuwRepresent the length of unique word；NdRepresent the data block
Length；Pi represents the spacing value of two neighboring pilot point；Pn represents the quantity of pilot point；
Step 4: each analog signal reaches receiving terminal by white Gaussian noise, it is converted into receiving respectively by A/D modular converters
The time-domain signal at end；
Time-domain signal r corresponding to m-th of numerical chracter in i-th of sub-block structurei,mIt is as follows：
<mrow>
<msub>
<mi>r</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>=</mo>
<msup>
<mi>e</mi>
<mrow>
<msub>
<mi>j&phi;</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
</mrow>
</msup>
<mo>&CenterDot;</mo>
<msub>
<mi>s</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>+</mo>
<msub>
<mi>n</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
</mrow>
For the phase noise of m-th of numerical chracter on i-th of sub-block of transmitting terminal, ni,mFor m on i-th of sub-block
White Gaussian noise component on individual time-domain signal；
Step 5: being directed to i-th of sub-block, receiving terminal carries out phase noise rough estimate using pilot linear interpolation, is somebody's turn to do
The rough estimate phase noise of each numerical chracter in sub-block；
First, the head and the tail numerical chracter phase noise in i-th of sub-block of rough estimate, it is specially：
For 16 first unique words of the sub-block, the phase noise average value of 16 symbols is taken as i-th of sub-block
First symbol of first unique word phase noise；Similarly, phase corresponding to 16 symbols of the tail unique word of the sub-block is made an uproar
Sound, the phase noise averaged as the tail unique word last symbol of the sub-block；
The phase noise of each symbol is calculated as follows in unique word：
<mrow>
<msubsup>
<mi>&Delta;&Phi;</mi>
<mrow>
<mi>u</mi>
<mo>,</mo>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
<mo>^</mo>
</msubsup>
<mo>=</mo>
<mfrac>
<mrow>
<msub>
<msup>
<mi>u</mi>
<mo>^</mo>
</msup>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>&CenterDot;</mo>
<msub>
<msup>
<mi>u</mi>
<mo>*</mo>
</msup>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
</mrow>
<mrow>
<mo>|</mo>
<msub>
<mi>u</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<msup>
<mo>|</mo>
<mn>2</mn>
</msup>
<mo>+</mo>
<msup>
<mi>&delta;</mi>
<mn>2</mn>
</msup>
</mrow>
</mfrac>
<mo>,</mo>
<mrow>
<mo>(</mo>
<mn>1</mn>
<mo>&le;</mo>
<mi>m</mi>
<mo>&le;</mo>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>,</mo>
<msub>
<mi>N</mi>
<mi>d</mi>
</msub>
<mo>+</mo>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>+</mo>
<mn>1</mn>
<mo>&le;</mo>
<mi>m</mi>
<mo>&le;</mo>
<msub>
<mi>N</mi>
<mi>d</mi>
</msub>
<mo>+</mo>
<mn>2</mn>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>)</mo>
</mrow>
<mo>)</mo>
</mrow>
δ2Represent the power of white Gaussian noise；
Then, the phase noise of remaining numerical chracter of i-th of sub-block of rough estimate, it is specially：
First, the phase noise of each pilot point in the sub-block is calculated, then, utilizes pilot point linear interpolation method, rough estimate
The phase noise of each numerical chracter between two neighboring pilot point；
The phase noise of each pilot point is calculated as follows：
<mfenced open = "" close = "">
<mtable>
<mtr>
<mtd>
<mrow>
<msubsup>
<mi>&Delta;&Phi;</mi>
<mrow>
<mi>p</mi>
<mo>,</mo>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
<mo>^</mo>
</msubsup>
<mo>=</mo>
<mfrac>
<mrow>
<msub>
<msup>
<mi>p</mi>
<mo>^</mo>
</msup>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>&CenterDot;</mo>
<msub>
<msup>
<mi>p</mi>
<mo>*</mo>
</msup>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
</mrow>
<mrow>
<mo>|</mo>
<msub>
<mi>p</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<msup>
<mo>|</mo>
<mn>2</mn>
</msup>
<mo>+</mo>
<msubsup>
<mi>&delta;</mi>
<mi>n</mi>
<mn>2</mn>
</msubsup>
</mrow>
</mfrac>
</mrow>
</mtd>
<mtd>
<mrow>
<mo>(</mo>
<mi>m</mi>
<mo>=</mo>
<mi>p</mi>
<mi>i</mi>
<mo>&CenterDot;</mo>
<mi>p</mi>
<mo>)</mo>
</mrow>
</mtd>
<mtd>
<mrow>
<mn>1</mn>
<mo>&le;</mo>
<mi>p</mi>
<mo>&le;</mo>
<mi>p</mi>
<mi>n</mi>
</mrow>
</mtd>
</mtr>
</mtable>
</mfenced>
1
Step 6: i-th of sub-block is directed to, using the rough estimate phase noise of each numerical chracter to respective time-domain signal
Phase noise compensation is carried out, obtains the data signal after each numerical chracter coarse compensation；
Data signal after m-th of numerical chracter coarse compensation of i-th of sub-blockIt is as follows：
<mrow>
<msubsup>
<mi>r</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
<mo>^</mo>
</msubsup>
<mo>=</mo>
<msub>
<mi>r</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>&CenterDot;</mo>
<msubsup>
<mi>&Delta;&Phi;</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
<mo>*</mo>
</msubsup>
</mrow>
Represent the conjugation of m-th of numerical chracter rough estimate phase noise of i-th of sub-block；
Step 7: carrying out SC-FDE demodulation to the data signal after each coarse compensation of i-th of sub-block, believe after being demodulated
Number；
Step 8: being carried out to signal of all sub-blocks after SC-FDE is demodulated from correcting, obtain the phase from after correcting and make an uproar
Sound.
2. the phase noise based on decision-feedback corrects benefit certainly in a kind of single-carrier frequency domain equalization system as claimed in claim 1
Compensation method, it is characterised in that in described step two, the length of each data block forms by 496 16QAM symbols；Each
The length of unique word is 16；Each data block forms a sub-block together with end to end unique word therewith；Each subnumber
There are 528 numerical chracters according to block.
3. the phase noise based on decision-feedback corrects benefit certainly in a kind of single-carrier frequency domain equalization system as claimed in claim 1
Compensation method, it is characterised in that in described step seven, SC-FDE demodulation includes FFT, MMSE frequency domain equalizations and IFFT.
4. the phase noise based on decision-feedback corrects benefit certainly in a kind of single-carrier frequency domain equalization system as claimed in claim 1
Compensation method, it is characterised in that described step eight specific implementation step is as follows：
Step 801, by after SC-FDE is demodulated signal carry out 16QAM demappings obtain binary bit stream；
Step 802, binary bit stream used into 16QAM map modulations, the 16QAM symbols after being compensated again；
Step 803, the 16QAM symbols after compensation are divided into data block, unique word, structure are inserted respectively in each data block head and the tail
Into the sub-block after compensation；
Step 804, for just compensation after i-th of sub-block, m-th of numerical chracterWith reference to the time-domain signal before rough estimate
ri,mPhase noise is tried to achieve from the feedback phase noise delta Φ in rectification module~ i,m；
Utilize the numerical chracter after first compensationWith time-domain signal ri,mAsk for i-th of sub-block, m-th of feedback phase noise delta
Φ~ i,m；
<mrow>
<msub>
<msup>
<mi>&Delta;&Phi;</mi>
<mo>~</mo>
</msup>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>=</mo>
<mfrac>
<mrow>
<msub>
<mi>r</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
</msub>
<mo>&CenterDot;</mo>
<msubsup>
<mi>r</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
<mrow>
<mo>~</mo>
<mo>*</mo>
</mrow>
</msubsup>
</mrow>
<mrow>
<mo>|</mo>
<msubsup>
<mi>r</mi>
<mrow>
<mi>i</mi>
<mo>,</mo>
<mi>m</mi>
</mrow>
<mo>~</mo>
</msubsup>
<msup>
<mo>|</mo>
<mn>2</mn>
</msup>
<mo>+</mo>
<msup>
<mi>&delta;</mi>
<mn>2</mn>
</msup>
</mrow>
</mfrac>
<mo>,</mo>
<mrow>
<mo>(</mo>
<mn>1</mn>
<mo>&le;</mo>
<mi>m</mi>
<mo>&le;</mo>
<msub>
<mi>N</mi>
<mi>d</mi>
</msub>
<mo>+</mo>
<mn>2</mn>
<mo>&CenterDot;</mo>
<msub>
<mi>N</mi>
<mrow>
<mi>u</mi>
<mi>w</mi>
</mrow>
</msub>
<mo>)</mo>
</mrow>
</mrow>
* conjugation is represented；
Step 805, in each pilot point of i-th of sub-block, four are obtained from correction threshold values after just compensating；
Four are respectively from correction threshold value：The phase minimum Φ of pilot point phase noisemin,p,i,m, pilot point phase noise
Phase maximum Φmax,p,i,m, pilot point phase noise amplitude minimum value | Φ |min,p,i,mWith the amplitude of pilot point phase noise
Maximum | Φ |max,p,i,m；
Step 806, the phase noise point using four mistaken verdicts from each feedback phase noise of correction threshold value searching；
Detailed process is as follows：
When the phase of certain feedback phase noise is less than the phase minimum Φ of pilot point phase noisemin,p,i,m, or more than pilot tone
The phase maximum Φ of point phase noisemax,p,i,m, or the feedback phase noise amplitude be less than pilot point phase noise width
It is worth minimum value | Φ |min,p,i,m, or amplitude is more than the amplitude maximum of pilot point phase noise | Φ |max,p,i,m, then the feedback
Phase noise is the phase noise point of mistaken verdict；
Step 807, the phase noise point for mistaken verdict, substituted according to alternative rule from corresponding phase noise value；Tool
Body process is as follows：
Substitute for the first time as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is that mistake is sentenced
During phase noise point certainly, substituted with the phase noise of previous numerical chracter；
Second of replacement is as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is that mistake is sentenced
During phase noise point certainly, replaced with the phase noise of previous numerical chracter and the phase noise average value of the latter numerical chracter
Generation；
Third time substitutes as follows：Since the 17th feedback phase noise, judge one by one, when the feedback phase noise is that mistake is sentenced
During phase noise point certainly, replaced with the phase noise of the first two numerical chracter and the phase noise average value of final two digits symbol
Generation；
By that analogy；
Step 808, using the time-domain signal before each phase noise after correction and each self-corresponding rough estimate, try to achieve and mended from correction
Data signal after repaying simultaneously carries out SC-FDE demodulation, return to step 801；
With the increase of iterations, the phase noise of the symbol of mistaken verdict is corrected, and finally gives the phase from after correcting
Noise.
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Cited By (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110138694A (en) * | 2019-03-08 | 2019-08-16 | 中山大学 | A kind of single carrier frequency domain equalization algorithm based on noise prediction |
| CN111884973A (en) * | 2020-07-14 | 2020-11-03 | 中国电子科技集团公司第五十四研究所 | Data receiving method for receiving end of single carrier frequency domain equalization system |
| CN113472712A (en) * | 2021-06-30 | 2021-10-01 | 中铁二院工程集团有限责任公司 | Phase noise suppression method |
| CN113467320A (en) * | 2021-07-21 | 2021-10-01 | 西安龙飞电气技术有限公司 | Weak signal detection algorithm based on gain compensation |
| CN114039829A (en) * | 2021-11-09 | 2022-02-11 | 北京邮电大学 | Phase recovery method and system based on pilot frequency assistance |
| CN114039673A (en) * | 2021-10-20 | 2022-02-11 | 清华大学 | Signal transmission method and signal transmission system |
| CN114465864A (en) * | 2022-02-17 | 2022-05-10 | 哲库科技(北京)有限公司 | Phase noise compensation method, phase noise compensation device, terminal, storage medium, and program product |
Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102265549A (en) * | 2008-09-22 | 2011-11-30 | 柯扎特·拉多·拉多莱斯库 | Network timing synchronization systems |
| CN102857468A (en) * | 2012-09-17 | 2013-01-02 | 哈尔滨工业大学 | SC-FDE (single carrier with frequency-domain equalization) system based on MAP (maximum a posterior) equalization and construction method of pilot frequency structure therein |
| CN103179062A (en) * | 2013-03-22 | 2013-06-26 | 电子科技大学 | Phase noise suppression method under low-complexity channel estimation of SC-FDE (single carrier-frequency domain equalization) system |
| WO2015183114A1 (en) * | 2014-05-30 | 2015-12-03 | Instituto Superior Tecnico | Receiver with joint equalization and phase-noise estimation and corresponding method |
| US20170111189A1 (en) * | 2015-10-14 | 2017-04-20 | Maxlinear Asia Singapore Private Limited | Wireless backhaul |
-
2017
- 2017-07-11 CN CN201710562231.5A patent/CN107438047B/en active Active
Patent Citations (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102265549A (en) * | 2008-09-22 | 2011-11-30 | 柯扎特·拉多·拉多莱斯库 | Network timing synchronization systems |
| CN102857468A (en) * | 2012-09-17 | 2013-01-02 | 哈尔滨工业大学 | SC-FDE (single carrier with frequency-domain equalization) system based on MAP (maximum a posterior) equalization and construction method of pilot frequency structure therein |
| CN102857468B (en) * | 2012-09-17 | 2015-05-13 | 哈尔滨工业大学 | SC-FDE (single carrier with frequency-domain equalization) system based on MAP (maximum a posterior) equalization and construction method of pilot frequency structure therein |
| CN103179062A (en) * | 2013-03-22 | 2013-06-26 | 电子科技大学 | Phase noise suppression method under low-complexity channel estimation of SC-FDE (single carrier-frequency domain equalization) system |
| WO2015183114A1 (en) * | 2014-05-30 | 2015-12-03 | Instituto Superior Tecnico | Receiver with joint equalization and phase-noise estimation and corresponding method |
| US20170111189A1 (en) * | 2015-10-14 | 2017-04-20 | Maxlinear Asia Singapore Private Limited | Wireless backhaul |
Cited By (12)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN110138694A (en) * | 2019-03-08 | 2019-08-16 | 中山大学 | A kind of single carrier frequency domain equalization algorithm based on noise prediction |
| CN111884973A (en) * | 2020-07-14 | 2020-11-03 | 中国电子科技集团公司第五十四研究所 | Data receiving method for receiving end of single carrier frequency domain equalization system |
| CN111884973B (en) * | 2020-07-14 | 2021-08-31 | 中国电子科技集团公司第五十四研究所 | Data receiving method for receiving end of single carrier frequency domain equalization system |
| CN113472712A (en) * | 2021-06-30 | 2021-10-01 | 中铁二院工程集团有限责任公司 | Phase noise suppression method |
| CN113472712B (en) * | 2021-06-30 | 2023-05-19 | 中铁二院工程集团有限责任公司 | Phase noise suppression method |
| CN113467320A (en) * | 2021-07-21 | 2021-10-01 | 西安龙飞电气技术有限公司 | Weak signal detection algorithm based on gain compensation |
| CN114039673A (en) * | 2021-10-20 | 2022-02-11 | 清华大学 | Signal transmission method and signal transmission system |
| CN114039673B (en) * | 2021-10-20 | 2023-09-12 | 清华大学 | Signal transmission method and signal transmission system |
| CN114039829A (en) * | 2021-11-09 | 2022-02-11 | 北京邮电大学 | Phase recovery method and system based on pilot frequency assistance |
| CN114039829B (en) * | 2021-11-09 | 2022-08-09 | 北京邮电大学 | Phase recovery method and system based on pilot frequency assistance |
| CN114465864A (en) * | 2022-02-17 | 2022-05-10 | 哲库科技(北京)有限公司 | Phase noise compensation method, phase noise compensation device, terminal, storage medium, and program product |
| CN114465864B (en) * | 2022-02-17 | 2024-01-12 | 哲库科技(北京)有限公司 | Phase noise compensation method, apparatus, terminal, storage medium, and program product |
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