CN113472712B - Phase noise suppression method - Google Patents
Phase noise suppression method Download PDFInfo
- Publication number
- CN113472712B CN113472712B CN202110741216.3A CN202110741216A CN113472712B CN 113472712 B CN113472712 B CN 113472712B CN 202110741216 A CN202110741216 A CN 202110741216A CN 113472712 B CN113472712 B CN 113472712B
- Authority
- CN
- China
- Prior art keywords
- phase noise
- pilot
- information
- noise information
- phase
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Active
Links
- 238000000034 method Methods 0.000 title claims abstract description 40
- 230000001629 suppression Effects 0.000 title claims abstract description 13
- 238000004891 communication Methods 0.000 claims abstract description 14
- 238000001914 filtration Methods 0.000 claims abstract description 9
- 238000004364 calculation method Methods 0.000 claims description 5
- 230000021615 conjugation Effects 0.000 claims description 3
- ODKSFYDXXFIFQN-UHFFFAOYSA-M argininate Chemical compound [O-]C(=O)C(N)CCCNC(N)=N ODKSFYDXXFIFQN-UHFFFAOYSA-M 0.000 claims description 2
- 230000000694 effects Effects 0.000 description 14
- 239000011159 matrix material Substances 0.000 description 12
- 238000009825 accumulation Methods 0.000 description 7
- 238000006243 chemical reaction Methods 0.000 description 6
- 230000005540 biological transmission Effects 0.000 description 3
- 238000003780 insertion Methods 0.000 description 3
- 230000037431 insertion Effects 0.000 description 3
- 238000013507 mapping Methods 0.000 description 3
- 230000004044 response Effects 0.000 description 3
- 238000001228 spectrum Methods 0.000 description 3
- 239000013256 coordination polymer Substances 0.000 description 2
- 230000000875 corresponding effect Effects 0.000 description 2
- 125000004122 cyclic group Chemical group 0.000 description 2
- 238000012545 processing Methods 0.000 description 2
- 238000011160 research Methods 0.000 description 2
- 230000009286 beneficial effect Effects 0.000 description 1
- 239000000969 carrier Substances 0.000 description 1
- 230000001427 coherent effect Effects 0.000 description 1
- 230000002596 correlated effect Effects 0.000 description 1
- 230000007547 defect Effects 0.000 description 1
- 238000011161 development Methods 0.000 description 1
- 238000010586 diagram Methods 0.000 description 1
- 238000005516 engineering process Methods 0.000 description 1
- 230000003287 optical effect Effects 0.000 description 1
- 238000012360 testing method Methods 0.000 description 1
- 230000009466 transformation Effects 0.000 description 1
Images
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2689—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation
- H04L27/2691—Link with other circuits, i.e. special connections between synchronisation arrangements and other circuits for achieving synchronisation involving interference determination or cancellation
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
- Y02D30/00—Reducing energy consumption in communication networks
- Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks
Landscapes
- Engineering & Computer Science (AREA)
- Computer Networks & Wireless Communication (AREA)
- Signal Processing (AREA)
- Noise Elimination (AREA)
Abstract
The invention discloses a phase noise suppression method, which comprises the following steps: s1, acquiring a received signal from a communication system, and recovering a data symbol from the received signal; s2, calculating pilot part phase noise information of reserved amplitude according to the recovered data symbols and pilot symbols inserted by a transmitting end of the communication system; s3, performing linear interpolation based on phase noise information of the pilot part with reserved amplitude to obtain phase noise information with the length of M; and S4, accumulating and filtering the phase noise information with the length of M, normalizing the filtered phase noise information to obtain a phase angle, and compensating the phase noise. With the adoption of the method, the performance improvement of the method is more obvious under the AWGN channel and the Rayleigh channel than the existing linear interpolation algorithm along with the increase of the pilot quantity, and the method has better phase noise estimation compensation performance.
Description
Technical Field
The invention relates to the technical field of phase noise processing of an OFDM system, in particular to a phase noise suppression method.
Background
Phase noise is essentially phase error noise in the received signal compared to the transmitted signal, which can cause random phase variations in the received signal of the DFT-S-OFDM system, resulting in reduced system performance. The impact of phase noise on the system can be generally divided into two parts: a first part, a common phase error (Common Phase Error, CPE), which part is embodied in a common phase rotation of constellation points in the constellation; and a second part, inter-Carrier-Interference (ICI), which is embodied in the constellation diagram, where constellation points diverge around. In general, CPE in phase noise is relatively easy to estimate and compensate because it is a common error part for each symbol; however, the second ICI portion of the phase noise is correlated with the transmitted symbols and is more random per data symbol, making phase noise estimation and compensation of that portion difficult. Since OFDM systems are widely used in practice, a great deal of literature has been studied on phase noise estimation and compensation techniques of OFDM systems, and the main algorithms can be roughly divided into three categories: first, only estimating and compensating CPE part in phase noise; a second class, ICI compensation algorithms that additionally take into account ICI components in phase noise; and the third class, iterative algorithm combining channel estimation and phase noise estimation compensation.
In the OFDM system, most of the literature adopts phase noise compensation based on a frequency domain, and a scholars propose phase noise compensation based on a time domain of the OFDM system, wherein a high-order FIR filter designed by utilizing a Remez algorithm is used for correcting estimated phase noise. The complexity of the adopted high-order FIR filter is higher, the latter adopts the accumulating and filtering with lower complexity to replace, and better phase noise compensation performance is obtained. Then, a learner expands an algorithm for correcting estimated phase noise by using an accumulation and filter as a low-pass filter to a DFT-S-OFDM system, obtains a better phase noise compensation effect, and analyzes two different pilot frequency insertion modes in the DFT-S-OFDM system: pilot frequency is inserted into data before pilot frequency and DFT module are inserted from frequency domain, so that the pilot frequency is inserted from frequency domain, and the phase noise compensation algorithm which has been proved by a great deal of researches and demonstration of predecessors in OFDM system can be utilized, but the pilot frequency insertion mode can improve peak-to-average power ratio of DFT-S-OFDM system; the pilot frequency is inserted before the DFT module, the receiving end carries out estimation compensation on the phase noise after demodulation of the IDFT module, and the pilot frequency insertion mode does not increase the peak-to-average ratio of the DFT-S-OFDM system. The method has the advantages that the Kalman filtering is applied to eliminate the phase noise problem in the communication system, and the Kalman filtering algorithm is applied to the DFT-S-OFDM system, so that a good phase noise estimation compensation effect is obtained.
In the existing phase noise estimation compensation algorithm, an iterative algorithm combining channel estimation and phase noise estimation compensation can better estimate and compensate the phase noise of a system, but in an actual communication system, the ICI part in the phase noise is not serious enough to be completely eliminated, so that the algorithm is less applied in the actual system due to higher complexity.
The nonlinear distortion of the radio frequency can bring phase noise interference to the OFDM system, so that the frequency spectrum of the signal output by the oscillator is expanded, and serious influence is brought to the system performance. With the rapid development of communication technology, the performance index requirements of the system for suppressing the phase noise in the system are more and more strict, so that the research of the phase noise related estimation compensation algorithm is more and more urgent. In the prior art, the invention patent publication No. (CN 201910059644.0) of a novel time domain volume Kalman phase noise compensation scheme in a coherent optical OFDM system discloses a phase noise compensation scheme of a CO-OFDM system, and particularly relates to a novel time domain volume Kalman phase noise compensation algorithm scheme. In the scheme, pilot frequency information is utilized to compensate CPE phase noise through an extended Kalman and linear interpolation algorithm, pre-judgment is carried out on a signal subjected to first-order compensation of the phase noise, and secondary symbol processing is carried out on the signal subjected to pre-judgment in a time domain. And combining the time domain signal processed by the secondary symbol, and carrying out a volume Kalman phase noise compensation algorithm on the judged data in the time domain to realize fine compensation of ICI phase noise. And carrying out iterative operation on the data after fine compensation, thereby improving the compensation effect.
However, the algorithm for ICI compensation with an accumulation and filter, where the input of the accumulation and filter is only the phase noise estimate information at the pilot, does not provide good source information for the input of the accumulation and filter.
Disclosure of Invention
The invention aims to overcome the defects in the prior art and provides a phase noise suppression method.
In order to achieve the above object, the present invention provides the following technical solutions:
a phase noise suppression method comprising the steps of:
s1, acquiring a received signal from a communication system, and recovering a data symbol from the received signal;
s2, calculating pilot part phase noise information of reserved amplitude according to the recovered data symbols and pilot symbols inserted by a transmitting end of the communication system;
s3, performing linear interpolation based on the phase noise information of the pilot part with the reserved amplitude to obtain phase noise information with the length of M;
and S4, accumulating and filtering the phase noise information with the length of M, normalizing the filtered phase noise information to obtain a phase angle, and compensating the phase noise.
As a preferred embodiment of the present invention, in step S1, the data symbol is represented as:
wherein ,
for the phase noise influence of the data symbols after the receiving end passes through the IDFT module, +.>
Is a gaussian noise vector of m×1, s= [ d (1), d (2),. The term, d (M)] T D (M) is a constellation symbol, m=1, 2.
As a preferred embodiment of the present invention, in step S2, a calculation formula for calculating the phase noise information of the pilot portion of the reserved amplitude is as follows:
wherein ,
is the phase noise information of pilot part of the reserved amplitude epsilon p(k) Index position indicating kth pilot, N p For inserting the number of pilot of the data part, < >>
Is the recovered data symbol, s= [ d (1), d (2),. The term, d (M)] T D (M) is a constellation symbol, m=1, 2,..m, [ ·] * Representing conjugation.
As a preferred embodiment of the present invention, in step S3, the calculation formula of the phase noise information with length M is:
wherein ,
is the first phase noise information of the pilot part of the reserved amplitude,/and>
is the +.about.th of the pilot part of the reserved amplitude>
Information of phase noise N p K=1, 2, N for the number of pilot inserted into the data portion p I is index position information, ε p(k) Index position indicating kth pilot,/-, etc.>
Is the pilot part phase noise information that preserves the amplitude.
As a preferred embodiment of the present invention, in step S4, the phase angle is an estimated phase noise phase angle
Estimated phase noise phase angle>
Represented as
Where, g is phase noise information, b is a vector of all 1, and arg { } is a principal value of argument of complex number.
Based on the same conception, the invention also provides a phase noise suppression device which comprises at least one processor and a memory which is in communication connection with the at least one processor; the memory stores instructions executable by the at least one processor to enable the at least one processor to perform the method of any one of the above.
Compared with the prior art, the invention has the beneficial effects that:
with the adoption of the method, the performance improvement of the method is more obvious under the AWGN channel and the Rayleigh channel than the existing linear interpolation algorithm along with the increase of the pilot quantity, and the method has better phase noise estimation compensation performance. And the ZC sequence, the Golay sequence and the 4-QAM sequence can exert better phase noise estimation compensation performance.
Description of the drawings:
FIG. 1 is a transmission model of a DFT-S-OFDM system with phase noise added in embodiment 1 of the present invention;
fig. 2 is a flowchart of a phase noise suppression method in embodiment 1 of the present invention;
FIG. 3 shows the compensation effect (AWGN channel) of the method of the present invention at different pilot numbers in example 1 of the present invention;
fig. 4 shows the compensation effect (rayleigh channel) of the method of the present invention at different pilot numbers in example 1 of the present invention;
FIG. 5 shows the compensation effect (AWGN channel) of the different pilot sequences according to the method of the invention in example 1;
fig. 6 shows the compensation effect (rayleigh channel) of the different pilot sequences according to the method of the present invention in example 1 of the present invention.
Detailed Description
The present invention will be described in further detail with reference to test examples and specific embodiments. It should not be construed that the scope of the above subject matter of the present invention is limited to the following embodiments, and all techniques realized based on the present invention are within the scope of the present invention.
Example 1:
the algorithm is directed to a DFT-S-OFDM (Discrete Fourier Transform-spread-Orthogonal Frequency Division Multiplexing) system. The DFT-S-OFDM system model in consideration of phase noise is shown in fig. 1, and the effect of phase noise on the transmitting system may be equivalent to that applied to a signal after radio frequency transmission, and the effect of phase noise on the receiving system may be equivalent to that applied to a received signal.
Matrix phi of phase angle of phase noise of transmitting end Tx And a receiving end phase noise phase angle matrix phi Rx Respectively defined as
φ Tx ＝diag(φ Tx (1)，φ Tx (2)，...，φ Tx (N)) (1)
φ Rx ＝diag(φ Rx (1)，φ Rx (2)，...，φ Rx (N)) (2)
Under the above background, a phase noise suppression method, a flowchart of which is shown in fig. 2, includes the following steps:
s1, acquiring a received signal from a communication system, and recovering a data symbol from the received signal;
s2, calculating pilot part phase noise information of reserved amplitude according to the recovered data symbols and pilot symbols inserted by a transmitting end of the communication system;
s3, performing linear interpolation based on the phase noise information of the pilot part with the reserved amplitude to obtain phase noise information with the length of M;
and S4, accumulating and filtering the phase noise information with the length of M, normalizing the filtered phase noise information to obtain a phase angle, and compensating the phase noise.
The method comprises the steps of S1, acquiring a received signal from a communication system, and recovering a data symbol from the received signal, wherein the method comprises the following steps:
let the information bit stream of the user be a bit vector c= [ c ] of length M x k 1 ，c 2 ，...，cM k ] T . Every k consecutive information bits { c (m-1)+1 ，c (m-1)+2 ，...，c (m-1)+k Modulated into one constellation symbol d (M), (m=1, 2,., M). Thus, the user's transmitted symbol can be written as s= [ d (1), d (2),] T, wherein [·]T Representing the transpose operation. Next, M-point DFT is performed on D to obtain s= [ S (0), S (1),] T i.e.
S＝F M s (3)
in the formulaFM The p-th row and q-th column elements of (a)
Representing an mxm DFT matrix. Similarly, an M-point IDFT matrix can be defined as F M H, wherein [·]H Representing the conjugate transpose operation.
The frequency domain symbol of the user is mapped to M corresponding sub-carriers by P, and then is transformed to the time domain by N-point IFFT to obtain the time domain transmitting symbol of the user as
And adding CP to x, and forming a time domain transmission signal of the user after parallel-serial conversion and D/A conversion.
Defining the time domain channel matrix from the user to the transmitting end as C. C is an nxn cyclic matrix, the first column of which is the user-to-sender channel impulse response sequence. After receiving the user signal, the receiving end obtains a time domain receiving signal y through A/D conversion, serial-parallel conversion and CP removal, which can be expressed as
Where z represents an n×1 time domain gaussian white noise vector, n×m matrix P represents a subcarrier mapping matrix of the user, and F represents FFT transformation.
Frequency domain symbol of user after frequency domain equalization Q process
Can be expressed as
Since the phase noise has steep low-pass characteristics and has a smaller spectrum bandwidth relative to the correlation bandwidth of the channel, the main energy in the phase noise power spectrum density is concentrated in the range of the coherence bandwidth of the channel, and the equation (6) can be simplified into
wherein ,
e jφ is the phase noise after combination, defined as
Wherein phi (n) =phi Tx (n)+φ Rx (n)，n＝1，2，...，N。
The frequency domain total phase noise matrix Φ can be expressed as
Substituted into (7) with
Thus, the recovered user sends the symbol as
in the formula
Is an mx1 gaussian noise vector.
From the foregoing, it can be seen that Φ is a cyclic matrix, the main diagonal elements of which are the same, and Φ is Φ=Φ 0 I N ++ (phi-phi) 0 I N ) Form of phi 0 Is the main diagonal element of the frequency domain total phase noise matrix phi, and the equation (10) can be expressed as
As can be derived from equation (11), the effect of phase noise on DFT-S-OFDM system is mainly manifested in two parts: a part is the common phase rotation brought by the CPE component of the phase noise to the system, i.e. the first term in equation (11); the other part is that phase noise causes interference of adjacent subcarriers of a symbol of the DFT-S-OFDM system, namely, the second term in the equation (11), and the part is ICI part of the phase noise.
Definition of the definition
A frequency response matrix affected by the time domain phase noise on the sub-carrier corresponding to the user, i.e
Formula (10) can be simplified as +.>
wherein ,
for the influence of phase noise on the data symbols after the receiving end passes through the IDFT module, i.e.
Since the DFT-S-OFDM system generally employs a centralized subcarrier mapping, it is possible to obtain a result that the phase noise interference suffered by the data symbol is equivalent to the time domain phase noise interference passing through the low-pass filter.
In reality, the DFT-S-OFDM system is mostly applied to the uplink, and if only the phase noise influence of the transmitting end is considered, the system will
And->
Are all gaussian noise distributed identically to z. The data symbols received by the receiver can therefore be expressed as the following relationship:
s2, calculating pilot part phase noise information of reserved amplitude according to the recovered data symbol and a pilot symbol inserted by a transmitting end of a communication system, wherein the method comprises the following steps:
in the DFT-S-OFDM system, the pilot is selected to be uniformly inserted into the DFT-input end, and is subjected to DFT conversion together with a transmitted data symbol, then is subjected to subcarrier mapping and then is subjected to IDFT conversion to be converted into a time domain to become a transmitted signal. Let the number of inserted pilot be N p ，ε p(k) The index position of the kth pilot is represented, and the calculation formula for calculating the phase noise information of the pilot part of the reserved amplitude is as follows:
wherein ,
is the phase noise information of pilot part of the reserved amplitude epsilon p(k) Index position indicating kth pilot, N p For inserting the number of pilot of the data part, < >>
Is the recovered data symbol, s= [ d (1), d (2),. The term, d (M)] T D (M) is a constellation symbol, m=1, 2,..m, [ ·] * Representing conjugation.
Conventional phase noise compensation CPE algorithms and belief interpolation algorithms require only phase noise phase angle information, which requires preserving amplitude information obtained from the received signal, because the algorithm can better estimate the reliability of phase noise per pilot symbol and automatically weight the higher reliability pilot symbols during filtering with a low pass filter.
S3, performing linear interpolation based on the phase noise information of the pilot part with the reserved amplitude to obtain phase noise information with the length of M, wherein the method comprises the following steps:
based on
Performing linear interpolation to obtain phase noise information g, # with length M>
wherein ,
is the first phase noise information of the pilot part of the reserved amplitude,/and>
is the +.about.th of the pilot part of the reserved amplitude>
Information of phase noise N p K=1, 2, N for the number of pilot inserted into the data portion p I is index position information, ε p(k) Index position indicating kth pilot,/-, etc.>
Is the pilot part phase noise information that preserves the amplitude.
And S4, accumulating and filtering the phase noise information with the length of M, normalizing the filtered phase noise information to obtain a phase angle, and compensating the phase noise.
The phase noise information g is subjected to an accumulation and filter to finish the operation of a low-pass filter, and the filtered phase noise information is normalized to obtain a phase angle, and the extracted phase information is the estimated phase noise phase angle
Assuming that the impulse response of the accumulation filter is b, i.e. b is a vector of all 1's, whose length is the window length of the accumulation filter, the estimated phase noise phase angle +.>
Can be expressed as
Where, represents a convolution operation.
Finally, using the estimated phase angle
The receiving end receives the symbol after phase noise compensation>
Can be expressed as
The compensation effect (AWGN channel) of the method of the present invention at different pilot numbers is shown in fig. 3; the compensation effect (rayleigh channel) of the method of the present invention at different pilot numbers is shown in fig. 4. On both channels, the bit error rate of the new algorithm is significantly reduced compared to the conventional linear interpolation algorithm for the same number of pilot. With the increase of the pilot quantity, the performance of the method is improved obviously under the AWGN channel and the Rayleigh channel compared with the linear interpolation algorithm. Therefore, the method of the invention has better phase noise estimation compensation performance.
The compensation effect (AWGN channel) of different pilot sequences based on the method of the present invention is shown in fig. 5; the compensation effect (Rayleigh channel) of different pilot sequences based on the method of the invention is shown in fig. 6, and in general, the ZC sequence, the Golay sequence and the 4-QAM sequence can exert better phase noise estimation compensation performance by adopting the method of the invention.
The execution sequence of the steps is one embodiment of the method of the present invention, and is not limited to being executed only in the sequence, but based on the method idea of the present invention, the execution sequence is changed and is within the protection scope of the present invention.
Claims (4)
1. A phase noise suppression method, comprising the steps of:
s1, acquiring a received signal from a communication system, and recovering a data symbol from the received signal;
s2, calculating pilot part phase noise information of reserved amplitude according to the recovered data symbols and pilot symbols inserted by a transmitting end of the communication system;
s3, performing linear interpolation based on the phase noise information of the pilot part with the reserved amplitude to obtain phase noise information with the length of M;
s4, accumulating and filtering the phase noise information with the length of M, normalizing the filtered phase noise information to obtain a phase angle, and compensating the phase noise;
in step S2, the calculation formula for calculating the phase noise information of the pilot part of the reserved amplitude is as follows:
wherein ,
is the phase noise information of pilot part of the reserved amplitude epsilon p(k) Index position indicating kth pilot, N p For inserting the number of pilot of the data part, < >>
Is the recovered data symbol, s= [ d (1), d (2),. The term, d (M)] T D (M) is a constellation symbol, m=1, 2,..m, [ ·] * Represents conjugation;
in step S3, the calculation formula of the phase noise information with length M is:
wherein ,
is the first phase noise information of the pilot part of the reserved amplitude,/and>
is the +.about.th of the pilot part of the reserved amplitude>
Information of phase noise N p K=1, 2, N for the number of pilot inserted into the data portion p I is index position information, ε p(k) Index position indicating kth pilot,/-, etc.>
Is the pilot part phase noise information that preserves the amplitude.
2. The phase noise suppression method according to claim 1, wherein in step S1, the data symbols are represented as:
wherein ,
for the phase noise influence of the data symbols after the receiving end passes through the IDFT module, +.>
Is a gaussian noise vector of m×1, s= [ d (1), d (2),. The term, d (M)] T D (M) is a constellation symbol, m=1, 2.
3. The method of phase noise suppression according to claim 1, wherein in step S4, the phase angle is an estimated phase noise phase angle
Estimated phase noise phase angle>
Represented as
Where, g is phase noise information, b is a vector of all 1, and arg { } is a principal value of argument of complex number.
4. A phase noise suppression device comprising at least one processor, and a memory communicatively coupled to the at least one processor; the memory stores instructions executable by the at least one processor to enable the at least one processor to perform the method of any one of claims 1-3.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110741216.3A CN113472712B (en) | 2021-06-30 | 2021-06-30 | Phase noise suppression method |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CN202110741216.3A CN113472712B (en) | 2021-06-30 | 2021-06-30 | Phase noise suppression method |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN113472712A CN113472712A (en) | 2021-10-01 |
| CN113472712B true CN113472712B (en) | 2023-05-19 |
Family
ID=77876944
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN202110741216.3A Active CN113472712B (en) | 2021-06-30 | 2021-06-30 | Phase noise suppression method |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN113472712B (en) |
Families Citing this family (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN114039829B (en) * | 2021-11-09 | 2022-08-09 | 北京邮电大学 | Phase recovery method and system based on pilot frequency assistance |
| CN115128563B (en) * | 2022-08-31 | 2022-11-29 | 中国科学院空天信息创新研究院 | Amplitude-phase error compensation method for satellite-borne synthetic aperture radar system |
| CN115955380A (en) * | 2022-12-30 | 2023-04-11 | 四川创智联恒科技有限公司 | Phase noise estimation and compensation method |
Citations (11)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| KR20020044505A (en) * | 2000-12-06 | 2002-06-15 | 윤종용 | Apparatus for receiving of OFDM signals and Method for recovering of signals by estimating channel |
| IL146266A0 (en) * | 1999-05-12 | 2002-07-25 | Qualcomm Inc | Amplitude and phase estimation method in a wireless communication system |
| CN101039291A (en) * | 2006-03-16 | 2007-09-19 | 中国科学院上海微系统与信息技术研究所 | Method and apparatus for correcting residual carrier frequency deviation, fixed phase and amplitude deviation |
| CN101595704A (en) * | 2007-01-16 | 2009-12-02 | 皇家飞利浦电子股份有限公司 | Be used for improving the system and method that frequency/phase errors is followed the tracks of in high-speed high frequency communication |
| WO2011140502A1 (en) * | 2010-05-07 | 2011-11-10 | Tyco Electronics Subsea Communications, Llc | Pilot symbol aided carrier phase estimation |
| WO2016148158A1 (en) * | 2015-03-17 | 2016-09-22 | Mitsubishi Electric Corporation | Method and receiver for decoding optical signal |
| CN107231216A (en) * | 2017-07-04 | 2017-10-03 | 电子科技大学 | Phase noise compensation suppressing method based on GAMP algorithms |
| CN107395282A (en) * | 2017-07-20 | 2017-11-24 | 浙江工业大学 | Phase noise compensation method of time domain unscented Kalman filtering large-linewidth CO-OFDM system |
| CN107438047A (en) * | 2017-07-11 | 2017-12-05 | 北京邮电大学 | The phase noise based on decision-feedback corrects compensation method certainly in a kind of single-carrier frequency domain equalization system |
| WO2019006597A1 (en) * | 2017-07-03 | 2019-01-10 | Huawei Technologies Co., Ltd. | Methods and apparatuses for quadrature amplitude modulation optimized for phase noise |
| CN109687912A (en) * | 2019-01-22 | 2019-04-26 | 重庆邮电大学 | A kind of novel time domain volume Kalman's phase noise compensation scheme in relevant optical OFDM system |
Family Cites Families (4)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US9473250B2 (en) * | 2014-09-03 | 2016-10-18 | Mitsubishi Electric Research Laboratories, Inc. | System and method for recovering carrier phase in optical communications |
| US10527663B2 (en) * | 2015-03-12 | 2020-01-07 | Texas Instruments Incorporated | Kalman filter for phase noise tracking |
| EP3406063B1 (en) * | 2016-03-31 | 2020-03-18 | Huawei Technologies Co., Ltd. | Method and apparatus for quadrature signal modulation |
| US10284303B2 (en) * | 2017-03-17 | 2019-05-07 | Nokia Of America Corporation | Feedforward decision-free pilot-aided phase noise estimation for coherent optical M-QAM systems |
-
2021
- 2021-06-30 CN CN202110741216.3A patent/CN113472712B/en active Active
Patent Citations (11)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| IL146266A0 (en) * | 1999-05-12 | 2002-07-25 | Qualcomm Inc | Amplitude and phase estimation method in a wireless communication system |
| KR20020044505A (en) * | 2000-12-06 | 2002-06-15 | 윤종용 | Apparatus for receiving of OFDM signals and Method for recovering of signals by estimating channel |
| CN101039291A (en) * | 2006-03-16 | 2007-09-19 | 中国科学院上海微系统与信息技术研究所 | Method and apparatus for correcting residual carrier frequency deviation, fixed phase and amplitude deviation |
| CN101595704A (en) * | 2007-01-16 | 2009-12-02 | 皇家飞利浦电子股份有限公司 | Be used for improving the system and method that frequency/phase errors is followed the tracks of in high-speed high frequency communication |
| WO2011140502A1 (en) * | 2010-05-07 | 2011-11-10 | Tyco Electronics Subsea Communications, Llc | Pilot symbol aided carrier phase estimation |
| WO2016148158A1 (en) * | 2015-03-17 | 2016-09-22 | Mitsubishi Electric Corporation | Method and receiver for decoding optical signal |
| WO2019006597A1 (en) * | 2017-07-03 | 2019-01-10 | Huawei Technologies Co., Ltd. | Methods and apparatuses for quadrature amplitude modulation optimized for phase noise |
| CN107231216A (en) * | 2017-07-04 | 2017-10-03 | 电子科技大学 | Phase noise compensation suppressing method based on GAMP algorithms |
| CN107438047A (en) * | 2017-07-11 | 2017-12-05 | 北京邮电大学 | The phase noise based on decision-feedback corrects compensation method certainly in a kind of single-carrier frequency domain equalization system |
| CN107395282A (en) * | 2017-07-20 | 2017-11-24 | 浙江工业大学 | Phase noise compensation method of time domain unscented Kalman filtering large-linewidth CO-OFDM system |
| CN109687912A (en) * | 2019-01-22 | 2019-04-26 | 重庆邮电大学 | A kind of novel time domain volume Kalman's phase noise compensation scheme in relevant optical OFDM system |
Non-Patent Citations (5)
| Title |
|---|
| Aviv Shalom，et. al..Improving the Phase Noise Tolerance of Subbanded DFT-Spread OFDM by Nonredundant Interleaving.《 Journal of Lightwave Technology》.2014,全文. * |
| Chao Li, et. al..Phase Noise Cancellation Polarization-Insensitive All-Optical Wavelength Conversion of DFT-S PDM-OFDM-8/16/32QAM Signals using Coherent Dual-Pumps.《2014 Asia Communications and Photonics Conference (ACP)》.2019,全文. * |
| 张浩元.PDM CO-OFDM系统中IQ不平衡和相位噪声抑制算法研究.《中国优秀博硕士学位论文全文数据库（硕士）信息科技辑》.2018,全文. * |
| 章青青.《中国优秀博硕士学位论文全文数据库（硕士）信息科技辑》.《中国优秀博硕士学位论文全文数据库（硕士）信息科技辑》.2019,全文. * |
| 胡凡.非协作数字通信系统群时延及相位噪声研究.《中国优秀博硕士学位论文全文数据库（硕士）信息科技辑》.2011,全文. * |
Also Published As
| Publication number | Publication date |
|---|---|
| CN113472712A (en) | 2021-10-01 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN113472712B (en) | Phase noise suppression method | |
| EP2887597A1 (en) | Channel estimation method and device in multiple-receiving antenna system | |
| US7539255B2 (en) | Using multiple pilot signals for timing error estimation in digital subscriber line communications | |
| CN109600332B (en) | CO-FBMC/OQAM modulation method and system | |
| US8619744B2 (en) | Reception method and receiver | |
| US20040076239A1 (en) | Apparatus and method for tracking residual frequency offset for single carrier-frequency domain equalizer system | |
| Fang et al. | Phase noise estimation and suppression for PDM CO-OFDM/OQAM systems | |
| JP4440303B2 (en) | Receiver | |
| KR101291683B1 (en) | Method of equalization for TDS-OFDM receiver and apparatus thereof | |
| CN103051578B (en) | With the iteration error propagation judgement OFDM channel estimation method that ICI eliminates | |
| CN102769599B (en) | A kind of novel signal of orthogonal frequency division multiplexing system processing method and device | |
| CN104641568A (en) | Power line communication transceiver and power line communication method | |
| CN112087266B (en) | Time-varying broadband Doppler compensation method based on EMD-WFFT | |
| KR20130016920A (en) | A receiver of communication system for orthogonal frequency division multiplexing and method for mitigate a phase noise in thereof | |
| US20150043683A1 (en) | Receiving device, receiving method, and receiving program | |
| US20170265202A1 (en) | Time domain pilot of single-carrier mimo system and synchronization method thereof | |
| KR20180052003A (en) | Method and Apparatus for Distortion Compensation of Subcarrier in Orthogonal Frequency Division Multiplexing System | |
| KR100431008B1 (en) | OFDM receiver using polar coordinate and method thereof | |
| CN116684033A (en) | Unmanned aerial vehicle system smart interference method and device | |
| CN115277318A (en) | Method and system for noise suppression of tester | |
| CN118041728B (en) | OCDM channel estimation method based on matching pursuit type reconstruction algorithm | |
| EP1296492B1 (en) | Multicarrier receiver with a sliding window Fourier transform and a Fourier transform | |
| KR20040105881A (en) | Blind equalizer of Multi-carrier transmission receiving system and a method thereof | |
| WO2022201387A1 (en) | Frequency offset estimation device, reception device, frequency offset estimation method, and program | |
| CN118041728A (en) | OCDM channel estimation method based on matching pursuit type reconstruction algorithm |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| PB01 | Publication | ||
| PB01 | Publication | ||
| SE01 | Entry into force of request for substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| GR01 | Patent grant | ||
| GR01 | Patent grant |