CN101667829B - 电压控制振荡器、单片微波集成电路及高频无线装置 - Google Patents
电压控制振荡器、单片微波集成电路及高频无线装置 Download PDFInfo
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Abstract
实现低相位噪声的电压控制振荡器。在具备在包含变容二极管(2)和控制电压端子(3)的可变谐振器的电压控制振荡器中,对该可变谐振器并联连接具有最长也是在高次谐波信号的1/4波长的奇数倍上相加高次谐波信号的1/16波长的长度而最短也是从高次谐波信号的1/4波长的奇数倍中减去高次谐波信号的1/16波长的长度的前端开路短截线(5)。通过该结构,在基波频率中,基波信号向前端开路短截线(5)和变容二极管(2)这两方传播,能实现高的Q值,而在高次谐波频率中,前端开路短截线(5)具有短路负载,高次谐波信号全部向前端开路短截线(5)传播,因此抑制高次谐波信号造成的控制电压(Vt)的变动。
Description
技术领域
本发明涉及电压控制振荡器、MMIC(单片微波集成电路)及高频无线装置,尤其涉及微波/毫米波区域中工作的电压控制振荡器、MMIC及高频无线装置。
背景技术
随着车载雷达或便携电话等高频无线装置的普及,对输出频率超过1GHz的振荡器的高性能化要求越来越高。振荡器指的是在电路内部产生高频电信号的振荡,并向外部发送高频电信号的电路。特别是设有用于改变输出频率的控制电压端子的振荡器被称为VCO(VoltageControlled Oscillator:电压控制振荡器)。在振荡器的内部装有用于放大高频电信号的晶体管等有源元件和用于产生特定频率的高频电信号的振荡的谐振器。为了实现输出可变功能,VCO主要内置具有变容二极管(可变电容)的可变谐振器,通过对变容二极管施加控制电压改变变容二极管的电容来改变输出频率。
在VCO中要重视的特性是相位噪声和输出频率。相位噪声是表示输出频率的稳定性的指标。在采用高频无线装置作为雷达或通信装置时对测距精度及通信误码率产生影响,因此相位噪声优选更低的值。
控制VCO的相位噪声的一种方法是改善谐振器的Q值(表示谐振器对于特定频率的电信号能够蓄积的能量的指标)。作为该方法之一例,报告了在谐振器使用多个短截线(stub)来作成具有高的Q值的谐振器的方法(例如,参照非专利文献1)。
此外,作为控制VCO的相位噪声的另一方法,有根据称为二次谐波信号、三次谐波信号、…的高次谐波信号抑制VCO内部的晶体管的谐振器侧端子上的电压发生变化的现象的方法(例如,参照非专利文献2)。
如此对抑制相位噪声的方法相关的提案数目很多,但是输出频率超过30GHz的VCO中,难以作成具有高的Q值的谐振器,得不到充分低的相位噪声特性。
此外,VCO最好直接输出高频无线装置处理的频率的信号。通过使用倍频器(frequency multiplier),也可使用输出比无线装置所处理的频率低的频率的信号的VCO,但是无线装置的结构变复杂,因此不利于低成本化。在发展无线装置的高频化的今天,希望提高VCO的输出频率。
非专利文献1:“A Low Phase Noise 19GHz-band VCO usina TwoDifferent Frequency Resonators”,IEEE MTT-S Int.Microwave Symp.Digest,pp.2189-2191,2003年
非专利文献2:“将高次谐波负载最优化的Ka波段二次谐波振荡器”信学技报,Vol.107,No.355,pp.29-32,2007年11月
随着输出频率的提高,原理上VCO的相位噪声增加即劣化。而且如果成为毫米波段以上的输出频率(超过30GHz),就难以作成具有高的Q值的谐振器,无法作成具有充分低的相位噪声特性的VCO。
在非专利文献1中记载的、使用多个谐振器的方法中,仅改善基波频率即振荡频率中的Q值,无法将高次谐波频率中电路的负载最优化。此外,在非专利文献2中记载的、抑制高次谐波信号产生的电压变动的方法中,仅考虑高次谐波频率中的电路负载,无法改善基波频率中的Q值。因此,在这些方法中,存在特别是输出频率超过大致30GHz的VCO中得不到充分低的相位噪声特性的问题。
发明内容
本发明为了解决上述问题构思而成,其目的在于得到即使在微波段(1GHz以上)或毫米波段(30GHz以上)的输出频率中,也可以实现低的相位噪声特性的电压控制振荡器(VCO)、MMIC(单片微波集成电路)及高频无线装置。
本发明的具备可变谐振器的电压控制振荡器,其特征在于:将至少一个前端开路短截线并联连接于所述可变谐振器上,所述前端开路短截线的长度最长也是在高次谐波信号的1/4波长的奇数倍上相加高次谐波信号的1/16波长的长度,而最短也是从高次谐波信号的1/4波长的奇数倍中减去高次谐波信号的1/16波长的长度。
此外,本发明的具备可变谐振器的电压控制振荡器,其特征在于:将至少一个前端短路短截线并联连接于所述可变谐振器上,所述前端短路短截线的长度最长也是在高次谐波信号的波长的整数倍上相加高次谐波信号的1/16波长的长度,而最短也是从高次谐波信号的波长的整数倍中减去高次谐波信号的1/16波长的长度。
(发明效果)
本发明的具备可变谐振器的电压控制振荡器,其特征在于:将至少一个前端开路短截线并联连接于所述可变谐振器上,所述前端开路短截线的长度最长也是在高次谐波信号的1/4波长的奇数倍上相加高次谐波信号的1/16波长的长度,而最短也是从高次谐波信号的1/4波长的奇数倍中减去高次谐波信号的1/16波长的长度。此外,本发明的具备可变谐振器的电压控制振荡器,其特征在于:将至少一个前端短路短截线并联连接于所述可变谐振器上,所述前端短路短截线的长度最长也是在高次谐波信号的波长的整数倍上相加高次谐波信号的1/16波长的长度,而最短也是从高次谐波信号的波长的整数倍中减去高次谐波信号的1/16波长的长度。从而,即使在微波段(1GHz以上)或毫米波段(30GHz以上)的输出频率中也可实现低的相位噪声特性。
附图说明
图1是表示本发明实施方式1的追加二次谐波中线路长度为λ/4的前端开路短截线的电压控制振荡器的结构的结构图。
图2是表示本发明实施方式2的设置二次谐波中线路长度为λ的前端短路短截线的电压控制振荡器的结构的结构图。
图3是表示本发明实施方式2的基波信号即38GHz的电场强度分布的说明图。
图4是表示本发明实施方式2的二次谐波信号即77GHz的电场强度分布的说明图。
图5是表示本发明实施方式2的二次谐波信号频率中谐振电路侧的阻抗和相位噪声的说明图。
图6是表示本发明实施方式3的设置具有二次谐波中线路长度为λ的线路和高频短路电容的偏置电路的电压控制振荡器的结构成的结构图。
图7是表示本发明实施方式4的设置二次谐波中成为短路负载的LCR电路的电压控制振荡器的结构的结构图。
图8是表示本发明实施方式4的设置二次谐波中成为短路负载的导波管电路的电压控制振荡器的结构的结构图。
图9是表示本发明实施方式5的设置二次谐波中成为短路负载的前端开路短截线的电压控制振荡器的结构的结构图。
图10是表示另一例本发明实施方式5的设置二次谐波中成为短路负载的前端开路短截线的电压控制振荡器的结构的结构图。
图11是表示另一例本发明实施方式5的设置二次谐波中成为短路负载的前端开路短截线及前端短路短截线的电压控制振荡器的结构的结构图。
图12是表示具备从实施方式1至5的电压控制振荡器的高频无线装置的结构的结构图。
(符号说明)
1晶体管;2变容二极管;3控制电压端子;4输出端子;5前端开路短截线;6前端短路短截线;7偏置电路,8 LCR电路;9导波管电路;20高频无线装置。
具体实施方式
实施方式1
图1是表示本发明实施方式1的VCO的结构的图。图1中示出串联正反馈结构的VCO,且示出振荡出频率为所希望频率的整数分之一的电信号(即基波信号)后从输出端子抽出高次谐波信号的高次谐波抽出型振荡器。1是晶体管,2是变容二极管,3是控制电压端子,4是信号的输出端子,5是相当于二次谐波信号的1/4波长的长度的前端开路短截线,12及13是线路,14是发射极线路,15是基波反射短截线,16是偏置电压端子。变容二极管2、线路12及控制电压端子3通过变容二极管2的可变电压的电容分量和线路12的电感分量来构成可变谐振器。可通过改变施加到控制电压端子3的控制电压Vt来改变输出频率。前端开路短截线5与上述可变谐振器并联连接。发射极线路14连接在晶体管1的发射极与接地之间。基波反射短截线15是例如相当于在电路内部振荡的基波的1/4波长的的前端开路短截线,如图1所示,连接于晶体管的输出侧安装的线路13上。
该VCO电路的结构为例如MMIC,此外也可采用MIC(微波集成电路)或离散(discrete)元件。衬底可以采用GaAs(砷化镓)、GaN(氮化镓)、InP(磷化铟)、Si等材料。
晶体管1的材料上没有限制,可以使用硅、砷化镓、氮化镓等。晶体管1的结构上也没有限制,可以使用双极型晶体管、场效应晶体管、高电子迁移率晶体管等,也可为真空管。
以下说明动作。电路内部的热噪声等噪声信号输入到晶体管1并得到放大,然后通过来自晶体管1的发射极线路14的反馈或来自基波反射短截线15的反射,经由线路13及晶体管1回到晶体管1的基极侧,再次输入到晶体管1得到放大。由此在VCO内部发生基波频率上的振荡,但晶体管1也会发生基波频率的2倍、3倍、…频率的高次谐波信号(二次谐波信号、三次谐波信号、…)。基波反射短截线15对于二次谐波信号而言是开路的,因此二次谐波信号向输出端子4对面的振荡器外部输出。基波信号不会比基波反射短截线15更靠近输出侧而传播,不会输出到振荡器外部。
若这些高次谐波信号传播到控制电压端子4,会使控制电压Vt变动,这样输出频率就会无意间变动。即输出频率的稳定性受损且增加相位噪声。为了抑制该控制电压Vt的变动而使基波信号通过,在晶体管1和线路12之间追加前端开路短截线5,吸收高次谐波信号。由于该前端开路短截线5而高次谐波信号无法在控制电压端子3中传播。另一方面,基波信号传播到变容二极管2,因此通过从外部改变控制电压Vt时的变容二极管2的电容变化,可以改变振荡频率。
在本实施方式中,前端开路短截线5的长度相当于二次谐波信号的1/4波长,因此基波频率中前端开路短截线5具有非短路且非开路的负载,通过来自晶体管1的发射极线路14的反馈或来自基波反射短截线15的反射而返回的基波信号传播到前端开路短截线5和变容二极管2这两方。因此,构成对基波使用多个短截线的谐振器,能够实现对基波高的Q值。这时基波与高次谐波在振荡器中具有6dB/oct的关系,因此能够减少基波与高次谐波这两方的相位噪声。另一方面,在二次谐波频率中前端开路短截线5具有短路负载,二次谐波信号全部向前端开路短截线5传播,因此在变容二极管2中不会传播二次谐波信号。因此,基于二次谐波信号的控制电压Vt的变动得到抑制,在具有变容二极管2的可变谐振电路中发生的相位噪声减少。此外,前端开路短截线5的连接部位不会产生二次谐波信号导致的电场变动,因此抑制了二次谐波信号产生的晶体管1的基极电压变动,而且减少了相位噪声。由以上能够实现低相位噪声的VCO。
图1中,作为一例,前端开路短截线5的线路长度为二次谐波信号的1/4波长,但也可为在二次谐波信号的1/4波长相加二次谐波信号的半波长整数倍后的长度。即,若设二次谐波信号的波长为λ,则由下述的式(1)定义的长度(二次谐波信号的1/4波长的奇数倍)即可。其原因是:相当于式(1)定义的长度的前端开路短截线,对于高次谐波具有短路负载,而对于基波具有非短路且非开路的负载。
(2n-1)λ/4(n=1、2、...) (1)
此外,前端开路短截线5的长度无需严格设定为上述式(1)的长度,可具有±λ/16左右的误差。其原因是:如果在该误差范围内,从针对谐振电路侧二次谐波负载阻抗的相位噪声大小计算结果,与严格按式(1)的长度设定时的相位噪声相比恶化止于0.8dB至1.4dB左右,能够充分期待位相噪声的抑制效果。
在本实施方式中,高次谐波信号设为二次谐波信号,但是如果三次谐波信号、四次谐波信号、...成为相位噪声的主要劣化因素,就可以使用设定为对于三次谐波信号、四次谐波信号、...的波长λ满足式(1)的线路长度的前端开路短截线,以在三次谐波频率、四次谐波频率、...中成为短路负载。此时即使有±λ/16的误差也能期待相位噪声的抑制效果。
还有,在图1的例中,示出只将一个前端开路短截线5并联连接于可变谐振器的例子,但不限于这种情况,也可将2以上的前端开路短截线5并联连接于可变谐振器。
通过以上结构,在本实施方式中,将一个以上的、最长也是在高次谐波信号的1/4波长的奇数倍上相加高次谐波信号的1/16波长的长度且最短也是从高次谐波信号的1/4波长的奇数倍减去高次谐波信号的1/16波长的长度的前端开路短截线5,并联连接于可变谐振器。上述前端开路短截线5具有在基波频率中非短路且非开路的负载,且具有在高次谐波频率中短路的负载,因此在基波频率中,基波信号传播到前端开路短截线5及变容二极管2这两方。即,构成使用多个短截线的谐振器,能够实现高的Q值。另一方面,在高次谐波频率中,前端开路短截线5具有短路负载,高次谐波信号全部传播到前端开路短截线5,因此高次谐波信号不会传播至变容二极管2,抑制高次谐波信号造成的控制电压Vt的变动。此外,在前端开路短截线5的连接部位中不会产生高次谐波信号造成的电场变动,因此抑制高次谐波信号造成的晶体管1的基极电压的变动。由以上可知,在本实施方式中,能够改善基波频率中的Q值,且能够抑制因施加到变容二极管及晶体管的电压的高次谐波信号而产生的变动,因此能够实现相位噪声低的VCO。
此外,图1中示出具有基波反射短截线15的高次谐波抽出型振荡器的例子,即使是不具有基波反射短截线15而输出基波的基波振荡器,也同样能实现相位噪声低的VCO。此外,即使在晶体管1的发射极侧或集电极侧连接上述可变谐振器的VCO中,在上述可变谐振器上并联连接前端开路短截线5就同样能够实现相位噪声低的VCO。
在本实施方式中,即使基波信号或高次谐波信号的频率不足1GHz,只要能够将前端开路短截线5的线路长度作成为式(1)表示的长度,就会起到与上述同样的效果。此外,图1中可变谐振器构成为具有变容二极管1和线路12,但由包含变容二极管的LCR电路构成也可。
实施方式2
图2是表示本发明实施方式2的VCO的结构的图。图2中,1~4和12~16与图1相同,6是长度相当于二次谐波信号的波长的前端短路短截线。与在二次谐波频率中具有短路负载的可变谐振器并联连接的短截线,采用前端短路短截线也能实现,对于二次谐波信号的波长λ,作成下述式(2)表示的线路长度(二次谐波信号的波长的整数倍)即可。
nλ(n=1、2、...) (2)
在大致不足1GHz的低频率中由式(2)表示的线路长度的前端短路短截线6在基波频率中也成为短路负载。因此,不会向包含变容二极管2的可变谐振器传播基波信号,不能改变振荡频率。另一方面,随着成为高频,因前端短路短截线6的线路中包含的寄生C分量或寄生L分量而由式(2)表示的线路长度不会成为基波信号的半波长的整数倍,因此式(2)的线路长度的前端短路短截线6在基波频率中具有非短路且非开路的负载。因此,在以大致1GHz以上的基波频率进行振荡的VCO中,能够使用式(2)表示的线路长度的前端短路短截线6,以取代实施方式1的前端开路短截线5。
本实施方式的VCO的工作原理基本上与实施方式1的VCO相同。在图3和图4中分别示出一例计算必须确认本实施方式的VCO的动作的、将电路结构作成MMIC并从晶体管基极端子向图2中的虚线框内输入38GHz基波信号及76GHz的二次谐波信号时的、图2中的虚线框内的电场分布。该计算是配置成在前端短路短截线6的中间连接偏置电压端子16的情况下进行,但是本质上与图2所示的VCO没有区别。
由图3所示的电场分布可知38GHz基波信号向前端短路短截线6及变容二极管2这两者传播。另一方面,由图4所示的电场分布可知76GHz的二次谐波信号仅向前端短路短截线6传播,不会向变容二极管2传播。而且,可知在晶体管基极端子中76GHz的二次谐波信号造成的电场成为零,即,基极电压不会变动。
表1中示出一例实施方式2的VCO的相位噪声的计算结果。由表1的结果可知在没有设置前端短路短截线6的情况下与设置的情况下,输出频率上没有大的变动,而通过追加前端短路短截线6,相位噪声得到抑制。此外,根据施加到控制电压端子3的电压,均可以做大致1GHz的频率变化。
[表1]
二次谐波的λ无前端短路短截线6 | 二次谐波的λ有前端短路短截线6 | |
输出频率 | 77.80GHz | 77.70GHz |
1MHz偏置的相位噪声 | -107.8dBc/Hz | -115.9dBc/Hz |
还有,在本实施方式中,也与实施方式1同样地,无需将前端短路短截线6严格设定为式(2)的长度,即使有±λ/16的误差也能期待相位噪声的抑制效果。图5中在50Ω的史密斯圆图(smith chart)中用点表示将前端短路短截线6的长度设为λ-λ/16、λ-λ/32、λ、λ+λ/32、λ+λ/16时的、从晶体管1的基极侧观看的谐振电路侧(可变谐振电路及前端短路短截线6)的二次谐波负载阻抗。此外,将针对谐振电路侧二次谐波负载阻抗的相位噪声大小的计算结果,以史密斯圆图上的0.2dB步长的等高线来表示。前端短路短截线6的长度为λ时,取最抑制相位噪声的最佳点即史密斯圆图的左端。随着长度从λ偏离,谐振电路侧的阻抗在史密斯圆图的外周上移动,可知相位噪声正在劣化。从计算结果可知当前端短路短截线6的长度成为λ±λ/16时,相位噪声从最佳点劣化0.8dB至1.4dB左右,但是该场合也能充分期待相位噪声的抑制效果。在使用前端开路短截线5的实施方式1中,针对谐振电路侧二次谐波负载阻抗的相位噪声大小的计算结果也图5相同。
此外,可以使用对于三次谐波信号、四次谐波信号、…的波长使线路长度满足式(2)的前端短路短截线,以在三次谐波频率、四次谐波频率、…中成为短路负载。此时即使具有±λ/16的误差也能期待相位噪声的抑制效果。
还有,在图2的例子中,示出只将一个前端短路短截线6并联连接于可变谐振器的例子,但并不限于此,可将2个以上的前端短路短截线6并联连接于可变谐振器。此外,前端短路短截线6的前端可经由以MIM(金属-绝缘体-金属)电容器为一例的电容,只使高频与接地端短路。
在本实施方式中,将1个以上的、最长也是在高次谐波信号的波长的整数倍上相加高次谐波信号的1/16波长的长度而最短也是从高次谐波信号的波长的整数倍减去高次谐波信号的1/16波长的长度的前端短路短截线,并联连接于可变谐振器。上述前端短路短截线6在基波频率中具有非短路且非开路的负载,且在高次谐波频率中具有短路的负载,因此在基波频率中,传播至前端短路短截线6及变容二极管2这两方。即,构成使用多个短截线的谐振器,能够实现高的Q值。另一方面,在高次谐波频率中,前端开路短截线5具有短路负载,高次谐波信号全部传播到前端短路短截线6,因此向变容二极管2不会传播高次谐波信号,抑制高次谐波信号造成的控制电压Vt的变动。此外,由于在前端短路短截线6的连接部位上不会产生高次谐波信号导致的电场变动,抑制高次谐波信号造成的晶体管1的基极电压的变动。由以上结构,在本实施方式中也与实施方式1同样,能够实现低相位噪声的VCO。
此外,在图2中示出具有基波反射短截线15的高次谐波抽出型振荡器的例子,但是不具有基波反射短截线15而输出基波的基波振荡器也同样能实现相位噪声低的VCO。此外,即使在可变谐振器连接到晶体管1的发射极侧或集电极侧的VCO中,只要前端短路短截线6并联连接于上述可变谐振器,也同样能实现相位噪声低的VCO。
图2中可变谐振器构成为具有变容二极管1和线路12,但是由包含变容二极管的LCR电路构成也可。
实施方式3
图6是表示本发明实施方式3的VCO的结构的图。图6中,1~4和12~15与图1相同,7是从连接部位经由电容(电容器)11到高频短路部位的线路长度相当于二次谐波信号的波长的偏置电路。
如在上述实施方式2中说明的那样,即便没有重新增加前端短路短截线6,在偏置电路中,通过从连接部位仅偏离满足上述式(2)的距离的部位上经由电容11进行短路,也能得到与上述实施方式2的前端短路短截线的追加同样的效果。
此外,在上述说明中,偏置电路7的线路长度相当于二次谐波信号的波长的长度情况下进行了说明,但并不限于这种场合,偏置电路7的线路长度相当于二次谐波信号的波长的整数倍的长度即可。此外,并不限于二次谐波信号而相当于三次谐波以上的高次谐波波长的整数倍的长度也可。
此外,即使偏置电路7的线路长度有±λ/16的误差也能期待相位噪声的抑制效果。
如上所述,在本实施方式中,从偏置电路连接部位经由电容器到接地部位的线路长度,最长也是在高次谐波信号的波长的整数倍上相加高次谐波信号的1/16波长的长度,而最短也是从高次谐波信号的波长的整数倍减去高次谐波信号的1/16波长的长度,将具有这种长度的偏置电路并联连接于可变谐振器,因此与实施方式2同样地,在本实施方式中也能实现低相位噪声的VCO。
实施方式4
图7是表示本发明实施方式4的VCO的结构的图。图7中,1~4和12~16与图1相同,8是在二次谐波频率中具有短路负载的LCR电路。
此外,图8是表示本发明的实施方式4的另一例VCO的结构的图。图8中,1~4和12~16与图1相同,9是在二次谐波频率中具有短路负载的导波管电路。
此外,LCR电路8和导波管电路9在高次谐波信号的频率中都是短路负载或接近短路负载的例如-30jΩ以上+30jΩ以下范围的负载。该范围的负载在50Ω的类型中使特性阻抗对应于实施方式2的前端短路短截线6时相当于λ±λ/16的范围,由图5可知,相位噪声收缩在从最佳点劣化0.8dB至1.4dB左右的范围,因此具有抑制相位噪声的效果。此外,由图5的史密斯圆图可知,即使设阻抗为在上述虚数分量外还具有0Ω以上15Ω以下的实数分量,相位噪声也收缩在从最佳点劣化0.8dB至1.4dB左右的范围,具有抑制相位噪声的效果。
在上述实施方式1或2中追加的电路,在基波频率中具有非短路且非开路的负载而在高次谐波频率中具有短路负载即可,无需为线路短截线。因而,如本实施方式所示,也可以使用LCR电路8或导波管电路9。
如上所述,在本实施方式中,至少将一个对于基本频率不短路且在高次谐波信号的频率中具有实数分量为0Ω以上15Ω以下且虚数分量为-30jΩ以上+30jΩ以下的负载的LCR电路8或导波管电路9,并联连接于可变谐振器,因此与上述实施方式2或3同样,能够实现低相位噪声的VCO。
实施方式5
如图9所示,也可以追加多个上述实施方式1、2、4中追加的电路。图9中示出连接3个长度相当于实施方式1中所示的二次谐波信号的1/4波长的前端开路短截线5、5A、5B的例子,便是可为实施方式2中所示的前端短路短截线6,也可为实施方式4中所示的LCR电路8或导波管电路9。此外,数目也不限于3个,可为适当的任意数目。
此外,如果多个次数的高次谐波信号成为相位噪声的劣化因素,则如图10所示,也可以使多个追加的电路相对于各自不同次数的高次谐波信号成为短路负载。在图10的例子中,追加了长度与实施方式1所示的三次谐波信号的1/4波长相当的前端开路短截线5;长度与三次谐波信号的1/4波长相当的前端开路短截线5C;以及长度与四次谐波信号的1/4波长相当的前端开路短截线5D,但这是一个例子,并不限定于此。基于相位噪声的劣化因素,适当选择组合。
再者,如图11所示,多个追加的电路可与前端短路短截线、前端开路短截线、LCR电路和导波管电路不同。在图11的例子中,设置了长度与四次谐波信号的波长相当的前端短路短截线6A;长度与三次谐波信号的1/4波长相当的前端开路短截线5E;以及长与度实施方式2中所示的二次谐波信号的波长相当的前端短路短截线6,但并不限定于此。对此也可根据相位噪声的劣化因素,适选择组合。
如上所述,在本实施方式中也与上述实施方式1、2、4同样,能够实现低位相噪声的VCO。
实施方式6
图12是具备实施方式1至5的电压控制振荡器的高频无线装置的结构例。高频无线装置20是雷达或便携电话等,使用微波或毫米波进行发送或接收或这两方的装置。
电压控制振荡器22以来自频率控制装置21的电压信号产生的频率进行振荡,用放大器23放大振荡信号,由输出天线2发送微波或毫米波。利用接收天线25接收微波或毫米波,基于频率控制装置26的电压信号,将电压控制振荡器27输出的振荡信号与来自接收天线25的接收信号,利用变频器28来进行频率变换后输出所希望的信号。
发送天线24和接收天线25可为一体。频率控制装置21和26、电压控制振荡器22和27分别为一也可。此外,发送部分和接收部分中的一个使用实施方式1至5的电压控制振荡器也可。
高频无线装置使用实施方式1至5的电压控制振荡器,从而能够发送相位噪声少的高品质的微波或毫米波。此外能够减少接收时的噪声。
Claims (7)
1.一种具备可变谐振器的电压控制振荡器,其特征在于:
将至少一个前端开路短截线并联连接于所述可变谐振器上,所述前端开路短截线的长度最长也是在高次谐波信号的1/4波长的奇数倍上相加高次谐波信号的1/16波长的长度,而最短也是从高次谐波信号的1/4波长的奇数倍中减去高次谐波信号的1/16波长的长度,
所述电压控制振荡器还具备:具有输入和输出的晶体管,
所述可变谐振器具有连接到所述晶体管的所述输入的第一端,
所述可变谐振器具有由施加给所述可变谐振器的电压所控制的可变的谐振频率,
所述前端开路短截线连接到所述晶体管的所述输入。
2.一种具备可变谐振器的电压控制振荡器,其特征在于:
将至少一个前端短路短截线并联连接于所述可变谐振器上,所述前端短路短截线的长度最长也是在高次谐波信号的波长的整数倍上相加高次谐波信号的1/16波长的长度,最短也是从高次谐波信号的波长的整数倍中减去高次谐波信号的1/16波长的长度,
所述电压控制振荡器还具备:具有输入和输出的晶体管,
所述可变谐振器具有连接到所述晶体管的所述输入的第一端,
所述可变谐振器具有由施加给所述可变谐振器的电压所控制的可变的谐振频率,
所述前端短路短截线连接到所述晶体管的所述输入。
3.一种具备可变谐振器的电压控制振荡器,其特征在于:
将偏置电路并联连接于所述可变谐振器上,所述偏置电路中,从偏置电路连接部位经由电容器到接地部位的线路长度,最长也是在高次谐波信号的波长的整数倍上相加高次谐波信号的1/16波长的长度,而最短也是从高次谐波信号的波长的整数倍中减去高次谐波信号的1/16波长的长度,
所述电压控制振荡器还具备:具有输入和输出的晶体管,
所述可变谐振器具有连接到所述晶体管的所述输入的第一端,
所述可变谐振器具有由施加给所述可变谐振器的电压所控制的可变的谐振频率,
所述偏置电路连接到所述晶体管的所述输入。
4.一种具备可变谐振器的电压控制振荡器,其特征在于:
将至少一个LCR电路并联连接于所述可变谐振器上,所述LCR电路相对于振荡频率不是短路,而在高次谐波信号的频率中,具有实数分量为0Ω以上15Ω以下、且虚数分量为-30jΩ以上+30jΩ以下的负载,
所述电压控制振荡器还具备:具有输入和输出的晶体管,
所述可变谐振器具有连接到所述晶体管的所述输入的第一端,
所述可变谐振器具有由施加给所述可变谐振器的电压所控制的可变的谐振频率,
所述LCR电路连接到所述晶体管的所述输入。
5.一种具备可变谐振器的电压控制振荡器,其特征在于:
将至少一个导波管电路并联连接于所述可变谐振器,所述导波管电路相对于振荡频率不是短路,而在高次谐波信号的频率中,具有实数分量为0Ω以上15Ω以下、且虚数分量为-30jΩ以上+30jΩ以下的负载,
所述电压控制振荡器还具备:具有输入和输出的晶体管,
所述可变谐振器具有连接到所述晶体管的所述输入的第一端,
所述可变谐振器具有由施加给所述可变谐振器的电压所控制的可变的谐振频率,
所述导波管电路连接到所述晶体管的所述输入。
6.一种单片微波集成电路,具备权利要求1至5中任一项所述的电压控制振荡器。
7.一种高频无线装置,具备权利要求1至5中任一项所述的电压控制振荡器。
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US9973940B1 (en) | 2017-02-27 | 2018-05-15 | At&T Intellectual Property I, L.P. | Apparatus and methods for dynamic impedance matching of a guided wave launcher |
US10298293B2 (en) | 2017-03-13 | 2019-05-21 | At&T Intellectual Property I, L.P. | Apparatus of communication utilizing wireless network devices |
CN108736838A (zh) * | 2017-04-16 | 2018-11-02 | 天津大学(青岛)海洋工程研究院有限公司 | 一种高次谐波可控的新型f类功放匹配电路 |
CN110337759B (zh) * | 2017-10-13 | 2022-05-13 | 株式会社友华 | 高频模块 |
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- 2009-06-30 CN CN200910159440.0A patent/CN101667829B/zh not_active Expired - Fee Related
- 2009-08-04 DE DE102009036098A patent/DE102009036098A1/de not_active Withdrawn
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JP2910830B2 (ja) * | 1995-09-19 | 1999-06-23 | 日本電気株式会社 | 誘電体共振発振器 |
US5886595A (en) * | 1996-05-01 | 1999-03-23 | Raytheon Company | Odd order MESFET frequency multiplier |
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US20100052799A1 (en) | 2010-03-04 |
JP2010062614A (ja) | 2010-03-18 |
CN101667829A (zh) | 2010-03-10 |
DE102009036098A1 (de) | 2010-03-25 |
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