CN101505140B - Trans-impedance amplifier with low noise and high gain-bandwidth product - Google Patents

Abstract

The invention provides an optical-receiver preamplifier for receiving analog or digital optical signals. The preamplifier comprises a high-gain amplifier A1, an input circuit, a negative-feedback impedor Zf, a low-gain amplifier A2 adjustable in gain and a feedback capacitor Cff, wherein the input circuit takes a photoelectric converter as a main component; the negative-feedback impedor Zf is connected with the reverse input end and the output end of the A1; the feedback capacitor Cff connects the output end of the A2 with the reverse input end of the A1; and the output end of the A1 can be directly connected with the input end of the A2 or can be connected with the input end of the A2 through a buffer. Output signals of the preamplifier are taken out from the output end of the A1 or the output end of the buffer, and can be directly subjected to subsequent signal processing or be further amplified and then subjected to subsequent signal processing.

Description

A kind of low noise high-gain-bandwidth product transimpedance amplifier

Technical field

The invention belongs to responsive electronics and sensor field in electronics and the information system; Be specifically related to a kind of low noise, high-gain-bandwidth product transimpedance amplifier; This amplifier is applicable to the optical receiver of analog or digital light signal; In the light signal that is specially adapted to receive, be extracted the corresponding faint analog or digital optical receiver of light intensity of signal, and bandwidth, highly sensitive, above-mentioned optical receiver that signal to noise ratio is high.

Background technology

Transimpedance amplifier (hereinafter to be referred as TIA) is the preamplifier that extensively adopts in the optical receiver; Its effect is light signal to be shone the low current signal that produces on the photoelectric device convert voltage signal into; Photoelectric device can be p-i-n type photodiode PIN; Or avalanche photodide APD etc., below be that the representative representative is described with PIN.Important performances such as the bandwidth of optical receiver, sensitivity, stability, dynamic range depend on the signal bandwidth B of TIA _SIG, signal gain A _SIG, the noise voltage gain A _N, unit bandwidth output noise level e _O, phase close is than (Closure Rate, Rate-of-Closure) or characteristics such as the abundant remaining PM of phase place (Phase Margin), dynamic range, therefore these characteristics also become is high-performance TIA leading indicator.

TIA has two kinds of single-ended imported and both-ends imported (or it is imported to be called difference).The imported common mode inhibition capacity of both-end is superior to single-ended imported; But the performance of aspects such as its open-loop gain, low-frequency cut-off frequency, noiseproof feature, dynamic range, bandwidth all is inferior to single-ended imported (seeing the description of United States Patent (USP) 5343160 and 6433638), so single-ended is the principal mode of TIA.Fig. 1-Fig. 4 is several kinds of forms of existing single-ended imported TIA.

Fig. 1 is the citation form of existing TIA, can be used as the basis of analyzing the TIA performance.Among the figure, the C that dotted line connects _fBe feedback resistance R _fParasitic capacitance, I _PhBe the output current of PIN, R _Sh, R _InBe PIN parallel resistance and amplifier input resistance, C _TBe that the total input capacitance of TIA (comprises PIN shunt capacitance C _PIN, amplifier input capacitance C _In, amplifier in wiring capacitance and other parasitic capacitance C _pDeng).In this citation form of Fig. 1, the signal bandwidth B of transimpedance amplifier TIA _SIG, signal gain A _SIG, the noise voltage gain A _NWith output noise level e _ORelation (descriptions of the document that sees reference [1]-[5]) shown in the formula of following (1)-(4) with circuit parameter:

$A_{\mathrm{SIG}}=\frac{V_O}{I_{\mathrm{ph}}}=-\frac{A_{\mathrm{<figure-callout\; id="1"\; label="OL"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">OL</figure-callout>}}}{\frac{1+A_{\mathrm{OL}}}{Z_f}+\frac{1}{Z_{\mathrm{in}}}}\&ap;\frac{-{\mathrm{<figure-callout\; id="1"\; label="R\; f"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">R</figure-callout>}}_f}{1+s{R}_f/(1+A_{\mathrm{OL}})((1+A_{\mathrm{OL}})C_f+C_T)}\&ap;\frac{-{\mathrm{<figure-callout\; id="1"\; label="R\; f"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">R</figure-callout>}}_f}{1+s{R}_f{C}_f}---\left(1\right)$

$B_{\mathrm{SIG}}=1/\left(2\mathrm{\&pi;}{\mathrm{\&tau;}}_{\mathrm{SIG}}\right)=\frac{1}{2\mathrm{\&pi;}\frac{{\mathrm{<figure-callout\; id="1"\; label="R\; f"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">R</figure-callout>}}_f}{1+A_{\mathrm{OL}}}((1+A_{\mathrm{OL}})C_f+C_T)}---\left(2\right)$

$A_N=\frac{1+s(C_T+C_f)R_f}{1+s{C}_f{R}_f}=\frac{1+s{C}_{\mathrm{\&Sigma;}}{R}_{\mathrm{<figure-callout\; id="1"\; label="f"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">f</figure-callout>}}}{1+s{C}_f{R}_f}---\left(3\right)$

$e_O=\sqrt{(2q{I}_{\mathrm{ph}}+4\mathrm{kT}/\left(R_{\mathrm{sh}}\right|\left|R_{\mathrm{in}}\right)+i_n)\&CenterDot;A_{\mathrm{SIG}}^2+e_n^2\&CenterDot;A_N^2+4k{\mathrm{TR}}_f}---\left(4\right)$

A in the formula _OLBe the open-loop gain of TIA, C _∑=C _T+ C _f, I _PhBe PIN average output current (comprising signal code and non-signal code), 2qI _PhBe I _PhThe power spectral density of the shot noise (shot noise) that produces, i _n, e _nBe the input current noise and the input voltage noise of amplifier, q=1.6 * 10 ^-19Coulomb is an electron charge, k=1.38 * 10 ^-23Joule/K is the graceful constant of Bohr thatch, and T is an absolute temperature.A _OL, A _SIG, A _NSee Fig. 5 with the relation of frequency.It should be noted that at A especially _NF at zero point _zWith limit f _pBetween, A _N(f) slope with 20dB/decade rises, and this has not only increased the contribution of input voltage noise to the output voltage noise, and occurs A easily _N(f) and A _OL(f) phase close between makes the TIA system unstable than the situation that equals 40dB/decade, specifies see reference document [1]-[4], [6], [7].

Variously know A by above _SIGAmplitude is with R _fAnd increase e _OWith (R _f) ^1/2And increase, therefore increase R _fHelp improving the signal to noise ratio of TIA output signal.Yet R _fIncrease will cause B _SIGDescend, and noise voltage rises when causing high frequency; But this adverse effect can be through reducing C _T, C _fOffset, increase R so the scheme of the existing TIA of improvement performance all concentrates on _fWith (or) reduction C _T, C _fInfluence this two aspect.

Increase R _fValue to be improving the TIA noiseproof feature, to improve TIA gain-bandwidth product, or at increase R _fThe scheme of the dynamic range of expansion TIA is a lot of under the prerequisite of value; With the patented technology is example; External like United States Patent (USP) 5343160,5455705,5521555,5532471,5889605,5982232,7221229,7330668; European patent EP 720729, International Patent Application WO/2006/017846 etc.; Domestic the patent of seeing the dynamic range of expansion TIA: 91306421.x, denomination of invention is " preamplifier for super-dynamic-range optical receiver ", but does not relate to R _fOr the noiseproof feature of TIA.

It is less relatively to the scheme of TIA performance impact to reduce feedback capacity or input stage equivalent capacity, mainly contains following several kinds:

1. adopt best feedback capacity

As shown in Figure 2, at the R of basic TIA _fThe feedback capacity with specific capacitance values is added at two ends, makes the noise voltage gain A _NAt its pole frequency f _p=1/ (2 π R _fC _f) time amplitude equal open-loop gain A _OLAmplitude when this frequency, i.e. A _N(f _p)=A _OL(f _p), this just can adopt the narrow as far as possible amplifier of bandwidth to meet the requirements of the TIA bandwidth, can make the abundant remaining of phase place of noise voltage gain reach 45 ° again, thereby the TIA working stability is unlikely to cause vibration.The best feedback capacity value that can reach this requirement is:

$C_{\mathrm{fop}}=\sqrt{\frac{C_T}{2\mathrm{\&pi;}{R}_f{f}_{\mathrm{GBW}}}}-C_{\mathrm{Rf}}---\left(5\right)$

F in the formula _GBWThe open-loop gain that is amplifier is 1 o'clock a frequency, C _RfBe R _fParasitic capacitance.Detailed description see reference document [1], [2].

2. adopt R-C offset-type feedback network

As shown in Figure 3, the R of employing Fig. 1, Fig. 2 _f-C _fDuring the parallel connection type feedback network, feedback impedance Z _fThe equivalent input impedance that causes at the TIA input is Z _f/ (1+A _OL), corresponding equivalent input resistance is R _Fin=R _f/ (1+A _OL), equivalent input capacitance is C _Fin=(1+A _OL) C _fDetails are referring to document [8]

When adopting R-C offset-type feedback network, at R _fC _f=R _cC _CCondition under (R _f＞＞R _C, C _C＞＞C _f), be equivalent to feedback loop cross-over connection R at TIA _Feff=R _f+ R _C≈ R _fPure resistance, insert C at input _fWith C _CThe pure capacitor C of series connection _Fin=C _fC _C/ (C _f+ C _C) ≈ C _fR _FeffThe equivalent input resistance that produces at input is R _Fin=(R _f+ R _c)/(1+A _OL) ≈ R _f/ (1+A _OL), identical with the situation of Fig. 1 or Fig. 2, but C at this moment _FinBe the 1/ (1+A of Fig. 1, Fig. 2 _OL), therefore this TIA can reduce C greatly _fTo the influence of TIA input circuit time constant, but can not reduce C _TInfluence.

3. adopt bootstrapping (bootstrap) input stage

As shown in Figure 4, the input stage of this TIA is a voltage follower, and PIN is connected in parallel on input, the output of voltage follower, and the ac potential at PIN two ends is identical, C _PINThere is not the interchange charge and discharge process, eliminated C _PINTo the TIA timeconstant _SIGInfluence, but can not eliminate other electric capacity (like C _f, C _In, C _pDeng) to τ _SIGInfluence.Specifically referring to United States Patent (USP) 4535233,5521555 and European patent EP 720729.

More than analyze explanation, existing TIA can not eliminate C simultaneously _PIN, C _f, C _In, C _pEtc. various electric capacity to the TIA timeconstant _SIGInfluence, thereby can not make full use of the open loop characteristic of amplifier, can not adopt the R of bigger resistance value _fImprove the noiseproof feature of TIA, therefore should further seek to eliminate the scheme of various electric capacity the influence of TIA time constant.

Referenced patent

United States Patent (USP): 4535233,5343160,5455705,5521555,5532471,5889605,5982232,6433638,7221229,7330668; European patent: EP720729; International patent application: WO/2006/017846; Chinese patent: 91306421.x; One Chinese patent application: 200610078460.1,200580026425.9.

List of references

[1].David?Westerman，Understand?and?Apply?the?Transimpedance?Amplifier，MobileHandset?Design?Line，08/08/2007，available：

http://www.mobilehandsetdesignline.com/howto/201400084

[2].National?Semiconductor?Application?Note?180，Design?Considerations?for?aTransimpedance?Amplifier，Maithil?Pachchigar，February?28，2008

[3].P?Wright，K?B?Ozanyan，S?J?Carey?and?H?McCann，Optimisation?of?the?Signal?toNoise?Performance?of?Photodiode?Receivers?in?Near-Infrared?Absorption?Tomography，Proc.3rd?World?Congress?on?Industrial?Process?Tomography(Banff，Canada，September?2003)，pp.219-225.

[4].Burr-Brown?Application?Bulletin，Noise?Analysis?of?FET?Transimpedance?Amplifier，February，1994.

[5].Toby?Whitley，Transimpetance?Feedback?Amplifier，Bath?University?Handout，available：

http://people.bath.ac.uk/tw236/TRANSIMPEDANCE％20FEEDBACK％20AMPLIFIER. doc

[6].Bonnie?Baker，Transimpedace-amplifier?stability?is?key?in?light-sensing?application，EDN，September?4，2008.

[7].Tim?Green，“Operational?Amplifier?Stability，Part?1?of?15：Loop?Stability?Basics，”Texas?Instruments，2005，AnalogZone，Acquistion-Zone.

[8].J.L.Hullett?and?T.V.Muoi，A?Feedback?Receiver?for?Optical?Transmission?Systems[J]，IEEE?Trans.Communication，1976，24(10)：1180-1185.

Summary of the invention

The objective of the invention is to improve the noiseproof feature of existing TIA, improve gain-bandwidth product, improve sensitivity, improve stability.

Technical scheme of the present invention has provided a kind of transimpedance type optical receiver preamplifier TIA; Comprise: a high-gain amplifier A1; Input circuit with optical-electrical converter; One is connected the reverse input end of said high-gain amplifier A1 and the degeneration impedance Zf of output, a low gain amplifier A2, the feedback capacity Cff of the output of a said low gain amplifier A2 of connection and the reverse input end of said high-gain amplifier A1; The output of said high-gain amplifier A1 is connected with the input of said low gain amplifier A2; The output signal of said preamplifier takes out from the output of said high-gain amplifier A1; Said output signal directly carries out follow-up signal to be handled, or carries out follow-up signal again after further amplifying and handle.

Wherein, The output of said high-gain amplifier A1 is connected with the input of a buffer Buffer; The input of said low gain amplifier A2 is connected with the output of buffer Buffer, and the output signal of said preamplifier takes out from the output of buffer Buffer.

Wherein, said low gain amplifier A2 is equipped with the Gain Adjustable device.

Wherein, said degeneration impedance Zf is composed in parallel by resistance R _ f and capacitor C f.

Wherein, said degeneration impedance Zf is made up of resistance R _ f, RC and capacitor C f, CC, and wherein an end of resistance R _ f is connected with the reverse input end of said high-gain amplifier A1 and the end of capacitor C f; The other end of resistance R _ f and the other end of capacitor C f and the end of resistance R C, the end of capacitor C C are connected, and the other end of resistance R C is connected with the output of said low gain amplifier A2, the other end earthing potential of capacitor C C.

Wherein, The low gain amplifier A2 of said buffer Buffer and Gain Adjustable is integrated; The low gain amplifier A2 of this incorporate buffer Buffer and Gain Adjustable is made up of a field-effect transistor JFET and two resistance R N, RP at least; Wherein the grid of field-effect transistor JFET is connected with the output of high-gain amplifier A1 among the said TIA, and source electrode is connected with resistance R P, and the other end of resistance R P is connected with signal ground; Drain electrode is connected with resistance R N, and the other end of resistance R N is connected with signal power source; The two ends of feedback impedance Zf are connected with the input of said high-gain amplifier A1, the source electrode of field-effect transistor JFET respectively among the said transimpedance amplifier TIA; The two ends of feedback capacity Cfi are connected with the input of said high-gain amplifier A1, the drain electrode of field-effect transistor JFET respectively among the transimpedance amplifier TIA, and the output signal of transimpedance amplifier TIA takes out from the source electrode of field-effect transistor JFET.

Wherein, the two ends of said feedback impedance Zf are connected with input, the output of said high-gain amplifier A1 respectively.

Wherein, The low gain amplifier A2 of said buffer Buffer and Gain Adjustable is integrated; The low gain amplifier A2 of this integrated buffer Buffer and Gain Adjustable is made up of a bipolar transistor BJT and two resistance R N, RP at least; Wherein the base stage of bipolar transistor BJT is connected with the output of high-gain amplifier A1 among the said transimpedance amplifier TIA, and emitter is connected with resistance R P, and the other end of resistance R P is connected with signal ground; Collector electrode is connected with resistance R N, and the other end of resistance R N is connected with signal power source; The two ends of feedback impedance Zf are connected with the input of said high-gain amplifier A1, the emitter of bipolar transistor BJT respectively among the said transimpedance amplifier TIA; The two ends of feedback capacity Cfi are connected with the input of said high-gain amplifier A1, the collector electrode of bipolar transistor BJT respectively among the transimpedance amplifier TIA, and the output signal of transimpedance amplifier TIA takes out from the emitter of bipolar transistor BJT.

Wherein, the two ends of said feedback impedance Zf are connected with input, the output of said high-gain amplifier A1 respectively.

The invention has the beneficial effects as follows:

(1) utilizes the positive feedback capacitor C according to transimpedance amplifier of the present invention _FfAt negative Miller (Miller) electric capacity that the TIA input produces, offset the original C of TIA input _PIN, C _In, C _p, and negative feedback C _fEquivalent input capacitance (1+A in the generation of TIA input _OL) C _f, to increase the signal bandwidth of TIA.

(2) can adopt bigger R according to transimpedance amplifier of the present invention than existing scheme resistance value _fImproving signal gain and the signal to noise ratio of TIA, but do not influence the bandwidth of TIA.

(3) utilize the positive feedback capacitor C according to transimpedance amplifier of the present invention _FfNegative Miller capacitance in that the TIA input produces changes the noise voltage gain A _N(ω) zero, pole location is eliminated A _N(ω) at the rising part of front end, make the noise voltage gain be always 0dB, reduce of the contribution of amplifier input noise voltage, and can guarantee the stability of TIA TIA output noise voltage.

(4) can be according to transimpedance amplifier of the present invention at R _fThe capacitor C of the big capacitance of two ends parallel connection _f, in order to the unsettled R of basic elimination capacitance _fThe influence of parasitic capacitance and wiring capacitance, but do not influence the bandwidth of TIA.

(5) can be according to transimpedance amplifier of the present invention easily through adjustment low gain booster amplifier A ₂Gain adjust the negative capacitance value that positive feedback electric capacity produces at the TIA input, with comprise R _fNumerical value such as parasitic capacitance, line layout electric capacity be difficult to predetermined influencing factor and be complementary in interior TIA input equivalent capacity.This fixed capacitor that just can adopt stable performance is as positive feedback electric capacity, and needn't adopt the variable capacitor of unstable properties.

Description of drawings

Below in conjunction with accompanying drawing the present invention is further specified.

Fig. 1 is the structural representation of the basic structure type transimpedance amplifier TIA of prior art;

Fig. 2 is the structural representation of the best feedback capacity type transimpedance amplifier TIA of prior art, and its purpose is to make the TIA working stability, is unlikely to cause vibration because of noise voltage;

Fig. 3 is the structural representation of the R-C compensation feedback network type transimpedance amplifier TIA of prior art, and its purpose is to reduce the influence of feedback capacity to TIA bandwidth and noiseproof feature;

Fig. 4 is the structural representation of the bootstrapping input stage type transimpedance amplifier TIA of prior art, and its purpose is to reduce the influence of PIN electric capacity to TIA bandwidth and noiseproof feature, and the voltage gain of buffer among the figure (Buffer) is 1;

Fig. 5 is the graph of relation between transimpedance amplifier TIA open-loop gain, signal gain, noise voltage gain and the frequency;

Fig. 6 is according to the structural representation (a) of the transimpedance amplifier TIA of embodiments of the invention 1 and schematic equivalent circuit (b) thereof;

Fig. 7 is the input voltage noise e according to the transimpedance amplifier TIA of embodiments of the invention 1 _nConvert equivalent input current noise I into _EnThe process sketch map;

Fig. 8 is the TIA performance comparison sheet according to the transimpedance amplifier TIA embodiment of embodiments of the invention 1 and prior art;

Fig. 9 is the structural representation according to the transimpedance amplifier TIA of embodiments of the invention 2;

Figure 10 is the structural representation according to the transimpedance amplifier TIA of embodiments of the invention 3;

Figure 11 is the structural representation (R according to the transimpedance amplifier TIA of embodiments of the invention 4 _N＜＜R _P).

Embodiment

Embodiment 1

Fig. 6 is according to the structural representation (a) of the transimpedance amplifier TIA of embodiment of the present invention 1 and schematic equivalent circuit (b) embodiment thereof.PIN is the optical-electrical converter that light signal is converted into current signal among the figure, A ₁Be main amplifier, A ₂Be booster amplifier, their open-loop gain also respectively note make A ₁, A ₂, and satisfy condition:

A ₁＞＞1，A ₂＜1，，A ₁·A ₂＝A _ff＞1；

Buffer is that voltage gain is 1 buffer, is not the indispensable part of present embodiment; R ' _InBe PIN parallel resistance R _ShWith amplifier input resistance R _InThe equivalent resistance that is in parallel, C _TBe that the total input capacitance of TIA (comprises PIN shunt capacitance C _PIN, amplifier input capacitance C _In, amplifier in wiring capacitance and other parasitic capacitance C _pDeng); R _fBe to connect A ₁Output N to A ₁The negative feedback resistor of input M, C _fBe R _fParasitic capacitance, C _FfBe to connect A ₂Output M ' is to A ₁The positive feedback electric capacity of input M (M ', the signal phase homophase located of 2 of M).Can know R by Fig. 6 (b) _f, C _fAt A ₁It is R that input M produces equivalent input resistance _Fin=R _f/ (1+A ₁), equivalent input capacitance is C _Fin=(1+A ₁) C _fWhen not having positive feedback electric capacity, A ₁The total equivalent input resistance R of input _∑With total equivalent input capacitance C _∑For:

R _∑f＝R′ _in‖R _f/(1+A ₁)≈R _f/(1+A ₁)(R′ _in＞＞R _f/(1+A ₁))

C _∑f＝C _T‖(1+A ₁)C _f＝C _T+(1+A ₁)C _f

The timeconstant of TIA input _SIGWith signal bandwidth B _SIGFor:

τ _SIG＝2πR _∑f·C _∑f≈2π[R _f/(1+A ₁)]·[(1+A ₁)C _f+C _T]＝2πR _f[C _f+C _T/(1+A ₁)]

$B_{\mathrm{SIG}}=1/{\mathrm{\&tau;}}_{\mathrm{SIG}}=\frac{1}{2\mathrm{\&pi;}{R}_f[C_f+C_T/(1+A_1)]}$

$B_{\mathrm{SIG}}=1/{\mathrm{\&tau;}}_{\mathrm{SIG}}=\frac{1}{2\mathrm{\&pi;}{R}_f[C_f+C_T/(1+A_1)]}$

Because this TIA has very big equivalent input capacitance C _∑, at this moment adopt the very R of high resistance for the noise ratio that improves TIA _f, just be difficult to obtain very wide signal bandwidth.

Adopt positive feedback capacitor C provided by the invention _FfThe time, C _FfTo produce the negative value Miller capacitance at TIA input M place:

$C_{\mathrm{ffin}}=-\frac{C_{\mathrm{ff}}}{A_{\mathrm{ff}}-1}$

Thereby TIA input M place:

R _∑ff＝R _∑f＝R′ _in‖R _f/(1+A ₁)，

$C_{\mathrm{\&Sigma;ff}}=C_{\mathrm{\&Sigma;f}}+C_{\mathrm{ffin}}=C_{\mathrm{\&Sigma;f}}-\frac{C_{\mathrm{ff}}}{A_{\mathrm{ff}}-1},$

Adjustment A _Ff(adjustment A ₂), make C _Ffin≈-C _{∑ f}, just can make the equivalent input capacitance of TIA be about zero.Even at this moment adopt the very R of high resistance _f, also can obtain very wide signal bandwidth.

C _FfCan also eliminate the noise voltage gain A _N(ω) at the rising part of front end, make A _N(ω) be always 0dB, reduce of the contribution of amplifier input noise voltage, and can guarantee that TIA does not vibrate TIA output noise voltage.

Derivation A _NProcess (ω) is seen Fig. 7 (a) and (b), (c), (d), Z ' among the figure _In=R ' _In‖ C _T, Z _f=R _f‖ C _fGet by (a):

$\frac{V_{\mathrm{no}}}{{\mathrm{<figure-callout\; id="1"\; label="A"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">A</figure-callout>}}_1}=-\frac{e_n(\frac{1}{{Z^{\&prime;}}_{\mathrm{in}}}+\frac{1}{Z_f})}{\frac{1}{{Z^{\&prime;}}_{\mathrm{in}}}+\frac{1+A_1}{Z_f}}$

Get by (b) and Miller equivalent electric circuit (c) thereof:

$\frac{V_{\mathrm{no}}}{{\mathrm{<figure-callout\; id="1"\; label="A"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">A</figure-callout>}}_1}=\frac{I_{\mathrm{En}}}{\frac{1}{{Z^{\&prime;}}_{\mathrm{in}}}+\frac{1+A_1}{Z_f}}$

If (c) be equivalent to (a), i.e. output voltage noise V in two circuit _NoEquate, then

$I_{\mathrm{En}}=-e_n(\frac{1}{{Z^{\&prime;}}_{\mathrm{in}}}+\frac{1}{Z_f})$

So I _EnThe V that causes _NoFor:

$V_{\mathrm{no}}=I_{\mathrm{En}}\&CenterDot;A_{\mathrm{SIG}}=-e_n(\frac{1}{{Z^{\&prime;}}_{\mathrm{in}}}+\frac{1}{Z_f})\frac{-{\mathrm{<figure-callout\; id="1"\; label="R\; f"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">R</figure-callout>}}_f}{1+s{R}_f/(1+A_1)((1+A_1)C_f+C_T)}$

At R ' _In＞＞1/ (j ω C _T), (1+A ₁C _fThe C of)＞＞ _TCondition under, Z ' _In=1/ (j ω C _T), A _SIG=-R _f/ (1+j ω R _fC _f), following formula becomes:

$V_{\mathrm{no}}=\frac{1+\mathrm{j\&omega;}{R}_f(C_f+C_T)}{1+\mathrm{j\&omega;}{R}_f{C}_f}{e}_n$

Thus noise voltage gain A by this _NFor:

$A_N=\frac{V_{\mathrm{no}}}{e_n}=\frac{1+s{R}_f(C_f+C_T)}{1+s{R}_f{C}_f}{e}_n$

The A that generally adopts in Here it is the document _NExpression formula, i.e. formula (3).

As Z ' _InIn capacitor C _TWith Z _f/ (1+A ₁) in electric capacity (1+A ₁) C _fAll by C _FfWhen the negative Miller capacitance that produces is offset, circuit (c) will become circuit (d), I _EnTo become

$I_{\mathrm{En}}=-e_n(\frac{1}{{R^{\&prime;}}_{\mathrm{in}}}+\frac{1}{R_f})=e_n\frac{{R^{\&prime;}}_{\mathrm{in}}+R_f}{{R^{\&prime;}}_{\mathrm{in}}{R}_f}\&ap;-e_n/R_f,({R^{\&prime;}}_{\mathrm{in}}>>R_f)$

So I _EnThe V that causes _NoBecome:

$V_{\mathrm{no}}=I_{\mathrm{En}}\&CenterDot;A_{\mathrm{SIG}}=-\frac{1}{R_f}\frac{-{\mathrm{<figure-callout\; id="1"\; label="R\; f"\; filenames="H2009100790428E00021.png,H2009100790428E00031.png"\; state="\{\{state\}\}">R</figure-callout>}}_f}{1+s{R}_f/(1+A_1)((1+A_1)C_f+C_T)}\&CenterDot;e_n$

In the following formula, because C _TWith (1+A ₁) C _fAll be cancelled, so V _No=e _n, promptly

$A_N\left(f\right)=\frac{V_{\mathrm{no}}}{e_n}=1$

In other words, noise voltage gain at this moment is equivalent to the A among Fig. 5 for 0dB _N(f) curve is a horizontal linear, A _N(f) and A _OL(f) phase close between than (rate-of-closure) is :-20dB/decade-0dB/decade=-20dB/decade, so TIA stable (document that sees reference [7]).

The TIA performance of embodiment example 1 and prior art TIA performance relatively see Fig. 8.With R among Fig. 8 _fThe performance of (relative value)=1.8 o'clock is an example, gain-bandwidth product of the TIA of embodiment example 1, be respectively the basic transimpedance type TIA that uses always and R-C compensation feedback network type TIA 9.02 times of gain-bandwidth product with 7.19 times.This means with back two kinds of TIA and compare, under the identical situation of bandwidth, the TIA of embodiment example 1 can adopt resistance value higher feedback resistance R _fThereby, can reduce R _fCorresponding equivalent input noise current, the signal to noise ratio of raising TIA; At R _fThe identical situation of resistance value under, the TIA of embodiment example 1 can have wideer bandwidth, thereby can receive the more light signal of broad frequency range.

Embodiment 2

Embodiment of the present invention 2 are referring to Fig. 9, PIN, A among the figure ₁, A ₂, Buffer, C _T, R ' _In, R _f, C _FfDefinition identical with embodiment 1, C _fBe greater than R _fThe fixed capacity of parasitic capacitance can make the equivalent input capacitance C of TIA _∑Can be because of R _fParasitic capacitance is unstable and influence the serviceability of TIA.According to the analysis of embodiment 1, adjustment A ₂Can make the equivalent input capacitance of TIA be about zero, thereby adopt the very R of high resistance _fThe time also can obtain very wide signal bandwidth, and can make noise voltage gain be 0dB, make TIA stable.

Embodiment 3

Embodiment of the present invention 3 are seen shown in Figure 10, PIN, A among the figure ₁, A ₂, Buffer, C _T, R ' _In, C _FfDefinition identical with embodiment 1, R _f, C _f, R _C, C _CThe common A that forms ₁Feedback network Z _f, and the R that satisfies condition _f＞＞R _c, C _c＞＞C _f, R _fC _f=R _CC _CDifferent with embodiment 1 and 2 is, at this moment feedback network Z _fThe equivalent input capacitance that produces at the TIA input is C _Fin=C _fC _c/ (C _f+ C _c) ≈ C _f, rather than (the 1+A in embodiment 1 and 2 ₁) C _f, this is with regard to the C of available smaller capacitive value _FfThe equivalent input capacitance of compensation TIA; The equivalence input resistance is R _Fin=(R _f+ R _c)/(1+A ₁) ≈ R _f/ (1+A ₁), identical with embodiment 1,2.Adjustment A ₂Can make the equivalent input capacitance of TIA be about zero equally, thereby adopt the very R of high resistance _fThe time also can obtain very wide signal bandwidth, and can make noise voltage gain be 0dB, make TIA stable.

Embodiment 4

Embodiment of the present invention 4 is with buffer Buffer in the embodiment 1,2,3 and booster amplifier A ₂The scheme that is integrated is shown in Figure 11 (a) and (b), (c), (d).JFET is a junction field effect transistor among the figure, and BJT is a bipolar transistor, and biasing circuit and feed circuit omit.Get R _N＜＜R _P, so that the input impedance of JFET level, BJT level is very high, output voltage and A ₁Output end voltage is basic identical, thereby the circuit function of JFET level, BJT level is identical with buffer.By figure, A ₂≈ R _N/ R _P＜＜1, but A ₁＞＞1, can guarantee A thus ₁A ₂=A _Ff＞1.C _FfBe connected between M, the M ', and this signal phase homophase of 2, so C _FfCan introduce the negative value Miller capacitance C that requires in the embodiment 1,2,3 at the input M place of TIA _Ffin=-C _Ff/ (A _Ff-1).Adjustment R _NCan adjust A ₂, and then adjustment A _Ff, C _Ffin, reach the purpose that the equivalent input capacitance that makes TIA is compensated.

Invention has been described according to specific exemplary embodiment here.It will be conspicuous under not departing from the scope of the present invention, carrying out suitable replacement to one skilled in the art or revise.Exemplary embodiment only is illustrative, rather than to the restriction of scope of the present invention, scope of the present invention is by appended claim definition.

Claims (9)

1. a transimpedance type optical receiver preamplifier TIA comprises: a high-gain amplifier A ₁, a input circuit with optical-electrical converter, one is connected said high-gain amplifier A ₁Reverse input end and the degeneration impedance Z of output _f, a low gain amplifier A ₂, one connects said low gain amplifier A ₂Output and said high-gain amplifier A ₁The feedback capacity C of reverse input end _FfSaid high-gain amplifier A ₁Output and said low gain amplifier A ₂Input connect, the output signal of said preamplifier is from said high-gain amplifier A ₁Output take out, said output signal directly carries out follow-up signal to be handled, or carries out follow-up signal again after further amplifying and handle.

2. transimpedance type optical receiver preamplifier TIA as claimed in claim 1 is characterized in that said high-gain amplifier A ₁Output be connected said low gain amplifier A with the input of a buffer Buffer ₂Input be connected with the output of buffer Buffer, the output signal of said preamplifier takes out from the output of buffer Buffer.

3. according to claim 1 or claim 2 transimpedance type optical receiver preamplifier TIA is characterized in that said low gain amplifier A ₂The Gain Adjustable device is equipped with.

4. according to claim 1 or claim 2 transimpedance type optical receiver preamplifier TIA is characterized in that said degeneration impedance Z _fBy resistance R _fWith capacitor C _fCompose in parallel.

5. according to claim 1 or claim 2 transimpedance type optical receiver preamplifier TIA is characterized in that said degeneration impedance Z _fBy resistance R _f, R _CWith capacitor C _f, C _CForm, wherein resistance R _fAn end and said high-gain amplifier A ₁Reverse input end and capacitor C _fAn end connect; Resistance R _fThe other end and capacitor C _fThe other end and resistance R _CAn end, capacitor C _CAn end connect resistance R _CThe other end and said low gain amplifier A ₂Input connect capacitor C _COther end earthing potential.

6. transimpedance type optical receiver preamplifier TIA as claimed in claim 1 is characterized in that the low gain amplifier A of said buffer Buffer and Gain Adjustable ₂Be integrated the low gain amplifier A of this incorporate buffer Buffer and Gain Adjustable ₂By a field-effect transistor JFET and at least two resistance R _N, R _PForm, wherein high-gain amplifier A among the grid of field-effect transistor JFET and the said TIA ₁Output connect source electrode and resistance R _PConnect resistance R _PThe other end be connected with signal ground, the drain electrode and resistance R _NConnect resistance R _NThe other end be connected with signal power source; Feedback impedance Z among the said transimpedance type optical receiver preamplifier TIA _fTwo ends respectively with said high-gain amplifier A ₁The source electrode of reverse input end, field-effect transistor JFET connect feedback capacity C among the transimpedance type optical receiver preamplifier TIA _FfTwo ends respectively with said high-gain amplifier A ₁The drain electrode of reverse input end, field-effect transistor JFET connect, the output signal of transimpedance type optical receiver preamplifier TIA takes out from the source electrode of field-effect transistor JFET.

7. transimpedance type optical receiver preamplifier TIA as claimed in claim 6 is characterized in that said feedback impedance Z _fTwo ends respectively with said high-gain amplifier A ₁Input, output connect.

8. transimpedance type optical receiver preamplifier TIA as claimed in claim 1 is characterized in that the low gain amplifier A of said buffer Buffer and Gain Adjustable ₂Be integrated the low gain amplifier A of this integrated buffer Buffer and Gain Adjustable ₂By a bipolar transistor BJT and at least two resistance R _N, R _PForm, wherein high-gain amplifier A among the base stage of bipolar transistor BJT and the said transimpedance type optical receiver preamplifier TIA ₁Inverse output terminal connect emitter and resistance R _PConnect; Feedback impedance Z among the said transimpedance type optical receiver preamplifier TIA _fTwo ends respectively with said high-gain amplifier A ₁The emitter of reverse input end, bipolar transistor BJT connect feedback capacity C among the transimpedance type optical receiver preamplifier TIA _FfTwo ends respectively with said high-gain amplifier A ₁The collector electrode of reverse input end, bipolar transistor BJT connect, the output signal of transimpedance type optical receiver preamplifier TIA takes out from the emitter of bipolar transistor BJT.

9. transimpedance type optical receiver preamplifier TIA as claimed in claim 8 is characterized in that said feedback impedance Z _fTwo ends respectively with said high-gain amplifier A ₁Reverse input, output connect.

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