WO2024114548A1 - 信息传输方法、通信设备及存储介质 - Google Patents

信息传输方法、通信设备及存储介质 Download PDF

Info

Publication number
WO2024114548A1
WO2024114548A1 PCT/CN2023/134170 CN2023134170W WO2024114548A1 WO 2024114548 A1 WO2024114548 A1 WO 2024114548A1 CN 2023134170 W CN2023134170 W CN 2023134170W WO 2024114548 A1 WO2024114548 A1 WO 2024114548A1
Authority
WO
WIPO (PCT)
Prior art keywords
value
constellation
model
equal
modulation symbol
Prior art date
Application number
PCT/CN2023/134170
Other languages
English (en)
French (fr)
Inventor
袁志锋
李志岗
马一华
李卫敏
Original Assignee
中兴通讯股份有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 中兴通讯股份有限公司 filed Critical 中兴通讯股份有限公司
Publication of WO2024114548A1 publication Critical patent/WO2024114548A1/zh

Links

Definitions

  • the embodiments of the present application relate to but are not limited to the field of communication technology, and in particular, to an information transmission method, a communication device, and a storage medium.
  • the wireless communication system needs to transmit information with a large number of terminals. Furthermore, if the spectrum efficiency of information transmission of each terminal is required not to be too low, it is also necessary to increase the order of the modulation method to improve the spectrum efficiency.
  • the commonly used high-order modulation method is Quadrature Amplitude Modulation (QAM), such as 16QAM, 32QAM, 64QAM and 256QAM.
  • QAM Quadrature Amplitude Modulation
  • these high-order modulation methods can only ensure performance when the channel estimation is relatively accurate. If the channel estimation error is large, the constellation diagram will be distorted during demodulation, which will lead to a decrease in demodulation performance, making it difficult to achieve high-spectrum-efficiency information transmission.
  • the massive terminals and the system transmit information, which will increase the pilot overhead at the terminal. If the pilot overhead is too large, it will be difficult to ensure the accuracy of the channel estimation. Therefore, it will also limit the demodulation performance of the base station or access point, and it is also difficult to achieve high-spectrum-efficiency information transmission. Therefore, how to support massive terminals to achieve high-spectrum-efficiency information transmission is an urgent problem to be solved.
  • the embodiments of the present application provide an information transmission method, a communication device, and a storage medium, which can support a large number of first communication nodes to achieve high-spectrum-efficiency information transmission.
  • an embodiment of the present application provides an information transmission method, which is applied to a first communication node.
  • the information transmission method includes:
  • the first number is greater than or equal to 1, and the data packet contains at least a modulation symbol
  • the complex forms corresponding to the 2*N1 constellation points in the first constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N1 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N1 e j( ⁇ + ⁇ ) ;
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N1 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N1 ;
  • the complex forms corresponding to the 4*N2 constellation points in the second constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N2 e j ⁇ , b 1 e j( ⁇ + ⁇ /2) ,b 2 e j( ⁇ + ⁇ /2) ,...,b N2 e j( ⁇ + ⁇ /2) , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N2 e j( ⁇ + ⁇ ) , b 1 e j( ⁇ +3 ⁇ /2) ,b 2 e j( ⁇ +3 ⁇ /2) ,...,b N2 e j( ⁇ +3 ⁇ /2) ;
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 , ..., a N2 and b 1 , b 2 , ..., b N2 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N2 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N2 ;
  • the complex forms corresponding to the 8*N3 constellation points in the third constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N3 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N3 e j( ⁇ + ⁇ ) ,
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N3 and b 1 , b 2 ,..., b N3 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N3 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N3 .
  • an embodiment of the present application provides an information transmission method, which is applied to a second communication node, and the information transmission method includes:
  • the first number is greater than or equal to 1, and the data packet contains at least a modulation symbol
  • the complex forms corresponding to the 2*N1 constellation points in the first constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N1 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N1 e j( ⁇ + ⁇ ) ;
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N1 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N1 ;
  • the complex forms corresponding to the 4*N2 constellation points in the second constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N2 e j ⁇ , b 1 e j( ⁇ + ⁇ /2) ,b 2 e j( ⁇ + ⁇ /2) ,...,b N2 e j( ⁇ + ⁇ /2) , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N2 e j( ⁇ + ⁇ ) , b 1 e j( ⁇ +3 ⁇ /2) ,b 2 e j( ⁇ +3 ⁇ /2) ,...,b N2 e j( ⁇ +3 ⁇ /2) ;
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 , ..., a N2 and b 1 , b 2 , ..., b N2 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N2 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N2 ;
  • the complex forms corresponding to the 8*N3 constellation points in the third constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N3 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N3 e j( ⁇ + ⁇ ) ,
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N3 and b 1 , b 2 ,..., b N3 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N3 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N3 .
  • an embodiment of the present application further provides a communication device, including: a memory, a processor, and a computer program stored in the memory and executable on the processor, wherein the processor implements the information transmission method as described above when executing the computer program.
  • the present application also provides a computer-readable storage medium storing computer-executable instructions.
  • the executable instructions are used to execute the information transmission method as described above.
  • an embodiment of the present application further provides a computer program product, comprising a computer program or computer instructions, wherein the computer program or the computer instructions are stored in a computer-readable storage medium, a processor of a computer device reads the computer program or the computer instructions from the computer-readable storage medium, and the processor executes the computer program or the computer instructions, so that the computer device performs the information transmission method as described above.
  • An embodiment of the present application includes: determining a first number of extremely sparse pilots, and then transmitting a data packet and the first number of extremely sparse pilots to a second communication node, wherein the first number is greater than or equal to 1, and the data packet at least includes a modulation symbol, and the modulation symbol is obtained by modulating M1+1 bits of information in the data packet according to a first constellation model, or by modulating M2+2 bits of information in the data packet according to a second constellation model, or by modulating M3+3 bits of information in the data packet according to a third constellation model, and M1 is an integer greater than or equal to 1, M2 is an integer greater than or equal to 1, and M3 is an integer greater than or equal to 0, that is, the first communication node transmits an extremely sparse pilot and a data packet containing the modulation symbol to the second communication node, so as to facilitate the transmission of the extremely sparse pilot and the data packet to the second communication node.
  • the second communication node can estimate part of the information of the wireless channel from the extremely sparse pilot, and further extract the channel information from the modulation symbol, without estimating all the information of the wireless channel from the extremely sparse pilot, thereby reducing the pilot overhead of the first communication node, and then improving the accuracy of the channel estimation, thereby improving the demodulation performance of the second communication node, and realizing high-spectral-efficiency information transmission; and because M1 is an integer greater than or equal to 1, M2 is an integer greater than or equal to 1, and M3 is an integer greater than or equal to 0, the modulation symbol can be obtained by modulating multiple bits of information in the data packet according to the first constellation model, the second constellation model, or the third constellation model, that is, the modulation symbol can carry multiple bits of information, thereby realizing high-order modulation, and then improving the spectrum efficiency of information transmission. Therefore, the embodiment of the present application can support a large number of first communication nodes to realize high-spectral-efficiency information transmission.
  • FIG1 is a constellation diagram corresponding to a 64QAM modulation symbol provided by an embodiment of the present application.
  • FIG2 is a flow chart of an information transmission method provided by an embodiment of the present application.
  • FIG3 is a schematic diagram of w mutually independent pilots provided by an embodiment of the present application.
  • FIG4 is a flow chart of an information transmission method provided by another embodiment of the present application.
  • FIG5 is a schematic diagram of a first constellation diagram model and a second constellation diagram model provided by an embodiment of the present application
  • FIG6 is a schematic diagram of a cross-shaped constellation diagram before and after channel rotation and scaling provided by an embodiment of the present application
  • FIG7 is a schematic diagram of dividing partitions on a two-dimensional plane coordinate system provided by an embodiment of the present application.
  • FIG8 is a schematic diagram of a cross-shaped constellation diagram provided by an embodiment of the present application.
  • FIG9 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG10 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG11 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG12 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG13 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG14 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG15 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG16 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG17 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG18 is a schematic diagram of a cross-shaped constellation diagram provided by another embodiment of the present application.
  • FIG19 is a schematic diagram of a PAM constellation diagram provided by an embodiment of the present application.
  • FIG20 is a schematic diagram of a PAM constellation diagram provided by another embodiment of the present application.
  • FIG21 is a schematic diagram of a third constellation diagram model provided by an embodiment of the present application.
  • FIG22 is a schematic diagram of a third constellation diagram model provided by another embodiment of the present application.
  • FIG23 is a schematic diagram of defining a physical resource block according to an embodiment of the present application.
  • FIG24 is a schematic diagram of defining a demodulation reference signal provided by an embodiment of the present application.
  • FIG25 is a schematic diagram of distinguishing different reference signal ports by OCC codes according to an embodiment of the present application.
  • FIG26 is a schematic diagram of defining a demodulation reference signal using an OCC code according to an embodiment of the present application.
  • FIG27 is a schematic diagram of another method of defining a demodulation reference signal using an OCC code according to an embodiment of the present application.
  • FIG28 is a schematic diagram of another method of defining a demodulation reference signal using an OCC code according to an embodiment of the present application.
  • FIG29 is a schematic diagram of a PRB-based transmission provided by an embodiment of the present application.
  • FIG30 is a schematic diagram of another method for defining a demodulation reference signal provided by an embodiment of the present application.
  • FIG31 is a schematic diagram of another method for defining a demodulation reference signal provided by an embodiment of the present application.
  • FIG32 is a schematic diagram of another method for defining a demodulation reference signal provided by an embodiment of the present application.
  • FIG33 is a schematic diagram of defining a reference signal provided by an embodiment of the present application.
  • FIG34 is a schematic diagram of another method for defining a reference signal provided by an embodiment of the present application.
  • FIG35 is a schematic diagram of another method for defining a reference signal provided by an embodiment of the present application.
  • FIG36 is a schematic diagram of generating a DMRS port provided by an embodiment of the present application.
  • FIG37 is a schematic diagram of another method for defining a reference signal provided by an embodiment of the present application.
  • FIG38 is a schematic diagram of another method for defining a reference signal provided by an embodiment of the present application.
  • Figure 39 is a schematic diagram of the structure of a communication device provided by an embodiment of the present application.
  • the present application provides an information transmission method, communication device and storage medium, wherein a first communication node can determine a first number of extremely sparse pilots, and then transmit a data packet and the first number of extremely sparse pilots to a second communication node, wherein the first number is greater than or equal to 1, and the data packet at least includes a modulation symbol, and the modulation symbol is obtained by modulating M1+1 bits of information in the data packet according to a first constellation diagram model, or by modulating M2+2 bits of information in the data packet according to a second constellation diagram model, or by modulating M3+3 bits of information in the data packet according to a third constellation diagram model, and M1 is an integer greater than or equal to 1, M2 is an integer greater than or equal to 1, and M3 is an integer greater than or equal to 0, that is, the first communication node transmits an extremely sparse pilot and the modulation symbol to the second communication node.
  • the second communication node can estimate part of the information of the wireless channel from the extremely sparse pilot, and further extract the channel information from the modulation symbol, without estimating all the information of the wireless channel from the extremely sparse pilot, thereby reducing the pilot overhead of the first communication node, and then improving the accuracy of the channel estimation, thereby improving the demodulation performance of the second communication node, and realizing high-spectral-efficiency information transmission; and because M1 is an integer greater than or equal to 1, M2 is an integer greater than or equal to 1, and M3 is an integer greater than or equal to 0, the modulation symbol can be obtained by modulating multiple bits of information in the data packet according to the first constellation model, the second constellation model, or the third constellation model, that is, the modulation symbol can carry multiple bits of information, thereby realizing high-order modulation, and then improving the spectrum efficiency of information transmission. Therefore, the embodiment of the present application can support a large number of first communication nodes to realize high-spectral-efficiency information transmission.
  • the wireless communication system needs to transmit information with a large number of terminals. Furthermore, if the spectrum efficiency of each terminal for information transmission is not too low, it is also necessary to increase the order of the modulation method to improve the spectrum efficiency.
  • the commonly used high-order modulation method is orthogonal amplitude modulation, such as 16QAM, 32QAM, 64QAM and 256QAM, and the constellation points in the constellation diagram are evenly distributed on the two-dimensional plane (i.e., the complex plane), so the two-dimensional signal space of the complex signal (i.e., the two-dimensional signal plane) can be fully utilized.
  • the communication signal can usually be represented by a complex number on the baseband, that is, the communication signal can be divided into an I-path signal and a Q-path signal, wherein the I-path signal is the real part and the Q-path signal is the imaginary part.
  • the constellation diagram includes 16 points, and the complex numbers corresponding to these 16 points include the following: 3+3j,3+j,3-j,3-3j, 1+3j,1+j,1-j,1-3j, -1+3j,-1+j,-1-j,-1-3j, -3+3j,-3+j,-3-j,-3-3j
  • the 16QAM constellation points are distributed relatively evenly.
  • the complex plane is equivalent to the two-dimensional plane, so the complex plane or the two-dimensional plane can also be called a two-dimensional complex plane, where the real part of the complex number is equivalent to the x-coordinate of the two-dimensional plane, and the imaginary part of the complex number is equivalent to the y-coordinate of the two-dimensional plane. Therefore, complex numbers can also be represented by points on a two-dimensional plane.
  • the complex number a+j*b can be represented by the coordinates (a, b) on the two-dimensional plane, where the coordinates (a, b) indicate that the x-coordinate on the two-dimensional plane is a and the y-coordinate is b.
  • the 16 points in the 16QAM constellation diagram can also be represented by 16 two-dimensional coordinates on a two-dimensional plane, where the 16 two-dimensional coordinates include the following: (3,3),(3,1),(3,-1),(3,-3), (1,3),(1,1),(1,-1),(1,-3), (-1,3),(-1,1),(-1,-1),(-1,-3), (-3,3),(-3,1),(-3,-1),(-3,-3)
  • the entire constellation diagram may be multiplied by a normalization factor (or scaling factor).
  • a normalization factor or scaling factor
  • the 16 complex numbers corresponding to the 16QAM modulation mode are all multiplied by the same normalization factor 1/sqrt(40).
  • the complex numbers corresponding to the 16 points included in the 16QAM constellation diagram after power normalization processing include the following: 1/sqrt(40)*[3+3j,3+j,3-j,3-3j,1+3j,1+j,1-j,1-3j,-1+3j,-1+j,-1-j,-1-3j,-3 +3j,-3+j,-3-j,-3-3j]
  • the coordinates corresponding to the 16 points in the power normalized 16QAM constellation diagram can be obtained by multiplying the 16 two-dimensional coordinates listed above by 1/sqrt(40), that is, multiplying the x-coordinate and y-coordinate of each two-dimensional coordinate by 1/sqrt(40).
  • high-order modulation methods such as 32QAM, 64QAM, 256QAM and other modulation methods
  • the constellation points of the constellation diagram are evenly distributed on the two-dimensional plane
  • the high-order modulation methods in the related technology can make full use of the two-dimensional signal space of the complex signal
  • the demodulation methods corresponding to these high-order modulation methods are not only simple, but also can guarantee performance. Therefore, these high-order modulation methods can approach the performance limit of transmission, that is, the Shannon limit, relatively simply and efficiently. Therefore, in scenarios where there is a certain demand for spectrum efficiency, these high-order modulation methods can be widely used.
  • these high-order modulation methods need to ensure performance when the channel estimation is relatively accurate. If the channel estimation error is large, the constellation diagram will be distorted when the base station (or access point) demodulates, that is, rotated and scaled, and the demodulation performance will decrease.
  • Orthogonal Frequency Division Multiplexing i.e., using OFDM subcarriers to transmit modulation symbols
  • OFDM Orthogonal Frequency Division Multiplexing
  • the modulation symbols carried on the OFDM subcarriers will be weighted with a complex weight (i.e., the frequency-selective channel will cause the modulation symbols carried on the subcarriers to be distorted); or, if there is a large synchronization error between the sender and the receiver (i.e., the first communication node and the second communication node), the timing deviation (i.e., time deviation) and frequency deviation (i.e., frequency deviation) will also cause the modulation symbols on the subcarriers to be weighted with a complex weight, i.e., the synchronization error will cause the modulation symbols to be distorted.
  • OFDM Orthogonal Frequency Division Multiplexing
  • the Doppler effect will also cause the modulation symbols on the subcarriers to be weighted with a complex weight, i.e., the synchronization error will cause the modulation symbols to be distorted.
  • the rotation and scaling amount of the modulation symbol i.e., the rotation amount and scaling amount
  • the rotation and scaling amount and scaling amount changes very quickly in scenarios with large synchronization errors
  • high-speed mobile scenarios such as high-speed mobile scenarios or satellite communication scenarios, etc.
  • a pilot with a very short time interval is required to accurately estimate the rotation and scaling amount of the modulation symbol between the pilots.
  • this will increase the pilot overhead and ultimately reduce the spectral efficiency of the transmission.
  • the pilot density in time does not meet the requirements, it is also difficult to accurately estimate the rotation and scaling amount of the modulation symbol, so the demodulation performance will also be reduced.
  • the receiving side i.e., the second communication node
  • the modulation symbol will be rotated and scaled.
  • the modulation symbols that have been slightly rotated and scaled will also seriously restrict the performance of high-order modulation methods.
  • each small dot in the figure corresponds to a modulation symbol, where the constellation diagram corresponding to the coordinate system on the left side of Figure 1 is the constellation diagram corresponding to the standard 64QAM modulation symbol; and the constellation diagram corresponding to the coordinate system on the right side of Figure 1 is the constellation diagram corresponding to the 64QAM modulation symbol weighted by a weight value (that is, a rotation and scaling amount), that is, the constellation diagram corresponding to the 64QAM modulation symbol that has undergone channel distortion.
  • a weight value that is, a rotation and scaling amount
  • the terminal usually does not establish a connection with the system when there is no need to transmit information (that is, the terminal is not connected to the system, or the terminal is disconnected from the system), that is, the terminal is in a disconnected state (wherein, Non Connected state, or Non RRC Connected state, or Connectionless state, or Connection-free state, or Disconnected state, etc. can all represent a disconnected state).
  • a disconnected state wherein, Non Connected state, or Non RRC Connected state, or Connectionless state, or Connection-free state, or Disconnected state, etc. can all represent a disconnected state.
  • the idle state (Idle state) or the inactive state (Inactive state) can be considered to be equivalent to the connectionless state, or the idle state (Idle state) or the inactive state (Inactive state) can also be considered to be a connectionless state.
  • the terminal When the terminal is originally in a disconnected state (i.e., has not yet entered a connected state, or has not yet established a connection with the system), if the relevant technology is used In the uplink information transmission scheme in the , in order to transmit information, the terminal must establish a connection with the system before transmission. After entering the connection state (also called the active state), the terminal can further apply for uplink transmission resources from the system (such as a base station or access point), and can only transmit information after obtaining resource authorization or resource scheduling from the system. It can be seen that in order to complete a traditional uplink information transmission, the terminal needs to complete many operations in advance, which will undoubtedly increase the power consumption generated by the terminal and the signaling overhead of the system.
  • the terminal transmits information with the system in a connectionless state, then before the information is transmitted, the terminal does not need to establish a connection or apply for dedicated transmission resources from the base station (or access point), that is, the terminal in a connectionless state does not need to notify the base station (or access point) before information transmission, but autonomously transmits information to the base station directly on a preset public transmission resource. Therefore, information transmission in a connectionless state can reduce the complexity of information transmission, reduce the power consumption generated by the terminal during information transmission, and the transmission delay of data to the base station (or access point), and also saves the signaling overhead required for information transmission.
  • the first communication node (such as a terminal) needs to autonomously select a pilot (or reference signal) from a preset pilot set.
  • a pilot or reference signal
  • this will lead to a problem, that is, since there is no central node to coordinate the pilots sent by different first communication nodes, different first communication nodes autonomously select pilots from a preset pilot set with a limited number of pilots, and the selected pilots may be the same, which will cause the problem of pilot collision.
  • high overload i.e., there are many first communication nodes sending data packets to the same second communication node
  • the probability of pilot collision is very high. Once the pilots of different first communication nodes collide, it is difficult for the second communication node to separate multiple first communication nodes through the pilots.
  • each pilot needs to have a signal in the entire transmission bandwidth and time.
  • each pilot cannot be distributed too sparsely in the entire transmission bandwidth and time, so that the channel information of the entire transmission bandwidth (such as wireless multipath channel, that is, frequency selective channel) and the frequency offset in the transmission time can be estimated. Therefore, to ensure the transmission performance of information transmission in a connectionless state, the use of the traditional pilot scheme will cause the overhead occupied by the pilot to increase exponentially, and the detection complexity will also increase significantly.
  • the present application proposes to support a large number of terminals to transmit information at a higher spectrum efficiency by combining extremely sparse pilots and modulation methods.
  • the scheme based on extremely sparse pilots allows the system to support more terminal access. Therefore, the scheme based on extremely sparse pilots is very suitable for large connection scenarios, such as scenarios where a large number of terminals directly transmit information in a disconnected state, and scenarios where a large number of users transmit information based on SPS.
  • the scheme based on extremely sparse pilots requires that the receiving side (i.e., the second communication node) be able to perform channel estimation based on the characteristics of the modulation symbols themselves, while the constellation diagram of the traditional high-order modulation method is too dense, which is not conducive to the receiving side extracting channel information through modulation symbols.
  • the present application proposes an information transmission method that can not only support high spectral effect application scenarios, but also reduce the complexity of the receiving side extracting channel information through modulation symbols, and improve the accuracy of the channel information extracted by the receiving side through modulation symbols.
  • FIG 2 is a flowchart of an information transmission method provided by an embodiment of the present application.
  • the information transmission method is applied to the first communication node, and the information transmission method may include but is not limited to step S110 and step S120.
  • Step S110 Determine a first number of extremely sparse pilots.
  • Step S120 Transmit the data packet and the first number of extremely sparse pilots to the second communication node.
  • the first number is greater than or equal to 1, and the data packet at least includes a modulation symbol.
  • the complex forms corresponding to the 2*N1 constellation points in the first constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N1 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N1 e j( ⁇ + ⁇ )
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N1 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N1 ;
  • the complex forms corresponding to the 4*N2 constellation points in the second constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N2 e j ⁇ , b 1 e j( ⁇ + ⁇ /2) ,b 2 e j( ⁇ + ⁇ /2) ,...,b N2 e j( ⁇ + ⁇ /2) , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N2 e j( ⁇ + ⁇ ) , b 1 e j( ⁇ +3 ⁇ /2) ,b 2 e j( ⁇ +3 ⁇ /2) ,...,b N2 e j( ⁇ +3 ⁇ /2) ;
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 , ..., a N2 and b 1 , b 2 , ..., b N2 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N2 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N2 ;
  • the complex forms corresponding to the 8*N3 constellation points in the third constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N3 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N3 e j( ⁇ + ⁇ ) ,
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N3 and b 1 , b 2 ,..., b N3 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N3 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N3 .
  • a first number of extremely sparse pilots can be determined, and then a data packet and the first number of extremely sparse pilots are transmitted to the second communication node together, wherein the first number is greater than or equal to 1, and the data packet at least includes a modulation symbol, and the modulation symbol is obtained by modulating M1+1 bits of information in the data packet according to the first constellation diagram model, or modulating M2+2 bits of information in the data packet according to the second constellation diagram model, or modulating M3+3 bits of information in the data packet according to the third constellation diagram model, and M1 is an integer greater than or equal to 1, M2 is an integer greater than or equal to 1, and M3 is an integer greater than or equal to 0, that is, the first communication node transmits an extremely sparse pilot and a data packet including the first number of extremely sparse pilots to the second communication node.
  • the data packet of the modulation symbol is so that the second communication node can estimate part of the information of the wireless channel from the extremely sparse pilot, and then further extract the channel information from the modulation symbol, without estimating all the information of the wireless channel from the extremely sparse pilot, thereby reducing the pilot overhead of the first communication node, and then improving the accuracy of the channel estimation, thereby improving the demodulation performance of the second communication node, and realizing high-spectral-efficiency information transmission; and because M1 is an integer greater than or equal to 1, M2 is an integer greater than or equal to 1, and M3 is an integer greater than or equal to 0, the modulation symbol can be obtained by modulating multiple bits of information in the data packet according to the first constellation model, the second constellation model, or the third constellation model, that is, the modulation symbol can carry multiple bits of information, thereby realizing high-order modulation, and then improving the spectrum efficiency of information transmission. Therefore, the embodiment of the present application can support a large number of first communication nodes to realize high-spectral-efficiency information transmission.
  • the pilot can be called a pilot signal, or a reference signal (RS), or a demodulation reference signal, or a preamble, and in terms of form, the pilot is usually a sequence or a string of symbols, so the pilot is also called a pilot sequence.
  • RS reference signal
  • the pilot is usually a sequence or a string of symbols, so the pilot is also called a pilot sequence.
  • each extremely sparse pilot includes a second number of non-zero value symbols, wherein the second number is greater than 0 and less than 5.
  • the second number of non-zero value symbols are carried on a third number of resource elements (REs) adjacent in the time-frequency domain, or carried on a third number of symbols in chronological order, or carried on a third number of resource elements on adjacent subcarriers in the frequency domain, wherein the third number is equal to the second number (i.e., the third number is greater than 0 and less than 5).
  • REs resource elements
  • each extremely sparse pilot has only one reference signal element (RSE), and the number of non-zero value symbols contained in the reference signal element is the second number.
  • one resource element can carry one symbol.
  • the second number of non-zero value symbols are carried on adjacent REs, it is equivalent to the second number of symbols being carried on adjacent time-frequency resources.
  • the second number of non-zero value symbols are carried on adjacent time-frequency resources, it is equivalent to the second number of non-zero value symbols being carried on adjacent REs.
  • the value of the second number is 4, the second number of non-zero value symbols constitutes a non-zero value symbol group [p1, p2, p3, p4], and the values of [p1, p2, p3, p4] include at least the various situations in Table 1, wherein different serial numbers in Table 1 correspond to different situations.
  • the symbol length of each extremely sparse pilot is greater than 24, that is, the total number of non-zero value symbols and zero value symbols of each extremely sparse pilot is greater than 24, and no specific limitation is made here.
  • the first number of extremely sparse pilots are independent of each other, that is, the first number of extremely sparse pilots are not associated or related, wherein the value of the first number can be greater than or equal to 1, that is, the value of the first number can be 1, 2 or other values, which are not specifically limited here.
  • the technology that contains 2 or more pilots in one transmission and the pilots are not associated or independent of each other is called independent multi-pilot technology, and the multiple independent pilots are called independent multi-pilots.
  • w extremely sparse pilots are included in one transmission, wherein the w extremely sparse pilots are respectively represented as P1, P2, ..., Pw, w can be a positive integer greater than 2, and the data packet contains the information of the w extremely sparse pilots, for example, the data packet contains the index number of the w extremely sparse pilots (that is, the index number of the extremely sparse pilot in the preset pilot set), so that once the data packet of a terminal is successfully decoded, the information of all extremely sparse pilots used by the terminal in this information transmission can be determined, so that the pilot signal can be interfered with.
  • the information transmission between the first communication node and the second communication node can adopt independent multi-pilot technology.
  • independent multi-pilot technology can be used to support more first communication nodes to transmit information.
  • independent multi-pilot technology and extremely sparse pilot technology can be combined, that is, multiple independent extremely sparse pilots are used to further reduce the probability of pilot collision and further increase the number of first communication nodes connected.
  • the base station can demodulate the corresponding first communication node in each round through multiple non-collision (i.e., independent of each other) extremely sparse pilots, and then reconstruct the data packet and extremely sparse pilot of the first communication node, and eliminate the data packet and extremely sparse pilot corresponding to the first communication node from the received signal, and iterate until all decomposable first communication nodes are demodulated to reduce the probability of pilot collision and further increase the number of connected first communication nodes.
  • non-collision i.e., independent of each other
  • the first number of extremely sparse pilots may be determined according to information in the data packet.
  • the first number of extremely sparse pilots can be determined based on one or more bits of information in the data packet.
  • an extremely sparse pilot can be determined based on one bit of information in the data packet; for another example, an extremely sparse pilot can be determined based on two bits of information in the data packet; for another example, both extremely sparse pilots are determined based on multiple bits of information in the data packet, and so on.
  • the embodiment of the present application does not limit the first number and the number of bit information.
  • each extremely sparse pilot is determined from a preset pilot set according to the bit information of the fourth number in the data packet, wherein the preset pilot set includes a fifth number of pilots, and the fourth number is in a logarithmic function relationship with the fifth number, and the logarithmic function is a logarithmic function with 2 as the base.
  • the fifth number is D
  • the fourth number is log 2 (D), which is not specifically limited here. It can be understood that the fifth number can be 64, 128 or more, which is not specifically limited here.
  • n includes 1, 2, ..., N1, that is, a n can be a 1 , a 2 , a 3 , or a N1 , etc.
  • d is a positive real number
  • ⁇ is a real number greater than or equal to 0, so that a 1 , a 2 , ..., a N1 constitutes an arithmetic progression.
  • the value of ⁇ can be At this time, when the value of d is 1, a n satisfies When the value of d is 1/2, so that a n satisfies
  • n includes 1, 2, ..., N2, that is, a n can be a 1 , a 2 , a 3 , or a N2 , etc.
  • b n can be b 1 , b 2 , b 3 , or b N2 , etc.
  • d is a positive real number, ⁇ and ⁇ are both real numbers greater than or equal to 0, so that a 1 , a 2 , ..., a N2 constitute an arithmetic progression, and b 1 , b 2 , ..., b N2 constitute an arithmetic progression.
  • the value of ⁇ can be At this time, when the value of d is 1, a n satisfies When the value of d is 1/2, so that a n satisfies
  • n includes 1, 2, ..., N3, that is, a n can be a 1 , a 2 , a 3 , or a N3 , etc.
  • b n can be b 1 , b 2 , b 3 , or b N3 , etc.
  • d is a positive real number, ⁇ and ⁇ are both real numbers greater than or equal to 0, so that a 1 , a 2 , ..., a N3 constitute an arithmetic progression, and b 1 , b 2 , ..., b N3 constitute an arithmetic progression.
  • the value of ⁇ can be At this time, when the value of d is 1, a n satisfies When the value of d is 1/2, so that a n satisfies
  • ⁇ is equal to 0.
  • ⁇ is greater than 0.
  • the value of d is such that the first constellation model is adopted.
  • the average power of the modulation symbol obtained by modulation is equal to a value of 1, that is, the value of d makes the mean of the square of the modulus of the constellation points in the first constellation diagram model is 1, that is, the value of d makes the average power of the first constellation diagram model is 1, and no specific limitation is made here.
  • a feasible implementation method is that when the modulation symbol is modulated according to the second constellation diagram model, the value of d is a value that makes the average power of the modulation symbol obtained by modulation using the second constellation diagram model equal to 1, that is, the value of d makes the mean of the square of the modulus of the constellation points in the second constellation diagram model is 1, that is, the value of d makes the average power of the second constellation diagram model is 1, and no specific limitation is made here.
  • a feasible implementation method is that when the modulation symbol is modulated according to the third constellation diagram model, the value of d is a value that makes the average power of the modulation symbol obtained by modulation using the third constellation diagram model equal to 1, that is, the value of d makes the mean of the square of the modulus of the constellation points in the third constellation diagram model is 1, that is, the value of d makes the average power of the third constellation diagram model is 1, and no specific limitation is made here.
  • FIG. 4 is an information transmission method provided by another embodiment of the present application.
  • the information transmission method is applied to the second communication node.
  • the information transmission method may include but is not limited to step S210.
  • Step S210 Receive a data packet and a first number of extremely sparse pilots sent by a first communication node.
  • the first number is greater than or equal to 1
  • the data packet includes at least a modulation symbol
  • the modulation symbol is obtained by modulating M1+1 bits of information in the data packet according to a first constellation model
  • the first constellation model includes 2*N1 constellation points
  • M1 is an integer greater than or equal to 1
  • the complex forms corresponding to the 2*N1 constellation points in the first constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N1 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N1 e j( ⁇ + ⁇ ) ;
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N1 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N1 ;
  • the complex forms corresponding to the 4*N2 constellation points in the second constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N2 e j ⁇ , b 1 e j( ⁇ + ⁇ /2) ,b 2 e j( ⁇ + ⁇ /2) ,...,b N2 e j( ⁇ + ⁇ /2) , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N2 e j( ⁇ + ⁇ ) , b 1 e j( ⁇ +3 ⁇ /2) ,b 2 e j( ⁇ +3 ⁇ /2) ,...,b N2 e j( ⁇ +3 ⁇ /2) ;
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 , ..., a N2 and b 1 , b 2 , ..., b N2 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N2 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N2 ;
  • the complex forms corresponding to the 8*N3 constellation points in the third constellation diagram model include the following: a 1 e j ⁇ ,a 2 e j ⁇ ,...,a N3 e j ⁇ , a 1 e j( ⁇ + ⁇ ) ,a 2 e j( ⁇ + ⁇ ) ,...,a N3 e j( ⁇ + ⁇ ) ,
  • ⁇ is the ratio of the circumference of a circle to its circumference; j is an imaginary number; the value of j is equal to e is a natural logarithm; ⁇ is a real number; a 1 , a 2 ,..., a N3 and b 1 , b 2 ,..., b N3 are all positive numbers and satisfy: 0 ⁇ a 1 ⁇ a 2 ⁇ ... ⁇ a N3 , 0 ⁇ b 1 ⁇ b 2 ⁇ ... ⁇ b N3 .
  • the second communication node can receive the data packet and the first number of extremely sparse pilots sent by the first communication node, wherein the first number is greater than or equal to 1, and the data packet at least includes a modulation symbol, and the modulation symbol is obtained by modulating M1+1 bits of information in the data packet according to the first constellation diagram model, or modulating M2+2 bits of information in the data packet according to the second constellation diagram model, or modulating M3+3 bits of information in the data packet according to the third constellation diagram model, and M1 is an integer greater than or equal to 1, M2 is an integer greater than or equal to 1, and M3 is an integer greater than or equal to 0, that is, the second communication node can estimate part of the information of the wireless channel from the extremely sparse pilot sent by the first communication node, and further Channel information is extracted from the modulation symbol without estimating all the information of the wireless channel from the extremely sparse pilot, thereby reducing the pilot overhead of the
  • the pilot can be called a pilot signal, or a reference signal (RS), or a demodulation reference signal, or a preamble, and in terms of form, the pilot is usually a sequence or a string of symbols, so the pilot is also called a pilot sequence.
  • RS reference signal
  • the pilot is usually a sequence or a string of symbols, so the pilot is also called a pilot sequence.
  • each extremely sparse pilot includes a second number of non-zero value symbols, wherein the second number is greater than 0 and less than 5.
  • the second number of non-zero value symbols are carried on a third number of resource elements (REs) adjacent in the time-frequency domain, or carried on a third number of symbols in chronological order, or carried on a third number of resource elements on adjacent subcarriers in the frequency domain, wherein the third number is equal to the second number (i.e., the third number is greater than 0 and less than 5).
  • REs resource elements
  • each extremely sparse pilot has only one reference signal element (RSE), and the number of non-zero value symbols contained in the reference signal element is the second number.
  • one resource element can carry one symbol.
  • the second number of non-zero value symbols are carried on adjacent REs, it is equivalent to the second number of symbols being carried on adjacent time-frequency resources.
  • the second number of non-zero value symbols are carried on adjacent time-frequency resources, it is equivalent to the second number of non-zero value symbols being carried on adjacent REs.
  • the value of the second number is 4, the second number of non-zero value symbols constitutes a non-zero value symbol group [p1, p2, p3, p4], and the values of [p1, p2, p3, p4] include at least the various situations in Table 1, wherein different serial numbers in Table 1 correspond to different situations.
  • the symbol length of each extremely sparse pilot is greater than 24, that is, the total number of non-zero value symbols and zero value symbols of each extremely sparse pilot is greater than 24, and no specific limitation is made here.
  • the first number of extremely sparse pilots are independent of each other, that is, the first number of extremely sparse pilots are not associated or related, wherein the value of the first number can be greater than or equal to 1, that is, the value of the first number can be 1, 2 or other values, which are not specifically limited here.
  • the technology that contains 2 or more pilots in one transmission and the pilots are not associated or independent of each other is called independent multi-pilot technology, and the multiple independent pilots are called independent multi-pilots.
  • w extremely sparse pilots are included in one transmission, wherein the w extremely sparse pilots are respectively represented as P1, P2, ..., Pw, w can be a positive integer greater than 2, and the data packet contains the information of the w extremely sparse pilots, for example, the data packet contains the index number of the w extremely sparse pilots (that is, the index number of the extremely sparse pilot in the preset pilot set), so that once the data packet of a terminal is successfully decoded, the information of all extremely sparse pilots used by the terminal in this information transmission can be determined, so that the pilot signal can be interfered with.
  • the information transmission between the first communication node and the second communication node can adopt independent multi-pilot technology.
  • independent multi-pilot technology can be used to support more first communication nodes for information transmission.
  • independent multi-pilot technology and extremely sparse pilot technology can be combined, that is, multiple independent and extremely sparse pilots are used to further reduce the probability of pilot collision and further increase the number of first communication nodes connected.
  • the base station can demodulate the corresponding first communication node in each round through multiple non-collision (i.e., independent of each other) extremely sparse pilots, and then reconstruct the data packet and extremely sparse pilot of the first communication node, and eliminate the data packet and extremely sparse pilot corresponding to the first communication node from the received signal, and iterate until all decomposable first communication nodes are demodulated to reduce the probability of pilot collision and further increase the number of connected first communication nodes.
  • non-collision i.e., independent of each other
  • the first number of extremely sparse pilots may be determined according to information in the data packet.
  • the first number of extremely sparse pilots can be determined based on one or more bits of information in the data packet.
  • an extremely sparse pilot can be determined based on one bit of information in the data packet; for example, an extremely sparse pilot can be determined based on two bits of information in the data packet; for example, both extremely sparse pilots can be determined based on multiple bits in the data packet.
  • Information determination, etc. the embodiment of the present application does not limit the first quantity and the number of bit information.
  • each extremely sparse pilot is determined from a preset pilot set according to the bit information of the fourth number in the data packet, wherein the preset pilot set includes a fifth number of pilots, and the fourth number is in a logarithmic function relationship with the fifth number, and the logarithmic function is a logarithmic function with 2 as the base.
  • the fifth number is D
  • the fourth number is log 2 (D), which is not specifically limited here. It can be understood that the fifth number can be 64, 128 or more, which is not specifically limited here.
  • n includes 1, 2, ..., N1, that is, a n can be a 1 , a 2 , a 3 , or a N1 , etc.
  • d is a positive real number
  • ⁇ is a real number greater than or equal to 0, so that a 1 , a 2 , ..., a N1 constitutes an arithmetic progression.
  • the value of ⁇ can be At this time, when the value of d is 1, a n satisfies When the value of d is 1/2, so that a n satisfies
  • n includes 1, 2, ..., N2, that is, a n can be a 1 , a 2 , a 3 , or a N2 , etc.
  • b n can be b 1 , b 2 , b 3 , or b N2 , etc.
  • d is a positive real number, ⁇ and ⁇ are both real numbers greater than or equal to 0, so that a 1 , a 2 , ..., a N2 constitute an arithmetic progression, and b 1 , b 2 , ..., b N2 constitute an arithmetic progression.
  • the value of ⁇ can be At this time, when the value of d is 1, a n satisfies When the value of d is 1/2, so that a n satisfies
  • n includes 1, 2, ..., N3, that is, a n can be a 1 , a 2 , a 3 , or a N3 , etc.
  • b n can be b 1 , b 2 , b 3 , or b N3 , etc.
  • d is a positive real number, ⁇ and ⁇ are both real numbers greater than or equal to 0, so that a 1 , a 2 , ..., a N3 constitute an arithmetic progression, and b 1 , b 2 , ..., b N3 constitute an arithmetic progression.
  • the value of ⁇ can be At this time, when the value of d is 1, a n satisfies When the value of d is 1/2, so that a n satisfies
  • ⁇ is equal to 0.
  • ⁇ is greater than 0.
  • a feasible implementation method is that when the modulation symbol is modulated according to the first constellation diagram model, the value of d is a value that makes the average power of the modulation symbol obtained by modulation using the first constellation diagram model equal to 1, that is, the value of d makes the mean of the square of the modulus of the constellation points in the first constellation diagram model is 1, that is, the value of d makes the average power of the first constellation diagram model is 1, and no specific limitation is made here.
  • a feasible implementation method is that when the modulation symbol is modulated according to the second constellation diagram model, the value of d is a value that makes the average power of the modulation symbol obtained by modulation using the second constellation diagram model equal to 1, that is, the value of d makes the mean of the square of the modulus of the constellation points in the second constellation diagram model is 1, that is, the value of d makes the average power of the second constellation diagram model is 1, and no specific limitation is made here.
  • a feasible implementation method is that when the modulation symbol is modulated according to the third constellation diagram model, the value of d is a value that makes the average power of the modulation symbol obtained by modulation using the third constellation diagram model equal to 1, that is, the value of d makes the mean of the square of the modulus of the constellation points in the third constellation diagram model is 1, that is, the value of d makes the average power of the third constellation diagram model is 1, and no specific limitation is made here.
  • FIG. 5 is a schematic diagram of a first constellation diagram model and a second constellation diagram model provided in an embodiment of the present application.
  • the second constellation diagram model may be a cross constellation diagram corresponding to the lower left coordinate system in FIG. 5 or a cross constellation diagram corresponding to the lower right coordinate system.
  • a cross-shaped constellation diagram wherein the cross-shaped constellation diagram is a constellation diagram in which half of the constellation points are located on a straight line passing through the zero point (i.e., the origin), and the other half of the constellation points are located on another straight line passing through the zero point (i.e., the origin), and the two straight lines are perpendicular to each other.
  • the cross-shaped constellation diagram has the advantages of high spectral efficiency and simple geometric shape.
  • the cross-shaped constellation diagram shown in FIG5 is a constellation diagram in a two-dimensional signal plane.
  • These cross-shaped constellations include 16 constellation points, each constellation point corresponds to a modulation symbol, and each modulation symbol can carry 4 bits of information, that is, 4 bits of information will be mapped (i.e., modulated) into a modulation symbol.
  • the constellation points in the cross-shaped constellation diagram in the lower left corner of FIG5 are distributed on the x-axis (i.e., I path) and the y-axis (i.e., Q path); the constellation points in the cross-shaped constellation diagram in the lower right corner of FIG5 are respectively distributed on a straight line in the 45° direction passing through the origin and a straight line in the 135° direction passing through the origin.
  • the cross-shaped constellation diagram in the lower right corner of FIG5 can be formed by rotating the cross-shaped constellation diagram in the lower left corner of FIG5 by 45°.
  • the second constellation model may also be other cross-shaped constellation diagrams besides the cross-shaped constellation diagrams shown in the lower left and lower right corners of FIG. 5 , and the embodiment of the present application does not specifically limit the form of the second constellation model.
  • the first constellation diagram model may be a PAM (Pulse Amplitude Modulation) constellation diagram corresponding to the coordinate system in the upper left corner of FIG5 or a PAM constellation diagram corresponding to the coordinate system in the upper right corner, i.e., a linear constellation diagram, wherein all constellation points of the PAM constellation diagram are on a straight line passing through the zero point (i.e., the origin).
  • the first constellation diagram model may also be other PAM constellation diagrams in addition to the PAM constellation diagrams shown in the upper left and upper right corners of FIG5, and the embodiment of the present application does not specifically limit the form of the first constellation diagram model.
  • each modulation symbol (that is, each constellation point) can carry multiple bits of information, that is, the effect of high-order modulation can be achieved, thereby achieving high spectral efficiency.
  • each modulation symbol can carry 4 bits of information, that is, 4 bits of information will be mapped (that is, modulated) into one modulation symbol; in another embodiment, each modulation symbol can carry 5 bits, that is, 5 bits of information will be mapped (that is, modulated) into one modulation symbol.
  • the linear constellation diagram i.e., the PAM constellation diagram
  • the cross constellation diagram and the 8-arm constellation diagram corresponding to the modulation symbols all have the advantage of simple geometric shapes. Even if the modulation symbols received by the receiving side (i.e., the second communication node) have been rotated and scaled by the channel, the constellation diagram corresponding to the modulation symbols is only a linear constellation diagram, a cross constellation diagram or an 8-arm constellation diagram that has been rotated and scaled, and the resulting geometric shape is still relatively simple.
  • Figure 6 is a schematic diagram of a cross constellation diagram before and after channel rotation and scaling, wherein the coordinate system on the left side of Figure 6 is a schematic diagram of a cross constellation diagram corresponding to the transmitted modulation symbol s (i.e., the modulation symbol s at the first communication node that has not undergone channel rotation and scaling), and the middle coordinate system of Figure 6 is a schematic diagram of a cross constellation diagram corresponding to the modulation symbol h*s (i.e., h multiplied by s, which can also be expressed as h ⁇ s or hs) received by the second communication node after rotation and scaling, wherein the complex number h is the rotation and scaling amount.
  • the transmitted modulation symbol s i.e., the modulation symbol s at the first communication node that has not undergone channel rotation and scaling
  • the middle coordinate system of Figure 6 is a schematic diagram of a cross constellation diagram corresponding to the modulation symbol h*s (i.e., h multiplied by s, which can also be
  • the constellation diagram shown on the right of FIG6 can be formed by adding a complex number corresponding to AWGN to the constellation point in the constellation diagram shown in the middle of FIG6, that is, the constellation point corresponding to the received modulation symbol (h*s+n) with AWGN will be distributed around the constellation point (h*s) corresponding to the constellation diagram shown in the middle of FIG6 according to the probability density of AWGN.
  • the color of the constellation point changes from dark to light from the middle to the edge, and the constellation point is a set of points formed by the corresponding modulation symbol being affected by AWGN.
  • the receiving side ie, the second communication node
  • the geometric shape of the cross-shaped constellation diagram as shown on the right side of FIG. 6 to estimate the amount of rotation and scaling of the constellation diagram, that is, to estimate h.
  • the two-dimensional plane (i.e., the two-dimensional signal plane) is divided into four partitions, and two typical methods can be used for partitioning.
  • the first partitioning method four quadrants are divided into four partitions, that is, the x-axis and the y-axis are used as partition lines, where the area filled with oblique lines is partition 1, the area filled with fine dots is partition 2, the area filled with vertical lines is partition 3, and the area filled with bricks is partition 4;
  • the four partitions in the second partitioning method are formed by rotating the four partitions in the first partitioning method by 45°, that is, the 45° ray emitted from the origin to the 45° ray emitted from the origin is the 45° ray emitted from the origin.
  • the area enclosed by the 135° ray emitted from the origin is partition 1, where partition 1 is filled with oblique lines; the area from the 135° ray emitted from the origin to the 225° ray emitted from the origin is partition 2, where partition 2 is filled with fine dots; the area from the 225° ray emitted from the origin to the 315° ray emitted from the origin is partition 3, where partition 3 is filled with vertical lines; the area from the 315° ray emitted from the origin to the 45° ray emitted from the origin is partition 4, where partition 4 is filled with bricks.
  • the two partitioning methods shown in Figure 7 are used to determine the partition to which a constellation point belongs.
  • the constellation points in each partition i.e., the modulation symbols corresponding to each constellation point
  • the number of constellation points in the partition i.e., the number of modulation symbols
  • a coordinate can be calculated, which is the center of the constellation point in the partition.
  • the cross-shaped constellation diagram shown in the middle of Figure 6 is a diagram of the center of the constellation point in Figure 6.
  • 6 is a schematic diagram of a constellation diagram formed by rotating and scaling the cross-shaped constellation diagram shown on the left side of FIG. 7. Taking the partition shown in the left coordinate system in FIG.
  • the constellation points in partition 3 are added up and then divided by the number of constellation points in the partition, and the constellation point center c3 of partition 3 can be obtained, that is, the position of the five-pointed star shown in the right coordinate system of FIG. 8;
  • the constellation points in partition 4 are added up and then divided by the number of constellation points in the partition, and the constellation point center c4 of partition 4 can be obtained, that is, the position of the six-pointed star shown in the right coordinate system of FIG. 8.
  • the rotation scaling amount of the entire constellation diagram can be obtained according to the constellation point centers of all partitions.
  • the two partitioning methods shown in FIG. 7 are usually required.
  • two rotation and scaling amounts of the constellation diagram are calculated according to the above estimation method respectively, and then the larger modulus value of the two rotation and scaling amounts is used as the rotation and scaling amount of the constellation diagram.
  • the constellation points in each partition i.e., the modulation symbols corresponding to each constellation point
  • the number of constellation points in the partition i.e., the number of modulation symbols.
  • a constellation point can be calculated, i.e., the constellation point center of the partition.
  • the rotation scaling of the entire constellation diagram is obtained through the constellation point centers of all partitions.
  • the following four different 2-partitioning methods can be used for partitioning.
  • the x-axis is used as the partition line to divide the two-dimensional signal plane into two partitions;
  • the y-axis is used as the partition line to divide the two-dimensional signal plane into two partitions;
  • the 45° straight line passing through the origin is used as the partition line to divide the two-dimensional signal plane into two partitions;
  • the 135° straight line passing through the origin is used as the partition line to divide the two-dimensional signal plane into two partitions.
  • the multiple modulation symbols obtained through the first constellation model, the second constellation model or the third constellation model can form a constellation with a simple geometric shape, and after the modulation symbols are distorted by rotation and scaling due to channel interference, the formed constellation still presents a simple geometric shape. Therefore, the information transmission method of the present application can compensate only through the shape characteristics of the constellation, thereby eliminating the need to increase pilot overhead to improve demodulation performance and ensure high-frequency spectrum efficiency.
  • the cross-shaped constellation diagram in FIG6 can be divided into two parts, and the constellation points of each part are on a straight line passing through the zero point (i.e., the origin).
  • the zero point i.e., the origin
  • the cross-shaped constellation diagram corresponding to the coordinate system on the left side of FIG6 half of the constellation points fall on the straight line of the x-axis, and the other half of the constellation points fall on the straight line of the y-axis.
  • the distance between adjacent constellation points on a straight line passing through the origin are equal, and the distance between two adjacent points is set to 2d, then among the four constellation points closest to the origin, the distance between adjacent constellation points is only That is to say, among the four constellation points closest to the origin, the distance between adjacent constellation points will be smaller than the distance between adjacent constellation points on the same straight line, that is, the four constellation points closest to the origin are more densely distributed. Therefore, the constellation diagram is more susceptible to AWGN interference, which will lead to a decrease in demodulation performance.
  • the cross-shaped constellation diagram can be divided into four parts.
  • the cross-shaped constellation diagram corresponding to the coordinate system on the left side of FIG. 6 can be divided into four parts according to the partition form shown in the coordinate system on the right side of FIG. 7 .
  • the constellation points of the first part are constellation points greater than 0 on the x-axis, that is, constellation points falling on the positive half axis of the x-axis, wherein the first part corresponds to partition 4 of the right coordinate system in Figure 7;
  • the constellation points of the second part are constellation points less than 0 on the x-axis, that is, constellation points falling on the negative half axis of the x-axis, wherein the second part corresponds to partition 2 of the right coordinate system in Figure 7;
  • the constellation points of the third part are constellation points greater than 0 on the y-axis, that is, constellation points falling on the positive half axis of the y-axis, wherein the third part corresponds to partition 1 of the right coordinate system in Figure 7;
  • the constellation points of the fourth part are constellation points less than 0 on the y-axis, that is, constellation points falling on the negative half axis of the y-axis, wherein the fourth part corresponds to partition 3 of the right coordinate system in Figure 7.
  • an offset ⁇ greater than 0 may be added to the constellation points of the 4 parts of the cross constellation diagram respectively, so that among the 4 constellation points closest to the origin, the distance between adjacent constellation points is greater than or equal to the distance between adjacent constellation points on the same straight line, that is, the constellation points of the 4 parts are all offset in a direction away from the origin, so as to avoid dense distribution of the 4 constellation points closest to the origin, so as to improve the demodulation performance.
  • the constellation points of the four parts of the cross constellation diagram may not be superimposed with an offset ⁇ greater than 0, that is, the offset of the constellation points of each part of the cross constellation diagram is 0, and in the cross constellation diagram in which the offsets of the constellation points of all parts are 0, the distances between adjacent constellation points on the same straight line are equal, so the average power of the constellation diagram is lower.
  • the value of ⁇ can be at this time, Among the 4 constellation points closest to the origin, the distance between adjacent constellation points is 2d, and the distance between adjacent constellation points on the same straight line is also 2d, that is, among the 4 constellation points closest to the origin, the distance between adjacent constellation points is equal to the distance between adjacent constellation points on the same straight line in each part.
  • the distance between adjacent constellation points in the four constellation points closest to the origin can be increased, so that the formed cross constellation diagram expands outward, avoiding the four constellation points closest to the origin from being too densely distributed, thereby reducing the impact of AWGN on the constellation points (i.e., modulation symbols) and improving the robustness of the cross constellation diagram.
  • an can be set to nd, d is a positive real number, so that in the four constellation points closest to the origin, the distance between adjacent constellation points is The distance between adjacent constellation points on the same straight line of each part is d, that is, among the four constellation points closest to the origin, the distance between adjacent constellation points is greater than the distance between adjacent constellation points on the same straight line of each part, that is, the constellation points of the four parts are all offset away from the origin to reduce the impact of AWGN on the constellation points (i.e., modulation symbols), thereby improving the demodulation performance of the second communication node.
  • an offset ⁇ greater than 0 is added to the constellation points of the four parts of the cross constellation diagram,
  • the second constellation model is a cross constellation
  • the distance between the constellation point a 1 e j0 (i.e., s1) and the constellation point b 1 e j(0+ ⁇ /2) (i.e., s3) is 2, i.e.
  • the distance between constellation point a 1 e j0 (i.e. s1) and constellation point a 2 e j0 i.e.
  • the coordinates of constellation point s1 are The coordinates of constellation point s2 are The coordinates of constellation point s3 are It is understandable that the two-dimensional coordinates of other constellation points can be calculated using the complex form corresponding to the constellation points in the second constellation diagram model, which will not be described in detail here.
  • the two-dimensional coordinates of the eight constellation points may include the following: (1+ ⁇ ,0),(3+ ⁇ ,0), (0,1+ ⁇ ),(0,3+ ⁇ ), (-(1+ ⁇ ),0),(-(3+ ⁇ ),0), (0,-(1+ ⁇ )),(0,-(3+ ⁇ ))
  • ⁇ can be expressed by a finite decimal, for example, No specific limitation is imposed here.
  • the two-dimensional coordinates of the eight constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor.
  • ⁇ satisfies the formula
  • the second constellation model is a cross constellation
  • the value of M2 1
  • the second constellation model is a cross constellation including 8 constellation points, wherein the value of d is 1, that is, the scaling amount of the cross constellation is 1; the value of ⁇ is ⁇ /4, that is, the rotation amount of the cross constellation is ⁇ /4, that is, half of the constellation points in the cross constellation fall on a straight line passing through the zero point (i.e., the origin) at an angle of 45° to the positive semi-axis of the x-axis, and the other half of the constellation points fall on a straight line passing through the zero point (i.e., the origin) at an angle of 135° to the positive semi-axis of the x-axis.
  • the coordinates of the constellation point s1 corresponding to a 1 e j ⁇ /4 are (1,1)
  • the coordinates of the constellation point s2 corresponding to a 2 e j ⁇ /4 are It is understandable that the two-dimensional coordinates of other constellation points can be calculated using the complex form corresponding to the constellation points in the second constellation diagram model, which will not be described in detail here.
  • ⁇ can also satisfy the formula
  • the two-dimensional coordinates of the eight constellation points may include the following:
  • the two-dimensional coordinates of the eight constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor.
  • the coordinates of the constellation point s1 can be determined to be
  • the coordinates of constellation point s2 are
  • the two-dimensional coordinates of other constellation points can be calculated using the complex form corresponding to the constellation points in the second constellation diagram model, which will not be described in detail here.
  • the two-dimensional coordinates of the 16 constellation points may include the following: (1+ ⁇ ,0),(3+ ⁇ ,0),(5+ ⁇ ,0),(7+ ⁇ ,0), (-(1+ ⁇ ),0),(-(3+ ⁇ ),0),(-(5+ ⁇ ),0),(-(7+ ⁇ ),0), (0,1+ ⁇ ),(0,3+ ⁇ ),(0,5+ ⁇ ),(0,7+ ⁇ ), (0,-(1+ ⁇ )),(0,-(3+ ⁇ )),(0,-(5+ ⁇ )),(0,-(7+ ⁇ ))
  • ⁇ can be expressed by a finite decimal, for example, No specific limitation is imposed here.
  • the two-dimensional coordinates of the 16 constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor.
  • the second constellation model is a cross constellation
  • the second constellation model is a cross constellation including 16 constellation points, wherein the value of d is 1, that is, the scaling amount of the cross constellation is 1; the value of ⁇ is ⁇ /4, that is, the rotation amount of the cross constellation is ⁇ /4, that is, the rotation amount of the cross constellation is ⁇ /4, that is, half of the constellation points in the cross constellation fall on a straight line that passes through the zero point (i.e., the origin) at an angle of 45° to the positive semi-axis of the x-axis, and the other half of the constellation points fall on a straight line that passes through the zero point (i.e., the origin) at an angle of 135° to the positive semi-axis of the x-axis.
  • ⁇ can also satisfy the formula
  • the two-dimensional coordinates of the 16 constellation points may include the following:
  • the distance between a 1 e j ⁇ /4 and a 2 e j ⁇ /4 is 2, that is, the distance between adjacent constellation points on the same straight line is 2.
  • the distance between a 1 e j ⁇ /4 and a 1 e j( ⁇ /4+ ⁇ /2) is 2, that is, the distance between the adjacent points of the 4 constellation points closest to the origin is 2. Therefore, the distance between the adjacent points of the 4 constellation points closest to the origin is equal to the distance between adjacent constellation points on the same straight line.
  • the two-dimensional coordinates of the 16 constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor.
  • the second constellation model is a cross constellation
  • the second constellation model is a cross constellation including 8 constellation points, wherein the value of d is 1, that is, the scaling amount of the cross constellation is 1; the value of ⁇ is 0, that is, the rotation amount of the cross constellation is 0, that is, the constellation points in the cross constellation are distributed on the x-axis (i.e., I path) and the y-axis (i.e., Q path).
  • the two-dimensional coordinates of the eight constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor (eg, 1/sqrt(10)).
  • a normalization factor eg, 1/sqrt(10)
  • the second constellation model is a cross constellation
  • the scaling amount of the cross constellation diagram is 1;
  • the value of ⁇ is ⁇ /4, that is, the rotation amount of the cross constellation diagram is ⁇ /4, that is, half of the constellation points in the cross constellation diagram fall on the straight line passing through the zero point (i.e., the origin) at an angle of 45° to the positive semi-axis of the x-axis, and the other half of the constellation points fall on the straight line passing through the zero point (i.e., the origin) at an angle of 135° to the positive semi-axis of the x-axis.
  • the two-dimensional coordinates of the eight constellation points may include the following:
  • the two-dimensional coordinates of the eight constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor (eg, 1/sqrt(10)).
  • a normalization factor eg, 1/sqrt(10)
  • the second constellation model is a cross constellation
  • the two-dimensional coordinates of the 16 constellation points may include the following: (1,0),(3,0),(5,0),(7,0), (-1,0),(-3,0),(-5,0),(-7,0), (0,1),(0,3),(0,5),(0,7), (0,-1),(0,-3),(0,-5),(0,-7)
  • the two-dimensional coordinates of the 16 constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor (eg, 1/sqrt(84)).
  • a normalization factor eg, 1/sqrt(84)
  • the second constellation model is a cross constellation
  • the second constellation model is a cross constellation including 16 constellation points, wherein the value of d is 1, that is, the scaling amount of the cross constellation is 1; the value of ⁇ is ⁇ /4, that is, the rotation amount of the cross constellation is ⁇ /4, that is, the rotation amount of the cross constellation is ⁇ /4, that is, half of the constellation points in the cross constellation fall on a straight line passing through the zero point (i.e., the origin) at an angle of 45° to the positive semi-axis of the x-axis, and the other half of the constellation points fall on a straight line passing through the zero point (i.e., the origin) at an angle of 135° to the positive semi-axis of the x-axis.
  • the two-dimensional coordinates of the 16 constellation points may include the following:
  • the two-dimensional coordinates of the 16 constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor (eg, 1/sqrt(84)).
  • a normalization factor eg, 1/sqrt(84)
  • the second constellation diagram model is a cross constellation diagram including 8 constellation points, wherein the value of d is 1, that is, the scaling amount of the cross constellation diagram is 1; the value of ⁇ is 0, that is, the rotation amount of the cross constellation diagram is 0, that is, the constellation points in the cross constellation diagram are distributed on the x-axis (i.e., I path) and the y-axis (i.e., Q path).
  • the coordinates of the constellation point s1 corresponding to a 1 e j0 are (1,0), and the coordinates of the constellation point s2 corresponding to a 2 e j0 are (2,0). It can be understood that the two-dimensional coordinates of other constellation points can be calculated using the complex form corresponding to the constellation points in the second constellation diagram model, which will not be repeated here.
  • the two-dimensional coordinates of the eight constellation points may include the following: (1,0),(2,0), (0,1),(0,2), (-1,0),(-2,0), (0,-1),(0,-2)
  • the two-dimensional coordinates of the eight constellation points corresponding to the second constellation model may be uniformly multiplied by a normalization factor (eg, 1/sqrt(5)).
  • the second constellation model is a cross constellation
  • the second constellation model is a cross constellation including 8 constellation points, wherein the value of d is 1, that is, the scaling amount of the cross constellation is 1; the value of ⁇ is ⁇ /4, that is, the rotation amount of the cross constellation is ⁇ /4, that is, half of the constellation points in the cross constellation fall on a straight line that passes through the zero point (i.e., the origin) at an angle of 45° to the positive semi-axis of the x-axis, and the other half of the constellation points fall on a straight line that passes through the zero point (i.e., the origin) at an angle of 135° to the positive semi-axis of the x-axis.
  • the second constellation diagram model may be a cross constellation diagram including 8 constellation points as shown in FIG18 , that is, the constellation diagram corresponding to ⁇ taking a value of ⁇ /4 may be obtained by rotating the constellation diagram corresponding to ⁇ taking a value of zero by 45°.
  • the two-dimensional coordinates of the 8 constellation points may include the following:
  • the first constellation model is a PAM constellation
  • the two-dimensional coordinates of the eight constellation points may include the following: (1,0),(3,0),(5,0),(7,0), (-1,0),(-3,0),(-5,0),(-7,0)
  • the two-dimensional coordinates of the eight constellation points corresponding to the first constellation model may be uniformly multiplied by a normalization factor (eg, 1/sqrt(84)).
  • a normalization factor eg, 1/sqrt(84)
  • the first constellation model is a PAM constellation
  • the two-dimensional coordinates of the four constellation points may include the following: (1,0),(2,0), (-1,0),(-2,0)
  • the two-dimensional coordinates of the four constellation points corresponding to the first constellation model may be uniformly multiplied by a normalization factor (eg, 1/sqrt(5)).
  • the coordinates corresponding to each constellation point in the second constellation diagram model may include the following: (a 1 cos ⁇ ,a 1 sin ⁇ ),(a 2 cos ⁇ ,a 2 sin ⁇ ),...,(a n cos ⁇ ,a n sin ⁇ ), (a 1 cos( ⁇ + ⁇ ),a 1 sin( ⁇ + ⁇ )),(a 2 cos( ⁇ + ⁇ ),a 2 sin( ⁇ + ⁇ )),...,(a n cos( ⁇ + ⁇ ),a n sin( ⁇ + ⁇ )), (a 1 cos( ⁇ +3 ⁇ /2),a 1 sin( ⁇ +3 ⁇ /2)),(a 2 cos( ⁇ +3 ⁇ /2),a 2 sin( ⁇ +3 ⁇ /2)),...,(a n cos( ⁇ +3 ⁇ /2),a n sin( ⁇ +3 ⁇ /2))
  • the coordinates corresponding to each constellation point in the second constellation diagram model can also be expressed as follows: (a 1 cos ⁇ ,a 1 sin ⁇ ),(a 2 cos ⁇ ,a 2 sin ⁇ ),...,(a n cos ⁇ ,a n sin ⁇ ) (-a 1 sin ⁇ ,a 1 cos ⁇ ),(-a 2 sin ⁇ ,a 2 cos ⁇ ),...,(-a n sin ⁇ ,a n cos ⁇ ) (-a 1 cos ⁇ ,-a 1 sin ⁇ ),(-a 2 cos ⁇ ,-a 2 sin ⁇ ),...,(-a n cos ⁇ ,-a n sin ⁇ ) (a 1 sin ⁇ ,-a 1 cos ⁇ ),(a 2 sin ⁇ ,-a 2 cos ⁇ ),...,(a n sin ⁇ ,-a n cos ⁇ )
  • the coordinates corresponding to each constellation point in the second constellation diagram model may include the following: (a 1 ,0),(a 2 ,0),...,(a n ,0), (0,a 1 ),(0,a 2 ),...,(0,a n ), (-a 1 ,0),(-a 2 ,0),...,(-a n ,0), (0,-a 1 ),(0,-a 2 ),...,(0,-a n )
  • the coordinates corresponding to each constellation point in the second constellation diagram model may include the following:
  • FIG. 21 is a schematic diagram of a third constellation diagram model provided by an embodiment of the present application, wherein the third constellation diagram model can be understood as an 8-arm constellation diagram, and the constellation points in the 8-arm constellation diagram corresponding to the coordinate system on the left side of FIG.
  • the 8-arm constellation diagram corresponding to the coordinate system on the right side of FIG. 21 can be formed by expanding the amplitude of the constellation points of the 4 arms (the rays passing through the 45° direction, 135° direction, 225° direction, and 315° direction of the origin) in the 8-arm constellation diagram corresponding to the middle coordinate system of FIG. 21 as a whole, so that the constellation points of the star constellation diagram can be more evenly distributed to improve the demodulation performance.
  • the constellation points in the 8-arm constellation diagram corresponding to the coordinate system on the left side of Figure 22 are respectively distributed on the straight line (including the ray of the positive half axis and the ray of the negative half axis) passing through the x-axis (i.e., I path), the straight line (including the ray of the positive half axis and the ray of the negative half axis) passing through the y-axis (i.e., Q path), the ray in the 45° direction passing through the origin, the ray in the 135° direction passing through the origin, the ray in the 225° direction passing through the origin, and the ray in the 315° direction passing through the origin.
  • the 8-arm constellation diagram corresponding to the coordinate system on the right side of Figure 22 can be formed by expanding the amplitudes of the constellation points of 4 arms (rays passing through the origin in the 45° direction, 135° direction, 225° direction, and 315° direction) in the 8-arm constellation diagram corresponding to the coordinate system on the left side of Figure 22 as a whole, which can make the constellation points of the star-shaped constellation diagram more evenly distributed to improve the demodulation performance.
  • the information transmission method provided in the above embodiment is exemplarily described below by taking extremely sparse pilot as an example.
  • a demodulation reference signal (DMRS) set may be defined, wherein the set includes 12 reference signals.
  • the demodulation reference signal may also be referred to as a demodulation reference signal port (DMRS ports), that is, a set including 12 demodulation reference signal ports (DMRS ports) may be defined.
  • FIG. 23 is a schematic diagram of defining a physical resource block provided by one embodiment.
  • a physical resource block can be defined, which contains 14 orthogonal frequency division multiplexing (or, Discrete Fourier Transform-Spread-Orthogonal Frequency Division Multiplexing (DFT-S-OFDM) or, Single-carrier Frequency-Division Multiple Access (SC-FDMA)) symbols in the time domain, and 12 subcarriers in the frequency domain.
  • DFT-S-OFDM Discrete Fourier Transform-Spread-Orthogonal Frequency Division Multiplexing
  • SC-FDMA Single-carrier Frequency-Division Multiple Access
  • the resource overhead occupied by the demodulation reference signal (DMRS) is 1/7.
  • the area except the reference signal (DMRS) area is the modulation symbol area.
  • FIG. 24 is a schematic diagram of defining a demodulation reference signal provided by one embodiment, 12 demodulation reference signals (DMRS) can be divided into three groups from the perspective of the occupied resource unit (RE) position, the non-zero value symbols (or non-zero signals, useful signals, etc.) of the first group of demodulation reference signals (DMRS) (i.e., the symbols are non-zero values) are carried on the first resource unit (RE), and the four DMRS ports can be distinguished by the OCC code; the non-zero value symbols of the second group of demodulation reference signals (DMRS) are carried on the second resource unit (RE), and the four DMRS ports can be distinguished by the OCC code; the non-zero value symbols of the third group of demodulation reference signals (DMRS) are carried on the third resource unit (RE), and the four DMRS ports can be distinguished by the OCC code.
  • DMRS demodulation reference signals
  • each group of demodulation reference signals takes a value of 0 (i.e., no signal) on the blank unfilled resource unit (RE).
  • RE blank unfilled resource unit
  • FIG25 is a schematic diagram of distinguishing different reference signal ports by OCC codes provided by an embodiment.
  • the 4 demodulation reference signal ports that is, the group of DMRS ports
  • the 4 demodulation reference signal ports can be separated by jointly using two long OCC codes [1,1], [1,-1] in the time domain and two long OCC codes [1,1], [1,-1] in the frequency domain, and different DMRS ports are generated by carrying different OCC codes on the first resource unit.
  • the situation of the 4 reference signals in the second group and the 4 reference signals in the third group in FIG24 is similar. Therefore, a total of 12 demodulation reference signals of the defined demodulation reference signal (DMRS) set, that is, 12 demodulation reference signal ports, can be obtained.
  • DMRS defined demodulation reference signal
  • Figure 26 is a schematic diagram of a demodulation reference signal defined by an OCC code provided by an embodiment.
  • the four demodulation reference signal ports can be separated by jointly using two long OCC codes [1,1], [1,-1] in the time domain and two long OCC codes [1,1], [1,-1] in the frequency domain.
  • the numbers in the shaded parts of the first to third columns are all 1 from top to bottom
  • the numbers in the shaded parts of the fourth column are all -1 from top to bottom
  • the numbers in the shaded parts of the fifth to seventh columns are 1, -1, 1, -1 from top to bottom, respectively
  • the numbers in the shaded parts of the eighth column are -1, 1, -1, 1 from top to bottom, respectively.
  • FIG27 is a schematic diagram of another method of defining a demodulation reference signal using an OCC code provided by one embodiment.
  • the four demodulation reference signal ports can be separated by jointly using two long OCC codes [1,1], [1,-1] in the time domain and two long OCC codes [1,1], [1,-1] in the frequency domain, wherein in FIG27, the numbers of the shaded parts of the first to third columns are all 1 from top to bottom, the numbers of the shaded parts of the fourth column are all -1 from top to bottom, the numbers of the shaded parts of the fifth to seventh columns are 1, -1, 1, -1 from top to bottom, respectively, and the numbers of the shaded parts of the eighth column are -1, 1, -1, 1 from top to bottom, respectively.
  • FIG28 is a schematic diagram of another demodulation reference signal using an OCC code provided by one embodiment.
  • 8 represents the first reference signal unit
  • 9 represents the second reference signal unit;
  • the four demodulation reference signal ports can be separated by jointly using two long OCC codes [1,1], [1,-1] in the time domain and two long OCC codes [1,1], [1,-1] in the frequency domain, wherein in FIG28, the numbers of the shaded parts of the first to third columns are all 1 from top to bottom, the numbers of the shaded parts of the fourth column are all -1 from top to bottom, the numbers of the shaded parts of the fifth to seventh columns are 1, -1, 1, -1 from top to bottom, respectively, and the numbers of the shaded parts of the eighth column are -1, 1, -1, 1 from top to bottom, respectively.
  • the reference signal carried on several resource elements (RE) adjacent in the time domain and frequency domain may be referred to as a reference signal element (RSE).
  • RSE reference signal element
  • the reference signal carried on four consecutive resource elements (RE) in the time and frequency domain is referred to as a reference signal element (RSE).
  • RSE reference signal element
  • the reference signal on the "grid" filled with a pattern is a reference signal element (RSE).
  • RSE the reference signal on the "grid" filled with a pattern
  • a reference signal element (RSE) has another characteristic, that is, when the reference signal is applied for channel estimation, each reference signal element (RSE) can estimate a channel value.
  • each reference signal has 2 reference signal elements (RSE) within 1 PRB bandwidth, so a channel value at 2 within 1 PRB bandwidth (that is, within 12 subcarrier bandwidth) can be estimated.
  • RSE reference signal elements
  • the channel values of 12 subcarriers within 1 PRB bandwidth can be interpolated through these reference signal element estimation values get.
  • FIG29 is a schematic diagram of a PRB-based transmission provided by one embodiment. If one transmission includes X PRBs, the 12 demodulation reference signals (or 12 demodulation reference signal ports) in the reference signal set are shown in FIG29, and each reference signal has 2 ⁇ X reference signal elements (RSEs), so the channel values at 2 ⁇ X equally spaced locations within the entire transmission bandwidth can be estimated, and then the channel values of all 12 ⁇ X subcarriers within the transmission bandwidth can be obtained by interpolation.
  • RSEs reference signal elements
  • 12 demodulation reference signals can be divided into three groups from the perspective of the occupied resource unit (RE) position.
  • Figure 30 is a schematic diagram of another method for defining demodulation reference signals provided by an embodiment, in which four demodulation reference signals of the first group can be distinguished by OCC codes
  • Figure 31 is a schematic diagram of another method for defining demodulation reference signals provided by an embodiment, in which four demodulation reference signals of the second group can be distinguished by OCC codes
  • Figure 32 is a schematic diagram of another method for defining demodulation reference signals provided by an embodiment, in which four demodulation reference signals of the third group can be distinguished by OCC codes.
  • the demodulation reference signal occupies a large amount of resources.
  • the distribution of the demodulation reference signal in the entire transmission bandwidth has a certain density and cannot be too sparse.
  • the density of the demodulation reference signal in the entire transmission bandwidth is 2 reference signals per PRB, or 2 reference signal elements (RSE) per PRB.
  • the reference signal overhead is 1/7, that is, the system pays 1/7 of the resources and can only design 12 demodulation reference signals.
  • the collision probability of the reference signals independently selected by any two terminals is 1/12, which shows that the collision probability is very high. Therefore, the demodulation reference signal will severely limit the number of terminals that transmit information in a connectionless state.
  • 12 demodulation reference signals can only support 12 terminals. It can be seen that the demodulation reference signal will severely limit the number of terminals that transmit information in an SPS scenario.
  • the reference signal also needs to estimate a certain frequency offset (Frequency Offset)
  • the resources occupied by each reference signal will continue to increase.
  • the density of each reference signal in the transmission signal will continue to increase.
  • the system pays 2/7 of the resources and can only design 12 demodulation reference signals.
  • Timing Offset the timing offset
  • the system pays 3/7 or even 4/7 of the overhead to design 12 demodulation reference signals.
  • a large overhead only a small number of reference signals (i.e., reference signal ports) can be obtained. It can be seen that the collision probability of the reference signals transmitted in the disconnected state is very high, and the number of terminals transmitting information in the SPS scenario will also be limited.
  • each group of demodulation reference signals has 3 reference signal units in each PRB bandwidth (such as 3 grids in a figurative way). Therefore, there can be 3 estimated values on each PRB, and there can be 3 ⁇ X estimated values on X PRBs. Then, the channel of all subcarriers of X PRBs can be obtained by linear interpolation.
  • This reference signal also occupies 1/7 of the transmission resource overhead, but only 8 demodulation reference signals (8 demodulation reference signal ports) can be separated, which is less than the number of demodulation reference signals defined above. It can be seen that the channel estimation capability is usually inversely proportional to the number of reference signals.
  • the problem faced by reference signals applied to connectionless transmission scenarios and SPS-based information transmission scenarios is that the reference signals must estimate the frequency-selective channel and time-frequency offset of the entire transmission channel and identify the terminal device. Therefore, the time-frequency resources occupied by the reference signals increase exponentially, which leads to a serious shortage of reference signals under certain resources, and in turn affects the number of terminals that can transmit information in connectionless transmission scenarios and SPS-based information transmission scenarios.
  • the main starting point of this application is to greatly reduce the task of reference signals, so that the resources occupied by each reference signal can be minimized, that is, the density of each reference signal in the transmission signal is made the sparsest, and then the number of reference signals can be maximized, and finally the number of terminals for information transmission in connectionless transmission scenarios and SPS-based information transmission scenarios can be increased.
  • the present application estimates the channel of the entire transmission bandwidth and the estimated time-frequency offset through data-based channel estimation technology (rather than reference signal-based) through the characteristics of the data itself, such as the geometric characteristics of the constellation diagram of the modulation symbol. That is to say, there is no need to estimate the channel and time-frequency offset of the entire transmission bandwidth through a reference signal.
  • data-based channel estimation technology rather than reference signal-based
  • the four partitioning methods in the above embodiment can be used to estimate the rotation and scaling of the constellation diagram, that is, the block flat fading channel can be estimated through the four partitioning methods in the above embodiment, and no further details will be given here.
  • the task of the reference signal is much smaller than that of the related scheme, so the resources occupied by each reference signal in the embodiment of the present application are less than the resources occupied by each reference signal in the related scheme. Therefore, under a certain overhead, the number of reference signals in the present application is more than the number of reference signals in the related scheme.
  • the R receiving antennas can provide very powerful spatial capabilities, thereby improving the performance of multi-terminal access.
  • the present application does not use the reference signal to estimate the channel within the entire transmission bandwidth, nor does it use it to estimate the time-frequency offset.
  • FIG. 33 is a schematic diagram of defining a reference signal, assuming that the reference signal occupies one OFDM symbol, and one transmission includes X PRBs of time-frequency resources, and one physical resource block (PRB) includes 14 OFDM (or DFT-S-OFDM or SC-FDMA) symbols in the time domain, and 12 subcarriers in the frequency domain.
  • the first OFDM symbol is used to carry the demodulation reference signal, that is, the first OFDM symbol is used as an extremely sparse pilot area, then 1/14 of the resources are used to transmit the reference signal, that is, 12 ⁇ X resource units (REs) are used to transmit the reference signal.
  • the area except the extremely sparse pilot area is the modulation symbol area.
  • FIG. 34 is a schematic diagram of another system-defined reference signal (i.e., extremely sparse pilot), each of which has a non-zero value symbol (non-zero signal, or useful signal) only on one RE resource unit, and no signal (i.e., the value is 0) on the remaining resource units, therefore, it can occupy a reference signal area of 1/14 overhead, and a total of 12 ⁇ X reference signals can be separated.
  • a system-defined reference signal i.e., extremely sparse pilot
  • FIG. 35 is a schematic diagram of another definition of a reference signal (i.e., an extremely sparse pilot port), each reference signal defined by the system has non-zero value symbols (or non-zero signals, or useful signals) only on 2 RE resource units, and 2 reference signals can be separated from each 2 RE through a 2-length OCC, and there is no signal on the remaining resource units (i.e., the value is 0), so it can occupy a reference signal area of 1/14 overhead, and a total of 12 ⁇ X reference signals can be separated.
  • a reference signal i.e., an extremely sparse pilot port
  • the resource overhead of each PRB is 1/14 overhead
  • Figure 36 is a schematic diagram of generating a DMRS port provided by an embodiment, by carrying different OCC codes (such as OCC code 1 and OCC code 2) on 2 REs to generate different DMRS ports, wherein, in Figure 36, RE is represented by two small squares filled with vertical lines.
  • OCC codes such as OCC code 1 and OCC code 2
  • FIG. 37 is a schematic diagram of another method for defining a reference signal provided by one embodiment, wherein the reference signal defined by the system occupies 2 OFDM symbols.
  • FIG. 38 is a schematic diagram of another definition of reference signals provided by one embodiment, each reference signal defined by the system has a non-zero value symbol (or a non-zero signal, or a useful signal) only on a group of adjacent 4 RE resource units, but there will be 4 reference signals multiplexing a group of adjacent 4 REs, wherein the 4 reference signals multiplexing the same group of adjacent 4 REs can be distinguished by the OCC code, therefore, when a transmission includes X PRBs, and each PRB occupies 1/7 of the reference signal area of the overhead, a total of 24 ⁇ X reference signals can be separated.
  • the number of reference signals is much larger than the number of reference signals in the NR system (the number of reference signals in the NR system is 8 or 12), and at the same time, it is explained that the number of extremely sparse pilots is proportional to the number of PRBs.
  • the extremely sparse pilot is a pilot with very few non-zero elements (i.e., non-zero value symbols) in the preset pilot set, for example, there are only 1-4 non-zero elements. Therefore, the solution for extremely sparse pilots can significantly increase the number of pilots without increasing the pilot overhead, thereby significantly reducing the probability of pilot collision.
  • the base station can estimate part of the information of the wireless channel from the extremely sparse pilot without estimating all the information of the wireless channel from the extremely sparse pilot, and the base station can further extract channel information from the modulation symbol, and then use the channel information to complete the equalization of the modulation symbol.
  • the extremely sparse pilot is only used for spatial merging but cannot be used for channel equalization. Therefore, the extremely sparse pilot can be considered as a spatial merging reference signal and is not specifically limited here.
  • the extremely sparse reference signals i.e., extremely sparse pilots
  • Figures 33, 34, 35 and 36 are all located at the first symbol, or the first symbol and the second symbol of the transmission resource
  • the present application does not limit the position of the extremely sparse reference signal.
  • the position of the extremely sparse reference signal may also be located in the middle of the transmission resource.
  • the information transmission in all the above embodiments is information in a broad sense, that is, the information can be business data or information used for system control, that is, signaling; or, the information can include bit data that needs to be transmitted, such as business bit data or signaling bit data, wherein different English expressions such as message, information, payload, etc. can all represent information.
  • the first communication node in all the above embodiments can be a terminal, for example, a mobile phone, a smart phone, a laptop computer, a PDA (Personal Digital Assistant), a PAD (tablet computer), a navigation device and other mobile terminals, and can also be an Internet of Things device terminal, etc., without specific limitation herein.
  • a terminal for example, a mobile phone, a smart phone, a laptop computer, a PDA (Personal Digital Assistant), a PAD (tablet computer), a navigation device and other mobile terminals, and can also be an Internet of Things device terminal, etc., without specific limitation herein.
  • the second communication node in all the above embodiments can be a base station, a receiver, an access point, etc., and no specific limitation is made here.
  • an embodiment of the present application further provides a communication device 100 , which includes at least one processor 101 and at least one memory 102 , and the memory 102 is used to store at least one program.
  • the processor 101 and the memory 102 may be connected via a bus or other means.
  • the memory 102 can be used to store non-transitory software programs and non-transitory computer executable programs.
  • the memory 102 may include a high-speed random access memory, and may also include a non-transitory memory, such as at least one disk storage device, a flash memory device, or other non-transitory solid-state storage device.
  • the memory 102 may optionally include a memory remotely arranged relative to the processor 101, and these remote memories may be connected to the processor 101 via a network. Examples of the above-mentioned network include, but are not limited to, the Internet, an intranet, a local area network, a mobile communication network, and combinations thereof.
  • the non-transitory software programs and instructions required to implement the information transmission method of the above embodiment are stored in the memory 102.
  • the information transmission method of the above embodiment is executed, for example, the method steps S110 to S120 in Figure 2 and the method step S210 in Figure 4 described above are executed.
  • the device embodiments described above are merely illustrative, and the units described as separate components may or may not be physically separated, that is, they may be located in one place or distributed on multiple network units. Some or all of the modules may be selected according to actual needs to achieve the purpose of the solution of this embodiment.
  • an embodiment of the present application also provides a computer-readable storage medium, which stores computer-executable instructions, and the computer-executable instructions are executed by a processor or controller, for example, by a processor in the above-mentioned device embodiment, so that the above-mentioned processor can execute the information transmission method in the above-mentioned embodiment, and execute the method steps S110 to S120 in Figure 2 described above and the method step S210 in Figure 4.
  • an embodiment of the present application also provides a computer program product, including a computer program or computer instructions, the computer program or computer instructions are stored in a computer-readable storage medium, the processor of a computer device reads the computer program or computer instructions from the computer-readable storage medium, and the processor executes the computer program or computer instructions, so that the computer device executes the information transmission method in the above embodiment, for example, executes method steps S110 to S120 in Figure 2 described above and method step S210 in Figure 4.
  • computer storage medium includes volatile and non-volatile, removable and non-removable media implemented in any method or technology for storing information (such as computer-readable instructions, data structures, program modules or other data).
  • Computer storage media include, but are not limited to, RAM, ROM, EEPROM, flash memory or other memory technologies, CD-ROM, digital versatile disks (DVD) or other optical disk storage, magnetic cassettes, magnetic tapes, disk storage or other magnetic storage devices, or any other medium that may be used to store desired information and may be accessed by a computer.
  • communication media typically embodies computer readable instructions, data structures, program modules, or other data in a modulated data signal such as a carrier wave or other transport mechanism, and may include any information delivery media.

Landscapes

  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
  • Power Engineering (AREA)

Abstract

本申请提供一种信息传输方法、通信设备及存储介质，该方法包括：可以确定第一数量的极稀疏导频，然后将数据包和第一数量的极稀疏导频一起传输至第二通信节点，其中，第一数量大于或者等于1，数据包至少包含调制符号，调制符号根据第一星座图模型对数据包中的M1+1个比特信息进行调制而得到，或者根据第二星座图模型对数据包中的M2+2个比特信息进行调制而得到，或者根据第三星座图模型对数据包中的M3+3个比特信息进行调制而得到，且M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数。

Description

信息传输方法、通信设备及存储介质
相关申请的交叉引用
本申请基于申请号为202211521420.5、申请日为2022年11月30日的中国专利申请提出，并要求该中国专利申请的优先权，该中国专利申请的全部内容在此引入本申请作为参考。
技术领域
本申请实施例涉及但不限于通信技术领域，尤其涉及一种信息传输方法、通信设备及存储介质。
背景技术
在无线通信系统与大量终端连接的场景下，无线通信系统需要与海量终端进行信息传输。进一步地，如果要求每个终端进行信息传输的频谱效率不能太低，则还需要通过提高调制方式的阶数以提高频谱效率。
相关技术中，通常采用的高阶调制方式为正交振幅调制(Quadrature Amplitude Modulation，QAM)，例如16QAM、32QAM、64QAM以及256QAM等方式，但是，这些高阶调制方式需要在信道估计较准确的情况下才能确保性能。如果信道估计误差较大，那么在解调时星座图会发生畸变，这将会导致解调性能下降，进而难以实现高谱效的信息传输。而且，在无线通信系统与大量终端连接的场景下，海量终端与系统进行信息传输，会增加终端处的导频开销，如果导频开销过大，将难以保证信道估计的准确性，因此，也会限制基站或者接入点的解调性能，同样难以实现高谱效的信息传输。因此，如何支持海量终端实现高谱效的信息传输，是亟待解决的一个问题。
发明内容
本申请实施例提供了一种信息传输方法、通信设备及存储介质，能够支持大量第一通信节点实现高谱效的信息传输。
第一方面，本申请实施例提供了一种信息传输方法，应用于第一通信节点，所述信息传输方法包括：
确定第一数量的极稀疏导频；
将数据包和所述第一数量的所述极稀疏导频一起传输至第二通信节点；
其中，所述第一数量大于或者等于1，所述数据包至少包含调制符号；
所述调制符号根据第一星座图模型对所述数据包中的M1+1个比特信息进行调制而得到，所述第一星座图模型包含2*N1个星座点，M1为大于或等于1的整数，N1和M1满足公式N1＝2M1；
所述第一星座图模型中的2*N1个所述星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN1ejθ,
a1ej(θ+π),a2ej(θ+π),…,aN1ej(θ+π)；
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN1均为正数，并且满足：0<a1<a2<…<aN1；
或者,
所述调制符号根据第二星座图模型对所述数据包中的M2+2个比特信息进行调制而得到，所述第二星座图模型包含4*N2个星座点，M2为大于或等于1的整数，N2和M2满足公式N2＝2M2；
所述第二星座图模型中的4*N2个所述星座点所对应的复数形式包括如下:
a1ejθ,a2ejθ,…,aN2ejθ,
b1ej(θ+π/2),b2ej(θ+π/2),…,bN2ej(θ+π/2),
a1ej(θ+π),a2ej(θ+π),…,aN2ej(θ+π),
b1ej(θ+3π/2),b2ej(θ+3π/2),…,bN2ej(θ+3π/2)；
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN2和b1,b2,…,bN2均为正数，并且满足：0<a1<a2<…<aN2，0<b1<b2<…<bN2；
或者,
所述调制符号根据第三星座图模型对所述数据包中的M3+3个比特信息进行调制而得到，所述第三星座图模型包含8*N3个星座点，M3为大于或等于0的整数，N3和M3满足公式N3＝2M3；
所述第三星座图模型中的8*N3个所述星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN3ejθ,



a1ej(θ+π),a2ej(θ+π),…,aN3ej(θ+π),


π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN3和b1,b2,…,bN3均为正数，并且满足：0<a1<a2<…<aN3，0<b1<b2<…<bN3。
第二方面，本申请实施例提供了一种信息传输方法，应用于第二通信节点，所述信息传输方法包括：
接收第一通信节点发送的数据包和第一数量的极稀疏导频；
其中，所述第一数量大于或者等于1，所述数据包至少包含调制符号；
所述调制符号根据第一星座图模型对所述数据包中的M1+1个比特信息进行调制而得到，所述第一星座图模型包含2*N1个星座点，M1为大于或等于1的整数，N1和M1满足公式N1＝2M1；
所述第一星座图模型中的2*N1个所述星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN1ejθ,
a1ej(θ+π),a2ej(θ+π),…,aN1ej(θ+π)；
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN1均为正数，并且满足：0<a1<a2<…<aN1；
或者,
所述调制符号根据第二星座图模型对所述数据包中的M2+2个比特信息进行调制而得到，所述第二星座图模型包含4*N2个星座点，M2为大于或等于1的整数，N2和M2满足公式N2＝2M2；
所述第二星座图模型中的4*N2个所述星座点所对应的复数形式包括如下:
a1ejθ,a2ejθ,…,aN2ejθ,
b1ej(θ+π/2),b2ej(θ+π/2),…,bN2ej(θ+π/2),
a1ej(θ+π),a2ej(θ+π),…,aN2ej(θ+π),
b1ej(θ+3π/2),b2ej(θ+3π/2),…,bN2ej(θ+3π/2)；
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN2和b1,b2,…,bN2均为正数，并且满足：0<a1<a2<…<aN2，0<b1<b2<…<bN2；
或者,
所述调制符号根据第三星座图模型对所述数据包中的M3+3个比特信息进行调制而得到，所述第三星座图模型包含8*N3个星座点，M3为大于或等于0的整数，N3和M3满足公式N3＝2M3；
所述第三星座图模型中的8*N3个所述星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN3ejθ,



a1ej(θ+π),a2ej(θ+π),…,aN3ej(θ+π),


π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN3和b1,b2,…,bN3均为正数，并且满足：0<a1<a2<…<aN3，0<b1<b2<…<bN3。
第三方面，本申请实施例还提供了一种通信设备，包括：存储器、处理器及存储在存储器上并可在处理器上运行的计算机程序，所述处理器执行所述计算机程序时实现如上所述的信息传输方法。
第四方面，本申请实施例还提供了一种计算机可读存储介质，存储有计算机可执行指令，所述计算机 可执行指令用于执行如上所述的信息传输方法。
第五方面，本申请实施例还提供了一种计算机程序产品，包括计算机程序或计算机指令，所述计算机程序或所述计算机指令存储在计算机可读存储介质中，计算机设备的处理器从所述计算机可读存储介质读取所述计算机程序或所述计算机指令，所述处理器执行所述计算机程序或所述计算机指令，使得所述计算机设备执行如上所述的信息传输方法。
本申请实施例包括：可以确定第一数量的极稀疏导频，然后将数据包和第一数量的极稀疏导频一起传输至第二通信节点，其中，第一数量大于或者等于1，数据包至少包含调制符号，调制符号根据第一星座图模型对数据包中的M1+1个比特信息进行调制而得到，或者根据第二星座图模型对数据包中的M2+2个比特信息进行调制而得到，或者根据第三星座图模型对数据包中的M3+3个比特信息进行调制而得到，且M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数，即是说，第一通信节点通过向第二通信节点传输极稀疏导频以及包含该调制符号的数据包，以便于第二通信节点可以从极稀疏导频中估计出无线信道的部分信息，再进一步从调制符号中提取出信道信息，而无需从极稀疏导频中估计出无线信道的全部信息，从而能够降低第一通信节点的导频开销，进而能够提高信道估计的准确性，从而能够提高第二通信节点的解调性能，实现高谱效的信息传输；又因为M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数，所以调制符号可以根据第一星座图模型、第二星座图模型或者第三星座图模型对数据包中的多个比特信息进行调制而得到，即该调制符号可以承载多个比特信息，从而实现高阶调制，进而能够提高信息传输的频谱效率。因此，本申请实施例能够支持大量第一通信节点实现高谱效的信息传输。
附图说明
图1是本申请一个实施例提供的64QAM调制符号所对应的星座图；
图2是本申请一个实施例提供的信息传输方法的流程图；
图3是本申请一个实施例提供的w个相互独立的导频的示意图；
图4是本申请另一个实施例提供的信息传输方法的流程图；
图5是本申请一个实施例提供的第一星座图模型和第二星座图模型的示意图；
图6是本申请一个实施例提供的在经过信道旋转缩放前后的十字形星座图的示意图；
图7是本申请一个实施例提供的在二维平面坐标系上划分分区的示意图；
图8是本申请一个实施例提供的十字形星座图的示意图；
图9是本申请另一个实施例提供的十字形星座图的示意图；
图10是本申请另一个实施例提供的十字形星座图的示意图；
图11是本申请另一个实施例提供的十字形星座图的示意图；
图12是本申请另一个实施例提供的十字形星座图的示意图；
图13是本申请另一个实施例提供的十字形星座图的示意图；
图14是本申请另一个实施例提供的十字形星座图的示意图；
图15是本申请另一个实施例提供的十字形星座图的示意图；
图16是本申请另一个实施例提供的十字形星座图的示意图；
图17是本申请另一个实施例提供的十字形星座图的示意图；
图18是本申请另一个实施例提供的十字形星座图的示意图；
图19是本申请一个实施例提供的PAM星座图的示意图；
图20是本申请另一个实施例提供的PAM星座图的示意图；
图21是本申请一个实施例提供的第三星座图模型的示意图；
图22是本申请另一个实施例提供的第三星座图模型的示意图；
图23是本申请一个实施例提供的定义物理资源块的示意图；
图24是本申请一个实施例提供的一种定义解调参考信号的示意图；
图25是本申请一个实施例提供的通过OCC码来区分出不同的参考信号端口的示意图；
图26是本申请一个实施例提供的一种采用OCC码定义解调参考信号的示意图；
图27是本申请一个实施例提供的另一种采用OCC码定义解调参考信号的示意图；
图28是本申请一个实施例提供的另一种采用OCC码定义解调参考信号的示意图；
图29是本申请一个实施例提供的一种基于PRB传输的示意图；
图30是本申请一个实施例提供的另一种定义解调参考信号的示意图；
图31是本申请一个实施例提供的另一种定义解调参考信号的示意图；
图32是本申请一个实施例提供的另一种定义解调参考信号的示意图；
图33是本申请一个实施例提供的一种定义参考信号的示意图；
图34是本申请一个实施例提供的另一种定义参考信号的示意图；
图35是本申请一个实施例提供的另一种定义参考信号的示意图；
图36是本申请一实施例提供的一种生成DMRS端口的示意图；
图37是本申请一个实施例提供的另一种定义参考信号的示意图；
图38是本申请一个实施例提供的另一种定义参考信号的示意图；
图39是本申请一个实施例提供的通信设备的结构示意图。
具体实施方式
为了使本申请的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本申请进行进一步详细说明。应当理解，此处所描述的具体实施例仅用以解释本申请，并不用于限定本申请。
需要说明的是，虽然在流程图中示出了逻辑顺序，但是在某些情况下，可以以不同于流程图中的顺序执行所示出或描述的步骤。说明书和权利要求书及上述附图的描述中，多个(或多项)的含义是两个以上，大于、小于、超过等理解为不包括本数，以上、以下、以内等理解为包括本数。如果有描述到“第一”、“第二”等只是用于区分技术特征为目的，而不能理解为指示或暗示相对重要性或者隐含指明所指示的技术特征的数量或者隐含指明所指示的技术特征的先后关系。
本申请提供了一种信息传输方法、通信设备及存储介质，第一通信节点可以确定第一数量的极稀疏导频，然后将数据包和第一数量的极稀疏导频一起传输至第二通信节点，其中，第一数量大于或者等于1，数据包至少包含调制符号，调制符号根据第一星座图模型对数据包中的M1+1个比特信息进行调制而得到，或者根据第二星座图模型对数据包中的M2+2个比特信息进行调制而得到，或者根据第三星座图模型对数据包中的M3+3个比特信息进行调制而得到，且M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数，即是说，第一通信节点通过向第二通信节点传输极稀疏导频以及包含该调制符号的数据包，以便于第二通信节点可以从极稀疏导频中估计出无线信道的部分信息，再进一步从调制符号中提取出信道信息，而无需从极稀疏导频中估计出无线信道的全部信息，从而能够降低第一通信节点的导频开销，进而能够提高信道估计的准确性，从而能够提高第二通信节点的解调性能，实现高谱效的信息传输；又因为M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数，所以调制符号可以根据第一星座图模型、第二星座图模型或者第三星座图模型对数据包中的多个比特信息进行调制而得到，即该调制符号可以承载多个比特信息，从而实现高阶调制，进而能够提高信息传输的频谱效率。因此，本申请实施例能够支持大量第一通信节点实现高谱效的信息传输。
值得注意的是，在无线通信系统与大量终端连接的场景下，无线通信系统需要与海量终端进行信息传输。进一步地，如果要求每个终端进行信息传输的频谱效率不能太低，则还需要通过提高调制方式的阶数以提高频谱效率。
相关技术中，通常采用的高阶调制方式为正交振幅调制，例如16QAM、32QAM、64QAM以及256QAM等方式，其星座图中的星座点在二维平面(即复平面)上分布比较均匀，因而可以比较充分地利用复信号的二维信号空间(即二维信号平面)。可以理解的是，通信信号在基带上通常可以用复数表示，即通信信号可以分为I路信号和Q路信号，其中，I路信号是实部，Q路信号是虚部。又因为，通信信号包括调制符号，因此，调制符号也可以用复数表示，也就是一个调制符号可以采用一个复数进行表示，例如，调制符号s可以表示为a+j*b，其中j是虚数，即j＝sqrt(-1)；a是s的实部，表示调制符号在I路传输，而b是s的虚部，表示调制符号在Q路传输。
以相关技术中的16QAM方式的星座图为例，该星座图包含16个点，这16个点所对应的复数包括如下：
3+3j,3+j,3-j,3-3j,
1+3j,1+j,1-j,1-3j,
-1+3j,-1+j,-1-j,-1-3j,
-3+3j,-3+j,-3-j,-3-3j
由该16个点所对应的复数可知，在实部取值范围为-3至3以及虚部取值范围为-3至3的二维平面(也可以称为复平面，或者复信号空间，或者二维信号空间)内，16QAM的星座点分布比较均匀。可以理解的是，复平面与二维平面等价，所以复平面或二维平面也可以称为二维复平面，其中，复数的实部与二维平面的x坐标等价，复数的虚部与二维平面的y坐标等价。因此复数也可以用二维平面上的点表示，例如，复数a+j*b可以用二维平面上的坐标(a,b)表示，其中，坐标(a,b)表示二维平面上的x坐标为a，y坐标为b。因此，16QAM星座图中的16个点除了可以用16个复数进行表示，还可以用二维平面上的16个二维坐标进行表示，其中，16个二维坐标包括如下：
(3,3),(3,1),(3,-1),(3,-3),
(1,3),(1,1),(1,-1),(1,-3),
(-1,3),(-1,1),(-1,-1),(-1,-3),
(-3,3),(-3,1),(-3,-1),(-3,-3)
在本实施例中，在需要对星座图整体进行功率归一化处理的情况下，可以将星座图整体乘以一个归一化因子(或称缩放因子)，例如，16QAM调制方式对应的16个复数都会乘以同一个归一化因子1/sqrt(40)，功率归一化处理后的16QAM星座图包含的16个点所对应的复数包括如下：
1/sqrt(40)*[3+3j,3+j,3-j,3-3j,1+3j,1+j,1-j,1-3j,-1+3j,-1+j,-1-j,-1-3j,-3
+3j,-3+j,-3-j,-3-3j]
功率归一化处理后的16QAM星座图包含的16个点所对应的坐标可以通过上述列出的16个二维坐标乘以1/sqrt(40)得到，也就是将每个二维坐标的x坐标和y坐标都乘以1/sqrt(40)。
可以理解的是，功率归一化只是让星座图整体缩小，缩小后的星座图中各个星座点仍然均匀分布。
对于其它高阶调制方式，例如32QAM，64QAM，256QAM等调制方式，均与16QAM调制方式相类似，其星座图的星座点在二维平面上均匀分布，所以相关技术中的高阶调制方式可以充分地利用复信号的二维信号空间，而且这些高阶调制方式所对应的解调方式不仅简单，而且能够保证性能，因此，这些高阶调制方式可以比较简单且高效地逼近传输的性能极限，即香农限。所以，在对频谱效率有一定需求的场景中，这些高阶调制方式能够得到广泛应用。但是，这些高阶调制方式需要在信道估计较准确的情况下才能确保性能。如果信道估计误差较大，那么基站(或者接入点)解调时星座图会发生畸变，即旋转缩放，此时，解调性能会下降。
具体地，以调制符号经正交频分复用(Orthogonal Frequency Division Multiplexing，OFDM)方式传输(即利用OFDM的子载波对调制符号进行传输)为例，经过多径信道或者频选信道后，承载在OFDM的子载波上的调制符号会被加权一个复数权值(也即频选信道会引发承载在子载波上的调制符号畸变)；或者，如果收发双方(即第一通信节点与第二通信节点双方)存在较大的同步误差，则定时偏差(即时偏)和频率偏差(即频偏)也会导致子载波上的调制符号被加权一个复数权值，也即同步误差会引发调制符号畸变。同样地，在高速移动场景或者卫星通信场景中，多普特效应也会导致子载波上的调制符号被加权一个复数权值，也即同步误差会引发调制符号畸变。由于在同步误差较大的场景、高速移动场景(例如高速移动场景或者卫星通信场景等)或者相位噪声较大的场景下，调制符号的旋转缩放量(即旋转量和缩放量)变化非常快，因此，需要时间间隔很短的导频才能较准确估计出导频之间的调制符号的旋转缩放量，但是，这会增加导频开销，最终导致传输的频谱效率下降。另外，如果导频在时间上的密度不满足要求，那么同样难以准确估计调制符号的旋转缩放量，所以解调性能同样会下降。
进一步地，这些发生在调制符号上的畸变会叠加。以调制符号s经OFDM方式传输为例，假设频选信道导致OFDM的子载波上的调制符号s的复数权值为g1，时频或频偏导致调制符号的复数权值为g2，若同时存在上述频选信道的情况和上述同步误差的情况，则相当于利用一个加权值h对调制符号进行加权，其中h＝g1*g2，也就是接收到的调制符号是y＝h*s+n＝g1*g2*s+n，其中，n是加性高斯白噪声(Additive White Gaussian Noise，AWGN)。接收侧(即第二通信节点)如果不能将调制符号上的畸变去除，也即不能将调制符号的加权值h均衡掉，那么调制符号会被旋转缩放。经过细微的旋转缩放的调制符号也会严重制约高阶调制方式的性能，如图1所示，图中的每个小圆点对应一个调制符号，其中图1中左边的坐标系所对应的星座图是标准的64QAM调制符号所对应的星座图；而图1中右边的坐标系所对应的星座图是通过一个加权值(也即一个旋转缩放量)加权后的64QAM调制符号所对应的星座图，也即经历了信道畸变的64QAM调制符号所对应的星座图。如果接收侧直接对如图1中右边的坐标系所对应的星座图进行解调，则即使接收侧的AWGN很小，解调性能也会受到影响。因此，在传统高谱效场景下，通常会利用导频去估计调制符号的复数权值(即畸变)，也就是会将接收到的调制符号y＝h*s+n＝g1*g2*s+n中的h估计出来，然后再将该加权值均衡掉，即将y除以h，也即y/h＝s+n/h，得到没有畸变且只受加性高斯白噪声影响的星座图s+n′，从而能够得到更好的解调性能，其中，n′＝n/h。但是，对于海量终端处于无连接状态(即终端与无线通信系统没有连接)时直接与无线通信系统进行信息传输的场景，以及海量终端基于半持续式调度(Semi-Persistent Scheduling，SPS)的信息传输场景等场景，难以通过导频准确估计出调制符号的复数权值(即畸变)，因此，高阶调制方式的性能会被严重制约。
此外，在无线通信系统与大量终端连接的场景下，海量终端与系统进行信息传输，会增加终端处的导频开销，这将难以保证信道估计的准确性，因此，也会限制基站(或者接入点)的解调性能。
下面对沿用传统的导频方案来实现在第一通信节点与第二通信节点之间的信息传输而导致导频开销的增加进行具体说明。
为了省电，终端在无需进行信息传输时，通常不会与系统建立连接(即终端与系统没有连接，或者终端与系统断开连接)，即终端处于无连接态(其中，Non Connected state，或Non RRC Connected state，或Connectionless state，或Connection-free state，或Disconnected state等均可表示无连接态)。可以理解的是，空闲态(Idle state)或非激活态(Inactive state)可认为与无连接态等价，或者，空闲态(Idle state)或非激活态(Inactive state)也可认为是一种无连接态。
当终端原来处于无连接态(即还没有进入连接态，或者还没有与系统建立连接)，如果沿用相关技术 中的上行进行信息传输方案，为了传输信息，终端必须在传输前与系统建立连接。在进入连接态(也可称激活态(Active state))后，终端才能进一步去向系统(如基站或接入点)申请上行传输资源，且在获得系统的资源授权或资源调度后才能进行信息传输。可见，终端要完成一次传统的上行进行信息传输，需要提前完成很多操作，这无疑会增加终端所产生的功耗，以及系统的信令开销。
反之，若终端在无连接状态下与系统进行信息传输，那么在信息传输之前，终端无需建立连接，也无需向基站(或者接入点)申请专用的传输资源，即允许处于无连接态的终端在信息传输之前无需通知基站(或者接入点)，而是自主地直接在一个预设的公共传输资源上向基站传输信息，因此，无连接状态下的信息传输可以降低信息传输的复杂度，减少终端在信息传输时所产生的功耗，以及数据达到基站(或者接入点)的传输时延，同时也节省了信息传输时所需的信令开销。
但是，在无连接状态下的信息传输的场景下，第一通信节点(比如终端)需要自主选择地从预设导频集合中选择导频(或者参考信号)，然而，这会导致一个问题，即由于没有一个中心节点对不同的第一通信节点所发送的导频进行统筹安排，因此，不同的第一通信节点从一个导频数量有限的预设导频集合里自主选择导频，会出现所选择的导频相同的情况，从而会产生导频碰撞的问题。在高过载(即向同一个第二通信节点发送数据包的第一通信节点很多)的场景下，出现导频碰撞的概率非常高。一旦不同的第一通信节点的导频发生碰撞，第二通信节点就难以通过导频分离出多个第一通信节点。
因此，为了减少导频碰撞的次数，需要定义更多的导频，即是说，在预设导频集合中的导频数量要尽可能多，而增加导频数量意味着导频所占用的开销也增加。另外，如果沿用传统的导频方案，即要求通过每个导频来估计信道和时频偏，以此完成调制符号的相关解调，所以每个导频在整个传输带宽以及时间内都需要有信号，换而言之，每个导频在整个传输带宽以及时间内不能分布得太稀疏，这样才能估计出整个传输带宽的信道信息(比如无线多径信道，也即频率选择性信道)以及传输时间内的频偏。所以，要确保无连接状态下的信息传输的传输性能，沿用传统的导频方案，会导致导频所占用的开销成倍增加，检测复杂度也会显著增加。
进一步地，相关技术中有一种上行信息传输的方式，即半持续式调度，其目的是降低传输的物理控制信令开销和时延，十分适用于周期性的业务。但是如果沿用SPS方式来实现大量终端的信息传输，为了提高周期预留资源的利用率，可以让一份SPS资源预留给多个终端共同使用，这就需要在一定的时频资源内定义大量的导频。同理，在一定的导频资源下，极稀疏导频可以使得可用的导频数量最大化，进而能够使得使用同一份SPS资源的终端的数量最大化，因此，极稀疏导频非常适用于海量终端基于半持续式调度的信息传输场景。
针对上述情况，本申请提出以极稀疏导频和调制方式相结合的方式支持海量终端在较高的频谱效率下进行信息传输。
可以理解的是，基于极稀疏导频的方案可以让系统支持更多的终端接入，因此，该基于极稀疏导频的方案非常适合大连接场景，例如，海量终端在无连接状态下直接传输信息的场景，以及海量用户基于SPS的信息传输场景。但是基于极稀疏导频的方案要求接收侧(即第二通信节点)能够通过调制符号自身的特点来进行信道估计，而传统的高阶调制方式的星座图过于密集，不利于接收侧通过调制符号提取信道信息。基于此，本申请提出一种信息传输方法，既能够支持高谱效应用场景，又能够降低接收侧通过调制符号提取信道信息的复杂度，以及提高接收侧通过调制符号所提取的信道信息的准确性。
参照图2，图2是本申请一个实施例提供的信息传输方法的流程图，该信息传输方法应用于第一通信节点，该信息传输方法可以包括但不限于步骤S110和步骤S120。
步骤S110：确定第一数量的极稀疏导频。
步骤S120：将数据包和第一数量的极稀疏导频一起传输至第二通信节点。
其中，第一数量大于或者等于1，数据包至少包含调制符号。
进一步地，调制符号根据第一星座图模型对数据包中的M1+1个比特信息进行调制而得到，第一星座图模型包括2*N1个星座点，M1为大于或者等于1的整数，N1和M1满足公式N1＝2M1；
第一星座图模型中的2*N1个星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN1ejθ,
a1ej(θ+π),a2ej(θ+π),…,aN1ej(θ+π)
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN1均为正数，并且满足：0<a1<a2<…<aN1；
或者,
调制符号根据第二星座图模型对数据包中的M2+2个比特信息进行调制而得到，第二星座图模型包含4*N2个星座点，M2为大于或等于1的整数，N2和M2满足公式N2＝2M2；
第二星座图模型中的4*N2个星座点所对应的复数形式包括如下:
a1ejθ,a2ejθ,…,aN2ejθ,
b1ej(θ+π/2),b2ej(θ+π/2),…,bN2ej(θ+π/2),
a1ej(θ+π),a2ej(θ+π),…,aN2ej(θ+π),
b1ej(θ+3π/2),b2ej(θ+3π/2),…,bN2ej(θ+3π/2)；
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN2和b1,b2,…,bN2均为正数，并且满足：0<a1<a2<…<aN2，0<b1<b2<…<bN2；
或者,
调制符号根据第三星座图模型对数据包中的M3+3个比特信息进行调制而得到，第三星座图模型包含8*N3个星座点，M3为大于或等于0的整数，N3和M3满足公式N3＝2M3；
第三星座图模型中的8*N3个星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN3ejθ,



a1ej(θ+π),a2ej(θ+π),…,aN3ej(θ+π),


π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN3和b1,b2,…,bN3均为正数，并且满足：0<a1<a2<…<aN3，0<b1<b2<…<bN3。
在本实施例中，通过采用包括上述步骤S110和步骤S120的信息传输方法，可以确定第一数量的极稀疏导频，然后将数据包和第一数量的极稀疏导频一起传输至第二通信节点，其中，第一数量大于或者等于1，数据包至少包含调制符号，调制符号根据第一星座图模型对数据包中的M1+1个比特信息进行调制而得到，或者根据第二星座图模型对数据包中的M2+2个比特信息进行调制而得到，或者根据第三星座图模型对数据包中的M3+3个比特信息进行调制而得到，且M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数,即是说，第一通信节点通过向第二通信节点传输极稀疏导频以及包含该调制符号的数据包，以便于第二通信节点可以从极稀疏导频中估计出无线信道的部分信息，再进一步从调制符号中提取出信道信息，而无需从极稀疏导频中估计出无线信道的全部信息，从而能够降低第一通信节点的导频开销，进而能够提高信道估计的准确性，从而能够提高第二通信节点的解调性能，实现高谱效的信息传输；又因为M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数，所以调制符号可以根据第一星座图模型、第二星座图模型或者第三星座图模型对数据包中的多个比特信息进行调制而得到，即该调制符号可以承载多个比特信息，从而实现高阶调制，进而能够提高信息传输的频谱效率。因此，本申请实施例能够支持大量第一通信节点实现高谱效的信息传输。
可以理解的是，导频可以称为导频信号，或参考信号(Reference Signal，RS)，或解调参考信号，或前导，而且从形式上看，导频通常是一条序列或一串符号，所以导频也称为导频序列。
一可行的实施方式，每个极稀疏导频均包括第二数量的非零值符号，其中，第二数量大于0且小于5。第二数量的非零值符号承载在时频域上相邻的第三数量的资源单元(Resource Elements，REs)上，或者承载在按时间先后顺序的第三数量的符号上，或者承载在频域相邻子载波上的第三数量的资源单元上，其中，第三数量与第二数量相等(即第三数量大于0且小于5)。可以理解的是，每个极稀疏导频只有一个参考信号单元(Reference Signal Element，RSE)，该参考信号单元包含的非零值符号的符号数量为第二数量。可以理解的是，对于OFDM而言，一个资源单元(RE)可以承载一个符号。另外，若第二数量的非零值符号承载在相邻的RE上，相当于该第二数量的符号承载在相邻的时频资源上。或者，对于OFDM而言，若第二数量的非零值符号承载在相邻的时频资源上，相当于该第二数量的非零值符号承载在相邻的RE上。
一可行的实施方式，第二数量的取值为4，第二数量的非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值至少包括如表1中的各种情况，其中，表1中的不同序号对应于不同的情况。
表1

一可行的实施方式，第二数量的取值为1，即一个非零值符号；或者，第二数量的取值为2，第二数量的非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值为[a1,a2]或者[b1,b2]，其中，[a1,a2]和[b1,b2]正交，即a1′*b1+a2′*b2＝0；或者，第二数量的取值为2，第二数量的非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值至少包括[p1,p2]＝[1,1]；[p1,p2]＝[1,-1]；[p1,p2]＝[1,j]；[p1,p2]＝[1,-j]等情况；或者，第二数量的取值为4，第二数量的非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值为[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]或者[d1,d2,d3,d4]，其中，[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]和[d1,d2,d3,d4]相互正交，即[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]和[d1,d2,d3,d4]中的任意两个相互正交，具体地，[a1,a2,a3,a4]与[b1,b2,b3,b4]相互正交，即a1′*b1+a2′*b2+a3′*b3+a4′*b4＝0；[a1,a2,a3,a4]与[c1,c2,c3,c4]相互正交，即a1′*c1+a2′*c2+a3′*c3+a4′*c4＝0，等等，在此不再赘述。
一可行的实施方式，每个极稀疏导频的符号长度均大于24，即每个极稀疏导频的非零值符号与零值符号的总数量大于24，在此不作具体限制。
一可行的实施方式，当第一数量的取值大于或者等于2时，第一数量的极稀疏导频之间相互独立，即第一数量的极稀疏导频之间不关联或者不相关，其中，第一数量的取值可以大于或者等于1，即第一数量的取值可以是1、2或者其他取值，在此不作具体限制。其中，将一次传输中包含2个或多个导频，并且导频之间无关联或者相互独立的技术称为独立多导频技术，该多个相互独立的导频称为独立多导频。如图3所示，在一次传输中包含w个极稀疏导频，其中，w个极稀疏导频分别表示为P1、P2、......、Pw，w可以为大于2的正整数，并且数据包中包含该w个极稀疏导频的信息，例如，该数据包中包含该w个极稀疏导频的索引号(即极稀疏导频在预设导频集合中的索引号)，这样，一旦某个终端的数据包译码成功，就可以确定该终端在此次信息传输中所使用的所有极稀疏导频的信息，从而可以对导频信号进行干扰消除。
可以理解的是，第一通信节点与第二通信节点的信息传输可以采用独立多导频技术，这样，在相同的导频开销下，不同的第一通信节点的独立多导频同时碰撞的概率会比传统单导频碰撞的概率小。因此，在 竞争式无连接状态(或者竞争式免调度方式)的传输场景下，可以采用独立多导频技术以支持更多的第一通信节点进行信息传输。进一步地，可以结合独立多导频技术和极稀疏导频技术，即采用多个相互独立的极稀疏导频，进一步减少导频碰撞的概率，进一步提升接入的第一通信节点的数量。
在一实施例中，当第二通信节点为基站时，通过基于迭代的接收机，基站每轮都可以通过不碰撞(即相互独立)的多个极稀疏导频解调出对应的第一通信节点，然后将重构出该第一通信节点的数据包和极稀疏导频，并从接收信号中将该第一通信节点对应的数据包和极稀疏导频消除，如此迭代直至解调出所有可解的第一通信节点，以减少导频碰撞的概率，进一步提升接入的第一通信节点的数量。
在一实施例中，当第一数量的取值大于或者等于2时，第一数量的极稀疏导频可以根据数据包中的信息而确定。
在另一实施例中，当第一数量的取值大于或者等于2时，第一数量的极稀疏导频可以根据数据包中的一个或者多个比特信息而确定，比如，一个极稀疏导频可以根据数据包中的一个比特信息确定；又如，一个极稀疏导频可以根据数据包中的两个比特信息确定；再如，两个极稀疏导频均根据数据包中的多个比特信息确定，等等，本申请实施例不对第一数量和比特信息的数量作限制。
在另一实施例中，当第一数量的取值大于或者等于2时，每个极稀疏导频均根据数据包中的第四数量的比特信息从预设导频集合中确定得到，其中，预设导频集合包括第五数量的导频，第四数量与第五数量成对数函数关系，对数函数为以2作为底数的对数函数。比如，假设第五数量是D个，则第四数量为log2(D)个，在此不作具体限制。可以理解的是，第五数量可以是64、128或者更多，在此不做具体限制。
一可行的实施方式，当调制符号根据第一星座图模型而调制得到，a1,a2,…,aN1均可由如下公式表示，即
an＝(2n-1+Δ)d；
其中，n的取值包括1,2,...,N1，即an可以是a1，也可以是a2，还可以是a3，或者aN1等等。d为正实数，Δ为大于或等于0的实数，使得a1,a2,…,aN1构成等差数列。
进一步地，当Δ的取值为0，且d的取值为1，使得an满足an＝2n-1；当Δ的取值为1，且d的取值为1/2，使得an满足an＝n；当Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
或者，Δ的取值可以为此时，当d的取值为1，使得an满足当d的取值为1/2，使得an满足
一可行的实施方式，当调制符号根据第二星座图模型而调制得到，a1,a2,…,aN2均可由如下公式表示：
an＝(2n-1+Δ)d；
b1,b2,…,bN2均可由如下公式表示：
bn＝an+β；
其中，n的取值包括1,2,...,N2，即an可以是a1，也可以是a2，还可以是a3，或者aN2等等。同样地，bn可以是b1，也可以是b2，还可以是b3或者bN2等等。d为正实数，Δ和β均为大于或等于0的实数，使得a1,a2,…,aN2构成等差数列，b1,b2,…,bN2构成等差数列。
进一步地，当Δ的取值为0，且d的取值为1，使得an满足an＝2n-1；当Δ的取值为1，且d的取值为1/2，使得an满足an＝n；当Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
或者，Δ的取值可以为此时，当d的取值为1，使得an满足当d的取值为1/2，使得an满足
一可行的实施方式，当调制符号根据第三星座图模型而调制得到，a1,a2,…,aN3均可由如下公式表示：
an＝(2n-1+Δ)d；
b1,b2,…,bN3均可由如下公式表示：
bn＝an+β；
其中，n的取值包括1,2,...,N3，即an可以是a1，也可以是a2，还可以是a3，或者aN3等等。同样地，bn可以是b1，也可以是b2，还可以是b3，或者bN3等等。d为正实数，Δ和β均为大于或等于0的实数，使得a1,a2,…,aN3构成等差数列，b1,b2,…,bN3构成等差数列。
进一步地，当Δ的取值为0，且d的取值为1，使得an满足an＝2n-1；当Δ的取值为1，且d的取值为1/2，使得an满足an＝n；当Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
或者，Δ的取值可以为此时，当d的取值为1，使得an满足当d的取值为1/2，使得an满足
一可行的实施方式，当调制符号根据第二星座图模型或者第三星座图模型而调制得到，β等于0。
一可行的实施方式，当调制符号根据第三星座图模型而调制得到，β大于0。
一可行的实施方式，当调制符号根据第一星座图模型而调制得到，d的取值为使采用第一星座图模型 进行调制得到的调制符号的平均功率等于1的数值，即d的取值使得第一星座图模型中的星座点的模平方的均值是1，也即d的取值使得第一星座图模型的平均功率是1，在此不作具体限制。
一可行的实施方式，当调制符号根据第二星座图模型而调制得到，d的取值为使采用第二星座图模型进行调制得到的调制符号的平均功率等于1的数值，即d的取值使得第二星座图模型中的星座点的模平方的均值是1，也即d的取值使得第二星座图模型的平均功率是1，在此不作具体限制。
一可行的实施方式，当调制符号根据第三星座图模型而调制得到，d的取值为使采用第三星座图模型进行调制得到的调制符号的平均功率等于1的数值，即d的取值使得第三星座图模型中的星座点的模平方的均值是1，也即d的取值使得第三星座图模型的平均功率是1，在此不作具体限制。
一可行的实施方式，θ的取值可以为0；或者，θ的取值可以为π/4，即满足公式θ＝π/4；或者，θ的取值满足公式θ＝π/8，在此不作具体限制。
另外，图4是本申请另一个实施例提供的一种信息传输方法，该信息传输方法应用于第二通信节点，该信息传输方法可以包括但不限于步骤S210。
步骤S210：接收第一通信节点发送的数据包和第一数量的极稀疏导频。
其中，第一数量大于或者等于1，数据包至少包含调制符号，调制符号根据第一星座图模型对数据包中的M1+1个比特信息进行调制而得到，第一星座图模型包含2*N1个星座点，M1为大于或等于1的整数，N1和M1满足公式N1＝2M1；
第一星座图模型中的2*N1个星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN1ejθ,
a1ej(θ+π),a2ej(θ+π),…,aN1ej(θ+π)；
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN1均为正数，并且满足：0<a1<a2<…<aN1；
或者,调制符号根据第二星座图模型对数据包中的M2+2个比特信息进行调制而得到，第二星座图模型包含4*N2个星座点，M2为大于或等于1的整数，N2和M2满足公式N2＝2M2；
第二星座图模型中的4*N2个星座点所对应的复数形式包括如下:
a1ejθ,a2ejθ,…,aN2ejθ,
b1ej(θ+π/2),b2ej(θ+π/2),…,bN2ej(θ+π/2),
a1ej(θ+π),a2ej(θ+π),…,aN2ej(θ+π),
b1ej(θ+3π/2),b2ej(θ+3π/2),…,bN2ej(θ+3π/2)；
π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN2和b1,b2,…,bN2均为正数，并且满足：0<a1<a2<…<aN2，0<b1<b2<…<bN2；
或者,调制符号根据第三星座图模型对数据包中的M3+3个比特信息进行调制而得到，第三星座图模型包含8*N3个星座点，M3为大于或等于0的整数，N3和M3满足公式N3＝2M3；
第三星座图模型中的8*N3个星座点所对应的复数形式包括如下：
a1ejθ,a2ejθ,…,aN3ejθ,



a1ej(θ+π),a2ej(θ+π),…,aN3ej(θ+π),


π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN3和b1,b2,…,bN3均为正数，并且满足：0<a1<a2<…<aN3，0<b1<b2<…<bN3。
在本实施例中，通过采用包括上述步骤S210的信息传输方法，第二通信节点可以接收第一通信节点发送的数据包和第一数量的极稀疏导频，其中，第一数量大于或者等于1，数据包至少包含调制符号，调制符号根据第一星座图模型对数据包中的M1+1个比特信息进行调制而得到，或者根据第二星座图模型对数据包中的M2+2个比特信息进行调制而得到，或者根据第三星座图模型对数据包中的M3+3个比特信息进行调制而得到，且M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数，即是说，第二通信节点可以从第一通信节点所发送的极稀疏导频中估计出无线信道的部分信息，再进一步 从调制符号中提取出信道信息，而无需从极稀疏导频中估计出无线信道的全部信息，从而能够降低第一通信节点的导频开销，进而能够提高信道估计的准确性，从而能够提高第二通信节点的解调性能，实现高谱效的信息传输；又因为M1为大于或等于1的整数，M2为大于或等于1的整数，M3为大于或等于0的整数，调制符号可以根据第一星座图模型、第二星座图模型或者第三星座图模型对数据包中的多个比特信息进行调制而得到，即该调制符号可以承载多个比特信息，从而实现高阶调制，进而能够提高信息传输的频谱效率。因此，本申请实施例能够支持大量第一通信节点实现高谱效的信息传输。
可以理解的是，导频可以称为导频信号，或参考信号(Reference Signal，RS)，或解调参考信号，或前导，而且从形式上看，导频通常是一条序列或一串符号，所以导频也称为导频序列。
一可行的实施方式，每个极稀疏导频均包括第二数量的非零值符号，其中，第二数量大于0且小于5。第二数量的非零值符号承载在时频域上相邻的第三数量的资源单元(Resource Elements，REs)上，或者承载在按时间先后顺序的第三数量的符号上，或者承载在频域相邻子载波上的第三数量的资源单元上，其中，第三数量与第二数量相等(即第三数量大于0且小于5)。可以理解的是，每个极稀疏导频只有一个参考信号单元(Reference Signal Element，RSE)，该参考信号单元包含的非零值符号的符号数量为第二数量。可以理解的是，对于OFDM而言，一个资源单元(RE)可以承载一个符号。另外，若第二数量的非零值符号承载在相邻的RE上，相当于该第二数量的符号承载在相邻的时频资源上。或者，对于OFDM而言，若第二数量的非零值符号承载在相邻的时频资源上，相当于该第二数量的非零值符号承载在相邻的RE上。
一可行的实施方式，第二数量的取值为1，即一个非零值符号；或者，第二数量的取值为2，第二数量的非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值为[a1,a2]或者[b1,b2]，其中，[a1,a2]和[b1,b2]正交，即a1′*b1+a2′*b2＝0；或者，第二数量的取值为2，第二数量的非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值至少包括[p1,p2]＝[1,1]；[p1,p2]＝[1,-1]；[p1,p2]＝[1,j]；[p1,p2]＝[1,-j]等情况；或者，第二数量的取值为4，第二数量的非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值为[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]或者[d1,d2,d3,d4]，其中，[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]和[d1,d2,d3,d4]相互正交，即[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]和[d1,d2,d3,d4]中的任意两个相互正交，具体地，[a1,a2,a3,a4]与[b1,b2,b3,b4]相互正交，即a1′*b1+a2′*b2+a3′*b3+a4′*b4＝0；[a1,a2,a3,a4]与[c1,c2,c3,c4]相互正交，即a1′*c1+a2′*c2+a3′*c3+a4′*c4＝0，等等，在此不再赘述。
一可行的实施方式，第二数量的取值为4，第二数量的非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值至少包括如表1中的各种情况，其中，表1中的不同序号对应于不同的情况。
一可行的实施方式，每个极稀疏导频的符号长度均大于24，即每个极稀疏导频的非零值符号与零值符号的总数量大于24，在此不作具体限制。
一可行的实施方式，当第一数量的取值大于或者等于2时，第一数量的极稀疏导频之间相互独立，即第一数量的极稀疏导频之间不关联或者不相关，其中，第一数量的取值可以大于或者等于1，即第一数量的取值可以是1、2或者其他取值，在此不作具体限制。其中，将一次传输中包含2个或多个导频，并且导频之间无关联或者相互独立的技术称为独立多导频技术，该多个相互独立的导频称为独立多导频。如图3所示，在一次传输中包含w个极稀疏导频，其中，w个极稀疏导频分别表示为P1、P2、......、Pw，w可以为大于2的正整数，并且数据包中包含该w个极稀疏导频的信息，例如，该数据包中包含该w个极稀疏导频的索引号(即极稀疏导频在预设导频集合中的索引号)，这样，一旦某个终端的数据包译码成功，就可以确定该终端在此次信息传输中所使用的所有极稀疏导频的信息，从而可以对导频信号进行干扰消除。
可以理解的是，第一通信节点与第二通信节点的信息传输可以采用独立多导频技术，这样，在相同的导频开销下，不同的第一通信节点的独立多导频同时碰撞的概率会比传统单导频碰撞的概率小。因此，在竞争式无连接状态(或者竞争式免调度方式)的传输场景下，可以采用独立多导频技术以支持更多的第一通信节点进行信息传输。进一步地，可以结合独立多导频技术和极稀疏导频技术，即采用多个相互独立的极稀疏导频，进一步减少导频碰撞的概率，进一步提升接入的第一通信节点的数量。
在一实施例中，当第二通信节点为基站时，通过基于迭代的接收机，基站每轮都可以通过不碰撞(即相互独立)的多个极稀疏导频解调出对应的第一通信节点，然后将重构出该第一通信节点的数据包和极稀疏导频，并从接收信号中将该第一通信节点对应的数据包和极稀疏导频消除，如此迭代直至解调出所有可解的第一通信节点，以减少导频碰撞的概率，进一步提升接入的第一通信节点的数量。
在一实施例中，当第一数量的取值大于或者等于2时，第一数量的极稀疏导频可以根据数据包中的信息而确定。
在另一实施例中，当第一数量的取值大于或者等于2时，第一数量的极稀疏导频可以根据数据包中的一个或者多个比特信息而确定，比如，一个极稀疏导频可以根据数据包中的一个比特信息确定；又如，一个极稀疏导频可以根据数据包中的两个比特信息确定；再如，两个极稀疏导频均根据数据包中的多个比特 信息确定，等等，本申请实施例不对第一数量和比特信息的数量作限制。
在另一实施例中，当第一数量的取值大于或者等于2时，每个极稀疏导频均根据数据包中的第四数量的比特信息从预设导频集合中确定得到，其中，预设导频集合包括第五数量的导频，第四数量与第五数量成对数函数关系，对数函数为以2作为底数的对数函数。比如，假设第五数量是D个，则第四数量为log2(D)个，在此不作具体限制。可以理解的是，第五数量可以是64、128或者更多，在此不做具体限制。
一可行的实施方式，当调制符号根据第一星座图模型而调制得到，a1,a2,…,aN1均可由如下公式表示，即
an＝(2n-1+Δ)d；
其中，n的取值包括1,2,...,N1，即an可以是a1，也可以是a2，还可以是a3，或者aN1等等。d为正实数，Δ为大于或等于0的实数，使得a1,a2,…,aN1构成等差数列。
进一步地，当Δ的取值为0，且d的取值为1，使得an满足an＝2n-1；当Δ的取值为1，且d的取值为1/2，使得an满足an＝n；当Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
或者，Δ的取值可以为此时，当d的取值为1，使得an满足当d的取值为1/2，使得an满足
一可行的实施方式，当调制符号根据第二星座图模型而调制得到，a1,a2,…,aN2均可由如下公式表示：
an＝(2n-1+Δ)d；
b1,b2,…,bN2均可由如下公式表示：
bn＝an+β；
其中，n的取值包括1,2,...,N2，即an可以是a1，也可以是a2，还可以是a3，或者aN2等等。同样地，bn可以是b1，也可以是b2，还可以是b3或者bN2等等。d为正实数，Δ和β均为大于或等于0的实数，使得a1,a2,…,aN2构成等差数列，b1,b2,…,bN2构成等差数列。
进一步地，当Δ的取值为0，且d的取值为1，使得an满足an＝2n-1；当Δ的取值为1，且d的取值为1/2，使得an满足an＝n；当Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
或者，Δ的取值可以为此时，当d的取值为1，使得an满足当d的取值为1/2，使得an满足
一可行的实施方式，当调制符号根据第三星座图模型而调制得到，a1,a2,…,aN3均可由如下公式表示：
an＝(2n-1+Δ)d；
b1,b2,…,bN3均可由如下公式表示：
bn＝an+β；
其中，n的取值包括1,2,...,N3，即an可以是a1，也可以是a2，还可以是a3，或者aN3等等。同样地，bn可以是b1，也可以是b2，还可以是b3，或者bN3等等。d为正实数，Δ和β均为大于或等于0的实数，使得a1,a2,…,aN3构成等差数列，b1,b2,…,bN3构成等差数列。
进一步地，当Δ的取值为0，且d的取值为1，使得an满足an＝2n-1；当Δ的取值为1，且d的取值为1/2，使得an满足an＝n；当Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
或者，Δ的取值可以为此时，当d的取值为1，使得an满足当d的取值为1/2，使得an满足
一可行的实施方式，当调制符号根据第二星座图模型或者第三星座图模型而调制得到，β等于0。
一可行的实施方式，当调制符号根据第三星座图模型而调制得到，β大于0。
一可行的实施方式，当调制符号根据第一星座图模型而调制得到，d的取值为使采用第一星座图模型进行调制得到的调制符号的平均功率等于1的数值，即d的取值使得第一星座图模型中的星座点的模平方的均值是1，也即d的取值使得第一星座图模型的平均功率是1，在此不作具体限制。
一可行的实施方式，当调制符号根据第二星座图模型而调制得到，d的取值为使采用第二星座图模型进行调制得到的调制符号的平均功率等于1的数值，即d的取值使得第二星座图模型中的星座点的模平方的均值是1，也即d的取值使得第二星座图模型的平均功率是1，在此不作具体限制。
一可行的实施方式，当调制符号根据第三星座图模型而调制得到，d的取值为使采用第三星座图模型进行调制得到的调制符号的平均功率等于1的数值，即d的取值使得第三星座图模型中的星座点的模平方的均值是1，也即d的取值使得第三星座图模型的平均功率是1，在此不作具体限制。
一可行的实施方式，θ的取值可以为0；或者，θ的取值可以为π/4，即满足公式θ＝π/4；或者，θ的取值满足公式θ＝π/8，在此不作具体限制。
如图5所示，图5是本申请一实施例提供的第一星座图模型和第二星座图模型的示意图，在一实施例中，第二星座图模型可以为如图5中左下角坐标系所对应的十字形星座图或者右下角坐标系所对应的十字 形星座图，其中，十字形星座图为一半数量的星座点位于一条经过零点(即原点)的直线上，另外一半数量的星座点位于另一条经过零点(即原点)的直线上，且这两条直线互相垂直的星座图，十字形星座图具有高谱效以及几何形状简单等优点。具体地，图5中所示的十字形星座图是二维信号平面中的星座图，这些十字形星座图均包括16个星座点，每个星座点对应一个调制符号，每个调制符号可以承载4个比特信息，即4个比特信息会映射(即调制)为一个调制符号。其中，图5中左下角的十字形星座图中的星座点分布在x轴(即I路)和y轴(即Q路)；图5中右下角的十字形星座图中的星座点分别分布在经过原点的45°方向的直线上和经过原点的135°方向直线上。图5中右下角的十字形星座图可以通过图5中左下角的十字形星座图旋转45°形成。可以理解的是，第二星座图模型还可以是除了如图5中左下角和右下角所示的十字形星座图以外的其它十字形星座图，本申请实施例并不对第二星座图模型的形式作具体限制。
如图5所示，在一实施例中，第一星座图模型可以为如图5中左上角坐标系所对应的PAM(Pulse Amplitude Modulation,脉冲幅度调制)星座图或者右上角坐标系所对应的PAM星座图，即线性星座图，其中，PAM星座图的全部星座点在一条经过零点(即原点)的直线上。可以理解的是，第一星座图模型还可以是除了如图5中左上角和右上角所示的PAM星座图以外的其它PAM星座图，本申请实施例并不对第一星座图模型的形式作具体限制。
详细来说，每个调制符号(也即每个星座点)可以承载多个比特信息，也就是可以实现高阶调制的效果，从而实现高谱效。在一实施例中，每个调制符号可以承载4个比特信息，也就是4个比特信息会映射(即调制)为一个调制符号；在另一实施例中，每个调制符号可以承载5个比特，也就是5个比特信息会映射(即调制)为一个调制符号。
可以理解的是，调制符号对应的线性星座图(即PAM星座图)、十字形星座图和8臂星座图都具备简单的几何形状的优点，即使接收侧(即第二通信节点)接收到的调制符号经过了信道的旋转缩放，调制符号所对应的星座图也仅是一个经过旋转缩放的线性星座图、十字形星座图或者8臂星座图，形成的几何形状依然比较简单。
由于线性星座图是相对较简单的星座图，通常会比十字形星座图的处理更简单，因此，下面以稍微复杂的十字形星座图为例来进行说明。
如图6所示，图6是在经过信道旋转缩放前后的十字形星座图的示意图，其中，图6左边坐标系是发射调制符号s(即第一通信节点处的未经过信道旋转缩放的调制符号s)对应的十字形星座图的示意图，图6中间坐标系是第二通信节点所接收到的经过旋转缩放后的调制符号h*s(即h乘以s，也可以表示为h·s或hs)对应的十字形星座图的示意图，其中复数h是旋转缩放量。
值得注意的是，图6中间坐标系是没有AWGN的接收调制符号(即第二通信节点所接收到的经过旋转缩放后的调制符号)对应的十字形星座图的示意图，图6右边坐标系是有AWGN的接收调制符号(y＝h*s+n)对应的十字形星座图的示意图。可以理解的是，图6右边所示的星座图可以通过在图6中间所示的星座图中的星座点加上AWGN对应的复数而形成，即带AWGN的接收调制符号(h*s+n)对应的星座点会在图6中间所示的星座图对应的星座点(h*s)的周围按照AWGN的概率密度进行分布。在图6右边的星座图中，星座点的颜色从中间到边缘由深到浅，该星座点是对应的调制符号被AWGN影响而形成的点的集合。由图6右边的星座图也可以看出，即使存在AWGN，接收调制符号对应的十字形星座图的大体形状依旧为十字形。因此，接收侧(即第二通信节点)可以利用如图6右边所示的十字形星座图的几何形状来估计该星座图所受到的旋转缩放量，也即将h估计出来。
下面对一种旋转缩放量估计方法进行详细地描述：
如图7所示，首先，将二维平面(即二维信号平面)分为4个分区，可以采用两种典型的方法进行分区。具体地，如图7中左边坐标系所示，在第一种分区方法中，以4个象限为4分区，也即以x轴和y轴为分区线，其中，采用斜线填充的区域是分区1，采用细点填充的区域是分区2，采用竖线填充的区域是分区3，采用砖块状填充的区域是分区4；如图7中右边坐标系所示，第二种分区方法中的4个分区由第一种分区方法中的4个分区旋转45°形成，也即以由原点发出的45°射线至由原点发出的135°射线所围成的区域为分区1，其中，分区1由斜线填充；以由原点发出的135°射线至由原点发出的225°射线的区域为分区2，其中，分区2由细点填充；以由原点发出的225°射线至由原点发出的315°射线的区域为分区3，其中，分区3由竖线填充；以由原点发出的315°射线至由原点发出的45°射线的区域为分区4，其中，分区4由砖块状填充。采用如图7所示的两种分区方法对一个星座点的所属分区进行判断，仅需要对星座点坐标进行一些简单的加减法即能够判断出该星座点所属的具体分区，而无需通过复杂的乘法运算来判断，可见，判断方法简便。除了可以采用上述两种分区方法之外，还可以选择其它分区方法将二维平面划分为4个分区，本申请实施例不作具体限制。
如图6至图8所示，在接收侧(即第二通信节点)将二维信号平面分为4个分区后，分别将每个分区里面的星座点(即每个星座点对应的调制符号)加起来，再除以该分区内星座点的数量(即调制符号的数量)，然后可以计算得出一个坐标，该坐标为该分区星座点的中心。图6中间所示的十字形星座图是将图 6左边所示的十字形星座图旋转缩放后所形成的星座图的示意图。以如图7中左边坐标系所示的分区为例，经过分区后，所有星座点被分成4部分，如图8中间坐标系和右边坐标系所示，将分区1里面的星座点加起来，再除以该分区内星座点的数量，可以得到分区1的星座点中心c1，即图8右边坐标系中所示的三角形的所处位置；同样地，将分区2里面的星座点加起来，再除以该分区内星座点的数量，可以得到分区2的星座点中心c2，即图8右边坐标系中所示的四边形的所处位置；将分区3里面的星座点加起来，再除以该分区内星座点的数量，可以得到分区3的星座点中心c3，即图8右边坐标系中所示的五角星的所处位置；将分区4里面的星座点加起来，再除以该分区内星座点的数量，可以得到分区4的星座点中心c4，即图8右边坐标系中所示的六角星的所处位置。
如图7和图8所示，根据所有分区的星座点中心可以得到整个星座图的旋转缩放量，具体地，以图8所示的一种分区方法为例，假设计算得到的4个分区的星座点中心分别为c1、c2、c3、c4，通过将分区2的星座点中心c2顺时针旋转90°得到c2′,也即c2′＝c2*(-j)；通过将分区3的星座点中心c3顺时针旋转180°得到c3′,也即c3′＝-c3；通过将分区4的星座点中心c4逆时针旋转90°得到c4′，也即c4′＝c4*j；然后根据c1、c2′、c3′、c4′可以对整个星座图所受到的旋转缩放量c进行估计，其中，该旋转缩放量c可以通过下式(1)表示，即：
c＝(c1+c2′+c3′+c4′)/4      (1)
在存在AWGN的情况下，尤其是在有些调制符号受到的AWGN较大的情况下，部分调制符号可能会发生越区切换的现象。为了更准确地估计旋转缩放量，通常需要用到图7中所示的两种分区方法，针对两种分区方法分别按照上述估计方法计算出星座图的两个旋转缩放量，然后将两个旋转缩放量中模值较大的一个模值作为该星座图的旋转缩放量。
如图5所示，对于如图5中左上角或右上角所示的线性星座图，可以只用2个分区去计算星座图的旋转缩放量，例如，接收侧(即第二通信节点)通过y轴将二维信号平面分为2个分区后，分别将每个分区里面的星座点(即每个星座点对应的调制符号)加起来，再除以该分区内星座点的数量(即调制符号的数量)，然后可以计算得到一个星座点，即该分区的星座点中心，再通过所有分区的星座点中心求得整个星座图的旋转缩放量，假设x轴右边(也即x＞＝0)为分区1，分区1的星座点中心为c1；x轴左边(也即x＜0)为分区2，分区2的星座点中心为c2，通过将分区2的星座点中心c2顺时针旋转90°得到c2′,也即c2′＝-c2；然后根据c1、c2′可以对整个星座图所受到的旋转缩放量c进行估计，其中，该旋转缩放量c可以通过下式(2)表示，即：
c＝(c1+c2′)/2      (2)
为了更准确地估计旋转缩放量，可以采用下面四种不同的2分区方法进行分区，在第一种分区方法中，以x轴为分区线，将二维信号平面分成2个分区；在第二种方法中，以y轴为分区线，将二维信号平面分成2个分区；在第三种分区方法中，以过原点的45°直线为分区线，将二维信号平面分成2个分区；在第四种分区方法中，以过原点的135°直线为分区线，将二维信号平面分成2个分区。
基于上述四种不同的2分区方法来计算得到四个旋转缩放量，然后将四个旋转缩放量中模值最大的一个作为该星座图的旋转缩放量，接收侧(即第二通信节点)估计到星座图所受到的旋转缩放量后，可以将星座图所经历的旋转缩放量均衡掉，得到没有畸变且只受AWGN影响的星座图。
因此，通过第一星座图模型、第二星座图模型或者第三星座图模型得到的多个调制符号能够形成一个几何形状简单的星座图，且在调制符号受到信道干扰发生旋转缩放的畸变后，所形成的星座图仍然呈现为一个简单的几何形状图形，所以，本申请的信息传输方法能够仅通过星座图所具备的形状特点进行补偿，从而无需增加导频开销以提高解调性能，保证高频率谱效。
针对上述实施例所提供的信息传输方法，下面以具体的示例进行详细的描述：
示例一：
参照图6，图6中的十字形星座图可以分成两部分，每一部分的星座点均处于一条经过零点(即原点)的直线上，例如，如图6左边坐标系所对应的十字形星座图，其中一半数量的星座点落在x轴的直线上，另外一半数量的星座点落在y轴的直线上。又如，图6中间坐标系所对应的十字形星座图中一半数量的星座点落在与x轴的正半轴成45°角的经过零点(即原点)的直线上，另一半数量的星座点落在与x轴的正半轴成135°角的经过零点(即原点)的直线上。
具体地，若一条经过原点的直线上的相邻星座点之间的距离是相等的，且相邻两点之间的距离设为2d，则在距离原点最近的4个星座点中，相邻的星座点之间的距离只有即是说，在距离原点最近的4个星座点中，相邻的星座点之间的距离会比处于同一直线上的相邻星座点的距离要小，即距离原点最近的4个星座点分布更密集，因此，星座图更容易受到AWGN干扰，这会导致解调性能下降。
另外，该十字形星座图可以分成4个部分，例如，如图6左边坐标系所对应的十字形星座图，可以按照图7右边坐标系所示的分区形式，将如图6左边坐标系所对应的十字形星座图中的星座点分成4个部分， 其中，第一部分的星座点为x轴上大于0的星座点，即落在x轴正半轴上的星座点，其中，第一部分对应于图7中右边坐标系的分区4；第二部分的星座点为x轴上小于0的星座点，即落在x轴负半轴上的星座点，其中，第二部分对应于图7中右边坐标系的分区2；第三部分的星座点为y轴上大于0的星座点，即落在y轴正半轴上的星座点，其中，第三部分对应于图7中右边坐标系的分区1；第四部分的星座点为y轴上小于0的星座点，即落在y轴负半轴上的星座点，其中，第四部分对应于图7中右边坐标系的分区3。
进一步地，为了避免上述问题(即在距离原点最近的4个星座点中，相邻的星座点之间的距离会比处于同一直线上的相邻星座点的距离要小)，可以对十字形星座图的4个部分的星座点分别加上一个大于0的偏移量Δ，使得在距离原点最近的4个星座点中，相邻的星座点之间的距离大于或等于同一直线上的相邻星座点的距离，即4个部分的星座点均向远离于原点的方向偏移，避免距离原点最近的4个星座点分布密集，以提高解调性能。
或者，该十字形星座图的4个部分的星座点可以不叠加一个大于0的偏移量Δ，即十字形星座图各个部分的星座点的偏移量为0，而在所有部分的星座点的偏移量均为0的十字形星座图中，同一直线上的相邻星座点之间的距离相等，因此，星座图的平均功率更低。具体地，在an＝(2n-1+Δ)d的情况下，Δ的取值可以为此时，在距离原点最近的4个星座点中，相邻的星座点之间的距离均为2d，且同一直线上的相邻星座点之间的距离也为2d，即在距离原点最近的4个星座点中，相邻的星座点之间的距离与各个部分中同一直线上相邻星座点之间的距离相等。
可以理解的是，为了提高该十字形星座图中距离原点最近的4个星座点的传输性能，可以增大在距离原点最近的4个星座点中相邻的星座点之间的距离，使得所形成的十字形星座图向外扩展，避免距离原点最近的4个星座点分布过于密集，从而降低AWGN对星座点(即调制符号)的影响，提高十字形星座图的鲁棒性。例如，可以令an＝nd，d为一个正实数，使得在距离原点最近的4个星座点中，相邻的星座点之间的距离为而各个部分的同一直线上的相邻星座点之间的距离为d，即在距离原点最近的4个星座点中，相邻的星座点之间的距离大于各个部分的同一直线上相邻星座点之间的距离，即4个部分的星座点均向远离于原点的方向偏移，以降低AWGN对星座点(即调制符号)的影响，从而提高第二通信节点的解调性能。但是，由于对十字形星座图的4个部分的星座点分别加上一个大于0的偏移量Δ，
会使得星座图的平均功率更大，因此，有些场景也会采用偏移量Δ为0的十字形星座图，在此不作具体限制。
示例二：
以Δ的取值满足公式为例，参照图9，当第二星座图模型为十字形星座图，在M2的取值为1的情况下，即利用第二星座图模型可以对数据包中的3个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝2，因此，第二星座图模型为包括8个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为0，即十字形星座图的旋转量为0，即是说，十字形星座图中的星座点分布在x轴(即I路)和y轴(即Q路)上。接着，由可知， 因此，星座点a1ej0(即s1)与星座点b1ej(0+π/2)(即s3)之间的距离为2，即其中，ej0＝cos0+jsin0＝1，ej(0+π/2)＝cos(0+π/2)+jsin(0+π/2)＝j，从而可以确定出同一直线上的相邻星座点之间的距离为2。同样地，星座点a1ej0(即s1)与星座点a2ej0(即s2)之间的距离为2，即离原点最近的4个星座点的相邻点之间的距离为2，因此，星座点s1与星座点s3之间的距离，跟星座点s1与星座点s2之间的距离相等。因此，距离原点最近的4个星座点的相邻点之间的距离，与同一直线上的相邻星座点之间的距离相等。其中，星座点s1的坐标为星座点s2的坐标为星座点s3的坐标为可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，在Δ取值等于的情况下，即如图9所示的8个星座点的二维坐标具体如下：



另外，该8个星座点的二维坐标可以包括如下：
(1+Δ,0),(3+Δ,0),
(0,1+Δ),(0,3+Δ),
(-(1+Δ),0),(-(3+Δ),0),
(0,-(1+Δ)),(0,-(3+Δ))
其中，Δ可以用有限位小数表示，例如，在此不作具体限制。
当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的8个星座点的二维坐标统一乘以一个归一化因子。
示例三：
以Δ的取值满足公式为例，参照图10，当第二星座图模型为十字形星座图，在M2的取值为1的情况下，即利用第二星座图模型可以对数据包中的3个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝2，因此，第二星座图模型为包括8个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为π/4，即十字形星座图的旋转量为π/4，即是说，十字形星座图中一半数量的星座点落在与x轴的正半轴成45°角的经过零点(即原点)的直线上，另一半数量的星座点落在与x轴的正半轴成135°角的经过零点(即原点)的直线上。
接着，由可知，则 而因此，与a1ejπ/4对应的星座点s1的坐标是(1,1)，与a2ejπ/4对应的星座点s2的坐标是可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，在Δ取值等于的情况下，即如图10所示的8个星座点的二维坐标具体如下：



可以理解的是，Δ的取值还可以满足公式其中，Δ可以用有限位小数表示，例如，Δ＝1-0.707＝0.293，在此不作具体限制。
当Δ的取值满足公式该8个星座点的二维坐标可以包括如下：



当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的8个星座点的二维坐标统一乘以一个归一化因子。
示例四：
参照图11，当第二星座图模型为十字形星座图，在M2取值为2的情况下，即利用第二星座图模型可以对数据包中的4个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝4，因此，第二星座图模型为包括16个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为0，即十字形星座图的旋转量为0，即是说，十字形星座图中的星座点分布在x轴(即I路)和y轴(即Q路)上。接着，由可知，由于ej0＝cos 0+j sin 0＝1，因此从而可以确定出a1ej0与a2ej0的距离是2，即同一直 线上的相邻星座点之间的距离为2。另外，根据bn＝an+β，在β取值为0的情况下可以得到bn＝an，即bn与an相等，由于ej(0+π/2)＝cos(0+π/2)+j sin(0+π/2)＝j，因此，从而可以确定出a1ej0与b1ej(0+π/2)的距离是2，即距离原点最近的4个星座点的相邻点之间的距离为2。因此，距离原点最近的4个星座点的相邻点之间的距离，与同一直线上的相邻星座点之间的距离相等。参照上述示例三中的计算方式可以确定星座点s1的坐标是星座点s2的坐标是其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，在Δ取值等于的情况下，即如图11所示的16个星座点的二维坐标具体如下：



另外，该16个星座点的二维坐标可以包括如下：
(1+Δ,0),(3+Δ,0),(5+Δ,0),(7+Δ,0),
(-(1+Δ),0),(-(3+Δ),0),(-(5+Δ),0),(-(7+Δ),0),
(0,1+Δ),(0,3+Δ),(0,5+Δ),(0,7+Δ),
(0,-(1+Δ)),(0,-(3+Δ)),(0,-(5+Δ)),(0,-(7+Δ))
其中，Δ可以用有限位小数表示，例如，在此不作具体限制。
当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的16个星座点的二维坐标统一乘以一个归一化因子。
示例五：
参照图12，当第二星座图模型为十字形星座图，在M2的取值为2的情况下，即利用第二星座图模型可以对数据包中的4个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝4，因此，第二星座图模型为包括16个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为π/4，即十字形星座图的旋转量为π/4，即十字形星座图的旋转量为π/4，即是说，十字形星座图中一半数量的星座点落在与x轴的正半轴成45°角的经过零点(即原点)的直线上，另一半数量的星座点落在与x轴的正半轴成135°角的经过零点(即原点)的直线上。接着，由an＝(2n-1+Δ)d可知，an＝2n-1+Δ，若也即由此得到 因此 所以与a1ejπ/4对应的星座点s1的坐标是(1,1)，与a2ejπ/4对应的星座点s2的坐标是可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，在Δ取值等于的情况下，即如图12所示的16个星座点的二维坐标具体下：



可以理解的是，Δ的取值还可以满足公式其中，Δ可以用有限位小数表示，例如，Δ＝1-0.707＝0.293，在此不作具体限制。
当Δ的取值满足公式该16个星座点的二维坐标可以包括如下：







由于从而可以确定出a1ejπ/4与a2ejπ/4的距离是2，即同一直线上的相邻星座点之间的距离为2。另外，由于因此，从而可以确定出a1ejπ/4与a1ej(π/4+π/2)的距离是2，即距离原点最近的4个星座点的相邻点之间的距离为2。因此，距离原点最近的4个星座点的相邻点之间的距离，与同一直线上的相邻星座点之间的距离相等。
当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的16个星座点的二维坐标统一乘以一个归一化因子。
示例六：
以Δ的取值满足公式Δ＝0为例，参照图13，当第二星座图模型为十字形星座图，在M2的取值为1的情况下，即利用第二星座图模型可以对数据包中的3个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝2，因此，第二星座图模型为包括8个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为0，即十字形星座图的旋转量为0，即是说，十字形星座图中的星座点分布在x轴(即I路)和y轴(即Q路)上。接着，由an＝(2n-1+Δ)d可知，an＝2n-1，因此可以确定a1＝1，a2＝3。由于ej0＝cos 0+j sin0＝1，因此，a1ej0＝1，a2ej0＝3，因此，a1ej0对应的星座点s1的坐标为(1,0)，a2ej0对应的星座点s2的坐标为(3,0)。可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，在Δ取值等于0的情况下，即如图13所示的8个星座点的二维坐标具体下：
(1,0),(3,0),
(0,1),(0,3),
(-1,0),(-3,0),
(0,-1),(0,-3)
当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的8个星座点的二维坐标统一乘以一个归一化因子(比如1/sqrt(10))。
示例七：
以Δ的取值满足公式Δ＝0为例，参照图14，当第二星座图模型为十字形星座图，在M2的取值为1的情况下，即利用第二星座图模型可以对数据包中的3个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝2，因此，第二星座图模型为包括8个星座点的十字形星座图，其中，d的取值为1， 即十字形星座图的缩放量为1；θ的取值为π/4，即十字形星座图的旋转量为π/4，即是说，十字形星座图中一半数量的星座点落在与x轴的正半轴成45°角的经过零点(即原点)的直线上，另一半数量的星座点落在与x轴的正半轴成135°角的经过零点(即原点)的直线上。接着，由an＝(2n-1+Δ)d可知，an＝2n-1，由此可得a1＝1，a2＝3，那么因此，与a1ejπ/4对应的星座点s1的坐标是与a2ejπ/4对应的星座点s2的坐标是可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，该8个星座点的二维坐标可以包括如下：



当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的8个星座点的二维坐标统一乘以一个归一化因子(比如1/sqrt(10))。
示例八：
参照图15，当第二星座图模型为十字形星座图，在M2的取值为2的情况下，即利用第二星座图模型可以对数据包中的4个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝4，因此，第二星座图模型为包括16个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为0，即十字形星座图的旋转量为0，即是说，十字形星座图中的星座点分布在x轴(即I路)和y轴(即Q路)上。
接着，由an＝(2n-1+Δ)d，可知，an＝2n-1+Δ，若Δ＝0，也即an＝2n-1，由此得到a1＝1，a2＝3。由于ej0＝cos0+jsin0＝1，从而可以确定出a1ej0＝ej0＝1，a2ej0＝3ej0＝3，因此，与a1ej0对应的星座点s1的坐标是(1,0)，与a2ej0对应的星座点s2的坐标是(3,0)。可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，该16个星座点的二维坐标可以包括如下：
(1,0),(3,0),(5,0),(7,0),
(-1,0),(-3,0),(-5,0),(-7,0),
(0,1),(0,3),(0,5),(0,7),
(0,-1),(0,-3),(0,-5),(0,-7)
当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的16个星座点的二维坐标统一乘以一个归一化因子(比如1/sqrt(84))。
示例九：
参照图16，当第二星座图模型为十字形星座图，在M2的取值为2的情况下，即利用第二星座图模型可以对数据包中的4个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝4，因此，第二星座图模型为包括16个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为π/4，即十字形星座图的旋转量为π/4，即十字形星座图的旋转量为π/4，即是说，十字形星座图中一半数量的星座点落在与x轴的正半轴成45°角的经过零点(即原点)的直线上，另一半数量的星座点落在与x轴的正半轴成135°角的经过零点(即原点)的直线上。接着，由an＝(2n-1+Δ)d，可知，an＝2n-1+Δ，若Δ＝0，也即an＝2n-1。由此得到a1＝1，a2＝3。那么 因此，与a1ejπ/4对应的星座点s1的坐标是与a2ejπ/4对应的星座点s2的坐标是可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形 式计算得到，在此不再赘述。
因此，该16个星座点的二维坐标可以包括如下：



当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的16个星座点的二维坐标统一乘以一个归一化因子(比如1/sqrt(84))。
示例十：
以Δ的取值满足公式Δ＝0为例，参照图17，当第二星座图模型为十字形星座图，在M2的取值为1的情况下，即利用第二星座图模型可以对数据包中的3个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝2，因此，第二星座图模型为包括8个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为0，即十字形星座图的旋转量为0，即是说，十字形星座图中的星座点分布在x轴(即I路)和y轴(即Q路)上。接着，由an＝2n-1可知，a1＝1，a2＝2，由于ej0＝cos0+jsin0＝1，因此可以确定出a1ej0＝1，a2ej0＝2，因此，a1ej0对应的星座点s1的坐标为(1,0)，a2ej0对应的星座点s2的坐标为(2,0)。可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，该8个星座点的二维坐标可以包括如下：
(1,0),(2,0),
(0,1),(0,2),
(-1,0),(-2,0),
(0,-1),(0,-2)
当需要对该第二星座图模型进行功率归一化时，可以将该第二星座图模型对应的8个星座点的二维坐标统一乘以一个归一化因子(比如1/sqrt(5))。
示例十一：
以Δ的取值满足公式Δ＝0为例，参照图18，当第二星座图模型为十字形星座图，在M2的取值为1的情况下，即利用第二星座图模型可以对数据包中的3个比特信息进行调制，得到一个调制符号，由于N2＝2M2，对应得出N2＝2，因此，第二星座图模型为包括8个星座点的十字形星座图，其中，d的取值为1，即十字形星座图的缩放量为1；θ的取值为π/4，即十字形星座图的旋转量为π/4，即是说，十字形星座图中一半数量的星座点落在与x轴的正半轴成45°角的经过零点(即原点)的直线上，另一半数量的星座点落在与x轴的正半轴成135°角的经过零点(即原点)的直线上。接着，由an＝2n-1可知，a1＝1，a2＝2，那么因此，与a1ejπ/4对应的星座点s1的坐标是与a2ejπ/4对应的星座点s2的坐标是可以理解的是，其它星座点的二维坐标可以采用第二星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，第二星座图模型中的8个星座点所对应的复数可以如下表示：



因此，第二星座图模型可以为如图18所示的包括8个星座点的十字形星座图，即是说，θ取值为π/4所对应的星座图可以通过θ取值为零所对应的星座图旋转45°得到。其中，该8个星座点的二维坐标可以包括如下：



示例十二：
参照图19，当第一星座图模型为PAM星座图，在M1的取值为2的情况下，即利用第一星座图模型可以对数据包中的3个比特信息进行调制，得到一个调制符号，由于N＝2M，对应得出N1＝4，因此，第一星座图模型为包括8个星座点的PAM星座图，其中，d的取值为1，即PAM星座图的缩放量为1；θ的取值为0，即PAM星座图的旋转量为0，即是说，PAM星座图中的星座点分布在x轴(即I路)。接着，由an＝(2n-1+Δ)d可知，若Δ＝0，也即an＝2n-1。由此得到a1＝1，a2＝3。那么a1ej0＝ej0＝1，a2ej0＝3ej0＝3，因此，与a1ej0对应的星座点s1的坐标是(1,0)，与a2ej0对应的星座点s2的坐标是(3,0)。可以理解的是，其它星座点的二维坐标可以采用第一星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
由此可知，该8个星座点的二维坐标可以包括如下：
(1,0),(3,0),(5,0),(7,0),
(-1,0),(-3,0),(-5,0),(-7,0)
当需要对该第一星座图模型进行功率归一化时，可以将该第一星座图模型对应的8个星座点的二维坐标统一乘以一个归一化因子(比如1/sqrt(84))。
示例十三：
以Δ的取值满足公式Δ＝0为例，参照图20，当第一星座图模型为PAM星座图，在M1的取值为1的情况下，即利用第一星座图模型可以对数据包中的2个比特信息进行调制，得到一个调制符号，由于N1＝2M1，对应得出N1＝2，因此，第一星座图模型为包括4个星座点的PAM星座图，其中，d的取值为1，即PAM星座图的缩放量为1；θ的取值为0，即PAM星座图的旋转量为0，即是说，PAM星座图中的星座点分布在x轴(即I路)上。接着，由an＝2n-1可知，a2＝2。那么a1ej0＝ej0＝1，a2ej0＝2ej0＝2，因此，与a1ej0对应的星座点s1的坐标是(1,0)，与a2ej0对应的星座点s2的坐标是(2,0)。可以理解的是，其它星座点的二维坐标可以采用第一星座图模型中星座点所对应的复数形式计算得到，在此不再赘述。
因此，该4个星座点的二维坐标可以包括如下：
(1,0),(2,0),
(-1,0),(-2,0)
当需要对该第一星座图模型进行功率归一化时，可以将该第一星座图模型对应的4个星座点的二维坐标统一乘以一个归一化因子(比如1/sqrt(5))。
可以理解的是，第二星座图模型中各个星座点所对应的坐标可以包括如下：
(a1cosθ,a1sinθ),(a2cosθ,a2sinθ),…,(ancosθ,ansinθ)，

(a1cos(θ+π),a1sin(θ+π)),(a2cos(θ+π),a2sin(θ+π)),…,(ancos(θ+π),ansin(θ+π)),
(a1cos(θ+3π/2),a1sin(θ+3π/2)),(a2cos(θ+3π/2),a2sin(θ+3π/2)),…,(ancos(θ
+3π/2),ansin(θ+3π/2))
根据三角函数公式，第二星座图模型中各个星座点所对应的坐标还可以表示如下：
(a1cosθ,a1sinθ),(a2cosθ,a2sinθ),…,(ancosθ,ansinθ)
(-a1sinθ,a1cosθ),(-a2sinθ,a2cosθ),…,(-ansinθ,ancosθ)
(-a1cosθ,-a1sinθ),(-a2cosθ,-a2sinθ),…,(-ancosθ,-ansinθ)
(a1sinθ,-a1cosθ),(a2sinθ,-a2cosθ),…,(ansinθ,-ancosθ)
其中，三角函数公式包括如下：
cos(θ+π/2)＝-sinθ
sin(θ+π/2)＝cosθ
cos(θ+π)＝-cosθ
sin(θ+π)＝-sinθ
cos(θ+3π/2)＝sinθ
sin(θ+3π/2)＝-cosθ
值得注意的是，在θ取值为0的情况下，即十字形星座图的旋转量为0，十字形星座图中的星座点分布在x轴和y轴上，因此，第二星座图模型中各个星座点所对应的坐标可以包括如下：
(a1,0),(a2,0),…,(an,0),
(0,a1),(0,a2),…,(0,an),
(-a1,0),(-a2,0),…,(-an,0),
(0,-a1),(0,-a2),…,(0,-an)
值得注意的是，在θ取值为π/4的情况下，即十字形星座图的旋转量为π/4，十字形星座图中的星座点分别分布在经过原点的45°方向的直线上和经过原点的135°方向的直线上，因此，第二星座图模型中各个星座点所对应的坐标可以包括如下：



示例十四：
参照图21，图21是本申请实施例提供的第三星座图模型的示意图，其中，第三星座图模型可以理解为8臂星座图，图21左边坐标系所对应的8臂星座图中的星座点分别分布在经过x轴(即I路)的直线(包括正半轴的射线以及负半轴的射线)上、经过y轴(即Q路)的直线(包括正半轴的射线以及负半轴的射线)上、经过原点的45°方向的射线上、经过原点的135°方向的射线上、经过原点的225°方向的射线上、经过原点的315°方向的射线上。对于第三星座图模型，当bn＝an，可以形成如图21中间坐标系所对应的8臂星座图，当bn＝an+β，可以形成如图21右边坐标系所对应的8臂星座图。
可以理解的是，图21右边坐标系所对应的8臂星座图可以通过将图21中间坐标系所对应的8臂星座图中的4个臂(经过原点的45°方向、135°方向、225°方向、315°方向的射线)的星座点的幅度整体往外扩展而形成，能够让星形星座图的星座点分布更均匀，以提高解调性能
示例十五：
参照图22，图22是本申请实施例提供的第三星座图模型的示意图，其中，第三星座图模型可以理解为8臂星座图，当M3＝0，调制符号根据第三星座图模型对数据包中的3个比特信息进行调制而得到，且由N3＝2M3，对应得出N3＝1，因此，第三星座图模型为包括8个星座点的8臂星座图。其中，图22左边坐标系所对应的8臂星座图中的星座点分别分布在经过x轴(即I路)的直线(包括正半轴的射线以及负半轴的射线)上、经过y轴(即Q路)的直线(包括正半轴的射线以及负半轴的射线)上、经过原点的45°方向的射线上、经过原点的135°方向的射线上、经过原点的225°方向的射线上、经过原点的315°方向的射线上。对于第三星座图模型，当bn＝an，可以形成如图22左边坐标系所对应的8臂星座图，当bn＝an+β，可以形成如图22中间或右边坐标系所对应的8臂星座图。
可以理解的是，图22右边坐标系所对应的8臂星座图可以通过将图22左边坐标系所对应的8臂星座图中的4个臂(经过原点的45°方向、135°方向、225°方向、315°方向的射线)的星座点的幅度整体往外扩展而形成，能够让星形星座图的星座点分布更均匀，以提高解调性能。
针对上述实施例所提供的信息传输方法，下面以极稀疏导频为例进行示例性说明。
在一实施例中，可以定义一种解调参考信号(DMRS)集合，其中该集合包含12个参考信号，解调参考信号也可称解调参考信号端口(DMRS ports)，即是说，可以定义的一种包含12个解调参考信号端口(DMRS ports)的集合。
在一实施例中，如图23所示，图23为一实施例提供的一种定义物理资源块的示意图，在图23中， 可以定义一个物理资源块(Physical Resource Block，PRB)，该物理资源块包含时域上的14个正交频分复用(或者，离散傅里叶变换扩展OFDM(Discrete Fourier Transform-Spread-Orthogonal Frequency Division Multiplexing，DFT-S-OFDM)或者，单载波频分多址(Single-carrier Frequency-Division Multiple Access，SC-FDMA))符号，以及频域上的12个子载波。其中每个小格子表示一个OFDM符号的一个子载波，通常也称为一个资源单元(Resource Element，RE)，因此该物理资源块(PRB)一共包含12×14＝168个资源单元(RE)。由于使用前2个OFDM符号来承载解调参考信号，即使用前2个OFDM符号作为参考信号(DMRS)区域，因此，解调参考信号(DMRS)占用的资源开销是1/7。其中，除参考信号(DMRS)区域外的区域为调制符号区域。
在一实施例中，如图24所示，图24为一实施例提供的一种定义解调参考信号的示意图，12个解调参考信号(DMRS)从占用资源单元(RE)位置来看可以分成三组，第一组解调参考信号(DMRS)的非零值符号(或称为非零信号、有用信号等)(即符号是非零值)承载在第一资源单元(RE)上，该4个DMRS端口可以通过OCC码区分；第二组解调参考信号(DMRS)的非零值符号承载在第二资源单元(RE)上，该4个DMRS端口可以通过OCC码区分；第三组解调参考信号(DMRS)的非零值符号承载在第三资源单元(RE)上，该4个DMRS端口可以通过OCC码区分。其中，在图24中，第一资源单元所在的小方格用竖直线填充，第二资源单元所在的小方格用横直线填充，第三资源单元所在的小方格用波浪线填充，每组解调参考信号在空白未填充图案的资源单元(RE)上取值为0(即没有信号)。由此可见，对于每组解调参考信号来说，并不是在参考信号区域的所有资源单元(RE)上都有信号。但是，对某个第一通信节点(比如终端)而言，即便其使用的参考信号端口只是在参考信号区域的一些RE上没有信号，但是仍旧不能利用没有信号的RE传输信息。因此，每组解调参考信号中的参考信号(或者参考信号端口)占用的资源开销也是1/7。
若每组解调参考信号的非零值符号承载在相同的资源单元(RE)上，则只能通过不同取值的非零值符号区分出不同的参考信号，比如可通过时域OCC码以及频域OCC码来区分出不同的参考信号端口。在一实施例中，如图25所示，图25为一实施例提供的通过OCC码来区分出不同的参考信号端口的示意图。以图24中的第一组的4个解调参考信号为例，可以通过联合使用时域上的两长OCC码[1,1]、[1,-1]，以及频域上的两长OCC码[1,1]、[1,-1]来分出该4个解调参考信号端口，即该组DMRS端口，通过在第一资源单元上承载不同的OCC码，来生成不同的DMRS端口。上述图24中的第二组中的4个参考信号和第三组中的4个参考信号的情况也类似。因此，一共可以得到定义的这种解调参考信号(DMRS)集合的12个解调参考信号，也即12个解调参考信号端口。
在一实施例中，如图26所示，图26为一实施例提供的一种采用OCC码定义解调参考信号的示意图，对于图24中的第一组中的4个解调参考信号，可以通过联合使用时域上的两长OCC码[1,1]、[1,-1]，以及频域上的两长OCC码[1,1]、[1,-1]来分出该4个解调参考信号端口，其中，在图26中，第一列至第三列的阴影部分的数字由上至下均是1，第四列的阴影部分的数字由上至下均是-1，第五列至第七列的阴影部分的数字由上至下分别是1、-1、1、-1，第八列的阴影部分的数字由上至下分别是-1、1、-1、1。
在一实施例中，如图27所示，图27为一实施例提供的另一种采用OCC码定义解调参考信号的示意图。对于图24中的第二组中的4个解调参考信号，可以通过联合使用时域上的两长OCC码[1,1]、[1,-1]，以及频域上的两长OCC码[1,1]、[1,-1]来分出该4个解调参考信号端口，其中，在图27中，第一列至第三列的阴影部分的数字由上至下均是1，第四列的阴影部分的数字由上至下均是-1，第五列至第七列的阴影部分的数字由上至下分别是1、-1、1、-1，第八列的阴影部分的数字由上至下分别是-1、1、-1、1。
在一实施例中，如图28所示，图28为一实施例提供的另一种采用OCC码的解调参考信号的示意图。8表示第一个参考信号单元，9表示第二个参考信号单元；对于图24中第三组的中的4个解调参考信号，可以通过联合使用时域上的两长OCC码[1,1]、[1,-1]，以及频域上的两长OCC码[1,1]、[1,-1]来分出该4个解调参考信号端口，其中，在图28中，第一列至第三列的阴影部分的数字由上至下均是1，第四列的阴影部分的数字由上至下均是-1，第五列至第七列的阴影部分的数字由上至下分别是1、-1、1、-1，第八列的阴影部分的数字由上至下分别是-1、1、-1、1。
本申请中可将时域及频域上相邻的若干个资源单元(RE)上承载的参考信号称为一个参考信号单元(Reference Signal Element，RSE)，例如图23至图28中，时频域上连续的4个资源单元(RE)上承载的参考信号称为一个参考信号单元(RSE)，形象地看，有图案填充下的“田字格”上的参考信号就是一个参考信号单元(RSE)。从信道估计的功能上看，一个参考信号单元(RSE)除了由时域或者频域上相邻的资源单元构成的特点外，还有一个特点，即在应用参考信号进行信道估计时，每个参考信号单元(RSE)可以估计一个信道值。
按此定义，如图23至图28所示，在定义的解调参考信号(DMRS)集合中，每一个参考信号在1个PRB带宽内都有2个参考信号单元(RSE)，因此可以估计出一个1个PRB带宽内(也即12个子载波带宽内)2处的信道值。另外，1个PRB带宽内12个子载波的信道值可以通过这些参考信号单元估计值插值 得到。
在一实施例中，如图29所示，图29为一实施例提供的一种基于PRB传输的示意图。如果一次传输包含X个PRB，则该参考信号集合中的12个解调参考信号(或者12个解调参考信号端口)如图29所示，每个参考信号有2×X个参考信号单元(RSE)，因此可以估计整个传输带宽内等间隔的2×X处的信道值，然后可以通过插值得到传输带宽内全部12×X个子载波的信道值。
基于上述实施例，12个解调参考信号(DMRS)从占用资源单元(RE)位置来看可以分成三组。其中，图30为一实施例提供的另一种定义解调参考信号的示意图，在图30中，可以通过OCC码区分的第一组的4个解调参考信号；图31为一实施例提供的另一种定义解调参考信号的示意图，在图31中，可以通过OCC码区分的第二组的4个解调参考信号；图32为一实施例提供的另一种定义解调参考信号的实现示意图，在图32中，可以通过OCC码区分的第三组的4个解调参考信号。
由此可见，为了估计每个接入的终端的整个传输带宽的信道，解调参考信号(或导频)占用的资源较大，换而言之，解调参考信号在整个传输带宽内的分布具有一定的密度，不能太稀疏。在如图20至图32所示的实施例中，解调参考信号在整个传输带宽内的密度为每PRB有2处参考信号，或者每PRB有2个参考信号单元(RSE)。
对系统而言，参考信号的开销是1/7，即是说，系统付出了1/7的资源，只能设计12个解调参考信号。对于无连接传输状态下的信息传输而言，任意2个终端自主选择的参考信号的碰撞概率是1/12，可见，碰撞概率很高。所以，解调参考信号会严重限制无连接状态下进行信息传输的终端的数量。对于SPS场景下的信息传输而言，12个解调参考信号只能支持12个终端，可见，解调参考信号会严重限制SPS场景下进行信息传输的终端的数量。
如果参考信号还需要估计一定的频偏(Frequency Offset)，则每个参考信号的占用资源还会继续增加，换而言之，每个参考信号在传输信号中的密度还会继续增加。例如，参考信号再在时域上重复一次，以估计频偏，那么参考信号占用的资源翻倍，即资源开销为2/7。即是说，为了估计频选信道和频偏，系统付出2/7的资源，也只能设计12个解调参考信号。如果进一步，系统还需要应付一定的时偏(Timing Offset)，则参考信号占用资源还会继续增加，例如，系统付出3/7甚至4/7的开销，才能设计出12个解调参考信号。如此大的开销，却只能得到少量的参考信号(即参考信号端口)，由此可见，在无连接状态下所传输的参考信号的碰撞概率很高，SPS场景下进行信息传输的终端的数量也会受限。
如果多径信道在频域变化得更快一些，即频选特性更加明显，则为了保证信道估计的精度，解调参考信号在频域上的密度会继续增加，每组解调参考信号在每个PRB带宽内有3个参考信号单元(如形象地看有3个田字格)，因此，每个PRB上可以有3处估计值，而X个PRB上可以有3×X处估计值，然后通过线性插值可以得到X个PRB的全部子载波的信道。该参考信号同样占传输资源的1/7开销，但是只能分出8个解调参考信号(8个解调参考信号端口)，比上述中定义的解调参考信号的数量还少。可见，信道估计能力通常和参考信号数量成反比。
因此，应用到无连接传输场景和基于SPS的信息传输场景中的参考信号，所面临的问题为参考信号既要估计整个传输信道的频选信道以及时频偏，又要识别终端设备，所以参考信号占用的时频资源成倍递增，这导致在一定资源下参考信号的数量严重不足，继而影响了可以在无连接传输场景和基于SPS的信息传输场景中进行信息传输的终端的数量。
本申请主要出发点是极大地减轻参考信号的任务，因此可以使得每个参考信号占用的资源最小化，即是说使得每个参考信号在传输信号中的密度最稀疏，继而可以使得参考信号数量最大化，最终可以增加在无连接传输场景和基于SPS的信息传输场景下进行信息传输的终端的数量。
具体而言，本申请通过基于数据的信道估计技术(而不是基于参考信号)，通过数据自身的特性，例如通过调制符号的星座图的几何特点，来估计整个传输带宽的信道以及估计时频偏，即是说，无需通过参考信号来估计整个传输带宽的信道和时频偏。以信道估计为例，为了简化描述，以块平衰(Block Flat Fading)信道为例，可以通过上述实施例中的四种分区方法来估计星座图所受到的旋转缩放量，即可以通过上述实施例中的四种分区方法对块平衰信道进行估计，在此不多做赘述。
因此，本申请实施例中，参考信号的任务比相关方案小得多，所以本申请实施例中每个参考信号占用的资源比相关方案中每个参考信号占用的资源少，因此，在一定的开销下，本申请的参考信号数量比相关方案中的参考信号数量多。
另一方面，当基站有多根接收天线时，例如，R根接收天线，理论上该R根接收天线可以提供很强大的空域能力，以此提高多终端接入的性能。为了获得到这个空域能力，本申请提出可以使用“极稀疏”的参考信号来估计各个终端信号所经历的空域信道hk＝[hk1,hk2,...,hkR]t，其中，t是转置运算符，然后利用该空域信道来得到空域合并权值，进而对R根接收天线的接收信号做空域合并。具体而言，对终端k的信号进行空域合并，得到调制符号sk＝hk′*y，其中，y＝[yk1,yk2,...,ykR]t是R根接收天线的接收信号，hk′是hk的共轭转置，然后接收机利用空域合并后的调制符号sk估计终端k的信号经历的整个传输带宽的信道以 及估计时频偏，接着对空域合并后的调制符号sk补偿信道和时频偏，最后对补偿信道和时频偏的调制符号进行解调译码。
所以，本申请不用参考信号来估计整个传输带宽内的信道，也不用其估计时频偏。
在一实施例中，图33为一种定义参考信号的示意图，假设参考信号占用1个OFDM符号，一次传输包含X个PRB的时频资源，一个物理资源块(PRB)包含时域上的14个OFDM(或者DFT-S-OFDM或者SC-FDMA)符号，以及频域上的12个子载波。其中，使用前1个OFDM符号来承载解调参考信号，即前1个OFDM符号作为极稀疏导频区域，则1/14的资源用于传输参考信号，即12×X个资源单元(RE)用于传输参考信号。其中，除极稀疏导频区域外的区域为调制符号区域。
在一实施例中，如图34所示，图34为另一种系统定义的参考信号(即极稀疏导频)的示意图，该每一个参考信号均只在一个RE资源单元上有非零值符号(非零信号，或有用信号)，而在其余资源单元上都没有信号(即取值为0)，因此，可以占1/14开销的参考信号区域，且总共可以分出12×X个参考信号。具体而言，如果有6个PRB，且每个PRB的资源开销均为1/14，则可以分出6×12＝72个参考信号，远大于NR系统中的参考信号数量(该NR系统中的参考信号数量为8个或者12个，该NR系统中的参考信号所占用的资源开销还是1/7)。
在一实施例中，如图35所示，图35为另一种定义参考信号(即极稀疏导频端口)的示意图，系统定义的每个参考信号均只在2个RE资源单元上有非零值符号(或者非零信号，或者有用信号)，每2个RE可以通过2长的OCC分出2个参考信号，其余资源单元上都没有信号(即取值为0)，因此，可以占1/14开销的参考信号区域，且总共可以分出12×X个参考信号。具体而言，如果有6个PRB，且每个PRB的资源开销均为1/14开销，则可以分出6×12＝72个参考信号，其参考信号的数量远大于NR系统中的参考信号数量(该NR系统中的参考信号数量为8个或者12个)。因此，极稀疏导频的非零值符号的数量与PRB的数量呈正比。
在一实施例中，如图36所示，图36为一实施例提供的一种生成DMRS端口的示意图，通过在2个RE上承载不同的OCC码(如OCC码1、OCC码2)，以生成不同的DMRS端口，其中，在图36中，RE用以竖直线填充的两个小方格表示。
在一实施例中，如图37所示，图37为一实施例提供的另一种定义参考信号的示意图，其中，该系统定义的参考信号占用2个OFDM符号。
在一实施例中，如图38所示，图38为一实施例提供的另一种定义参考信号的示意图，该系统定义的每个参考信号均只在一组相邻的4个RE资源单元上有非零值符号(或者非零信号，或者有用信号)，但是会有4个参考信号复用一组相邻的4个RE，其中复用同一组相邻的4个RE的4个参考信号可以通过OCC码以区分，因此，当一次传输包括X个PRB，且每个PRB均占用1/7开销的参考信号区域，则总共可以分出24×X个参考信号。具体而言，如果一次传输包括6个PRB，且每个PRB均占用1/7的资源开销，则可以分出6×24＝144个参考信号，因此，其参考信号的数量远大于NR系统中的参考信号数量(该NR系统中的参考信号数量为8个或者12个)，同时，说明了极稀疏导频的数量与PRB的数量呈正比。
值得注意的是，上述所有实施例中所示数值仅为示例性描述，不作具体限定，数值可以根据实际情况进行适应性调整。
由此可见，极稀疏导频为预设导频集合里的非零元素(即非零值符号)很少的导频，例如非零元素只有1-4个，因此，关于极稀疏导频的方案可以在不增加导频开销的情况下，显著增加导频的数量，继而显著减少导频碰撞的概率。另外，基站可以从极稀疏导频中估计出无线信道的部分信息，而无需从极稀疏导频中估计出无线信道的全部信息，而且基站可以从调制符号中进一步提取信道信息，进而利用该信道信息完成对调制符号的均衡。
需要说明的是，极稀疏导频仅仅用于空域合并，但并不能用于信道均衡，因此，极稀疏导频可以认为是一种空域合并参考信号，在此不作具体限制。
还需要说明的是，虽然图33、图34、图35和图36中所示极稀疏参考信号(即极稀疏导频)都位于传输资源的第一个符号，或者第一个符号和第二个符号，但本申请并不限制极稀疏参考信号的位置，例如极稀疏参考信号的位置也可以位于传输资源的中间。
可以理解的是，上述所有实施例中的信息传输，其中的信息是广义的信息，即信息可以是业务数据，也可以是用于系统控制的信息，即信令；或者，信息可以包括有需要传输的比特数据，例如业务比特数据或者信令比特数据，其中，message、information、payload等不同的英文表述均可表示信息。
可以理解的是，上述所有实施例中的第一通信节点可以是终端，例如，移动电话、智能电话、笔记本电脑、PDA(Personal Digital Assistant，个人数字助理)、PAD(平板电脑)、导航装置等移动终端，也可以是物联网设备终端等等，在此不作具体限制。
可以理解的是，上述所有实施例中的第二通信节点可以是基站、接收机以及接入点，等等，在此不作具体限制。
另外，参照图39，本申请的一个实施例还提供了一种通信设备100，该通信设备100包括至少一个处理器101及至少一个存储器102，存储器102用于存储至少一个程序。
处理器101和存储器102可以通过总线或者其他方式连接。
存储器102作为一种非暂态计算机可读存储介质，可用于存储非暂态软件程序以及非暂态性计算机可执行程序。此外，存储器102可以包括高速随机存取存储器，还可以包括非暂态存储器，例如至少一个磁盘存储器件、闪存器件、或其他非暂态固态存储器件。在一些实施方式中，存储器102可选包括相对于处理器101远程设置的存储器，这些远程存储器可以通过网络连接至该处理器101。上述网络的实例包括但不限于互联网、企业内部网、局域网、移动通信网及其组合。
实现上述实施例的信息传输方法所需的非暂态软件程序以及指令存储在存储器102中，当被处理器101执行时，执行上述实施例中的信息传输方法，例如，执行以上描述的图2中的方法步骤S110至S120以及图4中的方法步骤S210。
以上所描述的设备实施例仅仅是示意性的，其中作为分离部件说明的单元可以是或者也可以不是物理上分开的，即可以位于一个地方，或者也可以分布到多个网络单元上。可以根据实际的需要选择其中的部分或者全部模块来实现本实施例方案的目的。
此外，本申请的一个实施例还提供了一种计算机可读存储介质，该计算机可读存储介质存储有计算机可执行指令，该计算机可执行指令被一个处理器或控制器执行，例如，被上述设备实施例中的一个处理器执行，可使得上述处理器执行上述实施例中的信息传输方法，执行以上描述的图2中的方法步骤S110至S120以及图4中的方法步骤S210。
此外，本申请的一个实施例还提供了一种计算机程序产品，包括计算机程序或计算机指令，计算机程序或计算机指令存储在计算机可读存储介质中，计算机设备的处理器从计算机可读存储介质读取计算机程序或计算机指令，处理器执行计算机程序或计算机指令，使得计算机设备执行上述实施例中的信息传输方法，例如，执行以上描述的图2中的方法步骤S110至S120以及图4中的方法步骤S210。
本领域普通技术人员可以理解，上文中所公开方法中的全部或某些步骤、系统可以被实施为软件、固件、硬件及其适当的组合。某些物理组件或所有物理组件可以被实施为由处理器，如中央处理器、数字信号处理器或微处理器执行的软件，或者被实施为硬件，或者被实施为集成电路，如专用集成电路。这样的软件可以分布在计算机可读介质上，计算机可读介质可以包括计算机存储介质(或非暂时性介质)和通信介质(或暂时性介质)。如本领域普通技术人员公知的，术语计算机存储介质包括在用于存储信息(诸如计算机可读指令、数据结构、程序模块或其他数据)的任何方法或技术中实施的易失性和非易失性、可移除和不可移除介质。计算机存储介质包括但不限于RAM、ROM、EEPROM、闪存或其他存储器技术、CD-ROM、数字多功能盘(DVD)或其他光盘存储、磁盒、磁带、磁盘存储或其他磁存储装置、或者可以用于存储期望的信息并且可以被计算机访问的任何其他的介质。此外，本领域普通技术人员公知的是，通信介质通常包含计算机可读指令、数据结构、程序模块或者诸如载波或其他传输机制之类的调制数据信号中的其他数据，并且可包括任何信息递送介质。

Claims (34)

  1. 一种信息传输方法，应用于第一通信节点，所述信息传输方法包括：
    确定第一数量的极稀疏导频；
    将数据包和所述第一数量的所述极稀疏导频一起传输至第二通信节点；
    其中，所述第一数量大于或者等于1，所述数据包至少包含调制符号；
    所述调制符号根据第一星座图模型对所述数据包中的M1+1个比特信息进行调制而得到，所述第一星座图模型包含2*N1个星座点，M1为大于或等于1的整数，N1和M1满足公式N1＝2M1；
    所述第一星座图模型中的2*N1个所述星座点所对应的复数形式包括如下：
    a1ejθ,a2ejθ,…,aN1ejθ,
    a1ej(θ+π),a2ej(θ+π),…,aN1ej(θ+π)；
    π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN1均为正数，并且满足：0<a1<a2<…<aN1；
    或者,
    所述调制符号根据第二星座图模型对所述数据包中的M2+2个比特信息进行调制而得到，所述第二星座图模型包含4*N2个星座点，M2为大于或等于1的整数，N2和M2满足公式N2＝2M2；
    所述第二星座图模型中的4*N2个所述星座点所对应的复数形式包括如下:
    a1ejθ,a2ejθ,…,aN2ejθ,
    b1ej(θ+π/2),b2ej(θ+π/2),…,bN2ej(θ+π/2),
    a1ej(θ+π),a2ej(θ+π),…,aN2ej(θ+π),
    b1ej(θ+3π/2),b2ej(θ+3π/2),…,bN2ej(θ+3π/2)；
    π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN2和b1,b2,…,bN2均为正数，并且满足：0<a1<a2<…<aN2，0<b1<b2<…<bN2；
    或者,
    所述调制符号根据第三星座图模型对所述数据包中的M3+3个比特信息进行调制而得到，所述第三星座图模型包含8*N3个星座点，M3为大于或等于0的整数，N3和M3满足公式N3＝2M3；
    所述第三星座图模型中的8*N3个所述星座点所对应的复数形式包括如下：
    a1ejθ,a2ejθ,…,aN3ejθ,



    a1ej(θ+π),a2ej(θ+π),…,aN3ej(θ+π),


    π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN3和b1,b2,…,bN3均为正数，并且满足：0<a1<a2<…<aN3，0<b1<b2<…<bN3。
  2. 根据权利要求1所述的方法，其中：
    当所述调制符号根据所述第一星座图模型而调制得到，a1,a2,…,aN1均可由如下公式表示：
    an＝(2n-1+Δ)d；
    其中，n的取值包括1,2,...,N1；d为正实数，Δ为大于或等于0的实数；
    或者，当所述调制符号根据所述第二星座图模型而调制得到，a1,a2,…,aN2均可由如下公式表示：
    an＝(2n-1+Δ)d；
    b1,b2,…,bN2均可由如下公式表示：
    bn＝an+β；
    其中，n的取值包括1,2,...,N2；d为正实数，Δ和β均为大于或等于0的实数；
    或者，当所述调制符号根据所述第三星座图模型而调制得到，a1,a2,…,aN3均可由如下公式表示：
    an＝(2n-1+Δ)d；
    b1,b2,…,bN3均可由如下公式表示：
    bn＝an+β；
    其中，n的取值包括1,2,...,N3；d为正实数，Δ和β均为大于或等于0的实数。
  3. 根据权利要求2所述的方法，其中，Δ的取值为0，且d的取值为1，使得an满足an＝2n-1。
  4. 根据权利要求2所述的方法，其中，Δ的取值为1，且d的取值为1/2，使得an满足an＝n。
  5. 根据权利要求2所述的方法，其中，Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
  6. 根据权利要求2所述的方法，其中，Δ的取值为
  7. 根据权利要求6所述的方法，其中：
    d的取值为1，使得an满足
    或者，d的取值为1/2，使得an满足
  8. 根据权利要求2所述的方法，其中，当所述调制符号根据所述第二星座图模型或者所述第三星座图模型而调制得到，β等于0。
  9. 根据权利要求2所述的方法，其中，当所述调制符号根据所述第三星座图模型而调制得到，β大于0。
  10. 根据权利要求2所述的方法，其中：
    当所述调制符号根据所述第一星座图模型而调制得到，d的取值为使采用所述第一星座图模型进行调制得到的所述调制符号的平均功率等于1的数值；
    或者，
    当所述调制符号根据所述第二星座图模型而调制得到，d的取值为使采用所述第二星座图模型进行调制得到的所述调制符号的平均功率等于1的数值；
    或者，
    当所述调制符号根据所述第三星座图模型而调制得到，d的取值为使采用所述第三星座图模型进行调制得到的所述调制符号的平均功率等于1的数值。
  11. 根据权利要求1所述的方法，其中：
    θ的取值为0；
    或者，
    θ的取值满足公式θ＝π/4；
    或者，
    θ的取值满足公式θ＝π/8。
  12. 根据权利要求1所述的方法，其中，每个所述极稀疏导频均包括第二数量的非零值符号，所述第二数量大于0且小于5，所述第二数量的所述非零值符号承载在时频域上相邻的第三数量的资源单元上，或者承载在按时间先后顺序的第三数量的符号上，或者承载在频域相邻子载波上的第三数量的资源单元上，其中，所述第三数量与所述第二数量相等。
  13. 根据权利要求1或12所述的方法，其中，每个所述极稀疏导频的符号长度均大于24。
  14. 根据权利要求12所述的方法，其中：
    所述第二数量的取值为1；
    或者，
    所述第二数量的取值为2，所述第二数量的所述非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值为[a1,a2]或者[b1,b2]，其中，[a1,a2]和[b1,b2]正交；
    或者，
    所述第二数量的取值为2，所述第二数量的所述非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值至少包括如下的情况：
    [p1,p2]＝[1,1]；
    [p1,p2]＝[1,-1]；
    [p1,p2]＝[1,j]；
    [p1,p2]＝[1,-j]；
    或者，
    所述第二数量的取值为4，所述第二数量的所述非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值为[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]或者[d1,d2,d3,d4]，其中，[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]和[d1,d2,d3,d4]相互正交；
    或者，
    所述第二数量的取值为4，所述第二数量的所述非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值至少包括如下的情况：
    [p1,p2,p3,p4]＝[1,1,1,1]；
    [p1,p2,p3,p4]＝[1,1,-1,-1]；
    [p1,p2,p3,p4]＝[1,-1,1,-1]；
    [p1,p2,p3,p4]＝[1,-1,-1,1]；
    [p1,p2,p3,p4]＝[1,1,j,-j]；
    [p1,p2,p3,p4]＝[1,1,-j,j]；
    [p1,p2,p3,p4]＝[1,-1,j,j]；
    [p1,p2,p3,p4]＝[1,-1,-j,-j]；
    [p1,p2,p3,p4]＝[1,j,1,-j]；
    [p1,p2,p3,p4]＝[1,j,-1,j]；
    [p1,p2,p3,p4]＝[1,-j,1,j]；
    [p1,p2,p3,p4]＝[1,-j,-1,-j]；
    [p1,p2,p3,p4]＝[1,j,j,-1]；
    [p1,p2,p3,p4]＝[1,j,-j,1]；
    [p1,p2,p3,p4]＝[1,-j,j,1]；
    [p1,p2,p3,p4]＝[1,-j,-j,-1]；
    [p1,p2,p3,p4]＝[1,1,1,-1]；
    [p1,p2,p3,p4]＝[1,1,-1,1]；
    [p1,p2,p3,p4]＝[1,-1,1,1]；
    [p1,p2,p3,p4]＝[1,-1,-1,-1]；
    [p1,p2,p3,p4]＝[1,1,j,j]；
    [p1,p2,p3,p4]＝[1,1,-j,-j]；
    [p1,p2,p3,p4]＝[1,-1,j,-j]；
    [p1,p2,p3,p4]＝[1,-1,-j,j]；
    [p1,p2,p3,p4]＝[1,j,1,j]；
    [p1,p2,p3,p4]＝[1,j,-1,-j]；
    [p1,p2,p3,p4]＝[1,-j,1,-j]；
    [p1,p2,p3,p4]＝[1,-j,-1,j]；
    [p1,p2,p3,p4]＝[1,j,j,1]；
    [p1,p2,p3,p4]＝[1,j,-j,-1]；
    [p1,p2,p3,p4]＝[1,-j,j,-1]；
    [p1,p2,p3,p4]＝[1,-j,-j,1]；
    [p1,p2,p3,p4]＝[1,1,1,j]；
    [p1,p2,p3,p4]＝[1,1,-1,-j]；
    [p1,p2,p3,p4]＝[1,-1,1,-j]；
    [p1,p2,p3,p4]＝[1,-1,-1,j]；
    [p1,p2,p3,p4]＝[1,1,j,1]；
    [p1,p2,p3,p4]＝[1,1,-j,-1]；
    [p1,p2,p3,p4]＝[1,-1,j,-1]；
    [p1,p2,p3,p4]＝[1,-1,-j,1]；
    [p1,p2,p3,p4]＝[1,j,1,1]；
    [p1,p2,p3,p4]＝[1,j,-1,-1]；
    [p1,p2,p3,p4]＝[1,-j,1,-1]；
    [p1,p2,p3,p4]＝[1,-j,-1,1]；
    [p1,p2,p3,p4]＝[1,j,j,-j]；
    [p1,p2,p3,p4]＝[1,j,-j,j]；
    [p1,p2,p3,p4]＝[1,-j,j,j]；
    [p1,p2,p3,p4]＝[1,-j,-j,-j]；
    [p1,p2,p3,p4]＝[1,1,1,-j]；
    [p1,p2,p3,p4]＝[1,1,-1,j]；
    [p1,p2,p3,p4]＝[1,-1,1,j]；
    [p1,p2,p3,p4]＝[1,-1,-1,-j]；
    [p1,p2,p3,p4]＝[1,1,j,-1]；
    [p1,p2,p3,p4]＝[1,1,-j,1]；
    [p1,p2,p3,p4]＝[1,-1,j,1]；
    [p1,p2,p3,p4]＝[1,-1,-j,-j]；
    [p1,p2,p3,p4]＝[1,j,1,-1]；
    [p1,p2,p3,p4]＝[1,j,-1,1]；
    [p1,p2,p3,p4]＝[1,-j,1,1]；
    [p1,p2,p3,p4]＝[1,-j,-1,-1]；
    [p1,p2,p3,p4]＝[1,j,j,j]；
    [p1,p2,p3,p4]＝[1,j,-j,-j]；
    [p1,p2,p3,p4]＝[1,-j,j,-j]；
    [p1,p2,p3,p4]＝[1,-j,-j,j]。
  15. 根据权利要求1所述的方法，其中，所述第一数量的取值大于或者等于2时：
    所述第一数量的所述极稀疏导频之间相互独立；
    或者，
    所述第一数量的所述极稀疏导频根据所述数据包中的信息而确定；
    或者，
    所述第一数量的所述极稀疏导频根据所述数据包中的一个或者多个比特信息而确定；
    或者，
    每个所述极稀疏导频均根据所述数据包中的第四数量的比特信息从预设导频集合中确定得到，其中，所述预设导频集合包括第五数量的导频，所述第四数量与所述第五数量成对数函数关系，所述对数函数为以2为底数的对数函数。
  16. 根据权利要求1所述的方法，其中，所述第一数量的取值为1或者2。
  17. 一种信息传输方法，应用于第二通信节点，所述信息传输方法包括：
    接收第一通信节点发送的数据包和第一数量的极稀疏导频；
    其中，所述第一数量大于或者等于1，所述数据包至少包含调制符号；
    所述调制符号根据第一星座图模型对所述数据包中的M1+1个比特信息进行调制而得到，所述第一星座图模型包含2*N1个星座点，M1为大于或等于1的整数，N1和M1满足公式N1＝2M1；
    所述第一星座图模型中的2*N1个所述星座点所对应的复数形式包括如下：
    a1ejθ,a2ejθ,…,aN1ejθ,
    a1ej(θ+π),a2ej(θ+π),…,aN1ej(θ+π)；
    π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN1均为正数，并且满足：0<a1<a2<…<aN1；
    或者,
    所述调制符号根据第二星座图模型对所述数据包中的M2+2个比特信息进行调制而得到，所述第二星座图模型包含4*N2个星座点，M2为大于或等于1的整数，N2和M2满足公式N2＝2M2；
    所述第二星座图模型中的4*N2个所述星座点所对应的复数形式包括如下:
    a1ejθ,a2ejθ,…,aN2ejθ,
    b1ej(θ+π/2),b2ej(θ+π/2),…,bN2ej(θ+π/2),
    a1ej(θ+π),a2ej(θ+π),…,aN2ej(θ+π),
    b1ej(θ+3π/2),b2ej(θ+3π/2),…,bN2ej(θ+3π/2)；
    π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN2和b1,b2,…,bN2均为正数，并且满足：0<a1<a2<…<aN2，0<b1<b2<…<bN2；
    或者,
    所述调制符号根据第三星座图模型对所述数据包中的M3+3个比特信息进行调制而得到，所述第三星座图模型包含8*N3个星座点，M3为大于或等于0的整数，N3和M3满足公式N3＝2M3；
    所述第三星座图模型中的8*N3个所述星座点所对应的复数形式包括如下：
    a1ejθ,a2ejθ,…,aN3ejθ,



    a1ej(θ+π),a2ej(θ+π),…,aN3ej(θ+π),


    π为圆周率；j为虚数；j的值等于e为自然对数；θ是一个实数；a1,a2,…,aN3和b1,b2,…,bN3均为正数，并且满足：0<a1<a2<…<aN3，0<b1<b2<…<bN3。
  18. 根据权利要求17所述的方法，其中：
    当所述调制符号根据所述第一星座图模型而调制得到，a1,a2,…,aN1均可由如下公式表示：
    an＝(2n-1+Δ)d；
    或者，当所述调制符号根据所述第二星座图模型而调制得到，a1,a2,…,aN2均可由如下公式表示：
    an＝(2n-1+Δ)d；
    b1,b2,…,bN2均可由如下公式表示：
    bn＝an+β；
    其中，n的取值包括1,2,...,N2；d为正实数，Δ和β均为大于或等于0的实数；
    或者，当所述调制符号根据所述第三星座图模型而调制得到，a1,a2,…,aN3均可由如下公式表示：
    an＝(2n-1+Δ)d；
    b1,b2,…,bN3均可由如下公式表示：
    bn＝an+β；
    其中，n的取值包括1,2,...,N3；d为正实数，Δ和β均为大于或等于0的实数。
  19. 根据权利要求18所述的方法，其中，Δ的取值为0，且d的取值为1，使得an满足an＝2n-1。
  20. 根据权利要求18所述的方法，其中，Δ的取值为1，且d的取值为1/2，使得an满足an＝n。
  21. 根据权利要求18所述的方法，其中，Δ的取值为3，且d的取值为1/2，使得an满足an＝n+1。
  22. 根据权利要求18所述的方法，其中，Δ的取值为
  23. 根据权利要求22所述的方法，其中：
    d的取值为1，使得an满足
    或者，d的取值为1/2，使得an满足
  24. 根据权利要求18所述的方法，其中，当所述调制符号根据所述第二星座图模型或者所述第三星座图模型而调制得到，β等于0。
  25. 根据权利要求18所述的方法，其中，当所述调制符号根据所述第三星座图模型而调制得到，β大于0。
  26. 根据权利要求18所述的方法，其中：
    当所述调制符号根据所述第一星座图模型而调制得到，d的取值为使采用所述第一星座图模型进行调制得到的所述调制符号的平均功率等于1的数值；
    或者，
    当所述调制符号根据所述第二星座图模型而调制得到，d的取值为使采用所述第二星座图模型进行调制得到的所述调制符号的平均功率等于1的数值；
    或者，
    当所述调制符号根据所述第三星座图模型而调制得到，d的取值为使采用所述第三星座图模型进行调制得到的所述调制符号的平均功率等于1的数值。
  27. 根据权利要求17所述的方法，其中：
    θ的取值为0；
    或者，
    θ的取值满足公式θ＝π/4；
    或者，
    θ的取值满足公式θ＝π/8。
  28. 根据权利要求17所述的方法，其中，每个所述极稀疏导频均包括第二数量的非零值符号，所述第 二数量大于0且小于5，所述第二数量的所述非零值符号承载在时频域上相邻的第三数量的资源单元上，或者承载在按时间先后顺序的第三数量的符号上，或者承载在频域相邻子载波上的第三数量的资源单元上，其中，所述第三数量与所述第二数量相等。
  29. 根据权利要求17或28所述的方法，其中，每个所述极稀疏导频的符号长度均大于24。
  30. 根据权利要求28所述的方法，其中：
    所述第二数量的取值为1；
    或者，
    所述第二数量的取值为2，所述第二数量的所述非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值为[a1,a2]或者[b1,b2]，其中，[a1,a2]和[b1,b2]正交；
    或者，
    所述第二数量的取值为2，所述第二数量的所述非零值符号构成非零值符号对[p1,p2]，[p1,p2]的取值至少包括如下的情况：
    [p1,p2]＝[1,1]；
    [p1,p2]＝[1,-1]；
    [p1,p2]＝[1,j]；
    [p1,p2]＝[1,-j]；
    或者，
    所述第二数量的取值为4，所述第二数量的所述非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值为[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]或者[d1,d2,d3,d4]，其中，[a1,a2,a3,a4]、[b1,b2,b3,b4]、[c1,c2,c3,c4]和[d1,d2,d3,d4]相互正交；
    或者，
    所述第二数量的取值为4，所述第二数量的所述非零值符号构成非零值符号组[p1,p2,p3,p4]，[p1,p2,p3,p4]的取值至少包括如下的情况：
    [p1,p2,p3,p4]＝[1,1,1,1]；
    [p1,p2,p3,p4]＝[1,1,-1,-1]；
    [p1,p2,p3,p4]＝[1,-1,1,-1]；
    [p1,p2,p3,p4]＝[1,-1,-1,1]；
    [p1,p2,p3,p4]＝[1,1,j,-j]；
    [p1,p2,p3,p4]＝[1,1,-j,j]；
    [p1,p2,p3,p4]＝[1,-1,j,j]；
    [p1,p2,p3,p4]＝[1,-1,-j,-j]；
    [p1,p2,p3,p4]＝[1,j,1,-j]；
    [p1,p2,p3,p4]＝[1,j,-1,j]；
    [p1,p2,p3,p4]＝[1,-j,1,j]；
    [p1,p2,p3,p4]＝[1,-j,-1,-j]；
    [p1,p2,p3,p4]＝[1,j,j,-1]；
    [p1,p2,p3,p4]＝[1,j,-j,1]；
    [p1,p2,p3,p4]＝[1,-j,j,1]；
    [p1,p2,p3,p4]＝[1,-j,-j,-1]；
    [p1,p2,p3,p4]＝[1,1,1,-1]；
    [p1,p2,p3,p4]＝[1,1,-1,1]；
    [p1,p2,p3,p4]＝[1,-1,1,1]；
    [p1,p2,p3,p4]＝[1,-1,-1,-1]；
    [p1,p2,p3,p4]＝[1,1,j,j]；
    [p1,p2,p3,p4]＝[1,1,-j,-j]；
    [p1,p2,p3,p4]＝[1,-1,j,-j]；
    [p1,p2,p3,p4]＝[1,-1,-j,j]；
    [p1,p2,p3,p4]＝[1,j,1,j]；
    [p1,p2,p3,p4]＝[1,j,-1,-j]；
    [p1,p2,p3,p4]＝[1,-j,1,-j]；
    [p1,p2,p3,p4]＝[1,-j,-1,j]；
    [p1,p2,p3,p4]＝[1,j,j,1]；
    [p1,p2,p3,p4]＝[1,j,-j,-1]；
    [p1,p2,p3,p4]＝[1,-j,j,-1]；
    [p1,p2,p3,p4]＝[1,-j,-j,1]；
    [p1,p2,p3,p4]＝[1,1,1,j]；
    [p1,p2,p3,p4]＝[1,1,-1,-j]；
    [p1,p2,p3,p4]＝[1,-1,1,-j]；
    [p1,p2,p3,p4]＝[1,-1,-1,j]；
    [p1,p2,p3,p4]＝[1,1,j,1]；
    [p1,p2,p3,p4]＝[1,1,-j,-1]；
    [p1,p2,p3,p4]＝[1,-1,j,-1]；
    [p1,p2,p3,p4]＝[1,-1,-j,1]；
    [p1,p2,p3,p4]＝[1,j,1,1]；
    [p1,p2,p3,p4]＝[1,j,-1,-1]；
    [p1,p2,p3,p4]＝[1,-j,1,-1]；
    [p1,p2,p3,p4]＝[1,-j,-1,1]；
    [p1,p2,p3,p4]＝[1,j,j,-j]；
    [p1,p2,p3,p4]＝[1,j,-j,j]；
    [p1,p2,p3,p4]＝[1,-j,j,j]；
    [p1,p2,p3,p4]＝[1,-j,-j,-j]；
    [p1,p2,p3,p4]＝[1,1,1,-j]；
    [p1,p2,p3,p4]＝[1,1,-1,j]；
    [p1,p2,p3,p4]＝[1,-1,1,j]；
    [p1,p2,p3,p4]＝[1,-1,-1,-j]；
    [p1,p2,p3,p4]＝[1,1,j,-1]；
    [p1,p2,p3,p4]＝[1,1,-j,1]；
    [p1,p2,p3,p4]＝[1,-1,j,1]；
    [p1,p2,p3,p4]＝[1,-1,-j,-j]；
    [p1,p2,p3,p4]＝[1,j,1,-1]；
    [p1,p2,p3,p4]＝[1,j,-1,1]；
    [p1,p2,p3,p4]＝[1,-j,1,1]；
    [p1,p2,p3,p4]＝[1,-j,-1,-1]；
    [p1,p2,p3,p4]＝[1,j,j,j]；
    [p1,p2,p3,p4]＝[1,j,-j,-j]；
    [p1,p2,p3,p4]＝[1,-j,j,-j]；
    [p1,p2,p3,p4]＝[1,-j,-j,j]。
  31. 根据权利要求17所述的方法，其中，所述第一数量的取值大于或者等于2时：
    所述第一数量的所述极稀疏导频之间相互独立；
    或者，
    所述第一数量的所述极稀疏导频根据所述数据包中的信息而确定；
    或者，
    所述第一数量的所述极稀疏导频根据所述数据包中的一个或者多个比特信息而确定；
    或者，
    每个所述极稀疏导频均根据所述数据包中的第四数量的比特信息从预设导频集合中确定得到，其中，所述预设导频集合包括第五数量的导频，所述第四数量与所述第五数量成对数函数关系，所述对数函数为以2为底数的对数函数。
  32. 根据权利要求17所述的方法，其中，所述第一数量的取值为1或者2。
  33. 一种通信设备，包括：
    至少一个处理器；
    至少一个存储器，用于存储至少一个程序；
    当至少一个所述程序被至少一个所述处理器执行时实现如权利要求1至32中任意一项所述的信息传输方法。
  34. 一种计算机可读存储介质，存储有计算机可执行指令，所述计算机可执行指令用于执行如权利要求1至32中任意一项所述的信息传输方法。
PCT/CN2023/134170 2022-11-30 2023-11-24 信息传输方法、通信设备及存储介质 WO2024114548A1 (zh)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN202211521420.5 2022-11-30
CN202211521420.5A CN118118312A (zh) 2022-11-30 2022-11-30 信息传输方法、通信设备及存储介质

Publications (1)

Publication Number Publication Date
WO2024114548A1 true WO2024114548A1 (zh) 2024-06-06

Family

ID=91207529

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2023/134170 WO2024114548A1 (zh) 2022-11-30 2023-11-24 信息传输方法、通信设备及存储介质

Country Status (2)

Country Link
CN (1) CN118118312A (zh)
WO (1) WO2024114548A1 (zh)

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20110128842A1 (en) * 2009-11-27 2011-06-02 Sequans Communications Method for Estimating a Received Signal and Corresponding Device
CN105847192A (zh) * 2016-03-17 2016-08-10 上海交通大学 一种动态稀疏信道的联合估计方法
CN111726194A (zh) * 2019-03-19 2020-09-29 华为技术有限公司 传输数据的方法和通信装置
CN112887068A (zh) * 2019-11-30 2021-06-01 华为技术有限公司 数据传输方法、发送设备和接收设备
CN115004841A (zh) * 2020-01-23 2022-09-02 中兴通讯股份有限公司 用于无线通信中参考信号配置的方法、装置和系统

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20110128842A1 (en) * 2009-11-27 2011-06-02 Sequans Communications Method for Estimating a Received Signal and Corresponding Device
CN105847192A (zh) * 2016-03-17 2016-08-10 上海交通大学 一种动态稀疏信道的联合估计方法
CN111726194A (zh) * 2019-03-19 2020-09-29 华为技术有限公司 传输数据的方法和通信装置
CN112887068A (zh) * 2019-11-30 2021-06-01 华为技术有限公司 数据传输方法、发送设备和接收设备
CN115004841A (zh) * 2020-01-23 2022-09-02 中兴通讯股份有限公司 用于无线通信中参考信号配置的方法、装置和系统

Also Published As

Publication number Publication date
CN118118312A (zh) 2024-05-31

Similar Documents

Publication Publication Date Title
US11108608B2 (en) Information transmission method and apparatus
WO2017041297A1 (zh) 信息传输装置、方法以及通信系统
WO2018059488A1 (zh) 一种参考信号传输方法和装置
KR20200015753A (ko) 위상 잡음에 최적화된 직교 진폭 변조를 위한 방법 및 장치
WO2017016342A1 (zh) 一种多用户信息传输的调制方法、解调方法及装置
US20230246895A1 (en) Data modulation method and apparatus, device, and storage medium
US11546193B2 (en) Data compression method and apparatus
WO2018059350A1 (zh) 一种数据处理方法、装置和系统
WO2016082554A1 (zh) 叠加编码、解码方法、装置、发射机及接收机
US9426015B2 (en) Apparatus and method for symbol mapping in wireless communication system using multi tone frequency quadrature amplitude modulation
CN104283835A (zh) 一种正交幅度调制软比特解调方法及其装置
WO2024114548A1 (zh) 信息传输方法、通信设备及存储介质
WO2016184241A1 (zh) 一种多用户接入方法及装置
WO2020143612A1 (en) Semi-orthogonal multiple access communication of ppdu and acknowledgment
WO2024114590A1 (zh) 信息传输方法、通信设备及存储介质
WO2024114591A1 (zh) 信号调制方法、通信设备及存储介质
WO2012028087A1 (zh) Mu-mimo通信系统中数据调制的方法及装置
CN110505359B (zh) 无线广播通信系统中非均匀调制解调方法、系统、介质及终端
WO2020020330A1 (zh) 一种数据调制方法、装置及计算机存储介质
WO2024114603A1 (zh) 信息传输方法及装置、存储介质、电子装置
WO2019134689A1 (zh) 数据传输方法及装置、存储介质、电子装置
CN106789762B (zh) Fbmc导频信道估计方法及装置
WO2022161224A1 (zh) 数据发送方法、数据接收方法及相关装置
WO2023116613A1 (zh) 通信方法及装置
WO2023098398A1 (zh) 通信方法及装置