WO2023273184A1 - 矢量角控制方法 - Google Patents

矢量角控制方法 Download PDF

Info

Publication number
WO2023273184A1
WO2023273184A1 PCT/CN2021/137182 CN2021137182W WO2023273184A1 WO 2023273184 A1 WO2023273184 A1 WO 2023273184A1 CN 2021137182 W CN2021137182 W CN 2021137182W WO 2023273184 A1 WO2023273184 A1 WO 2023273184A1
Authority
WO
WIPO (PCT)
Prior art keywords
vector angle
link
output
control
current
Prior art date
Application number
PCT/CN2021/137182
Other languages
English (en)
French (fr)
Inventor
李武华
颜晔
田野
王宇翔
李成敏
李楚杉
何湘宁
Original Assignee
浙江大学
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from CN202110734368.0A external-priority patent/CN113437895B/zh
Priority claimed from CN202110734358.7A external-priority patent/CN113422533B/zh
Application filed by 浙江大学 filed Critical 浙江大学
Priority to JP2023514375A priority Critical patent/JP2023539674A/ja
Publication of WO2023273184A1 publication Critical patent/WO2023273184A1/zh

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration

Definitions

  • the invention belongs to the technical field of power electronic control, and in particular relates to a vector angle-based control method.
  • the three-phase converter is usually controlled by a proportional-integral controller (ie, PI controller) or a proportional resonant controller (ie, PR controller).
  • PI controller realizes the active power control of the three-phase current in the synchronous rotating coordinate system.
  • PR controller proportional resonant controller
  • the three-phase asymmetrical converter can use the PR controller to realize the active power control of the three-phase current in the static coordinate system.
  • the system control delay can reach millisecond level.
  • the present invention introduces a new control degree of freedom for the traditional PI controller or PR controller to increase the phase margin, so as to more effectively improve the system stability and dynamic performance.
  • the invention provides a vector angle control method.
  • the present invention provides a vector angle control method, which is used for vector angle proportional integral control or vector angle proportional resonance control.
  • the vector angle control method comprises the following steps:
  • the current value of axis and ⁇ axis, superscript T means transpose
  • the current error value is used as the input of the vector angle proportional integral link or the vector angle proportional resonance link, and the corresponding vector angle proportional integral link or the vector angle proportional resonance link output is obtained after calculation.
  • the vector angle proportional integral control is performed, the vector angle proportional integral
  • the calculation formula of the link output m dq_R is
  • m dq_R i dq_E ⁇ (K p +K i ⁇ e j ⁇ i /s), where K p is the proportional coefficient, K i is the integral coefficient, ⁇ i is the vector angle, s is the Laplacian operator, and j is The imaginary number unit, i dq_E is the current error value, when the vector angle proportional resonance control is performed, the calculation formula of the output M ⁇ _R of the vector angle proportional resonance control is
  • K p ⁇ and K p ⁇ are the proportional coefficients of the ⁇ -axis and the ⁇ -axis in the stationary coordinate system
  • K r ⁇ and K r ⁇ are the resonance coefficients of the ⁇ -axis and the ⁇ -axis in the stationary coordinate system, respectively
  • ⁇ 0 is the fundamental angular frequency
  • s is the Laplacian operator
  • ⁇ rp is the difference vector angle and the value is a compromise between -90° and 0°
  • I ⁇ _E is the current error value
  • the current sampling value is used as the input of the decoupling link, and the decoupling output is obtained after calculation;
  • the output of the vector angle proportional integral link or the vector angle proportional resonance link is added to the decoupling output as the input of the delay compensation link, and the output of the delay compensation link is used as the total output of the control loop;
  • the total output of the control loop is converted to the corresponding dq/abc coordinates or ⁇ /abc coordinates to obtain a three-phase modulated wave, which is compared with the carrier in the modulation and drive module to generate a drive signal to drive the converter topology to realize power conversion.
  • the current error value is used as the input of the vector angle proportional integral link, and before the output of the vector angle proportional integral link is obtained after calculation, the phase equalization link is also included processing, introducing the phase equalization angle, equalizing the current error i dq_E , and then taking the equalized current error as the input of the vector angle proportional integral link, and obtaining the output m dq_R of the vector angle proportional integral link after calculation, and the vector angle proportional integral link
  • the calculation formula of the decoupling output M ⁇ _D is:
  • L ⁇ and L ⁇ are the equivalent inductance values of the ⁇ -axis and ⁇ -axis in the stationary coordinate system, respectively, and ⁇ 0 is the fundamental angular frequency.
  • the calculation formula of the output M ⁇ of the delay compensation link is:
  • M ⁇ _RD M ⁇ _R +M ⁇ _D
  • n the compensation coefficient
  • T s the control cycle
  • ⁇ 0 the fundamental angular frequency
  • the present invention is aimed at the control of three-phase converters in the dq synchronous coordinate system.
  • the existing phase equalization scheme can only equalize the positive and negative bilateral phase margins, but cannot improve the overall phase margin.
  • the present invention proposes vector angle PI control.
  • a new control degree of freedom vector angle ⁇ i is introduced, which can realize the simultaneous improvement of positive phase margin and negative phase margin, thereby improving the performance under low carrier ratio conditions.
  • the stability margin and dynamic performance have achieved beneficial technical effects.
  • the carrier ratio is low, the positive and negative bilateral phase margins of the three-phase asymmetrical converter system under traditional PR control are low, and even unstable.
  • the present invention proposes the matrix vector angle PR control.
  • Figure 1 shows a schematic diagram of a power conversion circuit
  • Fig. 2 shows the general control block diagram provided according to the first embodiment of the present invention
  • FIG. 3 is a block diagram of a control loop with differential phase correction resonance control provided according to the first embodiment of the present invention
  • FIG. 4 is a block diagram showing the implementation of complex vectors in the real number domain in the control loop according to the first embodiment of the present invention
  • Fig. 5 shows the bilateral frequency-domain Bode diagram of the vector angle PI controller provided according to the first embodiment of the present invention
  • Fig. 6 shows the transient current waveform in the synchronous coordinate system of the traditional solution and the first embodiment solution
  • Fig. 7 shows the transient current waveforms in the synchronous coordinate system of the first embodiment scheme one and scheme two;
  • FIG. 8 is a general control block diagram provided according to the second embodiment of the present invention.
  • FIG. 9 is a block diagram of a control loop with matrix vector angle proportional resonance control provided according to a second embodiment of the present invention.
  • Fig. 10 shows according to the matrix vector angle PR controller that the second embodiment of the present invention provides the characteristic locus bilateral frequency domain Bode figure
  • FIG. 11 shows the transient current waveform diagrams of the conventional scheme and the scheme of the second embodiment.
  • FIG. 1 is a schematic diagram of a power conversion circuit, which is applicable to the first embodiment and the second embodiment described below.
  • 2-7 are schematic diagrams related to the vector angle proportional-integral control method provided by the first embodiment of the present invention.
  • 8-10 are schematic diagrams related to the vector angle proportional resonance control method provided by the second embodiment of the present invention.
  • the first embodiment of the present invention takes the current loop control of the general-purpose three-phase bridge inverter topology as an example, the three-phase current sampling obtains the AC side currents i a , i b , and i c , and obtains the static coordinates through abc/dq coordinate transformation The current id and i q under the system are used as the input of the control loop.
  • control loop outputs the modulation waves m d , m q in the static coordinate system, and the three-phase modulation waves ma , m b , m c are obtained through dq /abc coordinate transformation, and compared with the carrier in the modulation and drive module, The drive signal is generated to drive the converter topology to realize electric energy conversion.
  • the vector angle proportional integral control method provided by the first embodiment of the present invention includes a phase equalization link, a vector angle PI link and a delay compensation link.
  • the definition of a complex vector labeled dq is the same.
  • the control loop samples the corresponding idq from the controlled object, and outputs the modulated wave m dq to control the controlled object.
  • the steps of the vector angle-based proportional-integral control method corresponding to the control loop are as follows:
  • phase equalization link 3
  • i dq_B i dq_E
  • i dq_B i dq_E ⁇ e j ⁇ b
  • ⁇ b is the phase equalization angle, and its value can be zero, or usually can be set as half of the difference between the positive and negative bilateral phase margins to equalize the bilateral phase margins.
  • m dq_R i dq_B ⁇ (K p +K i ⁇ e j ⁇ i /s)
  • K p is a proportional coefficient
  • K i is an integral coefficient
  • ⁇ i is a vector angle proposed in this embodiment
  • s is a Laplacian operator
  • L is the inductance value of the AC side
  • ⁇ 0 is the fundamental angular frequency
  • m dq_RD After adding the output m dq_R of the vector angle PI and the decoupling output m dq_D , m dq_RD is obtained, which is used as the input of the delay compensation link, and the total output m dq of the control loop is obtained after calculation; the calculation formula of the delay compensation link is as follows :
  • m dq m dq_RD
  • m dq m dq_RD ⁇ e jnTs ⁇ ⁇ 0
  • T s is the control cycle
  • n is the compensation coefficient, which can be a typical value of 1.5, 0, or any other value.
  • the total output m dq of the control loop undergoes dq/abc coordinate transformation to obtain three-phase modulation waves ma , m b , m c , and compares them with the carrier wave in the modulation and drive module to generate a drive signal to drive the converter topology to realize power conversion .
  • the expression for the control loop includes the imaginary unit j, which represents the cross-coupling between the d-axis and the q-axis.
  • m d_D -i q ⁇ 0 L
  • m q_D i d ⁇ 0 L
  • m d_D and m q_D represent the decoupled output of the d-axis and q-axis, respectively.
  • a common control scheme is as follows: three-phase current sampling to obtain the AC side currents i a , i b , i c , and obtain the currents in the static coordinate system through abc/dq coordinate transformation
  • the current id and i q are used as the input of the control loop.
  • the specific implementation process of the control loop is the same as that described above, including phase equalization, PI, feedback decoupling, and delay compensation.
  • the traditional solution corresponds to the following formula:
  • the output of the above control loop is the modulation wave m d , m q in the synchronous coordinate system, and the three-phase modulation wave ma , m b , m c are obtained through the dq /abc coordinate transformation, and compared with the carrier in the modulation and drive module, The drive signal is generated to drive the converter topology to realize electric energy conversion.
  • this embodiment proposes the vector angle PI.
  • a new control degree of freedom vector angle ⁇ i is introduced, which can realize the simultaneous improvement of the positive phase margin and the negative phase margin, so that
  • ⁇ i is the vector angle proposed in this embodiment, and its value ranges from 0° to 90°. The larger the value, the stronger the phase leading ability on the right side of the pole. Considering that the value is too large, the bandwidth on the left side of the pole will be larger. , here, ⁇ i is selected as a compromise of 60°.
  • the phase lag of the positive frequency band is significantly reduced, and the corresponding phase margin of the positive end is significantly increased; and for the negative frequency band, if the crossover frequency is set to the left of -10Hz , the corresponding decrease of the phase margin of the negative terminal is smaller than the increase of the phase margin of the positive terminal. If it is further set to the left of -26Hz, the corresponding phase margin of the negative terminal also slightly increases. Therefore, the vector angle PI controller can realize the function of increasing the sum of bilateral phase margins.
  • the frequency carrier ratio is 5
  • the bandwidth f c is 60 Hz
  • the proportional coefficient K p is 2 ⁇ f c L
  • the integral coefficient K i is ⁇ f c K p /2.
  • the vector angle PI controller corrects the current loop from an unstable state to The phase margin is close to 45°, and the transient adjustment time is significantly reduced to a state of about 0.03s.
  • the second embodiment of the present invention takes the current loop control of the general-purpose three-phase bridge inverter topology as an example, in which the three-phase current sampling obtains the AC side currents i a , i b , and i c , and obtains the static The current i ⁇ and i ⁇ in the coordinate system are used as the input of the control loop.
  • the control loop outputs the modulation waves m ⁇ , m ⁇ in the static coordinate system, and the three-phase modulation waves ma , m b , m c are obtained through ⁇ /abc coordinate transformation, and compared with the carrier in the modulation and drive module,
  • the drive signal is generated to drive the converter topology to realize electric energy conversion.
  • the vector angle proportional resonance control method provided by the second embodiment of the present invention includes a matrix vector angle PR link, a feedback decoupling link and a delay compensation link.
  • the expressions of matrix and transfer function matrix are adopted, and the operation under this expression conforms to the operation rules of matrix.
  • the control loop samples the corresponding I ⁇ from the controlled object, and outputs the modulated wave M ⁇ to control the controlled object.
  • the steps of the proportional resonance control method based on the matrix vector angle corresponding to the control loop are as follows:
  • K p ⁇ and K p ⁇ are the proportional coefficients of the ⁇ -axis and the ⁇ -axis in the stationary coordinate system respectively
  • K r ⁇ and K r ⁇ are the resonance coefficients of the ⁇ -axis and the ⁇ -axis in the stationary coordinate system respectively
  • ⁇ rp is the second embodiment of the present invention
  • ⁇ 0 is the fundamental angular frequency
  • s is the Laplacian operator
  • L ⁇ and L ⁇ are the equivalent inductance values of the ⁇ -axis and ⁇ -axis in the stationary coordinate system, respectively.
  • M ⁇ _RD is obtained, which is used as the input of the delay compensation link, and the total output M ⁇ of the control loop is obtained after calculation; the calculation formula of the delay compensation link as follows:
  • T s is the control cycle
  • n is the compensation coefficient, which can be a typical value of 1.5, 0, or any other value.
  • the total output M ⁇ of the control loop undergoes ⁇ /abc coordinate transformation to obtain three-phase modulation waves ma , m b , m c , and compares them with the carrier wave in the modulation and drive module to generate a drive signal to drive the converter topology to realize power conversion .
  • a general control scheme is as follows: the three-phase current sampling obtains the AC side currents i a , i b , i c , and obtains the currents in the static coordinate system through abc/ ⁇ coordinate transformation
  • the current i ⁇ and i ⁇ are used as the input of the control loop.
  • the specific implementation process of the control loop is the same as that described above, including links such as PR, feedback decoupling, and delay compensation.
  • the PR link compared with the matrix-vector angle PR with additional degrees of freedom for adjusting the difference vector angle proposed in the second embodiment of the present invention, the corresponding formula of the traditional scheme is as follows:
  • the output of the above control loop is the modulation waves m ⁇ , m ⁇ in the static coordinate system .
  • the three-phase modulation waves ma , m b , m c are obtained , and compared with the carrier in the modulation and drive module,
  • the drive signal is generated to drive the converter topology to realize electric energy conversion.
  • the positive and negative bilateral phase margins of the system under traditional PR control are low, or even unstable.
  • the second embodiment of the present invention aims at the shortcomings of the traditional PR control scheme, and proposes the matrix vector angle PR, that is, on the traditional PR, a new control degree of freedom difference vector angle ⁇ rp is introduced, which can realize the positive phase margin and negative phase margin of the characteristic trajectory.
  • the phase margin is improved at the same time, thereby improving the stability margin and dynamic performance under low carrier ratio conditions.
  • ⁇ rp is the difference vector angle proposed by the second embodiment of the present invention, and its value is from -90° to 0°. The closer the value is to -90°, the easier the amplitude-frequency characteristic of the characteristic trajectory is. Distortion leads to system instability. The closer the value is to 0°, the less obvious the improvement effect of the phase margin of the characteristic trajectory is.
  • ⁇ rp is selected as a compromise of -60°.
  • the matrix vector angle PR controller can realize the function of increasing the positive and negative bilateral phase margins.
  • the frequency carrier ratio is 7
  • the bandwidth f c is 50Hz
  • the proportional coefficient K p ⁇ is 2 ⁇ f c L ⁇
  • K p ⁇ is 2 ⁇ f c L ⁇
  • the resonance coefficient K r ⁇ is K p ⁇ ⁇ c /4
  • K r ⁇ is K p ⁇ c / 4 .
  • the difference vector angle ⁇ rp is 0; in the solution of the second embodiment of the present invention, the difference vector angle ⁇ rp is -60°.
  • the stability margin and dynamic performance of the converter under the condition of low carrier ratio of the large-capacity converter can be improved, and beneficial technical effects have been achieved.
  • the present invention is not limited to the above-mentioned specific implementation manners. Those skilled in the art can adopt various other implementation manners according to the content disclosed in the present invention, such as replacing the feedback decoupling link with the feedforward decoupling link, converting the two-level current The topology is replaced by a three-level topology, etc. Accordingly, the claims are intended to cover all modifications within the true spirit and scope of the invention.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
  • Rectifiers (AREA)

Abstract

本发明提供一种矢量角控制方法，用于矢量角比例积分控制或者矢量角比例谐振控制。首先获得被控变流器中三相电流值，经过坐标变换得到同步/静止坐标系下的电流值，定义电流采样值；将电流参考值减去电流采样值，得到电流误差值；将电流误差值作为矢量角PI环节或者矢量角PR环节的输入，计算后得到对应的输出。将电流采样值作为解耦环节的输入，计算后得到解耦输出。将矢量角PI环节或者矢量角PR环节的输出与解耦输出相加，作为延迟补偿环节的输入，将延迟补偿环节的输出作为控制环总输出。控制环总输出经过对应的坐标变换得到三相调制波，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。

Description

矢量角控制方法 技术领域
本发明属于领域电力电子控制技术领域，尤其涉及一种基于矢量角的控制方法。
背景技术
三相大容量变流器作为能量变换装置，在电气化交通与船舶电力系统等工业领域得到日益广泛的应用。该类变流器通常工作于低载波比工况，控制与调制延迟显著，控制环稳定裕度不足，影响其动态性能。
三相变流器通常采用比例积分控制器(即PI控制器)或者比例谐振控制器(即PR控制器)进行控制。PI控制器在同步旋转坐标系下实现对三相电流的有功控制。但受限于通常小于1000Hz的大功率器件的开关频率，系统控制延迟可达毫秒级，利用复传递函数的数学工具进行建模与分析，其相位裕度与相应的动态性能严重不足。
在三相三线制大功率变流系统中，存在三相不对称的情况，如凸极同步电机的dq轴阻抗不对称、三相不对称负载、不对称故障状态等引入的三相不对称。三相不对称变流器可以采用PR控制器，在静止坐标系下实现对三相电流的有功控制。但受限于通常小于1000Hz的大功率器件的开关频率，系统控制延迟可达毫秒级，利用传递函数矩阵和特征轨迹分析等数学工具进行建模与分析，其相位裕度与相应的动态性能也严重不足。
发明内容
在大容量变流器所在的低载波比工况下，针对传统的PI控制器或者PR控制器，本发明引入新的调控自由度，来增加相位裕度，以更有效地提高系统稳定性与动态性能。为提高大容量变流器的动态性能，本发明提供一种矢量角控制方法。
为了实现本发明的一目的，本发明提供一种矢量角控制方法，用于矢量角比例积分控制或者矢量角比例谐振控制。矢量角控制方法包括如下步骤：
采样被控变流器中每一相的电流，当进行矢量角比例积分控制时，经过abc/dq坐标变换得到同步坐标系下的电流i d和i q，并定义电流采样值的复向量表示形式i dq＝i d+ji q，其中i d与i q分别为同步坐标系下d轴与q轴的电流值，j为虚数单位，当进行矢量角比例谐振控制时，经过abc/αβ坐标变换得到静止坐标系下的电流i α和i β，并定义电流采样值的二维列向量表示形式I αβ＝[i αi β] T，其中i α与i β分别为静止坐标系下α轴与β轴的电流值，上角标T为转置；
将电流参考值减去电流采样值，得到电流误差值；
将电流误差值作为矢量角比例积分环节或者矢量角比例谐振环节的输入，计算后得到对应的矢量角比例积分环节或者矢量角比例谐振环节输出，当进行矢量角比例积分控制时，矢量角比例积分环节的输出m dq_R的计算公式为
m dq_R＝i dq_E·(K p+K i·e jθi/s)，其中，K p为比例系数，K i为积分系数，θ i为矢量角，s为拉普拉斯算子，j为虚数单位，i dq_E为电流误差值，当进行矢量角比例谐振控制时，矢量角比例谐振控制的输出M αβ_R的计算公式为
Figure PCTCN2021137182-appb-000001
其中，K pα与K pβ分别为静止坐标系下α轴与β轴的比例系数，K rα与K rβ分别为静止坐标系下α轴与β轴的谐振系数，ω 0为基波角频率，s为拉普拉斯算子，θ rp是差值矢量角且取值在-90°到0°之间折中选取，I αβ_E为电流误差值；
将电流采样值作为解耦环节的输入，计算后得到解耦输出；
将矢量角比例积分环节或者矢量角比例谐振环节的输出与解耦输出相加，作为延迟补偿环节的输入，将延迟补偿环节的输出作为控制环总输出；
控制环总输出经过对应的dq/abc坐标或αβ/abc坐标变换得到三相调制波，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
于本发明的一实施例中，当进行矢量角比例积分控制时，在将电流误差值作为矢量角比例积分环节的输入，计算后得到矢量角比例积分环节的输出之前，还包括相位均衡环节的处理，引入相位均衡角，对电流误差i dq_E进行均衡，再将均衡后的电流误差作为矢量角比例积分环节的输入，计算后得到矢量角比例积分环节的输出m dq_R，矢量角比例积分环节的输出m dq_R的计算公式为m dq_R＝i dq_B·(K p+K i1·e jθi/s)，其中i dq_B＝i dq_E·e jθb，θ b是相位均衡角，i dq_B是相位均衡结果。
于本发明的一实施例中，当进行矢量角比例积分控制时，解耦输出m dq_D的计算公式为m dq_D＝i dq·jω 0L，其中L为交流侧电感值，ω 0为基波角频率。
于本发明的一实施例中，当进行矢量角比例积分控制时，延迟补偿环节的输出m dq的计算公式为m dq＝m dq_RD或m dq＝m dq_RD·e jnTs·ω0，其中m dq_RD＝m dq_R+m dq_D，T s为控制周期，n为补偿系数，ω 0为基波角频率。
于本发明的一实施例中，当进行矢量角比例谐振控制时，解耦输出M αβ_D的计算公式为
Figure PCTCN2021137182-appb-000002
其中，L α和L β分别为静止坐标系下α轴与β轴的等效电感值，ω 0为基波角频率。
于本发明的一实施例中，当进行矢量角比例谐振控制时，延迟补偿环节的的输出M αβ的计算公式为
M αβ＝M αβ_RD，或
Figure PCTCN2021137182-appb-000003
其中，M αβ_RD＝M αβ_R+M αβ_D，n为补偿系数，T s为控制周期，ω 0为基波角频率。
综上所述，本发明针对三相变流器在dq同步坐标系下的控制，既有的相位均衡方案只能均衡正负双边相位裕度，但不能提高总的相位裕度。本发明提出了矢量角PI控制，在传统PI控制上，引入新的调控自由度矢量角θ i，可实现正 相位裕度与负相位裕度的同时提升，从而提高低载波比工况下的稳定裕度与动态性能，取得了有益的技术效果。另一方面，在载波比较低时，传统PR控制下三相不对称变流器系统的正负双边相位裕度低，甚至会不稳定。本发明提出了矩阵矢量角PR控制，在传统PR控制上，引入新的调控自由度差值矢量角θ rp，可实现特征轨迹正相位裕度与负相位裕度的同时提升，从而提高低载波比工况下的稳定裕度与动态性能，取得了有益的技术效果。
附图说明
图1所示为功率变换电路的示意图；
图2所示为根据本发明第一实施例提供的总控制框图；
图3所示为根据本发明第一实施例提供的具有差异化相位校正谐振控制的控制环框图；
图4所示为根据本发明第一实施例提供的控制环中复向量在实数域的实现框图；
图5所示为根据本发明第一实施例提供的矢量角PI控制器的双边频域波特图；
图6所示为传统方案与第一实施例方案一在同步坐标系下暂态电流波形图；
图7所示为第一实施例方案一与方案二在同步坐标系下暂态电流波形图；
图8所示为根据本发明第二实施例提供的总控制框图；
图9所示为根据本发明第二实施例提供的具有矩阵矢量角比例谐振控制的控制环框图；
图10所示为根据本发明第二实施例提供的矩阵矢量角PR控制器的特征轨迹双边频域波特图；
图11所示为传统方案与第二实施例方案的暂态电流波形图。
具体实施方式
图1所示为功率变换电路的示意图，适用于如下所述的第一实施例和第二实施例。图2-图7所示为本发明第一实施例提供的矢量角比例积分控制方法相关的示意图。图8-10所示为本发明第二实施例提供的矢量角比例谐振控制方法相关的示意图。
本发明第一实施例以通用的三相桥式逆变拓扑的电流环控制为例，三相电流采样获得交流侧电流i a、i b、i c，并经过abc/dq坐标变换得到静止坐标系下的电流i d、i q，作为控制环的输入。接下来，控制环输出静止坐标系下的调制波m d、m q，经过dq/abc坐标变换得到三相调制波m a、m b、m c，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
本发明第一实施例提供的矢量角比例积分控制方法，包括相位均衡环节、矢量角PI环节和延迟补偿环节。这里采用复向量及复传递函数的表达方式。以静止坐标系下的电流采样值i dq为例，该复向量i dq＝i d+ji q，其中j为虚数单位，i d与i q分别代表d轴与q轴的电流值，其余下标含dq的复向量的定义与此相同。
控制环从被控对象采样获得相应的i dq，并输出调制波m dq来控制被控对象。在本发明的第一实施例中，该控制环对应的基于矢量角的比例积分控制方法的步骤如下：
1)采样被控变流器中每一相的电流，经过abc/dq坐标变换得到同步坐标系下的电流i d和i q，并定义电流采样值i dq＝i d+ji q，其中i d与i q分别为同步坐标系下d轴与q轴的电流值，i dq为复向量，j为虚数单位。
2)将电流参考值i dq_R减去电流采样值i dq，得到电流误差i dq_E。
3)电流误差i dq_E直接作为i dq_B，或作为相位均衡环节的输入，计算后得到i dq_B；所述的相位均衡环节的计算公式如下：
i dq_B＝i dq_E，或i dq_B＝i dq_E·e jθb
其中，θ b是相位均衡角，其值可为零，或通常可设为正负双边相位裕度的差值的一半，来均衡双边相位裕度。
4)均衡后的电流误差i dq_B作为矢量角PI环节的输入，计算后得到m dq_R；所述的矢量角PI环节的计算公式如下：
m dq_R＝i dq_B·(K p+K i·e jθi/s)
其中，K p为比例系数，K i为积分系数，θ i是本实施例所提出的矢量角，s为拉普拉斯算子。
5)将电流采样值i dq作为解耦环节的输入，计算后得到解耦输出m dq_D；所述的解耦环节的计算公式如下：
m dq_D＝i dq·jω 0L
其中，L为交流侧电感值，ω 0为基波角频率。
6)将矢量角PI的输出m dq_R与解耦输出m dq_D相加后得到m dq_RD，作为延迟补偿环节的输入，计算后得到控制环总输出m dq；所述的延迟补偿环节的计算公式如下：
m dq＝m dq_RD，或m dq＝m dq_RD·e jnTs·ω0
其中，T s为控制周期，n为补偿系数，其可为典型值1.5、0，或其余任意值。
7)控制环总输出m dq经过dq/abc坐标变换得到三相调制波m a、m b、m c，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
接下来简要阐述上述复向量在实数域的实现方式。控制环的表达式中包含虚数单位j，其代表d轴与q轴间的交叉耦合。其中反馈解耦环节包括j位于分子的项jω 0L，即m dq_D＝i dq·jω 0L，其在实数域的实现方式如图4中(a)所示，即：
m d_D＝-i q·ω 0L,m q_D＝i d·ω 0L
其中，m d_D和m q_D分别表示d轴和q轴的解耦输出。此外，矢量角PI环节、相位均衡环节(i dq_B＝i dq_E·e jθb)、延迟补偿环节(m dq＝m dq_RD·e jnTs·ω0)中包括指数函数。以上述指数函数的通用形式y dq＝u dq·e jθ为例，其在实数域的实现方式如图4中(b)所示，即：
y d＝u d·cosθ-u q·sinθ,y q＝u d·sinθ+u q·cosθ
其中，y dq与u dq仍采用前述的复向量的定义，即y d+jy q＝y dq，u d+ju q＝u dq，θ表示指数函数中用于使相位超前的角度。
下面给出本发明第一实施例的一个应用实例。
对于图1所示的三相功率变换电路，一种通用的控制方案为：三相电流采样获得交流侧电流i a、i b、i c，并经过abc/dq坐标变换得到静止坐标系下的电流i d、i q，作为控制环的输入。这里，控制环的具体实施过程与上文的表述相同，包括相位均衡、PI、反馈解耦、延迟补偿这些环节。对于PI环节，相比于本实施例提出的具有额外矢量角调控自由度的矢量角PI，传统方案对应如下公式：
m dq_R＝i dq_B·(K p+K i/s)
上述控制环的输出为同步坐标系下的调制波m d、m q，经过dq/abc坐标变换得到三相调制波m a、m b、m c，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
在大容量变流器对应的低载波比工况下，针对三相变流器在dq同步坐标系下的控制，既有的相位均衡方案只能均衡正负双边相位裕度，但不能提高总的相位裕度。本实施例针对传统的相位均衡方案的不足，提出了矢量角PI，在传统PI上，引入新的调控自由度矢量角θ i，可实现正相位裕度与负相位裕度的同时提升，从而提高低载波比工况下的稳定裕度与动态性能，具体分析如下。
利用复传递函数对改进前后的PI控制器进行分析，得到如图5所示的双边频域波特图。本图中，θ i是本实施例所提出的矢量角，其取值为0°到90°，值越大，在极点右侧相位超前能力越强，考虑到值过大会导致极点左侧带宽降低，这里，θ i选取为折中的60°。可以看出，在应用本实施例所提出的矢量角后，正频段的相位滞后显著减小，对应的正端相位裕度显著增加；而对于负频段，若将穿越频率设置在-10Hz左侧，对应的负端相位裕度的减小小于正端相位裕度的增大，若进一步设置在-26Hz左侧，对应的负端相位裕度也略有增加。因此，矢量角PI控制器可实现增大双边相位裕度之和的功能。
接下来进行传统方案与本实施例方案的时域对比分析。参数设置如下：频率载波比为5，带宽f c为60Hz，比例系数K p为2πf c L，积分系数K i为πf cK p/2。传统方案中，矢量角θ i与相位均衡角θ b均为0°；本实施例方案一与方案二中，矢量角θ i均为60°，而相位均衡角θ b，在方案一中仍为0，在方案二中选取为使双边相位裕度均衡的41.7°，即方案一对应i dq_B＝i dq_E，方案二对应i dq_B＝i dq_E·e jθb。
对比传统方案与本实施例方案一，如图6所示，当有功电流指令于0.03s从0pu跳变为1pu时，经典PI控制器下的电流响应呈发散失稳状态，而矢量角PI控制器下的电流响应可实现近似临界稳定。
进一步对比本实施例方案一与本实施例方案二，如图7所示，矢量角PI控制器在频率载波比为5、设计带宽为60Hz的条件下，将电流环由不稳定状态，校正为相位裕度接近45°、暂态调节时间显著降至约0.03s的状态。
因此，通过基于矢量角的比例积分控制，可提高大容量变流器低载波比工况下变流器的稳定裕度与动态性能，取得了有益的技术效果。
本发明第二实施例以通用的三相桥式逆变拓扑的电流环控制为例，其中三相电流采样获得交流侧电流i a、i b、i c，并经过abc/αβ坐标变换得到静止坐标系下的电流i α、i β，作为控制环的输入。接下来，控制环输出静止坐标系下的调制波m α、m β，经过αβ/abc坐标变换得到三相调制波m a、m b、m c，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
本发明第二实施例提供的矢量角比例谐振控制方法，包括矩阵矢量角PR环节、反馈解耦环节和延迟补偿环节。这里采用矩阵及传递函数矩阵的表达方式，该表达方式下的运算符合矩阵的运算规则。控制环从被控对象采样获得相应的I αβ，并输出调制波M αβ来控制被控对象。
在本发明的第二实施中，该控制环对应的基于矩阵矢量角的比例谐振控制方法的步骤如下：
1)采样被控变流器中每一相的电流，经过abc/αβ坐标变换得到静止坐标系下的电流i α和i β，并定义电流采样值I αβ＝[i αi β] T，其中i α与i β分别为静止坐标系下α轴与β轴的电流值，I αβ为二维列向量，上角标T为转置；
2)将电流参考值I αβ_R减去电流采样值I αβ，得到电流误差I αβ_E；
3)将电流误差I αβ_E作为矩阵矢量角PR环节的输入，计算后得到M αβ_R；所述的矩阵矢量角PR环节的计算公式如下：
Figure PCTCN2021137182-appb-000004
其中，K pα与K pβ分别为静止坐标系下α轴与β轴的比例系数，K rα与K rβ分别为静止坐标系下α轴与β轴的谐振系数，θ rp是本发明第二实施例所提出的差值矢量角，ω 0为基波角频率，s为拉普拉斯算子；
4)将电流采样值I αβ作为解耦环节的输入，计算后得到解耦输出M αβ_D；所述的解耦环节的计算公式如下：
Figure PCTCN2021137182-appb-000005
其中，L α与L β分别为静止坐标系下α轴与β轴的等效电感值。
5)将矩阵矢量角PR的输出M αβ_R与解耦输出M αβ_D相加后得到M αβ_RD，作为延迟补偿环节的输入，计算后得到控制环总输出M αβ；所述的延迟补偿环节的计算公式如下：
M αβ＝M αβ_RD，或
Figure PCTCN2021137182-appb-000006
其中，T s为控制周期，n为补偿系数，其可为典型值1.5、0，或其余任意值。
6)控制环总输出M αβ经过αβ/abc坐标变换得到三相调制波m a、m b、m c，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
下面给出本发明第二实施例的一个应用实例。
对于图1所示的三相功率变换电路，一种通用的控制方案为：三相电流采样获得交流侧电流i a、i b、i c，并经过abc/αβ坐标变换得到静止坐标系下的电流i α、i β，作为控制环的输入。这里，控制环的具体实施过程与上文的表述相同，包括PR、反馈解耦、延迟补偿这些环节。对于PR环节，相比于本发明第二实施例提出的具有额外差值矢量角调控自由度的矩阵矢量角PR，传统方案对应公式如下：
Figure PCTCN2021137182-appb-000007
上述控制环的输出为静止坐标系下的调制波m α、m β，经过αβ/abc坐标变换得到三相调制波m a、m b、m c，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
在大容量变流器对应的低载波比工况下，针对三相不对称变流器在αβ静止坐标系下的控制，传统PR控制下系统的正负双边相位裕度低，甚至不稳定。本发明第二实施例针对传统PR控制方案的不足，提出了矩阵矢量角PR，即在传统PR上，引入新的调控自由度差值矢量角θ rp，可实现特征轨迹正相位裕度与负相位裕度的同时提升，从而提高低载波比工况下的稳定裕度与动态性能，具体分析如下。
利用传递函数矩阵和特征轨迹分析等数学工具对改进前后的PR控制器进行分析，得到如图10所示的双边频域波特图，本图中，虚线表示传统方案，实线表示本发明第二实施例提出的方案，θ rp是本发明第二实施例所提出的差值矢量角，其取值为-90°到0°，取值越靠近-90°，特征轨迹幅频特性越容易畸变导致系统失稳，取值越靠近0°，特征轨迹相位裕度提升效果越不明显，这里θ rp选取为折中的-60°。可以看出，在应用本发明第二实施例所提出的矩阵矢量角PR后，负频段相位裕度由-4.2°增大到30.7°，正频段相位裕度由19.9°增大到27.5°。因此，矩阵矢量角PR控制器可实现增大正负双边相位裕度的功能。
接下来进行传统方案与本发明第二实施例方案的时域对比分析。参数设置如下：频率载波比为7，带宽f c为50Hz，比例系数K pα为2πf cL α，K pβ为2πf cL β，谐振系数K rα为K pαω c/4，K rβ为K pβω c/4。传统方案中，差值矢量角θ rp为0；本发明第二实施例方案中，差值矢量角θ rp为-60°。
对比传统方案与本发明第二实施例方案，如图11所示，当有功电流指令于0.1s从0pu跳变为1pu时，传统方案下的电流响应振荡多次后会逐渐发散，而本发明第二实施例方案下的电流响应快速收敛。
因此，通过基于矩阵矢量角的比例谐振控制，可提高大容量变流器低载波比工况下变流器的稳定裕度与动态性能，取得了有益的技术效果。
本发明不局限于上述具体实施方式，本领域的技术人员根据本发明公开的内容，可以采用多种其他实施方式，如将反馈解耦环节替换为前馈解耦环节、将两电平变流拓扑替换为三电平拓扑等。因而，权利要求书旨在涵盖本发明真正构思和范围内的所有变型。

Claims (6)

  1. 一种矢量角控制方法，用于矢量角比例积分控制或者矢量角比例谐振控制，其特征在于，所述矢量角控制方法包括如下步骤：
    采样被控变流器中每一相的电流，当进行矢量角比例积分控制时，经过abc/dq坐标变换得到同步坐标系下的电流i d和i q，并定义电流采样值的复向量表示形式i dq＝i d+ji q，其中i d与i q分别为同步坐标系下d轴与q轴的电流值，j为虚数单位，当进行矢量角比例谐振控制时，经过abc/αβ坐标变换得到静止坐标系下的电流i α和i β，并定义电流采样值的二维列向量表示形式I αβ＝[i αi β] T，其中i α与i β分别为静止坐标系下α轴与β轴的电流值，上角标T为转置；
    将电流参考值减去电流采样值，得到电流误差值；
    将电流误差值作为矢量角比例积分环节或者矢量角比例谐振环节的输入，计算后得到对应的矢量角比例积分环节或者矢量角比例谐振环节输出，当进行矢量角比例积分控制时，矢量角比例积分环节的输出m dq_R的计算公式为m dq_R＝i dq_E·(K p+K i·e jθi/s)，其中，K p为比例系数，K i为积分系数，θ i为矢量角，s为拉普拉斯算子，j为虚数单位，i dq_E为电流误差值，当进行矢量角比例谐振控制时，矢量角比例谐振控制的输出M αβ_R的计算公式为
    Figure PCTCN2021137182-appb-100001
    其中，K pα与K pβ分别为静止坐标系下α轴与β轴的比例系数，K rα与K rβ分别为静止坐标系下α轴与β轴的谐振系数，ω 0为基波角频率，s为拉普拉斯算子，θ rp是差值矢量角且取值在-90°到0°之间折中选取，I αβ_E为电流误差值；
    将电流采样值作为解耦环节的输入，计算后得到解耦输出；
    将矢量角比例积分环节或者矢量角比例谐振环节的输出与解耦输出相加，作为延迟补偿环节的输入，将延迟补偿环节的输出作为控制环总输出；
    控制环总输出经过对应的dq/abc坐标或αβ/abc坐标变换得到三相调制波，并在调制与驱动模块中与载波比较，生成驱动信号驱动变流拓扑，实现电能变换。
  2. 根据权利要求1所述的，其特征在于，当进行矢量角比例积分控制时，在将电流误差值作为矢量角比例积分环节的输入，计算后得到矢量角比例积分环节的输出之前，还包括相位均衡环节的处理，引入相位均衡角，对电流误差i dq_E进行均衡，再将均衡后的电流误差作为矢量角比例积分环节的输入，计算后得到矢量角比例积分环节的输出m dq_R，矢量角比例积分环节的输出m dq_R的计算公式为m dq_R＝i dq_B·(K p+K i1·e jθi/s)，其中i dq_B＝i dq_E·e jθb，θ b是相位均衡角，i dq_B是相位均衡结果。
  3. 根据权利要求1所述的矢量角控制方法，其特征在于，当进行矢量角比例积分控制时，解耦输出m dq_D的计算公式为m dq_D＝i dq·jω 0L，其中L为交流侧电感值，ω 0为基波角频率。
  4. 根据权利要求3所述的矢量角控制方法，其特征在于，当进行矢量角比例积分控制时，延迟补偿环节的输出m dq的计算公式为m dq＝m dq_RD或m dq＝m dq_RD·e jnTs·ω0，其中m dq_RD＝m dq_R+m dq_D，T s为控制周期，n为补偿系数，ω 0为基波角频率。
  5. 根据权利要求1所述的矢量角控制方法，其特征在于，当进行矢量角比例谐振控制时，解耦输出M αβ_D的计算公式为
    Figure PCTCN2021137182-appb-100002
    其中，L α和L β分别为静止坐标系下α轴与β轴的等效电感值，ω 0为基波角频率。
  6. 根据权利要求5所述的矢量角控制方法，其特征在于，当进行矢量角比例谐振控制时，延迟补偿环节的的输出M αβ的计算公式为
    M αβ＝M αβ_RD，或
    Figure PCTCN2021137182-appb-100003
    其中，M αβ_RD＝M αβ_R+M αβ_D，n为补偿系数，T s为控制周期，ω 0为基波角频率。
PCT/CN2021/137182 2021-06-30 2021-12-10 矢量角控制方法 WO2023273184A1 (zh)

Priority Applications (1)

Application Number Priority Date Filing Date Title
JP2023514375A JP2023539674A (ja) 2021-06-30 2021-12-10 ベクトル角制御方法

Applications Claiming Priority (4)

Application Number Priority Date Filing Date Title
CN202110734368.0A CN113437895B (zh) 2021-06-30 2021-06-30 一种矩阵矢量角比例谐振控制方法
CN202110734358.7 2021-06-30
CN202110734368.0 2021-06-30
CN202110734358.7A CN113422533B (zh) 2021-06-30 2021-06-30 一种矢量角比例积分控制方法

Publications (1)

Publication Number Publication Date
WO2023273184A1 true WO2023273184A1 (zh) 2023-01-05

Family

ID=84691020

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2021/137182 WO2023273184A1 (zh) 2021-06-30 2021-12-10 矢量角控制方法

Country Status (2)

Country Link
JP (1) JP2023539674A (zh)
WO (1) WO2023273184A1 (zh)

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN107395040A (zh) * 2017-06-13 2017-11-24 东南大学 并网变流器复矢量pi控制器解耦与延时补偿方法
CN111313732A (zh) * 2020-02-25 2020-06-19 浙江大学 一种正负双边频域不对称下差异化相位校正的谐振控制方法
CN113422533A (zh) * 2021-06-30 2021-09-21 浙江大学 一种矢量角比例积分控制方法
CN113437895A (zh) * 2021-06-30 2021-09-24 浙江大学 一种矩阵矢量角比例谐振控制方法

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN107395040A (zh) * 2017-06-13 2017-11-24 东南大学 并网变流器复矢量pi控制器解耦与延时补偿方法
CN111313732A (zh) * 2020-02-25 2020-06-19 浙江大学 一种正负双边频域不对称下差异化相位校正的谐振控制方法
CN113422533A (zh) * 2021-06-30 2021-09-21 浙江大学 一种矢量角比例积分控制方法
CN113437895A (zh) * 2021-06-30 2021-09-24 浙江大学 一种矩阵矢量角比例谐振控制方法

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
ZHOU SIZHAN, LIU JINJUN, ZHANG YAN, ZHOU LINYUAN, TU YIMING: "Unbalanced Current Control Strategy Based on Complex-coefficient Proportional Integrator Controller Implemented in Double Synchronous Reference Frames", PROCEEDINGS OF THE CSEE, ZHONGGUO DIANJI GONGCHENG XUEHUI, CN, vol. 37, no. 18, 20 September 2017 (2017-09-20), CN , pages 5399 - 5408+5539, XP093018709, ISSN: 0258-8013, DOI: 10.13334/j.0258-8013.pcsee.161851 *

Also Published As

Publication number Publication date
JP2023539674A (ja) 2023-09-15

Similar Documents

Publication Publication Date Title
WO2021169666A1 (zh) 一种差异化相位校正的谐振控制方法
CN110224431B (zh) 并网逆变器系统中减小锁相环影响的控制方法
Yuan et al. Current harmonics elimination control method for six-phase PM synchronous motor drives
CN113422533B (zh) 一种矢量角比例积分控制方法
CN112653342B (zh) 一种静止坐标系下的复矢量电流环解耦控制装置及方法
CN113437895B (zh) 一种矩阵矢量角比例谐振控制方法
CN112436769A (zh) 一种永磁同步电机低载波比运行的控制系统及其方法
CN113036784A (zh) 一种基于滞后环节的柔直高频振荡控制方法及系统
CN112186804A (zh) 一种孤岛微电网母线电压不平衡和谐波补偿方法及系统
Dai et al. Multiple current harmonics suppression for low-inductance PMSM drives with deadbeat predictive current control
WO2023273184A1 (zh) 矢量角控制方法
Guan et al. Stability analysis of matrix converter with constant power loads and LC input filter
Liu et al. Discrete-time complex-vector model based closed-loop current controller for CSC-fed PMAC machine systems with low carrier ratios
CN110707908A (zh) 一种基于自适应电流谐波抑制的逆变器电流控制系统
CN113224793B (zh) 微电网多逆变器并联自适应谐波阻抗重塑控制方法及系统
Su et al. Stator flux trajectory control with optimized pulse patterns based on voltage command feed-forward
CN113839595B (zh) 双三相永磁同步电机谐波和不平衡电流抑制方法
CN113258615B (zh) 并网逆变器频率自适应控制方法、装置、设备及存储介质
CN112838601B (zh) 基于锁相优化的柔直输电系统高频振荡抑制方法及系统
CN113467239B (zh) 一种基于矩阵相位平移补偿器的比例谐振控制方法
CN110391675B (zh) 一种模块化多电平换流器输出电流闭环控制方法
CN115133834B (zh) 高速双三相永磁同步电机的二自由度谐波电流控制方法
CN210608909U (zh) 一种基于自适应电流谐波抑制的逆变器电流控制系统
Yuan et al. A novel current vector decomposition controller design for six-phase permanent magnet synchronous motor
Wang et al. Suppression Strategy of Differential Mode Circulating Current in Parallel Three-Phase PWM Converter

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 21948109

Country of ref document: EP

Kind code of ref document: A1

ENP Entry into the national phase

Ref document number: 2023514375

Country of ref document: JP

Kind code of ref document: A

NENP Non-entry into the national phase

Ref country code: DE