WO2023221541A1 - 一种5g毫米波双频带双模混频器及无线通信终端 - Google Patents

一种5g毫米波双频带双模混频器及无线通信终端 Download PDF

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Publication number
WO2023221541A1
WO2023221541A1 PCT/CN2023/072081 CN2023072081W WO2023221541A1 WO 2023221541 A1 WO2023221541 A1 WO 2023221541A1 CN 2023072081 W CN2023072081 W CN 2023072081W WO 2023221541 A1 WO2023221541 A1 WO 2023221541A1
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Prior art keywords
mos transistor
dual
drain
millimeter wave
mos tube
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PCT/CN2023/072081
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English (en)
French (fr)
Inventor
赵晨曦
康凯
余益明
刘辉华
吴韵秋
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电子科技大学
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Priority to US18/262,503 priority Critical patent/US12184317B2/en
Publication of WO2023221541A1 publication Critical patent/WO2023221541A1/zh

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • H04B1/22—Circuits for receivers in which no local oscillation is generated
    • H04B1/24—Circuits for receivers in which no local oscillation is generated the receiver comprising at least one semiconductor device having three or more electrodes
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1441—Balanced arrangements with transistors using field-effect transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1466—Passive mixer arrangements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/16—Multiple-frequency-changing
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • H04B1/1607—Supply circuits
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
    • Y02D30/00—Reducing energy consumption in communication networks
    • Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks

Definitions

  • the invention belongs to the field of wireless communication technology, and in particular relates to a 5G millimeter wave dual-band dual-mode mixer and a wireless communication terminal.
  • radio frequency chips are required to simultaneously process radio frequency information near two frequencies of 27GHz and 39GHz to achieve dual-band transceiver communication.
  • the receiver chip it not only needs to be able to amplify the signals of two frequency bands at the same time, but also needs to adopt a suitable frequency conversion method to convert the RF signals of the two frequency bands to an intermediate frequency signal without interfering with each other.
  • a single-frequency point local oscillator frequency conversion method is used.
  • the common implementation scheme is mainly a modified form of the Hartley structure receiver, as shown in Figure 2.
  • the structure in the figure is a dual-band down-conversion scheme after the Hartley structure receiver has been changed.
  • the two radio frequency signals are input into the upper and lower mixers respectively.
  • the 33GHz local oscillator signal is divided into two channels with a phase of -90° and 0° through a quadrature generator, and is input into the upper and lower mixers respectively.
  • the 33GHz phase-90° signal minus the 27GHz phase 0° signal is mixed to obtain a 6GHz phase-90° IF signal; the 39GHz phase 0° signal minus the 33GHz phase-90° signal , obtain a 6GHz intermediate frequency signal with a phase of 90°.
  • the above mixer output passes through a 90° phase shifter, and two 6GHz IF signals with phases of -180° and 0° can be obtained.
  • the following mixer has a local oscillator signal due to The phase is 0°, so the 27GHz and 39GHz signals will be down-converted to a 6GHz signal with a phase of 0°. After that, the upper and lower signals are combined.
  • the intermediate frequency signals obtained by 27GHz frequency conversion When the two channels of information are added, the intermediate frequency signals obtained by 27GHz frequency conversion have opposite phases, so they cancel each other out and are suppressed.
  • the phases of the intermediate frequency signals obtained by 39GHz frequency conversion are all 0°, so they can be combined and output. ;
  • the intermediate frequency signals obtained by 27GHz frequency conversion When the two channels of information are subtracted, the intermediate frequency signals obtained by 27GHz frequency conversion have the same phase, so they can be combined and output.
  • the phases of the intermediate frequency signals obtained by 39GHz frequency conversion are all 0°, and they are canceled out after subtraction, so they are suppressed.
  • the layout area is large and the layout and wiring are complicated.
  • This solution requires the signal to be divided into upper and lower channels for mixing and phase shifting respectively. Each channel has its own matching network.
  • the layout of the entire downconversion module will occupy a large area.
  • the signal input to the local oscillator mixer usually needs to be differential, which will make the layout and routing of the layout very complicated.
  • the power requirement for the local oscillator signal is relatively large.
  • This solution needs to drive the upper and lower mixers at the same time, which is 3dBm higher than the original local oscillator power, and the on-chip orthogonal generator in the millimeter wave band will introduce larger losses.
  • the local oscillator power needs to Further increase by more than 3dBm. Therefore, this solution requires a large local oscillator signal power, which will increase the design pressure of the local oscillator link and increase power consumption.
  • Processing errors easily deteriorate the image suppression.
  • This solution relies heavily on the performance of the local oscillator quadrature generator and the intermediate frequency 90° phase shifter.
  • the capacitors, inductors or other coupling structures used are very sensitive to processing and can easily introduce errors during the processing.
  • additional additional phase shift and amplitude imbalance may be produced, and after the processing of the 90° phase shifter, additional additional phase shift and additional insertion loss may be introduced.
  • the problems and defects of the existing technology are: the layout area of the existing technology is large and the layout and wiring are complicated; at the same time, the power demand for the local oscillator signal is large, and the processing error easily worsens the image suppression.
  • the present invention provides a 5G millimeter wave dual-band dual-mode mixer and a wireless communication terminal.
  • the invention is implemented as follows: a 5G millimeter wave dual-band dual-mode mixer.
  • the 5G millimeter-wave dual-band dual-mode mixer is provided with a first MOS tube; the first MOS tube communicates with the second MOS tube through the drain.
  • the tube is connected to the source of the third MOS tube, and the first MOS tube is connected to the drain of the fourth MOS tube through the drain;
  • the second MOS transistor is connected to one end of the first capacitor through the gate, and the other end of the first capacitor is connected to the drain of the third MOS transistor;
  • the third MOS transistor is connected to one end of the second capacitor through the gate, and the other end of the second capacitor is connected to the drain of the second MOS transistor.
  • the first MOS tube is connected to the ground through the source, the first MOS tube is connected to the radio frequency input signal through the gate, the fourth MOS tube is connected to Vdd through the source, and the fourth MOS tube is connected to the bias voltage Vb1 through the gate.
  • the second MOS tube and the third MOS tube are respectively connected to the positive and negative terminals of the local oscillator signal through the gate, and the second MOS tube and the third MOS tube are respectively connected to the two ports of the primary coil of the transformer through the drain.
  • the transformer is connected to Vdd in series with the first inductor through the center tap of the primary coil, the center tap of the secondary coil is grounded, and the ports on both sides of the secondary coil are respectively connected to the sources of the fifth MOS tube and the sixth MOS tube.
  • the gate of the sixth MOS transistor is connected to the bias voltage Vb2, and the drain of the sixth MOS transistor is connected to the drain of the eighth MOS transistor.
  • the gate of the fifth MOS transistor is connected to the bias voltage Vb2, and the drain of the fifth MOS transistor is connected to the drain of the seventh MOS transistor, the gate of the seventh MOS transistor, and the gate of the eighth MOS transistor.
  • the seventh MOS transistor and the eighth MOS transistor are connected to Vdd through their sources.
  • the sixth MOS transistor is connected to one end of the fourth capacitor through its drain, and the other end of the fourth capacitor is connected in series with a second inductor and connected to the first input end of the single-pole double-throw switch.
  • the transformer is connected to one end of the third capacitor through the center tap of the primary coil, and the third capacitor The other end of the capacitor is connected to the second input end of the single-pole double-throw switch, and the output end of the single-pole double-throw switch is connected to the intermediate frequency output port.
  • Another object of the present invention is to provide a receiver chip for 5G millimeter wave communication.
  • the receiver chip for 5G millimeter wave communication includes the 5G millimeter wave dual-band dual-mode mixer.
  • the 5G millimeter wave communication receiver chip The receiver chip works simultaneously covering two millimeter wave frequency bands near 27 and 39GHz.
  • Another object of the present invention is to provide a wireless communication terminal equipped with the 5G millimeter wave dual-band dual-mode mixer according to any one of claims 1 to 8.
  • the invention uses a mixer core to mix the lower frequency band of the radio frequency dual-band with the fundamental wave of the local oscillator to generate a differential mode signal, and mix the higher frequency band with the second harmonic of the local oscillator to generate a common mode signal. , then extract the common mode signal through the center tap of the transformer, extract the differential mode signal through the coil coupling characteristics of the transformer, and separate the mixed intermediate frequencies of the two frequency bands. After the differential mode signal is converted into a single-ended signal through the active balun, the differential mode signal and the common mode signal are input into a single-pole double-throw switch respectively to select the output signal.
  • This invention uses the differential mode and common mode extraction method of the transformer to distinguish the frequency-converted information of the two frequency bands. Compared with the traditional discrimination method of a quadrature generator combined with a 90° phase shifter, the frequency deviation and impedance fluctuation caused by processing errors are better. It is less sensitive, so it can suppress the mutual interference between the two frequency bands more stably; at the same time, it avoids the larger layout area and more complicated layout of the traditional solution.
  • the present invention uses an active balun with common gate input to connect to the secondary coil of the transformer, which can maintain a balanced and similar input impedance within a wide intermediate frequency band and ensure the stability of the differential mode and common mode extraction of the transformer; it uses an LC network Connecting to the center tap of the primary coil of the transformer can form a high-order matching network with the transformer, so that the output impedance of the common-mode signal can be in a wider range. Very flat across the IF bandwidth.
  • the core part of the mixer of the present invention adopts a current injection active mixer with high gain, and the introduction of a neutralizing capacitor improves the stability of the core part of the mixer.
  • the main purpose of this invention is to solve the problem of how to realize down-conversion of information in frequency bands near 27GHz and 39GHz in the application of 5G millimeter wave communication technology, so as to effectively suppress the in-band image frequency without introducing large area and power consumption.
  • 5G 5G millimeter wave frequency bands
  • many countries have allocated two frequency bands near 27GHz and 39GHz for 5G communications.
  • China's 5G millimeter wave frequency bands are 24.25-27.5GHz and 37.5-42.5GHz.
  • the information of these two frequency bands is processed simultaneously and converted to a lower-frequency intermediate frequency signal. It is inevitable to suppress the image frequency in the frequency band to prevent it from interfering with the main signal.
  • the application of traditional image frequency suppression technology to the 5G millimeter wave band may cause problems such as excessive filter area, excessive local oscillator signal bandwidth, and increased link power consumption.
  • the present invention proposes a new structure downmixer, which is based on a single balanced mixer structure and has two mixing modes of local oscillator fundamental wave mixing and local oscillator second harmonic mixing, corresponding to 27GHz and 39GHz respectively.
  • the frequency bands represented by each frequency are then extracted by differential mode and common mode to output intermediate frequencies respectively.
  • the present invention only uses one mixer to realize the down-conversion of the two frequency band signals and suppresses the mutual interference between the two frequency band signals. , and will not introduce large power and area consumption.
  • the present invention improves the traditional single-balanced active mixer and uses the coupling method of the load transformer and the center tap extraction method to separately mix the differential mode signal mixed with the fundamental wave of the local oscillator and the second harmonic of the local oscillator.
  • the common-mode signal extraction of mixing corresponds to the down-conversion of the two frequency band signals of 5G communication, and then the active balun, LC matching network and single-pole double-throw switch are used to select different signal outputs, thus achieving a solution that can be applied to 5G Dual-mode downmixer for millimeter-wave dual-band receivers.
  • the radio frequency of the present invention covers the two frequency bands of 5G millimeter waves, the local oscillator bandwidth and the intermediate frequency bandwidth are relatively narrow and will not introduce a large link burden.
  • the present invention does not require the introduction of a quadrature generator, a second mixer and a 90° phase shifter. While reducing the local oscillator power demand, layout area and layout complexity, it can better Suppresses the interference between the two frequency bands, and the image frequencies are outside the frequency band, so it is more suitable for 5G millimeter Meewave applications.
  • the expected income and commercial value after the transformation of the technical solution of the present invention is:
  • the present invention is applied to 5G millimeter wave dual-band receiver chips, which can save chip area and reduce chip power consumption.
  • the technical solution of the present invention solves a technical problem that people have been eager to solve but have never succeeded: the traditional dual-band receiver frequency conversion module cannot be realized by a single mixer core, and often requires the cooperation of two mixers The completion of the local oscillator quadrature generator and 90° phase shifter will make the layout more complex and the layout area larger.
  • the present invention only requires one mixer core and does not require complex layout and large layout area.
  • Figure 1 is a schematic structural diagram of a 5G millimeter wave dual-band dual-mode mixer provided by an embodiment of the present invention
  • Figure 2 is a schematic circuit diagram of a dual-band down-conversion scheme with image suppression function provided by an embodiment of the present invention
  • Figure 3 is the principle of differential mode and common mode extraction by the transformer provided by the embodiment of the present invention.
  • Figure 3 Figure a, common mode; Figure b, differential mode;
  • the source of the first MOS transistor 1 is connected to the ground, the gate is connected to the radio frequency input signal, and the drain is connected to the second MOS transistor 2 and the third MOS transistor 2.
  • the sources of the three MOS tubes 3 are connected; the source of the fourth MOS tube 4 is connected to Vdd, the gate is connected to the bias voltage Vb1, and the drain is connected to the drain of the first MOS tube 1; the second MOS tube 2 and the third MOS tube
  • the gates of 3 are respectively connected to the positive and negative terminals of the local oscillator signal, and the drains are respectively connected to the two ports of the primary coil of transformer 16.
  • Transformer 16 is transformer Transformer1; the center tap of the primary coil of transformer 16 is connected in series with the first inductor 13. Enter Vdd, the center tap of the secondary coil is grounded, and the ports on both sides of the secondary coil are connected to the sources of the fifth MOS transistor 5 and the sixth MOS transistor 6 respectively; one end of the first capacitor 9 is connected to the gate of the second MOS transistor 2.
  • One end is connected to the drain of the third MOS transistor 3; one end of the second capacitor 10 is connected to the gate of the third MOS transistor 3, and the other end is connected to the drain of the second MOS transistor 2; the gate of the sixth MOS transistor 6 is connected to the bias voltage Vb2 , the drain is connected to the drain of the eighth MOS transistor 8; the gate of the fifth MOS transistor 5 is connected to the bias voltage Vb2, and the drain is connected to the drain of the seventh MOS transistor 7, the gate of the seventh MOS transistor 7 and the eighth The gate of the MOS transistor 8; the sources of the seventh MOS transistor 7 and the eighth MOS transistor 8 are connected to Vdd; one end of the fourth capacitor 12 is connected to the drain of the sixth MOS transistor 6, and one end is connected in series with the second inductor 14 to connect to SPDT.
  • the first input end of the switch 15; one end of the third capacitor 11 is connected to the center tap of the primary coil of the transformer 16, and one end is connected to the second input end of the single pole double throw switch 15.
  • the output end of the single pole double throw switch 15 is connected to the intermediate frequency output. port.
  • the working principle of the present invention is: the first MOS transistor 1, the second MOS transistor 2, the third MOS transistor 3, the fourth MOS transistor 4 and the first capacitor 9 and the second capacitor 10 constitute the core part of the mixer.
  • Current injection Type single balanced active mixer structure The first MOS transistor 1 serves as the transconductance stage of the mixer to provide gain for the mixer, and the fourth MOS transistor 4 serves as a current injection structure to increase the drain current of the first MOS transistor 1, further increasing the gain of the mixer.
  • the second MOS transistor 2 and the third MOS transistor 3 are switching transistors of the mixer.
  • the phase difference of the local oscillator signal connected to the gate is 180°. Since the local oscillator signal is large, signal transmission under the influence of nonlinearity needs to be considered.
  • V ds is the drain-source AC voltage difference of the second MOS transistor 2 or the third MOS transistor 3
  • V gs is the gate-source AC voltage difference of the second MOS transistor 2 or the third MOS transistor 3, which is affected by the local oscillator signal.
  • a i is the coefficient of the relevant nonlinearity.
  • V gs Acos( ⁇ LO t)
  • V gs -Acos( ⁇ LO t)
  • A is the local oscillator signal voltage amplitude
  • ⁇ LO is the local oscillator The angular frequency of the signal.
  • the primary term mixes with the fundamental wave of the local oscillator.
  • the second MOS transistor 2 and the third MOS transistor 3 are turned on and off in sequence to mix with the radio frequency signal entering from the source, and output b 1 cos ( ⁇ RF - from the drain respectively.
  • ⁇ LO ⁇ LO
  • -b 1 cos( ⁇ RF - ⁇ LO )
  • b 1 is the voltage amplitude of the two signals
  • ⁇ RF is the angular frequency of the radio frequency signal
  • the two signals are equal in amplitude and in phase, as Differential mode signal output.
  • the quadratic term mixes with the second harmonic of the local oscillator. Substitute it into the V gs expression. Since the square results of A and -A are the same, there is no phase difference.
  • the drain of the second MOS tube 2 or the third MOS tube 3 is mixed.
  • a signal of b 2 cos (2 ⁇ LO - ⁇ RF ) is generated at a high frequency and is output as a common mode signal.
  • b 2 is the voltage amplitude of this signal.
  • the gate DC voltage of the second MOS transistor 2 or the third MOS transistor 3 needs to be biased at a potential near the threshold voltage where the second harmonic is relatively large.
  • the first capacitor 9 and the second capacitor 10 are neutralizing capacitors and are used to ensure the working stability of the second MOS transistor 2 and the third MOS transistor 3 of the mixer switch.
  • Transformer 16 (Transformer1) is used as the load of the mixer core on the one hand to match the intermediate frequency signal of 4-6GHz, and on the other hand it is used to extract differential mode signals and common mode signals.
  • the extraction principle is shown in Figure 3.
  • the primary coil is black and is connected to the core of the mixer.
  • the secondary coil is gray and is connected to the sources of the fifth MOS tube 5 and the sixth MOS tube 6.
  • the common mode signal enters the primary coil, as shown in figure a in Figure 3
  • the two signals go through half a cycle of the primary coil respectively.
  • the center tap since the two signals are of equal amplitude and in phase, they are directly superimposed and strengthened, and are extracted from the center tap.
  • the common mode signal will not couple to the secondary coil and therefore will not be output from the secondary coil.
  • the differential mode signal enters the primary coil, as shown in figure b in Figure 3, when the two signals go through half a cycle to the center tap, since the two signals are equal in amplitude and in phase, they have a canceling effect.
  • the center tap of the primary coil See the AC ground so the signal doesn't flow out of the center tap.
  • the differential mode signal will travel in a large circle from one port of the primary coil to the other. This complete circle will couple the electromagnetic energy to the secondary coil and output it from the two ports of the secondary coil.
  • the first inductor 13 is the load of the common mode intermediate frequency signal
  • the third capacitor 11 is a DC blocking capacitor. The two form an LC matching network to introduce the 4-6 GHz intermediate frequency signal into the SPDT switch 15 .
  • the first MOS transistor 1 to the eighth MOS transistor 8, the fourth capacitor 12, and the second inductor 14 form an active balun.
  • the differential mode intermediate frequency signal enters the fifth MOS transistor 5 and the sixth MOS transistor respectively from the two output ports of the secondary coil.
  • the source of the MOS transistor 6, the fifth MOS transistor 5 and the sixth MOS transistor 6 have a common gate structure, and amplify and output the signal from the drain. After the signal output by the drain of the fifth MOS tube 5 enters the gate of the eighth MOS tube 8, it will undergo another inversion and be output from the drain of the eighth MOS tube 8. Therefore, the sixth MOS tube 6 and the eighth MOS tube The drain output signal of 8 is in phase and enters the fourth capacitor 12 after superposition.
  • the fourth capacitor 12 is a DC-blocking capacitor.
  • the second inductor 14 and the fourth capacitor 12 form a matching network to introduce the 4-6GHz intermediate frequency signal into the single-pole double-throw switch and at the same time adjust the input impedance balance of the active balun.
  • the size of the fifth MOS tube 5 is slightly larger than the sixth MOS tube 6, and the size of the seventh MOS tube 7 is slightly larger than the eighth MOS tube 8, so that the input impedances of the sources of the fifth MOS tube 5 and the sixth MOS tube 6 are similar.
  • the single-pole double-throw switch 15 is used to select whether the differential mode extraction or common mode extraction signal is output to the intermediate frequency.
  • the image frequency of the 27GHz radio frequency signal is 17GHz
  • the image frequency of the 39GHz radio frequency signal is 49GHz, both of which are outside the frequency band and can be filtered out through the matching network of the front stage.
  • the radio frequency end of the mixer can cover f if1 +f lo1 ⁇ f if2 +f lo2 and 2 ⁇ f lo1 -f if2 ⁇ 2 ⁇ f lo2 -f if1 two frequency bands.
  • the IF signal is 4-6GHz and the local oscillator signal is 20-24GHz, it can cover the two millimeter wave frequency bands of my country's 5G communication, 24.25-27.5GHz and 37.5-42.5GHz.
  • this section is an application example of the claimed technical solution in specific products or related technologies.
  • this invention It can be applied to receiver chips for 5G millimeter wave communications, so that the chip can work simultaneously covering two millimeter wave frequency bands near 27 and 39 GHz.
  • the present invention improves the traditional single-balanced active mixer and uses the coupling method of the load transformer and the center tap extraction method to separately mix the differential mode signal mixed with the fundamental wave of the local oscillator and the second harmonic of the local oscillator.
  • the common-mode signal extraction of mixing corresponds to the down-conversion of the two frequency band signals of 5G communication, and then the active balun, LC matching network and single-pole double-throw switch are used to select different signal outputs, thus achieving a solution that can be applied to 5G Dual-mode downmixer for millimeter-wave dual-band receivers.
  • the radio frequency of the present invention covers the two frequency bands of 5G millimeter waves, the local oscillator bandwidth and the intermediate frequency bandwidth are relatively narrow and will not introduce a large link burden.
  • the present invention does not require the introduction of a quadrature generator, a second mixer and a 90° phase shifter. While reducing the local oscillator power demand, layout area and layout complexity, it can better The interference between the two frequency bands is suppressed, and the image frequencies are outside the frequency band, so it is more suitable for 5G millimeter wave applications.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Superheterodyne Receivers (AREA)

Abstract

本发明属于无线通信技术领域，公开了一种5G毫米波双频带双模混频器及无线通信终端，所述5G毫米波双频带双模混频器中第一MOS管通过漏极与第二MOS管和第三MOS管的源极相连，第一MOS管通过漏极与第四MOS管的漏极连接；第二MOS管通过栅极与第一电容一端连接，第一电容另一端与第三MOS管的漏极连接；第三MOS管通过栅极与第二电容一端连接，第二电容另一端与第二MOS管的漏极连接。本发明基于单平衡混频器结构，具有本振基波混频和本振二次谐波混频两种混频模式，通过差模和共模的提取，输出中频，使用一个混频器实现了两个频带信号的下变频，抑制了两个频带信号之间的相互干扰，不会引入较大的功率、面积消耗。

Description

一种5G毫米波双频带双模混频器及无线通信终端 技术领域
本发明属于无线通信技术领域，尤其涉及一种5G毫米波双频带双模混频器及无线通信终端。
背景技术
目前，在5G毫米波技术的应用中，需要射频芯片同时处理27GHz和39GHz两个频率附近的射频信息，实现双频带的收发通信。对于接收机芯片，不仅需要能够同时放大两个频带的信号，还需要采取合适的变频方式，将两个频带的射频信号互不干扰地下变频到中频信号。以中国的5G毫米波频段24.25-27.5GHz和37.5-42.5GHz为例，采用单频点本振的变频方式，若本振频率小于24.25GHz，则双频带下变频后，中频的上限将超过42.5GHz-24.25GHz＝18.25GHz，中频过高增加中频链路的设计难度；若本振频率位于两个频带之间，则会引入镜频问题，在频谱上关于本振频率对称的两个频率会下变频到同一个频率的中频，造成两组信号相互混叠，影响了后续的信号解调。因此，对于双频带接收机的下变频，需要减轻中频和本振的带宽设计压力，同时，避免镜频问题。常见的实现方案主要是Hartley结构接收机的改变形式，如图2所示。图中的结构为Hartley结构接收机改变形式后的双频带下变频方案。以5G应用中27GHz和39GHz两个射频频率的输入为例；在射频信号到中频信号的变化中，27GHz信号为黑色，39GHz信号为灰色。两个射频信号分别输入上下两个混频器，33GHz的本振信号通过一个正交发生器分成相位-90°和0°的两路，分别输入上下两个混频器。对于上面的混频器，33GHz相位-90°的信号减去27GHz相位0°的信号，混频得到6GHz相位-90°的中频信号；39GHz相位0°的信号减去33GHz相位-90°的信号，得到6GHz相位90°的中频信号。上面的混频器输出经过一个90°移相器，可以得到两个相位分别为-180°和0°的6GHz中频信号。下面的混频器由于本振信号 相位为0°，所以27GHz和39GHz的信号都会下变频得到6GHz相位为0°的信号。之后上下两路信号进行合路，当两路信息相加时，27GHz变频得到的中频信号相位相反，所以相互抵消，被抑制了，39GHz变频得到的中频信号相位都是0°，所以可以合并输出；当两路信息相减时，27GHz变频得到的中频信号相位相同，所以可以合并输出，39GHz变频得到的中频信号相位都是0°，相减后抵消掉，所以被抑制了。同理类比到27GHz和39GHz附近频段内的其它频点，都可以使用此结构在下变频的同时，实现镜频抑制。上述的一种下变频方案虽然可以实现双频带下变频，并且能够镜像抑制，但是仍然存在很多问题，相应的缺点如下：(1)版图面积较大，布局布线较复杂。这种方案需要把信号分成上下两路，分别进行混频和移相，每一路有各自的匹配网络，结合本振端引入的正交发生器，整个下变频模块的版图占用面积会很大。版图绘制过程中需要保证上下两个三端口混频器的走线非常平衡，本振输入混频器的信号通常需要是差分的，这些都会导致版图的布局布线非常复杂。(2)对本振信号的功率需求较大。这一方案需要同时驱动上下两个混频器，比原本的本振功率高3dBm，并且片上做毫米波频段的正交发生器会引入较大的损耗，为了补偿这一损耗，本振功率需要进一步提高3dBm以上。因此，此方案需要较大的本振信号功率，这会增大本振链路的设计压力、增加功耗。(3)加工误差容易恶化镜频抑制度。这一方案非常依赖本振正交发生器和中频90°移相器的性能，这其中所使用的电容、电感或其它耦合结构，对工艺加工非常敏感，容易在加工过程中引入误差。正交发生器加工后可能产生额外的附加相移、幅度不平衡性，90°移相器加工后可能引入附加相移、附加插损。这些幅度、相位和误差会使得两路信号在输出端加减时无法做到完全抵消，进而恶化镜频抑制度。
通过上述分析，现有技术存在的问题及缺陷为：现有技术版图面积较大，布局布线较复杂；同时对本振信号的功率需求较大，加工误差容易恶化镜频抑制度。
发明内容
针对现有技术存在的问题，本发明提供了一种5G毫米波双频带双模混频器及无线通信终端。
本发明是这样实现的，一种5G毫米波双频带双模混频器，所述5G毫米波双频带双模混频器设置有第一MOS管；第一MOS管通过漏极与第二MOS管和第三MOS管的源极相连，第一MOS管通过漏极与第四MOS管的漏极连接；
第二MOS管通过栅极与第一电容一端连接，第一电容另一端与第三MOS管的漏极连接；
第三MOS管通过栅极与第二电容一端连接，第二电容另一端与第二MOS管的漏极连接。
进一步，所述第一MOS管通过源极接地，第一MOS管通过栅极接射频输入信号，第四MOS管通过源极接Vdd，第四MOS管通过栅极接偏置电压Vb1。
进一步，所述第二MOS管和第三MOS管通过栅极分别连接本振信号的正负端，第二MOS管和第三MOS管通过漏极分别接入变压器的初级线圈的两个端口。
进一步，所述变压器通过初级线圈的中心抽头串联第一电感后接入Vdd，次级线圈的中心抽头接地，次级线圈两侧的端口分别连接第五MOS管和第六MOS管的源极。
进一步，所述第六MOS管通过栅极接偏置电压Vb2，第六MOS管通过漏极接第八MOS管的漏极。
进一步，所述第五MOS管通过栅极接偏置电压Vb2，第五MOS管通过漏极接第七MOS管的漏极、第七MOS管的栅极和第八MOS管的栅极。
进一步，所述第七MOS管和第八MOS管通过源极接Vdd。
进一步，所述第六MOS管通过漏极与第四电容一端连接，第四电容另一端串联第二电感接入单刀双掷开关的第一个输入端。
进一步，所述变压器通过初级线圈的中心抽头与第三电容一端连接，第三 电容另一端连接单刀双掷开关的第二个输入端，单刀双掷开关的输出端接入中频输出端口。
本发明的另一目的在于提供一种5G毫米波通信的接收机芯片，所述5G毫米波通信的接收机芯片包含所述的5G毫米波双频带双模混频器，所述5G毫米波通信的接收机芯片同时工作覆盖27和39GHz附近的两个毫米波频带。
本发明的另一目的在于提供一种无线通信终端，所述无线通信终端安装有权利要求1～8任意一项所述的5G毫米波双频带双模混频器。
结合上述的技术方案和解决的技术问题，请从以下几方面分析本发明所要保护的技术方案所具备的优点及积极效果为：
第一、针对上述现有技术存在的技术问题以及解决该问题的难度，紧密结合本发明的所要保护的技术方案以及研发过程中结果和数据等，详细、深刻地分析本发明技术方案如何解决的技术问题，解决问题之后带来的一些具备创造性的技术效果。具体描述如下：
本发明通过混频器核心将射频双频带里频率较低的频带与本振的基波混频产生差模信号，将频率较高的频率与本振的二次谐波混频产生共模信号，再通过变压器的中心抽头提取共模信号，通过变压器的线圈耦合特性提取差模信号，将两个频带混频后的中频区分开。差模信号通过有源巴伦转换为单端信号后，与共模信号分别输入单刀双掷开关，以选择输出的信号。通过这种方式实现了5G毫米波双频带信号的下变频，并且镜像频率都在带外，避免了镜频抑制问题。本发明采用变压器的差模、共模提取方式区分两个频带变频后的信息，相比传统的正交发生器结合90°移相器的区分方式，对于加工误差带来的频偏和阻抗波动更不敏感，因此可以更稳定地抑制两个频带之间的相互干扰；同时避免了传统方案较大的版图面积和较复杂的版图走线布局。本发明采用共栅极输入的有源巴伦接入变压器的次级线圈，可以保住较宽中频频带内平衡且相近的输入阻抗，确保变压器差模共模提取时的稳定性；采用LC网络接入变压器初级线圈的中心抽头，可以和变压器组成高阶匹配网络，使得共模信号的输出阻抗在较宽 中频带宽内非常平坦。本发明的混频器的核心部分采用电流注入型有源混频器，增益较高，引入中和电容提高了混频器核心部分的稳定性。
本发明主要为了解决在5G毫米波通信技术的应用中，如何实现将27GHz和39GHz附近频段的信息实现下变频，在有效抑制带内镜像频率的同时，不会引入较大的面积和功率的消耗。随着5G的发展，许多国家将27GHz和39GHz附近的两个频段划分应用于了5G通信中，如，中国的5G毫米波频段为24.25-27.5GHz和37.5-42.5GHz。在5G毫米波接收机芯片中，同时处理这两个频带的信息，并变频到频率较低的中频信号，不可避免的需要抑制频带内的镜像频率，防止其干扰主信号。传统的镜像频率抑制技术应用于5G毫米波频段可能带来滤波器面积过大、本振信号带宽过宽、链路功耗增加等问题。本发明提出了一种新结构的下混频器，基于单平衡混频器结构，具有本振基波混频和本振二次谐波混频两种混频模式，分别对应27GHz和39GHz两个频率所代表的频段，再通过差模和共模的提取，分别输出中频，本发明仅使用一个混频器实现了两个频带信号的下变频，抑制了两个频带信号之间的相互干扰，并且不会引入较大的功率、面积消耗。
第二，把技术方案看做一个整体或者从产品的角度，本发明所要保护的技术方案具备的技术效果和优点，具体描述如下：
本发明通过对传统的单平衡有源混频器进行改进，通过负载变压器的耦合方式和中心抽头抽取方式，可以分别将与本振基波混频的差模信号和与本振二次谐波混频的共模信号提取，对应了5G通信的两个频带信号的下变频，再通过有源巴伦、LC匹配网络和单刀双掷开关以选择不同的信号输出，从而实现了可以应用于5G毫米波双频带接收机的双模下混频器。本发明在射频频率覆盖了5G毫米波的两个频带的同时，本振带宽和中频带宽相对较窄，不会引入较大的链路负担。本发明相较传统方案不需要引入正交发生器、第二个混频器和90°移相器，在减小了本振功率需求、版图面积和版图布局复杂度的同时，可以较好的抑制两个频带之间的干扰，并且镜像频率都在频带外，因此更适用于5G毫 米波的应用。
第三，作为本发明的权利要求的创造性辅助证据，还体现在以下几个重要方面：
(1)本发明的技术方案转化后的预期收益和商业价值为：本发明应用于5G毫米波双频带接收机芯片中，可以节约芯片面积，降低芯片功耗。
(2)本发明的技术方案解决了人们一直渴望解决、但始终未能获得成功的技术难题：传统双频带接收机变频模块无法通过单一的混频器核心实现，往往需要两个混频器配合本振正交发生器、90°移相器完成，会使得版图布局较为复杂、版图面积较大。本发明仅需要一个混频器核心，不需要复杂的版图布局和较大的版图面积。
附图说明
图1是本发明实施例提供的5G毫米波双频带双模混频器结构示意图；
图2是本发明实施例提供的带镜频抑制功能的双频带下变频方案电路示意图；
图3是本发明实施例提供的变压器进行差模共模提取的原理；
图3中：图a、共模；图b、差模；
图中：1、第一MOS管；2、第二MOS管；3、第三MOS管；4、第四MOS管；5、第五MOS管；6、第六MOS管；7、第七MOS管；8、第八MOS管；9、第一电容；10、第二电容；11、第三电容；12、第四电容；13、第一电感；14、第二电感；15、单刀双掷开关；16、变压器。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅仅用以解释本发明，并不用于限定本发明。
一、解释说明实施例。为了使本领域技术人员充分了解本发明如何具体实现，该部分是对权利要求技术方案进行展开说明的解释说明实施例。
如图1所示，本发明实施例提供的5G毫米波双频带双模混频器中第一MOS管1的源极接地，栅极接射频输入信号，漏极与第二MOS管2和第三MOS管3的源极相连；第四MOS管4的源极接Vdd，栅极接偏置电压Vb1，漏极连接第一MOS管1的漏极；第二MOS管2和第三MOS管3的栅极分别连接本振的信号的正负端，漏极分别接入变压器16的初级线圈的两个端口，变压器16为变压器Transformer1；变压器16初级线圈的中心抽头串联第一电感13后接入Vdd，次级线圈的中心抽头接地，次级线圈两侧的端口分别连接第五MOS管5和第六MOS管6的源极；第一电容9一端连接第二MOS管2的栅极，一端连接第三MOS管3的漏极；第二电容10一端连接第三MOS管3的栅极，一端连接第二MOS管2的漏极；第六MOS管6的栅极接偏置电压Vb2，漏极接第八MOS管8的漏极；第五MOS管5的栅极接偏置电压Vb2，漏极接第七MOS管7的漏极、第七MOS管7的栅极和第八MOS管8的栅极；第七MOS管7和第八MOS管8的源极接Vdd；第四电容12一端连接第六MOS管6的漏极，一端串联第二电感14接入单刀双掷开关15的第一个输入端；第三电容11一端连接变压器16的初级线圈的中心抽头，一端连接单刀双掷开关15的第二个输入端，单刀双掷开关15的输出端接入中频输出端口。
本发明的工作原理为：第一MOS管1、第二MOS管2、第三MOS管3、第四MOS管4和第一电容9、第二电容10构成混频器的核心部分，电流注入型单平衡有源混频器结构。第一MOS管1作为混频器的跨导级为混频器提供增益，第四MOS管4作为电流注入结构提高第一MOS管1的漏极电流，使得混频器的增益进一步提高。第二MOS管2、第三MOS管3为混频器的开关管，栅极所接本振信号相位差为180°，由于本振信号较大，所以需要考虑在非线性影响下的信号传递表达式：
其中，Vds是第二MOS管2或第三MOS管3的漏源交流电压差，Vgs是第二MOS管2或第三MOS管3的栅源交流电压差，受本振信号影响，ai是相关非线性的系数。对于第二MOS管2，Vgs＝Acos(ωLOt)，对于第三MOS管3，Vgs＝-Acos(ωLOt)，其中，A为本振信号电压幅度，ωLO是本振信号的角频率。一次项与本振的基波进行混频，第二MOS管2、第三MOS管3依次通断与从源极进入的射频信号进行混频，分别从漏极输出b1cos(ωRF-ωLO)和-b1cos(ωRF-ωLO)的两个信号，其中，b1为两个信号的电压幅度，ωRF是射频信号的角频率，两个信号等幅反相，作为差模信号输出。二次项与本振的二次谐波混频，代入Vgs表达式，由于A和-A的平方结果相同，所以没有相位差别，第二MOS管2或第三MOS管3的漏极混频产生b2cos(2ωLO-ωRF)的信号，作为共模信号输出，b2为此信号的电压幅度。为了确保二次混频的增益，第二MOS管2或第三MOS管3的栅极直流电压需要偏置在阈值电压附近二次谐波比较大的电位。第一电容9和第二电容10为中和电容，用来保证混频器开关管第二MOS管2和第三MOS管3的工作稳定性。
变压器16(变压器Transformer1)一方面用作混频器核心的负载，匹配4-6GHz的中频信号，另一方面用作差模信号和共模信号的提取，其提取原理如图3所示。在图3中，初级线圈为黑色，连接混频器核心，次级线圈为灰色，连接第五MOS管5、第六MOS管6的源极。当共模信号进入初级线圈，如图3中的图a所示，两个信号分别走过初级线圈的半周，在中心抽头处由于两者等幅同相，所以直接叠加加强，从中心抽头提取出来，而由于变压器16的对称性和平衡性，共模信号不会耦合到次级线圈，因此不会从次级线圈输出。当差模信号进入初级线圈器，如图3中的图b所示，当两个信号走过半周到中心抽头时，由于两个信号等幅反相，呈抵消的效果，在初级线圈的中心抽头处看到交流地，因此信号不会从中心抽头流出。差模信号会在初级线圈的一个端口到另一个端口走一个大圈，这个完整的圈会将电磁能量耦合到次级线圈，从次级线圈的两个端口输出。以本振22GHz为例，27GHz的射频信号可以混频产生 27GHz-22GHz＝5GHz的差模信号，这个信号会从变压器16的次级线圈耦合出来；39GHz的射频信号可以混频产生2×22GHz-39GHz＝5GHz的共模信号，通过初级线圈的中心抽头提取。由此，实现两个频带下混频后分别的差模、共模提取。第一电感13为共模中频信号的负载，第三电容11为阻隔直流的电容，两者形成LC匹配网络将4-6GHz的中频信号引入单刀双掷开关15。第一MOS管1～第八MOS管8和第四电容12、第二电感14构成有源巴伦，差模中频信号从次级线圈的两个输出端口分别进入第五MOS管5和第六MOS管6的源极，第五MOS管5和第六MOS管6呈共栅极结构，将信号从漏极放大输出。第五MOS管5的漏极输出的信号进入第八MOS管8的栅极后会再经过一次反相，从第八MOS管8的漏极输出，因此第六MOS管6和第八MOS管8的漏极输出信号同相，叠加后进入第四电容12。第四电容12为阻隔直流的电容，第二电感14与第四电容12组成匹配网络，将4-6GHz的中频信号引入单刀双掷开关，同时，调节有源巴伦的输入阻抗平衡性。第五MOS管5的尺寸略大于第六MOS管6，第七MOS管7的的尺寸略大于第八MOS管8，以使得第五MOS管5和第六MOS管6源极的输入阻抗相近。单刀双掷开关15用于选择差模提取或者共模提取的信号输出到中频。当本振信号22GHz时，27GHz射频信号的镜像频率为17GHz，39GHz射频信号的镜像频率为49GHz，都在频带外，可以通过前级的匹配网络滤除掉。这种通过差模共模提取进行基波混频和两次谐波混频切换的方式，使得本振在频率不变的情况下，只需要切换模式，即可将两个不同频带射频的信息下变到相同的中频，中频和本振的带宽设计压力大大降低。当中频带宽为fif1～fif2、本振范围为flo1～flo2时，则本混频器的射频端可以覆盖fif1+flo1～fif2+flo2和2×flo1-fif2～2×flo2-fif1两个频段。若中频信号取4-6GHz，本振信号取20-24GHz，即可覆盖我国5G通信的两个毫米波频段24.25-27.5GHz和37.5-42.5GHz。
二、应用实施例。为了证明本发明的技术方案的创造性和技术价值，该部分是对权利要求技术方案进行具体产品上或相关技术上的应用实施例。本发明 可以应用于5G毫米波通信的接收机芯片中，使得该芯片可以同时工作覆盖27和39GHz附近的两个毫米波频带。
三、实施例相关效果的证据。本发明实施例在研发或者使用过程中取得了一些积极效果，和现有技术相比的确具备很大的优势，下面内容结合试验过程的数据、图表等进行描述。
本发明通过对传统的单平衡有源混频器进行改进，通过负载变压器的耦合方式和中心抽头抽取方式，可以分别将与本振基波混频的差模信号和与本振二次谐波混频的共模信号提取，对应了5G通信的两个频带信号的下变频，再通过有源巴伦、LC匹配网络和单刀双掷开关以选择不同的信号输出，从而实现了可以应用于5G毫米波双频带接收机的双模下混频器。本发明在射频频率覆盖了5G毫米波的两个频带的同时，本振带宽和中频带宽相对较窄，不会引入较大的链路负担。本发明相较传统方案不需要引入正交发生器、第二个混频器和90°移相器，在减小了本振功率需求、版图面积和版图布局复杂度的同时，可以较好的抑制两个频带之间的干扰，并且镜像频率都在频带外，因此更适用于5G毫米波的应用。
以上所述，仅为本发明的具体实施方式，但本发明的保护范围并不局限于此，任何熟悉本技术领域的技术人员在本发明揭露的技术范围内，凡在本发明的精神和原则之内所作的任何修改、等同替换和改进等，都应涵盖在本发明的保护范围之内。

Claims (10)

  1. 一种5G毫米波双频带双模混频器，其特征在于，所述5G毫米波双频带双模混频器设置有：
    第一MOS管；
    第一MOS管通过漏极与第二MOS管和第三MOS管的源极相连，第一MOS管通过漏极与第四MOS管的漏极连接；
    第二MOS管通过栅极与第一电容一端连接，第一电容另一端与第三MOS管的漏极连接；
    第三MOS管通过栅极与第二电容一端连接，第二电容另一端与第二MOS管的漏极连接。
  2. 如权利要求1所述5G毫米波双频带双模混频器，其特征在于，所述第一MOS管通过源极接地，第一MOS管通过栅极接射频输入信号，第四MOS管通过源极接Vdd，第四MOS管通过栅极接偏置电压Vb1。
  3. 如权利要求1所述5G毫米波双频带双模混频器，其特征在于，所述第二MOS管和第三MOS管通过栅极分别连接本振信号的正负端，第二MOS管和第三MOS管通过漏极分别接入变压器的初级线圈的两个端口。
  4. 如权利要求3所述5G毫米波双频带双模混频器，其特征在于，所述变压器通过初级线圈的中心抽头串联第一电感后接入Vdd，次级线圈的中心抽头接地，次级线圈两侧的端口分别连接第五MOS管和第六MOS管的源极。
  5. 如权利要求4所述5G毫米波双频带双模混频器，其特征在于，所述第六MOS管通过栅极接偏置电压Vb2，第六MOS管通过漏极接第八MOS管的漏极。
  6. 如权利要求4所述5G毫米波双频带双模混频器，其特征在于，所述第五MOS管通过栅极接偏置电压Vb2，第五MOS管通过漏极接第七MOS管的漏极、第七MOS管的栅极和第八MOS管的栅极；
    所述第七MOS管和第八MOS管通过源极接Vdd。
  7. 如权利要求4所述5G毫米波双频带双模混频器，其特征在于，所述第 六MOS管通过漏极与第四电容一端连接，第四电容另一端串联第二电感接入单刀双掷开关的第一个输入端。
  8. 如权利要求3所述5G毫米波双频带双模混频器，其特征在于，所述变压器通过初级线圈的中心抽头与第三电容一端连接，第三电容另一端连接单刀双掷开关的第二个输入端，单刀双掷开关的输出端接入中频输出端口。
  9. 一种5G毫米波通信的接收机芯片，其特征在于，所述5G毫米波通信的接收机芯片包含权利要求1～8任意一项所述的5G毫米波双频带双模混频器，所述5G毫米波通信的接收机芯片同时工作覆盖27和39GHz附近的两个毫米波频带。
  10. 一种无线通信终端，其特征在于，所述无线通信终端安装有权利要求1～8任意一项所述的5G毫米波双频带双模混频器。
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Families Citing this family (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114826162B (zh) * 2022-05-18 2023-11-10 成都通量科技有限公司 一种5g毫米波双频带双模混频器及无线通信终端
CN115842521B (zh) * 2022-10-31 2024-02-20 电子科技大学 一种差分谐波混频器电路、芯片及接收机
CN116015332B (zh) * 2023-01-09 2024-06-25 成都通量科技有限公司 一种毫米波双频带镜像抑制接收机及接收方法
CN117394858B (zh) * 2023-12-08 2024-03-19 成都通量科技有限公司 一种降低回踢噪声的比较器、模数转换器及装置

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103187928A (zh) * 2011-12-29 2013-07-03 国民技术股份有限公司 一种能够改善低频闪烁噪声、高增益特性的方法及有源混频器
CN112368986A (zh) * 2018-09-19 2021-02-12 英特尔公司 具有高效能频谱滤波的超高数据速率数字毫米波发射机
CN113965167A (zh) * 2021-09-13 2022-01-21 天津大学 一种适用于5g通信系统的超宽带镜像抑制混频器
CN114826162A (zh) * 2022-05-18 2022-07-29 成都通量科技有限公司 一种5g毫米波双频带双模混频器及无线通信终端

Family Cites Families (16)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7509111B2 (en) * 2002-04-30 2009-03-24 Infineon Technologies Ag Integrated circuit having a mixer circuit
CN100555885C (zh) * 2002-12-11 2009-10-28 R.F.麦吉克公司 具有频带变换的集成交叉点切换器
JP2005269232A (ja) * 2004-03-18 2005-09-29 Renesas Technology Corp 周波数混合器
US7962114B2 (en) * 2007-01-12 2011-06-14 International Business Machines Corporation Drain-pumped sub-harmonic mixer for millimeter wave applications
EP2263308B1 (en) * 2008-03-25 2011-09-28 Telefonaktiebolaget L M Ericsson (PUBL) A combined mixer and balun design
US8224275B2 (en) * 2009-06-16 2012-07-17 Broadcom Corporation Area reduction techniques for saw-less receivers
CN102394566B (zh) * 2011-09-16 2014-07-09 复旦大学 一种带有自动最优偏置和谐波控制的吉尔伯特混频器
CN102457231B (zh) * 2011-12-16 2015-01-07 苏州云芯微电子科技有限公司 单平衡混频器
CN102832885B (zh) * 2012-09-07 2015-01-21 电子科技大学 一种低噪声可变增益混频器
US9237055B2 (en) * 2014-04-16 2016-01-12 University Of Macau ZigBee receiver exploiting an RF-to-BB current-reuse blixer and hybrid filter topology
CN106849876B (zh) * 2017-02-09 2019-10-25 中国科学技术大学 一种低功耗宽带射频混频器
CN107645300B (zh) * 2017-10-16 2020-04-10 中国科学院上海高等研究院 一种电流复用低功耗射频接收机
CN110336538B (zh) * 2019-04-16 2021-02-02 电子科技大学 一种cmos电容中和有源混频器
US11139779B2 (en) * 2019-10-21 2021-10-05 Samsung Electronics Co., Ltd. Dual band mixer
CN110995163B (zh) * 2019-11-26 2023-05-12 杭州电子科技大学 一种基波混频和谐波混频杂交型毫米波双频带无源混频器
US11271597B1 (en) * 2020-09-15 2022-03-08 Swiftlink Technologies Co., Ltd. Wideband transmitter for millimeter-wave wireless communication

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN103187928A (zh) * 2011-12-29 2013-07-03 国民技术股份有限公司 一种能够改善低频闪烁噪声、高增益特性的方法及有源混频器
CN112368986A (zh) * 2018-09-19 2021-02-12 英特尔公司 具有高效能频谱滤波的超高数据速率数字毫米波发射机
CN113965167A (zh) * 2021-09-13 2022-01-21 天津大学 一种适用于5g通信系统的超宽带镜像抑制混频器
CN114826162A (zh) * 2022-05-18 2022-07-29 成都通量科技有限公司 一种5g毫米波双频带双模混频器及无线通信终端

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