WO2023051520A1 - 开关电源的控制方法和开关电源 - Google Patents

开关电源的控制方法和开关电源 Download PDF

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Publication number
WO2023051520A1
WO2023051520A1 PCT/CN2022/121692 CN2022121692W WO2023051520A1 WO 2023051520 A1 WO2023051520 A1 WO 2023051520A1 CN 2022121692 W CN2022121692 W CN 2022121692W WO 2023051520 A1 WO2023051520 A1 WO 2023051520A1
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Prior art keywords
power supply
switching
switching power
voltage
tube
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PCT/CN2022/121692
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English (en)
French (fr)
Inventor
顾蝶芬
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力源科技有限公司
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Publication of WO2023051520A1 publication Critical patent/WO2023051520A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the embodiments of the present application relate to the technical field of power supplies, for example, to a method for controlling a switching power supply and the switching power supply.
  • non-isolated switching power supply In the field of power supply, non-isolated switching power supply has many advantages such as simple circuit, low cost and low space requirement.
  • the switching loss of the non-isolated switching power supply of the basic structure in the related art is too large, the efficiency is low, and the degree of electromagnetic interference (Electromagnetic Interference, EMI) is relatively high, which seriously limits the increase of the switching frequency of the power supply and hinders the miniaturization of the switching power supply. change.
  • EMI Electromagnetic Interference
  • ZVS Zero Voltage Switch
  • the non-isolated switching power supply that can realize ZVS needs to add several key control links and the corresponding hardware circuit structure.
  • the circuit structure of the switching power supply based on the ZVS design idea tends to be complicated. This not only increases the hardware cost of the switching power supply, but also increases the difficulty of controlling the power supply.
  • Embodiments of the present application provide a control method for a switching power supply and a switching power supply, so as to reduce switching losses of the switching power supply and improve EMI characteristics of the switching power supply without increasing the hardware cost of the switching power supply.
  • an embodiment of the present application provides a control method for a switching power supply
  • the switching power supply includes a first switching tube, a second switching tube, a filter inductor, and a control module; wherein the first switching tube is connected to a voltage Between the input terminal and the filter inductor, the second switch tube is connected between the first switch tube and the ground terminal, and the filter inductor is connected to the first switch tube and the voltage output of the switching power supply Between terminals, the control module is configured to control the first switch tube and the second switch tube to be turned on or off;
  • control methods include:
  • control module obtains the voltage waveform of the connection point between the first switching tube and the filter inductor
  • control module adjusts the cut-off current of the second switching tube until the voltage waveform of the connection point just returns to high;
  • the control module controls the first switch to be turned on, so as to realize the zero-voltage turn-on of the first switch.
  • the embodiment of the present application also provides a switching power supply, including a first switching tube, a second switching tube, a filter inductor, and a control module;
  • the first switch tube is configured to be turned on or off according to the main driving signal generated by the control module
  • the second switch tube is set to be turned on or off according to the synchronous driving signal generated by the control module;
  • the filter inductor is set to smooth the output current of the switching power supply
  • the control module is configured to obtain the voltage waveform of the connection point between the first switching tube and the filter inductor after the switching power supply is started; it is also configured to obtain zero voltage when the voltage waveform of the connection point does not achieve zero voltage When switching, adjust the cut-off current of the second switch tube until the voltage waveform of the connection point just returns to high; it is also set to control the conduction of the first switch tube when the voltage waveform of the connection point just returns to high.
  • FIG. 1 is a schematic structural diagram of a switching power supply provided by an embodiment of the present application
  • FIG. 2 is a flowchart of a control method for a switching power supply provided in an embodiment of the present application
  • FIG. 3 is a flow chart of another switching power supply control method provided by an embodiment of the present application.
  • FIG. 4 is a flow chart of another switching power supply control method provided by an embodiment of the present application.
  • FIG. 5 is a flow chart of another switching power supply control method provided by the embodiment of the present application.
  • FIG. 6 is a schematic waveform diagram of a switching power supply in a heavy-duty ZVS mode provided by an embodiment of the present application
  • FIG. 7 is a schematic waveform diagram of a switching power supply in a light-load ZVS mode provided by an embodiment of the present application.
  • Fig. 8 is a schematic structural diagram of another switching power supply provided by the embodiment of the present application.
  • FIG. 9 is a schematic waveform diagram of another switching power supply in the heavy-duty ZVS mode provided by the embodiment of the present application.
  • FIG. 10 is a schematic waveform diagram of another switching power supply in light-load ZVS mode provided by the embodiment of the present application.
  • Fig. 11 is a schematic structural diagram of another switching power supply provided by the embodiment of the present application.
  • Fig. 1 is a schematic structural diagram of a switching power supply provided by an embodiment of the present application
  • Fig. 2 is a flowchart of a control method of a switching power supply provided by an embodiment of the present application.
  • the embodiment of the present application can be applied to the power supply scene of any device with a non-isolated synchronous switching power supply structure with a basic architecture such as buck, boost, and buck-boost.
  • the control method of the switching power supply described in the embodiment of the present application can be It is limited to be implemented by the switching power supply in the embodiment of the present application as the execution subject, and the execution subject may be implemented by software and/or hardware.
  • Non-isolated switching power supply includes non-isolated synchronous switching power supply. Although there is a description of non-isolated switching power supply in this application, the control method of switching power supply described in this application is applied to non-isolated synchronous switching power supply.
  • FIG. 1 exemplarily shows the structure of a non-isolated switching power supply with a step-down architecture, which is not intended to limit the embodiment of the present application.
  • the switching power supply includes a first switching tube M1 , a second switching tube M2 , a filter inductor L and a control module IC.
  • the filter inductor L can also be used as an energy storage inductor.
  • the first switch tube M1 is connected between the voltage input terminal and the filter inductor L
  • the second switch tube M2 is connected between the first switch tube M1 and the ground terminal
  • the filter inductor L is connected to the voltage of the first switch tube M1 and the switching power supply
  • the control module IC is configured to control the first switching tube M1 and the second switching tube M2 to be turned on or off.
  • the control method includes:
  • control module After the switching power supply is started, the control module obtains a voltage waveform of a connection point between the first switching tube and the filter inductor.
  • control module adjusts the cut-off current of the second switch until the voltage waveform at the connection point just returns to high.
  • the cut-off current of the second switch tube M2 refers to the cut-off current of the second switch tube M2.
  • the fact that the voltage waveform of the connection point SW does not realize zero voltage switching means that the switching power supply does not realize ZVS. At this time, the switching loss of the switching power supply is too large, the efficiency is low, and the EMI is serious.
  • the adjustment process of the control module IC to the cut-off current of the second switching tube M2 can be increased, or can be decreased, or can be increased first and then decreased, or can be decreased first and then increased, or can be either An adjustment process of repeated oscillations. It can be understood that, the adjustment process of the cut-off current of the second switching tube M2 can be changed according to the setting and parameter selection adaptability of the switching power supply, which is not limited in this embodiment of the present application.
  • the voltage waveform of the connection point SW is exactly returned to high means that the voltage waveform of the connection point SW just returns to the high level state of the input voltage Vin, at this time, the voltage between the source and the drain of the first switching tube M1 The voltage difference is zero.
  • control module controls the first switching transistor to be turned on, so as to realize the zero-voltage turn-on of the first switching transistor.
  • the embodiment of the present application sets the control module IC to precisely control the conduction of the first switching tube M1 at the moment when the voltage waveform of the connection point SW just returns to high, thereby realizing the zero-voltage turn-on of the first switching tube M1 .
  • the switching power supply provided in this embodiment has a step-down non-isolated switching power supply structure, the first switching tube M1 and the second switching tube M2 cannot be turned on at the same time. Therefore, when the control module IC controls the first switch M1 to be turned on, the second switch M2 is turned off.
  • the control module IC obtains the voltage difference between the source and the drain of the first switching tube M1, that is, the difference between the input voltage and the voltage at the connection point SW, and when the difference reaches a preset voltage reference value, the control module IC
  • the first switching tube M1 is controlled to be turned off.
  • the preset voltage reference value corresponds to the peak current of the first switching tube M1, that is, the peak current of the filter inductor L.
  • Embodiments of the present application provide a control method for a switching power supply and a switching power supply.
  • the control method includes: after the switching power supply is started, the control module IC obtains the voltage waveform of the connection point SW between the first switching tube M1 and the filter inductor L; When the voltage waveform at point SW does not achieve zero-voltage switching, the control module IC adjusts the cut-off current of the second switch tube M2 until the voltage waveform at the connection point SW just returns to high; when the voltage waveform at the connection point SW just returns to high, the control module IC The IC controls the first switch M1 to turn on, so as to realize the zero-voltage turn-on of the first switch M1.
  • the embodiment of the present application does not add additional control links and circuit structures, and thus does not increase the circuit cost and control complexity of the switching power supply.
  • the control method of the non-isolated synchronous switching power supply of the basic structure in the related art is difficult to achieve ZVS, the switching loss is large, the efficiency is low, and the degree of EMI is high.
  • the cut-off current of the second switch tube M2 is adjusted by setting the control module IC, and when the voltage waveform of the connection point SW is just high, the first switch tube M1 is controlled to be turned on, and finally realized
  • the ZVS of the switching power supply can not only reduce the switching loss of the switching power supply, but also improve the EMI characteristics of the switching power supply.
  • the embodiment of the present application does not need to add additional control links and corresponding hardware circuit structures, so the circuit structure is simple, easy to control, and low in cost.
  • the switching power supply further includes a voltage divider circuit E, which is connected to the filter inductor L and the output voltage detection terminal S0 of the control module IC.
  • FIG. 3 is a flow chart of another method for controlling a switching power supply provided by an embodiment of the present application. As shown in Figure 3, the control method of the switching power supply provided by this embodiment includes:
  • control module After the switching power supply is started, the control module obtains a voltage waveform of a connection point between the first switching tube and the filter inductor.
  • control module adjusts the cut-off current of the second switching tube until the voltage waveform at the connection point just returns to high.
  • control module controls the first switching transistor to be turned on, so as to realize the zero-voltage turn-on of the first switching transistor.
  • the zero-voltage turn-on state of the first switching tube M1 may correspond to the zero-voltage turn-on of the switching power supply. It can be understood that, in the process of adjusting the cut-off current of the second switching tube M2, when the control module IC detects that the peak value of the voltage waveform at the connection point SW is close to returning to high, the control module IC controls the first switching tube M1 to turn on, and gradually realizes Zero-voltage turn-on of switching power supplies.
  • the control module adjusts the peak value of the current of the first switching tube, thereby realizing the switching power supply The stability loop control of the output voltage.
  • the first preset value is a certain current value
  • the setting method of the first preset value may be an initial setting of the switching power supply, or may be a user's own setting.
  • the switching current of the first switching tube refers to the current between the source and the drain of the first switching tube.
  • this embodiment can realize the stable loop control of the output voltage Vout of the switching power supply when the switching power supply is in a heavy load state and the load at the output end of the switching power supply changes, that is, the switching Heavy duty ZVS mode of the power supply.
  • the technical solution of this embodiment reduces the switching loss of the switching power supply and effectively improves the EMI characteristics of the switching power supply without increasing additional circuit costs and control complexity.
  • FIG. 4 is a flow chart of another method for controlling a switching power supply provided by an embodiment of the present application. As shown in Figure 4, the control method of the switching power supply provided by this embodiment includes:
  • control module After the switching power supply is started, the control module obtains a voltage waveform of a connection point between the first switching tube and the filter inductor.
  • control module adjusts the cut-off current of the second switching tube until the voltage waveform at the connection point just returns to high.
  • control module controls the first switching transistor to be turned on, so as to realize the zero-voltage turn-on of the first switching transistor.
  • the control module adjusts the peak value of the current of the first switching tube, thereby realizing the switching power supply The stability loop control of the output voltage.
  • the control module controls the first switch
  • the switching current of the tube is kept at the first preset value, and the frequency of the driving signal of the first switching tube is changed to enter the light-load ZVS mode to adjust the output voltage of the switching power supply.
  • the driving signal of the first switch tube is set to control the first switch tube M1 to be turned on or off.
  • the driving signal may be any pulse width or frequency modulation signal.
  • the load at the output end of the switching power supply changes, and the switching current of the first switching tube M1 is less than or equal to the first preset value means that due to the change in the load at the output end, the switching power supply changes from the heavy load state to light load condition.
  • the control module IC keeps the switching current of the first switching tube M1 at the first preset value, and the output voltage of the switching power supply is affected by changing the frequency of the driving signal of the first switching tube M1. Vout is adjusted, that is, the efficiency optimization of the switching power supply is realized through frequency adjustment.
  • S350 can be implemented in the following ways:
  • the control module IC keeps the switching current of the first switching tube M1 based on an appropriate anti-jitter threshold value. Near the first preset value, the peak current of the first switching tube M1 is further controlled to be in a steady state. Afterwards, the control module IC adjusts the output voltage Vout of the switching power supply in a frequency-adjusted light-load ZVS mode by adaptively changing the time interval during which the first switching tube M1 is turned on.
  • this embodiment can pass The frequency adjustment method realizes the optimization of the efficiency of the switching power supply under the light-load ZVS mode.
  • the technical solution of this embodiment can realize the real-time conversion between the light-load ZVS mode and the heavy-load ZVS mode according to the load change of the output end of the switching power supply without increasing the additional circuit cost and control complexity, not only The switching loss of the switching power supply is effectively reduced, and the EMI characteristic of the switching power supply is also improved.
  • the switching power supply further includes a snubber circuit Q, the snubber circuit Q is connected in parallel to both ends of the second switch tube M2, the snubber circuit Q includes a first resistor R1 and a first capacitor C1 connected in series; the voltage divider circuit E includes mutual The second resistor R2 and the third resistor R3 are connected in series.
  • FIG. 5 is a flow chart of another method for controlling a switching power supply provided by an embodiment of the present application. As shown in Figure 5, the control method of the switching power supply provided by this embodiment includes:
  • control module After the switching power supply is started, the control module obtains a voltage waveform of a connection point between the first switching tube and the filter inductor.
  • control module adjusts the cut-off current of the second switching tube until the voltage waveform at the connection point just returns to high.
  • control module controls the first switching transistor to be turned on, so as to realize the zero-voltage turn-on of the first switching transistor.
  • the control module adjusts the peak value of the current of the first switching tube, thereby realizing the switching power supply The stability loop control of the output voltage.
  • the control module controls the first switch
  • the switching current of the tube is kept at the first preset value, and the frequency of the driving signal of the first switching tube is changed to enter the light-load ZVS mode to adjust the output voltage of the switching power supply.
  • the load at the output end of the switching power supply changes, and the time interval between the zero-crossing off and on of the second switching tube is equal to or less than the second preset value means that due to the change in the load at the output end, the second switch
  • the time interval between the zero-crossing turn-off and turn-on of the tube M2 changes from greater than the second preset value to equal to or less than the second preset value, and the switching power supply changes from the light-load state to the heavy-load state again.
  • the control module IC needs to repeat S420-S440 to realize heavy-load ZVS regulation.
  • this embodiment can realize the heavy-load ZVS of the switching power supply again by adjusting the peak current of the first switching tube M1.
  • the technical solution of this embodiment can realize light-load ZVS frequency adjustment mode and heavy-load ZVS peak current adjustment mode according to the output load change of switching power supply without increasing additional circuit cost and control complexity.
  • the real-time conversion not only reduces the switching loss of the switching power supply, but also improves the EMI characteristics of the switching power supply.
  • the switching power supply includes a first switching tube M1 , a second switching tube M2 , a filter inductor L and a control module IC.
  • the first switching tube M1 is configured to be turned on or off according to the main driving signal Ig1 generated by the control module IC.
  • the second switching tube M2 is set to be turned on or off according to the synchronous driving signal Ig2 generated by the control module IC.
  • the filter inductor L is set to smooth the output current of the switching power supply.
  • the control module IC is set to obtain the voltage waveform of the connection point SW between the first switching tube M1 and the filter inductor L after the switching power supply is started;
  • the cut-off current of the second switching tube M2 is until the voltage waveform of the connection point SW just returns to high; it is also set to control the first switching tube M1 to conduct when the voltage waveform of the connecting point SW just returns to high, so as to realize the first switching tube M1
  • the first switching tube M1 and the second switching tube M2 may be, but not limited to, metal-oxide-semiconductor field-effect transistors (Metal-Oxide-Semiconductor Field-Effect Transistor, MOSFET). It can be understood that the type selection and structural parameters of the first switching tube M1 and the second switching tube M2 are related to the intended power supply effect, which is not limited in this embodiment of the present application.
  • MOSFET Metal-Oxide-Semiconductor Field-Effect Transistor
  • control module IC can also be configured to control the first switching tube M1 to turn on when the peak value of the voltage waveform at the connection point SW is close to returning to high after the switching power supply is started.
  • the switching power supply further includes a voltage dividing circuit E, and the voltage dividing circuit E is configured to generate a voltage dividing signal so that the control module IC obtains the output voltage Vout of the switching power supply.
  • the voltage dividing circuit E includes a second resistor R2 and a third resistor R3 connected in series.
  • the switching power supply further includes a snubber circuit Q, and the snubber circuit Q includes a first resistor R1 and a first capacitor C1 connected in series.
  • the snubber circuit Q is set to optimize the system switching loss, reduce the voltage and/or current peak of the switching tube, and improve the electromagnetic interference characteristics of the switching power supply.
  • the first resistor R1, the second resistor R2 and the third resistor R3 can be any kind of resistors, and the types and parameters of the above resistors can be adaptively adjusted according to the power supply effect to be obtained by the switching power supply. There are no restrictions on this. Exemplarily, the above resistors can all be chip resistors.
  • the first capacitor C1 can be any kind of capacitor, and the type and parameters of the capacitor can be adaptively adjusted according to the intended power supply effect of the switching power supply, which is not limited in this embodiment of the present application.
  • the first capacitor C1 may be a mica capacitor.
  • FIG. 1 the connection relationship of circuit elements of the non-isolated switching power supply of the step-down architecture provided by the embodiment of the present application is shown in FIG. 1 , which will not be repeated here. It can be understood that Vin in FIG. 1 represents the input voltage of the switching power supply, and Iin represents the current flowing through the filter inductor.
  • the working process of the switching power supply is as follows:
  • the control module IC After the switching power supply is started, the control module IC obtains the voltage waveform of the connection point SW. When the voltage waveform at the connection point SW does not achieve zero-voltage switching, the control module IC adjusts the cut-off current of the second switching tube M2 until the voltage waveform at the connection point SW just returns to high. When the voltage waveform at the connection point SW is exactly high, the control module IC controls the first switch M1 to be turned on, so as to realize the zero-voltage turn-on of the first switch M1 .
  • the control module IC adjusts the peak value of the current of the first switching tube M1 according to the output voltage Vout of the switching power supply, thereby realizing The stable loop control of the output voltage Vout of the switching power supply.
  • the control module IC After realizing the stable loop control of the output voltage Vout of the switching power supply, when the load at the output end of the switching power supply changes and the switching current of the first switching tube M1 is less than or equal to the first preset value, the control module IC based on the appropriate The anti-jitter threshold keeps the switching current of the first switching tube M1 close to the first preset value, and changes the frequency of the driving signal of the first switching tube M1 to enter the light-load ZVS mode to adjust the output voltage Vout of the switching power supply.
  • control module IC After entering the light-load ZVS mode to adjust the output voltage Vout of the switching power supply, when the load at the output terminal of the switching power supply changes, and the time interval between the zero-crossing off and on of the second switching tube M2 is equal to or less than the second preset When setting the value, the control module IC enters the heavy-duty ZVS mode again to adjust the peak current of the first switching tube M1.
  • control module IC in the embodiment of the present application can integrate a typical voltage reference module inside, or can use an external voltage reference module to realize the output voltage Vout of the switching power supply and the module reference voltage provided by the voltage reference module The comparison, and then realize the controllable adjustment of the output voltage Vout of the switching power supply in different states.
  • FIG. 6 is a schematic waveform diagram of a switching power supply in a heavy-duty ZVS mode provided by an embodiment of the present application.
  • the switching power supply and FIG. 6 for example, when the waveform of the voltage V SW at the connection point SW just returns to high, the first switch M1 is turned on, and the second switch M2 is turned off.
  • the switching current IS of the second switching tube M2 reaches the cut-off current IS_min , the second switching tube M2 is turned off.
  • FIG. 7 is a schematic waveform diagram of a switching power supply in a light-load ZVS mode provided by an embodiment of the present application.
  • the control module IC drives the first switching tube M1 according to the voltage signal fed back by the voltage divider circuit E, or the frequency of the first switching tube M1 driving signal The time interval for turning on or turning off M1 is adjusted.
  • the control module IC turns on the second switch tube M2 and adjusts the second switch tube.
  • the cut-off current I S_min of M2 makes the voltage waveform of the connection point SW just return to high; when the voltage waveform of SW just returns to high, the control module IC turns on the first switching tube M1, and when the switching current of the first switching tube M1 reaches the first preset value, or when the switching current of the first switching tube M1 reaches near the first preset value based on a certain anti-shake threshold, the first switching tube M1 is turned off, and the second switching tube M2 is turned on; when the second switching tube M2 When the switching current IS is at the current zero crossing point, the second switching tube M2 is turned off. So far, this embodiment completes one cycle of the light-load ZVS mode, and waits for the arrival of the next frequency or switching time interval.
  • the embodiment of the present application can realize the ZVS of the switching power supply under the heavy load state.
  • this embodiment can adjust the frequency In this way, the efficiency optimization of the switching power supply under the light-load ZVS state is realized.
  • this embodiment can enter the heavy-load ZVS again.
  • the technical solution of this embodiment can realize light-load ZVS frequency adjustment mode and heavy-load ZVS peak current adjustment mode according to the output load change of switching power supply without increasing additional circuit cost and control complexity.
  • the real-time conversion not only effectively reduces the switching loss of the switching power supply, but also helps to improve the EMI characteristics of the switching power supply.
  • the embodiment of the present application may, but not limited to, adjust the output voltage Vout of the switching power supply by adopting the light-load ZVS mode.
  • the control module IC Control the switching current of the first switching tube M1 to keep at the first preset value, and adopt the commonly used non-zero voltage turn-on method, and change the drive of the first switching tube M1 when the voltage waveform at the connection point SW has no return to high process
  • the signal frequency is used to adjust the output voltage Vout of the switching power supply.
  • the control module IC turns on the first switching tube M1; when the switching current of the first switching tube M1 When the first preset value is reached, or the switching current of the first switch tube M1 reaches near the first preset value based on a certain preset anti-shake threshold, the first switch tube M1 is turned off, and the second switch tube M2 is turned on; When the switching current IS of the second switching tube M2 is at the current zero crossing point, the second switching tube M2 is turned off. So far, this embodiment implements one cycle of the light-load non-ZVS mode, and waits for the next time that meets the frequency or time interval requirements.
  • FIG. 8 is a schematic structural diagram of another switching power supply provided by an embodiment of the present application.
  • the switching power supply shown in FIG. 8 is a non-isolated synchronous switching power supply with a boost architecture.
  • the circuit element connection relationship of the switching power supply provided by this embodiment is shown in FIG. 8 , which will not be repeated here.
  • Vin' in Figure 8 represents the input voltage of the non-isolated synchronous switching power supply of the boost architecture
  • L' represents the input inductance of the non-isolated synchronous switching power supply of the boost architecture
  • Iin' represents the non-isolated synchronous switching power supply of the boost architecture.
  • the inductor current of the synchronous switching power supply, Q' represents the sink circuit of the non-isolated synchronous switching power supply of the boost architecture.
  • the working process of the non-isolated synchronous switching power supply of the boost architecture is as follows:
  • the control module IC' After the switching power supply is started, the control module IC' obtains the voltage waveform of the connection point SX. When the voltage waveform at the connection point SX does not achieve zero-voltage switching, the control module IC' adjusts the cut-off current of the fourth switching tube M4 until the voltage waveform at the connection point SX is close to low or just low. When the voltage waveform at the connection point SX is just low, the control module IC' controls the third switching tube M3 to turn on.
  • the control module IC' adjusts the peak current of the third switching tube M3 according to the output voltage Vout' of the switching power supply, Furthermore, the stable loop control of the output voltage Vout' of the switching power supply is realized.
  • the control module IC' After realizing the stable loop control of the output voltage Vout' of the switching power supply, when the load at the output end of the switching power supply changes and the switching current of the third switching tube M3 is less than or equal to the third preset value, the control module IC' based on An appropriate anti-vibration threshold keeps the switching current of the third switching tube M3 near the third preset value, and changes the frequency of the driving signal of the third switching tube M3 to enter the light-load ZVS mode to adjust the output voltage of the switching power supply Vout'.
  • the control module IC' After entering the light-load ZVS mode to adjust the output voltage Vout' of the switching power supply, when the load at the output terminal of the switching power supply changes, and the time interval between the zero-crossing turn-off and turn-on of the fourth switching tube M4 is equal to or less than the fourth When the preset value is reached, the control module IC' enters the heavy-load ZVS mode again to adjust the peak current of the third switching tube M3.
  • the voltage waveform at the connection point SX just returns to low means that the voltage waveform at the connection point SX just returns to a low-level state of zero voltage. At this time, the voltage between the source and the drain of the third switching tube M3 The difference is zero.
  • FIG. 9 is a schematic waveform diagram of another switching power supply in the heavy-duty ZVS mode provided by the embodiment of the present application.
  • the third switching tube M3 is turned on, and the fourth switching tube M4 is turned on. is off.
  • the switching current I t of the fourth switching tube M4 reaches the cut-off current IT_min , the fourth switching tube M4 is turned off.
  • FIG. 10 is a schematic waveform diagram of another switching power supply in light-load ZVS mode provided by an embodiment of the present application.
  • the control module IC' drives the third switching tube M3 according to the voltage signal fed back by the voltage divider circuit E'
  • the frequency of the signal, or the time interval of turning on or off the third switch tube M3 is adjusted, and when the frequency of the driving signal of M3, or the time interval of turning on or off of M3 meets the requirements, the control module IC' turns on the fourth switch tube M4 , and adjust the cut-off current I T_min of the fourth switching tube M4 until the voltage waveform of the connection point SX just goes low; when the voltage waveform of SX just goes low, the control module IC' turns on the third switching tube M3, when the third switch When the switching current of the tube M3 reaches the third preset value, or when the switching
  • FIG. 11 is a schematic structural diagram of another switching power supply provided by the embodiment of the present application.
  • the switching power supply shown in FIG. 11 is a non-isolated synchronous switching power supply with a buck-boost architecture.
  • the connection relationship of circuit components of the switching power supply provided in this embodiment is shown in FIG. 11 , which will not be repeated here.
  • Vin indicates the input voltage of the non-isolated synchronous switching power supply of the buck-boost architecture
  • IC indicates the control module of the non-isolated synchronous switching power supply of the buck-boost architecture
  • Iin indicates the non-isolated synchronous switch of the buck-boost architecture
  • the inductor current of the power supply, Vout represents the output voltage of the non-isolated synchronous switching power supply of the buck-boost architecture.
  • the non-isolated synchronous switching power supply of the buck-boost architecture can be formed by docking part of the circuit structure of the non-isolated synchronous switching power supply of the boost and buck architecture, and the non-isolated synchronous switching power supply of the buck-boost architecture includes a boost mode, buck mode, and buck-boost mode.
  • the node waveform diagrams corresponding to the boost mode and the buck mode can be composed of multiple waveform diagrams of the non-isolated synchronous switching power supply of the aforementioned boost and buck architectures, and appear in the boost architecture or the buck mode respectively.
  • the buck-boost mode can be a composite mode of the boost mode and the buck mode.
  • the switching power supply is in the buck-boost mode, the waveforms on the nodes of the boost architecture and the buck architecture can appear simultaneously or alternately. This embodiment of the present application does not limit this.
  • the composite mode of the boost mode and the buck mode mentioned here means that the non-isolated synchronous switching power supply part of the boost architecture and the non-isolated synchronous switching power supply part of the step-down architecture in the buck-boost architecture non-isolated synchronous switching power supply are simultaneously or Alternately in the working state;
  • boost mode means that the non-isolated synchronous switching power supply part of the boost architecture in the buck-boost architecture non-isolated synchronous switching power supply is in the working state, while the non-isolated synchronous switching power supply part of the buck architecture is in the non-alternating switching state.
  • step-down mode means that the non-isolated synchronous switching power supply part of the boost architecture in the buck-boost architecture non-isolated synchronous switching power supply is in the state of non-alternating switching, while the non-isolated synchronous switching power supply part of the step-down architecture is in the working state.
  • the non-isolated synchronous switching power supply of the buck-boost architecture when the non-isolated synchronous switching power supply of the buck-boost architecture is in the boost mode, the first switching tube M1 is in a real-time on state, and the second switching tube M2 is in a real-time off state. Conversely, when the non-isolated synchronous switching power supply of the buck-boost architecture is in the buck mode, the fourth switching tube M4 is in a real-time on state, and the third switching tube M3 is in a real-time off state.

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Abstract

本申请实施例公开了一种开关电源的控制方法和开关电源。控制方法包括：开关电源启动后，控制模块获取第一开关管与滤波电感之间的连接点的电压波形；当连接点的电压波形未实现零电压切换时，控制模块调节第二开关管的截止电流，直至连接点的电压波形恰好归高；当连接点的电压波形恰好归高时，控制模块控制第一开关管导通，以实现第一开关管的零电压开启。

Description

开关电源的控制方法和开关电源
本申请要求在2021年9月28日提交中国专利局、申请号为202111143287.X的中国专利申请的优先权，该申请的全部内容通过引用结合在本申请中。
技术领域
本申请实施例涉及电源技术领域，例如涉及一种开关电源的控制方法和开关电源。
背景技术
在电源领域中，非隔离开关电源兼具电路简单、成本低廉和空间要求低等诸多优势，基于降压、升压以及升降压等基本架构的非隔离开关电源在手机、平板电脑和家用电器等电器设备的供电过程中应用广泛。
目前，相关技术中的基本架构的非隔离开关电源开关损耗过大，效率较低，电磁干扰(Electromagnetic Interference，EMI)程度偏高，因而严重限制了电源开关频率的提高，阻碍了开关电源的小型化。基于此，能够实现零电压开关(Zero Voltage Switch，ZVS)的非隔离开关电源应运而生。然而，与基本架构的非隔离开关电源相比，可实现ZVS的非隔离开关电源需要额外增加数个关键控制环节以及相应的硬件电路结构，基于ZVS设计思路的开关电源的电路结构趋于复杂，不仅提高了开关电源的硬件成本，还增加了电源的控制难度。
发明内容
本申请实施例提供一种开关电源的控制方法和开关电源，以在不增加开关电源硬件成本的前提下，降低开关电源的开关损耗，改善开关电源的EMI特 性。
第一方面，本申请实施例提供了一种开关电源的控制方法，所述开关电源包括第一开关管、第二开关管、滤波电感以及控制模块；其中，所述第一开关管连接于电压输入端和所述滤波电感之间，所述第二开关管连接于所述第一开关管和接地端之间，所述滤波电感连接于所述第一开关管和所述开关电源的电压输出端之间，所述控制模块设置为控制所述第一开关管和所述第二开关管导通或者关断；
所述控制方法包括：
所述开关电源启动后，所述控制模块获取所述第一开关管与所述滤波电感之间的连接点的电压波形；
当所述连接点的电压波形未实现零电压切换时，所述控制模块调节所述第二开关管的截止电流，直至所述连接点的电压波形恰好归高；
当所述连接点的电压波形恰好归高时，所述控制模块控制所述第一开关管导通，以实现所述第一开关管的零电压开启。
第二方面，本申请实施例还提供了一种开关电源，包括第一开关管、第二开关管、滤波电感以及控制模块；
所述第一开关管设置为根据所述控制模块产生的所述主驱动信号导通或者关断；
所述第二开关管设置为根据所述控制模块产生的同步驱动信号导通或者关断；
所述滤波电感设置为平滑所述开关电源的输出电流；
所述控制模块设置为在所述开关电源启动后，获取所述第一开关管与所述滤波电感之间的连接点的电压波形；还设置为当所述连接点的电压波形未实现零电压切换时，调节所述第二开关管的截止电流，直至所述连接点的电压波形恰好归高；还设置为当所述连接点的电压波形恰好归高时，控制所述第一开关管导通，以实现所述第一开关管的零电压开启；还设置为当所述开关电源的输 出端负载发生改变，且所述第一开关管的开关电流大于第一预设值时，根据所述开关电源的输出电压，调节所述第一开关管的电流峰值，进而实现所述开关电源的输出电压的稳定环路控制；还设置为在实现所述开关电源的输出电压的稳定环路控制之后，当所述开关电源的输出端负载发生改变，且所述第一开关管的开关电流小于或等于所述第一预设值时，控制所述第一开关管的开关电流保持在所述第一预设值，并改变所述第一开关管的驱动信号的频率，进入轻载ZVS模式以调节所述开关电源的输出电压；还设置为在进入轻载ZVS模式以调节开关电源的输出电压之后，当所述开关电源的输出端负载发生改变，且所述第二开关管的过零关断和开启之间的时间间隔等于或小于第二预设值时，再次进入重载ZVS模式，调节所述第一开关管的峰值电流。
附图说明
图1是本申请实施例提供的一种开关电源的结构示意图；
图2是本申请实施例提供的一种开关电源的控制方法的流程图；
图3是本申请实施例提供的另一种开关电源的控制方法的流程图；
图4是本申请实施例提供的又一种开关电源的控制方法的流程图；
图5是本申请实施例提供的又一种开关电源的控制方法的流程图；
图6是本申请实施例提供的一种重载ZVS模式下开关电源的波形示意图；
图7是本申请实施例提供的一种轻载ZVS模式下开关电源的波形示意图；
图8是本申请实施例提供的另一种开关电源的结构示意图；
图9是本申请实施例提供的另一种重载ZVS模式下开关电源的波形示意图；
图10是本申请实施例提供的另一种轻载ZVS模式下开关电源的波形示意图；
图11是本申请实施例提供的又一种开关电源的结构示意图。
具体实施方式
图1是本申请实施例提供的一种开关电源的结构示意图，图2是本申请实施例提供的一种开关电源的控制方法的流程图。本申请实施例可适用于具备降压、升压以及升降压等基本架构的非隔离同步开关电源结构的任意设备的电源供电场景，本申请实施例所述的开关电源的控制方法可以但不限于由本申请实施例中的开关电源作为执行主体来执行，该执行主体可以采用软件和/或硬件的方式实现。非隔离开关电源包括非隔离同步开关电源，虽然在本申请中有非隔离开关电源的表述，但是本申请所述的开关电源的控制方法应用于非隔离同步开关电源。
图1示例性示出了降压架构的非隔离开关电源的结构，不作为对本申请实施例的限定。参见图1，开关电源包括第一开关管M1、第二开关管M2、滤波电感L以及控制模块IC。其中，滤波电感L也可以作为储能电感。第一开关管M1连接于电压输入端和滤波电感L之间，第二开关管M2连接于第一开关管M1和接地端之间，滤波电感L连接于第一开关管M1和开关电源的电压输出端之间，控制模块IC设置为控制第一开关管M1和第二开关管M2导通或者关断。如图2所示，该控制方法包括：
S110、开关电源启动后，控制模块获取第一开关管与滤波电感之间的连接点的电压波形。
S120、当连接点的电压波形未实现零电压切换时，控制模块调节第二开关管的截止电流，直至连接点的电压波形恰好归高。
其中，第二开关管M2的截止电流是指第二开关管M2的关断电流。示例性地，连接点SW的电压波形未实现零电压切换意味着开关电源未实现ZVS。此时，开关电源的开关损耗偏大，效率较低且EMI严重。
控制模块IC对第二开关管M2截止电流的调节过程可以是增大，或者可 以是减小，或者可以是先增大后减小，或者可以是先减小后增大，或者可以是任一种反复震荡的调节过程。可以理解的是，上述第二开关管M2截止电流的调节过程可以根据开关电源的设置和参数选择适应性进行改变，本申请实施例对此不进行限制。
可以理解的是，连接点SW的电压波形恰好归高是指，连接点SW的电压波形恰好恢复到输入电压Vin的高电平状态，此时，第一开关管M1源极与漏极间的电压差为零。
S130、当连接点的电压波形恰好归高时，控制模块控制第一开关管导通，以实现第一开关管的零电压开启。
其中，当连接点SW的电压波形恰好归高意味着开关电源能够在该时刻刚好实现ZVS。基于此，本申请实施例通过设置控制模块IC在连接点SW的电压波形恰好归高的时刻，恰好控制第一开关管M1导通，进而实现了第一开关管M1的零电压开启。
可以理解的是，由于本实施例提供的开关电源具备降压架构的非隔离开关电源结构，因而第一开关管M1和第二开关管M2不能同时导通。因此，当控制模块IC控制第一开关管M1导通时，第二开关管M2关断。
示例性地，继续参见图1，可以理解的是，第一开关管M1的关断过程如下：
控制模块IC获取第一开关管M1的源极与漏极之间的电压差值，也即输入电压与连接点SW电压的差值，当该差值达到预设电压基准值时，控制模块IC控制第一开关管M1关断。可以理解的是，预设电压基准值对应于第一开关管M1的峰值电流，也即滤波电感L的峰值电流。
本申请实施例提供一种开关电源的控制方法和开关电源，控制方法包括：开关电源启动后，控制模块IC获取第一开关管M1与滤波电感L之间的连接点SW的电压波形；当连接点SW的电压波形未实现零电压切换时，控制模块IC调节第二开关管M2的截止电流，直至连接点SW的电压波形恰好归高；当 连接点SW的电压波形恰好归高时，控制模块IC控制第一开关管M1导通，以实现第一开关管M1的零电压开启。
与相关技术中基本架构的非隔离开关电源相比，本申请实施例未增加额外的控制环节及电路结构，因而未增加开关电源的电路成本和控制复杂程度。此外，相关技术中基本架构的非隔离同步开关电源的控制方法难以实现ZVS，开关损耗大，效率低，EMI程度高，而本申请实施例在第一开关管M1与滤波电感L之间连接点SW的电压波形未实现零电压切换时，通过设置控制模块IC调节第二开关管M2的截止电流，并在连接点SW的电压波形恰好归高时，控制第一开关管M1导通，最终实现了开关电源的ZVS，不仅能够降低开关电源的开关损耗，还能改善开关电源的EMI特性。
与相关技术中基于ZVS设计思路的非隔离开关电源相比，本申请实施例无需增加额外的控制环节及相应的硬件电路结构，因而电路结构简单，易于控制，成本低廉。
在上述实施例的基础上，以下对重载情况下开关电源输出电压Vout的调节方法进行说明，但不作为对本申请的限定。继续参见图1，开关电源还包括分压电路E，分压电路E连接于滤波电感L和控制模块IC的输出电压检测端S0。图3是本申请实施例提供的另一种开关电源的控制方法的流程图。如图3所示，本实施例提供的开关电源的控制方法包括：
S210、开关电源启动后，控制模块获取第一开关管与滤波电感之间的连接点的电压波形。
S220、当连接点的电压波形未实现零电压切换时，控制模块调节第二开关管的截止电流，直至连接点的电压波形恰好归高。
S230、当连接点的电压波形恰好归高时，控制模块控制第一开关管导通，以实现第一开关管的零电压开启。
其中，第一开关管M1的零电压开启状态可以对应于开关电源的零电压开启。可以理解的是，在调节第二开关管M2的截止电流过程中，当控制模块IC 检测到连接点SW电压波形的峰值接近归高时，控制模块IC控制第一开关管M1导通，逐步实现开关电源的零电压开启。
S240、当开关电源的输出端负载发生改变，且第一开关管的开关电流大于第一预设值时，根据开关电源的输出电压，控制模块调节第一开关管的电流峰值，进而实现开关电源的输出电压的稳定环路控制。
其中，第一预设值是某一电流值，第一预设值的设定方式可以是开关电源的初始设定，或者可以是用户自主设定。此处第一开关管的开关电流是指第一开关管源极与漏极之间的电流。
在上述实施例实现开关电源ZVS的基础上，本实施例能够在开关电源处于重载状态，且开关电源的输出端负载发生改变时，实现开关电源的输出电压Vout的稳定环路控制，即开关电源的重载ZVS模式。与相关技术相比，本实施例的技术方案在不增加额外电路成本和控制复杂程度的前提下，降低了开关电源的开关损耗，有效改善了开关电源的EMI特性。
在上述实施例的基础上，以下对重载转换为轻载情况下开关电源的控制方法进行说明，但不作为对本申请的限定。图4是本申请实施例提供的又一种开关电源的控制方法的流程图。如图4所示，本实施例提供的开关电源的控制方法包括：
S310、开关电源启动后，控制模块获取第一开关管与滤波电感之间的连接点的电压波形。
S320、当连接点的电压波形未实现零电压切换时，控制模块调节第二开关管的截止电流，直至连接点的电压波形恰好归高。
S330、当连接点的电压波形恰好归高时，控制模块控制第一开关管导通，以实现第一开关管的零电压开启。
S340、当开关电源的输出端负载发生改变，且第一开关管的开关电流大于第一预设值时，根据开关电源的输出电压，控制模块调节第一开关管的电流峰值，进而实现开关电源的输出电压的稳定环路控制。
S350、在实现开关电源的输出电压的稳定环路控制之后，当开关电源的输出端负载发生改变，且第一开关管的开关电流小于或等于第一预设值时，控制模块控制第一开关管的开关电流保持在第一预设值，并改变第一开关管的驱动信号的频率，进入轻载ZVS模式以调节开关电源的输出电压。
其中，第一开关管的驱动信号设置为控制第一开关管M1导通或关断。示例性地，该驱动信号可以是任一种脉冲宽度或频率调制信号。
可以理解的是，开关电源的输出端负载发生改变，且第一开关管M1的开关电流小于或等于第一预设值是指，由于输出端负载发生改变，使得开关电源由重载状态转换为轻载状态。
基于此，当开关电源转换为轻载状态时，控制模块IC使得第一开关管M1的开关电流保持在第一预设值，通过改变第一开关管M1的驱动信号频率对开关电源的输出电压Vout进行调节，即通过频率调节方式实现了开关电源的效率最优化。
示例性地，S350可以通过以下方式得以实现：
当开关电源的输出端负载发生改变，且第一开关管M1的开关电流小于或等于第一预设值时，控制模块IC基于适当的防抖动阀值使得第一开关管M1的开关电流保持在第一预设值附近，进而控制第一开关管M1的峰值电流处于稳态。之后，控制模块IC通过适应性改变第一开关管M1开启的时间间隔，以频率调节的轻载ZVS模式调整开关电源的输出电压Vout。
综上，在上述实施例实现重载状态下的开关电源ZVS，且开关电源的输出电压Vout保持稳态的基础上，当开关电源由重载状态转换为轻载状态时，本实施例能够通过频率调节方式实现轻载ZVS模式下开关电源的效率最优化。与相关技术相比，本实施例的技术方案在不增加额外电路成本和控制复杂程度的前提下，能够根据开关电源的输出端负载变化实现轻载ZVS模式与重载ZVS模式的实时转换，不仅有效降低了开关电源的开关损耗，还有利于改善了开关电源的EMI特性。
在上述实施例的基础上，以下对由重载变换至轻载状态后，再次转换为重载情况下的开关电源的控制方法进行说明，但不作为对本申请的限定。继续参见图1，开关电源还包括吸收电路Q，吸收电路Q并联于第二开关管M2的两端，吸收电路Q包括相互串联的第一电阻R1和第一电容C1；分压电路E包括相互串联的第二电阻R2和第三电阻R3。图5是本申请实施例提供的又一种开关电源的控制方法的流程图。如图5所示，本实施例提供的开关电源的控制方法包括：
S410、开关电源启动后，控制模块获取第一开关管与滤波电感之间的连接点的电压波形。
S420、当连接点的电压波形未实现零电压切换时，控制模块调节第二开关管的截止电流，直至连接点的电压波形恰好归高。
S430、当连接点的电压波形恰好归高时，控制模块控制第一开关管导通，以实现第一开关管的零电压开启。
S440、当开关电源的输出端负载发生改变，且第一开关管的开关电流大于第一预设值时，根据开关电源的输出电压，控制模块调节第一开关管的电流峰值，进而实现开关电源的输出电压的稳定环路控制。
S450、在实现开关电源的输出电压的稳定环路控制之后，当开关电源的输出端负载发生改变，且第一开关管的开关电流小于或等于第一预设值时，控制模块控制第一开关管的开关电流保持在第一预设值，并改变第一开关管的驱动信号的频率，进入轻载ZVS模式以调节开关电源的输出电压。
S460、在调节开关电源的输出电压之后，当开关电源的输出端负载发生改变，且第二开关管的过零关断和开启之间的时间间隔等于或小于第二预设值时，控制模块再次进入重载ZVS模式，调节第一开关管的峰值电流。
其中，开关电源的输出端负载发生改变，且第二开关管的过零关断和开启之间的时间间隔等于或小于第二预设值是指，由于输出端负载发生改变，使得第二开关管M2的过零关断和开启之间的时间间隔由大于第二预设值变换为等 于或小于第二预设值，开关电源由轻载状态再次转换为重载状态。可以理解的是，此时，控制模块IC需要重复S420～S440，以实现重载ZVS调节。
综上所述，在上述实施例实现重载状态下的开关电源ZVS，且开关电源的输出电压Vout保持稳态的基础上，当开关电源由重载状态转换为轻载状态时，本实施例能够通过频率调节方式实现轻载ZVS状态下开关电源的效率最优化。此外，当开关电源由轻载状态再次转换为重载状态时，本实施例能够通过调节第一开关管M1的峰值电流再次实现开关电源的重载ZVS。与相关技术相比，本实施例的技术方案在不增加额外电路成本和控制复杂程度的前提下，能够根据开关电源的输出端负载变化实现轻载ZVS频率调节方式与重载ZVS峰值电流调节方式的实时转换，不仅降低了开关电源的开关损耗，还改善了开关电源的EMI特性。
继续参见图1，开关电源包括第一开关管M1、第二开关管M2、滤波电感L以及控制模块IC。第一开关管M1设置为根据控制模块IC产生的主驱动信号Ig1导通或者关断。第二开关管M2设置为根据控制模块IC产生的同步驱动信号Ig2导通或者关断。滤波电感L设置为平滑开关电源的输出电流。
控制模块IC设置为在开关电源启动后，获取第一开关管M1与滤波电感L之间的连接点SW的电压波形；还设置为当连接点SW的电压波形未实现零电压切换时，调节第二开关管M2的截止电流，直至连接点SW的电压波形恰好归高；还设置为当连接点SW的电压波形恰好归高时，控制第一开关管M1导通，以实现第一开关管M1的零电压开启；还设置为当开关电源的输出端负载发生改变，且第一开关管M1的开关电流大于第一预设值时，根据开关电源的输出电压Vout，调节第一开关管M1的电流峰值，进而实现开关电源的输出电压Vout的稳定环路控制；还设置为在实现开关电源的输出电压Vout的稳定环路控制之后，当开关电源的输出端负载发生改变，且第一开关管M1的开关电流小于或等于第一预设值时，控制第一开关管M1的开关电流保持在第一预设值，并改变第一开关管M1的驱动信号的频率，进入轻载ZVS模式以调节开关 电源的输出电压Vout；还设置为在进入轻载ZVS模式以调节开关电源的输出电压之后，当开关电源的输出端负载发生改变，且第二开关管M2的过零关断和开启之间的时间间隔等于或小于第二预设值时，再次进入重载ZVS模式，调节第一开关管M1的峰值电流。
其中，示例性地，第一开关管M1和第二开关管M2可以但不限于是金属氧化物半导体场效应管(Metal-Oxide-Semiconductor Field-Effect Transistor，MOSFET)。可以理解的是，第一开关管M1和第二开关管M2的类别选择和结构参数与拟取得的供电效果相关，本申请实施例对此不进行限制。
示例性地，控制模块IC还可以设置为在开关电源启动后，当连接点SW的电压波形峰值接近归高时，控制第一开关管M1导通。
可以理解的是，在进入轻载ZVS模式以调节开关电源的输出电压之后，当开关电源的输出端负载发生改变，且第二开关管的过零关断和开启之间的时间间隔等于或小于第二预设值时，再次进入重载ZVS模式，调节第一开关管的峰值电流是指，在调节开关电源的输出电压Vout之后，控制模块IC还设置为当开关电源的输出端负载发生改变，且第二开关管M2的过零关断和开启之间的时间间隔由大于第二预设值变换为等于或小于第二预设值时，再次进入重载ZVS模式，调节第一开关管M1的峰值电流。
可选地，开关电源还包括分压电路E，分压电路E设置为生成分压信号，以使控制模块IC获取开关电源的输出电压Vout。
可选地，分压电路E包括相互串联的第二电阻R2和第三电阻R3。
可选地，开关电源还包括吸收电路Q，吸收电路Q包括相互串联的第一电阻R1和第一电容C1。吸收电路Q设置为优化系统开关损耗，降低开关管的电压和/或电流尖峰，以及改善开关电源的电磁干扰特性。
需要说明的是，第一电阻R1、第二电阻R2和第三电阻R3可以是任意一种电阻，上述电阻的种类和参数可以根据开关电源拟取得的供电效果进行适应性调整，本申请实施例对此不进行限制。示例性地，上述电阻均可以是贴片电 阻。
还需要说明的是，第一电容C1可以是任意一种电容，该电容的种类和参数可以根据开关电源拟取得的供电效果进行适应性调整，本申请实施例对此不进行限制。示例性地，第一电容C1可以是云母电容。
此外，本申请实施例所提供的降压架构的非隔离开关电源的电路元件连接关系如图1所示，在此不再进行赘述。可以理解的是，图1中的Vin表示开关电源的输入电压，Iin表示流过滤波电感的电流。
继续参见图1，示例性地，开关电源的工作过程如下：
开关电源启动后，控制模块IC获取连接点SW的电压波形。当连接点SW的电压波形未实现零电压切换时，控制模块IC调节第二开关管M2的截止电流，直至连接点SW的电压波形恰好归高。当连接点SW的电压波形恰好归高时，控制模块IC控制第一开关管M1导通，以实现第一开关管M1的零电压开启。当开关电源的输出端负载发生改变，且第一开关管M1的开关电流大于第一预设值时，根据开关电源的输出电压Vout，控制模块IC调节第一开关管M1的电流峰值，进而实现开关电源的输出电压Vout的稳定环路控制。在实现开关电源的输出电压Vout的稳定环路控制之后，当开关电源的输出端负载发生改变，且第一开关管M1的开关电流小于或等于第一预设值时，控制模块IC基于适当的防抖动阀值使得第一开关管M1的开关电流保持在第一预设值附近，并改变第一开关管M1的驱动信号的频率，进入轻载ZVS模式以调节开关电源的输出电压Vout。在进入轻载ZVS模式以调节开关电源的输出电压Vout之后，当开关电源的输出端负载发生改变，且第二开关管M2的过零关断和开启之间的时间间隔等于或小于第二预设值时，控制模块IC再次进入重载ZVS模式，调节第一开关管M1的峰值电流。
示例性地，本申请实施例中的控制模块IC可以在内部集成典型的电压基准模块，或者可以采用外置的电压基准模块，以实现开关电源的输出电压Vout与电压基准模块提供的模块基准电压的比较，进而实现不同状态下开关电源输 出电压Vout的可控调节。
图6是本申请实施例提供的一种重载ZVS模式下开关电源的波形示意图。参见上述开关电源的工作过程及图6，示例性地，当连接点SW的电压V SW的波形恰好归高时，第一开关管M1导通，第二开关管M2处于关断状态。此外，当第二开关管M2的开关电流I S达到截止电流I S_min时，第二开关管M2关断。
图7是本申请实施例提供的一种轻载ZVS模式下开关电源的波形示意图。参见上述开关电源的工作过程及图7，示例性地，在轻载ZVS模式下，控制模块IC根据分压电路E反馈的电压信号对第一开关管M1驱动信号的频率，或者第一开关管M1开启或关断的时间间隔进行调节，当M1驱动信号的频率，或者M1开启或关断的时间间隔满足预设值要求时，控制模块IC开启第二开关管M2，并调节第二开关管M2的截止电流I S_min，使得连接点SW的电压波形恰好归高；当SW的电压波形恰好归高时，控制模块IC开启第一开关管M1，当第一开关管M1的开关电流达到第一预设值，或者基于某一防抖阈值使第一开关管M1的开关电流到达第一预设值附近时，第一开关管M1关断，第二开关管M2开启；当第二开关管M2的开关电流I S处于电流过零点时，第二开关管M2关断，至此，本实施例完成了轻载ZVS模式的一个周期，并等待下一频率或者开关时间间隔满足时刻的到来。
本申请实施例能够实现重载状态下的开关电源ZVS，在开关电源的输出电压Vout保持稳态的基础上，当开关电源由重载状态转换为轻载状态时，本实施例能够通过频率调节方式实现轻载ZVS状态下开关电源的效率最优化。此外，当开关电源由轻载状态转换为重载状态时，本实施例能够再次进入重载ZVS。
与相关技术相比，本实施例的技术方案在不增加额外电路成本和控制复杂程度的前提下，能够根据开关电源的输出端负载变化实现轻载ZVS频率调节方式与重载ZVS峰值电流调节方式的实时转换，不仅有效降低了开关电源的 开关损耗，还有利于改善开关电源的EMI特性。
需要说明的是，本申请实施例可以但不限于采用轻载ZVS模式对开关电源的输出电压Vout进行调节。示例性地，在实现开关电源的输出电压的稳定环路控制之后，当开关电源的输出端负载发生改变，且第一开关管M1的开关电流小于或等于第一预设值时，控制模块IC控制第一开关管M1的开关电流保持在第一预设值，并采用常用的非零电压开启方式，在连接点SW的电压波形无归高过程的情况下，改变第一开关管M1的驱动信号频率，实现对开关电源输出电压Vout的调节。示例性地，当第一开关管M1的驱动信号频率，或者开启或关断的时间间隔满足频率或时间间隔要求时，控制模块IC开启第一开关管M1；当第一开关管M1的开关电流达到第一预设值，或者基于某一预设防抖阈值使得第一开关管M1的开关电流到达第一预设值附近时，第一开关管M1关断，第二开关管M2开启；当第二开关管M2的开关电流I S处于电流过零点时，第二开关管M2关断。至此，本实施例实现了轻载非ZVS模式的一个周期，并等待下一满足频率或时间间隔要求的时刻到来。
在上述实施例的基础上，示例性地，图8是本申请实施例提供的另一种开关电源的结构示意图。示例性地，图8所示的开关电源为升压架构的非隔离同步开关电源，本实施例所提供的开关电源的电路元件连接关系如图8所示，在此不再进行赘述。可以理解的是，图8中的Vin’表示升压架构的非隔离同步开关电源的输入电压，L’表示升压架构的非隔离同步开关电源的输入电感，Iin’表示升压架构的非隔离同步开关电源的电感电流，Q’表示升压架构的非隔离同步开关电源的吸收电路。
继续参见图8，示例性地，升压架构的非隔离同步开关电源的工作过程如下：
开关电源启动后，控制模块IC’获取连接点SX的电压波形。当连接点SX的电压波形未实现零电压切换时，控制模块IC’调节第四开关管M4的截止电流，直至连接点SX的电压波形接近归低或恰好归低。当连接点SX的电压波 形恰好归低时，控制模块IC’控制第三开关管M3开启。当开关电源的输出端负载发生改变，且第三开关管M3的开关电流大于第三预设值时，根据开关电源的输出电压Vout’，控制模块IC’调节第三开关管M3的电流峰值，进而实现开关电源的输出电压Vout’的稳定环路控制。在实现开关电源的输出电压Vout’的稳定环路控制之后，当开关电源的输出端负载发生改变，且第三开关管M3的开关电流小于或等于第三预设值时，控制模块IC’基于适当的防抖动阀值使得第三开关管M3的开关电流保持在第三预设值附近，并改变第三开关管M3的驱动信号的频率，进入轻载ZVS模式以调节开关电源的输出电压Vout’。在进入轻载ZVS模式以调节开关电源的输出电压Vout’之后，当开关电源的输出端负载发生改变，且第四开关管M4的过零关断和开启之间的时间间隔等于或小于第四预设值时，控制模块IC’再次进入重载ZVS模式，调节第三开关管M3的峰值电流。
可以理解的是，连接点SX的电压波形恰好归低是指，连接点SX的电压波形恰好恢复到零电压的低电平状态，此时，第三开关管M3源极与漏极间的电压差为零。
图9是本申请实施例提供的另一种重载ZVS模式下开关电源的波形示意图。参见上述升压架构的非隔离同步开关电源的工作过程及图9，示例性地，当连接点SX的电压V SX的波形恰好归低时，第三开关管M3导通，第四开关管M4处于关断状态。当第四开关管M4的开关电流I t达到截止电流I T_min时，第四开关管M4关断。
图10是本申请实施例提供的另一种轻载ZVS模式下开关电源的波形示意图。参见上述升压架构的非隔离同步开关电源的工作过程及图10，示例性地，在轻载ZVS模式下，控制模块IC’根据分压电路E’反馈的电压信号对第三开关管M3驱动信号的频率，或者第三开关管M3开启或关断的时间间隔进行调节，当M3驱动信号的频率，或者M3开启或关断的时间间隔满足要求时，控制模块IC’开启第四开关管M4，并调节第四开关管M4的截止电流I T_min， 直至连接点SX的电压波形恰好归低；当SX的电压波形恰好归低时，控制模块IC’开启第三开关管M3，当第三开关管M3的开关电流达到第三预设值，或者基于某一预设防抖阈值使得第三开关管M3的开关电流到达第三预设值附近时，第三开关管M3关断，第四开关管M4开启；之后，当第四开关管M4的开关电流I t处于电流过零点时，第四开关管M4关断，至此，本实施例完成了轻载ZVS模式的一个周期，并等待下一频率满足时刻的到来。
在上述实施例的基础上，示例性地，图11是本申请实施例提供的又一种开关电源的结构示意图。示例性地，图11所示的开关电源为升降压架构的非隔离同步开关电源，本实施例所提供的开关电源的电路元件连接关系如图11所示，在此不再进行赘述。参见图11，Vin”表示升降压架构的非隔离同步开关电源的输入电压，IC”表示升降压架构的非隔离同步开关电源的控制模块，Iin”表示升降压架构的非隔离同步开关电源的电感电流，Vout”表示升降压架构的非隔离同步开关电源的输出电压。
可以理解的是，升降压架构的非隔离同步开关电源可以由升压和降压架构的非隔离同步开关电源的部分电路结构对接而成，升降压架构的非隔离同步开关电源包括升压模式、降压模式和升降压模式。可以理解的是，升压模式和降压模式所对应的节点波形图可以由前述升压和降压架构的非隔离同步开关电源的多个波形图组成，并分别出现在升压架构或降压架构的节点上；升降压模式可以是升压模式与降压模式的复合模式，当开关电源处于升降压模式时，升压架构和降压架构的节点上的波形可以同时或交替出现，本申请实施例对此不进行限制。此处所述的升压模式与降压模式的复合模式是指，升降压架构非隔离同步开关电源中的升压架构非隔离同步开关电源部分与降压架构非隔离同步开关电源部分同时或交替处于工作状态；升压模式是指，升降压架构非隔离同步开关电源中的升压架构非隔离同步开关电源部分处于工作状态，而降压架构非隔离同步开关电源部分处于不交替开关的状态；降压模式是指，升降压架构非隔离同步开关电源中的升压架构非隔离同步开关电源部分处于不交替开关的状 态，而降压架构非隔离同步开关电源部分处于工作状态。
需要说明的是，当升降压架构的非隔离同步开关电源处于升压模式时，第一开关管M1处于实时开启状态，第二开关管M2处于实时关断状态。相反地，当升降压架构的非隔离同步开关电源处于降压模式时，第四开关管M4处于实时开启状态，第三开关管M3处于实时关断状态。

Claims (10)

  1. 一种开关电源的控制方法，所述开关电源包括第一开关管(M1)、第二开关管(M2)、滤波电感(L)以及控制模块(IC)；其中，所述第一开关管(M1)连接于电压输入端和所述滤波电感(L)之间，所述第二开关管(M2)连接于所述第一开关管(M1)和接地端之间，所述滤波电感(L)连接于所述第一开关管(M1)和所述开关电源的电压输出端之间，所述控制模块(IC)设置为控制所述第一开关管(M1)和所述第二开关管(M2)导通或者关断；
    所述控制方法包括：
    所述开关电源启动后，所述控制模块获取所述第一开关管与所述滤波电感之间的连接点的电压波形；
    响应于所述连接点的电压波形未实现零电压切换，所述控制模块调节所述第二开关管的截止电流，直至所述连接点的电压波形恰好归高；
    响应于所述连接点的电压波形恰好归高，所述控制模块控制所述第一开关管导通，以实现所述第一开关管的零电压开启。
  2. 根据权利要求1所述的方法，其中，所述开关电源还包括分压电路(E)，所述分压电路(E)连接于所述滤波电感(L)和所述控制模块(IC)的输出电压检测端(S0)；
    所述方法还包括：
    响应于所述开关电源的输出端负载发生改变，且所述第一开关管的开关电流大于第一预设值，根据所述开关电源的输出电压，所述控制模块调节所述第一开关管的电流峰值，进而实现所述开关电源的输出电压的稳定环路控制，控制模块采用重载零电压开关ZVS模式。
  3. 根据权利要求2所述的方法，在实现所述开关电源的输出电压的稳定环路控制之后，还包括：
    响应于所述开关电源的输出端负载发生改变，且所述第一开关管的开关电流小于或等于所述第一预设值，所述控制模块控制所述第一开关管的开关电流保持在所述第一预设值，并改变所述第一开关管的驱动信号的频率，进入轻载 ZVS模式以调节所述开关电源的输出电压。
  4. 根据权利要求3所述的方法，在进入轻载ZVS模式以调节开关电源的输出电压之后，还包括：
    响应于所述开关电源的输出端负载发生改变，且所述第二开关管的过零关断和开启之间的时间间隔等于或小于第二预设值，所述控制模块再次进入重载ZVS模式，调节所述第一开关管的峰值电流。
  5. 根据权利要求1-4任一项所述的方法，其中，所述开关电源还包括吸收电路(Q)，所述吸收电路(Q)并联于所述第二开关管(M2)的两端；所述吸收电路(Q)包括相互串联的第一电阻(R1)和第一电容(C1)。
  6. 根据权利要求2-5中任一项所述的方法，其中，所述分压电路(E)包括相互串联的第二电阻(R2)和第三电阻(R3)。
  7. 一种开关电源，包括第一开关管(M1)、第二开关管(M2)、滤波电感(L)以及控制模块(IC)；
    所述第一开关管(M1)设置为根据所述控制模块(IC)产生的主驱动信号导通或者关断；
    所述第二开关管(M2)设置为根据所述控制模块产生的同步驱动信号导通或者关断；
    所述滤波电感(L)设置为平滑所述开关电源的输出电流；
    所述控制模块(IC)设置为在所述开关电源启动后，获取所述第一开关管(M1)与所述滤波电感(L)之间的连接点(SW)的电压波形；还设置为当所述连接点(SW)的电压波形未实现零电压切换时，调节所述第二开关管(M2)的截止电流，直至所述连接点(SW)的电压波形恰好归高；还设置为当所述连接点(SW)的电压波形恰好归高时，控制所述第一开关管(M1)导通，以实现所述第一开关管(M1)的零电压开启；还设置为当所述开关电源的输出端负载发生改变，且所述第一开关管(M1)的开关电流大于第一预设值时，根据所述开关电源的输出电压(Vout)，调节所述第一开关管(M1)的电 流峰值，进而实现所述开关电源的输出电压(Vout)的稳定环路控制；还设置为在实现所述开关电源的输出电压(Vout)的稳定环路控制之后，当所述开关电源的输出端负载发生改变，且所述第一开关管(M1)的开关电流小于或等于所述第一预设值时，控制所述第一开关管(M1)的开关电流保持在所述第一预设值，并改变所述第一开关管(M1)的驱动信号的频率，进入轻载ZVS模式以调节所述开关电源的输出电压(Vout)；还设置为在进入轻载ZVS模式以调节开关电源的输出电压之后，当所述开关电源的输出端负载发生改变，且所述第二开关管(M2)的过零关断和开启之间的时间间隔等于或小于第二预设值时，再次进入重载ZVS模式，调节所述第一开关管(M1)的峰值电流。
  8. 根据权利要求7所述的开关电源，还包括分压电路(E)，所述分压电路(E)设置为生成分压信号，以使所述控制模块(IC)获取所述开关电源的输出电压(Vout)。
  9. 根据权利要求7或8所述的开关电源，还包括吸收电路(Q)，所述吸收电路(Q)包括相互串联的第一电阻(R1)和第一电容(C1)；所述吸收电路(Q)设置为优化系统开关损耗，降低开关管的电压和电流尖峰中的至少之一，以及改善所述开关电源的电磁干扰特性。
  10. 根据权利要求8或9所述的开关电源，其中，所述分压电路(E)包括相互串联的第二电阻(R2)和第三电阻(R3)。
PCT/CN2022/121692 2021-09-28 2022-09-27 开关电源的控制方法和开关电源 WO2023051520A1 (zh)

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Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN106787719A (zh) * 2016-12-05 2017-05-31 深圳信息职业技术学院 一种pwm/pfm双模式自动切换的降压型dc‑dc转换器
CN107026568A (zh) * 2017-05-19 2017-08-08 矽力杰半导体技术(杭州)有限公司 控制电路、控制方法及开关电源
CN108390567A (zh) * 2018-04-04 2018-08-10 广州金升阳科技有限公司 一种零电压开关同步整流Boost电路、零电压开关Boost电路及其控制方法
CN112134443A (zh) * 2020-09-17 2020-12-25 西安交通大学 一种基于临界导通模式的软开关实现及自适应控制方法
CN113890314A (zh) * 2021-09-28 2022-01-04 力源科技有限公司 一种开关电源的控制方法和开关电源

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN106787719A (zh) * 2016-12-05 2017-05-31 深圳信息职业技术学院 一种pwm/pfm双模式自动切换的降压型dc‑dc转换器
CN107026568A (zh) * 2017-05-19 2017-08-08 矽力杰半导体技术(杭州)有限公司 控制电路、控制方法及开关电源
CN108390567A (zh) * 2018-04-04 2018-08-10 广州金升阳科技有限公司 一种零电压开关同步整流Boost电路、零电压开关Boost电路及其控制方法
CN112134443A (zh) * 2020-09-17 2020-12-25 西安交通大学 一种基于临界导通模式的软开关实现及自适应控制方法
CN113890314A (zh) * 2021-09-28 2022-01-04 力源科技有限公司 一种开关电源的控制方法和开关电源

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