WO2022247382A1 - 一种同时同频全双工信号接收方法 - Google Patents

一种同时同频全双工信号接收方法 Download PDF

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WO2022247382A1
WO2022247382A1 PCT/CN2022/079387 CN2022079387W WO2022247382A1 WO 2022247382 A1 WO2022247382 A1 WO 2022247382A1 CN 2022079387 W CN2022079387 W CN 2022079387W WO 2022247382 A1 WO2022247382 A1 WO 2022247382A1
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self
signal
interference
interference cancellation
resampling
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PCT/CN2022/079387
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English (en)
French (fr)
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张昌明
余显斌
李雪敏
沈捷
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之江实验室
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Publication of WO2022247382A1 publication Critical patent/WO2022247382A1/zh
Priority to US18/075,430 priority Critical patent/US11664966B2/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/14—Two-way operation using the same type of signal, i.e. duplex
    • H04L5/1461—Suppression of signals in the return path, i.e. bidirectional control circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
    • H04B1/12—Neutralising, balancing, or compensation arrangements
    • H04B1/123—Neutralising, balancing, or compensation arrangements using adaptive balancing or compensation means
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
    • H04B1/40—Circuits
    • H04B1/50—Circuits using different frequencies for the two directions of communication
    • H04B1/52—Hybrid arrangements, i.e. arrangements for transition from single-path two-direction transmission to single-direction transmission on each of two paths or vice versa
    • H04B1/525—Hybrid arrangements, i.e. arrangements for transition from single-path two-direction transmission to single-direction transmission on each of two paths or vice versa with means for reducing leakage of transmitter signal into the receiver

Definitions

  • the invention relates to the field of wireless communication and signal processing, in particular to a simultaneous and same-frequency full-duplex signal receiving method.
  • Simultaneous same-frequency full-duplex technology means that the sending and receiving links of the device work on the same time slot and the same frequency band. Compared with the existing half-duplex technology (including time division duplex and frequency division duplex), it can theoretically be used Spectrum efficiency increased by 100%. With the increasing shortage of spectrum resources and the increasing demand for wireless transmission rates, the simultaneous same-frequency full-duplex technology has become one of the core technologies of B5G/6G, and has attracted widespread attention in academia and industry. However, simultaneous same-frequency full-duplex technology faces prominent self-interference, and its power may far exceed the useful signal power, and it is difficult to eliminate it to an ideal level. Because of this, the performance indicators of prototypes developed by various research institutions and enterprises It is still unable to meet the needs of market applications, and there is still a large gap between the increase in spectral efficiency and the ideal value.
  • FIG. 1 is a structural schematic diagram of a simultaneous and same-frequency full-duplex transceiver.
  • the digital-to-analog converter (DAC) converts the signal from the digital domain to the analog domain, and the frequency of the analog signal is moved to the required frequency band by the mixer, and then the The power amplifier (Power Amplifier, PA) is amplified and sent out from the antenna.
  • Power Amplifier, PA Power Amplifier
  • the signal received from the antenna is amplified by a low-noise amplifier (LNA) and then mixed
  • LNA low-noise amplifier
  • ADC Analog-to-Digital Converter
  • the signal enters the receiving baseband to reconstruct the self-interference signal and realize self-interference cancellation.
  • self-interference cancellation can also be realized in the antenna domain or radio frequency domain, but the self-interference cancellation capability of the antenna domain or radio frequency domain is limited, and the performance of the receiver is mainly guaranteed by the baseband self-interference cancellation.
  • the present invention will aim at the baseband part
  • the signal receiving process is innovated.
  • the sending end and the receiving end use the same local oscillator, that is, the co-local oscillator structure to realize spectrum shifting.
  • the independent local oscillator structure can also be used, but the co-local oscillator structure can avoid the relatively large frequency caused by phase noise. large adverse effects.
  • the existing simultaneous and same-frequency full-duplex technology usually processes self-interference cancellation and useful signal reception separately, as shown in FIG. 2 .
  • the self-interference reference signal is used to reconstruct the self-interference signal, and then the reconstruction result is subtracted from the received signal to realize self-interference cancellation.
  • Self-interference reconstruction is generally done in an adaptive way, including least mean square (Least Mean Square, LMS) algorithm, recursive least square (Recursive Least Square, RLS) algorithm, frequency-domain block least mean square (Frequency-domain Block Least Mean Square) , FBLMS) algorithm, etc.
  • the self-interference cancellation result is regarded as a useful signal, and resampling is realized at the best sampling point of the useful signal through a timing synchronization loop.
  • the timing synchronization loop includes resampling, timing error extraction, loop filtering, digitally controlled oscillator ( Numerical Controlled Oscillator, NCO).
  • NCO Numerical Controlled Oscillator
  • both the self-interference reference signal and the self-interference part of the received signal originate from the local device, and the clock information has been aligned, so there is no need to perform timing synchronization, and the useful signal source after self-interference cancellation From the peer device, the clock frequency may deviate from that of the local device. Therefore, timing synchronization is required to realize the resampling recovery of the optimal sampling point.
  • the signal after self-interference cancellation includes self-interference residual, useful signal and noise, which will be used as the driving signal for self-interference adaptive reconstruction, and the actual driving signal is only self-interference residual, so useful signal and noise will interfere with self-interference Eliminate adverse effects. Due to the existence of the useful signal, it is difficult to control the residual self-interference to a negligible level, so if the cancellation result is directly regarded as the useful signal, the sensitivity of the receiver will be significantly deteriorated compared to the half-duplex system. At the same time, in the case of poor self-interference cancellation capability, the timing synchronization loop will also be affected by self-interference residuals.
  • the present invention intends to design a new method for simultaneous and same-frequency full-duplex signal reception to enhance the ability of self-interference cancellation and improve the demodulation performance of useful signals, thereby promoting the application of simultaneous and same-frequency full-duplex technology.
  • the purpose of the present invention is to provide a simultaneous and same-frequency full-duplex signal receiving method, so as to enhance self-interference elimination capability and improve the receiving performance of useful signals.
  • a method for receiving simultaneous and same-frequency full-duplex signals comprising the following steps:
  • Step 1 Use the transmitted baseband signal as the self-interference reference signal, and subtract it from the received signal after adaptively reconstructing the self-interference to realize primary self-interference cancellation;
  • Step 2 pass the primary self-interference eliminated signal through the timing synchronization loop, realize the timing recovery at the best sampling point of the useful signal by resampling a, and pass the timing error signal in the timing synchronization loop through low-pass filtering, Control re-sampling b1 and re-sampling b2 to realize the best sampling point recovery of self-interference reference signal and received signal respectively;
  • Step 3 Use the resampled self-interference reference signal and the received signal to perform joint self-interference cancellation and equalization, and then complete the reception of useful signals through signal demodulation.
  • the primary self-interference cancellation adopts adaptive algorithms such as LMS, RLS, and FBLMS.
  • the timing synchronization loop includes resampling a, timing error extraction, loop filtering and numerically controlled oscillator; the resampling signal outputs a timing error signal after timing error extraction, and then loop filtering and numerically controlled oscillator Feedback to resampling a.
  • the loop filter uses a PI control filter.
  • timing error signal is firstly smoothed by low-pass filtering and processed by a numerically controlled oscillator, and then used to control the resampling b1 of the self-interference reference signal and the resampling b2 of the received signal.
  • the low-pass filtering is realized by using the I branch in the PI control filter.
  • the joint self-interference cancellation adopts adaptive algorithms such as LMS, RLS, and FBLMS.
  • the demodulation error is a decision error, including self-interference residual, equalization error and noise.
  • the present invention has the advantages of:
  • the signal after self-interference cancellation is used as the driving signal for self-interference adaptive reconstruction, including three parts: self-interference residual, useful signal and noise, but the actual driving signal is only self-interference residual, so the useful signal and noise will have adverse effects on self-interference cancellation, and the useful signal power is much greater than the noise power, which is the main factor restricting the ability of self-interference cancellation.
  • the simultaneous same-frequency full-duplex signal receiving method proposed by the present invention divides self-interference cancellation into two stages of primary self-interference cancellation and joint self-interference cancellation and equalization, and the timing synchronization of joint self-interference cancellation and equalization is changed from primary self-interference cancellation to joint self-interference cancellation and equalization Provide support.
  • the primary self-interference cancellation stage in the present invention does not put forward higher requirements for the self-interference cancellation capability.
  • the cancellation results only provide reference input for timing synchronization and are not used as a useful signal equalization solution. tuned input. Therefore, it is only necessary to ensure that the useful signal dominates after primary self-interference cancellation, and timing synchronization can realize timing recovery at the best sampling point of the useful signal, thereby greatly relieving the pressure of primary self-interference cancellation.
  • the joint implementation of self-interference cancellation and useful signal equalization of the present invention that is, joint self-interference cancellation and equalization, which is completed under the drive of demodulation error, can avoid the influence of useful signals on self-interference cancellation, thereby Enhance the ability of self-interference cancellation and improve the reception performance of useful signals.
  • the present invention passes the timing error extraction result through a low-pass filtering process with a bandwidth smaller than that of the loop filtering, and then controls the resampling of the self-interference reference signal and the received signal after passing through the NCO.
  • the accuracy of timing error information can be improved, thereby improving the timing synchronization performance of self-interference reference signals and received signals;
  • low-pass filtering is outside the timing synchronization loop , will not adversely affect the convergence speed of the loop, and the low-pass filter can be directly provided by the I branch of the loop filter, without an additional low-pass filter.
  • the peak value of the resampling error under the technical solution of the present invention is an order of magnitude lower than that of the prior art solution, thereby fully ensuring the use of high-order modulation methods that are sensitive to resampling errors, and Capable of eliminating self-interference to negligible levels.
  • Fig. 1 is a schematic diagram of the structure of a full-duplex communication transceiver with the same frequency at the same time;
  • Fig. 2 is a kind of block diagram of existing simultaneous same-frequency full-duplex signal reception realization
  • Fig. 3 is a kind of synchronous co-frequency full-duplex signal reception realization block diagram provided by the present invention.
  • Fig. 4 is a block diagram of an implementation of loop filtering and low-pass filtering in the timing synchronization of the present invention
  • Fig. 5 is the resampling error comparison figure of the present invention and prior art scheme
  • Fig. 6 is a comparison chart of the amount of self-interference cancellation between the present invention and the prior art solution
  • Fig. 7 is a comparison chart of MSE performance between the present invention and the prior art solution.
  • the present invention will start from the self-interference reference signal and the received signal, jointly perform self-interference cancellation and useful signal equalization, and use the demodulation error of the useful signal to drive the entire joint adaptive process.
  • the demodulation error does not contain the useful signal after convergence. Therefore, this signal receiving method can avoid the restriction of the self-interference cancellation capability by the useful signal.
  • the joint self-interference cancellation and equalization needs to align the clocks of the self-interference reference signal and the received signal to the best sampling point of the useful signal, so that the demodulation of the useful signal can be performed correctly.
  • the present invention first performs the primary self-interference cancellation, and then extracts the timing error from the primary self-interference cancellation result, which is used to control the re-sampling of the self-interference reference signal and the received signal, so as to realize the optimal sampling point of the useful signal.
  • Timing recovery, and then perform joint self-interference cancellation and equalization on the resampled self-interference reference signal and received signal the implementation method is shown in Figure 3, and the specific steps are as follows:
  • Step 1 Use the transmitted baseband signal as a self-interference reference signal, and subtract it from the received signal after adaptively reconstructing the self-interference, so as to realize primary self-interference cancellation.
  • the implementation of the primary self-interference cancellation is similar to the self-interference cancellation in the prior art solution in FIG. 2 , and the self-interference can be reconstructed by using adaptive algorithms such as LMS, RLS, and FBLMS.
  • the primary self-interference cancellation stage in the present invention does not put forward higher requirements on the self-interference cancellation capability, and the cancellation result only provides reference input for timing synchronization, and does not serve as an input for equalized demodulation of useful signals. Therefore, it is only necessary to ensure that the useful signal dominates after primary self-interference cancellation, and timing synchronization can realize timing recovery at the best sampling point of the useful signal, thereby greatly relieving the pressure of primary self-interference cancellation.
  • Step 2 pass the signal after primary self-interference elimination through the timing synchronization loop, realize the timing recovery at the best sampling point of the useful signal by resampling a, and pass the timing error signal in the timing synchronization loop through low-pass filtering, Re-sampling b1 and re-sampling b2 are controlled to recover the best sampling points of the self-interference reference signal and the received signal respectively.
  • the timing synchronization loop structure of the signal after primary self-interference cancellation is similar to the timing synchronization loop in the prior art solution, including resampling a, timing error extraction, loop filtering, and NCO.
  • the present invention passes the timing error extraction result through a low-pass filtering process with a bandwidth smaller than that of the loop filter, and then controls the resampling of the self-interference reference signal and the received signal after passing through the NCO, that is, resampling b1 and resampling b2.
  • the timing error source of resampling b1 and resampling b2 is the same as that of resampling a, so both can achieve timing recovery at the best sampling point of the useful signal.
  • the smoothing effect of the low-pass filter the accuracy of the timing error information can be improved, thereby improving the timing synchronization performance of the self-interference reference signal and the received signal.
  • the present invention can guarantee the convergence characteristic on the basis of improving the timing synchronization performance of the self-interference reference signal and the received signal.
  • the low-pass filter will not adversely affect the convergence speed of the loop outside the timing synchronization loop; on the other hand, after the loop converges, the timing error extraction results will also tend to be stable, self-interference reference The resampling process of the signal will also converge.
  • Loop filtering is generally implemented by a PI control filter, which includes a P branch that is more sensitive to input errors and an I branch that is insensitive to input errors.
  • the P branch is a proportional link
  • the I branch is an integral link.
  • the parameter factor k i of the I branch is usually much smaller than the parameter k p of the P branch, so the passband bandwidth of the I branch is much smaller than that of the loop filter itself. Therefore, the low-pass filter required for timing synchronization in the present invention can be directly provided by the I branch of the loop filter, as shown in FIG. 4 , and no additional low-pass filter is constructed in this way.
  • Step 3 Using the resampled self-interference reference signal and the received signal to perform joint self-interference cancellation and equalization, and then complete the reception of useful signals through signal demodulation.
  • the demodulation error is the decision error, which includes self-interference residual, equalization error and noise. After the system converges, these signals are much smaller than the useful signal itself. In theory, the self-interference cancellation process is not affected by the useful signal. Elimination Capabilities can be significantly improved over primary self-interference cancellation.
  • the channel response of the self-interference channel and the useful signal both use the classic rummer model, including two paths, the first path is the main path, the delay of the second path relative to the first path is 6.3ns, and the notch depth is 3dB, that is, the amplitude of the second path relative to the first path is 1-10 -3/20 , and the phase of the second path is randomly distributed (the difference between the self-interference channel and the useful signal channel response is determined by the second path phase representation).
  • the communication symbol rate is 100MHz, and the signal is shaped by a root-raised cosine waveform with a roll-off coefficient of 0.2, that is, the communication bandwidth is 120MHz.
  • Both primary self-interference cancellation and joint self-interference cancellation and equalization are realized by FBLMS algorithm, and the adaptive update step constant is set as 2 -12 .
  • Figure 5 shows that after the timing synchronization loop converges, the resampling error of the present invention and the prior art scheme at the best sampling point of the useful signal is compared, and its relative symbol period has been normalized.
  • the signal-to-interference ratio (Signal -to-Interference Ratio, SIR) is set to -20dB, that is, the self-interference is 20dB larger than the useful signal power
  • the Signal-to-Noise Ratio (SNR) is set to 20dB.
  • the peak value of the resampling error under the technical solution of the present invention is an order of magnitude lower than that of the prior art solution, thus fully ensuring the sensitivity to resampling errors The use of higher order modulation schemes.
  • Fig. 6 compares the self-interference cancellation performance of the present invention and the prior art solution. Under different SIR and SNR, it can be regarded as changing the power of the self-interference signal and noise under the premise that the power of the useful signal remains unchanged. It can be seen that since the present invention avoids the influence of useful signals on the self-interference cancellation process, the amount of self-interference cancellation in each case is greater than that of the prior art solution. As the SNR increases, the amount of self-interference cancellation under the technical solution of the present invention can be continuously improved, which is due to the continuous reduction of the power of noise affecting self-interference cancellation. However, as the SNR increases, the self-interference cancellation capability under the existing technical solutions is almost unchanged.
  • the useful signal and noise affect the self-interference cancellation process at the same time, and the useful signal power is much greater than the noise power, which restricts the self-interference cancellation capability.
  • the main factor is that although increasing the SNR can reduce the noise power, the useful signal power does not change, so the self-interference cancellation capability cannot be significantly improved.
  • the smaller the SIR the greater the corresponding self-interference cancellation, because the smaller the SIR corresponds to the greater the self-interference signal power, so the larger the self-interference cancellation.
  • Figure 7 shows the useful signal demodulation mean square error (Mean Square Error, MSE) performance corresponding to various situations in Figure 6, which reflects the reception performance of the useful signal.
  • MSE mean square error
  • the amount of self-interference cancellation is limited under the existing technical solution, especially when the SNR is high, the self-interference cancellation residual is more significant than the noise, so the MSE decreases relatively slowly in the area of high SNR. As the SIR changes, the MSE performance under the two schemes is less affected. This is because the self-interference cancellation amount corresponding to different SIRs also changes correspondingly, and the self-interference cancellation residual is almost unchanged.

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Abstract

本发明公开了一种同时同频全双工信号接收方法，该方法包括：将发送基带信号作为自干扰参考信号，构建自干扰后对接收信号执行初级自干扰消除；将初级自干扰消除信号通过定时同步环路，由重采样a实现在有用信号最佳采样点的定时恢复，并将定时同步环路中的定时误差信号通过低通滤波后，控制重采样b1和重采样b2分别实现对自干扰参考信号和接收信号的最佳采样点恢复；利用重采样后的自干扰参考信号和接收信号执行联合自干扰消除与均衡，再通过信号解调完成对有用信号的接收。通过以上方法，可以显著增强同时同频全双工的自干扰消除能力，提升有用信号的接收性能。

Description

一种同时同频全双工信号接收方法 技术领域
本发明涉及无线通信和信号处理领域，尤其涉及一种同时同频全双工信号接收方法。
背景技术
同时同频全双工技术是指设备的发送和接收链路工作在相同时隙相同频段上，相对于现有的半双工技术(包括时分双工和频分双工)，理论上可使频谱效率提升100％。在频谱资源日益短缺和无线传输速率需求不断增长的情况下，同时同频全双工技术成为B5G/6G的核心技术之一，在学术界和产业界得到了广泛关注。然而，同时同频全双工技术面临突出的自干扰，其功率可能远超有用信号功率，而且难以消除到理想水平，也正因如此，当前各研究机构和企业开发出的原型机的性能指标还无法满足市场应用需求，频谱效率提升量离理想值还有较大差距。
图1为同时同频全双工收发机结构示意图。发送端信号经过基带处理之后，由数字模拟变换器(Digital-to-Analog Converter,DAC)将信号从数字域转到模拟域，经混频器将模拟信号的频率搬移到所需频段，再由功率放大器(Power Amplifier，PA)放大后从天线发出。发送信号的一部分通过天线泄露或环境反射及散射进入接收端，形成自干扰信号并与接收有用信号混叠，从天线接收的信号经过低噪声放大器(Low-Noise Amplifier，LNA)放大后，由混频器和滤波器共同作用得到对应的模拟基带信号，再经模拟数字变换器(Analog-to-Digital Converter，ADC)采样得到数字信号后进入接收基带，其中，发送的基带信号也作为自干扰参考信号进入接收基带中，用以重建自干扰信号并实现自干扰消除。除了接收基带之外，也可以在天线域或射频域实现自干扰消除，但天线域或射频域的自干扰消除能力有限，接收机性能主要由基带自干扰消除保障，本发明将针对基带部分的信号接收过程进行创新。此外，图1中发送端和接收端采用一个相同的本振即共本振结构实现频谱搬移，实际系统中也可以采用独立本振的结构，但共本振结构可以避免因相位噪声而造成较大的不利影响。
现有同时同频全双工技术通常将自干扰消除和有用信号接收过程分开处理，如图2所示。自干扰参考信号用于重建自干扰信号，再将重建结果从接收信号中减去实现自干扰消除。自干扰重建一般通过自适应方式完成，包括最小均方(Least Mean Square，LMS)算法、递归最小二乘(Recursive Least Square，RLS)算法、频域块最小均方(Frequency-domain Block Least Mean Square，FBLMS)算法等。随后，将自干扰消除结果视作有用信号，通过定时同步环路在有用信号的最佳采样点上实现重采样，定时同步环路包括重采样、定时误差提取、环路滤 波、数控振荡器(Numerical Controlled Oscillator，NCO)。最后，对重采样后的信号执行均衡和解调，即完成了有用信号的接收过程，均衡一般也在解调误差的驱动下自适应完成。这里，在自干扰消除过程中，自干扰参考信号和接收信号中的自干扰部分都源自于本端设备，时钟信息已经对齐，因此不需要执行定时同步，而自干扰消除后的有用信号源自于对端设备，时钟频率与本端设备可能存在偏差，因此需通过定时同步来实现最佳采样点的重采样恢复。
现有技术方案的主要问题是自干扰消除能力较差。自干扰消除后的信号包括自干扰残余、有用信号和噪声，它们将共同作为自干扰自适应重建的驱动信号，而实际需要的驱动信号仅为自干扰残余，所以有用信号和噪声会对自干扰消除产生不利影响。由于有用信号的存在，自干扰残余很难控制到可忽略的水平，所以如果将消除结果直接视作有用信号，接收机灵敏度相对于半双工系统会受到显著恶化。与此同时，在自干扰消除能力较差的情况下，定时同步环路也会因自干扰残余而受影响。通过降低自适应调整的步径因子虽然可以在一定程度上改善自干扰消除效果，但会大大降低收敛速度。基于以上问题，本发明拟设计一种同时同频全双工信号接收的新型方法，以增强自干扰消除的能力，提升有用信号的解调性能，从而促进同时同频全双工技术的应用。
发明内容
本发明的目的在于提供一种同时同频全双工信号接收方法，以增强自干扰消除能力并提升有用信号的接收性能。
为了实现上述目的，本发明采用了如下技术方案：
一种同时同频全双工信号接收方法，包括以下步骤：
步骤1：将发送基带信号作为自干扰参考信号，自适应重建自干扰后从接收信号中减去，实现初级自干扰消除；
步骤2：将初级自干扰消除后的信号通过定时同步环路，由重采样a实现在有用信号最佳采样点的定时恢复，并将定时同步环路中的定时误差信号通过低通滤波后，控制重采样b1和重采样b2，分别实现对自干扰参考信号和接收信号的最佳采样点恢复；
步骤3：利用重采样后的自干扰参考信号和接收信号执行联合自干扰消除与均衡，再通过信号解调完成对有用信号的接收。
进一步的，初级自干扰消除采用LMS、RLS、FBLMS等自适应算法。
进一步的，所述的定时同步环路包括重采样a、定时误差提取、环路滤波和数控振荡器；重采样信号经定时误差提取后输出定时误差信号，再经环路滤波、数控振荡器之后反馈至重采样a。
进一步的，所述的环路滤波采用PI控制滤波器。
进一步的，所述的定时误差信号先经过低通滤波平滑和数控振荡器处理之后，再用于控制自干扰参考信号的重采样b1和接收信号的重采样b2。
进一步的，所述的低通滤波采用PI控制滤波器中的I支路实现。
进一步的，联合自干扰消除与均衡在解调误差的驱动下完成。
进一步的，所述的联合自干扰消除采用LMS、RLS、FBLMS等自适应算法。
进一步的，所述的解调误差为判决误差，包括自干扰残余、均衡误差和噪声。
与现有技术相比，本发明的优势在于：
1)传统方法中将自干扰消除后的信号作为自干扰自适应重建的驱动信号，包括了自干扰残余、有用信号和噪声三部分，而实际需要的驱动信号仅为自干扰残余，所以有用信号和噪声会对自干扰消除产生不利影响，而且有用信号功率远大于噪声功率，是制约自干扰消除能力的主要因素。本发明提出的同时同频全双工信号接收方法，将自干扰消除划分为初级自干扰消除和联合自干扰消除与均衡两个阶段，由初级自干扰消除为联合自干扰消除与均衡的定时同步提供支撑。
其中，初级自干扰消除与现有技术方案不同，本发明中初级自干扰消除阶段并不对自干扰消除能力提出较高需求，其消除结果仅为定时同步提供参考输入，并不作为有用信号均衡解调的输入。因此，只需保证在初级自干扰消除之后有用信号占据主导部分，定时同步就可以实现在有用信号最佳采样点的定时恢复，从而大大缓解了初级自干扰消除的实现压力。
相对于现有技术方案，本发明的自干扰消除与有用信号均衡联合执行，即联合自干扰消除与均衡，其在解调误差的驱动下完成，可以避免有用信号对自干扰消除的影响，从而增强自干扰消除能力，并提升有用信号的接收性能。
2)通过自干扰消除性能实验和有用信号解调均方误差性能实验，验证了本发明能有效避免有用信号对自干扰消除过程的影响，在各种情况下的自干扰消除量都大于现有技术方案，自干扰消除能力更强；且本发明技术方案的MSE在各种情况下都低于现有方案，对有用信号的接收性能更优。
3)本发明在传统的定时同步环路基础上，将定时误差提取结果经过一个带宽比环路滤波更小的低通滤波过程，再经NCO后控制自干扰参考信号和接收信号的重采样。一方面，由于低通滤波的平滑作用，定时误差信息的准确度可以得到提高，进而能够提升自干扰参考信号和接收信号的定时同步性能；另一方面，低通滤波在定时同步环路之外，不会对环路的收敛速度产生不利影响，且低通滤波可以直接由环路滤波的I支路提供，无需额外构建低通滤波器。
4)通过定时同步重采样实验验证，本发明技术方案下的重采样误差峰值相对于现有技术 方案低了一个数量级，从而充分保障了对重采样误差较敏感的高阶调制方式的使用，且能够将自干扰消除到可忽略的水平。
附图说明
图1为同时同频全双工通信收发机结构示意图；
图2为一种现有的同时同频全双工信号接收实现框图；
图3为本发明提供的一种同时同频全双工信号接收实现框图；
图4为本发明定时同步中环路滤波与低通滤波的一种实现方式框图；
图5为本发明与现有技术方案的重采样误差对比图；
图6为本发明与现有技术方案的自干扰消除量对比图；
图7为本发明与现有技术方案的MSE性能对比图。
具体实施方式
下面结合附图和实施例，对本发明做进一步的描述。
本发明将从自干扰参考信号和接收信号出发，联合执行自干扰消除与有用信号均衡，采用有用信号解调误差来驱动整个联合自适应过程，如此，在收敛之后解调误差不含有用信号，所以这种信号接收方式可以避免有用信号对自干扰消除能力的制约。联合自干扰消除与均衡需要自干扰参考信号与接收信号的时钟对齐到有用信号的最佳采样点，才能使有用信号解调能被正确执行。然而，在自干扰参考信号和接收信号中，有用信号都不是主要部分，无法通过这两个信号本身提取出定时误差后实现在有用信号最佳采样点的定时恢复。为此，本发明先执行初级自干扰消除，然后从初级自干扰消除结果中提取出定时误差，用于控制自干扰参考信号与接收信号的重采样，以实现在有用信号最佳采样点上的定时恢复，再对重采样后的自干扰参考信号和接收信号执行联合自干扰消除与均衡，实现方式如图3所示，具体步骤方案如下：
步骤一：将发送基带信号作为自干扰参考信号，自适应重建自干扰后从接收信号中减去，实现初级自干扰消除。
初级自干扰消除的实现方式类似于图2中现有技术方案中的自干扰消除，可以采用LMS、RLS、FBLMS等自适应算法重建自干扰。与现有技术方案不同，在本发明中初级自干扰消除阶段并不对自干扰消除能力提出较高需求，其消除结果仅为定时同步提供参考输入，并不作为有用信号均衡解调的输入。因此，只需保证在初级自干扰消除之后有用信号占据主导部分，定时同步就可以实现在有用信号最佳采样点的定时恢复，从而大大缓解了初级自干扰消除的 实现压力。
步骤二：将初级自干扰消除后的信号通过定时同步环路，由重采样a实现在有用信号最佳采样点的定时恢复，并将定时同步环路中的定时误差信号通过低通滤波后，控制重采样b1和重采样b2，分别实现对自干扰参考信号和接收信号的最佳采样点恢复。
初级自干扰消除后信号的定时同步环路结构与现有技术方案中的定时同步环路类似，包括重采样a、定时误差提取、环路滤波、NCO几个部分。本发明在此基础上，将定时误差提取结果经过一个带宽比环路滤波更小的低通滤波过程，再经NCO后控制自干扰参考信号和接收信号的重采样，即重采样b1和重采样b2。重采样b1和重采样b2的定时误差来源与重采样a相同，因此都能实现在有用信号最佳采样点的定时恢复。此外，由于低通滤波的平滑作用，定时误差信息的准确度可以得到提高，进而能够提升自干扰参考信号和接收信号的定时同步性能。
本发明在提升自干扰参考信号和接收信号的定时同步性能的基础上，可以使收敛特性得到保障。一方面，低通滤波在定时同步环路之外，不会对环路的收敛速度产生不利影响；另一方面，环路收敛后，定时误差提取结果也趋于稳定，自干扰参考信号和接收信号的重采样过程也将收敛。
环路滤波一般采用PI控制滤波器实现，其包含对输入误差较敏感的P支路以及对输入误差不敏感的I支路。其中，P支路为比例环节，I支路为积分环节，I支路的参数因子k i通常远小于P支路的参数因子k p，因此I支路的通带带宽远小于环路滤波器本身。因此，本发明中定时同步所需的低通滤波可以直接由环路滤波的I支路提供，如图4所示，采用这种方式不用额外构建低通滤波器。
步骤三：利用重采样后的自干扰参考信号和接收信号执行联合自干扰消除与均衡，再通过信号解调完成对有用信号的接收。
联合自干扰消除与均衡将在解调误差的驱动下完成，同样可以采用LMS、RLS、FBLMS等自适应算法实现。这里，解调误差即为判决误差，其包括自干扰残余、均衡误差和噪声，在系统收敛之后，这些信号都远小于有用信号本身，在理论上自干扰消除过程不受有用信号的影响，消除能力相对于初级自干扰消除可以得到显著提升。
为进一步说明本发明相对于现有方案的效果，下面将针对典型场景，给出性能仿真结果。这里，自干扰信道和有用信号的信道响应都采用经典的rummer模型，包括两条径，第一条径为主径，第二条径相对于第一条径的时延为6.3ns，notch深度为3dB，即第二条径相对于第一条径的幅度为1-10 -3/20，第二条径的相位随机分布(自干扰信道和有用信号信道响应的差异由第二条径的相位体现)。通信符号速率为100MHz，信号采用滚降系数为0.2的根升余弦 波形进行成型，即通信带宽为120MHz。初级自干扰消除和联合自干扰消除与均衡均采用FBLMS算法实现，自适应更新步径常数取为2 -12。此外，发送有用信号的对端设备时钟与本端设备时钟差异为10ppm，定时同步环路滤波器的参数设置为k p＝10 -2，k i＝10 -5，图3中低通滤波采用如图4所示的环路滤波的I支路实现，无需额外构建低通滤波器。
图5所示为定时同步环路收敛之后，本发明与现有技术方案在有用信号最佳采样点的重采样误差对比，其相对符号周期进行了归一化处理，这里，信干比(Signal-to-Interference Ratio，SIR)设置为-20dB，即自干扰比有用信号功率大20dB，信噪比(Signal-to-Noise Ratio，SNR)设置为20dB。从图中可以看到，在对定时误差提取结果进行低通滤波之后，本发明技术方案下的重采样误差峰值相对于现有技术方案低了一个数量级，从而充分保障了对重采样误差较敏感的高阶调制方式的使用。
图6比较了本发明与现有技术方案的自干扰消除性能，在不同SIR和SNR下，可以视作在有用信号功率不变的前提下改变自干扰信号和噪声的功率。可以看出，由于本发明避免了有用信号对自干扰消除过程的影响，在各种情况下的自干扰消除量都大于现有技术方案。随着SNR增大，本发明技术方案下的自干扰消除量能不断得到提升，这是由于影响自干扰消除的噪声的功率不断降低所致。然而，随着SNR增大，现有技术方案下自干扰消除能力几乎不变，此时有用信号和噪声同时影响自干扰消除过程，而且有用信号功率远大于噪声功率，是制约自干扰消除能力的主要因素，增大SNR虽然可以降低噪声功率，但有用信号功率却并不会发生变化，因此自干扰消除能力无法得到明显提升。此外，在两种方案下，SIR越小，所对应的自干扰消除量都更大，这是因为越小的SIR对应越大的自干扰信号功率，因此自干扰消除量也越大。
图7给出了图6中各种情况下对应的有用信号解调均方误差(Mean Square Error，MSE)性能，其反应了有用信号的接收性能。本发明技术方案的MSE在各种情况下都低于现有方案，特别是在SNR较大情况下优势更为明显。从图中可见，以dB为单位的MSE值随着SNR的增大而线性降低，说明了影响MSE性能的主要因素是噪声，在本发明技术方案下，自干扰被消除到可忽略的水平。现有技术方案下自干扰消除量有限，特别是SNR较高时自干扰消除残余相对于噪声较为显著，所以MSE在SNR较高的区域下降得相对缓慢。随着SIR的变化，两种方案下的MSE性能受影响较小，这是由于不同的SIR对应的自干扰消除量也相应变化，自干扰消除残余几乎不变。

Claims (7)

  1. 一种同时同频全双工信号接收方法，其特征在于，包括以下步骤：
    步骤1：将发送基带信号作为自干扰参考信号，自适应重建自干扰后从接收信号中减去，实现初级自干扰消除；
    步骤2：将初级自干扰消除后的信号通过定时同步环路，由重采样a实现在有用信号最佳采样点的定时恢复，并将定时同步环路中的定时误差信号通过低通滤波后，控制重采样b1和重采样b2，分别实现对自干扰参考信号和接收信号的最佳采样点恢复；
    步骤3：利用重采样后的自干扰参考信号和接收信号执行联合自干扰消除与均衡，再通过信号解调完成对有用信号的接收。
  2. 根据权利要求1所述的同时同频全双工信号接收方法，其特征在于，初级自干扰消除采用LMS、RLS或FBLMS自适应算法。
  3. 根据权利要求1所述的同时同频全双工信号接收方法，其特征在于，所述的定时同步环路包括重采样a、定时误差提取、环路滤波和数控振荡器；重采样信号经定时误差提取后输出定时误差信号，再经环路滤波、数控振荡器之后反馈至重采样a。
  4. 根据权利要求3所述的同时同频全双工信号接收方法，其特征在于，所述的定时误差信号先经过低通滤波平滑和数控振荡器处理之后，再用于控制自干扰参考信号的重采样b1和接收信号的重采样b2。
  5. 根据权利要求1所述的同时同频全双工信号接收方法，其特征在于，联合自干扰消除与均衡在解调误差的驱动下完成。
  6. 根据权利要求5所述的同时同频全双工信号接收方法，其特征在于，所述的联合自干扰消除采用LMS、RLS或FBLMS自适应算法。
  7. 根据权利要求5所述的同时同频全双工信号接收方法，其特征在于，所述的解调误差为判决误差，包括自干扰残余、均衡误差和噪声。
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