WO2022001943A1 - 纹波电流控制方法和装置、电子设备、计算机可读存储介质 - Google Patents

纹波电流控制方法和装置、电子设备、计算机可读存储介质 Download PDF

Info

Publication number
WO2022001943A1
WO2022001943A1 PCT/CN2021/102713 CN2021102713W WO2022001943A1 WO 2022001943 A1 WO2022001943 A1 WO 2022001943A1 CN 2021102713 W CN2021102713 W CN 2021102713W WO 2022001943 A1 WO2022001943 A1 WO 2022001943A1
Authority
WO
WIPO (PCT)
Prior art keywords
phase
voltage
certain
sign
phase voltage
Prior art date
Application number
PCT/CN2021/102713
Other languages
English (en)
French (fr)
Inventor
鄂本
周建平
林国仙
董秀锋
王恰
樊珊珊
崔玉龙
张伟
Original Assignee
中兴通讯股份有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 中兴通讯股份有限公司 filed Critical 中兴通讯股份有限公司
Priority to EP21833948.9A priority Critical patent/EP4170885A4/en
Priority to JP2022581584A priority patent/JP7418624B2/ja
Publication of WO2022001943A1 publication Critical patent/WO2022001943A1/zh

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/123—Suppression of common mode voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J3/00—Circuit arrangements for ac mains or ac distribution networks
    • H02J3/01—Arrangements for reducing harmonics or ripples
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4216—Arrangements for improving power factor of AC input operating from a three-phase input voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4233—Arrangements for improving power factor of AC input using a bridge converter comprising active switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/2173—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a biphase or polyphase circuit arrangement
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/40—Arrangements for reducing harmonics

Definitions

  • the present disclosure relates to, but is not limited to, the technical field of switching power supply conversion.
  • the input ripple current may exceed the standard range, and the input ripple current is too large, which will increase the loss of the rectifier and reduce the efficiency. , which affects the input and output indicators. Therefore, the input ripple current needs to be controlled.
  • the related methods of reducing the ripple current either increase the cost of the device, or are complicated in calculation and control.
  • the present disclosure provides a ripple current control method, which includes: if the phase voltage of a certain phase satisfies the phase-shift condition, determining the voltage of each corresponding switch tube according to a corresponding first carrier wave.
  • Drive signal wherein, the phase of the first carrier is 180° out of phase with the phase of a corresponding second carrier, and the second carrier is used to determine a corresponding phase voltage when the phase voltage of a certain phase does not meet the phase-shift condition.
  • the carrier of the drive signal of a switch includes: if the phase voltage of a certain phase satisfies the phase-shift condition, determining the voltage of each corresponding switch tube according to a corresponding first carrier wave.
  • the phase of the first carrier is 180° out of phase with the phase of a corresponding second carrier, and the second carrier is used to determine a corresponding phase voltage when the phase voltage of a certain phase does not meet the phase-shift condition.
  • the present disclosure provides an electronic device, comprising: at least one processor; and a memory, where at least one program is stored, and when the at least one program is executed by the at least one processor, the at least one process
  • the controller implements any of the ripple current control methods described herein.
  • the present disclosure provides a computer-readable storage medium on which a computer program is stored, and when the computer program is executed by a processor, implements any one of the ripple current control methods described herein.
  • FIG. 1 is a block diagram of a three-phase rectifier provided by the present disclosure
  • FIG. 2 is a block diagram of the composition of the power conversion unit of the three-phase rectifier provided by the present disclosure
  • FIG. 3 is a block diagram of the composition of the power conversion unit of the three-phase rectifier provided by the present disclosure
  • FIG. 4 is a block diagram of a control unit of a three-phase rectifier provided by the present disclosure
  • FIG. 5 is a waveform diagram of a three-phase phase voltage provided by the present disclosure.
  • FIG. 6 is a waveform diagram of a drive signal of a switch tube provided by the present disclosure.
  • FIG. 8 is a waveform diagram of a drive signal of a switch tube provided by the present disclosure.
  • FIG. 9 is a waveform diagram of a drive signal of a switch tube provided by the present disclosure.
  • FIG. 10 is a block diagram of a ripple current control device provided by the present disclosure.
  • Three-phase power factor correction (PFC, Power Factor Correction) rectifiers mainly include passive PFC rectifiers (ie passive PFC rectifiers) and active PFC rectifiers (ie active PFC rectifiers).
  • the passive PFC rectifier has the disadvantages of high harmonic content of the input current, and the output voltage is subject to the input voltage, which cannot keep up with the current development trend of the rectifier market. Therefore, active PFC rectifiers are widely used in current rectifier products.
  • the three-phase active PFC rectifier rectifies the input AC voltage into a DC voltage output while maintaining the phase synchronization of the three-phase input current and voltage without distortion.
  • the three-phase PFC rectifier currently used has various topologies.
  • the performance of the traditional uncontrolled rectifier or half-controlled rectifier device is far from meeting the current needs due to its own incomplete control.
  • the traditional two-level rectifier topology is also in line with the user's requirements for high voltage, high power and high performance. far cry.
  • the Vienna rectifier has the advantages of simple structure, low stress on switching tube devices, no danger of shoot-through and low harmonic content. Wide range of occasions; Totem (Totem) rectifier has the advantages of fewer components, low conduction loss, and high efficiency, and its application in three-phase rectifiers has also attracted more and more attention.
  • the input ripple current may exceed the standard range. If the input ripple current is too large, it will increase the loss of the rectifier, reduce the efficiency, and affect the input and output indicators. Therefore, the input ripple current needs to be controlled.
  • the common hardware methods for reducing the input ripple current include the following two methods: method 1, inductive coupling, this method needs to add a large number of components such as inductors and switch tubes, which will increase the cost of the device; method 2, The staggered parallel technology is adopted. This method divides each phase into two parallel branches, and then staggeredly drives them respectively, which increases the control complexity while increasing the cost of the device.
  • DSP Digital Signal Processing
  • ripple current control method and apparatus, electronic device, and computer-readable storage medium can realize the ripple current control of the three-phase rectifier and the three-phase inverter.
  • the following takes a three-phase rectifier as an example for description, and the implementation process of a three-phase inverter is similar.
  • FIG. 1 is a block diagram of a three-phase rectifier provided by the present disclosure. As shown in FIG. 1 , the three-phase rectifier includes a power conversion unit 101 , a sampling unit 102 , a control unit 103 and a driving unit 104 .
  • the power conversion unit 101 is configured to convert the AC voltage of the input power source into a DC voltage;
  • the sampling unit 102 is configured to measure the three-phase phase voltage of the input power source, the current of the three-phase inductor and the three-phase bus (BUS) voltage (that is, the voltage at both ends of the BUS capacitor) is sampled;
  • the control unit 103 is configured to calculate each switch tube according to the three-phase phase voltage of the input power, the current of the three-phase inductor and the three-phase BUS voltage sampled by the sampling unit 102
  • the drive signal is output to the drive unit 104;
  • the drive unit 104 is configured to drive the switch tube of the power conversion unit 101 on and off according to the drive signal, thereby controlling the output BUS voltage.
  • sampling unit 102 and the control unit 103 are implemented by the same DSP chip.
  • FIG. 2 is a block diagram of a power conversion unit of a three-phase rectifier provided by the present disclosure.
  • the three-phase Totem rectifier is taken as an example to give the topology of the power conversion unit, and the topology of the power conversion unit of other three-phase rectifiers is also within the protection scope of the present disclosure.
  • the power conversion unit of the three-phase Totem rectifier includes: three single-phase rectifier circuits, and the three single-phase rectifier circuits are connected in a star configuration.
  • Each single-phase rectifier circuit includes: an inductor, two pairs of switch tubes and a BUS capacitor.
  • the first-phase rectifier circuit includes: a first inductor L1, a first switch transistor Q11, a second switch transistor Q12, a third switch transistor Q13, a fourth switch transistor Q14, and a first BUS capacitor C1;
  • the phase rectifier circuit includes: a second inductance L2, a fifth switch tube Q21, a sixth switch tube Q22, a seventh switch tube Q23, an eighth switch tube Q24, and a second BUS capacitor C2;
  • the third phase rectifier circuit includes: a third inductance L3, the ninth switch transistor Q31, the tenth switch transistor Q32, the eleventh switch transistor Q33, the twelfth switch transistor Q34, and the third BUS capacitor C3.
  • One end of the first inductor L1 is connected to the first phase of the input power supply, and the other end of the first inductor L1 is connected to the first pole of the first switch tube Q11 and the second pole of the third switch tube Q13 at the same time, and the first switch tube Q11
  • the second pole of the second switch tube Q12 is connected to the second pole of the second switch tube Q12
  • the first pole of the second switch tube Q12 is connected to the second pole of the fourth switch tube Q14
  • the first pole of the third switch tube Q13 is connected to the fourth switch tube
  • the first pole of the transistor Q14 is connected
  • one end of the first BUS capacitor C1 is connected to the second pole of the second switch transistor Q12
  • the other end of the first BUS capacitor C1 is connected to the first pole of the fourth switch transistor Q14.
  • One end of the second inductor L2 is connected to the second phase of the input power supply, and the other end of the second inductor L2 is connected to the first pole of the fifth switch tube Q21 and the second pole of the seventh switch tube Q23 at the same time, and the fifth switch tube Q21
  • the second pole of the sixth switch tube Q22 is connected to the second pole of the sixth switch tube Q22
  • the first pole of the sixth switch tube Q22 is connected to the second pole of the eighth switch tube Q24
  • the first pole of the seventh switch tube Q23 is connected to the eighth switch tube.
  • the first pole of the transistor Q24 is connected
  • one end of the second BUS capacitor C2 is connected to the second pole of the sixth switch transistor Q22
  • the other end of the second BUS capacitor C2 is connected to the first pole of the eighth switch transistor Q24.
  • One end of the third inductor L3 is connected to the third phase of the input power supply, and the other end of the third inductor L3 is connected to the first pole of the ninth switch Q31 and the second pole of the eleventh switch Q33 at the same time, and the ninth switch Q33
  • the second pole of Q31 is connected to the second pole of the tenth switch Q32
  • the first pole of the tenth switch Q32 is connected to the second pole of the twelfth switch Q24
  • the first pole of the eleventh switch Q33 is connected to
  • the first pole of the twelfth switch transistor Q34 is connected
  • one end of the third BUS capacitor C3 is connected to the second pole of the tenth switch transistor Q32
  • the other end of the third BUS capacitor C3 is connected to the first pole of the twelfth switch transistor Q34 connect.
  • the third pole of the eleventh switch transistor Q33 and the third pole of the twelfth switch transistor Q34 are connected to the driving unit 104 .
  • first switch tube Q11 and the third switch tube Q13 are a pair of upper and lower tubes
  • second switch tube Q12 and the fourth switch tube Q14 are a pair of upper and lower tubes
  • the two switch tubes of the same pair of upper and lower tubes are driven.
  • the signals are complementary, the driving signals of the first switch Q11 and the third switch Q13 are the same, and the driving signals of the second switch Q12 and the fourth switch Q14 are the same.
  • the fifth switch tube Q21 and the seventh switch tube Q23 are a pair of upper and lower tubes
  • the sixth switch tube Q22 and the eighth switch tube Q24 are a pair of upper and lower tubes
  • the driving signals of the two switch tubes of the same pair of upper and lower tubes are complementary.
  • the driving signals of the switch Q21 and the seventh switch Q23 are the same
  • the driving signals of the sixth switch Q22 and the eighth switch Q24 are the same.
  • the ninth switch tube Q31 and the eleventh switch tube Q33 are a pair of upper and lower tubes
  • the tenth switch tube Q32 and the twelfth switch tube Q34 are a pair of upper and lower tubes
  • the driving signals of the two switch tubes of the same pair of upper and lower tubes are complementary
  • the driving signals of the ninth switch Q31 and the eleventh switch Q33 are the same
  • the driving signals of the tenth switch Q32 and the twelfth switch Q34 are the same.
  • the first electrode is the source electrode S
  • the second electrode is the drain electrode D
  • the third electrode is the gate electrode G.
  • the above-mentioned switch transistor is a metal oxide semiconductor (MOS, Metal Oxide Semiconductor) transistor, which is a high-frequency switch transistor.
  • MOS Metal Oxide Semiconductor
  • the above switch tube can also be implemented by other tubes, which is not specifically used to limit the protection scope of the embodiments of the present disclosure.
  • FIG. 3 is a block diagram of a power conversion unit of a three-phase rectifier provided by the present disclosure. It should be noted that only the three-phase Vienna rectifier is used as an example to give the topology of the power conversion unit in FIG. 3 , and the topology of the power conversion unit of other three-phase rectifiers is also within the protection scope of the present disclosure.
  • the power conversion unit of the three-phase Vienna rectifier includes: three single-phase rectifier circuits, and the three single-phase rectifier circuits are connected in a star shape.
  • Each single-phase rectifier circuit includes: an inductor, a pair of switch tubes, and a pair of power frequency tubes; the output end of the power conversion unit is connected to a positive BUS capacitor and a negative BUS capacitor, and the positive BUS capacitor and the negative BUS capacitor are connected in series to form a total BUS Capacitance, as the power conversion result of the power conversion unit.
  • the first-phase rectifier circuit includes: a fourth inductor L4, a thirteenth switch tube Q41, a fourteenth switch tube Q42, a first power frequency tube VT1, a fourth power frequency tube VT4, and a fourth BUS capacitor C4;
  • the second-phase rectifier circuit includes: the fifth inductor L5, the fifteenth switch tube Q51, the sixteenth switch tube Q52, the second power frequency tube VT2, the fifth power frequency tube VT5, and the fifth BUS capacitor C5;
  • the third The phase rectification circuit includes: a sixth inductor L6, a seventeenth switch tube Q61, an eighteenth switch tube Q62, a third power frequency tube VT3, a sixth power frequency tube VT6, a fourth BUS capacitor C4, and a fifth BUS capacitor C5.
  • One end of the fourth inductor L4 is connected to the first phase of the input power supply, and the other end of the fourth inductor L4 is connected to the second pole of the thirteenth switch tube Q41, the first pole of the thirteenth switch tube Q41 and the fourteenth switch
  • the first pole of the transistor Q42 is connected, and the second pole of the fourteenth switch transistor Q42 is connected to the star connection point N.
  • One end of the fifth inductor L5 is connected to the second phase of the input power supply, and the other end of the fifth inductor L5 is connected to the second pole of the fifteenth switch tube Q51, the first pole of the fifteenth switch tube Q51 and the sixteenth switch The first pole of the transistor Q52 is connected, and the second pole of the sixteenth switch transistor Q52 is connected to the star connection point N.
  • One end of the sixth inductor L6 is connected to the third phase of the input power supply, and the other end of the sixth inductor L6 is connected to the second pole of the seventeenth switch Q61, the first pole of the seventeenth switch Q61 and the eighteenth switch The first pole of the transistor Q62 is connected, and the second pole of the eighteenth switch transistor Q62 is connected to the star connection point N.
  • the third pole of the thirteenth switch Q41, the third pole of the fourteenth switch Q42, the third pole of the fifteenth switch Q51, the third pole of the sixteenth switch Q52, and the seventeenth switch Q61 The third pole of the eighteenth switch transistor Q62 is connected to the driving unit 104 .
  • One end of the fourth BUS capacitor C4 is connected to the cathode of the first power frequency tube VT1, the cathode of the second power frequency tube VT2 and the cathode of the third power frequency tube VT3 at the same time, and the other end of the fourth BUS capacitor C4 is connected to the star at the same time.
  • the point N is connected to one end of the fifth BUS capacitor C5, and the other end of the fifth BUS capacitor C5 is simultaneously connected to the anode of the fourth power frequency tube VT4, the fifth power frequency tube VT5 and the sixth power frequency tube VT6.
  • the anode of the first power frequency tube VT1 is connected to the other end of the fourth inductor L4 and the cathode of the fourth power frequency tube VT4 at the same time, and the anode of the second power frequency tube VT2 is simultaneously connected to the other end of the fifth inductor L5 and the fifth power frequency tube.
  • the cathode of the tube VT5 is connected, and the anode of the third power frequency tube VT3 is connected to the other end of the sixth inductor L6 and the cathode of the sixth power frequency tube VT6 at the same time.
  • the first electrode is the source electrode S
  • the second electrode is the drain electrode D
  • the third electrode is the gate electrode G.
  • the above-mentioned switch transistor is a MOS transistor, which is a high-frequency switch transistor.
  • the above switch tube can also be implemented by other tubes, which is not specifically used to limit the protection scope of the present disclosure.
  • the above-mentioned power frequency transistors are power diodes or MOS transistors.
  • the above-mentioned power frequency tube can also be implemented by other tubes, which is not specifically intended to limit the actual protection scope of the present disclosure.
  • the control unit 103 includes: an A/D conversion subunit 401 , a feedforward subunit 402 , a voltage equalization loop subunit 403 , a voltage loop subunit 404 , and a current loop subunit 405.
  • Pulse Width Modulation (PWM, Pulse Width Modulation) drives the subunit 406.
  • the A/D conversion subunit 401 is configured to convert the three-phase phase voltage of the input power source, the current of the three-phase inductor and the three-phase BUS voltage sampled by the sampling unit 102 into digital signals; restore the digital signals of the three-phase phase voltages The digital signal of the current of the three-phase inductor is restored to the actual current, and the digital signal of the three-phase BUS voltage is restored to the actual three-phase BUS voltage.
  • the feedforward subunit 402 is configured to calculate the feedforward duty cycle D 0 of each phase according to the phase locking angle, the actual three-phase phase voltage and the target voltage (ie, the target value of the BUS voltage), the feedforward duty cycle D 0 is the major part of the final duty cycle.
  • the voltage equalizing loop subunit 403 is configured to compare the magnitude of the absolute value of the actual three-phase BUS voltage, and make the difference between the two actual BUS voltages with the largest absolute value of the actual three-phase BUS voltage, and the difference is The value is used as the error input of the voltage equalization loop, and the voltage equalization adjustment duty ratio D 1 is calculated through the voltage equalization loop (including but not limited to the P regulator, the PI regulator and the PID regulator). 1 is a trim portion of the final duty cycle.
  • the voltage equalization loop subunit 403 is configured to combine the actual positive BUS voltage (ie the voltage across the fourth BUS capacitor C4) and the actual negative BUS voltage (ie the voltage across the fifth BUS capacitor C5). pressure difference as the loop error input, by comparing the output of the equalizing regulator adjusts the duty cycle D 1, the equalizing a regulating portion regulating a duty ratio D of the duty cycle for the final.
  • the voltage loop subunit 404 is configured to output the given current loop through the voltage loop, and output the difference between the current loop and the actual three-phase inductor current as the input of the current loop.
  • the voltage loop subunit 404 is configured to calculate the average value of the three-phase BUS voltage, and use the difference between the average value of the three-phase BUS voltage and the target voltage as the error input of the voltage loop, through the voltage loop (including but Not limited to P regulator, PI regulator and PID regulator) the output current loop is given, and the difference between the current and the actual three-phase inductor current is output as the input of the current loop.
  • the voltage loop including but Not limited to P regulator, PI regulator and PID regulator
  • the current loop subunit 405 is configured to output the current through the current loop to adjust the duty ratio D 2 , where the current regulation duty ratio D 2 is another fine-tuning part of the final duty ratio;
  • the PWM drive sub-unit 406 is configured to output the drive signal of each switch tube to the drive unit 104 according to the feedforward duty cycle D 0 , the voltage equalization regulation duty cycle D 1 , and the current regulation duty cycle D 2 .
  • the feedforward duty cycle D 0 , the voltage equalization regulation duty cycle D 1 , and the current regulation duty cycle D 2 are added to obtain the final duty cycle.
  • DQ11 is the waveform of the driving signal of the first switch tube Q11 of the three-phase Totem rectifier
  • DQ21 is the waveform of the drive signal of the fifth switch tube Q21 of the three-phase Totem rectifier
  • DQ31 is the ninth switch tube of the three-phase Totem rectifier.
  • the waveform of the driving signal of Q31; Da is the waveform of the driving signal of the thirteenth switch tube Q41 of the three-phase Vienna rectifier, Db is the waveform of the driving signal of the fifteenth switch tube Q51 of the three-phase Vienna rectifier, and Dc is the three-phase Vienna rectifier.
  • phase A ie the first phase
  • phase B is in the negative half cycle
  • the same phase of the triangular carrier corresponding to the three phases causes the driving signals corresponding to the three phases to have no interleaving relationship, so that the input ripple current cannot be obtained. to effective control.
  • FIG. 7 is a flowchart of a ripple current control method provided by the present disclosure.
  • the present disclosure provides a ripple current control method, which may include: Step 700: If the phase voltage of a certain phase satisfies the phase-shift condition, determine a certain corresponding first carrier wave respectively.
  • a corresponding driving signal for each switch wherein, the phase of the first carrier is 180° out of phase with the phase of a corresponding second carrier, and the second carrier is when the phase voltage of a certain phase does not meet the phase-shift condition It is used to determine the carrier of the driving signal of each corresponding switch tube.
  • the method further includes: if the phase voltage of a certain phase does not satisfy the phase-shift condition, determining the driving of each corresponding switch tube according to a corresponding modulation wave and the second carrier wave Signal.
  • the phase voltage of a certain phase meeting the phase-shift condition includes: the sign of the phase voltage of a certain phase is negative; the phase voltage of a certain phase not meeting the phase-shift condition includes: the phase voltage of a certain phase is negative. The sign is positive.
  • the waveform of the three-phase phase voltage is shown in Figure 5
  • a certain phase such as phase A
  • the first A carrier determines the driving signal of the corresponding switch tube, as shown in Figure 8
  • the sign of the phase voltage of the phase is positive, that is, the phase voltage of the phase is in the positive half cycle
  • the second carrier is used to determine the corresponding The driving signal of the switch tube; wherein, the phase of the first carrier and the phase of the second carrier differ by 180°.
  • the driving signal of each switch tube if the phase voltage of phase B is in the negative half cycle, the driving signal of each switch tube corresponding to B is determined according to the first carrier corresponding to B; if the phase voltage of phase C is in the negative half cycle half cycle, the driving signal of each switch tube corresponding to C is respectively determined according to the first carrier corresponding to C.
  • phase A ie the first phase
  • phase B is in the negative half cycle
  • C-phase ie the third phase
  • the driving signals are interleaved, which increases the number of switching states of the three-phase rectifier or three-phase inverter, thereby reducing the ripple current, that is, by simply judging whether the sign of the phase voltage of a certain phase is negative, and A simple phase-shifting technique achieves control of the ripple current.
  • the phase voltage of a certain phase meeting the phase-shift condition includes: the sign of the phase voltage of a certain phase is positive; the phase voltage of a certain phase not satisfying the phase-shift condition includes: the phase voltage of a certain phase is positive. The sign is negative.
  • phase A For example, assuming that the waveform of the three-phase phase voltage is shown in Figure 5, for a certain phase (such as phase A), if the sign of the phase voltage of this phase is positive, that is, the phase voltage of this phase is in a positive half cycle, then the first A carrier determines the drive signal of the corresponding switch; if the sign of the phase voltage of the phase is negative, that is, the phase voltage of the phase is in the negative half cycle, then the second carrier is used to determine the drive signal of the corresponding switch ; wherein, the phase of the first carrier and the phase of the second carrier differ by 180°.
  • the driving signal of each switch tube corresponding to B if the phase voltage of phase B is in the positive half cycle, the driving signal of each switch tube corresponding to B is determined according to the first carrier corresponding to B; if the phase voltage of phase C is in positive half cycle, the driving signal of each switch tube corresponding to C is respectively determined according to the first carrier corresponding to C.
  • phase A ie the first phase
  • phase B is in the negative half cycle
  • the C-phase that is, the third phase
  • the driving signal of the switch tube of phase A is interleaved with the driving signal of the switch tube of phase B and phase C,
  • the number of switching states of the three-phase rectifier or three-phase inverter is increased, thereby reducing the ripple current, that is, by simply judging whether the sign of the phase voltage of a certain phase is negative, and simple phase shifting technology
  • the control of the ripple current is realized.
  • Enhanced Pulse Width Modulator (EPWM, Enhanced Pulse Width Modulator) cannot perform phase shifting, so the corresponding carrier corresponding to EPWM cannot perform phase shifting, then the above judgment Whether the phase voltage of a certain phase satisfies the phase-shifting condition determines whether the driving signal of the corresponding switch tube is determined according to the first carrier or the second carrier.
  • EPWM Enhanced Pulse Width Modulator
  • the phase voltage of a certain phase that satisfies the phase-shift condition includes: the sign of the phase voltage of a certain phase is opposite to the sign of the phase voltage of the main phase; the phase voltage of a certain phase does not meet the phase-shift condition, including: the sign of the phase voltage of a certain phase is the same as that of the main phase.
  • the signs of the phase-to-phase voltages are the same; among them, the main phase is the phase that cannot phase-shift the corresponding carrier.
  • the waveform of the three-phase phase voltage is shown in Figure 5, for a certain phase (such as A phase), if the corresponding EWPM, the corresponding carrier wave cannot be phase-shifted, then this phase can be set as the main phase, do not phase shift the corresponding carrier; determine whether the other two corresponding carriers need to be phase shifted by judging whether the sign of the phase voltage of the other two phases is the same as the sign of the phase voltage of this phase.
  • a certain phase such as A phase
  • the sign of the phase voltage of phase A is positive, and the sign of the phase voltage of phase B is negative; or, the sign of the phase voltage of phase A is negative, and the phase voltage of phase B is negative.
  • the sign of is positive; then the drive signal of each switch corresponding to B is determined according to the first carrier corresponding to B; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase B is in negative half cycle; or, the phase voltage of phase A is in the negative half cycle, and the phase voltage of phase B is in the positive half cycle; then the drive signal of each switch tube corresponding to B is determined respectively according to the first carrier corresponding to B.
  • the second carrier of respectively determines the driving signal of each switch tube corresponding to B; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase B is in the positive half cycle; or, if the phase voltage of phase A is in the positive half cycle The negative half cycle, and the phase voltage of the B phase is in the negative half cycle; then the driving signal of each switch tube corresponding to B is determined according to the second carrier corresponding to B.
  • the first carrier of respectively determines the driving signal of each switch tube corresponding to C; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase C is in the negative half cycle; or, if the phase voltage of phase A is in the positive half cycle The negative half cycle, and the phase voltage of the C phase is in the positive half cycle; then the drive signal of each switch tube corresponding to C is determined respectively according to the first carrier corresponding to C.
  • the second carrier of respectively determines the driving signal of each switch tube corresponding to C; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase C is in the positive half cycle; or, if the phase voltage of phase A is in the positive half cycle Negative half cycle, and the phase voltage of C phase is in the negative half cycle; then the driving signal of each switch tube corresponding to C is determined respectively according to the second carrier corresponding to C.
  • the relationship between the three-phase voltages is Ua ⁇ 0 ⁇ Uc ⁇ Ub, as shown in the interval corresponding to the dotted line in Figure 5.
  • the phase voltage of phase A ie the first phase
  • the voltage of phase B is in the negative half cycle
  • the phase voltage (ie the second phase) and the phase voltage of the C phase are in the positive half cycle.
  • the driving signals of the switching tubes are interleaved, which increases the number of switching states of the three-phase rectifier or three-phase inverter, thereby reducing the ripple current. Whether the signs of the phase voltages are the same, and the simple phase-shifting technique realizes the control of the ripple current.
  • Enhanced Pulse Width Modulator (EPWM, Enhanced Pulse Width Modulator) cannot perform phase shifting, so the corresponding carrier corresponding to EPWM cannot perform phase shifting, then the above judgment Whether the phase voltage of a certain phase satisfies the phase-shifting condition determines whether the driving signal of the corresponding switch tube is determined according to the first carrier or the second carrier.
  • EPWM Enhanced Pulse Width Modulator
  • the phase voltage of a certain phase that satisfies the phase-shifting condition includes: the sign of the phase voltage of a certain phase is the same as the sign of the phase voltage of the main phase; the phase voltage of a certain phase does not meet the phase-shifting condition, including: the sign of the phase voltage of a certain phase is the same as that of the main phase voltage.
  • the signs of the phase voltages of the phases are opposite; among them, the main phase is the phase that cannot phase-shift the corresponding carrier.
  • the waveform of the three-phase phase voltage is shown in Figure 5, for a certain phase (such as A phase), if the corresponding EWPM, the corresponding carrier wave cannot be phase-shifted, then this phase can be set as the main phase, do not phase shift the corresponding carrier; determine whether the other two corresponding carriers need to be phase shifted by judging whether the sign of the phase voltage of the other two phases is the same as the sign of the phase voltage of this phase.
  • a certain phase such as A phase
  • the driving signal of each switch tube corresponding to B is determined according to the first carrier corresponding to B; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase B is in the positive half cycle; or, if the phase A phase voltage is in the positive half cycle; The phase voltage of phase B is in the negative half cycle, and the phase voltage of phase B is in the negative half cycle; then the driving signal of each switch tube corresponding to B is determined according to the first carrier corresponding to B;
  • the second carrier of respectively determines the driving signal of each switch tube corresponding to B; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase B is in the negative half cycle; or, if the phase voltage of phase A is in the positive half cycle The negative half cycle, and the phase voltage of the B phase is in the positive half cycle; then, the driving signal of each switch tube corresponding to B is determined respectively according to the second carrier corresponding to B.
  • the first carrier wave respectively determines the driving signal of each switch tube corresponding to C; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase C is in the positive half cycle; or, if the phase voltage of phase A is in the positive half cycle Negative half cycle, and the phase voltage of C phase is in the negative half cycle; then the drive signal of each switch tube corresponding to C is determined respectively according to the first carrier corresponding to C.
  • the second carrier of respectively determines the driving signal of each switch tube corresponding to C; that is, if the phase voltage of phase A is in the positive half cycle, and the phase voltage of phase C is in the negative half cycle; or, if the phase voltage of phase A is in the positive half cycle The negative half cycle, and the phase voltage of the C phase is in the positive half cycle; then the driving signal of each switch tube corresponding to C is determined respectively according to the second carrier corresponding to C.
  • the relationship between the three-phase voltages is 0 ⁇ Ua ⁇ Uc ⁇ Ub.
  • the phase voltage of phase A ie the first phase
  • the phase voltage of phase B ie the second phase
  • the phase C ie the first phase
  • the three-phase) phase voltages are all in the negative half cycle.
  • the drive signal of the switch tube of phase A is interleaved with the drive signals of the switch tubes of the other two phases, which improves the switching of the three-phase rectifier or the three-phase inverter.
  • the number of states thereby reducing the ripple current, that is, by simply judging whether the sign of the phase voltage of a certain phase is the same as the sign of the phase voltage of the main phase, and the simple phase shifting technique realizes the ripple current. control.
  • the corresponding second carrier when the phase voltage of a certain phase satisfies the phase-shift condition, the corresponding second carrier can be subjected to 180° phase-shift processing; when the phase voltage of a certain phase does not meet the phase-shift condition When the phase condition is met, the phase shift processing is not performed on the corresponding second carrier.
  • the method before judging whether the phase voltage of a certain phase satisfies the phase-shift condition, the method further includes: sampling the three-phase phase voltage, the current of the three-phase inductor and the three-phase bus voltage; The voltage, the current of the three-phase inductor, and the three-phase bus voltage calculate the modulation wave corresponding to the three-phase.
  • determining the drive signal of each corresponding switch tube according to a corresponding first carrier wave includes: determining each corresponding switch tube according to a corresponding modulation wave and the first carrier wave, respectively. tube drive signal.
  • determining the driving signal of each corresponding switch tube according to a certain corresponding second carrier wave includes: determining each corresponding switch tube according to a certain corresponding modulating wave and the second carrier wave respectively. tube drive signal.
  • how to calculate the modulated wave corresponding to the three phases according to the three-phase phase voltages, the currents of the three-phase inductors, and the three-phase bus voltages is the same as the functional description of the control unit in the foregoing embodiments, and will not be repeated here.
  • DQ11 is the waveform of the driving signal of the first switching transistor Q11 of the three-phase Totem rectifier
  • DQ21 is the driving signal of the fifth switching transistor Q21 of the three-phase Totem rectifier
  • DQ31 is the waveform of the drive signal of the ninth switch Q31 of the three-phase Totem rectifier
  • Da is the waveform of the drive signal of the thirteenth switch Q41 of the three-phase Vienna rectifier
  • Db is the fifteenth of the three-phase Vienna rectifier.
  • the waveform of the driving signal of the switch tube Q51, Dc is the waveform of the driving signal of the seventeenth switch tube Q61 of the three-phase Vienna rectifier.
  • the method further includes: driving the corresponding switch tube according to the driving signal of each switch tube respectively. On-off of the switch tube.
  • the method further includes: respectively driving the corresponding switch tube on and off according to the driving signal of each switch tube.
  • the ripple current control method when the phase voltage of a certain phase satisfies the phase-shift condition, and when the phase voltage of the phase does not satisfy the phase-shift condition, it is used to determine the driving signal of each corresponding switch tube.
  • the phase of the second carrier is shifted by 180°, and the first carrier is obtained. Based on the first carrier, the driving signal of each corresponding switch tube is determined respectively.
  • the The second carrier wave determines the drive signal of each switch tube corresponding to the other two phases, so that the drive signal of the switch tube of a certain phase is interleaved with the drive signals of the switch tubes of the other two phases, which improves the three-phase rectifier or three-phase inverter.
  • the number of switching states of the device is reduced, thereby reducing the ripple current. That is to say, the control of the ripple current is realized by simply judging whether the phase voltage of a certain phase satisfies the phase-shifting condition, and simple phase-shifting technology.
  • embodiments of the present disclosure provide an electronic device, including: at least one processor; and a memory, where at least one program is stored, and when the at least one program is executed by the at least one processor, the at least one processor implements any of the above A ripple current control method.
  • the processor is a device with data processing capability, which includes but is not limited to a central processing unit (CPU), etc.
  • the memory is a device with data storage capability, which includes but is not limited to random access memory (RAM, more specifically such as SDRAM) , DDR, etc.), read-only memory (ROM), charged erasable programmable read-only memory (EEPROM), flash memory (FLASH).
  • RAM random access memory
  • ROM read-only memory
  • EEPROM charged erasable programmable read-only memory
  • FLASH flash memory
  • the processor and memory are connected to each other through a bus, which in turn is connected to other components of the computing device.
  • embodiments of the present disclosure provide a computer-readable storage medium, where a computer program is stored thereon, and when the computer program is executed by a processor, any one of the above-mentioned ripple current control methods is implemented.
  • FIG. 10 is a block diagram of a ripple current control device according to an embodiment of the present disclosure.
  • an embodiment of the present disclosure provides a ripple current control device, including: a drive signal determination module 1001 configured to, if the phase voltage of a certain phase satisfies the phase-shift condition, according to a corresponding first A carrier determines the driving signal of each corresponding switch tube; wherein, the phase of the first carrier is 180° different from the phase of a corresponding second carrier, and the second carrier is the phase of a certain phase When the voltage does not meet the phase-shifting condition, it is used to determine the carrier of the driving signal of each corresponding switch tube.
  • the driving signal determination module 1001 is further configured to: if the phase voltage of a certain phase does not satisfy the phase-shift condition, determine the corresponding voltage of each switch tube according to a corresponding second carrier. drive signal.
  • the phase voltage of a certain phase meeting the phase-shift condition includes: the sign of the phase voltage of a certain phase is negative; the phase voltage of a certain phase not meeting the phase-shift condition includes: the phase voltage of a certain phase is negative. The sign is positive.
  • the phase voltage of a certain phase meeting the phase-shift condition includes: the sign of the phase voltage of a certain phase is positive; the phase voltage of a certain phase not satisfying the phase-shift condition includes: the phase voltage of a certain phase is positive. The sign is negative.
  • the phase voltage of a certain phase meeting the phase-shift condition includes: the sign of the phase voltage of a certain phase is opposite to the sign of the phase voltage of the main phase; the phase voltage of a certain phase does not satisfy the phase-shift condition includes: : The sign of the phase voltage of a certain phase is the same as the sign of the phase voltage of the main phase.
  • the phase voltage of a certain phase meeting the phase-shift condition includes: the sign of the phase voltage of a certain phase is the same as the sign of the phase voltage of the main phase; the phase voltage of a certain phase does not satisfy the phase-shift condition includes: : The sign of the phase voltage of a certain phase is opposite to the sign of the phase voltage of the main phase.
  • the apparatus further includes: a sampling module 1002, configured to sample the three-phase phase voltage, the current of the three-phase inductor, and the three-phase bus voltage; a modulated wave calculation module 1003, configured to The phase voltage, the current of the three-phase inductance and the three-phase bus voltage are used to calculate the modulation wave corresponding to the three phases; the driving signal determination module 1001 is further configured to: determine a corresponding modulation wave and the first carrier wave respectively according to a certain corresponding modulation wave. A drive signal for a switch.
  • a sampling module 1002 configured to sample the three-phase phase voltage, the current of the three-phase inductor, and the three-phase bus voltage
  • a modulated wave calculation module 1003 configured to The phase voltage, the current of the three-phase inductance and the three-phase bus voltage are used to calculate the modulation wave corresponding to the three phases
  • the driving signal determination module 1001 is further configured to: determine a corresponding modulation wave and the first carrier wave respectively according to a certain corresponding modulation wave
  • the apparatus further includes: a driving module 1004 configured to drive the corresponding switch transistors on and off according to the driving signal of each switch transistor.
  • the function of the sampling module 1002 can be implemented by the above-mentioned sampling unit 102
  • the function of the modulated wave calculation module 1003 can be implemented by the above-mentioned driving unit 104
  • the functions of the driving signal determination module 1001 and the driving module 1004 can be implemented by the above-mentioned driving unit unit 104 is implemented.
  • Such software may be distributed on computer-readable media, which may include computer storage media (or non-transitory media) and communication media (or transitory media).
  • computer storage media includes both volatile and nonvolatile implemented in any method or technology for storage of information, such as computer readable instructions, data structures, program modules or other data flexible, removable and non-removable media.
  • Computer storage media include, but are not limited to, RAM, ROM, EEPROM, flash memory or other memory technology, CD-ROM, digital versatile disk (DVD) or other optical disk storage, magnetic cartridges, magnetic tape, magnetic disk storage or other magnetic storage, or available with Any other medium that stores the desired information and can be accessed by a computer.
  • communication media typically embodies computer readable instructions, data structures, program modules, or other data in a modulated data signal such as a carrier wave or other transport mechanism, and can include any information delivery media, as is well known to those of ordinary skill in the art .

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Rectifiers (AREA)

Abstract

一种纹波电流控制方法和装置、电子设备、计算机可读存储介质，纹波电流控制方法包括：若某一相的相电压满足移相条件，根据某一相对应的第一载波分别确定某一相对应的每一个开关管的驱动信号；其中，第一载波的相位与某一相对应的第二载波的相位相差180°，第二载波为某一相的相电压不满足移相条件时用于确定某一相对应的每一个开关管的驱动信号的载波。

Description

纹波电流控制方法和装置、电子设备、计算机可读存储介质
相关公开的交叉引用
本公开要求2020年6月29提交给中国专利局的第202010616309.9号专利公开的优先权，其全部内容通过引用合并于此。
技术领域
本公开涉及但不限于开关电源变换技术领域。
背景技术
三相整流器由于硬件参数的选择，或者负载的差异、输入源的差异，可能会导致输入的纹波电流超出标准范围的情况，输入的纹波电流过大，会增大整流器的损耗、降低效率，影响输入输出指标。因此，需要对输入的纹波电流进行控制。相关的减小纹波电流的方法要么增加了器件的成本，要么计算繁琐、控制复杂。
发明内容
第一方面，本公开提供一种纹波电流控制方法，包括：若某一相的相电压满足移相条件，根据某一相对应的第一载波分别确定某一相对应的每一个开关管的驱动信号；其中，第一载波的相位与某一相对应的第二载波的相位相差180°，第二载波为某一相的相电压不满足移相条件时用于确定某一相对应的每一个开关管的驱动信号的载波。
第二方面，本公开提供一种电子设备，包括：至少一个处理器；存储器，存储器上存储有至少一个程序，当所述至少一个程序被所述至少一个处理器执行，使得所述至少一个处理器实现本文所述任意一种纹波电流控制方法。
第三方面，本公开提供一种计算机可读存储介质，计算机可读存储介质上存储有计算机程序，所述计算机程序被处理器执行时实现 本文所述任意一种纹波电流控制方法。
附图说明
图1为本公开提供的三相整流器的组成框图；
图2为本公开提供的三相整流器的功率变换单元的组成框图；
图3为本公开提供的三相整流器的功率变换单元的组成框图；
图4为本公开提供的三相整流器的控制单元的组成框图；
图5为本公开提供的三相相电压的波形图；
图6为本公开提供的开关管的驱动信号的波形图；
图7为本公开提供的一种纹波电流控制方法的流程图；
图8为本公开提供的开关管的驱动信号的波形图；
图9为本公开提供的开关管的驱动信号的波形图；
图10为本公开提供的一种纹波电流控制装置的组成框图。
具体实施方式
为使本领域的技术人员更好地理解本公开的技术方案，下面结合附图对本公开提供的纹波电流控制方法和装置、电子设备、计算机可读存储介质进行详细描述。
在下文中将参考附图更充分地描述示例实施方式，但是所述示例实施方式可以以不同形式来体现且不应当被解释为限于本文阐述的实施方式。反之，提供这些实施方式的目的在于使本公开透彻和完整，并将使本领域技术人员充分理解本公开的范围。
在不冲突的情况下，本公开各实施方式及实施方式中的各特征可相互组合。
如本文所使用的，术语“和/或”包括至少一个相关列举条目的任何和所有组合。
本文所使用的术语仅用于描述特定实施方式，且不意欲限制本公开。如本文所使用的，单数形式“一个”和“该”也意欲包括复数形式，除非上下文另外清楚指出。还将理解的是，当本说明书中使用术语“包括”和/或“由……制成”时，指定存在所述特征、整体、 步骤、操作、元件和/或组件，但不排除存在或添加至少一个其它特征、整体、步骤、操作、元件、组件和/或其群组。
除非另外限定，否则本文所用的所有术语(包括技术和科学术语)的含义与本领域普通技术人员通常理解的含义相同。还将理解，诸如那些在常用字典中限定的那些术语应当被解释为具有与其在相关技术以及本公开的背景下的含义一致的含义，且将不解释为具有理想化或过度形式上的含义，除非本文明确如此限定。
随着需求的不断提升和技术的不断进步，三相整流器产品的竞争优势主要体现在功率密度的提升、效率的提升以及成本的降低三大方面上。三相功率因数校正(PFC，Power Factor Correction)整流器主要包括无源PFC整流器(即被动式PFC整流器)和有源PFC整流器(即主动式PFC整流器)。无源PFC整流器有源具有输入电流的谐波含量高、输出电压受制于输入电压等缺点，无法跟上当前整流器市场的发展趋势。因此，当前整流器产品中有源PFC整流器的应用较为广泛。
三相有源PFC整流器将输入的交流电压整流成直流电压输出的同时，保持三相输入的电流与电压相位同步且无畸变。目前所应用的三相PFC整流器的拓扑类型多样。传统的不控整流或半控整流装置由于其自身的不完全可控，其性能已经远远难以满足当前的需要，传统的两电平整流器拓扑也与用户对高压、大功率，高性能的要求相距甚远。与其他拓扑相比，维也纳(Vienna)整流器具有结构简单、开关管器件应力小、无直通危险和谐波含量少等优点，在对输出功率要求大、功率密度要求高、功率因数改善要求严格的场合应用广泛；图腾(Totem)式整流器具有使用器件少、传导损耗低、效率高等优点，在三相整流器中的应用也引起了越来越多的关注。
然而无论是三相Vienna整流器、三相Totem整流器还是其他三相整流器，都无法避免对输入的纹波电流进行优化的相关问题。由于硬件参数的选择，或者负载的差异、输入源的差异，可能会导致输入的纹波电流超出标准范围的情况。输入的纹波电流过大，会增大整流器的损耗、降低效率，影响输入输出指标。因此，需要对输入的纹波 电流进行控制。
相关的减小输入的纹波电流的方法分为硬件方法和软件方法。
其中，常见的用于减小输入的纹波电流的硬件方法包括以下两种方法：方法一、电感耦合，该方法需增加大量的电感、开关管等元器件，会增加器件成本；方法二、采用交错并联技术，该方法将每一相分为并联的两个支路，再分别加以交错的驱动，在增加器件成本的同时，增加了控制复杂度。
其中，软件方法目前比较前沿的是在软件中应用空间矢量控制技术，通过复杂的数字逻辑计算，利用信号处理器进行控制。该方法计算繁琐、控制复杂，并且，对数字信号处理(DSP，Digital Signal Processing)控制器的要求更高，也会提升成本。
需要说明的是，本公开提供的纹波电流控制方法和装置、电子设备、计算机可读存储介质可以实现对三相整流器和三相逆变器的纹波电流的控制。以下以三相整流器为例进行说明，三相逆变器的实现过程类似。
图1为本公开提供的三相整流器的组成框图。如图1所示，三相整流器包括功率变换单元101、采样单元102、控制单元103和驱动单元104。
在一些示例性实施方式中，功率变换单元101，配置为将输入电源的交流电压转换成直流电压；采样单元102，配置为对输入电源的三相相电压、三相电感的电流和三相总线(BUS)电压(即BUS电容两端的电压)进行采样；控制单元103，配置为根据采样单元102采样的输入电源的三相相电压、三相电感的电流和三相BUS电压计算每一个开关管的驱动信号并输出给驱动单元104；驱动单元104，配置为根据驱动信号驱动功率变换单元101的开关管的通断，从而控制输出的BUS电压。
在一些示例性实施方式中，采样单元102和控制单元103采用同一个DSP芯片实现。
图2为本公开提供的三相整流器的功率变换单元的组成框图。图2中仅以三相Totem整流器为例给出了功率变换单元的拓扑，其 他三相整流器的功率变换单元的拓扑也在本公开的保护范围内。
如图2所示，三相Totem整流器的功率变换单元包括：三个单相整流电路，三个单相整流电路星型连接。每一个单相整流电路包括：一个电感、两对开关管和一个BUS电容。
如图2所示，第一相整流电路包括：第一电感L1、第一开关管Q11、第二开关管Q12、第三开关管Q13、第四开关管Q14、第一BUS电容C1；第二相整流电路包括：第二电感L2、第五开关管Q21、第六开关管Q22、第七开关管Q23、第八开关管Q24、第二BUS电容C2；第三相整流电路包括：第三电感L3、第九开关管Q31、第十开关管Q32、第十一开关管Q33、第十二开关管Q34、第三BUS电容C3。
第一电感L1的一端与输入电源的第一相相连，第一电感L1的另一端同时与第一开关管Q11的第一极和第三开关管Q13的第二极相连，第一开关管Q11的第二极和第二开关管Q12的第二极相连，第二开关管Q12的第一极和第四开关管Q14的第二极连接，第三开关管Q13的第一极和第四开关管Q14的第一极相连，第一BUS电容C1的一端与第二开关管Q12的第二极连接，第一BUS电容C1的另一端与第四开关管Q14的第一极连接。
第二电感L2的一端与输入电源的第二相相连，第二电感L2的另一端同时与第五开关管Q21的第一极和第七开关管Q23的第二极相连，第五开关管Q21的第二极和第六开关管Q22的第二极相连，第六开关管Q22的第一极和第八开关管Q24的第二极连接，第七开关管Q23的第一极和第八开关管Q24的第一极相连，第二BUS电容C2的一端与第六开关管Q22的第二极连接，第二BUS电容C2的另一端与第八开关管Q24的第一极连接。
第三电感L3的一端与输入电源的第三相相连，第三电感L3的另一端同时与第九开关管Q31的第一极和第十一开关管Q33的第二极相连，第九开关管Q31的第二极和第十开关管Q32的第二极相连，第十开关管Q32的第一极和第十二开关管Q24的第二极连接，第十一开关管Q33的第一极和第十二开关管Q34的第一极相连，第三BUS 电容C3的一端与第十开关管Q32的第二极连接，第三BUS电容C3的另一端与第十二开关管Q34的第一极连接。
第二开关管Q12的第一极、第四开关管Q14的第二极、第六开关管Q22的第一极、第八开关管Q24的第二极、第十开关管Q32的第一极、第十二开关管Q34的第二极同时与星型连接点连接。
第一开关管Q11的第三极、第二开关管Q12的第三极、第三开关管Q13的第三极、第四开关管Q14的第三极、第五开关管Q21的第三极、第六开关管Q22的第三极、第七开关管Q23的第三极、第八开关管Q24的第三极、第九开关管Q31的第三极、第十开关管Q32的第三极、第十一开关管Q33的第三极、第十二开关管Q34的第三极与驱动单元104连接。
需要说明的是，第一开关管Q11和第三开关管Q13为一对上下管，第二开关管Q12和第四开关管Q14为一对上下管，同一对上下管的两个开关管的驱动信号互补，第一开关管Q11和第三开关管Q13的驱动信号相同，第二开关管Q12和第四开关管Q14的驱动信号相同。
第五开关管Q21和第七开关管Q23为一对上下管，第六开关管Q22和第八开关管Q24为一对上下管，同一对上下管的两个开关管的驱动信号互补，第五开关管Q21和第七开关管Q23的驱动信号相同，第六开关管Q22和第八开关管Q24的驱动信号相同。
第九开关管Q31和第十一开关管Q33为一对上下管，第十开关管Q32和第十二开关管Q34为一对上下管，同一对上下管的两个开关管的驱动信号互补，第九开关管Q31和第十一开关管Q33的驱动信号相同，第十开关管Q32和第十二开关管Q34的驱动信号相同。
在一些示例性实施方式中，第一极为源极S，第二极为漏极D，第三极为栅极G。
在一些示例性实施方式中，上述开关管为金属氧化物半导体(MOS，Metal Oxide Semiconductor)管，为高频开关管。当然，上述开关管也可以采用其他的管实现，具体不用于限定本公开实施方式的保护范围。
图3为本公开提供的三相整流器的功率变换单元的组成框图。需要说明的是，图3中仅以三相Vienna整流器为例给出了功率变换单元的拓扑，其他三相整流器的功率变换单元的拓扑也在本公开的保护范围内。
如图3所示，三相Vienna整流器的功率变换单元包括：三个单相整流电路，三个单相整流电路星型连接。每一个单相整流电路包括：一个电感、一对开关管、一对工频管；功率变换单元的输出端连接一个正BUS电容和一个负BUS电容，正BUS电容和负BUS电容串联构成总BUS电容，作为功率变换单元的功率变换结果。
如图3所示，第一相整流电路包括：第四电感L4、第十三开关管Q41、第十四开关管Q42、第一工频管VT1、第四工频管VT4、第四BUS电容C4；第二相整流电路包括：第五电感L5、第十五开关管Q51、第十六开关管Q52、第二工频管VT2、第五工频管VT5、第五BUS电容C5；第三相整流电路包括：第六电感L6、第十七开关管Q61、第十八开关管Q62、第三工频管VT3、第六工频管VT6、第四BUS电容C4、第五BUS电容C5。
第四电感L4的一端与输入电源的第一相相连，第四电感L4的另一端与第十三开关管Q41的第二极相连，第十三开关管Q41的第一极和第十四开关管Q42的第一极相连，第十四开关管Q42的第二极和星型连接点N连接。
第五电感L5的一端与输入电源的第二相相连，第五电感L5的另一端与第十五开关管Q51的第二极相连，第十五开关管Q51的第一极和第十六开关管Q52的第一极相连，第十六开关管Q52的第二极和星型连接点N连接。
第六电感L6的一端与输入电源的第三相相连，第六电感L6的另一端与第十七开关管Q61的第二极相连，第十七开关管Q61的第一极和第十八开关管Q62的第一极相连，第十八开关管Q62的第二极和星型连接点N连接。
第十三开关管Q41的第三极、第十四开关管Q42的第三极、第十五开关管Q51的第三极、第十六开关管Q52的第三极、第十七开 关管Q61的第三极、第十八开关管Q62的第三极与驱动单元104连接。
第四BUS电容C4的一端同时与第一工频管VT1的阴极、第二工频管VT2的阴极和第三工频管VT3的阴极连接，第四BUS电容C4的另一端同时与星型连接点N和第五BUS电容C5的一端连接，第五BUS电容C5的另一端同时与第四工频管VT4的阳极、第五工频管VT5的阳极和第六工频管VT6的阳极连接。
第一工频管VT1的阳极同时与第四电感L4的另一端和第四工频管VT4的阴极连接，第二工频管VT2的阳极同时与第五电感L5的另一端和第五工频管VT5的阴极连接，第三工频管VT3的阳极同时与第六电感L6的另一端和第六工频管VT6的阴极连接。
需要说明的是，同一相的两个开关管的驱动信号相同。
在一些示例性实施方式中，第一极为源极S，第二极为漏极D，第三极为栅极G。
在一些示例性实施方式中，上述开关管为MOS管，为高频开关管。当然，上述开关管也可以采用其他的管实现，具体不用于限定本公开的保护范围。
在一些示例性实施方式中，上述工频管为功率二极管或MOS管。当然，上述工频管也可以采用其他的管实现，具体不用于限定本公开实的保护范围。
在一些示例性实施方式中，如图4所示，控制单元103包括：A/D转换子单元401、前馈子单元402、均压环路子单元403、电压环子单元404、电流环子单元405、脉冲宽度调制(PWM，Pulse Width Modulation)驱动子单元406。
其中，A/D转换子单元401，配置为将采样单元102采样的输入电源的三相相电压、三相电感的电流和三相BUS电压转换为数字信号；将三相相电压的数字信号还原成实际的三相相电压，将三相电感的电流的数字信号还原成实际的电流，将三相BUS电压的数字信号还原成实际的三相BUS电压。
前馈子单元402，配置为根据锁相角、实际的三相相电压和目标 电压(即BUS电压的目标值)计算每一相的前馈占空比D 0，该前馈占空比D 0为最终占空比的主要部分。
对于三相Totem整流器，均压环路子单元403，配置为比较实际的三相BUS电压的绝对值的大小，将实际的三相BUS电压的绝对值最大的两个实际的BUS电压做差，差值作为均压环路的误差输入，通过均压环路(包括但不限于P调节器、PI调节器以及PID调节器)计算均压调节占空比D 1，该均压调节占空比D 1为最终占空比的一个微调部分。
对于三相Vienna整流器，均压环路子单元403，配置为将实际的正BUS电压(即第四BUS电容C4两端的电压)和实际的负BUS电压(即第五BUS电容C5两端的电压)的差值作为均压环路的误差输入，通过比较调节器输出均压调节占空比D 1，该均压调节占空比D 1为最终占空比的一个微调部分。
对于三相Totem整流器，电压环子单元404，配置为通过电压环输出电流环的给定，与实际的三相电感的电流之差输出作为电流环的输入。
对于三相Vienna整流器，电压环子单元404，配置为计算三相BUS电压的平均值，将三相BUS电压的平均值与目标电压的差值作为电压环的误差输入，通过电压环(包括但不限于P调节器、PI调节器以及PID调节器)输出电流环的给定，与实际的三相电感的电流之差输出作为电流环的输入。
电流环子单元405，配置为通过电流环输出电流调节占空比D 2，该电流调节占空比D 2为最终占空比的另一个微调部分；
PWM驱动子单元406，配置为根据前馈占空比D 0、均压调节占空比D 1、电流调节占空比D 2输出每一个开关管的驱动信号给驱动单元104。示例性地，假设三相相电压的波形如图5所示，将前馈占空比D 0、均压调节占空比D 1、电流调节占空比D 2相加，得到最终占空比；根据最终占空比计算每一相相电压的比较值，即调制波Va、Vb、Vc，如图6所示，将每一相的调制波与该相对应的三角载波进行比较得到该相的开关管的驱动信号的波形。图6中DQ11为三相 Totem整流器的第一开关管Q11的驱动信号的波形，DQ21为三相Totem整流器的第五开关管Q21的驱动信号的波形，DQ31为三相Totem整流器的第九开关管Q31的驱动信号的波形；Da为三相Vienna整流器的第十三开关管Q41的驱动信号的波形，Db为三相Vienna整流器的第十五开关管Q51的驱动信号的波形，Dc为三相Vienna整流器的第十七开关管Q61的驱动信号的波形。
假设某一时刻，三相相电压的关系为Ua<0<Uc<Ub，如图5中的虚线对应的区间所示，此时刻，A相(即第一相)电压处于负半周，B相(即第二相)电压和C相(即第三相)电压处于正半周，三相对应的三角载波的同相位导致三相对应的驱动信号没有交错关系，从而使得输入的纹波电流得不到有效控制。
图7为本公开提供的一种纹波电流控制方法的流程图。
第一方面，参照图7，本公开提供了一种纹波电流控制方法，可包括：步骤700、若某一相的相电压满足移相条件，根据某一相对应的第一载波分别确定某一相对应的每一个开关管的驱动信号；其中，第一载波的相位与某一相对应的第二载波的相位相差180°，第二载波为某一相的相电压不满足移相条件时用于确定某一相对应的每一个开关管的驱动信号的载波。
在一些示例性实施方式中，该方法还包括：若某一相的相电压不满足移相条件，根据某一相对应的调制波和第二载波确定某一相对应的每一个开关管的驱动信号。
在一些示例性实施方式中，某一相的相电压满足移相条件包括：某一相的相电压的符号为负；某一相的相电压不满足移相条件包括：某一相的相电压的符号为正。例如，假设三相相电压的波形如图5所示，对于某一相(例如A相)，如果该相的相电压的符号为负，也就是该相的相电压处于负半周，那么采用第一载波确定该相对应的开关管的驱动信号，如图8所示；如果该相的相电压的符号为正，也就是该相的相电压处于正半周，那么采用第二载波确定该相对应的开关管的驱动信号；其中，第一载波的相位和第二载波的相位相差180°。
需要说明的是，通过判断某一相的相电压是否满足移相条件来 决定根据第一载波还是第二载波确定该相对应的开关管的驱动信号的方式中，三相的相电压的判断和处理过程是完全独立的，不需要考虑其他两相的相电压是否处于负半周的，也就是说，如果A相的相电压处于负半周，则根据A相对应的第一载波分别确定A相对应的每一个开关管的驱动信号；如果B相的相电压处于负半周，则根据B相对应的第一载波分别确定B相对应的每一个开关管的驱动信号；如果C相的相电压处于负半周，则根据C相对应的第一载波分别确定C相对应的每一个开关管的驱动信号。
假设某一时刻，三相相电压的关系为Ua<0<Uc<Ub，如图5中的虚线对应的区间所示，此时刻，A相(即第一相)电压处于负半周，B相(即第二相)电压和C相(即第三相)电压处于正半周，采用本公开的方式后，如图8所示，某一相的开关管的驱动信号与其他两相的开关管的驱动信号交错，提升了三相整流器或三相逆变器的开关状态数，从而减小了纹波电流，也就是说，通过简单的判断某一相的相电压的符号是否为负，以及简单的移相技术实现了对纹波电流的控制。
在一些示例性实施方式中，某一相的相电压满足移相条件包括：某一相的相电压的符号为正；某一相的相电压不满足移相条件包括：某一相的相电压的符号为负。例如，假设三相相电压的波形如图5所示，对于某一相(例如A相)，如果该相的相电压的符号为正，也就是该相的相电压处于正半周，那么采用第一载波确定该相对应的开关管的驱动信号；如果该相的相电压的符号为负，也就是该相的相电压处于负半周，那么采用第二载波确定该相对应的开关管的驱动信号；其中，第一载波的相位和第二载波的相位相差180°。
需要说明的是，通过判断某一相的相电压是否满足移相条件来决定根据第一载波还是第二载波确定该相对应的开关管的驱动信号的方式中，三相的相电压的判断和处理过程是完全独立的，不需要考虑其他两相的相电压是否处于正半周的，也就是说，如果A相的相电压处于正半周，则根据A相对应的第一载波分别确定A相对应的每一个开关管的驱动信号；如果B相的相电压处于正半周，则根据B 相对应的第一载波分别确定B相对应的每一个开关管的驱动信号；如果C相的相电压处于正半周，则根据C相对应的第一载波分别确定C相对应的每一个开关管的驱动信号。
假设某一时刻，三相相电压的关系为Ua<0<Uc<Ub，如图5中的虚线对应的区间所示，此时刻，A相(即第一相)电压处于负半周，B相(即第二相)电压和C相(即第三相)电压处于正半周，采用本公开的方式后，A相的开关管的驱动信号与B相和C相的开关管的驱动信号交错，提升了三相整流器或三相逆变器的开关状态数，从而减小了纹波电流，也就是说，通过简单的判断某一相的相电压的符号是否为负，以及简单的移相技术实现了对纹波电流的控制。
在一些示例性实施方式中，存在以下情况：增强型脉冲宽度调制器(EPWM，Enhanced Pulse Width Modulator)不能进行移相，那么EPWM对应的相对应的载波也就不能进行移相，那么上述通过判断某一相的相电压是否满足移相条件来决定根据第一载波还是第二载波确定该相对应的开关管的驱动信号的方式则无法实现，本公开提出另一种实现方式，即某一相的相电压满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相反；某一相的相电压不满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相同；其中，主相为不能对对应的载波进行移相的相。例如，假设三相相电压的波形如图5所示，对于某一相(例如A相)，如果该相对应EWPM，则该相对应的载波不能进行移相，那么可以将该相设置为主相，不对该相对应的载波进行移相；通过判断其他两相的相电压的符号是否与该相的相电压的符号是否相同来决定是否需要对其他两相对应的载波进行移相。
示例性地，如图9所示，如果A相的相电压的符号为正，且B相的相电压的符号为负；或者，A相的相电压的符号为负，且B相的相电压的符号为正；那么根据B相对应的第一载波分别确定B相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且B相的相电压处于负半周；或者，A相的相电压处于负半周，且B相的相电压处于正半周；那么根据B相对应的第一载 波分别确定B相对应的每一个开关管的驱动信号。
如果A相的相电压的符号为正，且B相的相电压的符号为正；或者，A相的相电压的符号为负，且B相的相电压的符号为负；那么根据B相对应的第二载波分别确定B相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且B相的相电压处于正半周；或者，A相的相电压处于负半周，且B相的相电压处于负半周；那么根据B相对应的第二载波分别确定B相对应的每一个开关管的驱动信号。
如果A相的相电压的符号为正，且C相的相电压的符号为负；或者，A相的相电压的符号为负，且C相的相电压的符号为正；那么根据C相对应的第一载波分别确定C相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且C相的相电压处于负半周；或者，A相的相电压处于负半周，且C相的相电压处于正半周；那么根据C相对应的第一载波分别确定C相对应的每一个开关管的驱动信号。
如果A相的相电压的符号为正，且C相的相电压的符号为正；或者，A相的相电压的符号为负，且C相的相电压的符号为负；那么根据C相对应的第二载波分别确定C相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且C相的相电压处于正半周；或者，A相的相电压处于负半周，且C相的相电压处于负半周；那么根据C相对应的第二载波分别确定C相对应的每一个开关管的驱动信号。
假设某一时刻，三相相电压的关系为Ua<0<Uc<Ub，如图5中的虚线对应的区间所示，此时刻，A相(即第一相)相电压处于负半周，B相(即第二相)相电压和C相(即第三相)相电压处于正半周，采用本公开的方式后，如图9所示，A相的开关管的驱动信号与其他两相的开关管的驱动信号交错，提升了三相整流器或三相逆变器的开关状态数，从而减小了纹波电流，也就是说，通过简单的判断某一相的相电压的符号与主相的相电压的符号是否相同，以及简单的移相技术实现了对纹波电流的控制。
在一些示例性实施方式中，存在以下情况：增强型脉冲宽度调制器(EPWM，Enhanced Pulse Width Modulator)不能进行移相，那么EPWM对应的相对应的载波也就不能进行移相，那么上述通过判断某一相的相电压是否满足移相条件来决定根据第一载波还是第二载波确定该相对应的开关管的驱动信号的方式则无法实现，本公开提出另一种实现方式，即某一相的相电压满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相同；某一相的相电压不满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相反；其中，主相为不能对对应的载波进行移相的相。例如，假设三相相电压的波形如图5所示，对于某一相(例如A相)，如果该相对应EWPM，则该相对应的载波不能进行移相，那么可以将该相设置为主相，不对该相对应的载波进行移相；通过判断其他两相的相电压的符号是否与该相的相电压的符号是否相同来决定是否需要对其他两相对应的载波进行移相。
示例性地，如果A相的相电压的符号为正，且B相的相电压的符号为正；或者，A相的相电压的符号为负，且B相的相电压的符号为负；那么根据B相对应的第一载波分别确定B相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且B相的相电压处于正半周；或者，A相的相电压处于负半周，且B相的相电压处于负半周；那么根据B相对应的第一载波分别确定B相对应的每一个开关管的驱动信号；
如果A相的相电压的符号为正，且B相的相电压的符号为负；或者，A相的相电压的符号为负，且B相的相电压的符号为正；那么根据B相对应的第二载波分别确定B相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且B相的相电压处于负半周；或者，A相的相电压处于负半周，且B相的相电压处于正半周；那么根据B相对应的第二载波分别确定B相对应的每一个开关管的驱动信号。
如果A相的相电压的符号为正，且C相的相电压的符号为正；或者，A相的相电压的符号为负，且C相的相电压的符号为负；那 么根据C相对应的第一载波分别确定C相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且C相的相电压处于正半周；或者，A相的相电压处于负半周，且C相的相电压处于负半周；那么根据C相对应的第一载波分别确定C相对应的每一个开关管的驱动信号。
如果A相的相电压的符号为正，且C相的相电压的符号为负；或者，A相的相电压的符号为负，且C相的相电压的符号为正；那么根据C相对应的第二载波分别确定C相对应的每一个开关管的驱动信号；也就是说，如果A相的相电压处于正半周，且C相的相电压处于负半周；或者，A相的相电压处于负半周，且C相的相电压处于正半周；那么根据C相对应的第二载波分别确定C相对应的每一个开关管的驱动信号。
假设某一时刻，三相相电压的关系为0<Ua<Uc<Ub，此时刻，A相(即第一相)相电压、B相(即第二相)相电压和C相(即第三相)相电压均处于负半周，采用本公开的方式后，A相的开关管的驱动信号与其他两相的开关管的驱动信号交错，提升了三相整流器或三相逆变器的开关状态数，从而减小了纹波电流，也就是说，通过简单的判断某一相的相电压的符号与主相的相电压的符号是否相同，以及简单的移相技术实现了对纹波电流的控制。
需要说明的是，在具体实现时，可以在某一相的相电压满足移相条件时，对该相对应的第二载波进行180°的移相处理；在某一相的相电压不满足移相条件时，则不对该相对应的第二载波进行移相处理。
在一些示例性实施方式中，判断某一相的相电压是否满足移相条件之前，该方法还包括：对三相相电压、三相电感的电流和三相总线电压进行采样；根据三相相电压、三相电感的电流和三相总线电压计算三相对应的调制波。
相应的，根据某一相对应的第一载波分别确定某一相对应的每一个开关管的驱动信号包括：根据某一相对应的调制波和第一载波分别确定某一相对应的每一个开关管的驱动信号。
相应的，根据某一相对应的第二载波分别确定某一相对应的每一个开关管的驱动信号包括：根据某一相对应的调制波和第二载波分别确定某一相对应的每一个开关管的驱动信号。
在一些示例性实施方式中，如何根据三相相电压、三相电感的电流和三相总线电压计算三相对应的调制波与前述实施方式控制单元的功能描述相同，这里不再赘述。
在一些示例性实施方式中，如图8和图9所示，DQ11为三相Totem整流器的第一开关管Q11的驱动信号的波形，DQ21为三相Totem整流器的第五开关管Q21的驱动信号的波形，DQ31为三相Totem整流器的第九开关管Q31的驱动信号的波形；Da为三相Vienna整流器的第十三开关管Q41的驱动信号的波形，Db为三相Vienna整流器的第十五开关管Q51的驱动信号的波形，Dc为三相Vienna整流器的第十七开关管Q61的驱动信号的波形。
在一些示例性实施方式中，根据某一相对应的第一载波分别确定某一相对应的每一个开关管的驱动信号之后，该方法还包括：分别根据每一个开关管的驱动信号驱动对应的开关管的通断。
在一些示例性实施方式中，分别确定某一相对应的每一个开关管的驱动信号之后，该方法还包括：分别根据每一个开关管的驱动信号驱动对应的开关管的通断。
本公开提供的纹波电流控制方法，在某一相的相电压满足移相条件时，将该相的相电压不满足移相条件时用于确定该相对应的每一个开关管的驱动信号的第二载波的相位偏移180°，得到第一载波，基于第一载波分别确定该相对应的每一个开关管的驱动信号，若此时其他两相的相电压不满足移相条件，则采用第二载波分别确定其他两相对应的每一个开关管的驱动信号，使得某一相的开关管的驱动信号与其他两相的开关管的驱动信号交错，提升了三相整流器或三相逆变器的开关状态数，从而减小了纹波电流，也就是说，通过简单的判断某一相的相电压是否满足移相条件，以及简单的移相技术实现了对纹波电流的控制。
第二方面，本公开实施方式提供一种电子设备，包括：至少一 个处理器；存储器，存储器上存储有至少一个程序，当至少一个程序被至少一个处理器执行，使得至少一个处理器实现上述任意一种纹波电流控制方法。
其中，处理器为具有数据处理能力的器件，其包括但不限于中央处理器(CPU)等；存储器为具有数据存储能力的器件，其包括但不限于随机存取存储器(RAM，更具体如SDRAM、DDR等)、只读存储器(ROM)、带电可擦可编程只读存储器(EEPROM)、闪存(FLASH)。
在一些实施方式中，处理器、存储器通过总线相互连接，进而与计算设备的其它组件连接。
第三方面，本公开实施方式提供一种计算机可读存储介质，计算机可读存储介质上存储有计算机程序，所述计算机程序被处理器执行时实现上述任意一种纹波电流控制方法。
图10为本公开实施方式提供的一种纹波电流控制装置的组成框图。
第四方面，参照图10，本公开实施方式提供一种纹波电流控制装置，包括：驱动信号确定模块1001，配置为若某一相的相电压满足移相条件，根据某一相对应的第一载波分别确定所述某一相对应的每一个开关管的驱动信号；其中，第一载波的相位与某一相对应的第二载波的相位相差180°，第二载波为某一相的相电压不满足移相条件时用于确定某一相对应的每一个开关管的驱动信号的载波。
在一些示例性实施方式中，驱动信号确定模块1001还配置为：若某一相的相电压不满足移相条件，根据某一相对应的第二载波确定某一相对应的每一个开关管的驱动信号。
在一些示例性实施方式中，某一相的相电压满足移相条件包括：某一相的相电压的符号为负；某一相的相电压不满足移相条件包括：某一相的相电压的符号为正。
在一些示例性实施方式中，某一相的相电压满足移相条件包括：某一相的相电压的符号为正；某一相的相电压不满足移相条件包括：某一相的相电压的符号为负。
在一些示例性实施方式中，某一相的相电压满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相反；某一相的相电压不满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相同。
在一些示例性实施方式中，某一相的相电压满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相同；某一相的相电压不满足移相条件包括：某一相的相电压的符号与主相的相电压的符号相反。
在一些示例性实施方式中，所述装置还包括：采样模块1002，配置为对三相相电压、三相电感的电流和三相总线电压进行采样；调制波计算模块1003，配置为根据三相相电压、三相电感的电流和三相总线电压计算三相对应的调制波；驱动信号确定模块1001还配置为：根据某一相对应的调制波和第一载波分别确定某一相对应的每一个开关管的驱动信号。
在一些示例性实施方式中，所述装置还包括：驱动模块1004，配置为分别根据每一个开关管的驱动信号驱动对应的开关管的通断。
需要说明的是，采样模块1002的功能可以采用上述采样单元102来实现，调制波计算模块1003的功能可以采用上述驱动单元104来实现，驱动信号确定模块1001和驱动模块1004的功能可以采用上述驱动单元104来实现。
上述纹波电流控制装置的具体实现过程与上述纹波电流控制方法的具体实现过程相同，这里不再赘述。
本领域普通技术人员可以理解，上文中所公开方法中的全部或某些步骤、系统、装置中的功能模块/单元可以被实施为软件、固件、硬件及其适当的组合。在硬件实施方式中，在以上描述中提及的功能模块/单元之间的划分不一定对应于物理组件的划分；例如，一个物理组件可以具有多个功能，或者一个功能或步骤可以由若干物理组件合作执行。某些物理组件或所有物理组件可以被实施为由处理器，如中央处理器、数字信号处理器或微处理器执行的软件，或者被实施为硬件，或者被实施为集成电路，如专用集成电路。这样的软件可以分 布在计算机可读介质上，计算机可读介质可以包括计算机存储介质(或非暂时性介质)和通信介质(或暂时性介质)。如本领域普通技术人员公知的，术语计算机存储介质包括在用于存储信息(诸如计算机可读指令、数据结构、程序模块或其它数据)的任何方法或技术中实施的易失性和非易失性、可移除和不可移除介质。计算机存储介质包括但不限于RAM、ROM、EEPROM、闪存或其它存储器技术、CD-ROM、数字多功能盘(DVD)或其它光盘存储、磁盒、磁带、磁盘存储或其它磁存储器、或者可以用于存储期望的信息并且可以被计算机访问的任何其它的介质。此外，本领域普通技术人员公知的是，通信介质通常包含计算机可读指令、数据结构、程序模块或者诸如载波或其它传输机制之类的调制数据信号中的其它数据，并且可包括任何信息递送介质。
本文已经公开了示例实施方式，并且虽然采用了具体术语，但它们仅用于并仅应当被解释为一般说明性含义，并且不用于限制的目的。在一些实例中，对本领域技术人员显而易见的是，除非另外明确指出，否则可单独使用与特定实施方式相结合描述的特征、特性和/或元素，或可与其它实施方式相结合描述的特征、特性和/或元件组合使用。因此，本领域技术人员将理解，在不脱离由所附的权利要求阐明的本公开的范围的情况下，可进行各种形式和细节上的改变。

Claims (10)

  1. 一种纹波电流控制方法，包括：
    若某一相的相电压满足移相条件，根据所述某一相对应的第一载波分别确定所述某一相对应的每一个开关管的驱动信号；
    其中，所述第一载波的相位与所述某一相对应的第二载波的相位相差180°，所述第二载波为所述某一相的相电压不满足移相条件时用于确定所述某一相对应的每一个开关管的驱动信号的载波。
  2. 根据权利要求1所述的方法，还包括：
    若所述某一相的相电压不满足移相条件，根据所述某一相对应的第二载波确定所述某一相对应的每一个开关管的驱动信号。
  3. 根据权利要求1-2任一项所述的方法，其中，所述某一相的相电压满足移相条件包括：所述某一相的相电压的符号为负；所述某一相的相电压不满足移相条件包括：所述某一相的相电压的符号为正。
  4. 根据权利要求1-2任一项所述的方法，其中，所述某一相的相电压满足移相条件包括：所述某一相的相电压的符号为正；所述某一相的相电压不满足移相条件包括：所述某一相的相电压的符号为负。
  5. 根据权利要求1-2任一项所述的方法，其中，所述某一相的相电压满足移相条件包括：所述某一相的相电压的符号与主相的相电压的符号相反，所述某一相的相电压不满足移相条件包括：所述某一相的相电压的符号与所述主相的相电压的符号相同。
  6. 根据权利要求1-2任一项所述的方法，其中，所述某一相的相电压满足移相条件包括：所述某一相的相电压的符号与主相的相电压的符号相同；所述某一相的相电压不满足移相条件包括：所述某一相的相电压的符号与主相的相电压的符号相反。
  7. 根据权利要求1-2任一项所述的方法，判断某一相的相电压是否满足移相条件之前，还包括：对三相相电压、三相电感的电流和三相总线电压进行采样；根据所述三相相电压、所述三相电感的电流和所述三相总线电压计算三相对应的调制波；
    所述根据某一相对应的第一载波分别确定某一相对应的每一个 开关管的驱动信号包括：根据所述某一相对应的调制波和所述第一载波分别确定所述某一相对应的每一个开关管的驱动信号。
  8. 根据权利要求1-2任一项所述的方法，所述分别确定某一相对应的每一个开关管的驱动信号之后，该方法还包括：
    分别根据每一个所述开关管的驱动信号驱动对应的开关管的通断。
  9. 一种电子设备，包括：
    至少一个处理器；
    存储器，所述存储器上存储有至少一个程序，当所述至少一个程序被所述至少一个处理器执行，使得所述至少一个处理器实现根据权利要求1-8任意一项所述的纹波电流控制方法。
  10. 一种计算机可读存储介质，所述计算机可读存储介质上存储有计算机程序，所述计算机程序被处理器执行时实现根据权利要求1-8任意一项所述的纹波电流控制方法。
PCT/CN2021/102713 2020-06-29 2021-06-28 纹波电流控制方法和装置、电子设备、计算机可读存储介质 WO2022001943A1 (zh)

Priority Applications (2)

Application Number Priority Date Filing Date Title
EP21833948.9A EP4170885A4 (en) 2020-06-29 2021-06-28 CURRENT RIPPLE LIMITING METHOD AND APPARATUS, ELECTRONIC DEVICE AND COMPUTER-READABLE STORAGE MEDIUM
JP2022581584A JP7418624B2 (ja) 2020-06-29 2021-06-28 リップル電流制御方法および装置、電子機器、コンピュータ可読記憶媒体

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
CN202010616309.9A CN114094803B (zh) 2020-06-29 2020-06-29 纹波电流控制方法和装置、电子设备、计算机可读存储介质
CN202010616309.9 2020-06-29

Publications (1)

Publication Number Publication Date
WO2022001943A1 true WO2022001943A1 (zh) 2022-01-06

Family

ID=79317445

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/CN2021/102713 WO2022001943A1 (zh) 2020-06-29 2021-06-28 纹波电流控制方法和装置、电子设备、计算机可读存储介质

Country Status (4)

Country Link
EP (1) EP4170885A4 (zh)
JP (1) JP7418624B2 (zh)
CN (1) CN114094803B (zh)
WO (1) WO2022001943A1 (zh)

Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0600635A2 (en) * 1992-11-30 1994-06-08 Hitachi, Ltd. Parallel-connection multiple inverter system and control method therefor
CN101895222A (zh) * 2010-06-02 2010-11-24 黑龙江科技学院 基于反相交叉的多载波tpwm调制方法
CN104052320A (zh) * 2014-06-17 2014-09-17 华为技术有限公司 一种pwm调制方法及装置
CN106100430A (zh) * 2016-08-23 2016-11-09 合肥工业大学 三相五电平逆变器低共模电压调制的载波实现方法
JP2019216507A (ja) * 2018-06-11 2019-12-19 東芝三菱電機産業システム株式会社 多段変換器の制御装置
CN110663163A (zh) * 2017-06-15 2020-01-07 雷诺股份公司 用于控制三相维也纳式整流器的方法

Family Cites Families (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP5109354B2 (ja) * 2006-12-06 2012-12-26 株式会社豊田自動織機 モータインバータ装置及びその制御方法
AU2008364881B2 (en) 2008-12-01 2013-02-21 Mitsubishi Electric Corporation Alternating current-direct current converting apparatus and apparatus for driving electric machinery
US8659925B2 (en) 2010-10-22 2014-02-25 Hamilton Sundstrand Corporation Three-level active rectification pulse width modulation control
EP2600517A1 (en) * 2011-12-02 2013-06-05 Hamilton Sundstrand Corporation Three-level active rectification pulse width modulation control
US10075097B2 (en) 2013-11-28 2018-09-11 Mitsubishi Electric Corporation Power conversion device and AC electric-vehicle drive system
CN106385196A (zh) * 2016-09-27 2017-02-08 华中科技大学 一种基于电流纹波实时预测模型的三电平电压源变开关频率控制方法

Patent Citations (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0600635A2 (en) * 1992-11-30 1994-06-08 Hitachi, Ltd. Parallel-connection multiple inverter system and control method therefor
CN101895222A (zh) * 2010-06-02 2010-11-24 黑龙江科技学院 基于反相交叉的多载波tpwm调制方法
CN104052320A (zh) * 2014-06-17 2014-09-17 华为技术有限公司 一种pwm调制方法及装置
CN106100430A (zh) * 2016-08-23 2016-11-09 合肥工业大学 三相五电平逆变器低共模电压调制的载波实现方法
CN110663163A (zh) * 2017-06-15 2020-01-07 雷诺股份公司 用于控制三相维也纳式整流器的方法
JP2019216507A (ja) * 2018-06-11 2019-12-19 東芝三菱電機産業システム株式会社 多段変換器の制御装置

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
See also references of EP4170885A4 *

Also Published As

Publication number Publication date
CN114094803B (zh) 2024-03-12
EP4170885A1 (en) 2023-04-26
JP7418624B2 (ja) 2024-01-19
CN114094803A (zh) 2022-02-25
JP2023532139A (ja) 2023-07-26
EP4170885A4 (en) 2023-11-29

Similar Documents

Publication Publication Date Title
CN109495001B (zh) 模块化并联三电平Vienna整流器、控制系统及方法
WO2019120244A1 (zh) 谐振变换电路及其控制方法
CN108039822B (zh) 一种双有源全桥直流变换器的瞬时电流控制方法
US20220077769A1 (en) Power Factor Correction Circuit, Control Method and Electrical Appliance
CN107567680B (zh) 具有有源转换器的变速驱动器
WO2020082762A1 (zh) 三电平整流器共模电压抑制pwm方法、调制器及系统
CN112054694B (zh) 基于最小电流应力的双向变换器优化控制方法及装置
US20220029555A1 (en) Power conversion circuit, inverter, and control method
US20190280615A1 (en) Modulation method and apparatus based on three-phase neutral point clamped inverter
WO2023098217A1 (zh) 一种两电平型三相整流矫正器及其控制方法
CN110768536B (zh) 一种双有源桥电路损耗控制方法
CN106549591B (zh) 一种三电平t型逆变器死区消去及死区补偿联合方法
WO2022001943A1 (zh) 纹波电流控制方法和装置、电子设备、计算机可读存储介质
WO2022227972A1 (zh) 一种电压控制方法、装置、家电设备、计算机存储介质和程序
CN112910283B (zh) 模块化并联整流器的共模电压和环流同时抑制方法及系统
CN116131646A (zh) 多电平载波混叠pwm调制策略获取方法及电路控制方法
CN107947610B (zh) 应用于柔性直流输电系统的mmc模块拓扑结构及其调制方法
CN112054696A (zh) 基于最小回流功率的多电平变换器优化控制方法及装置
CN219833990U (zh) 一种逆变电源装置
WO2021110172A1 (zh) 功率变换器的bus电压均衡调节方法、功率变换器、存储介质及电子装置
CN112187083B (zh) 一种三电平风电变流器的功率损耗计算方法
CN113315379B (zh) 一种基于非对称调制的双有源桥式变换器混合控制方法
CN117856650A (zh) 光伏储能系统的功率控制与共模抑制方法及系统
CN206481233U (zh) 一种新型五电平逆变器
TW202420713A (zh) 飛跨電容多電平整流器及其控制方法

Legal Events

Date Code Title Description
121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 21833948

Country of ref document: EP

Kind code of ref document: A1

ENP Entry into the national phase

Ref document number: 2022581584

Country of ref document: JP

Kind code of ref document: A

ENP Entry into the national phase

Ref document number: 2021833948

Country of ref document: EP

Effective date: 20230118

NENP Non-entry into the national phase

Ref country code: DE