WO2020208789A1 - Dispositif d'entraînement de moteur, ventilateur électrique, aspirateur électrique et sèche-mains - Google Patents

Dispositif d'entraînement de moteur, ventilateur électrique, aspirateur électrique et sèche-mains Download PDF

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Publication number
WO2020208789A1
WO2020208789A1 PCT/JP2019/015855 JP2019015855W WO2020208789A1 WO 2020208789 A1 WO2020208789 A1 WO 2020208789A1 JP 2019015855 W JP2019015855 W JP 2019015855W WO 2020208789 A1 WO2020208789 A1 WO 2020208789A1
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Prior art keywords
phase
motor
voltage
unit
phase motor
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PCT/JP2019/015855
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English (en)
Japanese (ja)
Inventor
遥 松尾
裕次 ▲高▼山
和徳 畠山
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三菱電機株式会社
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Priority to CN201980094945.5A priority Critical patent/CN113647011A/zh
Priority to JP2021513124A priority patent/JP7170848B2/ja
Priority to PCT/JP2019/015855 priority patent/WO2020208789A1/fr
Publication of WO2020208789A1 publication Critical patent/WO2020208789A1/fr

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/04—Single phase motors, e.g. capacitor motors

Definitions

  • the present invention relates to a motor drive device for driving a single-phase motor, an electric blower, an electric vacuum cleaner, and a hand dryer.
  • the single-phase PM motor has a brushless structure that does not use a brush, which is a mechanical structure, as compared with a DC motor with a brush, so that brush wear does not occur. Due to this feature, the single-phase PM motor can ensure a long life and high reliability. Further, the single-phase PM motor is a highly efficient motor because a secondary current does not flow through the rotor as compared with an induction motor.
  • the single-phase PM motor has the following advantages as compared with the three-phase PM motor having different numbers of phases.
  • (1) In the case of a three-phase PM motor, a three-phase inverter is required, whereas in a single-phase PM motor, a single-phase inverter may be used.
  • (2) When a full-bridge inverter generally used as a three-phase inverter is used, six switching elements are required, whereas in the case of a single-phase PM motor, four switching elements are required even if a full-bridge inverter is used. Can be configured with.
  • (3) Due to the features of (1) and (2), the single-phase PM motor can be miniaturized as compared with the three-phase PM motor.
  • Patent Document 1 discloses a technique relating to a drive system for a single-phase PM motor.
  • Patent Document 1 switching control synchronized with the sensor cycle is realized based on the signal of the position sensor that detects the position of the rotor.
  • the position information of the rotor is indispensable for such control, and it is generally necessary to mount a position sensor using a Hall element inside the motor in order to acquire the position information.
  • mounting the position sensor is due to the increase in costs associated with the parts of the position sensor and the mounting process, design restrictions due to the integration of the board equipped with the position sensor and the motor, and the position deviation when the position sensor is mounted. There were problems such as the impact on control.
  • the present invention has been made in view of the above, and an object of the present invention is to obtain a motor driving device capable of estimating the position of a rotor from a physical quantity detected when driving a single-phase motor.
  • the motor drive device detects an inverter that outputs an AC voltage to a single-phase motor having a permanent magnet and a physical quantity that represents the operating state of the single-phase motor. It is provided with a detection unit for detecting a current including a frequency component of 2n + 1 times the fundamental wave of the AC voltage output by the inverter to the single-phase motor from the physical quantity detected by the detection unit. n is a natural number.
  • the motor drive device has an effect that the position of the rotor can be estimated from the physical quantity detected when driving the single-phase motor.
  • the figure which shows the example of the circuit configuration of the inverter shown in FIG. A block diagram showing a functional part that generates a PWM signal among the functional parts of the control unit shown in FIG.
  • a block diagram showing a configuration example of the carrier comparison unit shown in FIG. A time chart showing a waveform example of a main part in the carrier comparison part shown in FIG.
  • the figure which shows the other configuration example of the detection current signal processing part which concerns on Embodiment 1. A flowchart showing an operation of estimating the position of the rotor of the single-phase motor in the motor drive device according to the first embodiment.
  • FIG. 1 is a diagram showing a configuration example of a motor drive system 1 including a motor drive device 2 according to the first embodiment of the present invention.
  • the motor drive system 1 shown in FIG. 1 includes a single-phase motor 12, a motor drive device 2, a battery 10, a voltage sensor 20, and a switch 102.
  • the motor drive device 2 supplies AC power to the single-phase motor 12 to drive the single-phase motor 12.
  • the battery 10 is a DC power source that supplies DC power to the motor drive device 2.
  • the voltage sensor 20 is a detection unit that detects a physical quantity representing an operating state of the single-phase motor 12. Specifically, the voltage sensor 20 detects the DC voltage Vdc output from the battery 10 to the motor drive device 2.
  • the single-phase motor 12 is used as a rotary electric machine for rotating an electric blower (not shown).
  • the single-phase motor 12 and the electric blower are mounted on devices such as a vacuum cleaner and a hand dryer.
  • the voltage sensor 20 detects the DC voltage V dc , but the detection target of the voltage sensor 20 is not limited to the DC voltage V dc output from the battery 10.
  • the detection target of the voltage sensor 20 may be the inverter output voltage, which is the output voltage of the motor drive device 2.
  • Inverter output voltage is synonymous with “motor applied voltage” described later.
  • the motor drive device 2 includes an inverter 11, a control unit 25, and a drive signal generation unit 32.
  • the inverter 11 is connected to the single-phase motor 12 and outputs an AC voltage to the single-phase motor 12.
  • a capacitor (not shown) may be inserted between the battery 10 and the inverter 11 for voltage stabilization.
  • the control unit 25 controls the AC voltage output by the inverter 11.
  • the motor drive device 2 includes a current detection unit 22 for detecting the current flowing through the single-phase motor 12, that is, the motor current.
  • the current detection unit 22 is a detection unit that detects a physical quantity representing an operating state of the single-phase motor 12. Specifically, the current detection unit 22 detects the current value of the current flowing through the single-phase motor 12.
  • the current detection unit 22 may be arranged anywhere as long as the current flowing through the single-phase motor 12 can be detected.
  • the current detection unit 22 may be arranged in series with the wiring of the single-phase motor 12, may be arranged in series with the switching element of the inverter 11, or may be arranged in series with the switching element of the inverter 11. Alternatively, it may be placed on the ground line.
  • the current detection method in the current detection unit 22 a method of calculating the current value from Ohm's law by inserting a resistor having a known resistance value and detecting the voltage value, a detection method using a transformer, and a Hall effect are used. There are some detection methods, but any method can be used as long as the current can be detected. In this embodiment, a method of inserting a transformer current sensor in series with the wiring of the single-phase motor 12 will be described.
  • the current detection unit 22 detects the current in the motor drive device 2, the motor drive device 2 may be controlled as described later by using a current converted into a voltage.
  • the inverter 11 is assumed to be a single-phase inverter, it may be any one capable of driving the single-phase motor 12.
  • the control unit 25 the DC voltage V dc detected by the voltage sensor 20, current I m detected by the current detection unit 22, the command value output from the protection signal and the switch 102 are input.
  • Examples of the command value include an effective current command value Ip * due to torque, a rotation speed command value ⁇ *, and the like.
  • Control unit 25 includes a DC voltage V dc, the current I m, based on the command value, to generate a PWM (Pulse Width Modulation) signal Q1, Q2, Q3, Q4. That is, it can be said that the control unit 25 controls the AC voltage output from the inverter 11 to the single-phase motor 12 according to the physical quantities detected by the current detection unit 22 and the voltage sensor 20.
  • PWM Pulse Width Modulation
  • the control unit 25 can estimate the position of the rotor 12a in the rotation direction of the single-phase motor 12 by using the physical quantities detected by the current detection unit 22 and the voltage sensor 20.
  • the control unit 25 controls the AC voltage output from the inverter 11 to the single-phase motor 12 according to a physical quantity, that is, an estimated position of the rotor 12a.
  • the physical quantity detected by the current detection unit 22 and the voltage sensor 20 represents the position of the rotor 12a in the rotation direction of the single-phase motor 12.
  • the switch 102 is, for example, a physical switch, such as a changeover switch for strong operation or weak operation at hand, which is used for a vacuum cleaner or the like.
  • a physical switch such as a changeover switch for strong operation or weak operation at hand, which is used for a vacuum cleaner or the like.
  • the switch 102 is not limited to the physical switch, and may be a process on software in the case of a configuration in which the command value is automatically switched according to the usage time and the state.
  • the drive signal generation unit 32 generates drive signals S1, S2, S3, S4 for driving the switching element of the inverter 11 based on the PWM signals Q1, Q2, Q3, Q4 output from the control unit 25.
  • the drive signal generation unit 32 converts the PWM signals Q1, Q2, Q3, Q4 output from the control unit 25 into drive signals S1, S2, S3, S4 for driving the inverter 11 and outputs the PWM signals to the inverter 11. To do.
  • the drive signal generation unit 32 may have a structure built in the inverter 11 or may be integrated with the control unit 25, and is shown as an example in FIG.
  • An example of the single-phase motor 12 is a brushless motor.
  • the single-phase motor 12 is a brushless motor
  • a plurality of permanent magnets are arranged in the circumferential direction on the rotor 12a of the single-phase motor 12. That is, the single-phase motor 12 has a permanent magnet. These plurality of permanent magnets are arranged so that the magnetizing directions are alternately reversed in the circumferential direction, and form a plurality of magnetic poles of the rotor 12a.
  • a winding (not shown) is wound around the stator 12b of the single-phase motor 12. An alternating current flows through the winding.
  • the current flowing through the winding of the single-phase motor 12 is appropriately referred to as "motor current".
  • the number of magnetic poles of the rotor 12a is assumed to be four, but the number of magnetic poles of the rotor 12a may be other than four poles.
  • FIG. 2 is a diagram showing an example of the circuit configuration of the inverter 11 shown in FIG.
  • the inverter 11 has a plurality of bridge-connected switching elements 51, 52, 53, 54.
  • the switching elements 51 and 52 form the first leg 5A.
  • the switching elements 53 and 54 form the second leg 5B.
  • the switching element 53 and the switching element 54 are connected in series.
  • the switching elements 51 and 53 are located on the high potential side, and the switching elements 52 and 54 are located on the low potential side.
  • the high potential side is generally referred to as an "upper arm” and the low potential side is generally referred to as a "lower arm”.
  • the switching element 51 of the first leg 5A may be referred to as the "upper arm first element”
  • the switching element 53 of the second leg 5B may be referred to as the "upper arm second element”.
  • the switching element 52 of the first leg 5A may be referred to as a "lower arm first element”
  • the switching element 54 of the second leg 5B may be referred to as a "lower arm second element".
  • connection point 6A between the switching element 51 and the switching element 52 and the connection point 6B between the switching element 53 and the switching element 54 form an AC end in the bridge circuit.
  • a single-phase motor 12 and a current detection unit 22 for detecting the current flowing through the single-phase motor 12 are connected between the connection point 6A and the connection point 6B.
  • the current detection unit 22 may be inserted anywhere as long as the current flowing through the single-phase motor 12 can be detected, and the current detection method in the current detection unit 22 is not limited.
  • MOSFET Metal-Oxide-Semiconductor Field-Effective Transistor
  • FET Field-Effective Transistor
  • the switching element 51 is formed with a body diode 51a connected in parallel between the drain and the source of the switching element 51.
  • the switching element 52 is formed with a body diode 52a connected in parallel between the drain and the source of the switching element 52.
  • the switching element 53 is formed with a body diode 53a connected in parallel between the drain and the source of the switching element 53.
  • a body diode 54a connected in parallel between the drain and the source of the switching element 54 is formed in the switching element 54.
  • Each of the plurality of body diodes 51a, 52a, 53a, 54a is a parasitic diode formed inside the MOSFET and is used as a freewheeling diode.
  • a freewheeling diode may be connected separately, and an IGBT (Insulated Gate Bipolar Transistor) may be used instead of the MOSFET.
  • IGBT Insulated Gate Bipolar Transistor
  • the plurality of switching elements 51, 52, 53, 54 are not limited to MOSFETs formed of silicon-based materials, and may be MOSFETs formed of silicon carbide, gallium nitride-based materials, or wide bandgap semiconductors such as diamond. At least one of the plurality of switching elements 51, 52, 53, 54 may be formed of a wide bandgap semiconductor.
  • wide bandgap semiconductors have higher withstand voltage and heat resistance than silicon semiconductors. Therefore, by using the wide bandgap semiconductors for the plurality of switching elements 51, 52, 53, 54, the withstand voltage resistance and the allowable current density of the switching elements are increased, and the semiconductor module incorporating the switching elements can be miniaturized.
  • the wide bandgap semiconductor has high heat resistance, it is possible to reduce the size of the heat dissipation part for dissipating the heat generated by the semiconductor module, and the heat dissipation structure for dissipating the heat generated by the semiconductor module is simple. It is possible to change.
  • FIG. 3 is a block diagram showing a functional part that generates PWM signals Q1, Q2, Q3, and Q4 among the functional parts of the control unit 25 shown in FIG.
  • the carrier comparison unit 38 is input with the advance angle controlled advance phase ⁇ v and the reference phase ⁇ e used when generating the voltage command V ref described later.
  • the reference phase ⁇ e is a phase obtained by converting the rotor mechanical angle ⁇ m , which is an angle from the reference position of the rotor 12a, into an electric angle, and is an estimated phase in the case of position sensorless drive.
  • the "advance angle phase” represents the “advance angle” which is the "advance angle” of the voltage command V ref in terms of phase.
  • the "advance angle” here is a phase difference between the motor applied voltage applied to the winding of the stator 12b and the motor induced voltage induced in the winding of the stator 12b.
  • the “advance angle” takes a positive value when the motor applied voltage is ahead of the motor induced voltage.
  • the carrier comparison unit 38 in addition to the advance phase ⁇ v and the reference phase ⁇ e , the carrier generated by the carrier generation unit 33, the DC voltage V dc, and the voltage which is the amplitude value of the voltage command V ref. Amplitude command V * is input.
  • the carrier comparison unit 38 generates PWM signals Q1, Q2, Q3, and Q4 based on the carrier, the advance phase ⁇ v , the reference phase ⁇ e , the DC voltage V dc, and the voltage amplitude command V * .
  • FIG. 4 is a block diagram showing a configuration example of the carrier comparison unit 38 shown in FIG. FIG. 4 shows a detailed configuration of the carrier comparison unit 38A and the carrier generation unit 33, which are examples of the carrier comparison unit 38.
  • a carrier frequency f C [Hz] which is a carrier frequency
  • f C [Hz] is set in the carrier generation unit 33.
  • f C [Hz] a triangular wave carrier that moves up and down between "0" and "1" is shown as an example of the carrier waveform.
  • the PWM control of the inverter 11 includes synchronous PWM control and asynchronous PWM control. In the case of synchronous PWM control, it is necessary to synchronize the carrier with the advance phase ⁇ v . On the other hand, in the case of asynchronous PWM control, it is not necessary to synchronize the carriers with the advance phase ⁇ v .
  • the carrier comparison unit 38A includes an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38d, a multiplication unit 38f, an addition unit 38e, a comparison unit 38g, a comparison unit 38h, and an output inversion. It has a unit 38i and an output inversion unit 38j.
  • the absolute value calculation unit 38a calculates the absolute value
  • the division unit 38b divides the absolute value
  • the output of the division unit 38b is the modulation factor.
  • the battery voltage which is the output voltage of the battery 10, fluctuates as the current continues to flow. Therefore, in the carrier comparison unit 38A, the value of the modulation factor can be adjusted by dividing the absolute value
  • the multiplication unit 38c calculates a sine value of “ ⁇ e + ⁇ v ” obtained by adding the advance phase ⁇ v to the reference phase ⁇ e .
  • the multiplication unit 38c multiplies the sine value of “ ⁇ e + ⁇ v ” by the modulation factor which is the output of the division unit 38b.
  • the multiplication unit 38d multiplies the voltage command V ref , which is the output of the multiplication unit 38c, by “1/2”.
  • the addition unit 38e adds "1/2" to the output of the multiplication unit 38d.
  • the multiplication unit 38f multiplies the output of the addition unit 38e by "-1".
  • the output of the addition unit 38e is input to the comparison unit 38g as a positive voltage command V ref1 for driving the two switching elements 51, 53 of the upper arm among the plurality of switching elements 51, 52, 53, 54. ..
  • the output of the multiplication unit 38f is input to the comparison unit 38h as a negative voltage command V ref2 for driving the two switching elements 52 and 54 of the lower arm.
  • the comparison unit 38g compares the positive voltage command V ref1 with the amplitude of the carrier.
  • the output of the output inversion unit 38i which is the inverted output of the comparison unit 38g, is the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g is the PWM signal Q2 to the switching element 52.
  • the comparison unit 38h compares the negative voltage command V ref2 with the carrier amplitude.
  • the output of the output inversion unit 38j, which is the inverted output of the comparison unit 38h is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54.
  • the output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time.
  • the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.
  • FIG. 5 is a time chart showing a waveform example of a main part in the carrier comparison unit 38A shown in FIG.
  • FIG. 5 shows the waveform of the positive voltage command V ref1 output from the addition unit 38e, the waveform of the negative voltage command V ref2 output from the multiplication unit 38f, and the waveforms of the PWM signals Q1, Q2, Q3, and Q4. And the waveform of the inverter output voltage are shown.
  • PWM signal Q1 is “high (High)” when “low (Low)” next when the positive voltage command V ref1 is greater than the carrier, the positive voltage command V ref1 is smaller than the carrier.
  • the PWM signal Q2 is an inverted signal of the PWM signal Q1.
  • PWM signal Q3 is “high (High)” when “low (Low)” becomes when negative voltage instruction V ref2 is larger than the carrier, the negative-side voltage instruction V ref2 smaller than the carrier.
  • the PWM signal Q4 is an inverted signal of the PWM signal Q3. As described above, the circuit shown in FIG. 4 is configured with “Low Active", but even if each signal is configured with "High Active” having opposite values. Good.
  • the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2.
  • These voltage pulses are applied from the inverter 11 to the single-phase motor 12 as the motor applied voltage.
  • Bipolar modulation and unipolar modulation are known as modulation methods used by the carrier comparison unit 38A to generate PWM signals Q1, Q2, Q3, and Q4.
  • Bipolar modulation is a modulation method that outputs a voltage pulse that changes with a positive or negative potential for each cycle of the voltage command V ref .
  • Unipolar modulation is a modulation method that outputs a voltage pulse that changes at three potentials for each cycle of the voltage command V ref , that is, a voltage pulse that changes at a positive potential, a negative potential, and a zero potential.
  • the waveform shown in FIG. 5 is due to unipolar modulation.
  • any modulation method may be used. In applications where it is necessary to control the motor current waveform to a more sinusoidal wave, it is preferable to employ unipolar modulation having a lower harmonic content than bipolar modulation.
  • the waveforms shown in FIG. 5 show the switching elements 51 and 52 constituting the first leg 5A and the switching elements 53 and 54 constituting the second leg 5B in the section of the half cycle T / 2 of the voltage command V ref. It is obtained by a method of switching operation of the four switching elements of. This method is called “both-side PWM" because the switching operation is performed by both the positive side voltage command V ref1 and the negative side voltage command V ref2 .
  • the switching operation of the switching elements 51 and 52 is suspended, and in the other half cycle of the one cycle T of the voltage command V ref , the switching operation is suspended.
  • one-sided PWM will be described.
  • FIG. 6 is a block diagram showing another configuration example of the carrier comparison unit 38 shown in FIG.
  • FIG. 6 shows an example of a PWM signal generation circuit by the above-mentioned “one-sided PWM”, and specifically, a detailed configuration of a carrier comparison unit 38B and a carrier generation unit 33, which are examples of the carrier comparison unit 38, is shown. Has been done.
  • the configuration of the carrier generation unit 33 shown in FIG. 6 is the same as or equivalent to that shown in FIG.
  • the configuration of the carrier comparison unit 38B shown in FIG. 6 the same or equivalent components as the carrier comparison unit 38A shown in FIG. 4 are designated by the same reference numerals.
  • the carrier comparison unit 38B includes an absolute value calculation unit 38a, a division unit 38b, a multiplication unit 38c, a multiplication unit 38k, an addition unit 38m, an addition unit 38n, a comparison unit 38g, a comparison unit 38h, and an output inversion. It has a unit 38i and an output inversion unit 38j.
  • the absolute value calculation unit 38a calculates the absolute value
  • the division unit 38b divides the absolute value
  • the multiplication unit 38c calculates a sine value of “ ⁇ e + ⁇ v ” obtained by adding the advance phase ⁇ v to the reference phase ⁇ e .
  • the multiplication unit 38c multiplies the sine value of “ ⁇ e + ⁇ v ” by the modulation factor which is the output of the division unit 38b.
  • the multiplication unit 38k multiplies the voltage command V ref , which is the output of the multiplication unit 38c, by “-1”.
  • the addition unit 38m adds “1” to the voltage command V ref which is the output of the multiplication unit 38c.
  • the addition unit 38n adds “1” to the output of the multiplication unit 38k, that is, the inverted output of the voltage command V ref .
  • the output of the addition unit 38m is input to the comparison unit 38g as the first voltage command V ref3 for driving the two switching elements 51, 53 of the upper arm among the plurality of switching elements 51, 52, 53, 54. ..
  • the output of the addition unit 38n is input to the comparison unit 38h as a second voltage command V ref4 for driving the two switching elements 52 and 54 of the lower arm.
  • the comparison unit 38g compares the first voltage command V ref3 with the amplitude of the carrier.
  • the output of the output inversion unit 38i which is the inverted output of the comparison unit 38g, is the PWM signal Q1 to the switching element 51, and the output of the comparison unit 38g is the PWM signal Q2 to the switching element 52.
  • the comparison unit 38h compares the second voltage command V ref 4 with the amplitude of the carrier.
  • the output of the output inversion unit 38j which is the inverted output of the comparison unit 38h, is the PWM signal Q3 to the switching element 53, and the output of the comparison unit 38h is the PWM signal Q4 to the switching element 54.
  • the output inverting unit 38i does not turn on the switching element 51 and the switching element 52 at the same time.
  • the output inverting unit 38j does not turn on the switching element 53 and the switching element 54 at the same time.
  • FIG. 7 is a time chart showing a waveform example of a main part in the carrier comparison unit 38B shown in FIG.
  • FIG. 7 shows the waveform of the first voltage command V ref3 output from the adder 38m , the waveform of the second voltage command V ref4 output from the adder 38n, and the waveforms of the PWM signals Q1, Q2, Q3, and Q4. And the waveform of the voltage applied to the motor are shown.
  • the waveform portion of the first voltage command V ref3 whose amplitude value is larger than the peak value of the carrier and the second voltage command V ref 4 whose amplitude value is larger than the peak value of the carrier.
  • the corrugated portion is represented by a flat straight line.
  • PWM signal Q1 is "low (Low)” next when the first voltage command V ref3 is larger than the carrier, the first voltage command V ref3 is “high (High)” when less than the carrier.
  • the PWM signal Q2 is an inverted signal of the PWM signal Q1.
  • PWM signal Q3, the second voltage command V ref4 is “high (High)” when “low (Low)” becomes when larger than the carrier, the second voltage command V ref4 smaller than the carrier.
  • the PWM signal Q4 is an inverted signal of the PWM signal Q3.
  • the circuit shown in FIG. 6 is configured with “Low Active", but even if each signal is configured with "High Active” having opposite values. Good.
  • the waveform of the inverter output voltage shows a voltage pulse due to the difference voltage between the PWM signal Q1 and the PWM signal Q4 and a voltage pulse due to the difference voltage between the PWM signal Q3 and the PWM signal Q2.
  • These voltage pulses are applied from the inverter 11 to the single-phase motor 12 as the motor applied voltage.
  • the waveform of the inverter output voltage is unipolar modulation that changes at three potentials for each cycle of the voltage command V ref .
  • bipolar modulation may be used instead of unipolar modulation, but unipolar modulation is preferably adopted in applications where it is necessary to control the motor current waveform to a more sinusoidal wave.
  • FIG. 8 is for calculating the advance phase ⁇ v input to the carrier comparison unit 38A shown in FIG. 4 and the carrier comparison unit 38B shown in FIG. 6 among the functional parts of the control unit 25 shown in FIG. It is a block diagram which shows the functional structure of.
  • FIG. 9 is a diagram showing an example of a method of calculating the advance angle phase ⁇ v in the control unit 25 according to the first embodiment.
  • FIG. 10 is a time chart used to explain the relationship between the voltage command V ref shown in FIGS. 4 and 6 and the advance phase ⁇ v .
  • the function of calculating the advance phase ⁇ v can be realized by the detection current signal processing unit 41, the position estimation unit 42, the rotation speed calculation unit 43, and the advance phase calculation unit 44.
  • Detection current signal processing unit 41 performs signal processing for removing noise with respect to the current I m that is detected by the current detection unit 22.
  • the position estimation unit 42 estimates the position of the rotor 12a of the single-phase motor 12 from the current Im from which noise has been removed by the detection current signal processing unit 41. The detailed configuration and operation of the detection current signal processing unit 41 and the position estimation unit 42 will be described later.
  • the rotation speed calculation unit 43 calculates the rotation speed ⁇ of the single-phase motor 12 based on the estimated position of the rotor 12a of the single-phase motor 12 estimated by the position estimation unit 42. Further, the rotation speed calculation unit 43 calculates a reference phase ⁇ e obtained by converting the rotor mechanical angle ⁇ m , which is an angle from the reference position of the rotor 12a, into an electric angle. In the example of FIG. 10, the portion of the edge where the estimated position falls is used as the reference position of the rotor 12a.
  • the advance angle phase calculation unit 44 calculates the advance angle phase ⁇ v based on the rotation speed ⁇ and the reference phase ⁇ e calculated by the rotation speed calculation unit 43.
  • the rotation speed N is shown on the horizontal axis
  • the advance phase ⁇ v is shown on the vertical axis.
  • the advance phase ⁇ v can be determined by using a function in which the advance phase ⁇ v increases with respect to the increase in the rotation speed N.
  • the advance phase ⁇ v is determined by the linear function of the first order, but the linear function is not limited to the first order. A function other than the linear linear function of the first order may be used as long as the advance phase ⁇ v becomes the same or increases as the rotation speed N increases.
  • the 10 shows a state in which the rotor mechanical angles ⁇ m when the rotor 12a is rotated in the clockwise direction are 0 °, 45 °, 90 °, 135 ° and 180 °.
  • the rotor 12a of the single-phase motor 12 is provided with four magnets, and four teeth 12b1 are provided on the outer circumference of the rotor 12a.
  • the position estimation unit 42 estimates the position of the rotor 12a according to the rotor mechanical angle ⁇ m .
  • the rotation speed calculation unit 43 calculates the reference phase ⁇ e converted into the electric angle based on the estimated position of the rotor 12a of the single-phase motor 12 estimated by the position estimation unit 42.
  • the voltage command V ref having the same phase as the reference phase ⁇ e is output.
  • the amplitude of the voltage command V ref at this time is determined based on the voltage amplitude command V * described above.
  • the voltage command V ref advanced by ⁇ / 4 which is a component of the advance phase ⁇ v from the reference phase ⁇ e, is output.
  • IPMSM Interior Permanent Magnet Synchronous Motor
  • the inductance component of the winding changes according to the rotation angle due to its structure.
  • IPMSM three-phase embedded permanent magnet synchronous motor
  • a high frequency is superimposed on the applied voltage command
  • the inductance value is measured from the high frequency current flowing through the winding, and the position can be estimated.
  • high frequencies are superimposed in this method, there is a problem that high frequency sounds, torque ripples, and the like are generated.
  • FIG. 11 is a diagram showing a configuration example of a single-phase embedded permanent magnet synchronous motor, which is the single-phase motor 12 according to the first embodiment.
  • the single-phase structure IPMSM has a structure in which the inductance value between the terminals of the winding changes with respect to the rotation angle of the rotor 12a. Therefore, the motor drive device 2 can estimate the position of the rotor 12a of the single-phase motor 12 from the current in the same manner as the position estimation described above from the change in the inductance value.
  • a three-phase IPMSM a method of controlling the rotation speed, current, etc. of the motor from the rotating coordinates is common.
  • the rotating coordinates no change in the inductance value of the winding due to the salient pole ratio of the three-phase IPMSM is observed. Therefore, when the three-phase IPMSM is used, the position is estimated by the method of superimposing a high frequency wave faster than the rotational angular velocity as described above.
  • the single-phase IPMSM a change in the inductance value of the winding is observed even on the rotating coordinates.
  • FIG. 11 shows a two-pole rotor 12a, it does not mean that the number of poles is limited.
  • the single-phase motor 12 is assumed to be a single-phase IPMSM as shown in FIG.
  • the magnetic flux is not uniformly distributed with respect to the rotation angle due to the polarity of the rotor 12a. Therefore, in the single-phase motor 12, the inductance component of the coil changes according to the rotation angle as shown in FIG.
  • FIG. 12 is a diagram showing changes in the inductance value of the single-phase motor 12 of the motor drive system 1 according to the first embodiment. The change in the inductance value is as shown in the following equation (1).
  • Lamp is the amplitude of the inductance value
  • Lbias is the offset of the inductance value
  • ⁇ e is the reference phase. As shown in FIG. 12, L bias corresponds to the inductance value when there is no salient pole.
  • the current flowing through the winding of the single-phase motor 12 is as follows. First, the voltage V m applied from the inverter 11 to the single-phase motor 12 is as shown in the following equation (2).
  • V m is the inverter application voltage
  • R is the resistance of the windings of the single-phase motor
  • I n is the current flowing through the windings of the single-phase motor
  • e m is the induced voltage Is. Equation (2) is transformed as follows.
  • Vamp is the voltage AC amplitude applied to the inverter
  • eamp is the induced voltage AC amplitude
  • ⁇ v is the advance phase
  • V m -e m of formula (6) can be expressed as the following equation (9).
  • the equation (12) can be expressed as the equation (13).
  • FIG. 13 shows the result of a simple frequency analysis of the first term of the formula (13).
  • Figure 13 is a diagram showing a result of the current I n flowing in the single-phase motor 12 in accordance with the first embodiment was simplified manner frequency analysis.
  • the fundamental wave 1f component sinx is shown in the spectrum of # 1
  • the 2f component sin2x is shown in the spectrum of # 2
  • 1 / (sin2x + offset) obtained by dividing the waveform of the 2f component with an offset is shown in the spectrum of # 3.
  • Shown. sine / (sin2x + offset) is shown in the spectrum of # 4, and this is assumed to be the first term of equation (13).
  • the frequency component included in the current I n can be readily understood to include odd multiples of the fundamental wave from the spectrum # 4 (2n + 1).
  • This (2n + 1) f component includes a phase component of the inductance L.
  • the position estimation unit 42 by detecting the phase performs signal processing by Fourier transform to be included in the current I n (2n + 1) f of the signal, it is possible to estimate the position of the rotor 12a.
  • n is a natural number.
  • the third-order frequency component is used when the phase is detected accurately. It is good to use. However, depending on the signal processing method and the configuration of the current detection circuit, it is not always possible to accurately detect the third-order frequency component. Therefore, there is no limitation as to which component of the odd-numbered harmonics such as the 5th and 7th harmonics is used.
  • the motor drive device 2 can improve the accuracy of phase detection by performing phase detection on each of a plurality of odd-numbered multiple spectra and averaging them. For example, the phase is detected from the 3rd, 5th, and 7th harmonics and averaged. At this time, since the motor drive device 2 has a large power of the spectrum having a low order as described above, even if the averaging is weighted for each order in order to strongly reflect the detection result of the spectrum having a low order. Good.
  • the position is estimated by extracting the frequency component of the odd multiple (2n + 1) of the fundamental wave of the motor current, it is necessary to detect the frequency component of the odd multiple (2n + 1) of the fundamental wave when detecting the motor current. ..
  • various methods such as a method using a detector such as a current sensor and a method of calculating the current from the voltage drop due to the shunt resistor can be considered.
  • a low-pass filter (LPF: Low Pass Filter) is generally used to remove noise components.
  • LPF Low Pass Filter
  • FIG. 14 is a diagram showing a configuration example of the detection current signal processing unit 41 according to the first embodiment.
  • FIG. 15 is a diagram showing another configuration example of the detection current signal processing unit 41 according to the first embodiment. Since the fundamental frequency of the winding current changes in proportion to the rotation speed ⁇ of the motor, the harmonics of odd multiples also change in proportion to the rotation speed of the motor. Therefore, as shown in FIG. 14, the detection current signal processing unit 41 prepares a plurality of LPFs having different cutoff frequencies, and selects the current signal to be detected according to the current rotation speed of the motor by the selector circuit 411. .. Alternatively, as shown in FIG.
  • the detection current signal processing unit 41 may use one LPF having a cutoff frequency three times the maximum rotation speed fmax of the motor in advance. In any method, the detection current signal processing unit 41 may detect at least the third harmonic component of the fundamental wave and the harmonics of odd multiples of the current flowing through the winding.
  • FIG. 16 is a flowchart showing an operation of estimating the position of the rotor 12a of the single-phase motor 12 in the motor drive device 2 according to the first embodiment.
  • the motor drive device 2, the current detection unit 22 detects the current I m (step ST1).
  • Position estimating unit 42, the current I m which noise has been removed by the detection current signal processing unit 41 acquires a current including a third-order harmonic component to the fundamental wave 1f (step ST3).
  • the position estimation unit 42 performs a Fourier transform on the current including the third-order harmonic component to extract the phase of the third-order component (step ST4).
  • the position estimation unit 42 estimates the position of the rotor 12a of the single-phase motor 12 from the extracted phase (step ST5).
  • the operations after the rotation speed calculation unit 43 are as described above.
  • the LPF has been described as a filter used by the detection current signal processing unit 41, the present invention is not limited to this. As long as the detection current signal processing unit 41 can detect the fundamental wave of the current and the harmonics of odd multiples such as a bandpass filter, the form of the filter does not matter. Further, in the detection current signal processing unit 41, the filter may be configured as an analog circuit on the circuit or as a digital circuit inside the microcomputer.
  • the motor drive device 2 directly detects the voltage applied to the motor and detects the frequency phase of an odd multiple of the fundamental wave. You may estimate the position.
  • the detection target of the voltage sensor 20 may be the inverter output voltage, which is the output voltage of the motor drive device 2. That is, the control unit 25 detects a voltage containing a frequency component 2n + 1 times the fundamental wave of the AC voltage output by the inverter 11 to the single-phase motor 12 from the AC voltage output to the single-phase motor 12, and 2n + 1. The phase is calculated from the double frequency component, and the position of the rotor 12a of the single-phase motor 12 is estimated.
  • the position estimation method using the inductance component having the salient pole ratio by the above single-phase IPMSM structure can estimate the position without using the position sensor, and unlike the method using the induced voltage, it can be used even when the motor is at low speed. It is a big merit to be able to exert the effect.
  • FIG. 17 is a diagram showing an example of a hardware configuration that realizes the control unit 25 included in the motor drive device 2 according to the first embodiment.
  • the control unit 25 is realized by the processor 201 and the memory 202.
  • the processor 201 is a CPU (Central Processing Unit, central processing unit, processing unit, arithmetic unit, microprocessor, microprocessor, processor, DSP (Digital Signal Processor)), or system LSI (Large Scale Integration).
  • the memory 202 includes RAM (Random Access Memory), ROM (Read Only Memory), flash memory, EPROM (Erasable Programmable Read Only Memory), EEPROM (registered trademark) (Registered Trademark) (Electrically General Memory), and EEPROM (Registered Trademark).
  • RAM Random Access Memory
  • ROM Read Only Memory
  • flash memory flash memory
  • EPROM Erasable Programmable Read Only Memory
  • EEPROM registered trademark
  • a semiconductor memory can be exemplified.
  • the memory 202 is not limited to these, and may be a magnetic disk, an optical disk, a compact disk, a mini disk, or a DVD (Digital Versaille Disc).
  • Embodiment 2 In the second embodiment, an application example of the motor drive device 2 described in the first embodiment will be described.
  • FIG. 18 is a diagram showing a configuration example of an electric blower 64 including the motor drive device 2 according to the second embodiment.
  • the electric blower 64 includes the motor drive device 2 described in the first embodiment, and the propeller 69 is attached to the single-phase motor 12 driven by the motor drive device 2.
  • the electric blower 64 has a structure in which the motor driving device 2 rotates the single-phase motor 12 to send out or suck the wind.
  • FIG. 19 is a diagram showing a configuration example of a vacuum cleaner 61 including the electric blower 64 according to the second embodiment.
  • the vacuum cleaner 61 includes a battery 67 corresponding to the battery 10 shown in FIG. 1, a motor driving device 2 shown in FIG. 1, and an electric blower 64 driven by a single-phase motor 12 shown in FIG. .. Further, the vacuum cleaner 61 includes a dust collecting chamber 65, a sensor 68, a suction port 63, an extension pipe 62, and an operation unit 66.
  • the user who uses the vacuum cleaner 61 has an operation unit 66 and operates the vacuum cleaner 61.
  • the motor drive device 2 of the vacuum cleaner 61 drives the electric blower 64 using the battery 67 as a power source. By driving the electric blower 64, dust is sucked from the suction port 63. The sucked dust is collected in the dust collecting chamber 65 via the extension pipe 62.
  • the vacuum cleaner 61 is a product in which the rotation speed of the single-phase motor 12 fluctuates from 0 [rpm] to over 100,000 [rpm].
  • a single-phase motor 12 drives a product that rotates at high speed, a high carrier frequency is required. Therefore, in the conventional current detection method, it is difficult to adjust the A / D conversion timing, the switching time is shortened, and the detection becomes more difficult. Therefore, the control method according to the first embodiment described above is suitable.
  • the control unit 25 suspends the switching operation between the upper arm first element and the lower arm first element in one half cycle of the voltage command cycle. In the other half cycle of the voltage command cycle, the switching operation between the upper arm second element and the lower arm second element is suspended. As a result, an increase in switching loss is suppressed, and an efficient vacuum cleaner 61 can be realized.
  • the switching elements 51, 52, 53, 54 of the inverter 11 are formed of a wide bandgap semiconductor, so that the heat dissipation parts can be simplified to reduce the size and weight. it can.
  • FIG. 20 is a diagram showing a configuration example of a hand dryer 90 including the electric blower 64 according to the second embodiment.
  • the hand dryer 90 includes a casing 91, a hand detection sensor 92, a water receiving portion 93, a drain container 94, a cover 96, a sensor 97, an intake port 98, and an electric blower 64.
  • the sensor 97 is either a gyro sensor or a motion sensor.
  • the hand dryer 90 when the hand is inserted into the hand insertion portion 99 at the upper part of the water receiving portion 93, the water is blown off by the blown air by the electric blower 64, and the blown water is collected by the water receiving portion 93. After that, it is stored in the drain container 94.
  • the hand dryer 90 is a product in which the motor rotation speed fluctuates from 0 [rpm] to over 100,000 [rpm], similar to the vacuum cleaner 61 shown in FIG. Therefore, also in the hand dryer 90, the control method according to the above-described embodiment is suitable, and the same effect as that of the vacuum cleaner 61 can be obtained.
  • the motor drive device 2 is applied to an electric device equipped with a motor.
  • Electrical equipment equipped with motors includes incinerators, crushers, dryers, dust collectors, printing machines, cleaning machines, confectionery machines, tea making machines, woodworking machines, plastic extruders, cardboard machines, packaging machines, hot air generators, and OA.
  • Equipment, electric blowers, etc. The electric blower is a blower means for transporting an object, collecting dust, or for general blowing and exhausting.
  • the configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is one of the configurations without departing from the gist of the present invention. It is also possible to omit or change the part.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

Le dispositif d'entraînement de moteur comprend : un onduleur (11) pour délivrer en sortie une tension alternative à un moteur monophasé (12) ayant un aimant permanent ; une unité de détection pour détecter une quantité physique indiquant l'état de fonctionnement du moteur monophasé (12) ; et une unité de commande (25) pour détecter, à partir de la quantité physique détectée par l'unité de détection, un courant comprenant une composante de fréquence de (2n +1) fois l'onde fondamentale de la tension alternative qui est délivrée au moteur monophasé (12) par l'onduleur (11), n étant un nombre naturel.
PCT/JP2019/015855 2019-04-11 2019-04-11 Dispositif d'entraînement de moteur, ventilateur électrique, aspirateur électrique et sèche-mains WO2020208789A1 (fr)

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CN201980094945.5A CN113647011A (zh) 2019-04-11 2019-04-11 马达驱动装置、电动鼓风机、电动吸尘器以及干手器
JP2021513124A JP7170848B2 (ja) 2019-04-11 2019-04-11 モータ駆動装置、電動送風機、電気掃除機及びハンドドライヤ
PCT/JP2019/015855 WO2020208789A1 (fr) 2019-04-11 2019-04-11 Dispositif d'entraînement de moteur, ventilateur électrique, aspirateur électrique et sèche-mains

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Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPS6218955A (ja) * 1985-05-24 1987-01-27 エヌ・ベ−・フイリツプス・フル−イランペンフアブリケン 単相同期モ−タ
JPH1023765A (ja) * 1996-07-05 1998-01-23 Matsushita Refrig Co Ltd Ｐｗｍ方式電圧形インバータ
WO2018229874A1 (fr) * 2017-06-13 2018-12-20 三菱電機株式会社 Dispositif d'entraînement de moteur, souffleur électrique d'air, aspirateur électrique, et séchoir à mains

Family Cites Families (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2017077579A1 (fr) * 2015-11-02 2017-05-11 三菱電機株式会社 Dispositif d'entraînement de moteur, machine à vide électrique et sèche-mains
WO2018073869A1 (fr) * 2016-10-17 2018-04-26 三菱電機株式会社 Dispositif d'excitation de moteur, ventilateur électrique, aspirateur électrique et sèche-mains
JP6671516B2 (ja) * 2017-01-25 2020-03-25 三菱電機株式会社 モータ駆動装置、電動送風機、電気掃除機及びハンドドライヤ

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPS6218955A (ja) * 1985-05-24 1987-01-27 エヌ・ベ−・フイリツプス・フル−イランペンフアブリケン 単相同期モ−タ
JPH1023765A (ja) * 1996-07-05 1998-01-23 Matsushita Refrig Co Ltd Ｐｗｍ方式電圧形インバータ
WO2018229874A1 (fr) * 2017-06-13 2018-12-20 三菱電機株式会社 Dispositif d'entraînement de moteur, souffleur électrique d'air, aspirateur électrique, et séchoir à mains

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