WO2019218389A1 - 基于虚拟电压注入的感应电机无速度传感器驱动控制方法 - Google Patents

基于虚拟电压注入的感应电机无速度传感器驱动控制方法 Download PDF

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WO2019218389A1
WO2019218389A1 PCT/CN2018/088315 CN2018088315W WO2019218389A1 WO 2019218389 A1 WO2019218389 A1 WO 2019218389A1 CN 2018088315 W CN2018088315 W CN 2018088315W WO 2019218389 A1 WO2019218389 A1 WO 2019218389A1
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axis
motor
speed
induction motor
rotor
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PCT/CN2018/088315
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English (en)
French (fr)
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孙伟
徐殿国
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华中科技大学
哈尔滨工业大学
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Priority to US16/772,160 priority Critical patent/US11081992B2/en
Publication of WO2019218389A1 publication Critical patent/WO2019218389A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/04—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for very low speeds
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H02P21/0021—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control using different modes of control depending on a parameter, e.g. the speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P23/00—Arrangements or methods for the control of AC motors characterised by a control method other than vector control
    • H02P23/12—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P23/00—Arrangements or methods for the control of AC motors characterised by a control method other than vector control
    • H02P23/14—Estimation or adaptation of motor parameters, e.g. rotor time constant, flux, speed, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
    • H02P6/183—Circuit arrangements for detecting position without separate position detecting elements using an injected high frequency signal
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/01—Asynchronous machines

Definitions

  • the invention belongs to the field of motor control, and particularly relates to a speed sensorless driving control method for an induction motor based on virtual voltage injection.
  • Induction Motor relies on electromagnetic induction to cause the rotor to induce current to realize electromechanical energy conversion.
  • An AC motor is essentially an asynchronous motor.
  • Motor speed detection devices often use speed sensors.
  • the installation of these speed sensors increases the cost of the motor control system.
  • the speed sensor is not suitable for harsh environments such as humidity, vibration and electromagnetic noise interference, thus limiting its application range. Therefore, the speed sensorless technology, that is, how to quickly and accurately estimate the motor speed value through the known speed control system parameters, has become another hot spot in today's research.
  • the speed sensorless drive control system also has drawbacks: the motor has low load carrying capacity at low speed and low speed power generation instability.
  • the object of the present invention is to solve the technical problem of the instability of the prior art induction motor speed sensorless control system at low synchronous speed and zero synchronous speed.
  • the present invention provides a speed sensorless driving control method for an induction motor based on virtual voltage injection, which is based on an existing induction motor speed sensorless driving system, and the stator of the motor in an ⁇ coordinate system Voltage command input values u s ⁇ , u s ⁇ and flux observer observer voltage input values Add a virtual voltage injection module between them, or the stator voltage command input values u sd , u sq and flux linkage observer stator voltage input values in the dq coordinate system Adding a virtual voltage injection module implementation, the method includes the following steps:
  • This operation is equivalent to injecting u s ⁇ _inj and u s ⁇ _inj on the basis of u s ⁇ and u s ⁇ , where Make it satisfy
  • u s ⁇ _inj is the virtual voltage injection value under the ⁇ axis
  • u s ⁇ _inj is the virtual voltage injection value under the ⁇ axis
  • u sd_inj is the virtual voltage injection value under the d-axis
  • u sq_inj is the virtual voltage injection value under the q-axis
  • the ⁇ coordinate system is a 2-phase stationary coordinate system
  • the dq coordinate system is a 2-phase synchronous rotation coordinate system.
  • the virtual voltage injection module is implemented by an adder, a multiplier, or a combination thereof.
  • step S1 the formula for calculating the proportional relationship k in step S1 is as follows:
  • p is a constant greater than zero, according to the stability of the induction motor speed and torque;
  • R r is the rotor resistance of the induction motor;
  • L m is the mutual inductance of the induction motor;
  • L r is the rotor side inductance of the induction motor.
  • step S3 The dynamic mathematical model of the constructed flux observer is as follows:
  • stator current observation values under the ⁇ -axis, ⁇ -axis, d-axis, and q-axis are respectively;
  • ⁇ e is the synchronous rotation speed;
  • R s and R r are the stator resistance and rotor resistance of the motor respectively;
  • L s , L r , and L m are respectively Motor stator side inductance, motor rotor side inductance and motor mutual inductance.
  • step S4 based on When constructing a dynamic mathematical model of the flux observer, the rotor speed of the induction motor in step S4 Calculated as follows:
  • step S4 calculates the rotor speed of the induction motor in step S4 as follows:
  • step S4 Calculated as follows:
  • step S4 Calculated as follows:
  • step S5 includes the following steps:
  • the flux linkage current command gives the output of the module As a command of the d-axis current PI adjustment module;
  • the two-phase currents i U and i V of the induction motor sampled by the current sensor are input to the 3-phase stationary coordinate/2-phase stationary coordinate transformation module, and then output To the 2-phase synchronous rotating coordinate/2-phase stationary coordinate transformation module, finally obtain the d-axis current i sd and the q-axis current i sq in the 2-phase synchronous rotating coordinate system, and obtain the d-axis current and the q-axis current as d respectively.
  • Feedback amount of the shaft current PI regulator and the q-axis current PI regulator, and the corresponding flux linkage current command with Perform current PI control after doing the difference;
  • the output to the voltage space vector pulse width modulation module generates a switching signal capable of controlling the switching devices S A , S B , and S C , thereby achieving the purpose of controlling the speed and torque of the induction motor.
  • an embodiment of the present invention provides a computer readable storage medium, where the computer readable storage medium stores a computer program, and when the computer program is executed by a processor, implementing the virtual Induction motor with voltage injection has no speed sensor drive control method.
  • the present invention inputs the values u s ⁇ , u s ⁇ and the flux observer through the stator voltage command of the motor in the ⁇ coordinate system.
  • Stator voltage input value Add a virtual voltage injection module between them, or the stator voltage command input values u sd , u sq and flux linkage observer stator voltage input values in the dq coordinate system
  • a virtual voltage injection module is added between them to achieve:
  • the induction motor controlled by the speed sensorless drive system of the induction motor can output 150% of the rated motor torque at zero synchronous speed or low synchronous speed.
  • the induction motor controlled by the speed sensorless drive system of the induction motor runs stably for a long time at 0% motor rated torque and zero rotor speed, and can be operated for a long time. Normal acceleration starts.
  • the induction motor controlled by the speed sensorless drive system of the induction motor can perform the motor speed with any acceleration/deceleration time under the condition that the load is 150% of the rated torque of the motor. Positive and negative switching.
  • FIG. 1 is a schematic structural diagram of a speed sensorless driving control system for an induction motor based on virtual voltage injection according to a first embodiment of the present invention.
  • FIG. 2 is a schematic structural diagram of a speed sensorless driving control system for an induction motor based on virtual voltage injection according to a second embodiment of the present invention.
  • FIG. 3 is a graph showing the performance of the rotor speed of the induction motor in different stages according to an embodiment of the present invention.
  • FIG. 4 is a graph showing changes of a U-phase stator current and a motor rotor speed command value with time according to an embodiment of the present invention.
  • FIG. 1 is a schematic structural diagram of a speed sensorless driving control system for an induction motor based on virtual voltage injection according to a first embodiment of the present invention.
  • the hardware part of the induction motor speed sensorless drive system includes: a three-phase voltage source inverter and an induction motor.
  • the three-phase AC power supply is subjected to uncontrolled rectification to obtain a DC bus voltage U DC , which is supplied to a voltage source type inverter, and an inverter is used to control the induction motor to perform torque and speed control.
  • Voltage and current sensors are included in the three-phase voltage source inverter.
  • the software part of the induction motor speed sensorless drive system includes: 3-phase stationary coordinate/2-phase stationary coordinate transformation module, 2-phase synchronous rotation coordinate/2-phase stationary coordinate transformation module, voltage space vector pulse width modulation module, current PI (Proportion Integration The adjustment module, the speed PI adjustment module, the flux linkage current command given module, the rotational speed command given module, the flux linkage and the speed observer module and the virtual voltage injection module.
  • the control mode of the induction motor is mainly divided into VF control, vector control and direct torque control, and the embodiment of the invention preferably has a vector control strategy.
  • the invention mainly relates to a virtual voltage injection module, and the other modules are functional modules of the induction motor speed sensorless vector control, which are common knowledge in the art.
  • the virtual voltage injection module is implemented by an adder, a multiplier, or a combination thereof.
  • Embodiment 1 The control system inputs the values u s ⁇ , u s ⁇ and the flux observer stator voltage input value by the stator voltage command of the motor in the ⁇ coordinate system.
  • a virtual voltage injection module is added between the implementations. The control method of the entire system is described below, including steps S1 to S5.
  • p is a constant greater than zero, according to the stability of the induction motor speed and torque;
  • R r is the rotor resistance of the induction motor;
  • L m is the mutual inductance of the induction motor;
  • L r is the rotor side inductance of the induction motor.
  • This operation is equivalent to injecting u s ⁇ _inj and u s ⁇ _inj on the basis of u s ⁇ and u s ⁇ , where Make it satisfy
  • u s ⁇ _inj is the virtual voltage injection value under the ⁇ axis
  • u s ⁇ _inj is the virtual voltage injection value under the ⁇ axis
  • stator current observation values under the ⁇ -axis and ⁇ -axis are respectively;
  • R s and R r are the stator resistance and rotor resistance of the motor respectively;
  • L s , L r and L m are respectively the stator side inductance of the motor, the rotor side inductance of the motor and the mutual inductance of the motor. .
  • k p and k i are the proportional link gain and the integral link gain of the speed observer respectively; i s ⁇ and i s ⁇ are the actual measured values of the stator current under the ⁇ axis and the ⁇ axis, respectively; Observed values of stator current under the ⁇ -axis and ⁇ -axis, respectively; The values of the rotor flux linkage under the ⁇ -axis and the ⁇ -axis are respectively; S 1 is Time integral value.
  • step S5 includes the following steps:
  • the flux linkage current command gives the output of the module As a command of the d-axis current PI adjustment module;
  • the two-phase currents i U and i V of the induction motor sampled by the current sensor are input to the 3-phase stationary coordinate/2-phase stationary coordinate transformation module, and then output To the 2-phase synchronous rotating coordinate/2-phase stationary coordinate transformation module, finally obtain the d-axis current i sd and the q-axis current i sq in the 2-phase synchronous rotating coordinate system, and obtain the d-axis current and the q-axis current as d respectively.
  • Feedback amount of the shaft current PI regulator and the q-axis current PI regulator, and the corresponding flux linkage current command with Perform current PI control after doing the difference;
  • the output to the voltage space vector pulse width modulation module generates a switching signal capable of controlling the switching devices S A , S B , and S C , thereby achieving the purpose of controlling the speed and torque of the induction motor.
  • the second embodiment control system inputs the values u sd , u sq and the flux linkage observer stator voltage input values by the stator voltage command of the motor in the dq coordinate system.
  • a virtual voltage injection module is added between the implementations. The control method of the entire system is described below, including steps S1 to S5.
  • This operation is equivalent to injecting u sd_inj and u sq_inj based on u sd and u sq , where Make it satisfy
  • u sd_inj is the virtual voltage injection value under the d-axis
  • u sq_inj is the virtual voltage injection value under the q-axis
  • the stator flux linkage observation values under the d-axis and q-axis are respectively; Observed values of the rotor flux linkage under the d-axis and q-axis, respectively;
  • the stator current observation values are d-axis and q-axis respectively;
  • ⁇ e is the synchronous speed;
  • R s and R r are the stator resistance and rotor resistance of the motor respectively;
  • L s , L r and L m are respectively the stator side inductance of the motor and the motor Rotor side inductance and motor mutual inductance.
  • k p and k i are the proportional link gain and the integral link gain of the speed observer respectively; i sd and i sq are the actual measured values of the stator current under the d-axis and q-axis, respectively; The stator current observation values under the d-axis and q-axis are respectively; The rotor flux linkage observations under the d-axis and q-axis are respectively; S 2 is The time integral value, i sq is the actual measured value of the stator current under the q-axis, ⁇ s is the slip speed, R r is the rotor resistance of the motor, L r , L m are the inductance of the rotor side of the motor and the mutual inductance of the motor, respectively, S 3 represents the pair Time integral.
  • the flux linkage current command gives the output of the module As a command of the d-axis current PI adjustment module;
  • the two-phase currents i U and i V of the induction motor sampled by the current sensor are input to the 3-phase stationary coordinate/2-phase stationary coordinate transformation module, and then output To the 2-phase synchronous rotating coordinate/2-phase stationary coordinate transformation module, finally obtain the d-axis current i sd and the q-axis current i sq in the 2-phase synchronous rotating coordinate system, and obtain the d-axis current and the q-axis current as d respectively.
  • Feedback amount of the shaft current PI regulator and the q-axis current PI regulator, and the corresponding flux linkage current command with Perform current PI control after doing the difference;
  • the output to the voltage space vector pulse width modulation module generates a switching signal capable of controlling the switching devices S A , S B , and S C , thereby achieving the purpose of controlling the speed and torque of the induction motor.
  • FIG. 3 is a graph showing the performance of the rotor speed of the induction motor in different stages according to an embodiment of the present invention.
  • the motor rotor speed can be stabilized at zero synchronous speed of the motor and 0% load torque, zero synchronous speed and 150% load torque and low synchronous speed and 150% load torque.
  • FIG. 4 is a graph showing changes in U-phase stator current and motor rotor speed value of a motor according to an embodiment of the present invention.
  • the rotor speed of the motor can be kept stable from -120 rpm to 120 rpm at 150% load torque.

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Abstract

本发明公开了基于虚拟电压注入的感应电机无速度传感器驱动控制方法，首先在电机磁链和转速观测器中注入虚拟电压信号，使得电机磁链和和转速观测器的输入与电机的指令输入存在差异；其次以任一种电机磁链和转速观测器为基础对电机磁链旋转角度和电机转子转速进行估计，并以某种控制策略(如矢量控制)驱动感应电机正常运行；然后，依照本方法设计出的、仅注入在电机磁链和转速观测器中的信号，能够保证在感应电机无速度传感器控制系统驱动下的感应电机，在电机低同步转速以及电机零同步转速运行时输出150％的额定转矩，并长时间保持其稳定。

Description

基于虚拟电压注入的感应电机无速度传感器驱动控制方法 【技术领域】
本发明属于电机控制领域，具体涉及基于虚拟电压注入的感应电机无速度传感器驱动控制方法。
【背景技术】
感应电机(Induction Motor，简称IM)依靠电磁感应作用使转子感应电流从而实现机电能量转换的一种交流电机，其实质上是一种异步电机(Asynchronous Motor)。
电机转速检测装置多采用速度传感器，这些速度传感器的安装增加了电机控制系统的成本，此外，速度传感器不适用于潮湿、震荡和电磁噪声干扰大等恶劣环境，因而限制了其应用范围。于是，无速度传感器技术，也就是如何通过已知的调速系统参数快速而准确地估算出电机的转速值，已成为当今研究的又一个热点。但无速度传感器驱动控制系统也存在缺陷：电机低速运行带载能力弱及低速发电不稳定性。
为保证感应电机无速度传感器驱动系统在电机同步转速为零或电机转子转速为零时能够长时间带载稳定运行，近年来人们做了很多研究，主要包括低频电流信号注入法、高频电流/电压信号注入法、检测零序电流谐波估算转子位置法等。但是上述方法要求感应电机具有明显的磁场异向性，并依赖电机设计，存在转矩波动、噪声等问题。尚无企业或研究机构能够在不对电机进行信号注入的情况下，实现感应电机无速度传感器驱动系统在零同步转速下的带载稳定运行。
【发明内容】
针对现有技术的缺陷，本发明的目的在于解决现有感应电机无速度传感器控制系统在低同步转速和零同步转速下的不稳定的技术问题。
为实现上述目的，第一方面，本发明提供了基于虚拟电压注入的感应电机无速度传感器驱动控制方法，该方法基于现有的感应电机无速度传感器驱动系统，通过在αβ坐标系下电机的定子电压指令输入值u sα、u sβ和磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000001
之间增加一虚拟电压注入模块，或者，在dq坐标系下电机的定子电压指令输入值u sd、u sq和磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000002
之间增加一虚拟电压注入模块实现，该方法包括如下步骤：
S1.基于感应电机的参数计算k，k为虚拟电压注入模块中的比例关系；
S2.将αβ坐标系下电机的定子电压指令输入值u sα、u sβ分别乘以比例关系k，获得αβ坐标系下磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000003
或者，将dq坐标系下电机的定子电压指令输入值u sd、u sq分别乘以比例关系k，获得dq坐标系下磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000004
该操作相当于在u sα和u sβ的基础上，注入了u sα_inj和u sβ_inj，其中
Figure PCTCN2018088315-appb-000005
使得满足
Figure PCTCN2018088315-appb-000006
式中，u sα_inj为α轴下的虚拟电压注入值，u sβ_inj为β轴下的虚拟电压注入值；
或者，相当于在u sd和u sq的基础上，注入了u sd_inj和u sq_inj，其中
Figure PCTCN2018088315-appb-000007
使得满足
Figure PCTCN2018088315-appb-000008
式中，u sd_inj为d轴下的虚拟电压注入值，u sq_inj为q轴下的虚拟电压注入值；
S3.基于
Figure PCTCN2018088315-appb-000009
或
Figure PCTCN2018088315-appb-000010
构建磁链观测器的动态数学模型；
S4.基于磁链观测器的动态数学模型，使用转速观测器观测感应电机转 子转速
Figure PCTCN2018088315-appb-000011
使用磁链观测器观测转子磁链的旋转角度
Figure PCTCN2018088315-appb-000012
S5.观测出的转子转速
Figure PCTCN2018088315-appb-000013
被用于转速PI调节模块和磁链观测器，观测出的转子磁链旋转角度
Figure PCTCN2018088315-appb-000014
用于2相同步旋转坐标/2相静止坐标变换模块，从而实现感应电机无速度传感器转速和转矩的控制；
其中，αβ坐标系是2相静止坐标系，dq坐标系是2相同步旋转坐标系。
具体地，虚拟电压注入模块通过加法器、乘法器或其组合实现。
具体地，步骤S1中比例关系k的计算公式如下：
Figure PCTCN2018088315-appb-000015
其中，p为大于零的常数，根据感应电机转速和转矩的稳定程度来取值；R r为感应电机转子电阻；L m为感应电机互感；L r为感应电机转子侧电感。
具体地，步骤S3中基于
Figure PCTCN2018088315-appb-000016
构建的磁链观测器的动态数学模型如下：
Figure PCTCN2018088315-appb-000017
其中：
Figure PCTCN2018088315-appb-000018
Figure PCTCN2018088315-appb-000019
Figure PCTCN2018088315-appb-000020
Figure PCTCN2018088315-appb-000021
基于
Figure PCTCN2018088315-appb-000022
构建的磁链观测器的动态数学模型如下：
Figure PCTCN2018088315-appb-000023
其中：
Figure PCTCN2018088315-appb-000024
Figure PCTCN2018088315-appb-000025
Figure PCTCN2018088315-appb-000026
Figure PCTCN2018088315-appb-000027
其中，
Figure PCTCN2018088315-appb-000028
分别为α轴、β轴、d轴、q轴下的定子磁链观测值；
Figure PCTCN2018088315-appb-000029
分别为α轴、β轴、d轴、q轴下的转子磁链观测值；
Figure PCTCN2018088315-appb-000030
分别为α轴、β轴、d轴、q轴下的定子电流观测值；ω e为同步转速；R s、R r分别为电机定子电阻和转子电阻；L s、L r、L m分别为电机定子侧电感、电机转子侧电感和电机互感。
具体地，基于
Figure PCTCN2018088315-appb-000031
构建磁链观测器的动态数学模型时，步骤S4中感应电机转子转速
Figure PCTCN2018088315-appb-000032
计算公式如下：
Figure PCTCN2018088315-appb-000033
基于
Figure PCTCN2018088315-appb-000034
构建磁链观测器的动态数学模型时，步骤S4中感应电机转子转速
Figure PCTCN2018088315-appb-000035
计算公式如下：
Figure PCTCN2018088315-appb-000036
其中，k p、k i分别为转速观测器的比例环节增益和积分环节增益；i sα、i sβ、i sd、i sq分别为α轴、β轴、d轴、q轴下定子电流实际测量值；
Figure PCTCN2018088315-appb-000037
Figure PCTCN2018088315-appb-000038
分别为α轴、β轴、d轴、q轴下定子电流观测值；
Figure PCTCN2018088315-appb-000039
分别为α轴、β轴、d轴、q轴下的转子磁链观测值；S 1、S 2分别为
Figure PCTCN2018088315-appb-000040
的时间积分值。
具体地，基于
Figure PCTCN2018088315-appb-000041
构建磁链观测器的动态数学模型时，步骤S4中旋转角度
Figure PCTCN2018088315-appb-000042
计算公式如下：
Figure PCTCN2018088315-appb-000043
基于
Figure PCTCN2018088315-appb-000044
构建磁链观测器的动态数学模型时，步骤S4中旋转角度
Figure PCTCN2018088315-appb-000045
计算公式如下：
Figure PCTCN2018088315-appb-000046
Figure PCTCN2018088315-appb-000047
其中，
Figure PCTCN2018088315-appb-000048
分别为α轴、β轴、d轴下的转子磁链观测值；i sq为q轴下定子电流实际测量值，ω s为滑差转速，R r为电机转子电阻，L r、L m分别为电机转子侧电感和电机互感，S 3代表对
Figure PCTCN2018088315-appb-000049
的时间积分。
具体地，步骤S5包括以下步骤：
S501，观测出的感应电机转子转速
Figure PCTCN2018088315-appb-000050
作为转速PI调节模块的反馈量，与对应的转速指令
Figure PCTCN2018088315-appb-000051
做差后进行转速PI控制；
S502，观测出的磁链旋转角度
Figure PCTCN2018088315-appb-000052
被用于2相同步旋转坐标/2相静止坐标变换模块中的坐标变换计算；
S503，转速PI调节模块的输出
Figure PCTCN2018088315-appb-000053
作为q轴电流PI调节模块的指令，磁链电流指令给定模块的输出
Figure PCTCN2018088315-appb-000054
作为d轴电流PI调节模块的指令；由电流传感器采样得到的感应电机两相电流i U、i V，输入3相静止坐标/2相静止坐标 变换模块，然后输出
Figure PCTCN2018088315-appb-000055
至2相同步旋转坐标/2相静止坐标变换模块，最后得到2相同步旋转坐标系下的d轴电流i sd和q轴电流i sq，并将得到的d轴电流和q轴电流分别作为d轴电流PI调节器和q轴电流PI调节器的反馈量，与对应的磁链电流指令
Figure PCTCN2018088315-appb-000056
和
Figure PCTCN2018088315-appb-000057
做差后进行电流PI控制；
S504，d轴和q轴电流PI调节模块的输出u sd和u sq至2相同步旋转坐标/2相静止坐标变换模块，其将2相同步旋转坐标系下的电机输入电压指令转换为2相静止坐标系下的电机输入电压指令
Figure PCTCN2018088315-appb-000058
S505，将
Figure PCTCN2018088315-appb-000059
输出至电压空间矢量脉宽调制模块，产生能够控制开关器件S A、S B、S C的开关信号，进而达到控制感应电机转速和转矩的目的。
第二方面，本发明实施例提供了一种计算机可读存储介质，所述计算机可读存储介质上存储有计算机程序，所述计算机程序被处理器执行时实现上述第一方面所述的基于虚拟电压注入的感应电机无速度传感器驱动控制方法。
总体而言，通过本发明所构思的以上技术方案与现有技术相比，具有以下有益效果：本发明通过在αβ坐标系下电机的定子电压指令输入值u sα、u sβ和磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000060
之间增加一虚拟电压注入模块，或者，在dq坐标系下电机的定子电压指令输入值u sd、u sq和磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000061
之间增加一虚拟电压注入模块，从而实现了：
(1)在不对电机本体进行信号注入的情况下，实现感应电机无速度传感器驱动系统控制的感应电机在零同步转速或低同步转速下均能够输出150％电机额定转矩。
(2)在不对电机本体进行信号注入的情况下，实现感应电机无速度传感器驱动系统控制的感应电机在0％电机额定转矩和零转子转速下长时间 稳定运行，并能够在长时间运行后正常加速启动。
(3)在不对电机本体进行信号注入的情况下，实现感应电机无速度传感器驱动系统控制的感应电机在负载为150％电机额定转矩不变的情况下，以任意加减速时间进行电机转速的正反转切换。
附图说明
图1为本发明实施例一提供的基于虚拟电压注入的感应电机无速度传感器驱动控制系统结构示意图。
图2为本发明实施例二提供的基于虚拟电压注入的感应电机无速度传感器驱动控制系统结构示意图。
图3为本发明实施例提供的不同阶段感应电机转子转速性能曲线图。
图4为本发明实施例提供的电机U相定子电流、电机转子转速指令值随时间变化的曲线图。
【具体实施方式】
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅仅用以解释本发明，并不用于限定本发明。
图1为本发明实施例一提供的基于虚拟电压注入的感应电机无速度传感器驱动控制系统结构示意图。
感应电机无速度传感器驱动系统的硬件部分包括：三相电压源型逆变器和感应电机。三相交流电源经过不控整流得到直流母线电压U DC，供给电压源型逆变器，并利用逆变器来控制感应电机进行转矩和转速的控制。三相电压源型逆变器中包括电压和电流传感器。
感应电机无速度传感器驱动系统的软件部分包括：3相静止坐标/2相静止坐标变换模块，2相同步旋转坐标/2相静止坐标变换模块，电压空间矢量脉宽调制模块，电流PI(Proportion Integration)调节模块，转速 PI调节模块，磁链电流指令给定模块，转速指令给定模块，磁链和转速观测器模块和虚拟电压注入模块。
感应电机的控制方式主要分为VF控制、矢量控制和直接转矩控制，本发明实施例优选矢量控制策略。本发明主要涉及虚拟电压注入模块，其他模块为感应电机无速度传感器矢量控制的功能性模块，为本领域公知常识。虚拟电压注入模块通过加法器、乘法器或其组合实现。
实施例一控制系统通过在αβ坐标系下电机的定子电压指令输入值u sα、u sβ和磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000062
之间增加一虚拟电压注入模块实现，下面介绍整个系统的控制方法，包括步骤S1至步骤S5。
S1.基于感应电机的参数计算k，k为虚拟电压注入模块中的比例关系，计算公式如下：
Figure PCTCN2018088315-appb-000063
其中，p为大于零的常数，根据感应电机转速和转矩的稳定程度来取值；R r为感应电机转子电阻；L m为感应电机互感；L r为感应电机转子侧电感。
比例关系k被计算出来后，在电机控制器中以常数形式存在，其值不随电机各参数的变化而变化。
S2.将αβ坐标系下电机的定子电压指令输入值u sα、u sβ分别乘以比例关系k，获得αβ坐标系下磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000064
该操作相当于在u sα和u sβ的基础上，注入了u sα_inj和u sβ_inj，其中
Figure PCTCN2018088315-appb-000065
使得满足
Figure PCTCN2018088315-appb-000066
式中，u sα_inj为α轴下的虚拟电压注入值，u sβ_inj为β轴下的虚拟电压注入值；
S3.基于
Figure PCTCN2018088315-appb-000067
构建磁链观测器的动态数学模型，动态数学模型如下：
Figure PCTCN2018088315-appb-000068
其中：
Figure PCTCN2018088315-appb-000069
Figure PCTCN2018088315-appb-000070
Figure PCTCN2018088315-appb-000071
Figure PCTCN2018088315-appb-000072
其中，
Figure PCTCN2018088315-appb-000073
分别为α轴、β轴下的定子磁链观测值；
Figure PCTCN2018088315-appb-000074
分别为α轴、β轴下的转子磁链观测值；
Figure PCTCN2018088315-appb-000075
分别为α轴、β轴下的定子电流观测值；R s、R r分别为电机定子电阻和转子电阻；L s、L r、L m分别为电机定子侧电感、电机转子侧电感和电机互感。
S4.基于磁链观测器的动态数学模型，使用转速观测器观测感应电机转子转速
Figure PCTCN2018088315-appb-000076
使用磁链观测器观测转子磁链的旋转角度
Figure PCTCN2018088315-appb-000077
Figure PCTCN2018088315-appb-000078
Figure PCTCN2018088315-appb-000079
其中，k p、k i分别为转速观测器的比例环节增益和积分环节增益；i sα、i sβ分别为α轴、β轴下定子电流实际测量值；
Figure PCTCN2018088315-appb-000080
分别为α轴、β轴下定子电流观测值；
Figure PCTCN2018088315-appb-000081
分别为α轴、β轴下的转子磁链观测值；S 1分别为
Figure PCTCN2018088315-appb-000082
的时间积分值。
S5.观测出的转子转速
Figure PCTCN2018088315-appb-000083
被用于转速PI调节模块和磁链观测器，观测出的转子磁链旋转角度
Figure PCTCN2018088315-appb-000084
用于2相同步旋转坐标/2相静止坐标变换模块，从而实现感应电机无速度传感器转速和转矩的控制。
具体地，步骤S5包括以下步骤：
S501，观测出的感应电机转子转速
Figure PCTCN2018088315-appb-000085
作为转速PI调节模块的反馈量，与对应的转速指令
Figure PCTCN2018088315-appb-000086
做差后进行转速PI控制；
S502，观测出的磁链旋转角度
Figure PCTCN2018088315-appb-000087
被用于2相同步旋转坐标/2相静止坐标变换模块中的坐标变换计算；
S503，转速PI调节模块的输出
Figure PCTCN2018088315-appb-000088
作为q轴电流PI调节模块的指令，磁链电流指令给定模块的输出
Figure PCTCN2018088315-appb-000089
作为d轴电流PI调节模块的指令；由电流传感器采样得到的感应电机两相电流i U、i V，输入3相静止坐标/2相静止坐标变换模块，然后输出
Figure PCTCN2018088315-appb-000090
至2相同步旋转坐标/2相静止坐标变换模块，最后得到2相同步旋转坐标系下的d轴电流i sd和q轴电流i sq，并将得到的d轴电流和q轴电流分别作为d轴电流PI调节器和q轴电流PI调节器的反馈量，与对应的磁链电流指令
Figure PCTCN2018088315-appb-000091
和
Figure PCTCN2018088315-appb-000092
做差后进行电流PI控制；
S504，d轴和q轴电流PI调节模块的输出u sd和u sq至2相同步旋转坐标/2相静止坐标变换模块，其将2相同步旋转坐标系下的电机输入电压指令转换为2相静止坐标系下的电机输入电压指令
Figure PCTCN2018088315-appb-000093
S505，将
Figure PCTCN2018088315-appb-000094
输出至电压空间矢量脉宽调制模块，产生能够控制开关器件S A、S B、S C的开关信号，进而达到控制感应电机转速和转矩的目的。
如图2所示，实施例二控制系统通过在在dq坐标系下电机的定子电压指令输入值u sd、u sq和磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000095
之间增加一虚拟电压注入模块实现，下面介绍整个系统的控制方法，包括步骤S1至步骤S5。
S1.基于感应电机的参数计算k，k为虚拟电压注入模块中的比例关系，计算公式如下：
Figure PCTCN2018088315-appb-000096
S2.将dq坐标系下电机的定子电压指令输入值u sd、u sq分别乘以比例关系k，获得dq坐标系下磁链观测器定子电压输入值
Figure PCTCN2018088315-appb-000097
该操作相当于在u sd和u sq的基础上，注入了u sd_inj和u sq_inj，其中
Figure PCTCN2018088315-appb-000098
使得满足
Figure PCTCN2018088315-appb-000099
式中，u sd_inj为d轴下的虚拟电压注入值，u sq_inj为q轴下的虚拟电压注入值；
S3.基于
Figure PCTCN2018088315-appb-000100
构建磁链观测器的动态数学模型；，动态数学模型如下：
Figure PCTCN2018088315-appb-000101
其中：
Figure PCTCN2018088315-appb-000102
Figure PCTCN2018088315-appb-000103
Figure PCTCN2018088315-appb-000104
Figure PCTCN2018088315-appb-000105
其中，
Figure PCTCN2018088315-appb-000106
分别为d轴、q轴下的定子磁链观测值；
Figure PCTCN2018088315-appb-000107
分别为d轴、q轴下的转子磁链观测值；
Figure PCTCN2018088315-appb-000108
分别为d轴、q轴下的定子电流 观测值；ω e为同步转速；R s、R r分别为电机定子电阻和转子电阻；L s、L r、L m分别为电机定子侧电感、电机转子侧电感和电机互感。
S4.基于磁链观测器的动态数学模型，使用转速观测器观测感应电机转子转速
Figure PCTCN2018088315-appb-000109
使用磁链观测器观测转子磁链的旋转角度
Figure PCTCN2018088315-appb-000110
Figure PCTCN2018088315-appb-000111
Figure PCTCN2018088315-appb-000112
Figure PCTCN2018088315-appb-000113
其中，k p、k i分别为转速观测器的比例环节增益和积分环节增益；i sd、i sq分别为d轴、q轴下定子电流实际测量值；
Figure PCTCN2018088315-appb-000114
分别为d轴、q轴下定子电流观测值；
Figure PCTCN2018088315-appb-000115
分别为d轴、q轴下的转子磁链观测值；S 2为
Figure PCTCN2018088315-appb-000116
的时间积分值，i sq为q轴下定子电流实际测量值，ω s为滑差转速，R r为电机转子电阻，L r、L m分别为电机转子侧电感和电机互感，S 3代表对
Figure PCTCN2018088315-appb-000117
的时间积分。
S5.观测出的转子转速
Figure PCTCN2018088315-appb-000118
被用于转速PI调节模块和磁链观测器，观测出的转子磁链旋转角度
Figure PCTCN2018088315-appb-000119
用于2相同步旋转坐标/2相静止坐标变换模块，从而实现感应电机无速度传感器转速和转矩的控制。
S501，观测出的感应电机转子转速
Figure PCTCN2018088315-appb-000120
作为转速PI调节模块的反馈量，与对应的转速指令
Figure PCTCN2018088315-appb-000121
做差后进行转速PI控制；
S502，观测出的磁链旋转角度
Figure PCTCN2018088315-appb-000122
被用于2相同步旋转坐标/2相静止坐标变换模块中的坐标变换计算；
S503，转速PI调节模块的输出
Figure PCTCN2018088315-appb-000123
作为q轴电流PI调节模块的指令，磁链电流指令给定模块的输出
Figure PCTCN2018088315-appb-000124
作为d轴电流PI调节模块的指令；由电流传 感器采样得到的感应电机两相电流i U、i V，输入3相静止坐标/2相静止坐标变换模块，然后输出
Figure PCTCN2018088315-appb-000125
至2相同步旋转坐标/2相静止坐标变换模块，最后得到2相同步旋转坐标系下的d轴电流i sd和q轴电流i sq，并将得到的d轴电流和q轴电流分别作为d轴电流PI调节器和q轴电流PI调节器的反馈量，与对应的磁链电流指令
Figure PCTCN2018088315-appb-000126
和
Figure PCTCN2018088315-appb-000127
做差后进行电流PI控制；
S504，d轴和q轴电流PI调节模块的输出u sd和u sq至2相同步旋转坐标/2相静止坐标变换模块，其将2相同步旋转坐标系下的电机输入电压指令转换为2相静止坐标系下的电机输入电压指令
Figure PCTCN2018088315-appb-000128
S505，将
Figure PCTCN2018088315-appb-000129
输出至电压空间矢量脉宽调制模块，产生能够控制开关器件S A、S B、S C的开关信号，进而达到控制感应电机转速和转矩的目的。
图3为本发明实施例提供的不同阶段感应电机转子转速性能曲线图。
如图3所示，在电机零同步转速并且0％负载转矩、零同步转速并且150％负载转矩和低同步转速并且150％负载转矩下电机转子转速能够保持稳定。
图4为本发明实施例提供的电机U相定子电流、电机转子转速值随时间变化的曲线图。
如图4所示，在150％负载转矩下，电机转子转速由-120rpm穿越至120rpm时，能够保持稳定。
以上，仅为本申请较佳的具体实施方式，但本申请的保护范围并不局限于此，任何熟悉本技术领域的技术人员在本申请揭露的技术范围内，可轻易想到的变化或替换，都应涵盖在本申请的保护范围之内。因此，本申请的保护范围应该以权利要求的保护范围为准。

Claims (8)

  1. 基于虚拟电压注入的感应电机无速度传感器驱动控制方法，该方法基于现有的感应电机无速度传感器驱动系统，通过在αβ坐标系下电机的定子电压指令输入值u sα、u sβ和磁链观测器定子电压输入值
    Figure PCTCN2018088315-appb-100001
    之间增加一虚拟电压注入模块，或者，在dq坐标系下电机的定子电压指令输入值u sd、u sq和磁链观测器定子电压输入值
    Figure PCTCN2018088315-appb-100002
    之间增加一虚拟电压注入模块实现，其特征在于，该方法包括如下步骤：
    S1.基于感应电机的参数计算k，k为虚拟电压注入模块中的比例关系；
    S2.将αβ坐标系下电机的定子电压指令输入值u sα、u sβ分别乘以比例关系k，获得αβ坐标系下磁链观测器定子电压输入值
    Figure PCTCN2018088315-appb-100003
    或者，将dq坐标系下电机的定子电压指令输入值u sd、u sq分别乘以比例关系k，获得dq坐标系下磁链观测器定子电压输入值
    Figure PCTCN2018088315-appb-100004
    该操作相当于在u sα和u sβ的基础上，注入了u sα_inj和u sβ_inj，其中
    Figure PCTCN2018088315-appb-100005
    使得满足
    Figure PCTCN2018088315-appb-100006
    式中，u sα_inj为α轴下的虚拟电压注入值，u sβ_inj为β轴下的虚拟电压注入值；
    或者，相当于在u sd和u sq的基础上，注入了u sd_inj和u sq_inj，其中
    Figure PCTCN2018088315-appb-100007
    使得满足
    Figure PCTCN2018088315-appb-100008
    式中，u sd_inj为d轴下的虚拟电压注入值，u sq_inj为q轴下的虚拟电压注入值；
    S3.基于
    Figure PCTCN2018088315-appb-100009
    或
    Figure PCTCN2018088315-appb-100010
    构建磁链观测器的动态数学模型；
    S4.基于磁链观测器的动态数学模型，使用转速观测器观测感应电机转子转速
    Figure PCTCN2018088315-appb-100011
    使用磁链观测器观测转子磁链的旋转角度
    Figure PCTCN2018088315-appb-100012
    S5.观测出的转子转速
    Figure PCTCN2018088315-appb-100013
    被用于转速PI调节模块和磁链观测器，观测出的转子磁链旋转角度
    Figure PCTCN2018088315-appb-100014
    用于2相同步旋转坐标/2相静止坐标变换模块，从而实现感应电机无速度传感器转速和转矩的控制；
    其中，αβ坐标系是2相静止坐标系，dq坐标系是2相同步旋转坐标系。
  2. 如权利要求1所述的驱动控制方法，其特征在于，虚拟电压注入模块通过加法器、乘法器或其组合实现。
  3. 如权利要求1或2所述的驱动控制方法，其特征在于，步骤S1中比例关系k的计算公式如下：
    Figure PCTCN2018088315-appb-100015
    其中，p为大于零的常数，根据感应电机转速和转矩的稳定程度来取值；R r为感应电机转子电阻；L m为感应电机互感；L r为感应电机转子侧电感。
  4. 如权利要求1或2所述的驱动控制方法，其特征在于，步骤S3中基于
    Figure PCTCN2018088315-appb-100016
    构建的磁链观测器的动态数学模型如下：
    Figure PCTCN2018088315-appb-100017
    其中：
    Figure PCTCN2018088315-appb-100018
    Figure PCTCN2018088315-appb-100019
    Figure PCTCN2018088315-appb-100020
    Figure PCTCN2018088315-appb-100021
    基于
    Figure PCTCN2018088315-appb-100022
    构建的磁链观测器的动态数学模型如下：
    Figure PCTCN2018088315-appb-100023
    其中：
    Figure PCTCN2018088315-appb-100024
    Figure PCTCN2018088315-appb-100025
    Figure PCTCN2018088315-appb-100026
    Figure PCTCN2018088315-appb-100027
    其中，
    Figure PCTCN2018088315-appb-100028
    分别为α轴、β轴、d轴、q轴下的定子磁链观测值；
    Figure PCTCN2018088315-appb-100029
    分别为α轴、β轴、d轴、q轴下的转子磁链观测值；
    Figure PCTCN2018088315-appb-100030
    分别为α轴、β轴、d轴、q轴下的定子电流观测值；ω e为同步转速；R s、R r分别为电机定子电阻和转子电阻；L s、L r、L m分别为电机定子侧电感、电机转子侧电感和电机互感。
  5. 如权利要求1或2所述的驱动控制方法，其特征在于，基于
    Figure PCTCN2018088315-appb-100031
    构建磁链观测器的动态数学模型时，步骤S4中感应电机转子转速
    Figure PCTCN2018088315-appb-100032
    计算公式如下：
    Figure PCTCN2018088315-appb-100033
    基于
    Figure PCTCN2018088315-appb-100034
    构建磁链观测器的动态数学模型时，步骤S4中感应电机转 子转速
    Figure PCTCN2018088315-appb-100035
    计算公式如下：
    Figure PCTCN2018088315-appb-100036
    其中，k p、k i分别为转速观测器的比例环节增益和积分环节增益；i sα、i sβ、i sd、i sq分别为α轴、β轴、d轴、q轴下定子电流实际测量值；
    Figure PCTCN2018088315-appb-100037
    Figure PCTCN2018088315-appb-100038
    分别为α轴、β轴、d轴、q轴下定子电流观测值；
    Figure PCTCN2018088315-appb-100039
    Figure PCTCN2018088315-appb-100040
    分别为α轴、β轴、d轴、q轴下的转子磁链观测值；S 1、S 2分别为
    Figure PCTCN2018088315-appb-100041
    的时间积分值。
  6. 如权利要求1或2所述的驱动控制方法，其特征在于，基于
    Figure PCTCN2018088315-appb-100042
    构建磁链观测器的动态数学模型时，步骤S4中旋转角度
    Figure PCTCN2018088315-appb-100043
    计算公式如下：
    Figure PCTCN2018088315-appb-100044
    基于
    Figure PCTCN2018088315-appb-100045
    构建磁链观测器的动态数学模型时，步骤S4中旋转角度
    Figure PCTCN2018088315-appb-100046
    计算公式如下：
    Figure PCTCN2018088315-appb-100047
    Figure PCTCN2018088315-appb-100048
    其中，
    Figure PCTCN2018088315-appb-100049
    分别为α轴、β轴、d轴下的转子磁链观测值；i sq为q轴下定子电流实际测量值，ω s为滑差转速，R r为电机转子电阻，L r、L m分别为电机转子侧电感和电机互感，S 3代表对
    Figure PCTCN2018088315-appb-100050
    的时间积分。
  7. 如权利要求1或2所述的驱动控制方法，其特征在于，步骤S5包括以下步骤：
    S501，观测出的感应电机转子转速
    Figure PCTCN2018088315-appb-100051
    作为转速PI调节模块的反馈量，与对应的转速指令
    Figure PCTCN2018088315-appb-100052
    做差后进行转速PI控制；
    S502，观测出的磁链旋转角度
    Figure PCTCN2018088315-appb-100053
    被用于2相同步旋转坐标/2相静止坐标变换模块中的坐标变换计算；
    S503，转速PI调节模块的输出
    Figure PCTCN2018088315-appb-100054
    作为q轴电流PI调节模块的指令，磁链电流指令给定模块的输出
    Figure PCTCN2018088315-appb-100055
    作为d轴电流PI调节模块的指令；由电流传感器采样得到的感应电机两相电流i U、i V，输入3相静止坐标/2相静止坐标变换模块，然后输出
    Figure PCTCN2018088315-appb-100056
    至2相同步旋转坐标/2相静止坐标变换模块，最后得到2相同步旋转坐标系下的d轴电流i sd和q轴电流i sq，并将得到的d轴电流和q轴电流分别作为d轴电流PI调节器和q轴电流PI调节器的反馈量，与对应的磁链电流指令
    Figure PCTCN2018088315-appb-100057
    和
    Figure PCTCN2018088315-appb-100058
    做差后进行电流PI控制；
    S504，d轴和q轴电流PI调节模块的输出u sd和u sq至2相同步旋转坐标/2相静止坐标变换模块，其将2相同步旋转坐标系下的电机输入电压指令转换为2相静止坐标系下的电机输入电压指令
    Figure PCTCN2018088315-appb-100059
    S505，将
    Figure PCTCN2018088315-appb-100060
    输出至电压空间矢量脉宽调制模块，产生能够控制开关器件S A、S B、S C的开关信号，进而达到控制感应电机转速和转矩的目的。
  8. 一种计算机可读存储介质，其特征在于，所述计算机可读存储介质上存储有计算机程序，所述计算机程序被处理器执行时实现如权利要求1至7任一项所述的基于虚拟电压注入的感应电机无速度传感器驱动控制方法。
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CN115001342A (zh) * 2022-07-08 2022-09-02 中国长江三峡集团有限公司 感应电机的转速估算方法、装置及系统
CN115001342B (zh) * 2022-07-08 2023-07-21 中国长江三峡集团有限公司 感应电机的转速估算方法、装置及系统
CN116846281A (zh) * 2023-07-14 2023-10-03 郑州轻工业大学 基于带补偿磁链观测器的异步电机无速度传感器控制方法
CN116846281B (zh) * 2023-07-14 2024-06-04 郑州轻工业大学 基于带补偿磁链观测器的异步电机无速度传感器控制方法

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