WO2019192234A1 - 一种零电压开关Boost电路及其控制方法 - Google Patents

一种零电压开关Boost电路及其控制方法 Download PDF

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Publication number
WO2019192234A1
WO2019192234A1 PCT/CN2019/070642 CN2019070642W WO2019192234A1 WO 2019192234 A1 WO2019192234 A1 WO 2019192234A1 CN 2019070642 W CN2019070642 W CN 2019070642W WO 2019192234 A1 WO2019192234 A1 WO 2019192234A1
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Prior art keywords
circuit
zero voltage
switch
boost
inductor
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PCT/CN2019/070642
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English (en)
French (fr)
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袁源
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广州金升阳科技有限公司
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H02M1/342—Active non-dissipative snubbers
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a power electronic circuit, in particular to a zero voltage switch Boost circuit and a control method thereof.
  • FIG. 1 A conventional power supply block diagram is shown in FIG. 1.
  • the bridge rectifier circuit 11 converts an AC input into a DC voltage supply Boost converter 20, which provides a power factor correction function to meet industry standards or simply to fluctuate
  • the large voltage is converted to a stable voltage or a voltage range having a smaller variation range
  • the DC/DC converter of the subsequent stage converts the output voltage of the Boost converter to the voltage required by the load and performs isolation.
  • the AC input range is 85-265VAC
  • the Boost converter has a regulation voltage of 400V or the regulation range is 200-400V.
  • the narrower the output voltage range of the front-stage Boost circuit the simpler the design of the rear-stage DC/DC converter and the better the performance that can be achieved.
  • a conventional Boost converter 20 is shown in FIG. 2.
  • the main switch 22 enters an on state under the action of the control circuit 26, one end of the inductor 21 is shorted to ground through the main switch 22, and the inductor 21 The other end is connected to the power supply, so the input voltage Vin will cause the current in the inductor 21 to rise.
  • the rectifier diode 23 is reverse biased to be in an off state, and the output filter capacitor 24 supplies power to the load.
  • the main switch 22 is turned off by the control circuit 26, the inductor current cannot be abruptly changed, and the energy stored in the inductor 21 during the on-time of the main switch 22 can be supplied to the load through the rectifier diode 23.
  • the output filter capacitor 24 maintains the output voltage substantially constant.
  • the on-time and off-time of the main switch 22 are determined by the control circuit 26 to ensure that the output voltage is a set voltage value.
  • a conventional Boost circuit for implementing ZVS is a synchronous rectification Boost circuit as shown in FIG. 6-1, which adopts PWM control, and its working timing waveform is shown in FIG. 6-2, and the output voltage and the synchronous rectifier diode are used to realize Boost.
  • the negative excitation of the boost inductor uses the negative inductor current to implement the main switch ZVS.
  • the condition for the operation is that the circuit operates in DCM mode, that is, the inductor current crosses zero.
  • the circuit is suitable for occasions with low power, and the negative current is obviously increased at light load, and the light load efficiency is not high.
  • FIG. 7 Another conventional control mode synchronous rectification Boost converter and its control block diagram are shown in Figure 7.
  • the negative current of the boost boost inductor, the input voltage Vin and the output voltage Vout are detected for control, wherein Vout is detected to control
  • the output voltage is stable, the Vin is detected to set the minimum value of the boost inductor negative current, and the boost inductor negative current is detected to control the turn-off of the synchronous rectifier diode for PFM control.
  • the working frequency is higher at light load and the light load efficiency is lower; the operating frequency is higher when the high voltage is light load, and the light load efficiency is lower.
  • the present invention provides a Boost circuit for zero voltage switching and a control method thereof, to solve the problem of excessive loss of high frequency operation switching, to solve the reverse recovery problem of the inductor current continuous rectifier diode, and also to solve the hard switching.
  • the resulting EMI problem at the same time, in order to improve the light load efficiency, the present invention also provides a light load control method to further improve the overall efficiency of the Boost converter. And the control method automatically adapts to the DCM (current interrupt mode) and CCM (current continuous mode) operating modes.
  • a zero voltage switch Boost circuit comprising a boost inductor, a main switch tube, a rectifier diode, an output filter capacitor and a control circuit, and a zero voltage switch circuit;
  • One end of the boost inductor is connected to the input voltage +, the other end of the boost inductor is connected to the drain of the main switch, the source of the main switch is connected to the input voltage - the gate of the main switch is connected to one output of the control circuit;
  • the output voltage Vout of the output filter capacitor is supplied to the subsequent stage load;
  • the control circuit generates a feedback voltage signal according to the output voltage Vout and adjusts the duty ratio of the main switch according to the feedback voltage signal;
  • the input end of the zero voltage switch circuit Connected to the other end of the boost inductor, the output of the zero voltage switching circuit is connected to the anode of the rectifier diode, and the control terminal of the zero voltage switching circuit is connected to the other output of the control circuit;
  • the rectifier diode The cathode is connected to the positive terminal of the output filter capacitor.
  • the zero voltage switch Boost circuit further includes a CS current detecting circuit, wherein the input end of the CS current detecting circuit is connected to the source of the main switching tube for detecting the current of the source of the main switching tube; the CS current detecting The output of the circuit is connected to the control circuit.
  • the zero voltage switching circuit comprises a resonant inductor, an auxiliary switching tube, a resonant capacitor, one end of the resonant inductor is used as an input end of the zero voltage switching circuit, and the other end of the resonant inductor is used as an output end of the zero voltage switching circuit;
  • the source of the tube is connected to one end of the resonant inductor, the drain of the auxiliary switch is connected to the anode of the resonant capacitor, and the cathode of the resonant capacitor is connected to the other end of the resonant inductor;
  • the gate of the auxiliary switch is used as the control terminal of the zero voltage switching circuit.
  • the zero voltage switching circuit further comprises an auxiliary diode, the cathode of the auxiliary diode is connected to the anode of the resonant capacitor, and the anode of the auxiliary diode is connected to the cathode of the resonant capacitor.
  • control circuit controls the main switch tube and the auxiliary switch tube to be complementary drive control.
  • the main switch tube and the auxiliary switch tube are MOS tubes or IGBTs, and the auxiliary diodes are Schottky diodes.
  • a zero voltage switch Boost circuit includes a boost inductor, a main switch tube, an output filter capacitor, a synchronous rectifier switch tube, and a zero voltage switch circuit;
  • One end of the boost inductor is connected to the input voltage +, the other end of the boost inductor is connected to the drain of the main switch, the source of the main switch is connected to the input voltage - the gate of the main switch is connected to one output of the control circuit;
  • the output voltage Vout of the output filter capacitor is supplied to the subsequent stage load;
  • the control circuit generates a feedback voltage signal according to the output voltage Vout and adjusts the duty ratio of the main switch according to the feedback voltage signal;
  • the input end of the zero voltage switch circuit Connected to the drain of the main switch, the output of the zero voltage switch circuit is connected to the source of the synchronous rectifier switch, and the control end of the zero voltage switch circuit is connected to the other output of the control circuit;
  • the drain of the rectifier switch is connected to the anode of the output filter capacitor, and the gate of the synchronous rectifier switch is connected to the third output of the control circuit.
  • the zero voltage switching circuit comprises a resonant inductor, an auxiliary switching tube, a resonant capacitor, one end of the resonant inductor is used as an input end of the zero voltage switching circuit, and the other end of the resonant inductor is used as an output end of the zero voltage switching circuit;
  • the source of the tube is connected to one end of the resonant inductor, the drain of the auxiliary switch is connected to the anode of the resonant capacitor, and the cathode of the resonant capacitor is connected to the other end of the resonant inductor;
  • the gate of the auxiliary switch is used as the control terminal of the zero voltage switching circuit.
  • the main switch tube and the auxiliary switch tube are complementarily driven, and the auxiliary switch tube and the synchronous rectifier switch tube are synchronously driven.
  • the zero voltage switching circuit further comprises an auxiliary diode, the cathode of the auxiliary diode is connected to the anode of the resonant capacitor, and the anode of the auxiliary diode is connected to the cathode of the resonant capacitor.
  • the object of the present invention is achieved by integrating an auxiliary resonant circuit in a rectifier circuit of a main power circuit.
  • the zero voltage of the auxiliary switch is realized by the current of the boost boost inductor.
  • a control method for a zero voltage switch Boost circuit the CS current detecting circuit detects a peak current of the main switch tube, and keeps the minimum peak current from being controlled from decreasing when the peak current is reduced to a set reference value, When the input voltage rises or the load continues to decrease, causing the main switch tube peak current to continue to decrease, the output of the Boost circuit is stabilized by lowering the operating frequency of the main switch; and when the operating frequency of the main switch reaches the minimum operating frequency Then enter the frequency hopping work.
  • the zero-voltage Boost circuit of the present invention can simply implement complementary zero-voltage turn-on of the main switch and the auxiliary switch by using complementary driving.
  • this simple control implementation only the output voltage is controlled to be controlled.
  • the on-time of the main switch can be.
  • the voltage stress before the auxiliary switch tube is turned on is very small, so the auxiliary switch tube can be turned on immediately after the main tube is turned off, that is, the dead time can be short; but the voltage stress before the main switch tube is turned on is Vout, which needs It takes a certain time to draw zero, so it is necessary to leave a proper dead time between the auxiliary switch and the main switch.
  • the present invention proposes a method for implementing frequency reduction control on the basis of complementary control, sampling the main on the basis of the sampling output voltage.
  • the peak current of the switch tube when the load decreases, the peak current of the main switch tube decreases, and when the load decreases to a certain extent, the circuit enters the DCM mode.
  • the minimum peak current control that is, the output of the converter is stabilized by reducing the operating frequency of the switching tube when the load continues to decrease, and When the working frequency reaches the minimum operating frequency, it enters the frequency hopping mode.
  • This control can ensure the light load and frequency reduction while still achieving the ZVS (zero voltage turn-on) of the main switch and the auxiliary switch, which is beneficial to the module. Light load efficiency and EMI performance.
  • the present invention has the following beneficial effects:
  • the ZVS operation is not limited by the working mode, and the ZVS of the main switch tube and the auxiliary switch tube can be realized in both the CCM mode and the DCM mode;
  • the implementation of ZVS does not affect the current of the Boost boost inductor.
  • the current of the Boost boost inductor does not cross zero and can be used for PFC control.
  • the Boost boost inductor When applied to the synchronous rectification Boost circuit, the Boost boost inductor can achieve zero voltage turn-on of the main switch without the need of negative excitation.
  • Figure 1 is a conventional power supply block diagram
  • FIG. 2 is a schematic diagram of a conventional diode rectified Boost converter
  • FIG. 3 is a schematic diagram and a control block diagram of a circuit according to Embodiment 1 of the present invention.
  • Figure 4 is a working modal view of Embodiment 1 of the present invention.
  • Figure 5 is an operation waveform of Embodiment 1 according to the present invention.
  • Figure 6-1 is a conventional synchronous rectification Boost converter and its control block diagram
  • Figure 6-2 is a control timing diagram of the synchronous rectification Boost converter shown in Figure 6-1;
  • 8-1 is a schematic diagram and a control block diagram of a circuit according to Embodiment 2 of the present invention.
  • FIG. 8-2 is a schematic diagram of a frequency change curve according to Embodiment 2 of the present invention.
  • Embodiment 3 of the present invention is a schematic circuit diagram of Embodiment 3 of the present invention.
  • Figure 10 is a schematic diagram of the circuit of Embodiment 4 of the present invention.
  • the zero voltage switch Boost circuit 30 includes a boost inductor 31, a main switch 32, a rectifier diode 33, and an output.
  • the filter capacitor 34 and the control circuit 36 output the voltage Vout at both ends of the output filter capacitor 34 to supply power to the load 35.
  • the control circuit 36 generates a feedback voltage signal according to the output voltage Vout and adjusts the duty ratio of the main switch 32 according to the feedback voltage signal.
  • the zero voltage switch Boost circuit 30 further includes a zero voltage switching circuit 40.
  • the input end of the zero voltage switching circuit 40 is connected to the other end of the boost inductor 31, and the output terminal and the rectification
  • the anode of diode 33 is connected and the control terminal is connected to the other output of the control circuit.
  • the zero-voltage switching circuit 40 includes a resonant inductor 41, an auxiliary switching transistor 42, an auxiliary diode 43, and a resonant capacitor 44.
  • One end of the resonant inductor 41 is connected to the other end of the boosting inductor 31, and the other end of the resonant inductor 41 is connected to the rectifier diode 33.
  • the cathode of the rectifier diode 33 is connected to the anode of the output filter capacitor 34;
  • the source of the auxiliary switch 42 is connected to one end of the resonant inductor 41, and the drain of the auxiliary switch 42 is connected to the anode of the resonant capacitor 44 and the cathode of the auxiliary diode 43 respectively.
  • the cathode of the resonant capacitor 44 is connected to the anode of the auxiliary diode 43 and to the other end of the resonant inductor 41.
  • Ds1 and Cs1 are the parasitic diodes and parasitic capacitances of the main switch 32, respectively, which are not present in the actual circuit; likewise, Ds2 and Cs2 are the parasitic diodes and parasitic capacitances of the auxiliary switch 42 respectively.
  • the main switch tube 32 and the auxiliary switch tube 42 are fully controlled semiconductor switches;
  • the main switch tube 32 and the auxiliary switch tube 42 are the MOS tubes shown in FIG. 3;
  • the main switch tube 32 and the auxiliary switch tube 42 are IGBTs;
  • the main switch tube 32 and the auxiliary switch tube 42 are SiC MOS tubes or GaN MOS tubes;
  • the auxiliary diode 43 is a Schottky diode.
  • FIG. 4 is a main operation mode in the switching operation process according to Embodiment 1 of the present invention
  • FIG. 5 is a main waveform in the switching operation process according to Embodiment 1 of the present invention, and the working waveform is briefly described in comparison with the working mode.
  • Mode1 (t0-t1): At time t0, the auxiliary switch tube 42 is in the off-off state, and the main switch tube 32 is switched from the on state to the off-state. Since the inductor current cannot be abruptly changed, the resonant inductor 41 current can be considered as a short time. Unchanged, a part of the current of the boosting inductor 31 charges the drain-source junction capacitance Cs1 of the main switching transistor 32, causing Vds1 to rise rapidly; another part of the current discharges the drain-source junction capacitance Cs2 of the auxiliary switching transistor 42, resulting in Vds2 quickly drops to 0;
  • Mode2(t1-t2) At time t1, the drain-source junction capacitance Cs2 of the auxiliary switching transistor 42 is discharged to 0V, the auxiliary diode 42 body diode Ds2 is turned on, and a part of the current of the boosting inductor 31 continues.
  • the drain-source junction capacitance Cs1 of the main switching transistor 32 is charged, and the other portion is charged to the resonant capacitor 44 (Cr) through the body diode Ds2 of the auxiliary switching transistor 42 while the voltage of the resonant capacitor 44 is VCr
  • the resonant inductor is excited across the resonant inductor 41, that is, the drain-source junction capacitance Cs1 of the main switch 32 is connected in parallel with the resonant capacitor 44 (Cr) to resonate with the resonant inductor 41;
  • Mode3 (t2-t3): At time t2, the auxiliary switch tube 42 is turned on at zero voltage, and does not affect the resonance process that is going on at this time. At time t3, the resonant capacitor 44 (Cr) voltage resonates to 0 V, and the resonance The current of the inductor 41 reaches a maximum value;
  • Mode 4 (t3-t4): during this period of time, the auxiliary diode 43 is turned on, and the current of the resonant inductor 41 continues to flow through the auxiliary diode 43 and the auxiliary switch tube 42 and remains substantially unchanged;
  • Mode5(t4-t5) At time t4, the auxiliary switch tube 42 is turned off, a part of the current of the resonant inductor 41 charges the drain-source junction capacitor Cs2 of the auxiliary switch tube 42, and the other part is drained to the main switch tube 32.
  • the source junction capacitor Cs1 is discharged, that is, the drain-source junction capacitor Cs1 of the main switch transistor 32 is connected in parallel with the drain-source junction capacitor Cs2 of the auxiliary switch transistor 42 to resonate with the resonant inductor 41;
  • Mode6 (t5-t6): At time t5, the drain-source voltage Vds2 of the auxiliary switching transistor 42 rises to Vout, the drain-source voltage Vds1 of the main switching transistor 32 is reduced to 0V, and the body diode Ds1 of the main switching transistor 32 is turned on.
  • the voltage applied across Lr is -Vout, iLr decreases linearly, and the voltage applied across Lp is Vin, and iLp increases linearly;
  • Mode7 (t6-t7) t6 the main switch 32 zero voltage conduction, does not affect the previous working process, iLr continues to linearly decrease, iLp continues to increase linearly, until iLp> iLr, the current flowing through the main switch 32 becomes a positive current;
  • Mode8 (t7-t8): At time t7, the current of the resonant inductor 41 linearly decreases to 0, the rectifier diode 33 has a zero current turn-off, and the auxiliary switch transistor 42 drain-source junction capacitance Cs2 is connected in series with the resonant capacitor 44 (Cr) and Lr Resonance, at time t8, the drain-source voltage Vds2 of the auxiliary switching transistor 42 resonates to 0V, and the resonant inductor 41 current reaches a negative minimum value;
  • Mode9 (t8-t9): At time t8, the auxiliary diode 42 body diode Ds2 is turned on, and the resonant capacitor 44 (Cr) and Lr are in series resonance. At time t9, the current of the resonant inductor 41 resonates to 0, and the resonant capacitor voltage reaches one. Relatively stable value;
  • Mode10(t9-t10) After the resonant capacitor voltage reaches a relatively stable value, Lr will resonate slightly with the drain-source junction capacitor Cs2 of the auxiliary switching transistor 42, which is ignored here and does not appear in the typical operating waveform; t10 The main switch 32 is switched from the on state to the off state again, starting another cycle.
  • FIG. 8-1 shows a schematic diagram and a control block diagram according to a second embodiment of the present invention. Unlike the first embodiment, a CS current is added between the source of the main switching transistor 32 and the control circuit. Detection circuit.
  • This embodiment is mainly embodied in the light load control, the sampling output voltage Vout controls the stability of the output voltage, and the peak current of the main switch tube is sampled to realize the light load control.
  • the main switch tube peak current is reduced, and when the load is reduced to a certain extent, the circuit enters DCM mode operation.
  • the minimum peak current is kept from being controlled to no longer decrease, that is, the minimum peak current control, that is, the output of the converter is stabilized by reducing the operating frequency of the switching tube when the load continues to decrease, and When the working frequency reaches the minimum working frequency, it enters the Burst working mode.
  • This control can ensure the ZVS of the main switch and the auxiliary switch while the light load is reduced, which is beneficial to improve the light load efficiency of the module.
  • Figure 8-2 shows the corresponding frequency control curve.
  • FIG. 9 is a schematic diagram showing a third embodiment of the present invention, that is, the zero voltage switching circuit 40 of the present invention is also applicable to a synchronously rectified Boost circuit, which differs from the first embodiment in that the rectifier diode 33 is replaced.
  • the synchronous rectification switch 81 has an input end connected to the drain of the main switch 32, an output connected to the source of the synchronous rectification switch 81, and a drain connection output of the synchronous rectification switch 81.
  • the embodiment further includes a control circuit 86.
  • the control circuit 86 outputs three driving signals to control the switches of the main switch 32, the auxiliary switch 42 and the synchronous rectifier switch 81.
  • the main switch 32 and the auxiliary switch 42 are driven separately.
  • the auxiliary switching tube 42 is driven in synchronization with the synchronous rectification switching tube 81.
  • the working principle of this embodiment can refer to the working principle of Embodiment 1, and will not be described in detail herein.
  • the Boost boost inductor 31 current is continuous (current is not zero)
  • the zero-voltage turn-on of the main switch and the synchronous rectifier switch can also be realized
  • the Boost boost inductor 31 is interrupted (current zero-crossing)
  • the boost inductor will generate a negative current under the action of the output voltage.
  • FIG 10 shows a schematic diagram of a fourth embodiment in accordance with the present invention.
  • This embodiment differs from Embodiment 1 in that the auxiliary diode 43 in parallel with the resonant capacitor is eliminated, and the complementary drive control is also employed, when the Boost boost inductor current When it is continuous, this embodiment is consistent with the working principle of the embodiment 1, and the zero voltage opening of the main switch tube and the auxiliary switch tube can also be realized, which will not be described in detail herein.

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Abstract

一种零电压开关同步整流Boost电路、零电压开关的Boost电路（30）及其控制方法，在主功率电路的整流电路中串入辅助谐振电路（40），当主功率开关管（32）关断后，利用Boost升压电感（31）的电流实现辅助开关管（42）的零电压导通，然后利用谐振电容（44）与谐振电感（41）的谐振实现谐振电感（41）电流的快速增加并且促使谐振电感（41）的电流大于Boost升压电感（31）的电流，并利用与谐振电容（44）并联的二极管（43）对谐振电感（41）电流的续流，以保证谐振电容（44）的电压不会反向以及谐振电感（41）的电流在达到最大值后不会快速减小。当辅助开关管（42）关断后，利用谐振电流与Boost升压电感（31）电流的差值来实现主开关管（32）的零电压开通。

Description

一种零电压开关Boost电路及其控制方法 技术领域
本发明涉及电力电子电路，尤其涉及一种零电压开关Boost电路及其控制方法。
背景技术
一种传统的电源框图如图1所示，桥式整流电路11将AC输入转换成直流电压供给Boost变换器20，所述Boost变换器提供功率因素矫正功能以满足行业标准或者单纯地将波动较大的电压变换到一个稳定的电压或者变化范围较小的一个电压范围，后级的DC/DC变换器将所述Boost变换器的输出电压变换到负载所需的电压并实现隔离的作用。通常情况，AC输入范围为85-265VAC，Boost变换器的稳压值为400V或者稳压范围为200-400V。前级Boost电路的输出电压范围越窄，后级DC/DC变换器设计越简单、能够达到的性能越好。
一种传统的Boost变换器20如图2所示，当主开关管22在控制电路26的作用下进入导通状态时，电感21的一端通过主开关管22短接到地，而所述电感21的另一端与电源正相连，因此输入电压Vin将使电感21中的电流上升。同时，在主开关22的导通时间内，整流二极管23反向偏置而处于截止状态，输出滤波电容24给负载供电。当主开关管22被控制电路26关断时，电感电流不能突变，在主开关管22导通时间内储存在电感21中的能量能够通过整流二极管23供给负载。输出滤波电容24维持输出电压基本为常量。主开关管22的导通时间和关断时间由控制电路26决定，以保证输出电压为某个设定的电压值。
随着电力电子技术的发展进步，电力电子电路朝着高频化、小型化方向发展。因工作频率的提高，电源模块可以使用更小的器件，拥有更小的体积。对于上述传统的Boost变换器20，主开关管22的开关过程存在开关损耗，并且在电感电流连续时，整流二极管23存在反向恢复损耗，因而高频工作将引起上述传统的Boost变换器20的开关损耗的明显增加并导致模块效率的显著降低，由于变换器损耗的增加、体积的减小，模块发热严重，可靠性显著降低。零电压开关的变换器可以大大降低开关损耗，因而，零电压开关的变换器越来越引起人们的关注。
一种现有的实现ZVS的Boost电路为如图6-1所示的同步整流Boost电路，采取PWM控制，图6-2所示为其工作时序波形，利用输出电压和同步整流二极管来实现Boost升 压电感的负向励磁，利用负向的电感电流来实现主开关管ZVS，实现该工作过程的条件是电路工作在DCM模式，即电感电流过零。该电路适合功率较小的场合，并且轻载时负向电流明显增加，轻载效率不高。
现有的另一种典型控制方式的同步整流Boost变换器及其控制框图如图7所示，检测Boost升压电感的负电流、输入电压Vin与输出电压Vout来做控制，其中检测Vout来控制输出电压稳定，检测Vin来设置升压电感负向电流的最小值，检测升压电感负向电流来控制同步整流二极管的关断，为PFM控制。轻载时工作频率更高，轻载效率偏低；高压轻载时工作频率更高，高压轻载效率更低。
发明内容
有鉴如此，本发明提供一种零电压开关的Boost电路及其控制方法，以解决高频工作开关损耗过大的问题，解决电感电流连续整流二极管存在的反向恢复问题，同时也解决硬开关所产生的EMI问题。同时为了提高轻载效率，本发明还提供一种轻载控制方法，以进一步提高Boost变换器的整体效率。并且所述控制方法自动适应DCM(电流断续模式)与CCM(电流连续模式)工作模式。
一种零电压开关Boost电路，包含升压电感、主开关管、整流二极管、输出滤波电容以及控制电路，还包含零电压开关电路；
升压电感的一端连接输入电压+，升压电感的另一端连接主开关管的漏极，主开关管的源极连接输入电压-，主开关管的的栅极连接控制电路的一路输出端；输出滤波电容的两端输出电压Vout给后级负载供电；所述控制电路根据输出电压Vout产生反馈电压信号并根据反馈电压信号调节主开关管的占空比；所述零电压开关电路的输入端与升压电感的另一端相连，所述零电压开关电路的输出端与所述整流二极管的阳极相连，所述零电压开关电路的控制端与控制电路的另一路输出端相连；所述整流二极管的阴极连接输出滤波电容的正极。
优选的，零电压开关Boost电路还包括一个CS电流检测电路，所述CS电流检测电路的输入端连接主开关管的源极，用于检测主开关管的源极的电流；所述CS电流检测电路的输出端连接控制电路。
优选的，所述的零电压开关电路包含谐振电感、辅助开关管、谐振电容，谐振电感的一端作为零电压开关电路的输入端，谐振电感的另一端作为零电压开关电路的输出端；辅助开关管的源极连接谐振电感的一端，辅助开关管的漏极连接谐振电容的正极， 谐振电容的负极连接谐振电感的另一端；辅助开关管的栅极作为零电压开关电路的控制端。
优选的，所述的零电压开关电路还包括辅助二极管，辅助二极管的阴极连接谐振电容的正极，辅助二极管的阳极连接谐振电容的负极。
优选的，控制电路对主开关管与辅助开关管的控制为互补驱动控制。
优选的，主开关管与辅助开关管为MOS管或IGBT，所述辅助二极管为肖特基二极管。
一种零电压开关Boost电路，包括升压电感、主开关管、输出滤波电容、同步整流开关管，还包含零电压开关电路；
升压电感的一端连接输入电压+，升压电感的另一端连接主开关管的漏极，主开关管的源极连接输入电压-，主开关管的的栅极连接控制电路的一路输出端；输出滤波电容的两端输出电压Vout给后级负载供电；所述控制电路根据输出电压Vout产生反馈电压信号并根据反馈电压信号调节主开关管的占空比；所述零电压开关电路的输入端与主开关管的漏极相连，所述零电压开关电路的输出端与同步整流开关管的源极相连，所述零电压开关电路的控制端与控制电路的另一路输出端相连；所述同步整流开关管的漏极连接输出滤波电容的正极，所述同步整流开关管的栅极与控制电路的第三路输出端相连。
优选的，所述的零电压开关电路包含谐振电感、辅助开关管、谐振电容，谐振电感的一端作为零电压开关电路的输入端，谐振电感的另一端作为零电压开关电路的输出端；辅助开关管的源极连接谐振电感的一端，辅助开关管的漏极连接谐振电容的正极，谐振电容的负极连接谐振电感的另一端；辅助开关管的栅极作为零电压开关电路的控制端。
优选的，主开关管与辅助开关管互补驱动，辅助开关管与同步整流开关管同步驱动。
优选的，所述的零电压开关电路还包括辅助二极管，辅助二极管的阴极连接谐振电容的正极，辅助二极管的阳极连接谐振电容的负极。
就电路本身而言，本发明目的是这样实现的，在主功率电路的整流电路中串入辅助谐振电路，当主功率开关管关断后，利用Boost升压电感的电流实现辅助开关管的零电压导通，然后利用谐振电容与谐振电感的谐振实现谐振电感电流的快速增加并且促使谐振电感的电流大于Boost升压电感的电流，并利用与谐振电容并联的二极管对谐振电感 电流的续流，以保证谐振电容的电压不会反向以及谐振电感的电流在达到最大值后不会快速减小。当辅助开关管关断后，利用谐振电流与Boost升压电感电流的差值来实现主开关管的零电压开通。
一种零电压开关Boost电路的控制方法：CS电流检测电路检测主开关管的峰值电流，当所述峰值电流减小到设定的参考值时从控制上保持这个最小峰值电流不再减小，当输入电压升高或者负载继续减小造成主开关管峰值电流有继续减小的趋势时，通过降低主开关管的工作频率来稳定Boost电路的输出；而当主开关管的工作频率达到最小工作频率时则进入跳频工作。
就控制而言，本发明的零电压Boost电路可以简单地采用互补驱动来实现主开关管与辅开关管的零电压开通的功能，在这种简单的控制实现中，只需采样输出电压来控制主开关管的导通时间即可。辅助开关管导通前的电压应力很小，因而主管关断后就可立即开通辅助开关管，也就是说这个死区时间可以很短；但是主开关管导通前的电压应力为Vout，需要一定的时间才能抽到零，因此辅开关管关断与主开关管导通之间需要留合适的死区时间。
就轻载控制而言，为了提高本发明零电压Boost电路在轻载时的效率，本发明提出了一种在互补控制的基础上实现降频控制的方法，在采样输出电压的基础上采样主开关管的峰值电流，当负载减小时，主开关管峰值电流减小，当负载减小到一定程度时电路进入DCM模式工作。当峰值电流减小到一定程度时从控制上保持这个最小峰值电流不再减小，即最小峰值电流控制，也就是说负载继续减小时通过降低开关管的工作频率来稳定变换器的输出，而当工作频率达到最小工作频率时则进入跳频(Burst)工作模式，该控制能保证轻载降频的同时仍然实现主开关管与辅开关管的ZVS(零电压开通)，有利于提升模块的轻载效率和EMI性能。
与现有技术相比，本发明具有如下有益效果：
(1)实现ZVS工作不受工作模式的限制，在CCM模式与DCM模式下均能实现主开关管与辅助开关管的ZVS；
(2)应用简单；
(3)整流二极管电流自然过零关断，不存在反向恢复问题；
(4)在互补驱动下也很容易实现轻载降频工作，轻载效率更高；
(5)ZVS的实现不影响Boost升压电感的电流，Boost升压电感的电流不会过零，可 以用于PFC控制；
(6)当应用于同步整流Boost电路时，Boost升压电感无需负向励磁就能实现主开关管的零电压开通。
附图说明
图1为传统的电源框图；
图2为传统的二极管整流Boost变换器原理图；
图3为本发明实施例1电路原理图及控制框图；
图4为本发明施例1的工作模态图；
图5为根据本发明施例1的工作波形；
图6-1为一种现有的同步整流Boost变换器及其控制框图；
图6-2为图6-1所示同步整流Boost变换器的控制时序图；
图7为现有的具有另一种控制方式的同步整流Boost变换器及其控制框图；
图8-1为本发明实施例2电路原理图及控制框图；
图8-2为本发明实施例2的频率变化曲线示意图；
图9为本发明实施例3电路原理图；
图10为本发明实施例4电路原理图。
具体实施方式
第一实施例
图3所示为依据本发明的零电压开关Boost电路实施例1原理图，与传统Boost电路类似，所述零电压开关Boost电路30包含升压电感31、主开关管32、整流二极管33、输出滤波电容34以及控制电路36，输出滤波电容34的两端输出电压Vout给负载35供电；所述控制电路36根据输出电压Vout产生反馈电压信号并根据反馈电压信号调节主开关管32的占空比；升压电感31的一端连接输入电压+，升压电感31的另一端连接主开关管32的漏极，主开关管32的源极连接输入电压-，主开关管32的的栅极连接控制电路的一路输出端。与传统Boost电路不同的是，所述零电压开关Boost电路30还包含零电压开关电路40，所述零电压开关电路40的输入端与升压电感31的另一端相连，输出端与所述整流二极管33的阳极相连，控制端与控制电路的另一路输出端相连。
所述零电压开关电路40包含谐振电感41、辅助开关管42、辅助二极管43、谐振电 容44，谐振电感41的一端连接升压电感31的另一端，谐振电感41的另一端连接整流二极管33的阳极，整流二极管33的阴极连接输出滤波电容34的正极；辅助开关管42的源极连接谐振电感41的一端，辅助开关管42的漏极分别连接谐振电容44的正极和辅助二极管43的阴极，谐振电容44的负极连接辅助二极管43的阳极并且连接到谐振电感41的另一端。
图3中Ds1和Cs1分别为主开关管32的寄生二极管和寄生电容，实际电路中并不存在；同样，Ds2和Cs2分别为辅助开关管42的寄生二极管和寄生电容。
所述主开关管32与辅助开关管42为全控型半导体开关；
优选地，主开关管32与辅助开关管42为图3所示MOS管；
优选地，主开关管32与辅助开关管42为IGBT；
优选地，主开关管32与辅助开关管42为SiC MOS管或GaN MOS管；
优选地，辅助二极管43为肖特基二极管。
图4所示为本发明实施例1开关工作过程中的主要工作模态，图5所示为本发明实施例1开关工作过程中的主要波形，现对照工作模态对工作波形做简要说明。
Mode1(t0-t1)：t0时刻，辅助开关管42处于关断截止状态，主开关管32从导通状态切换到关断截止状态，因电感电流不能突变，短时间内可认为谐振电感41电流不变，升压电感31的电流一部分给所述主开关管32的漏源极结电容Cs1充电，导致Vds1快速上升；另一部分电流给所述辅助开关管42漏源极结电容Cs2放电，导致Vds2快速下降到0；
Mode2(t1-t2)：t1时刻，所述辅助开关管42漏源极结电容Cs2被放电到0V，所述辅助开关管42体二极管Ds2导通，所述升压电感31的电流的一部分继续给所述主开关管32的漏源极结电容Cs1充电，另一部分通过所述辅助开关管42的体二极管Ds2给所述谐振电容44(Cr)充电，同时，所述谐振电容44的电压VCr加在所述谐振电感41两端给谐振电感励磁，也即所述主开关管32的漏源极结电容Cs1与所述谐振电容44(Cr)并联后与所述谐振电感41谐振；
Mode3(t2-t3)：t2时刻，所述辅助开关管42零电压导通，不影响此时正在进行的谐振过程，t3时刻，所述谐振电容44(Cr)电压谐振到0V，所述谐振电感41的电流达到最大值；
Mode4(t3-t4)：此时间段内，辅助二极管43导通，所述谐振电感41的电流通过 所述辅助二极管43与辅助开关管42续流并保持基本不变；
Mode5(t4-t5)：t4时刻，辅助开关管42关断，谐振电感41的电流的一部分给所述辅助开关管42漏源极结电容Cs2充电，另一部分给所述主开关管32的漏源极结电容Cs1放电，也即所述主开关管32的漏源极结电容Cs1与所述辅助开关管42漏源极结电容Cs2并联后与所述谐振电感41谐振；
Mode6(t5-t6)：t5时刻，辅助开关管42漏源极电压Vds2上升到Vout，主开关管32的漏源极压Vds1减小到0V，所述主开关管32的体二极管Ds1导通，加在Lr两端的电压为-Vout，iLr线性减小，加在Lp两端电压为Vin，iLp线性增加；
Mode7(t6-t7)t6时刻，主开关管32零电压导通，不影响此前的工作过程，iLr继续线性减小，iLp继续线性增加，直到iLp>iLr时，流过主开关管32的电流变为正电流；
Mode8(t7-t8)：t7时刻，谐振电感41的电流线性减小到0，整流二极管33零电流关断，辅助开关管42漏源极结电容Cs2与谐振电容44(Cr)串联后与Lr谐振，t8时刻，所述辅助开关管42漏源极电压Vds2谐振到0V，谐振电感41电流达到负的最小值；
Mode9(t8-t9)：t8时刻，辅助开关管42体二极管Ds2导通，谐振电容44(Cr)与Lr串联谐振，t9时刻，所述谐振电感41的电流谐振到0，谐振电容电压达到一个相对稳定值的值；
Mode10(t9-t10)：谐振电容电压达到一个相对稳定的值后Lr将与辅助开关管42漏源极结电容Cs2小幅度谐振，在这里被忽略掉而没有出现在典型工作波形中；t10时刻，主开关管32再次从导通状态切换到关断截止状态，开始另一个循环过程。
第二实施例
图8-1示出了根据本发明的第二实施例的原理图及控制框图，与第一实施例不同的是，在主开关管32的源极与控制电路之间，增加了一个CS电流检测电路。
此实施例主要体现在轻载控制上，采样输出电压Vout控制输出电压的稳定，采样主开关管峰值电流来实现轻载控制。当负载减小时，主开关管峰值电流减小，当负载减小到一定程度时电路进入DCM模式工作。当峰值电流减小到一定程度时从控制上保持这个最小峰值电流不再减小，即最小峰值电流控制，也就是说负载继续减小时通过降低开关管的工作频率来稳定变换器的输出，而当工作频率达到最小工作频率时则进入Burst 工作模式，该控制能保证轻载降频的同时时仍然实现主开关管与辅开关管的ZVS，有利于提升模块的轻载效率。图8-2所示为对应的频率控制变化曲线示意图。
第三实施例
图9示出了依据本发明的第三实施例的原理图，也就是说本发明的零电压开关电路40同样适用于同步整流的Boost电路，与实施例1的区别在于整流二极管33换成了同步整流开关管81，所述零电压开关电路40的输入端与主开关管32的漏极相连、输出端与同步整流开关管81的源极相连，而同步整流开关管81的漏极连接输出滤波电容34的正极。另外本实施例还包含控制电路86，控制电路86输出三路驱动信号来控制主开关管32、辅助开关管42以及同步整流开关管81的开关，其中主开关管32与辅助开关管42互补驱动，辅助开关管42与同步整流开关管81同步驱动。
本实施例的工作原理可以参照实施例1的工作原理，此处不再详细说明。特别说明的是当Boost升压电感31电流连续时(电流不过零)也能实现主开关管与同步整流开关管的零电压开通，而当Boost升压电感31电流断续时(电流过零)，升压电感将在输出电压的作用下负向励磁而产生负电流，在此情况下当辅助开关管与同步整流开关管关断时升压电感的负向励磁电流与谐振电感的电流将同时作用来抽取主开关管32的漏源极结电容电荷，因而这并不影响主开关管零定压的实现。
第四实施例
图10示出了依据本发明的第四实施例的原理图，此实施例与实施例1的区别在于少了与谐振电容并联的辅助二极管43，同样采取互补驱动控制，当Boost升压电感电流连续时，此实施例与实施例1工作原理一致，同样可以实现主开关管与辅开关管的零电压开通，这里不再做详细说明。
以上公开的仅为本发明的具体实施例，但是本发明并非局限于此，任何本领域的技术人员在未脱离本发明的核心思想的前提下对本发明进行的若干修饰均应该落在本发明权利要求的保护范围之类。

Claims (11)

  1. 一种零电压开关Boost电路，包含升压电感(31)、主开关管(32)、整流二极管(33)、输出滤波电容(34)以及控制电路(36)，其特征在于：还包含零电压开关电路(40)；
    升压电感(31)的一端连接输入电压+，升压电感(31)的另一端连接主开关管(32)的漏极，主开关管(32)的源极连接输入电压-，主开关管(32)的的栅极连接控制电路的一路输出端；输出滤波电容(34)的两端输出电压Vout给后级负载供电；所述控制电路(36)根据输出电压Vout产生反馈电压信号并根据反馈电压信号调节主开关管(32)的占空比；所述零电压开关电路(40)的输入端与升压电感(31)的另一端相连，所述零电压开关电路(40)的输出端与所述整流二极管(33)的阳极相连，所述零电压开关电路(40)的控制端与控制电路的另一路输出端相连；所述整流二极管(33)的阴极连接输出滤波电容(34)的正极。
  2. 根据权利要求1所述的一种零电压开关Boost电路，其特征在于：还包括一个CS电流检测电路，所述CS电流检测电路的输入端连接主开关管(32)的源极，用于检测主开关管(32)的源极的电流；所述CS电流检测电路的输出端连接控制电路。
  3. 根据权利要求1或2所述的一种零电压开关Boost电路，其特征在于：所述的零电压开关电路(40)包含谐振电感(41)、辅助开关管(42)、谐振电容(44)，谐振电感(41)的一端作为零电压开关电路(40)的输入端，谐振电感(41)的另一端作为零电压开关电路(40)的输出端；辅助开关管(42)的源极连接谐振电感(41)的一端，辅助开关管(42)的漏极连接谐振电容(44)的正极，谐振电容(44)的负极连接谐振电感(41)的另一端；辅助开关管(42)的栅极作为零电压开关电路(40)的控制端。
  4. 根据权利要求3所述一种零电压开关Boost电路，其特征在于：所述的零电压开关电路(40)还包括辅助二极管(43)，辅助二极管(43)的阴极连接谐振电容(44)的正极，辅助二极管(43)的阳极连接谐振电容(44)的负极。
  5. 根据权利要求4所述的一种零电压开关Boost电路，其特征在于：控制电路对主开关管(32)与辅助开关管(42)的控制为互补驱动控制。
  6. 根据权利要求5所述的一种零电压开关Boost电路，其特征在于：主开关管(32)与辅助开关管(42)为MOS管或IGBT，所述辅助二极管(43)为肖特基二极管。
  7. 一种零电压开关Boost电路，包括升压电感(31)、主开关管(32)、输出滤波电容(34)、同步整流开关管(81)，其特征在于：还包含零电压开关电路(40)；
    升压电感(31)的一端连接输入电压+，升压电感(31)的另一端连接主开关管(32)的漏极，主开关管(32)的源极连接输入电压-，主开关管(32)的的栅极连接控制电路的一路输出端；输出滤波电容(34)的两端输出电压Vout给后级负载供电；所述控制电路(36)根据输出电压Vout产生反馈电压信号并根据反馈电压信号调节主开关管(32)的占空比；所述零电压开关电路(40)的输入端与主开关管(32)的漏极相连，所述零电压开关电路(40)的输出端与同步整流开关管(81)的源极相连，所述零电压开关电路(40)的控制端与控制电路的另一路输出端相连；所述同步整流开关管(81)的漏极连接输出滤波电容(34)的正极，所述同步整流开关管(81)的栅极与控制电路的第三路输出端相连。
  8. 根据权利要求7所述的一种零电压开关Boost电路，其特征在于：所述的零电压开关电路(40)包含谐振电感(41)、辅助开关管(42)、谐振电容(44)，谐振电感(41)的一端作为零电压开关电路(40)的输入端，谐振电感(41)的另一端作为零电压开关电路(40)的输出端；辅助开关管(42)的源极连接谐振电感(41)的一端，辅助开关管(42)的漏极连接谐振电容(44)的正极，谐振电容(44)的负极连接谐振电感(41)的另一端；辅助开关管(42)的栅极作为零电压开关电路(40)的控制端。
  9. 根据权利要求8所述的一种零电压开关Boost电路，其特征在于：主开关管(32)与辅助开关管(42)互补驱动，辅助开关管(42)与同步整流开关管(81)同步驱动。
  10. 根据权利要求9所述的一种零电压开关Boost电路，其特征在于：所述的零电压开关电路(40)还包括辅助二极管(43)，辅助二极管(43)的阴极连接谐振电容(44)的正极，辅助二极管(43)的阳极连接谐振电容(44)的负极。
  11. 一种零电压开关Boost电路的控制方法，其特征在于：
    CS电流检测电路检测主开关管(32)的峰值电流，当所述峰值电流减小到设定的参考值时从控制上保持这个最小峰值电流不再减小，当输入电压升高或者负载继续减小造成主开关管(32)峰值电流有继续减小的趋势时，通过降低主开关管(32)的工作频率来稳定Boost电路的输出；而当主开关管(32)的工作频率达到最小工作频率时则进入跳频工作。
PCT/CN2019/070642 2018-04-04 2019-01-07 一种零电压开关Boost电路及其控制方法 WO2019192234A1 (zh)

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