WO2019129270A1 - 一种提高同步整流原边反馈反激式电源动态性能的方法 - Google Patents

一种提高同步整流原边反馈反激式电源动态性能的方法 Download PDF

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WO2019129270A1
WO2019129270A1 PCT/CN2018/125484 CN2018125484W WO2019129270A1 WO 2019129270 A1 WO2019129270 A1 WO 2019129270A1 CN 2018125484 W CN2018125484 W CN 2018125484W WO 2019129270 A1 WO2019129270 A1 WO 2019129270A1
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module
mode
state
dynamic
output
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PCT/CN2018/125484
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English (en)
French (fr)
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徐申
王炜
林峰
何波涌
苏巍
孙伟锋
时龙兴
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无锡华润上华科技有限公司
东南大学
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Publication of WO2019129270A1 publication Critical patent/WO2019129270A1/zh
Priority to US16/915,524 priority Critical patent/US11394306B2/en

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K5/00—Manipulating of pulses not covered by one of the other main groups of this subclass
    • H03K5/22—Circuits having more than one input and one output for comparing pulses or pulse trains with each other according to input signal characteristics, e.g. slope, integral
    • H03K5/24—Circuits having more than one input and one output for comparing pulses or pulse trains with each other according to input signal characteristics, e.g. slope, integral the characteristic being amplitude
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/165—Indicating that current or voltage is either above or below a predetermined value or within or outside a predetermined range of values
    • G01R19/16528—Indicating that current or voltage is either above or below a predetermined value or within or outside a predetermined range of values using digital techniques or performing arithmetic operations
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
    • H02M3/33523—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0006—Arrangements for supplying an adequate voltage to the control circuit of converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0012—Control circuits using digital or numerical techniques
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0025—Arrangements for modifying reference values, feedback values or error values in the control loop of a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/1566—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with means for compensating against rapid load changes, e.g. with auxiliary current source, with dual mode control or with inductance variation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33584—Bidirectional converters

Definitions

  • the present invention relates to an isolated switching power converter, and more particularly to a method for improving the dynamic performance of a synchronous rectification primary feedback flyback power supply with heavy load and light load, which is suitable for a synchronous rectified flyback power supply during high power operation.
  • Switching power supplies are typically used as power supplies for all types of electrical equipment to convert unregulated AC or DC input voltages into regulated AC or DC output voltages. Since the switching power supply needs to be adapted to different operating conditions, the performance requirements for the dynamic response of the power supply are getting higher and higher. Good dynamic effects require small voltage changes and voltage recovery times.
  • the primary side feedback (PSR) flyback converter is widely used in the field of switching power supplies because of its simple structure, low cost, and flexible control. There is a serious imbalance between the dynamic response performance of the PSR flyback converter and the actual demand, especially in the case of large load switching, the output voltage will fluctuate greatly. In order to improve the dynamic response performance, researchers of PSR flyback converters have proposed many innovative methods.
  • a digital multi-mode control algorithm has been proposed. Different load ranges use different working modes, and PWM is combined with PFM. Faster, smoother switching is achieved when the load is switched over a small range.
  • the control modes of the algorithm can only be switched in order, and the response time is long, the response speed is slow, and the overshoot voltage is large when the load is switched over a large range.
  • Chong Wang et al. proposed a hybrid control method.
  • This control method adds two dynamic working modes: LTH mode and HTL mode by adding a dynamic process detection module in the system, in addition to the steady state multi-mode.
  • LTH mode LTH mode
  • HTL mode control is performed using a fixed cycle and a fixed duty cycle.
  • Vo_ref the slope of Vo is used to judge the load point, so that the system enters the correct steady state mode of operation.
  • This scheme effectively reduces the overshoot voltage and response time under DCM and improves the dynamic performance of the converter, but it also has shortcomings.
  • the recovery time is too long and takes 53ms.
  • a method of improving the dynamic performance of a synchronous rectification primary feedback flyback power supply is provided.
  • a method for improving the dynamic performance of a synchronous rectifier primary feedback flyback power supply In the primary side feedback main topology structure, the secondary side adopts a synchronous rectification structure, and based on digital multi-mode control, a sampling module, a multi-mode judgment module, and a digital PID are set.
  • the control system composed of the module, the high dynamic control module, the load point judgment module and the PWM drive module forms a closed loop with the controlled switching power supply;
  • the high dynamic control module and the load point judgment module do not work;
  • the control module outputs a high dynamic Duty_SR_dynamic duty signal to the PWM drive module, the drive module is turned off by the PWM primary main switch, and a fixed period Ts_ HTL, fixed duty ratio D HTL manner Controlling the switch of the secondary synchronous rectifier tube, extracting the energy of the load capacitance of the output terminal to the primary side during the time when the synchronous rectifier is turned on, so that the energy stored in the load capacitor is rapidly decreased, so that the output voltage rapidly drops with a fixed slope, sampling The module always samples the output voltage.
  • the output voltage sample value Vo_sample drops to its rated value V O_REF , the system jumps from the dynamic working mode to the steady state operating mode;
  • the load point judging module receives the output voltage value Vo_sample outputted by the sampling module in the dynamic working mode, obtains the falling time ⁇ t of the output voltage by counting the switching period, and the output voltage change amount ⁇ Vo_sample is fixed, and the slope of the output voltage is obtained Kdown
  • the slope Kdown of the output voltage is kept constant, and the slope Kdown is different at different load points. According to the slope, Kdown corresponds to the load at this time, and the current load point is determined, and then the dynamic process is determined according to the current load size.
  • the system should be in the DPFM steady state operating mode and the duty cycle Ts_judge, the primary peak current Ipeak (the primary peak current is fixed in the DPFM mode), giving the jump state determination signal state_judge and the switching period Ts_judge at this time, and
  • the switching period Ts_judge is input to the PWM driving module, and the state determining signal state_judge is input to the multi-mode judging module, after which the system enters the steady state working mode and adopts the traditional digital PID control;
  • the digital PID compensation module calculates the control voltage compensation amount V PI according to the error signal err outputted by the sampling module and the state signal state output by the multi-mode judging module, and the multi-mode judging module according to Control the voltage compensation amount VPI to perform steady-state multi-mode switching.
  • the output includes a steady state value of CCM, PWM, PFM, DPWM, and DPFM.
  • the PWM drive module uses peak current control in steady state.
  • the period of the primary side main switching tube is calculated according to the control voltage compensation amount VPI, and the duty ratio of the secondary side synchronous rectifier tube is calculated according to the current output voltage sampling value Vo_sample, and the period of the secondary side synchronous rectifier tube and the main The switch tube is consistent;
  • the duty cycle signal output by the PWM module is controlled by the drive module to obtain the PWM waveform of the main switching tube of the switching power supply and the synchronous rectifier of the secondary side to achieve constant voltage output.
  • the application also provides a control system and a control method for a primary feedback flyback switching power supply.
  • 1 is a block diagram showing the system structure of the control method of the present application.
  • 2 is an internal structure diagram of a sampling module
  • 3 is an internal structure diagram of a multi-mode judging module
  • Figure 5 is a diagram showing the internal structure of the load point judging module
  • Figure 6 is a diagram showing the internal structure of a digital PID module
  • Figure 7 is a diagram showing the internal structure of the PWM driving module
  • FIG. 8 is a schematic diagram of an application of a heavy-duty cut light load HTL mode
  • Figure 9 is a key waveform diagram of output voltage, primary current and secondary current when the load is switched to light load mode
  • Figure 11 shows the dynamic results before the high dynamics algorithm of the present application is used when the load is switched from 4 ⁇ to 800 ⁇ ;
  • Figure 12 shows the dynamic results of the high dynamics algorithm of this application when the load is switched from 4 ⁇ to 800 ⁇ .
  • the present application proposes a method for improving the dynamic performance of a synchronous rectification primary feedback flyback power supply.
  • the overshoot of the output voltage can be limited.
  • significantly shorten the dynamic recovery time significantly improve the dynamic performance, does not cause system instability in multi-mode control, has a good effect on reducing voltage overshoot and reducing dynamic recovery time, making the system The dynamic performance is even better.
  • FIG. 1 is a system diagram of an overall implementation of a digital multi-mode control algorithm for improving the dynamic performance of a heavy-duty switching light load of a flyback converter according to the present application.
  • a synchronous rectification structure is adopted, and based on the digital multi-mode control, a sampling module, a multi-mode judging module, a digital PID compensation module, and a dynamic control module are set (in this embodiment, the dynamic mode judgment signal is 1
  • the load point judgment module, the PWM drive module and the controlled switching power supply form a closed loop system.
  • the main switch tube by controlling the secondary side synchronous rectifier tube to fix the duty cycle switch, causes the output voltage to rapidly drop
  • the output voltage value Vo_sample outputted by the sampling module is controlled by The calculation module calculates the output voltage drop slope Kdown, and according to the slope, the load size at this time is obtained, and according to the current load size, the DPFM steady state operation mode and the duty cycle Ts_judge and the primary peak value of the system should be determined at the end of the dynamic process.
  • the current Ipeak (the primary peak current is fixed in the DPFM mode) gives the jump-out state determination signal state_judge and the switching period Ts_judge at this time, and inputs the switching period Ts_judge to the PWM driving module, and inputs the state determination signal state_judge to the multi-mode.
  • the judgment module enters the steady state thereafter and adopts the traditional digital PID control; the PWM drive module turns off the primary main switch tube in the dynamic mode HTL state, receives the signal Duty_SR_dynamic, and controls the vice by a fixed period and a fixed duty ratio.
  • the PWM of the main synchronous switch and the secondary synchronous rectifier is controlled by the switch of the synchronous rectifier; the PWM drive module adopts peak current control in steady state.
  • FIG. 2 is a diagram showing the internal structure of the sampling module.
  • the sampling module collects information on the resistor divider voltage Vsense on the primary auxiliary winding, and includes two comparators COMP1, COMP2, a waveform real-time analysis module and a subtractor, and the positive terminal of the comparator COMP1 is connected to the resistor divider on the auxiliary winding.
  • Vsense the sampling signal Vsample output by the real-time analysis module of the negative-end connection waveform
  • the positive terminal of the comparator COMP2 is connected to Vsense
  • the negative terminal is connected to the 0 level
  • the waveform real-time analysis module outputs the sampling signal Vsample and the output voltage according to the results of the two comparators.
  • the voltage waveform Vsense after the resistor divider on the primary auxiliary winding is the input of the sampling module.
  • the two comparators COMP1 and COMP2 compare Vsense with Vsample and 0 voltage respectively, and obtain two signals comp_sample and comp_zero respectively.
  • the analysis module analyzes the output signals of the two comparators to obtain the current output voltage sample value Vo_sample, and adjusts the value of the sampling signal Vsample to prepare for the next sampling; the Vo_sample and the reference value of the output voltage V O_REF ( The fixed value is subtracted to obtain the error value err of the output voltage.
  • the multi-mode judging module receives the output voltage value Vo_sample collected by the sampling module, and includes two comparators COMP3, COMP4, a dynamic mode judging module, a steady state mode judging module and a multi-mode state judging module, and the positive end of the comparator COMP3 is connected.
  • the set Vo_sample upper limit value is Vomax, the negative end is connected to Vo_sample; the positive end of the comparator COMP4 is connected to Vo_sample, the negative end is connected to the set rated voltage V O_REF , and the dynamic mode judging module outputs the dynamic mode judgment according to the results of the two comparators.
  • Signal mode_dynamic is an internal structural diagram of a multi-mode control module.
  • the control voltage compensation amount V PI and the state signal state are input to the steady state mode judging module, and the steady state mode judging module outputs the steady state signal state_steady.
  • FIG 4 is a structural diagram of a high dynamic control module.
  • FIG. 5 is an internal structural diagram of a load point judging module. It includes a comparator, a parameter calculation module, a slope calculation module, and a pop-out mode determination module.
  • the positive terminal of the comparator COMP is connected to Vo_sample
  • the negative terminal is connected to the set rated voltage V O_REF
  • the comparator outputs the reference voltage comparison signal Comp_V REF
  • the input of the parameter calculation module is the dynamic mode determination signal Mode_dynamic and the output voltage sampling value Vo_sample
  • the output is The output voltage variation ⁇ Vo_sample and the corresponding time ⁇ t
  • the inputs of the slope calculation module are ⁇ Vo_sample, ⁇ t and Comp_V REF
  • the output is the voltage drop slope Kdown
  • the input of the jump-out mode determination module is Kdown
  • the output is the jump-out state determination signal state_judge and
  • the switching period is Ts_judge.
  • state HTL
  • the parameter calculation module counts by the counter to obtain ⁇ t, and according to the input Vo_sample value, the variation ⁇ Vo_sample of the output voltage in the ⁇ t time is obtained.
  • the slope calculation module analyzes the output of the comparator Comp_V REF to capture the time when Vo_sample drops to V O_REF , and then calculates the slope Kdown of the output voltage.
  • the subsequent jump-out mode determination module determines the operation of the system after jumping out of the dynamic according to the value of Kdown.
  • the mode state_judge and the switching period value Ts_judge of the first cycle, the state judgment signal state_judge and the switch cycle signal Ts_judge after the jumpout are valid only when jumping out of the dynamic mode.
  • FIG. 6 is an internal structural diagram of a digital PID compensation module. It includes a PI parameter selection module and a digital PI compensation algorithm module.
  • the input of the PI parameter selection module is the state signal state
  • the output is the PI parameters Kp and Ki
  • the input of the digital PI compensation algorithm module is Kp, Ki and the error signal err
  • the output is the control voltage compensation amount V PI .
  • This application uses the PID compensation algorithm, but only adopts P adjustment and I adjustment, and does not use D adjustment.
  • different parameters Kp and Ki are selected, and input with the current output voltage error value err to the following digital PI compensation algorithm module, and the control voltage compensation amount of the next cycle is calculated.
  • V PI In dynamic mode, the digital PID compensation module does not work.
  • FIG. 7 is a diagram showing the internal structure of the PWM drive module.
  • the utility model comprises a gate module, a main switch tube cycle Ts calculation module, a main switch tube duty control module and a synchronous rectifier SR tube duty control module.
  • the state signal State is input to the gating module to obtain two signals: a steady state signal State_HTL and a dynamic state signal State_steady.
  • the strobe module determines whether the current working mode state is in the steady state operating mode state_steady or the dynamic working mode state_HTL.
  • the gate module output is State_HTL
  • the inputs are V PI , State_steady and Ts_judge, and the output is the main switch cycle signal Ts.
  • the subsequent calculation of the switching period value Ts is the same as the steady state mode.
  • the peak current control is adopted in the steady state mode, and the period Ts of the main switching tube in the PFM and DPFM modes is calculated according to the digital compensation amount V PI , and the remaining steady state operating modes are fixed.
  • the period of the secondary synchronous rectifier (SR tube) is equal to the period of the primary switching transistor.
  • the value Vo_sample calculates the duration Ton_sr of the SR tube opening.
  • the period of the secondary synchronous rectifier (SR tube) is equal to the period of the primary switching transistor.
  • the value Vo calculates the duration Ton_sr of the SR tube opening.
  • the dynamic control method proposed in the present application can turn off the primary switching tube of the primary side when the output voltage exceeds the upper limit voltage, and control the switch of the secondary synchronous rectifier tube by using a fixed period and a fixed duty ratio, in synchronous rectification During the time when the tube is turned on, the energy of the load capacitance of the output terminal is extracted to the primary side, so that the output voltage is rapidly decreased, the overshoot voltage is greatly reduced, and the dynamic recovery time is greatly reduced to be shortened to 2.5 ms.
  • the dynamic control method proposed in the present application calculates the slope of the output voltage reaching the upper limit and then falls to the rated value in the heavy load cut light load mode, and obtains the load size according to the one-to-one monotonic relationship between the slope and the load, when jumping out After the HTL mode, it jumps to the steady state working state of the corresponding load point. After the jump, the energy and the load steady state consumption are not much different, eliminating the subsequent voltage oscillation and reducing the dynamic recovery time.
  • the slope is used to determine the magnitude of the load after the jump, which can avoid the large voltage resonance caused by the large difference between the energy and the steady-state consumption of the load after the jump, so that the oscillation caused by the heavy load cut light load mode and the normal mode is more stable.
  • This application is based on the digital multi-mode control method, which increases the heavy-duty cut light-load mode—the HTL mode and the load point judgment method, and does not affect the stability of the general multi-mode control loop.
  • This application can be applied to the switching power supply circuit structure of the synchronous rectification structure, and has versatility, reusability and portability.
  • the input voltage range of the flyback converter is 90-265V, the output voltage is constant voltage 20V, the current is 5A, the transformer inductance is 417 ⁇ H, the transformer turns ratio is 45:8:4, and the clock frequency is 20MHz.
  • the relationship between the output load and the voltage drop slope Kdown and the corresponding operating state is as shown in Table 1.
  • the HTL mode when the heavy load is cut lightly, it can be seen from the schematic diagram that when the output voltage is greater than Vomax, the HTL mode is employed. If the PID adjustment is used, as shown by the thick dotted line, the voltage will still rise after the output voltage rises to Vomax, and the dynamic recovery time is also very long.
  • the HTL mode when the output voltage is greater than the Vomax, the HTL mode is used immediately, and the shutdown is performed.
  • the primary switching tube of the primary side controls the switch of the secondary synchronous rectifier tube by means of fixed period and fixed duty ratio. When the synchronous rectifier is opened, the energy of the output capacitor of the output terminal is extracted to the primary side, so that the output voltage is rapid.
  • the output load can be obtained by the slope size, so that the operating mode energy jumping out of the HTL mode is similar to the load power consumption, and the resonance introduced by the subsequent energy mismatch is removed, as shown by the solid line;
  • the HTL mode is jumped out, if the working state is not suitable, the input energy is low, and the voltage resonance is introduced, as shown by the thin dotted line.
  • Figure 9 is a key waveform diagram of the output voltage, primary current, secondary current, and auxiliary winding waveform Vsense when the load is switched to light load mode.
  • the output voltage is greater than Vomax, enter the HTL mode.
  • Primary switch is turned off, only the secondary-side synchronous rectifier at a fixed cycle Ts_ HTL, fixed duty ratio D HTL way switch.
  • the secondary current Is increases linearly in the reverse direction to form a negative current, and reaches the maximum value of the secondary negative current when the synchronous rectifier is turned off.
  • the energy of the output terminal load capacitor is stored in the magnetizing inductance, causing the output voltage to drop rapidly.
  • the sampling is performed within the time when the synchronous rectifier is turned on, and the waveform of the auxiliary winding Vsense is also as shown.
  • the energy stored in the magnetizing inductance is coupled to the primary side, forming a negative current on the primary side, and returning energy to the input voltage network through the body diode of the primary side main switching tube, thereby gradually reducing the negative current of the primary side. As small as zero, the energy extracted from the secondary side load capacitor has been released.
  • the above process is repeated by controlling the switches of the secondary side synchronous rectifier.
  • FIG. 10 is an overall flow chart of the high dynamic control algorithm of the present application.
  • the voltage waveform on the auxiliary winding is first collected, and then the mode judgment is performed to determine whether the system enters the dynamic mode. If the system enters the dynamic mode, so that the primary switch is turned off, a fixed cycle Ts_ HTL, the duty ratio D HTL fixed manner synchronous rectifier switching control, the output voltage drops with a fixed slope (constant load, a voltage The falling slope does not change). At the same time, the output voltage is sampled by the auxiliary winding waveform Vsense during the period in which the synchronous rectifier is turned on, and when the output voltage drops to the rated voltage, the load point judging module is entered.
  • the slope Kdown of the falling is calculated according to the time ⁇ t of the output voltage drop, thereby determining the corresponding load point, and then jumping to the corresponding steady state working mode according to the load size.
  • the system's high dynamic control mode ends, enters the steady state mode, the output voltage is stable at the rated value, and the digital PID compensation algorithm is used for the traditional digital multi-mode control.
  • Figure 11 shows the dynamic results before the high-dynamic algorithm of this application is used when the load is switched from 4 ⁇ to 800 ⁇ . It can be seen from the figure that the overshoot voltage of the output voltage is 3.150V when using the traditional digital multi-mode control algorithm.
  • the dynamic recovery time is 64.32ms, the overshoot voltage is large and the recovery time is long. Moreover, when the output voltage drops to the rated value, there will be voltage fluctuations, and the dynamic effect is poor.
  • FIG. 12 is a dynamic result of using the high dynamic algorithm of the present application when the load is switched from 4 ⁇ to 800 ⁇ , which is an embodiment of the present application. It can be seen from the figure that when the high dynamic control algorithm is used, the overshoot voltage of the output voltage is 1.343V, and the dynamic recovery time is 1.888ms, which effectively reduces the overshoot voltage and greatly shortens the dynamic recovery time. Moreover, when the output voltage drops to the rated value, there is no voltage fluctuation, and the load point is judged accurately, which greatly improves the dynamic effect of the heavy load and the light load.

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Abstract

一种提高同步整流原边反馈反激式电源动态性能的方法，针对已有的小功率DCM下的动态控制算法存在的不足，尤其是重载切轻载情况下，动态恢复时间过长的缺陷，通过控制变换器原边开关管的关断和副边同步整流管以固定占空比开关，抽取输出端负载电容的能量至原边，使输出电压迅速下降，同时一直以固定的频率对输出电压进行采样，当输出电压降低至额定值时，根据负载大小判断出此时系统对应的跳出高动态模式后的稳态模式，然后系统进入稳态模式。上述方法能够将输出电压缩短在2.5ms内，并且系统在离开动态模式进入稳态模式时输出电压纹波很小，显著提高了大功率下反激变换器的动态性能。

Description

一种提高同步整流原边反馈反激式电源动态性能的方法 技术领域
本申请涉及隔离式开关电源变换器，尤其涉及一种提高同步整流原边反馈反激式电源重载切轻载动态性能的方法，适用于大功率工作时的同步整流反激电源。
背景技术
这里的陈述仅提供与本申请有关的背景信息，而不必然地构成示例性技术。
开关电源通常作为各类用电设备的电源，起到将未调整的交流或直流输入电压变换为调整后的交流或直流输出电压。由于开关电源需要适应于不同的工作条件，对电源的动态响应的性能要求越来越高。好的动态效果要求有小的电压变化以及电压恢复时间。
原边反馈(PSR)反激变换器因为其结构简单、成本低、控制灵活等优点被广泛应用在开关电源领域。PSR反激变换器动态响应性能和实际需求之间存在严重的不平衡，尤其是在大负载切换的情况下，输出电压会出现大幅度波动。为了提高动态响应性能，PSR反激变换器的研究者们提出了很多创新的方法。
有人提出了数字多模式控制算法，不同负载范围采用不同的工作模式，PWM与PFM相结合。当负载在小范围内切换时，能够实现较快、平滑切换。但是该算法各个控制模式之间只能按顺序切换，负载大范围切换时响应时间长、响应速度慢、过冲电压大。
为了能够提高全负载范围内的动态响应性能，Chong Wang等人提出了一种混合的控制方式。该控制方式通过在系统加入一个动态过程检测模块，除了稳态的多模式，另外设置了两个动态工作模式：LTH模式与HTL模式。在LTH模式与HTL模式内，均采用固定周期、固定占空比的方式进行控制。当输出电压接近理想值Vo_ref时，利用Vo的斜率进行负载点的判断，使系统进入正确的稳态工作模式。该方案有效地降低了DCM下的过冲电压和响应时间，提高了变换器的动态性能，但是也存在不足之处。满载切极轻载时，恢复时间过长，需要53ms。
另外，有的控制方法为了加快动态响应的速度，会提高PI参数来加快补偿，以此来提高动态效果，但在多模式控制对提高动态性能效果改善不大。
发明内容
根据本申请的各种实施例，提供一种提高同步整流原边反馈反激式电源动态性能的方法。
一种提高同步整流原边反馈反激式电源动态性能的方法，在原边反馈主拓扑结构下，副边采用同步整流的结构，基于数字多模式控制，设置采样模块、多模式判断模块、数字PID模块、高动态控制模块、负载点判断模块和PWM驱动模块构成的控制系统与被控的开关电源构成闭环；
1)采样模块对原边辅助绕组上的电阻分压Vsense进行采样，得到当前的输出电压值Vo_sample和输出电压与其额定值V O_REF的误差信号err＝V O_REF－Vo_sample；
2)多模式判断模块接收由采样模块采集到的输出电压值Vo_sample，输出模式判断信号mode_dynamic给高动态控制模块和负载点判断模块，判断系统进入动态工作模式或稳态工作模式中的哪一种；若输出电压值Vo_sample≥设定的输出电压上限值Vomax，多模式判断模块输出的模式判断信号mode_dynamic＝1，说明系统发生了较大范围的负载切换，系统处于由重载切到轻载的状态，系统进入动态工作模式，即HTL模式，动态工作模式下，数字PID模块不工作；否则，多模式判断模块输出的模式判断信号mode_dynamic＝0，系统进入稳态工作模式，稳态工作模式下，高动态控制模块和负载点判断模块均不工作；
3)在动态工作模式下，高动态控制模块输出占空比信号Duty_SR_dynamic给PWM驱动模块，通过PWM驱动模块关断原边主开关管，并采用固定周期Ts_ HTL、固定占空比D HTL的方式控制副边同步整流管的开关，在同步整流管开启的时间内，抽取输出端负载电容的能量至原边，使负载电容存储的能量迅速下降，从而使输出电压以固定的斜率迅速下降，采样模块一直对输出电压进行采样，当输出电压采样值Vo_sample下降至其额定值V O_REF时，系统从动态工作模式跳转至稳态工作模式；
4)负载点判断模块在动态工作模式下接收采样模块输出的输出电压值Vo_sample，通过对开关周期的计数得到输出电压的下降时间Δt，而输出电压变化量ΔVo_sample固定，则得到输出电压的斜率Kdown，输出电压的下降的斜率Kdown保持不变，而不同负载点下斜率Kdown不同，根据该斜率Kdown对应出此时的负载大小，判断出当下的负载点，进而根据当下的负载大小判断出动态过程结束时系统应处于的DPFM稳态工作模式及工作周期Ts_judge、原边峰值电流Ipeak(DPFM模式下原边峰值电流是固定的)，给出跳出状态判断信号state_judge和此时的开关周期Ts_judge，并将开关周期Ts_judge输入到PWM驱动模块，将状态判断信号state_judge输入到多模式判断模块，此后系 统便进入稳态工作模式，采用传统的数字PID控制；
5)稳态工作模式下，数字PID补偿模块根据采样模块输出的误差信号err和多模式判断模块输出的状态信号state，采用PID补偿算法，计算出控制电压补偿量V PI，多模式判断模块根据控制电压补偿量VPI的大小进行稳态多模式的切换，输出包括CCM、PWM、PFM、DPWM、DPFM中的一种稳态的state值，PWM驱动模块在稳态时，采用峰值电流控制，在不同的状态信号state下，根据控制电压补偿量VPI计算原边主开关管的周期，根据当前的输出电压采样值Vo_sample计算副边同步整流管的占空比，副边同步整流管的周期与主开关管一致；
6)PWM模块输出的占空比信号通过驱动模块得到调控开关电源主开关管以及副边同步整流管的PWM波形，实现恒压输出。
本申请还提供一种原边反馈反激式开关电源的控制系统及控制方法。
一种原边反馈反激式开关电源的控制系统，所述开关电源包括变压器原边侧、变压器副边侧及变压器辅助绕组；所述系统包括：采样模块，设置为对所述辅助绕组串联的采样电阻的电压Vsense进行采样，并根据Vsense得到所述副边侧的输出电压采样值Vo_sample，以及Vo_sample与所述副边侧的输出电压额定值V O_REF的误差信号err＝V O_REF－Vo_sample；多模式判断模块，接收由所述采样模块输出的Vo_sample，并输出模式判断信号mode_dynamic；所述多模式判断模块预设有输出电压上限值Vomax，若Vo_sample≥Vomax，则mode_dynamic＝1；若Vo_sample<Vomax,则mode_dynamic＝0；动态控制模块，接收mode_dynamic，当mode_dynamic＝1时代表所述系统进入动态工作模式，输出副边同步整流管的占空比控制信号Duty_SR_dynamic；负载点判断模块，接收mode_dynamic和采样模块输出的Vo_sample，根据Vo_sample的变化趋势判断Vo_sample下降至V O_REF的时刻，并在该时刻输出跳出状态判断信号state_judge和开关周期Ts_judge；所述多模式判断模块设置为输出状态信号state，且在接收到state_judge时输出对应稳态工作模式的state值，代表所述系统进入稳态工作模式；数字PID模块，接收state和err，根据state选择PID补偿算法，并根据err、预设的PI参数Kp和Ki得到控制电压补偿量V PI并输出，所述多模式判断模块根据VPI调整state的值；及PWM驱动模块，在接收到Duty_SR_dynamic时，控制所述原边侧的原边主开关管关断，并采用固定周期Ts_ HTL、固定占空比D HTL的方式控制所述副边侧的副边同步整流管的开关；所述PWM驱动模块还在接收到Ts_judge时，由所述动态工作模式变为稳态工作模式，所述稳态工作模式下PWM驱动模块以Ts_judge为首个周期的周期值对所述原边主开关管进行开关控制，同时根据Vo_sample计算所述副边同步整流管的占空比，对所述副边同步整流管进行开关控制，且所述副边同步整流管 的周期与所述原边主开关管一致，所述首个周期后根据state和V PI采用峰值电流控制方式对所述原边主开关管进行控制。
一种原边反馈反激式开关电源的控制方法，所述开关电源包括变压器原边侧、变压器副边侧及变压器辅助绕组，所述方法包括：对所述辅助绕组串联的采样电阻的电压Vsense进行采样；根据Vsense得到所述副边侧的输出电压采样值Vo_sample，以及Vo_sample与所述副边侧的输出电压额定值V O_REF的误差信号err＝V O_REF－Vo_sample；在Vo_sample≥Vomax时，系统进入动态工作模式，否则系统进入稳态工作模式；在动态工作模式下，PWM驱动模块关断所述原边侧的原边主开关管，并采用固定周期Ts_ HTL、固定占空比D HTL的方式控制所述副边侧的副边同步整流管的开关，当Vo_sample下降至V O_REF时，系统从所述动态工作模式跳转至所述稳态工作模式；在稳态工作模式下，数字PID模块采用PID补偿算法，并根据err、预设的PI参数Kp和Ki得到控制电压补偿量V PI，所述PWM驱动模块采用峰值电流控制，根据V PI计算所述原边主开关管的周期，根据Vo_sample计算所述副边同步整流管的占空比，对所述原边主开关管和副边同步整流管进行开关控制，且所述副边同步整流管的周期与所述原边主开关管一致。
本申请的一个或多个实施例的细节在下面的附图和描述中提出。本申请的其它特征、目的和优点将从说明书、附图以及权利要求书变得明显。
附图说明
为了更清楚地说明本申请实施例或示例性技术中的技术方案，下面将对实施例或示例性技术描述中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本申请的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他实施例的附图。
图1是本申请控制方法的系统结构框图；
图2是采样模块内部结构图；
图3是多模式判断模块内部结构图；
图4是高动态控制模块内部结构图；
图5是负载点判断模块内部结构图；
图6是数字PID模块内部结构图；
图7是PWM驱动模块内部结构图；
图8是重载切轻载HTL模式的应用示意图；
图9为重载切到轻载模式时输出电压、原边电流和副边电流的关键波形图；
图10为本申请高动态控制算法的整体流程图；
图11为负载从4Ω切换到800Ω时，未采用本申请的高动态算法前的动态结果；
图12为负载从4Ω切换到800Ω时，采用了本申请的高动态算法后的动态结果。
具体实施方式
下面将结合附图和实施例，对本申请的技术方案进行清楚、完整的描述。
为克服现有技术的局限和不足，本申请提出了一种提高同步整流原边反馈反激式电源动态性能的方法，当负载由重载向轻载切换时，可以限制输出电压的过冲在一定的范围内，并明显缩短动态恢复时间，显著提高动态性能，在多模式控制中不会引起系统的不稳定，对减小电压过冲，减小动态回复时间有很好的效果，使得系统的动态性能更优秀。
如图1为本申请提出的提高反激变换器重载切轻载动态性能的数字多模式控制算法的整体实现系统图。在原边反馈主拓扑结构下，采用了同步整流的结构，基于数字多模式控制，设置采样模块、多模式判断模块、数字PID补偿模块、动态控制模块(本实施例中在动态模式判断信号为1时进入动态模式，因此为高动态控制模块)、负载点判断模块、PWM驱动模块与被控的开关电源构成闭环系统。
采样模块对原边辅助绕组上的电阻分压Vsense进行信息采集，得到当前的输出电压采样值Vo_sample和输出电压与其额定值V O_REF的误差信号err(err＝V O_REF-Vo_sample)；多模式判断模块接收由采样模块采集到的输出电压值Vo_sample，若Vo_sample≥Vomax，输出当前动态模式判断信号mode_dynamic＝1，表明系统处于由重载切到轻载的状态(HTL模式)，并将动态模式判断信号mode_dynamic输入到高动态控制模块和负载点判断模块，进行高动态控制；高动态控制模块接收多模式判断模块输出的动态模式判断信号mode_dynamic，当mode_dynamic＝1，系统进入HTL模式时，关断原边的主开关管，通过控制副边同步整流管以固定占空比开关，使输出电压迅速下降，输出副边同步整流管的占空比信号Duty_SR_dynamic给PWM驱动模块；负载点判断模块在HTL工作模式下接收采样模块输出的输出电压值Vo_sample，由斜率计算模块计算输出电压下降斜率Kdown，根据该斜率对应得出此时的负载大小，进而根据当下的负载大小判断出动态过程结束时系统应处于的DPFM稳态工作模式及工作周期Ts_judge、原边峰值电流Ipeak(DPFM模式下原边峰值电流是固定的)，给出跳出状态判断信号state_judge和此时的开关周期Ts_judge，并将开关周期Ts_judge输入到PWM驱动模块， 将状态判断信号state_judge输入到多模式判断模块，此后便进入稳态，采用传统的数字PID控制；PWM驱动模块在动态模式HTL状态下，关断原边主开关管，接收信号Duty_SR_dynamic，采用固定周期、固定占空比的方式控制副边同步整流管的开关，得到调控主开关管和副边同步整流管的PWM波形；PWM驱动模块在稳态时，采用峰值电流控制。
图2为采样模块内部结构图。采样模块对原边辅助绕组上的电阻分压Vsense进行信息采集，包含两个比较器COMP1、COMP2、一个波形实时分析模块和一个减法器，比较器COMP1的正端连接辅助绕组上的电阻分压Vsense，负端连接波形实时分析模块输出的采样信号Vsample；比较器COMP2的正端连接Vsense，负端连接0电平，波形实时分析模块根据两个比较器的结果，输出采样信号Vsample和输出电压采样值Vo_sample，Vo_sample连接减法器的负端，输出电压额定值V O_REF连接减法器的正端得到误差信号err(err＝V O_REF-Vo_sample)。原边辅助绕组上电阻分压后的电压波形Vsense为采样模块的输入，通过两个比较器COMP1、COMP2，将Vsense与Vsample、0电压分别进行比较，分别得到comp_sample与comp_zero两个信号，波形实时分析模块通过对两个比较器的输出信号进行分析，得到当前的输出电压采样值Vo_sample，并调节采样信号Vsample的值，为下一次采样做好准备；将Vo_sample与输出电压的参考值V O_REF(固定值)做减法，得到输出电压的误差值err。
图3为多模式控制模块的内部结构图。多模式判断模块接收由采样模块采集到的输出电压值Vo_sample，包含两个比较器COMP3、COMP4、一个动态模式判断模块、稳态模式判断模块和多模式状态判断模块，比较器COMP3的正端连接设定的Vo_sample上限值Vomax，负端连接Vo_sample；比较器COMP4的正端连接Vo_sample，负端连接设定的额定电压V O_REF，动态模式判断模块根据两个比较器的结果，输出动态模式判断信号mode_dynamic。控制电压补偿量V PI和状态信号state输入稳态模式判断模块，稳态模式判断模块输出稳态状态信号state_steady。将采样模块采集到的输出电压值Vo_sample与设定的上限值Vomax(固定值)、参考值V O_REF通过比较器分别进行比较，比较结果输入动态模式判断模块进行分析，若Vo_sample>Vomax，则mode_dynamic＝1，进入动态模式，state＝HTL。在动态模式下，当Vo_sample下降至V O_REF时，mode_dynamic＝0，跳出动态模式，具体跳转到哪个稳态模式，由state_judge信号决定，state＝state_judge。在稳态模式时，mode_dynamic＝0，稳态模式判断模块根据当下的状态state和数字补偿量V PI的大小进行模式判断，判断出接下来的工作模式state_steady，并赋值给state。
图4为高动态控制模块的结构图。高动态控制模块接收模式判断模块输出 的动态模式判断信号Mode_dynamic，若Mode_dynamic＝1，进入动态模式，因为动态模式内，原边主开关管一直关断，没有能量输入，所以只需要控制副边同步整流管。而动态模式内，由代码控制产生一个固定周期Ts_ HTL、固定占空比D HTL的信号控制同步整流管，因而只输出副边同步整流管的栅极PWM信号Duty_SR_dynamic。
图5为负载点判断模块的内部结构图。包括一个比较器、参数计算模块、斜率计算模块和跳出模式判断模块。比较器COMP的正端连接Vo_sample，负端连接设定的额定电压V O_REF，比较器输出参考电压比较信号Comp_V REF，参数计算模块的输入为动态模式判断信号Mode_dynamic和输出电压采样值Vo_sample，输出为输出电压变化量ΔVo_sample和对应的时间Δt，斜率计算模块的输入为ΔVo_sample、Δt和Comp_V REF，输出为电压下降斜率Kdown，跳出模式判断模块的输入为Kdown，输出为跳出状态判断信号state_judge和此时的开关周期Ts_judge。动态模式内，state＝HTL，参数计算模块通过计数器进行计数，从而得到Δt，根据输入的Vo_sample值，得到输出电压在Δt时间内的变化量ΔVo_sample。斜率计算模块通过分析比较器的输出Comp_V REF，捕捉到Vo_sample下降至V O_REF的时刻，然后计算出输出电压的斜率Kdown，后续的跳出模式判断模块根据Kdown的值，判断出跳出动态后系统的工作模式state_judge以及第一个周期的开关周期值Ts_judge，跳出状态判断信号state_judge和跳出后开关周期信号Ts_judge只在跳出动态模式时有效。
图6为数字PID补偿模块的内部结构图。包括PI参数选择模块和数字PI补偿算法模块。PI参数选择模块的输入为状态信号state，输出为PI参数Kp和Ki，数字PI补偿算法模块的输入为Kp、Ki和误差信号err，输出为控制电压补偿量V PI。本申请采用PID补偿算法，但只采用了P调节与I调节，未用D调节。在稳态模式下，根据不同的工作模式state，选择不同的参数Kp、Ki，与当前的输出电压误差值err一起输入到后面的数字PI补偿算法模块，计算得到下一周期的控制电压补偿量V PI。动态模式下，数字PID补偿模块不工作。
图7为PWM驱动模块的内部结构图。包括一个选通模块、主开关管周期Ts计算模块、主开关管占空比控制模块和同步整流SR管占空比控制模块。状态信号State输入到选通模块，得到两个信号：稳态状态信号State_HTL和动态状态信号State_steady。选通模块判断当下的工作模式state处于稳态工作模式state_steady还是动态工作模式state_HTL。当选通模块输出为State_HTL时，主开关管占空比控制模块的输入为State_HTL和0，输出为主开关管动态占空比信号duty＝Duty_dynamic。同步整流SR管占空比控制模块的输入为State_HTL和由高动态控制模块输出的Duty_SR_dynamic，输出同步整流SR管占空比信号Duty_SR＝Duty_SR_dynamic；当输出为State_steady时，主开关管周期Ts计 算模块的输入为V PI、State_steady和Ts_judge，输出为主开关管周期信号Ts。主开关管占空比控制模块的输入为Ts和原边峰值电流Ipeak，输出为主开关管稳态占空比信号duty＝Duty_steady。同步整流SR管占空比控制模块的输入为State_steady和Vo_sample，输出SR管占空比信号Duty_SR＝Duty_SR_steady。
动态模式HTL下，主开关管一直关断，主管的栅极输入的PWM信号duty＝0；副边同步整流管栅极输入的PWM信号Duty_SR＝Duty_SR_dynamic，该信号由高动态控制模块给出。
在工作状态由动态跳转至稳态后，主开关管的第一个开关周期的周期值Ts＝Ts_judge，Ts_judge由负载点判断模块给出。而后续的开关周期值Ts的计算，和稳态模式一样。
稳态模式下采用峰值电流控制，PFM、DPFM模式下的主开关管的周期Ts根据数字补偿量V PI计算得到，其余稳态工作模式周期固定。当周期计数器的值为0时，主开关管打开，duty＝1；当原边电流达到峰值电流后，主开关管关闭，duty＝0，直到周期计数器的值为0时再次变为1，如此反复。
稳态下副边同步整流管(SR管)的周期与原边主开关管周期相等，当主开关管关断后，SR管打开，duty_SR＝1；SR管占空比控制模块根据当前的输出电压值Vo_sample计算出SR管打开的持续时间Ton_sr，当duty_SR＝1的时间达到该时间后，SR管关断，duty_SR＝0，如此反复。
稳态下副边同步整流管(SR管)的周期与原边主开关管周期相等，当主开关管关断后，SR管打开，duty_SR＝1；SR管占空比控制模块根据当前的输出电压值Vo计算出SR管打开的持续时间Ton_sr，当duty_SR＝1的时间达到该时间后，SR管关断，duty_SR＝0，如此反复。
本申请的优点及显著效果：
1、本申请提出的动态控制方法，能够在输出电压超出上限电压时，关断原边的主开关管，采用固定周期、固定占空比的方式控制副边同步整流管的开关，在同步整流管开启的时间内，抽取输出端负载电容的能量至原边，使输出电压迅速下降，使过冲电压大幅度减小，动态恢复时间大幅度减小，缩短至2.5ms内。
2、本申请提出的动态控制方法在重载切轻载模式中计算输出电压达到上限后下降至额定值的斜率，并根据斜率与负载的一一对应单调性质的关系得到负载的大小，当跳出HTL模式后，跳到对应负载点的稳态工作状态，跳变后能量与负载稳态消耗相差不大，消除了后续的电压振荡，减小动态恢复时间。另外通过斜率来确定跳出后的负载大小，能够避免跳变后能量与负载稳态消耗相差较大引起的大的电压谐振，使得重载切轻载模式与正常模式产生的振荡，电路更稳定。
3、本申请基于数字多模式控制方式，增加重载切轻载模式——HTL模式及负载点判断的方法，对一般的多模式控制环路的稳定性不会产生影响。
4、本申请能适用于同步整流结构的开关电源电路结构，具备通用性，可复用性和可移植性。
反激变换器实例的输入电压范围为90～265V，输出电压为恒压20V，电流最大为5A，变压器电感大小为417μH，变压器匝比为45:8:4，时钟频率为20MHz。负载点判断时，输出负载与电压下降斜率Kdown及对应工作状态的关系如表1所示。
表1 HTL模式下不同斜率Kdown对应跳转模式
Figure PCTCN2018125484-appb-000001
以上是该20V/5A的原边反馈反激变换器的设计实例的具体参数。
参看图8，在重载切轻载时，从该示意图可以看到当输出电压大于Vomax时，采用HTL模式。若采用PID调节则如粗虚线所示，在输出电压上升到Vomax后电压仍然会有所上升，动态恢复时间也很长；采用HTL模式，当输出电压大于Vomax时，立刻采用HTL模式，关断原边的主开关管，采用固定周期、固定占空比的方式控制副边同步整流管的开关，在同步整流管开启的时间内，抽取输出端负载电容的能量至原边，使输出电压迅速下降，当输出电压下降至额定值时，可以通过斜率大小得到输出负载大小，使得跳出HTL模式的工作模式能量与负载功耗相近，去掉后续能量不吻合引入的谐振，如实线所示；可以看到若跳出HTL模式后，若工作状态不合适导致其输入能量偏低，引入电压谐振，则如细虚线所示。
图9为重载切到轻载模式时输出电压、原边电流、副边电流和辅助绕组波形Vsense组成的关键波形图。当输出电压大于Vomax时，进入HTL模式。原边开关管关断，只有副边同步整流管以固定周期Ts_ HTL、固定占空比D HTL的方式进行开关。
当同步整流管开启时，副边电流Is反向线性增加，形成负电流，并在同步整流管关断时达到副边负电流的最大值。在此过程中抽取输出端负载电容的能量存储在励磁电感中，使输出电压迅速下降。并在同步整流管开启的时间内进行采样，辅助绕组Vsense波形也如图所示。
当同步整流管关断时，励磁电感中存储的能量耦合到原边，在原边形成负电流，通过原边主开关管的体二极管将能量返回给输入电压网络，使原边的负电流逐渐减小至零，则从副边负载电容上抽取的能量已经释放完毕。
通过控制副边同步整流管的开关，重复上述过程。以固定周期Ts_ HTL、固定占空比D HTL的方式对同步整流管进行开关控制，使输出端负载电容上存储的能量迅速减少，从而使输出电压迅速下降至额定值，进而跳转至相应的稳态工作模式。
图10为本申请高动态控制算法的整体流程图，如图所示，首先采集辅助绕组上的电压波形，接下来进行模式判断，判断系统是否进入动态模式。如果系统进入动态模式，则使原边开关管关断，以固定周期Ts_ HTL、固定占空比D HTL的方式对同步整流管进行开关控制，使输出电压以固定斜率下降(负载不变，电压下降斜率不变)。同时在同步整流管开启的时间段内通过辅助绕组波形Vsense对输出电压进行采样，当输出电压下降至额定电压时，进入负载点判断模块。根据输出电压下降的时间Δt计算出其下降的斜率Kdown，从而判断出对应的负载点，然后根据负载大小跳转至相应的稳态工作模式。这时系统的高动态控制模式结束，进入稳态工作模式，输出电压稳定在额定值，后续采用数字PID补偿算法进行传统的数字多模式控制。
图11为负载从4Ω切换到800Ω时，未采用本申请的高动态算法前的动态结果，从图中可以看出，采用传统的数字多模式控制算法时，输出电压的过冲电压为3.150V，动态恢复时间为64.32ms，过冲电压大且恢复时间长。而且当输出电压下降至额定值后还会有电压波动，动态效果较差。
图12为负载从4Ω切换到800Ω时，采用了本申请的高动态算法后的动态结果，此为本申请实施例。从图中可以看出，采用高动态控制算法时，输出电压的过冲电压为1.343V，动态恢复时间为1.888ms，有效地减小了过冲电压，且极大地缩短了动态恢复时间。而且当输出电压下降至额定值后没有电压波动，负载点判断准确，极大地提高了重载切轻载的动态效果。

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  1. 一种提高同步整流原边反馈反激式电源动态性能的方法，其中，在原边反馈主拓扑结构下，副边采用同步整流的结构，基于数字多模式控制，设置采样模块、多模式判断模块、数字PID模块、高动态控制模块、负载点判断模块和PWM驱动模块构成的控制系统与被控的开关电源构成闭环；
    1)采样模块对原边辅助绕组上的电阻分压Vsense进行采样，得到当前的输出电压值Vo_sample和输出电压与其额定值V O_REF的误差信号err＝V O_REF－Vo_sample；
    2)多模式判断模块接收由采样模块采集到的输出电压值Vo_sample，输出模式判断信号mode_dynamic给高动态控制模块和负载点判断模块，判断系统进入动态工作模式或稳态工作模式中的哪一种；若输出电压值Vo_sample≥设定的输出电压上限值Vomax，多模式判断模块输出的模式判断信号mode_dynamic＝1，说明系统发生了较大范围的负载切换，系统处于由重载切到轻载的状态，系统进入动态工作模式，即HTL模式，动态工作模式下，数字PID模块不工作；否则，多模式判断模块输出的模式判断信号mode_dynamic＝0，系统进入稳态工作模式，稳态工作模式下，高动态控制模块和负载点判断模块均不工作；
    3)在动态工作模式下，高动态控制模块输出占空比信号Duty_SR_dynamic给PWM驱动模块，通过PWM驱动模块关断原边主开关管，并采用固定周期Ts _HTL、固定占空比D HTL的方式控制副边同步整流管的开关，在同步整流管开启的时间内，抽取输出端负载电容的能量至原边，使负载电容存储的能量迅速下降，从而使输出电压以固定的斜率迅速下降，采样模块一直对输出电压进行采样，当输出电压采样值Vo_sample下降至其额定值V O_REF时，系统从动态工作模式跳转至稳态工作模式；
    4)负载点判断模块在动态工作模式下接收采样模块输出的输出电压值Vo_sample，通过对开关周期的计数得到输出电压的下降时间Δt，而输出电压变化量ΔVo_sample固定，则得到输出电压的斜率Kdown，输出电压的下降的斜率Kdown保持不变，而不同负载点下斜率Kdown不同，根据该斜率Kdown对应出此时的负载大小，判断出当下的负载点，进而根据当下的负载大小判断出动态过程结束时系统应处于的DPFM稳态工作模式及工作周期Ts_judge、原边峰值电流Ipeak，给出跳出状态判断信号state_judge和此时的开关周期Ts_judge，并将开关周期Ts_judge输入到PWM驱动模块，将状态判断信号state_judge输入到多模式判断模块，此后系统便进入稳态工作模式，采用传统的数字PID控制；
    5)稳态工作模式下，数字PID补偿模块根据采样模块输出的误差信号err和多模式判断模块输出的状态信号state，采用PID补偿算法，计算出控制电压补偿量V PI，多模式判断模块根据控制电压补偿量V PI的大小进行稳态多模式的切换，输出包括CCM、PWM、PFM、DPWM、DPFM中的一种稳态的state值，PWM驱动模块在稳态时，采用峰值电流控制，在不同的状态信号state下，根据控制电压补偿量V PI计算原边主开关管的周期，根据当前的输出电压采样值Vo_sample计算副边同步整流管的占空比，副边同步整流管的周期与主开关管一致；
    6)PWM模块输出的占空比信号通过驱动模块得到调控开关电源主开关管以及副边同步整流管的PWM波形，实现恒压输出。
  2. 根据权利要求1所述的提高同步整流原边反馈反激式电源动态性能的方法，其特征在于，4)中所述输出电压的下降斜率Kdown，是通过开关周期的计数得到输出电压下降的变化ΔVo_sample所消耗的时间Δt，计算输出电压的下降斜率Kdown＝ΔVo_sample/Δt。
  3. 根据权利要求1所述的提高同步整流原边反馈反激式电源动态性能的方法，其特征在于，所述采样模块包含两个比较器COMP1和COMP2以及波形实时分析模块和减法器，比较器COMP1和COMP2的正端均连接辅助绕组上的电阻分压Vsense，比较器COMP2的负端连接0电平，比较器COMP1和COMP2的输出连接波形实时分析模块，波形实时分析模块输出的采样信号Vsample连接比较器COMP1的负端，波形实时分析模块输出的电压采样值Vo_sample连接减法器的负端，减法器的正端连接输出电压额定值V O_REF，减法器输出误差信号err＝V O_REF－Vo_sample。
  4. 根据权利要求1所述的提高同步整流原边反馈反激式电源动态性能的方法，其特征在于，所述多模式判断模块包含两个比较器COMP3和COMP4以及动态模式判断模块、稳态模式判断模块和多模式状态判断模块；比较器COMP3的正端连接设定的Vo_sample的上限值Vomax，比较器COMP3的负端和COMP4的正端均连接采样模块中波形实时分析模块输出的电压采样值Vo_sample，比较器COMP4的负端连接设定的额定电压V O_REF，比较器COMP3和COMP4的输出连接动态模式判断模块，动态模式判断模块输出动态模式判断信号Mode_dynamic连接多模式状态判断模块的一个输入端；稳态模式判断模块的输入信号分别是原边峰值电流的数字补偿量V PI和多模式状态判断模块输出的状态信号state，稳态模式判断模块输出稳态状态信号State_steady连接多模式状态判断模块的另一个输入端，多模式状态判断模块的第三个输入端连接负载点判断模块输出的跳出状态判断信号state_judge，多模式状态判断模块的输出状态信号state连接至稳态模式判断模块的输入端。
  5. 根据权利要求1所述的提高同步整流原边反馈反激式电源动态性能的方法，其特征在于，所述高动态控制模块接收模式判断模块中动态模式判断模块输出的动态模式判断信号Mode_dynamic，输出占空比信号Duty_SR_dynamic给PWM驱动模块，通过PWM驱动模块关断原边主开关管，并由代码控制产生一个固定周期Ts _HTL、固定占空比D HTL的信号来控制副边同步整流管的开关。
  6. 根据权利要求1所述的提高同步整流原边反馈反激式电源动态性能的方法，其特征在于，所述负载点判断模块包括比较器COMP、参数计算模块、斜率计算模块和跳出模式判断模块；比较器COMP正端连接输出电压采样值Vo_sample，比较器COMP负端连接设定的额定电压V O_REF，参数计算模块的输入为动态模式判断信号Mode_dynamic和输出电压采样值Vo_sample，比较器输出的参考电压比较信号Comp_V REF以及参数计算模块输出的输出电压变化量ΔVo_sample及其对应的时间Δt均连接斜率计算模块，斜率计算模块输出电压下降斜率Kdown连接至跳出模式判断模块，跳出模式判断模块输出跳出状态判断信号state_judge和此时的开关周期Ts_judge分别给多模式判断模块和PWM驱动模块。
  7. 根据权利要求1所述的提高同步整流原边反馈反激式电源动态性能的方法，其特征在于，所述数字PID模块包括PI参数选择模块和数字PI补偿算法模块；PI参数选择模块的输入为状态信号state，PI参数选择模块的输出为PI参数Kp和Ki，数字PI补偿算法模块的输入为PI参数选择模块输出的PI参数Kp、Ki和采样模块输出的误差信号err，数字PI补偿算法模块输出控制电压补偿量V PI。
  8. 根据权利要求1所述的提高同步整流原边反馈反激式电源动态性能的方法，其特征在于，所述PWM驱动模块包括选通模块、主开关管周期Ts计算模块、主开关管占空比控制模块和同步整流管占空比控制模块；状态信号State输入到选通模块，选通模块输出稳态状态信号State_HTL和动态状态信号State_steady两者之一；当选通模块输出为State_HTL时，主开关管占空比控制模块的输入为选通模块输出的State_HTL和0，主开关管占空比控制模块的输出为主开关管动态占空比信号duty＝Duty_dynamic；同步整流管占空比控制模块的输入为选通模块输出的State_HTL和高动态控制模块输出的Duty_SR_dynamic，同步整流管占空比控制模块的输出为同步整流管占空比信号Duty_SR＝Duty_SR_dynamic；当选通模块输出为State_steady时，主开关管周期Ts计算模块的输入为VPI、State_steady和Ts_judge，输出为主开关管周期信号Ts；主开关管占空比控制模块的输入为主开关管周期Ts和原边峰值电流Ipeak，输出为主开关管稳态占空比信号duty＝Duty_steady；此时同步整流 管占空比控制模块的输入为State_steady和Vo_sample，输出同步整流管占空比信号Duty_SR＝Duty_SR_steady。
  9. 一种原边反馈反激式开关电源的控制系统，所述开关电源包括变压器原边侧、变压器副边侧及变压器辅助绕组；
    所述系统包括：
    采样模块，设置为对所述辅助绕组串联的采样电阻的电压Vsense进行采样，并根据Vsense得到所述副边侧的输出电压采样值Vo_sample，以及Vo_sample与所述副边侧的输出电压额定值V O_REF的误差信号err＝V O_REF－Vo_sample；
    多模式判断模块，接收由所述采样模块输出的Vo_sample，并输出模式判断信号mode_dynamic；所述多模式判断模块预设有输出电压上限值Vomax，若Vo_sample≥Vomax，则mode_dynamic＝1；若Vo_sample<Vomax,则mode_dynamic＝0；
    动态控制模块，接收mode_dynamic，当mode_dynamic＝1时代表所述系统进入动态工作模式，输出副边同步整流管的占空比控制信号Duty_SR_dynamic；
    负载点判断模块，接收mode_dynamic和采样模块输出的Vo_sample，根据Vo_sample的变化趋势判断Vo_sample下降至V O_REF的时刻，并在该时刻输出跳出状态判断信号state_judge和开关周期Ts_judge；所述多模式判断模块设置为输出状态信号state，且在接收到state_judge时输出对应稳态工作模式的state值，代表所述系统进入稳态工作模式；
    数字PID模块，接收state和err，根据state选择PID补偿算法，并根据err、预设的PI参数Kp和Ki得到控制电压补偿量V PI并输出，所述多模式判断模块根据V PI调整state的值；及
    PWM驱动模块，在接收到Duty_SR_dynamic时，控制所述原边侧的原边主开关管关断，并采用固定周期Ts _HTL、固定占空比D HTL的方式控制所述副边侧的副边同步整流管的开关；所述PWM驱动模块还在接收到Ts_judge时，由所述动态工作模式变为稳态工作模式，所述稳态工作模式下PWM驱动模块以Ts_judge为首个周期的周期值对所述原边主开关管进行开关控制，同时根据Vo_sample计算所述副边同步整流管的占空比，对所述副边同步整流管进行开关控制，且所述副边同步整流管的周期与所述原边主开关管一致，所述首个周期后根据state和V PI采用峰值电流控制方式对所述原边主开关管进行控制。
  10. 根据权利要求9所述的系统，其特征在于，在所述动态工作模式下，所述数字PID模块不工作；在所述稳态工作模式下，所述动态控制模块和负载 点判断模块均不工作。
  11. 根据权利要求9所述的系统，其特征在于，所述采样模块包括：
    比较器COMP1，正端连接所述采样电阻以获取Vsense；
    比较器COMP2，正端连接所述采样电阻以获取Vsense，负端连接0电平；
    波形实时分析模块，输入端连接所述比较器COMP1的输出端和比较器COMP2的输出端，所述波形实时分析模块的第一输出端输出Vo_sample，所述波形实时分析模块的第二输出端连接所述比较器COMP1的负端、输出采样信号Vsample；及
    减法器，负端连接所述波形实时分析模块的第一输出端，正端设置为获取V O_REF，输出端输出err。
  12. 根据权利要求9所述的系统，其特征在于，所述多模式判断模块包括：
    比较器COMP3，正端设置为获取Vomax，负端设置为获取Vo_sample；
    比较器COMP4，负端设置为获取V O_REF，正端设置为获取Vo_sample；
    动态模式判断模块，输入端连接所述比较器COMP3的输出端和比较器COMP4的输出端，输出端输出动态模式判断信号mode_dynamic；
    稳态模式判断模块，设置为获取V PI和state，输出端输出稳态状态信号state_steady；及
    多模式状态判断模块，第一输入端连接所述动态模式判断模块的输出端，第二输入端连接所述稳态模式判断模块的输出端，第三输入端连接所述负载点判断模块以获取state_judge，输出端输出state。
  13. 根据权利要求12所述的系统，其特征在于，所述多模式状态判断模块输出的state根据大小的不同分别对应CCM、PWM、PFM、DPWM、DPFM模式。
  14. 根据权利要求9所述的系统，其特征在于，所述负载点判断模块包括：
    比较器COMP，正端连接所述波形实时分析模块的第一输出端，负端设置为获取V O_REF，输出端输出参考电压比较信号Comp_V REF；
    参数计算模块，设置为获取mode_dynamic和Vo_sample，并输出输出电压采样值的变化量ΔVo_sample及其对应的时间Δt；
    斜率计算模块，设置为获取ΔVo_sample、Δt及Comp_V REF，并根据ΔVo_sample和Δt计算输出电压的斜率Kdown；及
    跳出模式判断模块，连接所述斜率计算模块的输出端，并输出Ts_judge和state_judge。
  15. 根据权利要求9所述的系统，其特征在于，所述数字PID模块包括：
    PI参数选择模块，设置为获取state，并根据state选择对应的PI参数Kp和Ki输出；
    数字PI补偿算法模块，设置为获取Kp、Ki、及err并根据Kp、Ki、及err得到V PI。
  16. 根据权利要求9所述的系统，其特征在于，所述PWM驱动模块包括：
    选通模块，设置为获取state，根据state的值对应选择稳态状态信号State_HTL或动态状态信号State_steady输出；
    主开关管周期Ts计算模块，设置为获取V PI、State_steady和Ts_judge，据此计算所述原边主开关管的周期信号Ts并输出；
    主开关管占空比控制模块，在所述系统处于稳态工作模式时，获取Ts和原边峰值电流Ipeak，输出duty为主开关管稳态占空比信号Duty_steady；在所述系统处于动态工作模式时，所述主开关管占空比控制模块的输入包括State_HTL，输出duty为主开关管动态占空比信号Duty_dynamic；及
    同步整流管占空比控制模块，在所述系统处于稳态工作模式时，获取State_steady和Vo_sample，输出为Duty_SR_steady；在所述系统处于动态工作模式时，获取State_HTL和Duty_SR_dynamic，输出为Duty_steady。
  17. 一种原边反馈反激式开关电源的控制方法，所述开关电源包括变压器原边侧、变压器副边侧及变压器辅助绕组，所述方法包括：
    对所述辅助绕组串联的采样电阻的电压Vsense进行采样；
    根据Vsense得到所述副边侧的输出电压采样值Vo_sample，以及Vo_sample与所述副边侧的输出电压额定值V O_REF的误差信号err＝V O_REF－Vo_sample；
    在Vo_sample≥Vomax时，系统进入动态工作模式，否则系统进入稳态工作模式；
    在动态工作模式下，PWM驱动模块关断所述原边侧的原边主开关管，并采用固定周期Ts _HTL、固定占空比D HTL的方式控制所述副边侧的副边同步整流管的开关，当Vo_sample下降至V O_REF时，系统从所述动态工作模式跳转至所述稳态工作模式；
    在稳态工作模式下，数字PID模块采用PID补偿算法，并根据err、预设的PI参数Kp和Ki得到控制电压补偿量V PI，所述PWM驱动模块采用峰值电流控制，根据V PI计算所述原边主开关管的周期，根据Vo_sample计算所述副边同步整流管的占空比，对所述原边主开关管和副边同步整流管进行开关控制，且所述副边同步整流管的周期与所述原边主开关管一致。
  18. 根据权利要求17所述的方法，其特征在于，还包括在动态工作模式下：
    获取Vo_sample，并根据其变化量ΔVo_sample和下降时间Δt，得到输出电压的斜率Kdown；
    根据Kdown得到当前的负载点；
    根据所述负载点得到动态工作模式转稳态工作模式时系统应处于的开关周期Ts_judge、原边峰值电流Ipeak；及
    将Ts_judge和Ipeak输入到PWM驱动模块，所述稳态工作模式下PWM驱动模块是以Ts_judge为首个周期的周期值对所述原边主开关管进行开关控制。
  19. 根据权利要求17所述的方法，其特征在于，所述系统从所述动态工作模式跳转至所述稳态工作模式，是跳转至DPFM稳态工作模式。
  20. 根据权利要求17所述的方法，其特征在于，在稳态工作模式下，所述副边侧的输出是恒压输出。
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