WO2016149105A1 - Construction de convertisseur de puissance flexible à circuits de régulation et réseaux de commutation - Google Patents

Construction de convertisseur de puissance flexible à circuits de régulation et réseaux de commutation Download PDF

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Publication number
WO2016149105A1
WO2016149105A1 PCT/US2016/022040 US2016022040W WO2016149105A1 WO 2016149105 A1 WO2016149105 A1 WO 2016149105A1 US 2016022040 W US2016022040 W US 2016022040W WO 2016149105 A1 WO2016149105 A1 WO 2016149105A1
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Prior art keywords
regulating
switching
circuit
switching network
power
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PCT/US2016/022040
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English (en)
Inventor
David Giuliano
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Arctic Sand Technologies, Inc.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
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Publication date
Application filed by Arctic Sand Technologies, Inc. filed Critical Arctic Sand Technologies, Inc.
Priority to KR1020177029575A priority Critical patent/KR102671328B1/ko
Priority to CN201680027105.3A priority patent/CN107580748B/zh
Priority to DE112016001188.1T priority patent/DE112016001188T5/de
Priority to CN202211455045.9A priority patent/CN115864826A/zh
Priority to JP2017567041A priority patent/JP7015172B2/ja
Publication of WO2016149105A1 publication Critical patent/WO2016149105A1/fr

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/06—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider
    • H02M3/07—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using resistors or capacitors, e.g. potential divider using capacitors charged and discharged alternately by semiconductor devices with control electrode, e.g. charge pumps

Definitions

  • This disclosure relates to power supplies, and in particular to power converters.
  • Switch- mode power converters regulate the output voltage or current by switching energy storage elements (i.e. inductors and capacitors) into different electrical configurations using a switch network.
  • Switched capacitor converters are switch-mode power converters that primarily use capacitors to transfer energy. In such converters, the number of capacitors and switches increases as the transformation ratio increases.
  • Switches in the switch network are usually active devices that are implemented with transistors. The switch network may be integrated on a single or on multiple monolithic semiconductor substrates, or formed using discrete devices.
  • Typical DC-DC converters perform voltage transformation and output regulation. This is usually done in a single-stage converter such as a buck converter. However it is possible to split these two functions into two specialized stages, namely a transformation stage, such as a switching network, and a separate regulation stage, such as a regulating circuit.
  • the transformation stage transforms one voltage into another, while the regulation stage ensures that the voltage and/or current output of the transformation stage maintains desired characteristics.
  • the invention features an apparatus for processing electric power.
  • Such an apparatus includes a power-converter having a power path for power flow between first and second power-converter terminals maintained at corresponding first and second voltages during operation thereof. The second voltage is less than the first.
  • a first regulating-circuit and a switching network are both on the power path.
  • the switching network includes a first charge-storage-element, and first and second switching-network- terminals.
  • the first regulating-circuit includes a first magnetic-storage-element and a first-regulating-circuit terminal.
  • the power path includes the first-regulating-circuit terminal, the first switching-network-terminal, and the second switching-network- terminal with the first-regulating-circuit terminal being connected to the first switching- network-terminal.
  • the switching network transitions between first and second switch- configurations. In the first charge accumulates in the first charge-storage-element at a first rate, and in the second switch configuration charge is depleted from the first charge- storage-element at a second rate.
  • the first magnetic-storage-element constrains both of these rates. In some cases, the constraint is such that the rates are equal, whereas in others, the rates are different.
  • Some embodiments also include a second regulating-circuit disposed on the path.
  • the second regulating-circuit includes a second-regulating-circuit terminal that is also on the power path. This second-regulating-circuit terminal connects to the second switching-network-terminal.
  • the switching network further includes a second charge- storage-element. Placing the switching network in the first switch-configuration depletes charge from the second charge-storage-element at a first rate. Placing the switching network in the second configuration accumulates charge in the second charge-storage- element at a second rate. The first magnetic-storage element constrains both of these rates.
  • the second regulating circuit includes a second magnetic-storage-element and a switch connected to the second magnetic-storage-element, the switch being controllable to switch between at least two switching configurations. Also among these embodiments are those in which the second regulating-circuit further includes a feedback loop for controlling operation of the switch in response to a measured output of the power converter.
  • the first magnetic-storage-element includes a filter.
  • the filter has a resonant frequency.
  • the third regulating-circuit connects to the switching network and has an inductor coupled to an inductor
  • the second regulating-circuit includes an inductor that is coupled to the inductor of the third regulating-circuit.
  • the third regulating-circuit connects to the switching network and both the second and third regulating circuits include inductors that share the same inductor core.
  • the inductors can be coupled such that the product of voltage and current at both inductors has the same sign or opposite signs.
  • the switching network includes a reconfigurable switching network.
  • a reconfigurable switching network is one that has a set of switch configurations ⁇ cti, o3 ⁇ 4 ...a n ⁇ where n>2 and the switching network is able to transition between o3 ⁇ 4 and a numerology for all m, n.
  • it includes a multi-phase switching-network. In yet others, it includes a multi-phase multiple stage switching network, or a multiple stage switching network. Still other embodiments have switching networks that include a cascade multiplier.
  • the invention can also be implemented with many kinds of regulating circuit. These include bidirectional regulating-circuits, multi-phase regulating-circuits, switch- mode power converters, resonant power converters, a buck converter, a boost converter, a buck/boost converter, a linear regulator, a Cuk converter, a fly-back converter, a forward converter, a half -bridge converter, a full-bridge converter, a magnetic-storage element, and a magnetic filter.
  • regulating circuits include bidirectional regulating-circuits, multi-phase regulating-circuits, switch- mode power converters, resonant power converters, a buck converter, a boost converter, a buck/boost converter, a linear regulator, a Cuk converter, a fly-back converter, a forward converter, a half -bridge converter, a full-bridge converter, a magnetic-storage element, and a magnetic filter.
  • the switching network receives charge at an input thereof and outputs the charge at an output thereof. In these embodiments, transport of charge from the input to the output is carried out in more than one switching cycle.
  • fly-back converter that include a quasi-resonant fly-back converter, an active-clamp fly-back converter, an interleaved flyback converter, or a two-switch fly-back converter.
  • a forward converter that feature a forward converter are those that include a multi-resonant forward converter, an active-clamp forward converter, an interleaved forward converter, or a two-switch forward converter.
  • a half-bridge converter examples include those that include an asymmetric half-bridge converter, a multi-resonant half-bridge converter, or a LLC resonant half-bridge.
  • the switching network is an AC switching network.
  • These include embodiments with a power- factor correction circuit connected to the AC switching network.
  • the power- factor correction circuit is between the AC switching network and the first regulating-circuit.
  • the power-converter varies switch configurations of the switching network at a frequency that is different from a frequency at which switching configuration of at least one of the first and second regulating-circuits is varied.
  • the switching network includes an asymmetric cascade multiplier having a plurality of DC nodes, each of which is available to deliver power at a voltage that is a multiple of the first voltage.
  • Yet other embodiments include a power-management integrated circuit into which the first regulating circuit is incorporated.
  • the power path includes a power-path section that extends out of the power-management integrated circuit and into the switching network.
  • Switches that have different physical areas.
  • switch-widths of the switches are selected such that a time constant of charge transfer between charge-storage elements of the switching network is greater than or equal to a switching frequency at which the switching network changes state.
  • the switching network is configured such that, at the switching- frequency of the switching network, increasing resistance of the switches reduces loss associated with current flowing within the switching network.
  • the various components of the apparatus need not share the same ground. In fact, one ground can float relative to the other.
  • the first-regulating-circuit receives a first voltage difference and the second power-converter terminal outputs a second voltage difference.
  • the first voltage difference is a difference between a first voltage and a second voltage that is less than the first voltage; the second voltage difference is a difference between a third voltage and a fourth voltage that is less than the third voltage.
  • a difference between the fourth voltage and the second voltage is non-zero.
  • the first-regulating-circuit receives a DC voltage difference, and the power converter receives an AC voltage difference.
  • the DC voltage is a difference between a first voltage and a second voltage that is less than the first voltage;
  • the AC voltage difference is a difference between a time-varying voltage and a constant voltage. A difference between the constant voltage and the second voltage is non-zero.
  • the invention features a method for causing a power converter to process electric power.
  • a method includes, on a power path for power flow between a first power-converter terminal and a second power-converter terminal, connecting a first-regulating-circuit terminal of a first regulating-circuit to a first switching-network-terminal of a first switching-network, placing the first switching- network in a configuration for allowing charge to accumulate in the first charge-storage- element of the first switching-network, using energy stored in a magnetic field by a first magnetic-storage-element in the first regulating-circuit, constraining a rate of charge accumulation in a first charge-storage-element of the first switching-network, using the switches in the first switching-network, placing the first switching-network in a configuration for allowing charge to be depleted from the first charge-storage-element of the first switching-network, and, using energy stored by the first magnetic-storage- element in the first regulating-circuit
  • Some practices further include connecting a second-regulating-circuit terminal of a second regulating-circuit to a second switching-network-terminal of the first switching- network, and using the second regulating-circuit, maintaining the first power-converter terminal at a first voltage, thereby maintaining the second power-converter terminal at a second voltage that is lower than the first voltage, using switches in the first switching- network.
  • Other practices include, while constraining a rate of charge depletion from the first charge-storage-element, constraining a rate of charge accumulation in a second charge-storage-element, and, while constraining a rate of charge accumulation into the first charge-storage-element, constraining a rate of charge depletion from the second charge-storage-element.
  • Yet other practices include controlling a switch connected to a magnetic-storage element of the second regulating-circuit in response to measured output of the power converter.
  • the first magnetic-storage-element includes a filter.
  • this filter has a resonant frequency.
  • a second regulating circuit includes including a third regulating-circuit that is connected to the switching network.
  • the third regulating-circuit includes an inductor
  • the first regulating-circuit includes an inductor that is coupled to the inductor of the third regulating-circuit.
  • the two inductors can be positively or negatively coupled.
  • Some practices include constraining the rate of change so that the first rate and the second rate are equal. Others include constraining the rate of change so that the first rate and the second rate are unequal.
  • practices of the invention contemplate a variety of switching networks.
  • practices of the invention include selecting the switching network to be reconfigurable switching-network, selecting it to be a multi-phase switching-network, selecting it to be a multi-phase series-parallel switching-network, selecting it to be a multi-phase multiple-stage switching network, selecting it to be a cascade multiplier, or selecting it to be a multiple stage switching network.
  • regulating circuits can be used in different practices.
  • practices of the invention include selecting a regulating circuit to be bidirectional, to be multi-phase, to be a switch-mode power converter, to be a resonant power converter, to be a magnetic-storage element, or to be a magnetic filter.
  • Other practices include selecting the switching network to be an AC switching network. Among these are practices that include controlling a power-factor of an output of the AC switching network. These include practices that include connecting a power- factor correction circuit between the AC switching network and the first regulating-circuit.
  • Yet other practices include varying switch configurations of the switching network at a frequency that is different from a frequency at which switching
  • regulating circuits can be used for at least one of the first and second regulating circuits. These include a bidirectional regulating-circuit, a multi-phase regulating-circuit, a switch-mode power converter, a resonant power converter, a buck converter, a boost converter, a buck/boost converter, a linear regulator, a Cuk converter, a fly-back converter, a forward converter, a half-bridge converter, a full- bridge converter, a magnetic-storage element, and a magnetic filter.
  • Practices that rely on a fly-back converter include those that rely on a quasi- resonant fly-back converter, an active-clamp fly-back converter, an interleaved fly-back converter, or a two-switch fly-back converter.
  • Practices that rely on a forward converter include those that rely on a multi-resonant forward converter, an active-clamp forward converter, an interleaved forward converter, or a two-switch forward converter.
  • Practices that rely on a half-bridge converter include those that rely on an asymmetric half-bridge converter, a multi-resonant half-bridge converter, or a LLC resonant half-bridge.
  • the invention features a non-transitory computer-readable medium that stores a data structure that is to be operated upon by a program executable on a computer system.
  • the data structure When operated upon by such a program, the data structure causes at least a portion of a process for fabricating an integrated circuit.
  • This integrated circuit includes circuitry described by the data structure.
  • Such circuitry includes a switching network that has been configured to be used with a power-converter having a path for power flow between a first power-converter terminal and a second power-converter terminal. During the power-converter's operation of the power-converter, the first power- converter terminal is maintained at a first voltage and the second power-converter terminal is maintained at a second voltage that is lower than the first voltage.
  • the power- converter includes a first regulating-circuit and the above-mentioned switching network, both of which are disposed on the path.
  • the switching network includes switches, and first and second switching-network-terminals.
  • the first regulating-circuit includes a first magnetic-storage-element and a first-regulating-circuit terminal.
  • the power path includes the first-regulating-circuit terminal, the first switching-network- terminal, and the second switching-network-terminal.
  • the first-regulating-circuit terminal is to be connected to the first switching-network-terminal, and the switching network is configured to transition between first and second switch-configurations.
  • the invention also includes circuitry that is described by the foregoing data structure.
  • Such circuitry includes a switching network having first and second switching terminals, and configured for disposition, along with first and second regulating circuits, at least one of which includes a magnetic storage element, on a power-flow path between first and second power converter terminals of a power converter, the first and second power converter terminals of which are maintained at corresponding first and second voltages, the second voltage being lower than the first voltage.
  • the switching network is configured to transition between switch configurations during each of which an amount of charge in a charge-storage element in the power converter changes at a rate that is constrained by the magnetic storage element.
  • the power path includes a first-regulating- circuit terminal associated with the first regulating circuit and connected to the first switching network terminal.
  • FIG. 1 A shows a DC-DC converter with a separate regulating circuit and switching
  • FIG. IB shows a bidirectional version of FIG. 1A
  • FIGS. 2-4 show DC-DC converters with alternate configurations of regulating circuits and switching networks
  • FIG. 5 shows a particular implementation of the power converter illustrated in FIG. 4;
  • FIGS. 6A and 6B show embodiments with multiple regulating circuits;
  • FIG. 7 shows an RC circuit
  • FIG 8 shows a model of a switched capacitor DC-DC converter
  • FIGS. 9A and 9B show a series-parallel SC converter operating in charge phase
  • FIG. 10 shows a series pumped symmetric cascade multiplier with diodes
  • FIG. 11 shows a parallel pumped symmetric cascade multiplier with diodes
  • FIG. 12 shows charge pump signals
  • FIG. 13 shows a two-phase symmetric series pumped cascade multiplier with switches
  • FIG. 14 shows a two-phase symmetric parallel pumped cascade multiplier with switches
  • FIG. 15 shows four different cascade multipliers along with corresponding half-wave versions
  • FIG. 16 shows output impedance of a switched capacitor converter as a function of
  • FIG. 17 shows a particular implementation of the DC-DC converter illustrated in FIG. IB with a full-wave adiabatically charged switching network
  • FIG. 18 shows the DC-DC converter illustrated in FIG. 17 during phase A
  • FIG. 19 shows the DC-DC converter illustrated in FIG. 17 during phase B
  • FIG. 20 shows various waveforms associated with a 4: 1 adiabatically charged converter
  • FIG. 21 shows adiabatic charging of series connected stages
  • FIG. 22 shows a particular implementation of the power converter illustrated in FIG. 21;
  • FIG. 23 shows an AC voltage rectified using a reconfigurable switched capacitor stage;
  • FIG. 24 shows an AC-DC power converter architecture;
  • FIG. 25 shows a particular implementation of the AC-DC converter illustrated in FIG. 24;
  • FIG. 26 shows the AC-DC converter illustrated in FIG. 25 during the positive portion of the AC cycle;
  • FIG. 27 shows the AC-DC converter illustrated in FIG. 25 during the negative portion of the AC cycle
  • FIG. 28 shows an AC-DC power converter architecture with power-factor correction
  • FIGS. 29 and 30 show particular implementations of the DC-DC converter illustrated in FIGS. 1A-1B;
  • FIGS. 31 and 32 show particular implementations of the DC-DC converter illustrated in FIG. 3;
  • FIGS. 33 and 34 show particular implementations of the DC-DC converter illustrated in FIG. 2;
  • FIGS. 35 and 36 show particular implementations of the DC-DC converter illustrated in FIG. 4.
  • FIG. 37 shows an implementation of a DC-DC converter similar to that shown in FIG. 6B.
  • FIG. 1 A shows a converter 10 having a switching network 12A connected to a voltage source 14 at an input end thereof. An input of a regulating circuit 16A is then connected to an output of the switching network 12A. A load 18A is then connected to an output of the regulating circuit 16A. Power flows between the voltage source 14 and the load 18A in the direction indicated by the arrows.
  • Embodiments described herein rely at least in part on the recognition that in a multi-stage DC-DC converter, the various constituent components can be made essentially modular and can be mixed and matched in a variety of different ways. These constituent components include switching networks and regulating circuits, the latter being made to function either as regulators or magnetic filters by simply varying the duty cycle. This modularity simplifies the assembly of such converters. As such, the configuration shown in FIG. 1 A represents only one of multiple ways to configure one or more switching networks 12A with one or more regulating circuits 16A.
  • FIG. IB shows a bidirectional version of FIG. 1A, where power can flow along a power- flow path either from a voltage source 14 to a load 18A or from the load 18A to the voltage source 14, as indicated by the arrows.
  • switching networks 12A, 12B and regulating circuits 16A, 16B There are two fundamental elements described in connection with the following embodiments: switching networks 12A, 12B and regulating circuits 16A, 16B. Assuming series connected elements of the same type are combined, there are a total of four basic building blocks. These are shown in FIGS. 1 A-4.
  • the embodiments disclosed herein include at least one of the four basic building blocks shown in FIGS. 1 A-4. More complex converter can be realized by combining the fundamental building blocks.
  • a controller not shown for clarity, will control and coordinate operation of the overall system.
  • Additional embodiments further contemplate the application of object-oriented programming concepts to the design of DC-DC converters by enabling switching networks 12A, 12B and regulating circuits 16A, 16B to be "instantiated" in a variety of different ways, so long as their inputs and outputs continue to match in a way that facilitates modular assembly of DC-DC converters having various properties.
  • the switching network 12A is instantiated as a switched charge-storage network of charge-storage elements, such as capacitors.
  • charge-storage elements such as capacitors.
  • a switched charge-storage network is also known as a switched capacitor network when the charge-storage elements are capacitors.
  • a particularly useful switched capacitor network is an adiabatically charged version of a full-wave cascade multiplier. However, diabatically charged versions can also be used.
  • charge periodically accumulates and is depleted from the charge-storage elements in a switched charge-storage network.
  • changing the charge on a capacitor adiabatically means causing an amount of charge stored in that capacitor to change by passing the charge through a non-capacitive element.
  • a positive adiabatic change in charge on the capacitor is considered adiabatic charging while a negative adiabatic change in charge on the capacitor is considered adiabatic discharging.
  • non-capacitive elements include inductors, magnetic-storage elements, such as magnetic filters, resistors, and combinations thereof.
  • a capacitor can be charged adiabatically for part of the time and diabatically for the rest of the time. Such capacitors are considered to be adiabatically charged. Similarly, in some cases, a capacitor can be discharged adiabatically for part of the time and diabatically for the rest of the time. Such capacitors are considered to be adiabatically discharged.
  • Diabatic charging includes all charging that is not adiabatic and diabatic discharging includes all discharging that is not adiabatic.
  • an adiabatically charged switching network is a switching network 12A having at least one capacitor that is both adiabatically charged and adiabatically discharged.
  • a diabatically charged switching network is a switching network 12A that is not an adiabatically charged switching network.
  • the regulating circuit 16A can be instantiated by circuitry that plays a role in somehow constraining the electrical characteristics of the system in some desirable way. For example, such a circuit might constrain the characteristic to be at some value or range of values, or constrain it to change at some rate, or constraint it to change in some direction.
  • a common example would be a regulator that constrains an output voltage or current to be at a particular value, or to be within some range of values.
  • a buck converter when combined with an appropriate feedback loop, would be an attractive candidate for such a role due to its high efficiency and speed.
  • Such a converter is also advantageous because of its ability to seamlessly transition from constraining an output voltage to be some desired value to constraining a rate of charge transfer within a switching network 12 A to be within some desired range, effectively functioning as a magnetic filter, by adjustment of its duty cycle.
  • Suitable regulating circuits 16A include boost converters, buck/boost converters, fly-back converters, forward converters, half-bridge converters, full-bridge converters, Cuk converters, resonant converters, and linear regulators.
  • the fly-back converter can be a quasi-resonant fly-back converter, an active-clamp fly-back converter, an interleaved fly-back converter, or a two-switch fly-back converter.
  • the forward converter can be a multi-resonant forward converter, an active-clamp forward converter, an interleaved forward converter, or a two-switch forward converter.
  • the half- bridge converter can be an asymmetric half-bridge converter, a multi-resonant half-bridge converter, or a LLC resonant half-bridge.
  • a voltage source 14 provides an input to a first switching network 12A, which is instantiated as a switched capacitor network.
  • the output of the first switching network 12A is a lower voltage than the input voltage that is provided to a regulating circuit 16A (e.g. a buck, a boost, or a buck/boost converter).
  • This regulating circuit 16A provides a regulated input voltage to a second switching network 12B, such as another switched capacitor network.
  • a high- voltage output of this second switching network 12B is then applied to a load 18 A.
  • An embodiment such as that shown in FIG. 2 can be configured to regulate the load 18 A or to regulate the voltage source 14 depending on the direction of energy flow along the power-flow path.
  • a low- voltage source 14 connects to an input of a regulating circuit 16 A, the output of which is provided to an input of a switching network 12Ato be boosted to a higher DC value. The output of the switching network is then provided to a load 18 A.
  • An embodiment such as that shown in FIG. 3 can be used to regulate the voltage source 14 or the load 18 A depending on the direction of energy flow along the power- flow path.
  • FIG. 4 another embodiment of a converter 100 includes a first regulating circuit 16A connected to an input 102 thereof and a second regulating circuit 16B connected to an output 104 thereof. Between the first and second regulating circuits 16A, 16B is a switching network 12A having an input 202 and an output 204.
  • the switching network 12A includes charge-storage elements 210 interconnected by switches 212. These charge-storage elements 210 are divided into first and second groups 206, 208.
  • either one of the regulating circuits 1 A, 16B can be a buck converter, which can be either configured to control a voltage or to function as a magnetic filter, a boost converter, a buck/boost converter, a fly-back converter, a Cuk converter, a resonant converter, or a linear regulator.
  • the regulating circuits 16A, 16B can be operated at a duty cycle required to achieve a desired result. For example, in the case of a buck converter, the duty cycle can be adjusted so that the buck converter's main switch maintains an indefinitely extended connection to its magnetic-storage element while its accompanying synchronous rectifier remain open indefinitely.
  • one of the two regulating circuits 16A, 16B can be replaced by a magnetic filter, thus avoiding the need for additional switches.
  • a magnetic filter includes a magnetic- storage element, such as an inductor, that resists rapid changes in current and thus promotes adiabatic charging of capacitors in the switching network 12A.
  • the switching network 12A can be a bidirectional switched capacitor network such as that shown in FIG. 5.
  • the switched capacitor network in FIG. 5 features a first capacitor 20 and a second capacitor 22 in parallel.
  • a first switch 24 selectively connects one of the first and second capacitors 20, 22 to a first regulating circuit 16A
  • a second switch 26 selectively connects one of the first and second capacitors 20, 22 to a second regulating circuit 16B.
  • the first and second regulating circuits 16 A, 16B can be operated at variable duty cycles.
  • one of the regulating circuits 16 A, 16B can be replaced by a magnetic filter having an inductor that resists rapid changes in current and thus promotes adiabatic charging of capacitors within the switching network 12A.
  • Both the first and second switches 24, 26 can be operated at high frequency, thus facilitating the adiabatic charging and discharging of the first and second capacitors 20, 22.
  • FIG. 5 has a two-phase switching network 12A.
  • other types of switching networks 12A can be used instead.
  • first, second, and third regulating circuits 16A, 16B, 16C which could be incorporated into one or more separate power management ICs, are provided at an output of a first switching network 12A for driving first, second, and third loads 18A, 18B, 18C.
  • a second switching network 12B is provided between the third load 18C and the third regulating circuit 16C thus creating a pathway similar to that shown in FIG. 2.
  • FIG. 6A provides an example of how the modular construction of regulating circuits and switching networks facilitates the ability to mix and match components to provide flexibility in DC-DC converter construction.
  • FIG. 6B the configuration shown in FIG. 6A has been reversed: first, second, and third regulating circuits 16A, 16B, 16C in FIG. 6A are replaced with first, second, and third switching networks 12A, 12B, 12C in FIG. 6B; and first and second switching networks 12A, 12B in FIG. 6A are replaced with fourth and third regulating circuits 16D, 16C in FIG. 6B.
  • first and second regulating circuits 16 A, 16B in the form of magnetic filters, that have been added to constrain charge transfer within the first and second switching networks 12A, 12B.
  • the first and second regulating circuits 16A, 16B are implemented by buck converters with appropriately selected duty cycles.
  • the first and second regulating circuits 16 A, 16B have an inductor that shares the same core, thus coupling them together. This provides a way to save space in the circuit's overall footprint.
  • a switched capacitor (SC) DC-DC power converter includes a network of switches and capacitors. By cycling the network through different topological states using these switches, one can transfer energy from an input to an output of the SC network.
  • Some converters known as “charge pumps,” can be used to produce high voltages in FLASH and other reprogrammable memories.
  • FIG. 7 shows a capacitor C initially charged to some value Vc( ).
  • Vc( ) the switch S is closed.
  • a brief surge of current flows as the capacitor C charges to its final value of V in .
  • the energy loss incurred while charging the capacitor can be found by calculating the energy dissipated in resistor R, which is
  • the total energy loss incurred in charging the capacitor is independent of its resistance R. In that case, the amount of energy loss is equal to
  • a switched capacitor converter can be modeled as an ideal transformer, as shown in FIG. 8, with a finite output resistance R 0 that accounts for the power loss incurred in charging or discharging of the energy transfer capacitors, as shown in FIG. 8. This loss is typically dissipated in the ON resistance of the MOSFETs and equivalent series resistance of the capacitors.
  • the output voltage of the switched capacitor converter is given by
  • R 0 is sensitive to the series resistance of the MOSFETs and capacitors, but is not a function of the operating frequency.
  • R 0 of the converter operating in the fast-switching limit is a function of parasitic resistance.
  • the switching period is much longer than the RC time constant ⁇ of the energy transfer capacitors.
  • ⁇ the time constant
  • This systemic energy loss arises in part because the root mean square (RMS) of the charging and discharging current is a function of the RC time constant. If the effective resistance R e ffof the charging path is reduced (i.e. reduced RC), the RMS current increases and it so happens that the total charging energy loss is independent of R e ff.
  • RMS root mean square
  • a switched capacitor network It is desirable for a switched capacitor network to have a common ground, large transformation ratio, low switch stress, low DC capacitor voltage, and low output resistance.
  • More useful topologies are: Ladder, Dickson, Series-Parallel, Fibonacci, and Doubler.
  • FIGS. 9A and 9B show a 2: 1 series-parallel switched capacitor converter operating in charge phase and in discharge phase, respectively.
  • the capacitors are in series.
  • the capacitors are in parallel.
  • FIGS. 10 and 11 Other useful topologies are cascade multiplier topologies, as shown in FIGS. 10 and 11.
  • a source is located at V ⁇ and a load is located at V 2 .
  • packets of charge are pumped along a diode chain as the coupling capacitors are successively charged and discharged.
  • clock signals v clk and v cik with amplitude v pump are 180 degrees out of phase.
  • the coupling capacitors can either be pumped in series or in parallel.
  • the foregoing topologies are suitable for stepping up voltage, they can also be used to step down voltage by switching the location of the source and the load.
  • the diodes can be replaced with controlled switches such as MOSFETs and BJTs.
  • the foregoing cascade multipliers are half-wave multipliers in which charge is transferred during one phase of the of the clock signal. This causes a discontinuous input current.
  • Both of these cascade multipliers can be converted into full-wave multipliers by connecting two half-wave multipliers in parallel and running the half-wave multipliers 180 degrees out of phase.
  • FIG. 13 shows a full-wave symmetric series pumped cascade multiplier version
  • FIG. 14 shows a full- wave symmetric parallel pumped cascade multiplier version.
  • the switches in FIG. 13 and FIG. 14 are bidirectional. As a result, in both of these cascade multipliers, power can flow either from the source to the load or from the load to the source.
  • Asymmetric multipliers can also be converted into full-wave multipliers
  • FIG. 15 shows four different step-down versions of full-wave multipliers along with their corresponding half-wave versions. Furthermore, it is possible to combine N phases in parallel and run them 180 degrees/N out of phase to reduce output voltage ripple and increase output power handling capability.
  • the asymmetric multipliers have a special property: they contain DC nodes that are at voltage levels that are multiples of 2 . These DC nodes can serve as tap points for delivering or drawing power. They also provide convenient locations at which to reference V ⁇ . This permits one to separate the grounds.
  • the basic building blocks in the modular architecture shown FIGS. 1A-4 can either be connected as independent entities or coupled entities.
  • switching networks and regulating circuits are tightly coupled, it is possible to prevent and/or reduce the systemic energy loss mechanism of the switching networks through adiabatic charging.
  • This generally includes using the regulating circuit to control the charging and discharging of the capacitors in the switching network.
  • the output voltage of the regulating circuit and thus the total converter can be regulated in response to external stimuli.
  • One approach to regulating the output voltage is by controlling the average DC current in a magnetic-storage element, such as that found in a magnetic filter.
  • a desirable feature of the regulating circuit is to constrain the root mean square (RMS) current through the capacitors in the switching network to be below some limit.
  • RMS root mean square
  • a regulating circuit achieves such a constraint by using either resistive or magnetic-storage elements.
  • resistive elements would consume power so their use is less desirable. Therefore, embodiments described herein rely on a magnetic-storage element with optional switches in the regulating circuit.
  • the regulating circuit limits the RMS current by forcing the capacitor current through the magnetic-storage element in the regulating circuit that has an average DC current.
  • the switches are operated so as to maintain an average DC current through the magnetic-storage element. This can be achieved by varying the duty cycle of a switch in series with the magnetic-storage element. In one embodiment, the duty cycle approaches zero so that at least one switch is effectively always on. In the limiting case, at least one switch can be eliminated altogether.
  • the regulating circuit may limit both the RMS charging current and the RMS discharging current of at least one capacitor in the switching network.
  • One single regulating circuit may limit the current in or out of the switching network by sinking and/or sourcing current. Therefore, there are four fundamental configurations, which are shown in FIGS. 1 A-4. Assuming power flows from the source to load then, in FIG. 1 A, the regulating circuit 16A may sink both the charging and discharging current of the switching network 12A. In FIG. 3, the regulating circuit 16A may source both the charging and discharging current of the switching network 12A. In FIG. 4, the regulating circuit 16A may source the charging current of the switching network 12 A and the regulating circuit 16B may sink the discharging current of the same switching network 12A and vice- versa.
  • the regulating circuit 16A may source both the charging and discharging current of the switching network 12B while also sinking both the charging and discharging current of the switching network 12A. Furthermore, if both the switching networks 12A, 12B and the regulating circuits 16A, 16B allow power to flow in both directions then bidirectional power flow is possible (source to load and load to source).
  • One embodiment relies on at least partially adiabatically charging full-wave cascade multipliers.
  • a particularly preferred switching network because of its superior fast- switching limit impedance, the ease with which it can be scaled up in voltage, and its low switch stress, is the cascade multiplier.
  • the coupling capacitors are typically pumped with a clocked voltage source v clk & v ⁇ .
  • the coupling capacitors are pumped with a clocked current source i clk & instead, then the RMS charging and discharging current in the coupling capacitor may be limited.
  • the capacitors are at least partially charged adiabatically thus lowering, if not eliminating, the l/2C*AVc 2 loss that is associated with a switched capacitor converter when operated in the slow- switching limit. This has the effect of lowering the output impedance to the fast-switching limit impedance. As shown by the black dotted line in FIG. 16, which depicts adiabatic operation, under full adiabatic charging, the output impedance would no longer be a function of switching frequency.
  • an adiabatically charged switched capacitor converter can operate at a much lower switching frequency than a conventionally charged switched capacitor converter, but at higher efficiency.
  • an adiabatically charged switched capacitor converter can operate at the same frequency and with the same efficiency as a conventionally charged switched capacitor converter, but with much smaller coupling capacitors, for example, between four and ten times smaller.
  • FIG. 17 shows a step-down converter consistent with the architecture shown in FIG. IB.
  • the switching network 12A is adiabatically charged using the regulating circuit 16 A.
  • the clocked current sources i clk & are emulated by four switches and the regulating circuit 16A.
  • the output capacitor Co has also been removed so as to allow V x to swing.
  • the regulating circuit 16A is a boost converter that behaves as constant source with a small AC ripple. Any power converter that has a non-capacitive input impedance at the frequency of operation would have allowed adiabatic operation.
  • switch-mode power converters are attractive candidates due to their high efficiency, linear regulators are also practical.
  • the act of closing switches labeled “1” charges the capacitors C 4 , C 5 , and C while discharging the capacitors d, C 2 , and C 3 .
  • the act of closing switches labeled “2” has the complementary effect.
  • the first topological state (phase A) is shown in FIG. 18, where all switches labeled “1” are closed and all switches labeled "2” are opened.
  • the second topological state (phase B) is shown in FIG. 19, where all switches labeled "2" are closed and all switches labeled "1” are opened.
  • the regulating circuit 16A limits the RMS charging and discharging current of each capacitor.
  • the capacitor C 3 is discharged through the magnetic filtering element in the regulating circuit 16A during phase A, while the capacitor C 3 is charged through the magnetic filtering element in the regulating circuit 16A during phase B, clearly demonstrating the adiabatic concept.
  • all of the active components are implemented with switches so the converter can process power in both directions.
  • FIG. 20 A few representative node voltages and currents are shown in FIG. 20. There is a slight amount of distortion on the rising and falling edges of the two illustrated currents (/ >i and Ip2), but for the most part, the currents resemble two clocks 180 degrees out of phase.
  • adiabatic charging occurs in cascade multipliers only if at least one end of a switch stack is not loaded with a large capacitance, as is the case in this embodiment, where the Vx node is only loaded down by the regulating circuit 16A.
  • the switches shown in FIG. 17 will transition between states at some switching frequency.
  • One way to constrain the RMS current is to correctly choose the resistances of the switches.
  • the resistances should be high enough so that the RC time constant of the charge transfer between the capacitors is similar to, or longer than, the switching frequency.
  • the switching network 12A can be forced into the fast-switching limit region.
  • the regulating circuit 16 A allows us to reduce the resistance of the switches while still operating adiabatically. Therefore, the switches can be optimally sized for the highest efficiency without worrying about constraining the RMS current since it is handled by the regulating circuit 16A (or optionally a magnetic filter).
  • the optimal size for each switch is chosen by balancing the resistive and capacitive losses in each switch at a given switching frequency and at a given current.
  • the modular architecture with the basic building blocks shown in FIGS. 1A-4 may be expanded to cover a wider range of applications, such as high- voltage DC, AC- DC, buck-boost, and multiple output voltages. Each of these applications includes separating the transformation, regulation, and possibly magnetic filtering functions. Extension of the architecture can also incorporate adiabatically charged switched capacitor converters. In many switched capacitor converters, the number of capacitors and switches increases linearly with the transformation ratio. Thus, a large number of capacitors and switches are required if the transformation ratio is large. Alternatively, a large
  • transformation ratio can be achieved by connecting numerous low gain stages in series as depicted in FIG. 21.
  • the transformation ratio of the total switch capacitor stack ( V in /V x ) is as follows:
  • transformation ratio can be easily changed by bypassing one or more stages.
  • Adiabatic charging of a preceding series-connected switching network only occurs if the following switching network controls the charging and discharging current of the preceding stage.
  • FIG. 22 shows a converter with a first switching network 12A connected in series with a second switching network 12D consistent with the architecture shown in FIG. 21.
  • Both the first and second switching networks 12A, 12D are two-phase cascade multipliers.
  • switches labeled “1” and “2” are always in complementary states and switches labeled "7" and “8” are always in complementary states.
  • switches labeled "1" and “2" are always in complementary states and switches labeled "7" and “8” are always in complementary states.
  • switches labeled "1” are open and all switches labeled "2" are closed.
  • In a second switched-state all switches labeled "1” are closed and all switches labeled "2" are opened.
  • closing switches 1 charges capacitors Ci, C 2 , C 3 , while discharging capacitors C 4 , C 5 , C and closing switches 2 has the complementary effect.
  • closing switches 7 charges capacitors C 7 , Cg, C while discharging capacitors Ci 0 , Cn , Ci 2 and closing switches 8 has the complementary effect.
  • the power converter provides a total step-down of 32: 1, assuming the first regulating circuit 16 A is a buck converter with a nominal step-down ratio of 2: 1.
  • the switches in the first switching network 12A will need to block 8 volts while the switches in the second switching network 12D will need to block 2 volts.
  • the modular architecture with the basic building blocks shown in FIGS. 1 A-4 may be configured to handle an AC input voltage as well.
  • One of the main attributes of switched capacitor converters is their ability to operate efficiency over a large input range by reconfiguring the switched capacitor network. If the AC wall voltage (i.e. 60 Hz & 120 VRMS) can be thought of as a slow moving DC voltage, then a front-end switched capacitor stage 13A, also known as an AC switching network, should be able to unfold the time-varying input voltage into a relatively stable DC voltage.
  • FIG. 23 A diagram of a 120 VRMS AC waveform over a single 60 Hz cycle overlaid with the unfolded DC voltage is shown in FIG. 23.
  • the AC switching network 13 A has different configurations (1/3, 1/2, 1/1) at its disposal along with an inverting stage. It was also designed to keep the DC voltage under 60 V.
  • a regulating circuit 16A shown in FIG. 24, to produce a final output voltage. It may also be necessary to place another switching network between the AC switching network 13A and the regulating circuit 16A to further condition the voltage. If this is the case, then the caveats for series-connected stages hold true since the AC switching network 13A is a special purpose switching network.
  • Some form of magnetic or electric isolation is also common in AC-DC converters for safety reasons.
  • voltages: VAC, VDC, and Vo are purposely defined as being agnostic to a common ground.
  • FIG. 25 shows an AC-DC converter corresponding to the architecture shown in FIG. 24.
  • the AC switching network 13A is a synchronous AC bridge rectifier followed by a reconfigurable two-phase step-down cascade multiplier with three distinct conversion ratios (1/3, 1/2, 1/1) while the regulating circuit 16A is a synchronous buck converter.
  • switches labeled "7" and "8" are always in complementary states. During the positive portion of the AC cycle (0 to ⁇ radians) all switches labeled "7” are closed while all switches labeled "8” are opened as shown in FIG. 26. Similarly, during the negative portion of the AC cycle ( ⁇ to 2 ⁇ radians) all switches labeled "8” are closed while all switches labeled "7” are opened as shown in FIG. 27.
  • switches 1A - IE and switches 2 A - 2E may be selectively opened and closed as shown in Table 1 to provide three distinct conversion ratios of: 1/3, 1/2, and 1.
  • the AC switching network 13A is provided with a digital clock signal CLK.
  • a second signal CLKB is also generated, which may simply be the complement of CLK (i.e., is high when CLK is low and low when CLK is high), or which may be generated as a non-overlapping complement.
  • the AC switching network 13A provides a step-down ratio of one-third (1/3).
  • the AC switching network 13A provides a step-down ratio of one-half (1/2).
  • the AC switching network 13A provides a step-down ratio of one.
  • Power factor is a dimensionless number between 0 and 1 that defines a ratio of the real power flowing to apparent power.
  • a common way to control the harmonic current and thus boost the power factor is by using an active power factor corrector, as shown in FIG. 28.
  • a power-factor correction circuit 17A causes the input current to be in phase with the line voltage, thus causing reactive power consumption to be zero.
  • FIGS. 29-36 show specific implementations of power converters that conform to the architectural diagrams shown in FIGS. 1 A-4.
  • a regulating circuit or multiple regulating circuits which may include magnetic filters, may limit both the RMS charging current and the RMS discharging current of at least one capacitor in each switching network so all of these switching networks are adiabatically charged switching networks.
  • decoupling capacitors 9A or 9B are present, then the ability of the regulating circuit to limit the RMS charging and discharging current may be diminished.
  • Capacitors 9A and 9B are optional and to keep the output voltage fairly constant, a capacitor Co is used. All of the stages share a common ground. However, this need not be case.
  • a regulating circuit 16A is implemented as a fly-back converter, then the ground can be separated easily. Even a switching network 12 A can have separate grounds through capacitive isolation. Furthermore, for simplicity, the switching network in each implementation has a single conversion ratio. However, reconfigurable switching networks that provide power conversion at multiple distinct conversion ratios may be used instead.
  • switches labeled "1" and “2" are always in complementary states. Thus, in a first switched-state, all switches labeled “1” are open and all switches labeled “2" are closed. In a second switched-state, all switches labeled “1” are closed and all switches labeled “2” are opened. Similarly, switches labeled "3" and "4" are in
  • switches labeled "5" and “6” are in complementary states
  • switches labeled “7” and “8” are in complementary states.
  • the regulating circuits operate at higher switching frequencies than the switching networks. However, there is no requirement on the switching frequencies between and amongst the switching networks and regulating circuits.
  • FIG. 29 shows a step-up converter corresponding to the architecture shown in FIG. 1 A.
  • the switching network 12 A is a two-phase step-up cascade multiplier with a conversion ratio of 1 :3 while the regulating circuit 16A is a two-phase boost converter.
  • closing switches 1 and opening switches 2 charges capacitors C 3 and C 4 while discharging capacitors C and C 2 .
  • opening switches 1 and closing switches 2 charges the capacitors Ci and C 2 while discharging the capacitors C 3 and C 4 .
  • FIG. 30 shows a bidirectional step-down converter corresponding to the architecture shown in FIG. IB.
  • the switching network 12A is a two- phase step-down cascade multiplier with a conversion ratio of 4: 1 while the regulating circuit 16A is a synchronous buck converter.
  • closing switches 1 and opening switches 2 charges capacitors Cj, C 2 , and C 3 while discharging capacitors C 4 , C 5 , and C 6 .
  • opening switches 1 and closing switches 2 charges the capacitors C 4 , C 5 , and C 6 while discharging the capacitors C ⁇ , C 2 , and C 3 . All of the active components are implemented with switches so the converter can process power in both directions.
  • FIG. 31 shows a step-up converter consistent with the architecture shown in FIG. 3.
  • the regulating circuit 16A is a boost converter while the switching network 12A is a two-phase step-up series-parallel SC converter with a conversion ratio of 1 : 2.
  • closing switches 1 charges a capacitor C 2 while discharging a capacitor Ci.
  • Closing switches 2 has the complementary effect.
  • FIG. 32 shows a bidirectional up-down converter consistent with the architecture shown in FIG. 3.
  • the regulating circuit 16A is a synchronous four switch buck-boost converter while the switching network 12A is a two-phase step-up cascade multiplier with a conversion ratio of 1 :4.
  • closing switches 1 charges capacitors C 4 , C 5 , and C 6 while discharging capacitors Ci, C 2 , and C 3 .
  • Closing switches 2 has the complementary effect. All of the active components are implemented with switches so the converter can process power in both directions.
  • FIG. 33 shows an inverting up-down converter consistent with the architecture shown in FIG. 2.
  • the first switching network 12A is a step-down series-parallel SC converter with a conversion ratio of 2: 1 ;
  • the first regulating circuit 16A is a buck/boost converter;
  • the second switching network 12B is a step-up series- parallel SC converter with a conversion ratio of 1 :2.
  • closing switches 1 charges a capacitor Ci while closing switches 2 discharges the capacitor C ⁇ .
  • closing switches 7 discharges a capacitor C 2 while closing switches 8 charges the capacitor C 2 .
  • FIG. 34 shows a bidirectional inverting up-down converter consistent with the architecture shown in FIG. 2.
  • the first switching network 12A is a two-phase step-down series-parallel SC converter with a conversion ratio of 2:1 ;
  • the first regulating circuit 16A is a synchronous buck/boost converter;
  • the second switching network 12B is a two-phase step-up series-parallel SC converter with a conversion ratio of 1 :2.
  • closing switches 1 charges a capacitor while discharging a capacitor C 2 .
  • Closing switches 2 has the complementary effect.
  • closing switches 7 charges a capacitor C 4 while discharging a capacitor C 3 .
  • Closing switches 8 has the complementary effect. All of the active components are implemented with switches so the converter can process power in both directions.
  • FIG. 35 shows an up-down converter consistent with the block diagram shown in FIG. 4.
  • the first regulating circuit 16A is a boost converter
  • the first switching network 12A is a two-phase step-up series-parallel SC converter with a conversion ratio of 1 :2
  • the second regulating circuit 16B is a boost converter.
  • closing switches 1 charges capacitors Ci and C 2 while simultaneously discharging capacitors C 3 and C 4 .
  • Closing switches 2 has the complementary effect.
  • FIG. 36 shows a bidirectional up-down converter consistent with the block diagram shown in FIG. 4.
  • the first regulating circuit 16A is a synchronous boost converter
  • the first switching network 12A is a two-phase fractional step-down series-parallel SC converter with a conversion ratio of 3:2
  • the second regulating circuit 16B is a synchronous buck converter.
  • closing switches 1 charges capacitors C 3 and C 4 while simultaneously discharging capacitors Ci and C 2 .
  • Closing switches 2 has the complementary effect. All of the active components are implemented with switches so the converter can process power in both directions.
  • Adjusting the duty cycle of the second regulating circuit 16B so that switch 6 remains closed for extended periods allows an inductor L 2 to promote adiabatic charge transfer between capacitors in the first switching network 12A.
  • the switches 5, 6 can be dispensed with, thus reducing the overall chip area required to implement the second regulating circuit 16B.
  • FIG. 37 shows a step-down converter that conforms substantially to the
  • a fourth regulating circuit 16D has coupled inductors Li, L 2 .
  • the fourth regulating circuit 16D regulates first and second switching networks 12A, 12B in parallel that operate 90° out of phase.
  • the task of constraining charge transfer among the four capacitors C 0 of the first and second switching network 12A, 12B is shared by first and second regulating circuits 16A, 16B that also share coupled inductors L 3 , L 4 . If the coupling factor of the coupled inductors L 3 , L 4 is set properly, the ripple current through these inductors can be reduced.
  • the topology of the regulating circuit can be any type of power converter with the ability to regulate the output voltage, including, but without limitation, synchronous buck, three-level synchronous buck, SEPIC, magnetic filters, and soft switched or resonant converters.
  • the switching networks can be realized with a variety of switched capacitor topologies, depending on desired voltage
  • a computer accessible storage medium includes a database representative of one or more components of the converter.
  • the database may include data representative of a switching network that has been optimized to promote low-loss operation of a charge pump.
  • a computer accessible storage medium may include any non- transitory storage media accessible by a computer during use to provide instructions and/or data to the computer.
  • a computer accessible storage medium may include storage media such as magnetic or optical disks and semiconductor memories.
  • a database representative of the system may be a database or other data structure that can be read by a program and used, directly or indirectly, to fabricate the hardware comprising the system.
  • the database may be a behavioral-level description or register-transfer level (RTL) description of the hardware functionality in a high level design language (HDL) such as Verilog or VHDL.
  • the description may be read by a synthesis tool that may synthesize the description to produce a netlist comprising a list of gates from a synthesis library.
  • the netlist comprises a set of gates that also represent the functionality of the hardware comprising the system.
  • the netlist may then be placed and routed to produce a data set describing geometric shapes to be applied to masks.
  • the masks may then be used in various semiconductor fabrication steps to produce a semiconductor circuit or circuits corresponding to the system.
  • the database may itself be the netlist (with or without the synthesis library) or the data set.

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Abstract

L'invention concerne un appareil de traitement de puissance électrique qui comprend un convertisseur de puissance ayant un chemin pour le transit de puissance entre des première et seconde bornes de convertisseur de puissance. Pendant le fonctionnement, les première et seconde bornes de convertisseur de puissance sont maintenues à des première et seconde tensions respectives. Un circuit de régulation et un réseau de commutation sont disposés sur le chemin. Le premier circuit de régulation comprend un élément de stockage magnétique et une borne de premier circuit de régulation. La borne de premier circuit de régulation est connectée à la première borne de réseau de commutation. Le réseau de commutation effectue une transition entre une première configuration de commutation et une seconde configuration de commutation. Dans la première configuration de commutation, une charge s'accumule dans le premier élément de stockage de charge à une première vitesse. Inversement, dans la seconde configuration de commutation, une charge est épuisée dans le premier élément de stockage de charge à une seconde vitesse. Ces vitesses sont contraintes par l'élément de stockage magnétique.
PCT/US2016/022040 2015-03-13 2016-03-11 Construction de convertisseur de puissance flexible à circuits de régulation et réseaux de commutation WO2016149105A1 (fr)

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CN201680027105.3A CN107580748B (zh) 2015-03-13 2016-03-11 具有调节电路和开关网络的灵活的电能转换器结构
DE112016001188.1T DE112016001188T5 (de) 2015-03-13 2016-03-11 Konstruktion flexibler Stromrichter mit Regelkreisen und Schaltnetzen
CN202211455045.9A CN115864826A (zh) 2015-03-13 2016-03-11 具有调节电路和开关网络的灵活的电能转换器结构
JP2017567041A JP7015172B2 (ja) 2015-03-13 2016-03-11 調整回路及びスイッチングネットワークを備えるフレキシブルな電力変換器構造

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CN107580748A (zh) 2018-01-12
JP2018508178A (ja) 2018-03-22
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DE112016001188T5 (de) 2018-03-08
KR20180004116A (ko) 2018-01-10

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