WO2016127479A1 - 一种导航信号数据导频联合跟踪方法及装置 - Google Patents

一种导航信号数据导频联合跟踪方法及装置 Download PDF

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WO2016127479A1
WO2016127479A1 PCT/CN2015/075671 CN2015075671W WO2016127479A1 WO 2016127479 A1 WO2016127479 A1 WO 2016127479A1 CN 2015075671 W CN2015075671 W CN 2015075671W WO 2016127479 A1 WO2016127479 A1 WO 2016127479A1
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data
carrier
phase
pilot
signal
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PCT/CN2015/075671
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English (en)
French (fr)
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魏蛟龙
唐祖平
石建峰
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华中科技大学
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Priority to EP15881640.5A priority Critical patent/EP3258293B1/en
Priority to JP2017520470A priority patent/JP6389331B2/ja
Publication of WO2016127479A1 publication Critical patent/WO2016127479A1/zh
Priority to US15/586,254 priority patent/US10649095B2/en

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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S19/00—Satellite radio beacon positioning systems; Determining position, velocity or attitude using signals transmitted by such systems
    • G01S19/01—Satellite radio beacon positioning systems transmitting time-stamped messages, e.g. GPS [Global Positioning System], GLONASS [Global Orbiting Navigation Satellite System] or GALILEO
    • G01S19/13—Receivers
    • G01S19/24—Acquisition or tracking or demodulation of signals transmitted by the system
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S19/00—Satellite radio beacon positioning systems; Determining position, velocity or attitude using signals transmitted by such systems
    • G01S19/01—Satellite radio beacon positioning systems transmitting time-stamped messages, e.g. GPS [Global Positioning System], GLONASS [Global Orbiting Navigation Satellite System] or GALILEO
    • G01S19/13—Receivers
    • G01S19/24—Acquisition or tracking or demodulation of signals transmitted by the system
    • G01S19/29—Acquisition or tracking or demodulation of signals transmitted by the system carrier including Doppler, related
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S19/00—Satellite radio beacon positioning systems; Determining position, velocity or attitude using signals transmitted by such systems
    • G01S19/01—Satellite radio beacon positioning systems transmitting time-stamped messages, e.g. GPS [Global Positioning System], GLONASS [Global Orbiting Navigation Satellite System] or GALILEO
    • G01S19/13—Receivers
    • G01S19/24—Acquisition or tracking or demodulation of signals transmitted by the system
    • G01S19/30—Acquisition or tracking or demodulation of signals transmitted by the system code related

Definitions

  • the invention belongs to the technical field of navigation signal tracking, and more particularly to a navigation signal data pilot joint tracking method and device.
  • GNSS Global Navigation Satellite System
  • the navigation signal system introduces many new technologies. Compared with traditional single data channel navigation signals, the introduction of pilot channels is the biggest feature of modern navigation signal systems. Analysis of the GPS and Galileo ICD documents revealed that most of the newly designed navigation signals introduced pilot components, including: GPS L1C, GPS L2C, GPS L5C, Galileo E1 OS, Galileo E5, Galileo E6.
  • the introduction of the pilot channel greatly improves the tracking performance of the navigation signal. Since the pilot signal has no modulated data bits, there is no problem of data bit flipping, so when tracking the pilot signal, a longer coherent integration time can be used, thereby improving the tracking accuracy.
  • the carrier tracking of the pilot channel can use a pure phase-locked loop (Pure PLL) that is sensitive to data bit flipping. However, since the data channel has a data bit flip, its carrier tracking can only be used to be insensitive to data bit flipping.
  • the Costas loop The dynamic traction range of the pure phase-locked loop is 360°, the corresponding phase noise 1-sigma experience threshold is 45°, and the dynamic traction range of the Costas loop is only 180°, and the corresponding phase noise 1-sigma experience threshold is 15°.
  • the tracking sensitivity of the pure phase-locked loop is better than that of the Costas ring.
  • this single-pilot tracking method brings another problem: the signal of the data channel does not participate in tracking, resulting in energy loss. Therefore, how to combine data and pilots in the tracking process has become a research hotspot of modern navigation signal tracking.
  • the present invention provides a universal navigation signal data pilot joint tracking method and apparatus with low complexity, high tracking accuracy and high tracking sensitivity, which is used for a pilot channel with pilot channels. Navigation signal tracking and provides good tracking performance.
  • a navigation signal data pilot joint tracking method including the following steps:
  • Step 1 multiplies the digital intermediate frequency signal by the local carrier to complete carrier stripping, and the local carrier adopts two branches of the same direction and orthogonality;
  • Step 2 The carrier stripped signal is multiplied by the data baseband signal and the pilot baseband signal respectively to complete the code stripping, and the baseband signal adopts three branches of advance, immediate and hysteresis, wherein the baseband signal includes the data baseband signal and the Said pilot baseband signal;
  • the signal after step 3 code stripping is processed by integration and clearing to obtain the coherent integration result of each branch.
  • the coherent integration result of the pilot channel includes: in-phase leading I PE , in-phase instant I PP , in-phase lag I PL , positive
  • the intersection of pre-existing Q PE , quadrature instantaneous Q PP and quadrature lag Q PL , the coherent integration results of the data channel include: in-phase leading I DE , in-phase instant I DP , in-phase lag I DL , orthogonal leading Q DE , orthogonal instant Q DP and orthogonal lag Q DL ;
  • Step 5 calculating an optimal power weighting factor by using a power comparison between the data channel and the pilot channel Wherein P 1 and P 2 represent signal power of the data channel and the pilot channel, respectively;
  • Step 6 calculates a probability weighting factor ⁇ 2 by using the coherent integration result of each instant branch, and the probability weighting factor ⁇ 2 is a weighting factor that is monotonic to the correct probability of the data inversion detection;
  • Step 7 calculates the in-phase instant I P and quadrature instantaneous Q P coherent integration results as follows:
  • I P I PP +Flip ⁇ I DP ⁇ 1
  • Q P Q PP +Flip ⁇ Q DP ⁇ 1
  • the in-phase instant I P and the orthogonal real-time Q P coherent integration result are phase-detected to obtain a phase-detecting output. And output the phase detector And multiplying the probability weighting factor ⁇ 2 , and then performing loop pilot filtering to obtain a data pilot joint carrier adjustment amount, wherein the data pilot and the carrier adjustment amount are fed back to a carrier digital control oscillator that controls the local carrier, The carrier digitally controlled oscillator is adjusted to implement tracking of the data pilot joint signal carrier;
  • Step 8 calculates the coherent integration results of the data pilot combined with the in-phase leading I E , the in-phase lag I L , the orthogonal leading Q E and the orthogonal lag Q L as follows:
  • I E I PE +Flip ⁇ I DE ⁇ 1
  • I L I PL +Flip ⁇ I DL ⁇ 1
  • the in-phase lead I E , the in-phase lag I L , the orthogonal lead Q E and the orthogonal lag Q L coherent integration result are phase-detected to obtain a phase-detection result ⁇ , and the phase-detection result ⁇ Multiplying by the probability weighting factor ⁇ 2 , and then loop filtering to obtain a data pilot joint code adjustment amount, the code adjustment amount being fed back to a code digital control oscillator for controlling the baseband signal, and the code digital control oscillator Adjustments are made to achieve tracking of the baseband signal.
  • a navigation signal data pilot joint tracking apparatus including: a carrier stripper, a code stripper, a local carrier generator, a baseband signal generator, a carrier digital controlled oscillator, and a code digital control oscillation. , integration and clearer, data flip detector, probability weighting factor calculator, power weighting factor calculator, data pilot joint carrier adjustment estimator, and data pilot joint code adjustment estimator, wherein:
  • the carrier digitally controlled oscillator is configured to control a local carrier generator to generate a local carrier; the code digital control oscillator is configured to control a baseband signal generator to generate a baseband signal, where the baseband signal includes a data baseband signal and a pilot baseband signal;
  • the carrier stripper is configured to implement carrier stripping of the local carrier and the digital intermediate frequency signal; the code stripper is configured to implement code stripping of the signal after carrier stripping and the baseband signal; and the integration and clearer is used to obtain a code Coherent integration result of the signal after stripping; the coherent integration result of the instant branch in the coherent integration result is sent to the data inversion detector to obtain a data inversion detection output; the coherent integration result of the instant branch is also sent to the
  • the probability weighting factor calculator obtains a probability weighting factor; the power weighting factor calculator is used to compare the power of the data channel and the pilot channel to obtain a power weighting factor; and the coherent integration result of the instant branch is also sent to the data pilot joint
  • the problem of data bit flipping is solved,
  • the coherent integration result of the data channel and the coherent result of the pilot channel can be coherently accumulated, so that the tracking accuracy can be optimized under a high carrier-to-noise ratio environment.
  • the introduction of the probability weighting factor makes the discriminator output result weighted according to the maximum likelihood probability, thus ensuring good tracking performance and improving tracking sensitivity in a low carrier-to-noise ratio environment.
  • the introduction of these two innovative structures is performed after the coherent integration, which belongs to the low-speed processing and does not increase the complexity of the high-speed processing part, so that the entire tracking device is relatively simple and the implementation complexity is low.
  • FIG. 1 is a flowchart of a method for jointly tracking a navigation signal data pilot according to the present invention
  • FIG. 2 is a block diagram of an overall implementation of an embodiment of a data pilot joint tracking method according to the present invention
  • FIG. 3 is a block diagram of a data inversion detection implementation of the present invention.
  • FIG. 4 is a block diagram of an implementation of estimating a data pilot joint carrier adjustment amount according to the present invention.
  • FIG. 5 is a block diagram of an implementation of estimating a data pilot joint code adjustment amount according to the present invention.
  • FIG. 6 is an overall block diagram of a data pilot joint tracking apparatus according to the present invention.
  • the digital intermediate frequency signal is multiplied by a local carrier controlled by a carrier controlled oscillator (NCO) to complete carrier stripping; carrier stripping
  • the subsequent signals are respectively multiplied by the data baseband signal and the pilot baseband signal controlled by the code NCO to complete the code stripping; the signal after the code stripping is processed by integration and clearing to obtain the coherent integration result of each branch; using the coherent integration result
  • Data rollover detection and probability weighting factor calculation; power weighting factor is calculated by power comparison of data channel and pilot channel; data flip detection result, power weighting factor and summary Rate weighting factor auxiliary data pilot combined with carrier adjustment amount estimation, obtaining carrier adjustment amount; data inversion detection result, power weighting factor and probability weighting factor auxiliary data pilot joint code adjustment amount estimation, obtaining code adjustment amount; carrier adjustment amount control carrier
  • the NCO realizes tracking of the data signal of the data pilot joint signal; the code adjustment amount control code NCO realizes tracking of the baseband signal.
  • FIG. 2 is a block diagram showing an overall implementation of an embodiment of a data pilot joint tracking method according to the present invention.
  • the local carrier of the embodiment of the present invention adopts two branches of the same direction (I) and orthogonal (Q), and the baseband signal adopts three branches of lead (E), instant (P), and lag (L).
  • the tracking starting point of the embodiment of the present invention is a digital intermediate frequency signal, and AD conversion and down conversion of the radio frequency analog signal are not considered.
  • the signal system is a navigation signal with a pilot channel, and the mathematical expression of the received digital intermediate frequency signal is as shown in the following formula (1):
  • P 1 represents data channel power
  • d(t) represents data bit symbol at current time t
  • c d represents data channel spreading code
  • ⁇ represents signal delay
  • ⁇ IF represents signal angular frequency
  • Data representing a baseband signal and a pilot phase difference between the baseband signal
  • P 2 represents a pilot channel power
  • c p represents the pilot channel spread code
  • n (t) represents the mean 0 N 0 is the power spectral density of white noise .
  • the mathematical expressions of the received signals are not intended to limit the invention, and any signal containing a data channel and a pilot channel with a constant phase difference therebetween does not depart from the scope of the present application.
  • Step 1 Since the data pilot is jointly tracked, the data channel and the pilot channel signal should be the same frequency and the phase difference is constant, so the data channel and the pilot channel share the carrier NCO 1 and the code NCO 6.
  • the carrier NCO 1 controls the cosine map 2 and the sine map 3, respectively.
  • the cosine map 2 generates an in-phase carrier signal
  • the sinusoid map 3 generates a quadrature carrier signal.
  • the in-phase carrier signal and the digital intermediate frequency signal are multiplied by the multiplier 4 under the control of the sampling pulse to obtain the in-phase branch signal I
  • the orthogonal carrier signal and the digital intermediate frequency signal are multiplied by the multiplier 5 under the control of the sampling pulse to obtain the orthogonal branch signal.
  • Step 2 The code NCO 6 shared by the pilot channel and the data channel respectively drives the pilot baseband signal generator 7 and the data baseband signal generator 8 to generate a leading pilot baseband signal and a leading data baseband signal.
  • the leading pilot baseband signal generates an instantaneous pilot baseband signal and a delayed pilot baseband signal under the action of the delay 9.
  • the leading data baseband signal produces an instantaneous data baseband signal and a delayed data baseband signal by the delayer 10.
  • Step 3 The pilot channel leads, instantaneously, and lags the three baseband signals and phase shift respectively.
  • the subsequent in-phase branch signal I is multiplied by the multipliers 11, 12, and 13 and then accumulated by the integration and the clearers 23, 24, and 25 to obtain pilot in-phase, immediate, and lag coherent integration results (ie, I PE , I PP , I PL );
  • Pilot channel leading, immediate, and lag three baseband signals and phase shift The subsequent quadrature tributary signal Q is multiplied by multipliers 14, 15, and 16 and then accumulated by the integrating and clearing units 26, 27, and 28 to obtain pilot orthogonal lead, instantaneous, and lag coherent integration results (ie, Q PE , Q PP , Q PL );
  • the data channel leads, instantaneously, and lags three baseband signals respectively multiplied by the in-phase tributary signal I through the multipliers 20, 21, and 22, and then accumulated by the integration and the decimators 34, 33, 32 to obtain data in-phase advance, instant and Lag coherent integration results (ie I DE , I DP , I DL );
  • the data channel lead, instantaneous, and lag three baseband signals are respectively multiplied by the orthogonal tributary signal Q by the multipliers 17, 18, 19, and then accumulated by the integration and clearers 31, 30, 29 to obtain data orthogonal advance, Instant and lag coherent integration results (ie Q DE , Q DP , Q DL ).
  • Step 4 Using the above-mentioned instant branch coherent integration results I PP , Q PP , I DP , Q DP for data flip detection, the block diagram of data flip detection is shown in Figure 3. I PP and I DP are multiplied by a multiplier 35, Q PP and Q DP are multiplied by a multiplier 36, the two multiplied results are added by an adder 37, and the result of the addition is taken by the symbol extractor 38.
  • sign(x) means that when x is greater than or equal to 0, the output is +1, and when x is less than zero, the output is -1, that is, the data flip detection output is positive and negative 1, and the output Flip is positive 1.
  • Time indicates that the data channel is in phase with the pilot channel.
  • a negative one indicates that the data channel is 180° out of phase with the pilot channel.
  • Step 5 Calculation of the power weighting factor. Since the data pilot power ratio is not 1:1 in the actual signal system, the influence of power needs to be considered when coherently accumulating the data pilot coherent integration results.
  • the power weighting factor ⁇ 1 directly affects the data pilot joint tracking accuracy.
  • the embodiment of the present invention provides an optimized power weighting factor, as shown in the following formula (2):
  • P 1 and P 2 represent data channel and pilot channel signal power, respectively.
  • Step 6 Calculation of the probability weighting factor.
  • the detection result is not 100% correct, but is correct with a certain probability.
  • the result of high correct probability it can be considered that the output of the discriminator has a high probability of reliability, so it should be output with a large weight; on the contrary, for the result of low correct probability, the probability that the output of the discriminator is reliable is low. , so it should be output with a small weight.
  • T p represents the coherent integration time
  • d represents the current data bit
  • n DIP , n DQP , n PIP , n PQP respectively represent the noise normalized after the coherent integration of each branch.
  • ⁇ is the angle between vector 1 (I DP , Q DP ) and vector 2 (I PP , Q PP ).
  • This weighting factor based on the maximum likelihood probability is not unique, and there are many approximation methods. Any weighting factor that is the same as the monotonicity of the correct probability of data inversion detection does not depart from the scope of the present application.
  • Step 7 The block diagram of the data pilot joint carrier adjustment estimation is shown in Figure 4.
  • the instantaneous branch coherent integration results ie, I PP , Q PP , I DP , Q DP
  • I DP and Q DP are multiplied by the data flip detector output Flip by multipliers 39, 40, respectively, to eliminate the effects of data flipping.
  • the result of the multiplication is multiplied by the power weighting factor ⁇ 1 by the multipliers 41 and 42, respectively, and the result is coherently accumulated by the adders 43, 44 and I PP and I DP to obtain I P and Q P , respectively .
  • I P and Q P are then phase-detected by the phase detector 45 to obtain a phase-detecting output.
  • phase-locked loop phase detector can be used, for example: four-quadrant arctangent arctan(I P , Q P ).
  • Phase detection output The multiplier 46 multiplies the probability weighting factor ⁇ 2 and performs filtering processing by the loop filter 47 to obtain a final carrier adjustment amount.
  • the carrier adjustment amount is fed back to the carrier NCO 1 in FIG. 2, and the carrier NCO 1 is adjusted to implement tracking of the data pilot joint signal carrier.
  • Step 8 The block diagram of the data pilot joint code adjustment amount estimation is shown in FIG. 5.
  • the coherent integration results of the leading and trailing branches ie, I PE , Q PE , I PL , Q PL , I DE , Q DE , I DL , Q DL ) are used to estimate the amount of data pilot joint code adjustment.
  • I DE , Q DE , I DL , and Q DL are multiplied by the data flip detector output Flip by multipliers 49, 50, 51, 52, respectively, to eliminate the influence of data flipping.
  • the multiplied results are then multiplied by a power weighting factor ⁇ 1 by multipliers 53, 54, 55, 56, respectively, and the results are passed through adders 57, 58, 59, 60 and I PE , Q PE , I PL , Q, respectively.
  • the PL rows are coherently accumulated to obtain joint coherent integration results I E , Q E , I L , Q L .
  • N represents the number of non-coherent accumulations.
  • the leading non-coherent accumulation result E S and the lag non-coherent accumulation result L S are further phase-detected by the phase detector 63 to obtain a phase-detection result ⁇ .
  • the non-coherent over period lag squared phase detector is selected and normalized.
  • the phase detector is used only to maintain the integrity of the present invention, and other implementations do not depart from the scope of the present application. Its mathematical expression is shown in the following formula (9):
  • phase discrimination result ⁇ is multiplied by the probability weighting factor ⁇ 2 by the multiplier 64, and then subjected to filtering processing by the loop filter 65 to obtain a final code adjustment amount.
  • the code adjustment amount is fed back to the code NCO 6 in FIG. 2, and the code NCO 6 is adjusted to achieve tracking of the baseband signal.
  • the carrier NCO 104 controls the local carrier generator 102 to generate a local carrier
  • the code NCO 105 controls the baseband signal generator 103 to generate a baseband signal. Note that the baseband signal is required.
  • the baseband signal generated by the generator 103 includes both a data baseband signal and a pilot baseband signal.
  • the local carrier and the digital intermediate frequency signal are subjected to carrier stripping by the carrier stripper 100, and the carrier stripped signal and the baseband signal are subjected to code stripping by the code stripper 101.
  • the signal after the code stripping is further obtained by the integration and clearer 106 to obtain the coherent integration result, wherein the coherent integration result of the instant branch is sent to the data inversion detector 107 to obtain the data inversion detection output Flip, and the implementation principle of the data inversion detector 107 is as follows.
  • Figure 3 shows.
  • the coherent integration result of the instant branch is also fed to the probability weighting factor calculator 108 to obtain a probability weighting factor ⁇ 2 whose implementation principle is defined by equation (7).
  • the power weighting factor ⁇ 1 is obtained by the power weighting factor calculator 109.
  • the instant branch coherent integration result is also sent to the data pilot joint carrier adjustment estimator 110.
  • a typical implementation is shown in FIG.
  • the data pilot joint carrier adjustment estimator 110 obtains a carrier adjustment amount through which the carrier is obtained.
  • the adjustment amount controls the carrier NCO 104 to implement tracking of the data pilot joint signal carrier.
  • the integration and cleaner 106 obtains the coherent integration result of the lead branch and the lag branch in the coherent integration result and sends it to the data pilot joint code adjustment amount estimator 111.
  • a typical implementation is shown in FIG. 5, and the data guide
  • the frequency joint code adjustment amount estimator 111 obtains a code adjustment amount, and the baseband signal is tracked by the code adjustment amount control code NCO105.

Abstract

一种导航信号数据导频联合跟踪方法及装置，所述方法包括：数据导频联合中频信号与载波数字控制振荡器(NCO)控制的本地载波相乘，完成载波剥离；载波剥离后的信号分别与码NCO控制的数据基带信号和导频基带信号相乘，完成码剥离；码剥离后的信号通过积分和清零处理，得到各支路相干积分结果；利用相干积分结果实现数据翻转检测和概率加权因子计算；数据翻转检测结果和概率加权因子辅助数据导频联合载波调整量估计，得到载波调整量；数据翻转检测结果和概率加权因子辅助数据导频联合码调整量估计，得到码调整量；载波调整量控制载波NCO，实现对数据导频联合载波信号的跟踪；码调整量控制码NCO，实现对数据导频联合基带信号的跟踪。

Description

一种导航信号数据导频联合跟踪方法及装置 【技术领域】
本发明属于导航信号跟踪技术领域，更具体地，涉及一种导航信号数据导频联合跟踪方法及装置。
【背景技术】
随着全球导航卫星系统(GNSS)的持续建设，导航信号体制引入了许多新的技术。与传统单数据通道的导航信号相比，导频通道的引入是现代化的导航信号体制的最大特点。对GPS和Galileo的ICD文档进行分析后可以发现，大多数新设计的导航信号均引入了导频分量，其中包括：GPS L1C、GPS L2C、GPS L5C、Galileo E1 OS、Galileo E5、Galileo E6。
导频通道的引入极大的提升了导航信号的跟踪性能。由于导频信号没有调制数据位，不存在数据位翻转的问题，所以在跟踪导频信号时，可以使用更长的相干积分时间，从而提高跟踪精度。除此之外，导频通道的载波跟踪可以使用对数据位翻转敏感的纯锁相环(Pure PLL)，而数据通道由于有数据位的翻转，其载波跟踪只能使用对数据位翻转不敏感的科斯塔斯环(Costas loop)。纯锁相环的动态牵引范围是360°，对应的相位噪声1-sigma经验门限为45°，而科斯塔斯环的动态牵引范围只有180°，对应的相位噪声1-sigma经验门限为15°，由此可见，纯锁相环的跟踪灵敏度要优于科斯塔斯环。但是这种单导频的跟踪方式却会带来另外一个问题：数据通道的信号没有参与跟踪，造成能量损失。所以在跟踪过程中如何将数据和导频联合成为现代化导航信号跟踪的研究热点。
对于导航信号数据导频联合跟踪，目前国内外已经有了许多研究成果。主要可以分为两类：一种是数据导频非相干累加，另一种是数据导频相干累加。在参考文献[1]“Trade-Off Between Data Rate and Signal Power  Split in GNSS Signal Design”中提到了一种数据导频非相干联合跟踪的方法，该方法在数据通道采用科斯塔斯环，在导频通道采用纯锁相环，然后将两个环路的鉴别器输出按照一定权重相加，得到一个可供数据通道和导频通道共用的鉴别器输出。但是这种非相干的累加方法会带来平方损耗，使得跟踪精度并不是最优。除此之外，数据通道采用科斯塔斯环，其牵引范围小于导频通道的纯锁相环，这种联合方式相对单导频的跟踪灵敏度的提升并不明显。
在参考文献[2]“Dual Channel Optimizations of Tracking Schemes for E1 CBOC Signal”中，提出了一种数据导频相干累加的方法，该方法通过对数据位的估计，遍历数据通道与导频通道的可能组合方式，然后通过一定的判决方式确定最终的组合方式，将组合后的相干积分结果用于跟踪。这种方法在高载噪比的时候，能达到最佳的跟踪性能，但是在低载噪比的时候性能不佳。因此，有必要为带导频通道的导航信号提供一种性能良好且结构简单的数据导频联合跟踪的方法。
【发明内容】
针对现有技术的以上缺陷或改进需求，本发明提供一种通用的、实现复杂度低、跟踪精度高、跟踪灵敏度高的导航信号数据导频联合跟踪方法及装置，用于带导频通道的导航信号的跟踪，并提供良好的跟踪性能。
为实现上述目的，按照本发明的一个方面，提供一种导航信号数据导频联合跟踪方法，包括以下步骤：
步骤1将数字中频信号与本地载波相乘，完成载波剥离，所述本地载波采用同向和正交两个支路；
步骤2载波剥离后的信号分别与数据基带信号和导频基带信号相乘，完成码剥离，基带信号采用超前、即时和滞后三个支路，其中所述基带信号包括所述数据基带信号和所述导频基带信号；
步骤3码剥离后的信号通过积分和清零处理，得到各个支路的相干积 分结果，其中，导频通道的相干积分结果包括：同相超前IPE、同相即时IPP、同相滞后IPL、正交超前QPE、正交即时QPP以及正交滞后QPL，数据通道的相干积分结果包括：同相超前IDE、同相即时IDP、同相滞后IDL、正交超前QDE、正交即时QDP以及正交滞后QDL；
步骤4利用各即时支路的相干积分结果实现数据翻转检测，得到数据翻转检测输出Flip＝sign(IPP×IDP+QPP×QDP)，其中，sign表示取符号位函数，sign(x)表示当x大于等于0时输出为+1，当x小于零时输出为-1；
步骤5利用所述数据通道和所述导频通道的功率对比计算最优功率加权因子
Figure PCTCN2015075671-appb-000001
其中，P1和P2分别表示所述数据通道和所述导频通道信号功率；
步骤6利用各即时支路的相干积分结果计算概率加权因子α2，所述概率加权因子α2为与所述数据翻转检测正确概率单调性相同的权重因子；
步骤7计算得到同相即时IP和正交即时QP相干积分结果如下所示：
IP＝IPP+Flip×IDP×α1 QP＝QPP+Flip×QDP×α1
然后将所述同相即时IP和所述正交即时QP相干积分结果进行鉴相，得到鉴相输出
Figure PCTCN2015075671-appb-000002
并将所述鉴相输出
Figure PCTCN2015075671-appb-000003
和所述概率加权因子α2相乘，然后经环路滤波得到数据导频联合载波调整量，所述数据导频联合载波调整量反馈到控制所述本地载波的载波数字控制振荡器，对所述载波数字控制振荡器进行调整，实现对数据导频联合信号载波的跟踪；
步骤8计算得到数据导频联合同相超前IE、同相滞后IL、正交超前QE和正交滞后QL相干积分结果如下所示：
IE＝IPE+Flip×IDE×α1 IL＝IPL+Flip×IDL×α1
QE＝QPE+Flip×QDE×α1 QL＝QPL+Flip×QDL×α1
然后将所述同相超前IE、所述同相滞后IL、所述正交超前QE和所述正交滞后QL相干积分结果进行鉴相，得到鉴相结果Δτ，所述鉴相结果Δτ和概 率加权因子α2相乘，然后经环路滤波得到数据导频联合码调整量，所述码调整量反馈到控制所述基带信号的码数字控制振荡器，对所述码数字控制振荡器进行调整，实现对所述基带信号的跟踪。
按照本发明的另一方面，提供一种导航信号数据导频联合跟踪装置，包括：载波剥离器、码剥离器、本地载波生成器、基带信号生成器、载波数字控制振荡器、码数字控制振荡器、积分和清零器、数据翻转检测器、概率加权因子计算器、功率加权因子计算器、数据导频联合载波调整量估计器以及数据导频联合码调整量估计器，其中：
所述载波数字控制振荡器用于控制本地载波生成器生成本地载波；所述码数字控制振荡器用于控制基带信号生成器生成基带信号，所述基带信号包括数据基带信号和导频基带信号；所述载波剥离器用于实现所述本地载波与数字中频信号的载波剥离；所述码剥离器用于实现载波剥离后的信号与所述基带信号的码剥离；所述积分和清零器用于得到码剥离后的信号的相干积分结果；所述相干积分结果中的即时支路的相干积分结果送入所述数据翻转检测器得到数据翻转检测输出；所述即时支路的相干积分结果还送入所述概率加权因子计算器得到概率加权因子；所述功率加权因子计算器用于对比数据通道和导频通道的功率，得到功率加权因子；所述即时支路的相干积分结果还送入数据导频联合载波调整量估计器得到载波调整量，通过所述载波调整量控制所述载波数字控制振荡器，实现对数据导频联合信号载波的跟踪；所述相干积分结果中的超前支路和滞后支路的相干积分结果送入所述数据导频联合码调整量估计器得到码调整量，通过所述码调整量控制所述码数字控制振荡器，实现对所述基带信号的跟踪。
总体而言，通过本发明所构思的以上技术方案与现有技术相比，具有以下有益效果：
由于本发明数据翻转检测器的引入，解决了数据位翻转的问题，使得 数据通道的相干积分结果和导频通道的相干结果可以进行相干累加，从而使得跟踪精度在高载噪比环境下能够达到最优。概率加权因子的引入，使得鉴别器输出结果按照最大似然概率加权，从而在低载噪比的环境下能够保证良好的跟踪性能，提高跟踪灵敏度。而且这两种创新性的结构引入都是在相干积分之后进行，属于低速处理，没有增加高速处理部分的复杂度，使得整个跟踪设备比较简单，实现复杂度低。
【附图说明】
图1为本发明导航信号数据导频联合跟踪方法的流程图；
图2为本发明数据导频联合跟踪方法的一实施例的整体实现框图；
图3为本发明数据翻转检测实现框图；
图4为本发明数据导频联合载波调整量估计实现框图；
图5为本发明数据导频联合码调整量估计实现框图；
图6为本发明数据导频联合跟踪装置的整体框图。
【具体实施方式】
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅仅用以解释本发明，并不用于限定本发明。此外，下面所描述的本发明各个实施方式中所涉及到的技术特征只要彼此之间未构成冲突就可以相互组合。
图1所示为本发明导航信号数据导频联合跟踪方法的流程图，数字中频信号与载波数字控制振荡器(numerically controlled oscillator，以下简称NCO)控制的本地载波相乘，完成载波剥离；载波剥离后的信号分别与码NCO控制的数据基带信号和导频基带信号相乘，完成码剥离；码剥离后的信号通过积分和清零处理，得到各个支路相干积分结果；利用该相干积分结果实现数据翻转检测和概率加权因子计算；利用数据通道和导频通道的功率对比计算功率加权因子；数据翻转检测结果、功率加权因子和概 率加权因子辅助数据导频联合载波调整量估计，得到载波调整量；数据翻转检测结果、功率加权因子和概率加权因子辅助数据导频联合码调整量估计，得到码调整量；载波调整量控制载波NCO，实现对数据导频联合信号载波信号的跟踪；码调整量控制码NCO，实现对基带信号的跟踪。
为了描述方便，下文给出了一种典型实施例，但是该实施例仅仅是为了说明的目的而举的示例，而不是用来进行限制。本领域技术人员应当理解，凡在本申请的教导和权利要求保护范围下所作的任何修改、等同替换等，均应包含在本申请要求保护的范围内。
图2所示为本发明数据导频联合跟踪方法的一实施例的整体实现框图。本发明实施例的本地载波采用同向(I)和正交(Q)两个支路，基带信号采用超前(E)、即时(P)、滞后(L)三个支路。本发明实施例的跟踪起点为数字中频信号，对射频模拟信号的AD转换和下变频不予考虑。针对的信号体制是带导频通道的导航信号，接收的数字中频信号数学表达式如下公式(1)所示：
Figure PCTCN2015075671-appb-000004
其中：P1表示数据通道功率；d(t)表示当前时刻t的数据比特符号；cd表示数据通道扩频码；τ表示信号时延；ωIF表示信号角频率；
Figure PCTCN2015075671-appb-000005
表示载波初始相位；
Figure PCTCN2015075671-appb-000006
表示数据基带信号和导频基带信号之间的相位差；P2表示导频通道功率；cp表示导频通道扩频码；n(t)表示均值为0功率谱密度为N0的白噪声。这里给出接收信号的数学表达式并不是为了限定本发明，凡是含有数据通道和导频通道并且二者之间相位差恒定的信号都不脱离本发明的申请范围。
如图2所示的典型实施例的详细步骤如下：
步骤1：由于是数据导频联合跟踪，数据通道和导频通道信号应该同频并且相位差恒定，所以数据通道和导频通道共用载波NCO 1和码NCO 6。如 图2所示，载波NCO 1分别控制余弦映射表2和正弦映射表3。余弦映射表2产生同相载波信号，正弦映射表3产生正交载波信号。同相载波信号和数字中频信号在采样脉冲控制下通过乘法器4相乘得到同相支路信号I，正交载波信号和数字中频信号在采样脉冲控制下通过乘法器5相乘得到正交支路信号Q。
步骤2：导频通道和数据通道共用的码NCO 6分别驱动导频基带信号发生器7和数据基带信号发生器8生成超前的导频基带信号和超前的数据基带信号。超前的导频基带信号在延时器9的作用下产生即时导频基带信号和滞后导频基带信号。超前的数据基带信号在延时器10的作用下产生即时的数据基带信号和滞后的数据基带信号。
步骤3：导频通道超前、即时、滞后三路基带信号分别与相位移动
Figure PCTCN2015075671-appb-000007
后的同相支路信号I通过乘法器11、12、13相乘，再通过积分和清零器23、24、25累加，得到导频同相超前、即时和滞后相干积分结果(即IPE、IPP、IPL)；
导频通道超前、即时、滞后三路基带信号分别与相位移动
Figure PCTCN2015075671-appb-000008
后的正交支路信号Q通过乘法器14、15、16相乘，再通过积分和清零器26、27、28累加，得到导频正交超前、即时和滞后相干积分结果(即QPE、QPP、QPL)；
数据通道超前、即时、滞后三路基带信号分别与同相支路信号I通过乘法器20、21、22相乘，再通过积分和清零器34、33、32累加，得到数据同相超前、即时和滞后相干积分结果(即IDE、IDP、IDL)；
数据通道超前、即时、滞后三路基带信号分别与正交支路信号Q通过乘法器17、18、19相乘，再通过积分和清零器31、30、29累加，得到数据正交超前、即时和滞后相干积分结果(即QDE、QDP、QDL)。
步骤4：利用上述即时支路相干积分结果IPP、QPP、IDP、QDP进行数据翻转检测，数据翻转检测的实现框图如图3所示。IPP和IDP通过乘法器35相乘，QPP和QDP通过乘法器36相乘，两个相乘结果通过加法器37相加， 再由符号提取器38对相加的结果取符号，得到数据翻转检测输出Flip：
Flip＝sign(IPP×IDP+QPP×QDP)
其中，sign表示取符号位函数，sign(x)表示当x大于等于0时输出为+1，当x小于零时输出为-1，即数据翻转检测输出为正负1，输出Flip为正1时表示数据通道与导频通道同相，为负1时表示数据通道与导频通道相差180°。
步骤5：功率加权因子的计算。由于实际的信号体制中，数据导频功率比并不是1:1，所以将数据导频相干积分结果进行相干累加时需要考虑功率的影响。功率加权因子α1会直接影响数据导频联合跟踪精度，本发明实施例给出最优化的功率加权因子，如下公式(2)所示：
Figure PCTCN2015075671-appb-000009
其中，P1和P2分别表示数据通道和导频通道信号功率。选取该加权因子，数据导频联合的跟踪精度能够达到最优。
步骤6：概率加权因子计算。对于数据翻转的检测，由于有噪声的存在，所以检测结果并不是百分之百正确，而是以一定的概率正确。对于正确概率高的结果，可以认为本次鉴别器的输出可靠的概率高，所以应该以较大的权重输出；相反，对于正确概率低的结果，可以认为本次鉴别器的输出可靠的概率低，所以应该以较小的权重输出。
根据上述公式(1)，数据通道和导频通道即时支路的相干积结果的数学表达式如下所示：
Figure PCTCN2015075671-appb-000010
Figure PCTCN2015075671-appb-000011
Figure PCTCN2015075671-appb-000012
Figure PCTCN2015075671-appb-000013
其中，Tp表示相干积分时间；d表示当前数据位；nDIP、nDQP、nPIP、nPQP分别表示各支路相干积分后归一化的噪声。
定义θ为矢量1(IDP，QDP)和矢量2(IPP，QPP)的夹角。没有信号存在时，由于nDIP、nDQP、nPIP、nPQP是零均值的白噪声，所以θ应该在0～180°内均匀分布。有信号存在时，假设d＝1，则两个矢量信号部分应该是同相，则θ角度越小的概率越高。根据前面数据翻转检测器的原理，当θ<90°时，判断d＝1，此时θ角度越小，判断正确的概率越高，而θ角度越接近90°，判断正确的概率越低；假设d＝-1，则两个矢量信号部分应该是反相，则θ角度越大的概率越高。根据前面数据翻转检测器的原理，当θ>90°时，判断d＝-1，此时θ角度越接近180°，判断正确的概率越高，而θ角度越接近90°，判断正确的概率越低。实际上在θ＝90°时，可以认为由于有噪声的存在，此时已经完全无法判断d的正负，这种结果完全不可靠，应以权重0输出。但是实际的计算这种最大似然概率权重的过程非常复杂，不利于工程实现，凡是与数据翻转检测正确概率单调性相同的权重因子都可以作为近似的概率加权因子，近似度越高，跟踪结果越好。在本发明实施例中用一个和最大似然概率权重单调性相同且近似度较高的权重因子来替换。定义概率加权因子α2如下公式(7)所示：
Figure PCTCN2015075671-appb-000014
这种基于最大似然概率的加权因子并不唯一，还有许多近似的方法，任何与数据翻转检测正确概率单调性相同的权重因子均不脱离本申请的范围。
步骤7：数据导频联合载波调整量估计实现框图如图4所示。即时支路相干积分结果(即IPP、QPP、IDP、QDP)用来进行数据导频联合载波调整量 估计。IDP和QDP分别通过乘法器39、40与数据翻转检测器输出Flip相乘，消除数据翻转的影响。相乘的结果再分别通过乘法器41、42与功率加权因子α1相乘，其结果再分别通过加法器43、44与IPP和IDP进行相干累加得到IP和QP。IP和QP再通过鉴相器45进行鉴相，得到鉴相输出
Figure PCTCN2015075671-appb-000015
由于消除了数据位翻转，可以使用纯锁相环鉴相器，例如：四象限反正切arctan(IP，QP)。鉴相输出
Figure PCTCN2015075671-appb-000016
通过乘法器46与概率加权因子α2相乘，再通过环路滤波器47进行滤波处理，得到最终的载波调整量。载波调整量反馈到图2中的载波NCO 1，对载波NCO 1进行调整，实现对数据导频联合信号载波的跟踪。
步骤8：数据导频联合码调整量估计实现框图如图5所示。超前支路和滞后支路的相干积分结果(即IPE、QPE、IPL、QPL、IDE、QDE、IDL、QDL)用来进行数据导频联合码调整量估计。IDE、QDE、IDL、QDL分别通过乘法器49、50、51、52与数据翻转检测器输出Flip相乘，消除数据翻转的影响。相乘的结果再分别通过乘法器53、54、55、56与功率加权因子α1相乘，其结果再分别通过加法器57、58、59、60与IPE、QPE、IPL、QPL行相干累加得到联合相干积分结果IE、QE、IL、QL。码相位鉴别器有许多不同的实现方法，为了保持发明的完整性，给出一种非相干超前减滞后鉴别器，其它的实现方式不脱离本申请的范围。联合相干积分结果IE、QE和IL、QL分别通过非相干累加器61、62进行非相干累加得到超前非相干累加结果ES和滞后非相干累加结果LS，其数学表达式如下公式(8)所示：
Figure PCTCN2015075671-appb-000017
其中，N表示非相干累加次数。超前非相干累加结果ES和滞后非相干累加结果LS再通过鉴相器63进行鉴相，得到鉴相结果Δτ。这里选取非相干超期减滞后平方鉴相器，并做归一化处理，采用该鉴相器只是为了保持本发明的完整性，其它的实现方式不脱离本申请的范围。其数学表达式如下公式(9)所示：
Figure PCTCN2015075671-appb-000018
鉴相结果Δτ通过乘法器64与概率加权因子α2相乘，再通过环路滤波器65进行滤波处理，得到最终的码调整量。码调整量反馈到图2中的码NCO 6，对码NCO 6进行调整，实现对基带信号的跟踪。
图6所示为本发明数据导频联合跟踪装置的整体框图，载波NCO 104控制本地载波生成器102生成本地载波，码NCO 105控制基带信号生成器103生成基带信号，需要注意的是，基带信号生成器103生成的基带信号既包括数据基带信号，也包括导频基带信号。本地载波与数字中频信号通过载波剥离器100实现载波剥离，载波剥离后的信号与基带信号通过码剥离器101实现码剥离。码剥离后的信号再通过积分和清零器106得到相干积分结果，其中即时支路的相干积分结果送入数据翻转检测器107，得到数据翻转检测输出Flip，数据翻转检测器107的实现原理如图3所示。即时支路的相干积分结果还送入概率加权因子计算器108，得到概率加权因子α2，其实现原理由公式(7)定义。利用数据通道和导频通道的功率对比，通过功率加权因子计算器109，得到功率加权因子α1。即时支路相干积分结果还送入数据导频联合载波调整量估计器110，其一种典型的实施例如图4所示，数据导频联合载波调整量估计器110得到载波调整量，通过该载波调整量控制载波NCO 104，实现对数据导频联合信号载波的跟踪。积分和清零器106得到相干积分结果中的超前支路和滞后支路的相干积分结果送入数据导频联合码调整量估计器111，其一种典型的实施例如图5所示，数据导频联合码调整量估计器111得到码调整量，通过该码调整量控制码NCO105，实现对基带信号的跟踪。
本领域的技术人员容易理解，以上所述仅为本发明的较佳实施例而已，并不用以限制本发明，凡在本发明的精神和原则之内所作的任何修改、等同替换和改进等，均应包含在本发明的保护范围之内。

Claims (9)

  1. 一种导航信号数据导频联合跟踪方法，其特征在于，包括：
    步骤1将数字中频信号与本地载波相乘，完成载波剥离，所述本地载波采用同向和正交两个支路；
    步骤2载波剥离后的信号分别与数据基带信号和导频基带信号相乘，完成码剥离，基带信号采用超前、即时和滞后三个支路，其中所述基带信号包括所述数据基带信号和所述导频基带信号；
    步骤3码剥离后的信号通过积分和清零处理，得到各个支路的相干积分结果，其中，导频通道的相干积分结果包括：同相超前IPE、同相即时IPP、同相滞后IPL、正交超前QPE、正交即时QPP以及正交滞后QPL，数据通道的相干积分结果包括：同相超前IDE、同相即时IDP、同相滞后IDL、正交超前QDE、正交即时QDP以及正交滞后QDL；
    步骤4利用各即时支路的相干积分结果实现数据翻转检测，得到数据翻转检测输出Flip＝sign(IPP×IDP+QPP×QDP)，其中，sign表示取符号位函数，sign(x)表示当x大于等于0时输出为+1，当x小于零时输出为-1；
    步骤5利用所述数据通道和所述导频通道的功率对比计算最优功率加权因子
    Figure PCTCN2015075671-appb-100001
    其中，P1和P2分别表示所述数据通道和所述导频通道信号功率；
    步骤6利用各即时支路的相干积分结果计算概率加权因子α2，所述概率加权因子α2为与所述数据翻转检测正确概率单调性相同的权重因子；
    步骤7计算得到同相即时IP和正交即时QP相干积分结果如下所示：
    IP＝IPP+Flip×IDP×α1   QP＝QPP+Flip×QDP×α1
    然后将所述同相即时IP和所述正交即时QP相干积分结果进行鉴相，得到鉴相输出
    Figure PCTCN2015075671-appb-100002
    并将所述鉴相输出
    Figure PCTCN2015075671-appb-100003
    和所述概率加权因子α2相乘，然后经 环路滤波得到数据导频联合载波调整量，所述数据导频联合载波调整量反馈到控制所述本地载波的载波数字控制振荡器，对所述载波数字控制振荡器进行调整，实现对数据导频联合信号载波的跟踪；
    步骤8计算得到数据导频联合同相超前IE、同相滞后IL、正交超前QE和正交滞后QL相干积分结果如下所示：
    IE＝IPE+Flip×IDE×α1   IL＝IPL+Flip×IDL×α1
    QE＝QPE+Flip×QDE×α1   QL＝QPL+Flip×QDL×α1
    然后将所述同相超前IE、所述同相滞后IL、所述正交超前QE和所述正交滞后QL相干积分结果进行鉴相，得到鉴相结果Δτ，所述鉴相结果Δτ和概率加权因子α2相乘，然后经环路滤波得到数据导频联合码调整量，所述码调整量反馈到控制所述基带信号的码数字控制振荡器，对所述码数字控制振荡器进行调整，实现对所述基带信号的跟踪。
  2. 如权利要求1所述的方法，其特征在于，所述步骤1中接收的所述数字中频信号含有所述数据通道和所述导频通道并且二者之间相位差恒定。
  3. 如权利要求1所述的方法，其特征在于，所述步骤1中接收的所述数字中频信号的表达式如下所示：
    Figure PCTCN2015075671-appb-100004
    其中，P1表示数据通道功率；d(t)表示当前时刻t的数据比特符号；cd表示数据通道扩频码；τ表示信号时延；ωIF表示信号角频率；
    Figure PCTCN2015075671-appb-100005
    表示载波初始相位；
    Figure PCTCN2015075671-appb-100006
    表示所述数据基带信号和所述导频基带信号之间的相位差；P2表示导频通道功率；cp表示导频通道扩频码；n(t)表示均值为0功率谱密度为N0的白噪声。
  4. 如权利要求1-3中任一项所述的方法，其特征在于，所述步骤3中所述导频通道超前、即时、滞后三路基带信号分别与同相支路信号相乘， 再分别进行累加，得到导频同相超前、即时和滞后相干积分结果；所述导频通道超前、即时、滞后三路基带信号还分别与正交支路信号相乘，再分别进行累加，得到导频正交超前、即时和滞后相干积分结果；
    所述数据通道超前、即时、滞后三路基带信号分别与同相支路信号相乘，再分别进行累加，得到数据同相超前、即时和滞后相干积分结果；所述数据通道超前、即时、滞后三路基带信号还分别与正交支路信号相乘，再分别进行累加，得到数据正交超前、即时和滞后相干积分结果。
  5. 如权利要求1-3中任一项所述的方法，其特征在于，所述步骤4中所述数据翻转检测输出Flip为正1时表示所述数据通道与所述导频通道同相，为负1时表示所述数据通道与所述导频通道相差180°。
  6. 如权利要求1-3中任一项所述的方法，其特征在于，所述步骤6中所述概率加权因子为
    Figure PCTCN2015075671-appb-100007
  7. 如权利要求1所述的方法，其特征在于，所述步骤7中使用纯锁相环鉴相器对所述同相即时IP和所述正交即时QP相干积分结果进行鉴相。
  8. 如权利要求1所述的方法，其特征在于，所述步骤8中联合相干积分结果IE、QE和IL、QL分别进行非相干累加得到超前非相干累加结果ES和滞后非相干累加结果LS，其数学表达式如下所示：
    Figure PCTCN2015075671-appb-100008
    其中，N表示非相干累加次数；所述超前非相干累加结果ES和所述滞后非相干累加结果LS再进行鉴相，得到所述鉴相结果Δτ。
  9. 一种导航信号数据导频联合跟踪装置，其特征在于，包括：载波剥离器、码剥离器、本地载波生成器、基带信号生成器、载波数字控制振荡器、码数字控制振荡器、积分和清零器、数据翻转检测器、概率加权因子计算器、功率加权因子计算器、数据导频联合载波调整量估计器以及数据 导频联合码调整量估计器，其中：
    所述载波数字控制振荡器用于控制本地载波生成器生成本地载波；所述码数字控制振荡器用于控制基带信号生成器生成基带信号，所述基带信号包括数据基带信号和导频基带信号；所述载波剥离器用于实现所述本地载波与数字中频信号的载波剥离；所述码剥离器用于实现载波剥离后的信号与所述基带信号的码剥离；所述积分和清零器用于得到码剥离后的信号的相干积分结果；所述相干积分结果中的即时支路的相干积分结果送入所述数据翻转检测器得到数据翻转检测输出；所述即时支路的相干积分结果还送入所述概率加权因子计算器得到概率加权因子；所述功率加权因子计算器用于对比数据通道和导频通道的功率，得到功率加权因子；所述即时支路的相干积分结果还送入数据导频联合载波调整量估计器得到载波调整量，通过所述载波调整量控制所述载波数字控制振荡器，实现对数据导频联合信号载波的跟踪；所述相干积分结果中的超前支路和滞后支路的相干积分结果送入所述数据导频联合码调整量估计器得到码调整量，通过所述码调整量控制所述码数字控制振荡器，实现对所述基带信号的跟踪。
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